hift corresponds to a large number of code chips, often as many as half the code period. Consider once again using the DLL of Figure 2.1 to PN track the sum of s(t) of (2.96) and ni(t) of (2.2). If the PN generator of the DLL is arbitrarily selectd to match c1(t), then, since, by the assumption of code independence, C2t-Tct-,8) (2.97) and the fact that the inphase and quadrature self-noise processes can be neglected, we obtain a loop error signal identical to (2.7) except that S is replaced by S/2 in both its signal and noise components. Slated another way, the DLL tracks the channel corresponding to C1(t - T1) (which contains half the total power) as if the other channel were absent. As such, a 3 dB signal- to-noise radio penalty is paid relative to that for a PN spread binary signal- to-noise ratio penalty is paid relative to that for a PN spread binary data modulation as in (2.1). This 3 dB penalty in signal-to-noise ratio does not, however, directly translate into a 3 dB increase in mean-squared tracking jitter due to the non-linear dependence of the loop squaring loss on S/N° (see (2.29) and (2.30)). Once having obtained proper code alignment for C1 2 t,), the despreading code for C2(t-T1- - TD) can be derived by an appropri- ate shift of the local PN generator provided that the two codes were gener- ated as suggested above. If indeed the two codes are totally unrelated, i.e., they were generated by separate PN generators with, in general, completely Pseudonoise Code Tracking in Direct-Sequence Receivers different tap connections and code periods, then two separate DLLs must be employed in the receiver, each having a local PN generator matched to one of the two codes. Again, the performance of each DLL would behave as per the above discussion. Finally, if the two codes have different chip rates in addition to different code structure, one must again employ separate DLLs for each code where now, in addition to having different local PN generators, the two loops would also have different voltage-controlled clo