ed, with an under-compensation of 1020ps/nm that in our case means an uncompensated dispersion of 141 ps. Taking into account that the transmit- ted pulses are 60 ps long residual dispersion requires per channel compensation. Coherent detection renders available both amplitude and phase of the received signal SO that very smart electronic processing is possible, simultaneously targeting the equaliza- tion of several impairments [51,78]. Here we will use a suboptimal design: that is, we will assume that the ISI equalizer (caring for chromatic dispersion, PMD, and any form of nonlinear ISI), the LO frequency locker, and the decision device are separated circuits that do not exchange information, but for the link between decision and frequency lock. As a matter of fact, decision driven frequency lock is operated SO that the decision device provides a feedback to the frequency lock control signal generator. A wide review of all the available alternatives that is very useful to analyze other design possibilities is reported in [77]. As far as the ISI equalizer is concerned, we will assume to use a small number of states MLSE with the objective to compensate all the ISI deriving from the interplay between residual dispersion and SPM. Such an MLSE can be based on a small number of sample per bit; in literature it is clearly demonstrated that a number of samples greater than 4-5 per bit interval is of small util- ity if the number of states of the MLSE is not huge [78] and three samples seems to be the optimum value for small states MLSE [79]. Telecommunication Networks Thus a sampling rate of 50 Gsamples/s is assumed, which is within the reach of very fast 32 nm CMOS electronics, as demonstrated by literature on predistortion compensation (see Chapter 5). A very general method for the evaluation of the performances of an MLSE in the case of multilevel transmission is reported in [79]. This can be applied in our case with suit- able modification that takes into account both the different receiver and the modulation forma