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Patent US20080037614 - GPS m-code receiver tracking system - Google PatentsSearch Images Maps Play YouTube News Gmail Drive More »Sign inPatentsAn m-code GPS receiver receives m-code GPS communication signals having a multimodal autocorrelation, using an m-code mode identifier unambiguously determining a mode value of one of the m-code modal peaks coherently aligned to a coherent unimodal detected envelope, based on sequential probability estimation...http://www.google.com/patents/US20080037614?utm_source=gb-gplus-sharePatent US20080037614 - GPS m-code receiver tracking systemAdvanced Patent SearchPublication numberUS20080037614 A1Publication typeApplicationApplication numberUS 11/500,801Publication dateFeb 14, 2008Filing dateAug 8, 2006Priority dateAug 8, 2006Also published asUS7899109, US7944997, US20100046589Publication number11500801, 500801, US 2008/0037614 A1, US 2008/037614 A1, US 20080037614 A1, US 20080037614A1, US 2008037614 A1, US 2008037614A1, US-A1-20080037614, US-A1-2008037614, US2008/0037614A1, US2008/037614A1, US20080037614 A1, US20080037614A1, US2008037614 A1, US2008037614A1InventorsRandal K. DouglasOriginal AssigneeDouglas Randal KExport CitationBiBTeX, EndNote, RefManPatent Citations (20), Referenced by (15), Classifications (6), Legal Events (2) External Links: USPTO, USPTO Assignment, EspacenetGPS m-code receiver tracking system
[0001] The invention relates to the field of communications and navigation systems. More particularly, the present invention relates to tracking GPS m-code signals for use in communications and navigation systems.
[0002] The GPS m-code signal is a symmetric multimodal signal having a plurality of modes or peaks in a multimodal autocorrelation function of the m-code signal. The steep gradients of the m-code autocorrelation function about the peaks allow for precise code phase tracking with conventional square law code tracking loops and other code tracking schemes. The m-code autocorrelation function has seven modes, all indistinguishable with respect to conventional early, prompt, and late correlator tap arrangements. A problem with tracking of the m-code multimodal autocorrelation function is that code phase error estimates can be ambiguous at to which one of the peaks is being tracked.
[0003] A weak-lock GPS signal tracking system uses a GPS signal-tracking filter. The GPS signal-tracking filter is an extended Kalman filter with high data rate of 1000 Hz, measurements that include early, prompt, and late inphase and quadrature correlations. States of the GPS signal-tracking filter include a carrier replica phase error, a carrier replica Doppler error, and a code replica phase error. In the weak-lock configuration, the states of the GPS signal-tracking filter, which are directly related to code replica, carrier replica phase, and Doppler errors are applied to both replicas at a very low rate of 1.0 Hz as a correction. The steep gradients of the m-code autocorrelation function allow precise estimates of the code replica phase error but only with respect to a neighboring m-code autocorrelation peak. Tracking of a modal peak to achieve precise m-code replica phase error estimation introduces ambiguity with respect to which one of the autocorrelation peaks is currently being tracked.
[0004] An m-code envelope processor has a unimodal autocorrelation function with a single mode coincident with a center peak of the m-code autocorrelation function. An m-code envelope has only one correlation peak so tracking the m-code signal about the center peak is unambiguous. However, an m-code envelope does a low-grade gradient as compared to the steep gradient of the m-code autocorrelation function so tracking the m-code signal using only an m-code envelope is not precise, in a local sense, as with conventional early, prompt, and late m-code correlation taps.
[0005] M-code envelope processors fall into two classes. One class of m-code processor uses an m-code envelope signal tracking system to form an envelope using the m-code signal. An m-code processor can use conventional or specially configured harmonic m-code correlation taps. The m-code envelope processor with harmonic m-code correlators forms a smooth, monotonic, low-grade gradient envelope but the harmonic m-code correlators are not a conventional design adding complexity and cost to the m-code signal tracking system.
[0006] The second class of m-code processor is a coherent code envelope processor. The coherent code envelope processor uses a coherently tracked unimodal code, such as the c/a-code or p(y)-code, modulated on the same carrier. Coherently tracked means replicas for all codes, the m-code and other unimodal codes, are assigned such that the replica phase errors are the same. An ultra tightly coupled GPS tracking system provides coherent code tracking because replicas for all codes are computed using only a reference navigation solution and satellite ephemeredes. Coherently tracked codes do not have the steep gradient of the m-code autocorrelation function so tracking, in the local sense, is not as precise as with conventional early, prompt, and late m-code correlation taps. The m-code tracking processors suffer from either accurate but ambiguous tracking, or, unambiguous but inaccurate tracking. These and other problems are solved or reduced using the present invention.
[0007] An object of the invention is to provide a system for improved code phase tracking in m-code spread spectrum communication systems.
[0008] Another object of the invention is to provide a system for unambiguously tracking the GPS m-code signal using a low-grade gradient detected coherent envelope and for accurate tracking of the GPS m-code signal about a steep modal peak of the m-code autocorrelation function.
[0009] A further object of the invention is to provide a system for generating m-code phase tracking errors for m-code phase tracking of GPS m-code spread spectrum communication signals using an extended Kalman filter and a sequential probability ratio test for improved locking in a GPS navigation system.
[0010] The present invention can be implemented in a receiving communication system that code phase tracks a multimodal spread spectrum signal. An exemplar multimodal signal is the GPS m-code spread spectrum signal. The decorrelation of the multimodal chipping code is the m-code autocorrelation function having a plurality of peaks or modes. In a preferred form, an m-code envelope encompassing the peaks is first determined for determining which one of the modes is currently aligned to the current code phase for removing ambiguity as to which mode is being tracked and for updating the code phase to the center mode. The correlation function is then tracked about the steep gradient center peak of the correlation for maintaining accurate code phase lock. In the preferred form, the invention uses envelope detection with the m-code correlation taps of m-code correlators to achieve unambiguous m-code phase error detection using the unimodal peak of the envelope. The unambiguous m-code phase error detection is then combined with accurate tracking of the steep gradients of the multimodal m-code autocorrelation function for precise and unambiguous tracking of the m-code phase error. The envelope detection can be m-code envelope detection or coherent envelope detection, such as c/a-code envelope detection. In a first aspect of the invention, the m-code envelope can be generated directly from the peaks of the m-code correlation. Alternatively, as a second aspect of the invention, the envelope can be generated from or is an autocorrelation of a coherently received chipping code signal such as the preferred GPS c/a-code signal having an autocorrelation function that encompasses and envelops the multimodal peaks of the n-code autocorrelation. By first determining the code phase error of the coherently received spread spectrum signal, the code phase error is coherently aligned to the center mode of the multimodal signal. After alignment, the center mode code phase error is accurately tracked for maintaining the correct code phase of the multimodal signal. Both aspects of the invention are directed to n-code mode resolution. The first aspect of the invention includes r-code envelope detection using conventional m-code correlation taps. The second aspect uses a coherently tracked unimodal code, such as the GPS c/a-code or p-code. With either approach, an extended Kalman filter can be used to form a statistically independent sequence of residuals that is biased by the m-code mode. The system uses an extended Kalman filter and sequential probability ratio test m-code replica phase error estimation well suited for m-code spread spectrum communication systems such as GPS navigation systems. Statistical computations identify the correct mode and then to detect mode slips. A Wald sequential probability ratio test generates a conditional probability for each mode that the mode is the correct one. A Shiryayev sequential probability ratio test generates conditional probabilities that the mode has slipped from the nominal mode to another. The m-code modes are combined using the conditional probabilities as weights to form an m-code mode estimate. The m-code mode estimate is fed back to the extended Kalman filter that uses the m-code mode estimate to form an observation model. The computational burden is reduced because the system uses only one Kalman filter. These and other advantages will become more apparent from the following detailed description of the preferred embodiment.
[0011] FIG. 1 is a block diagram of a GPS inertial navigation system.
[0012] FIG. 2 is a block diagram of a weak-lock m-code envelope navigation processor.
[0013] FIG. 3 is a block diagram of an ultratight m-code envelope tracking system.
[0014] FIG. 4 is a block diagram of an adaptive m-code envelope tracking system.
[0015] FIG. 5 is a block diagram of an m-code envelope detector.
[0016] FIG. 6 is a block diagram of an m-code envelope-tracking filter.
[0017] FIG. 7 is a block diagram of an m-code mode identifier.
[0018] FIG. 8 is a block diagram of an ultratight coherent m-code and c/a-code navigation processor.
[0019] FIG. 9 is a block diagram of a coherent m-code and c/a-code tracking system.
[0020] FIG. 10 is a block diagram of a coherent m-code and c/a-code tracking filter.
[0021] An embodiment of the invention is described with reference to the figures using reference designations as shown in the figures. Referring to FIG. 1, a global positioning system (GPS) inertial navigation system includes a navigation system 10 receiving GPS signals and inertial measurement unit samples from a sensor assembly 12 and provides position and velocity data to a control and display unit 14. The navigation system 10 functions as a GPS inertial navigation system tracking GPS signals from a plurality of in-view satellites, not shown. The sensor assembly 12 includes an antenna 16 for receiving and providing received GPS signals and includes an inertial measurement unit (IMU) 18 providing the IMU sample signals, both of which are communicated to the navigation system 10. The navigation system 10 includes a downconverter 20 for frequency downconversion of the received GPS signals using a reference oscillator 24 providing an fo internal frequency reference, and includes an analog to digital (A/D) converter 22 communicating digitized GPS samples to a navigation processor 26. The navigation processor 26 also receives the INU samples and provides the position and velocity data to the control and display unit 14. The navigation system can be hosted on a vehicle for receiving a GPS signal from a satellite through the vehicular antenna 16.
[0022] Referring to FIGS. 1 and 2, and more particularly to FIG. 2, the navigation system 10 may be a weak-lock m-code envelope navigation processor that includes a navigation processor 26 receiving the digitized GPS samples 38. The samples 38 are communicated to mixers 34 and 36 for providing inphase (I) and quadrature (Q) signals respectively using cosine and sine demodulation signals from a carrier numerically controlled oscillator (NCO) 32. The NCO 32 is driven by a reference oscillator 30 feeding an fo reference signal to the mixers 34 and 36 for demodulation of the samples. The fo reference signal is also fed to a timer 56 for driving a GPS satellite position and velocity calculator 54. The cosine and sine demodulation signals are replica carrier signals for carrier demodulating the received GPS samples 38 that are also fed to an ephemeredes demodulator 58 for providing ephemerid data to the GPS satellite position and velocity calculator 54. The calculator 54 provides position and velocity data P-V of a transmitting satellite to a delta pseudorange and delta rate calculator 51 for providing delta pseudo range data to an m-code clock NCO 52 for sampling I and Q sample signals 38. The I and Q samples 38 are received by a correlator 40. The correlator 40 also receives early (CE), prompt (CP), late (CL), early-early (CEE), early-late (CEL), prompt-early (CPE), prompt-late (CPL), late-early (CLE), and late-late (CLL) code replica signals from an m-code chipping code generator 44 receiving a chipping code clocking signal from an m-code clock NCO 52. The chipping m-code may be a direct sequence spread spectrum code. The correlator 40 may operate, for example, at 1 KHz. The correlator 40 provides inphase and quadrature sample signals. The inphase and quadrature sample signals preferably include early Ime, Qme, prompt Imp, Qmp, late Iml, Qml, envelope early-early Ivee, Qvee, envelope early late Ivel, Qvel, envelope prompt-early Ivpe, Qvpe, envelope prompt-late Ivpl, Qvpl, envelope late-early Ivle, Qvle, and envelope late-late Ivll, Qvll. The correlated sample signals are communicated at a fixed rate, for example, at 1K Hertz into an adaptive m-code envelope tracking system 42. The adaptive m-code envelope tracking system 42 has, for example, a one Hertz system output 48. The system output 48 preferably includes a carrier phase error signal φe, a carrier phase error rate signal {dot over (φ)}e, a carrier frequency error rate signal {umlaut over (φ)}e, an m-code global range error signal rm, an amplitude state am signal, and an envelope state covariance matrix Pv. The m-code global range error rm is the desired unambiguous and accurate m-code code phase error. The carrier phase error rate {dot over (φ)}e, and the m-code global range error signal rm are communicated to the carrier NCO 32 for adjusting the cosine and sine downconverting carrier signals to the mixers 34 and 36 for coherent demodulation of the GPS samples 38.
[0023] The adaptive m-code envelope tracking system 42 receives I and Q sampled outputs from the 1KHz correlator 40. The adaptive m-code envelope tracking system 42 drives the NCOs 32 and 52, and the code generator 44, while providing an integrated error vector including signals φe, {dot over (φ)}e, {umlaut over (φ)}e, rm and am, and the envelope state covariance Pv as part of a system output 48. The φe, {dot over (φ)}e, and {umlaut over (φ)}e signals may, in the context of inertial navigation, respectively correspond to pseudorange, pseudorange rate, and pseudorange acceleration. The carrier frequency error {dot over (φ)}e is the first time rate of change derivative of the carrier phase error φe. The carrier frequency rate error {umlaut over (φ)}e is the second time rate of change derivative of the carrier phase error φe. The adaptive m-code envelope tracking system 42 also computes the signal amplitude state am, m-code global range error rm, and the envelope state covariance Pv as part of a measurement vector. The adaptive m-code envelope tracking system 42 computes the envelope state covariance Pv indicating the uncertainties in the measurement vector of errors for a number of samples m over a major cycle epoch time. The adaptive m-code envelope tracking system 42 can be used as a part of a weak-lock navigation processor using local carrier and code tracking loops for improved code and carrier tracking. The adaptive m-code envelope tracking system 42 can further be used in an ultratight method for GPS pseudorange computation tightly coupled with the correlation process to improve the ability to maintain tracking lock upon the received GPS signals.
[0024] In the exemplar weak-lock m-code envelope navigation processor, the adaptive m-code envelope tracking system 42 and GPS navigation calculator 46 provide the system outputs 48. The system outputs 48 include the receiver position and velocity output P/V. The m-code global range error rm is communicated to the m-code generator 44 in a local code phase tracking loop. Concurrently, the carrier frequency error {dot over (φ)}e is fed to the carrier NCO 32 in a carrier-tracking loop. An m-code prompt code phase φm from the m-code generator 44 and a carrier Doppler frequency ω1 from the carrier NCO 32 are communicated to the GPS navigation calculator 46. The carrier Doppler frequency ω1 may be for example, an L1 band Doppler frequency offset signal. The GPS navigation calculator 46 computes the best estimate of the navigation state vector solution defined by the vehicular position and velocity data P/V. The vehicular position and velocity data P/V is communicated to a delta pseudorange and delta rate calculator 51. The GPS navigation calculator 46 also computes user clock error estimates Cee communicated to the delta pseudorange and delta rate calculator 51. The delta pseudorange and delta rate calculator 51 receives the position and velocity data P-V of the satellite and the position and velocity data P/V of a vehicle hosting the navigation processor. The delta pseudorange and delta rate calculator 51 provides the delta pseudorange rate to the carrier NCO 32 and the delta pseudorange to the m-code NCO 52. The m-code NCO 52 also receives the fo reference clock for generating an m-code clock signal that is fed to the m-code generator 44. That is, in the weak-lock m-code envelope navigation processor, the GPS navigation calculator 46 provides the best estimate of the navigation state vector solution. The navigation state vector solution is indicated by the position and velocity data P/V communicated to the 100 Hz delta pseudorange and delta rate calculator 51 that, in turn, provides delta pseudorange rate data to the carrier NCO 32 and provides delta pseudorange data to the code NCO 52. The delta pseudorange and delta rate calculator 51 receives calculated GPS satellite position and velocity P-V data from a 100 Hz GPS satellite position and velocity calculator 54 and receives the clock error estimates Cee. The term P-V references the GPS satellite calculated position and velocity data. The clock error estimates Cee include a clock phase error and clock frequency error. The clock frequency error is the time derivative of the clock phase error. The calculated delta pseudorange is equal to the change in the geometric line-of-sight range to a satellite adjusted by the clock phase error estimates Cee and certain signal propagation compensation terms, such as the nominal ionospheric and tropospheric delays. The delta pseudorange is a change in a geometric range taken with respect to the one Hertz adaptive m-code envelope tracking system 42 updates applied to the carrier NCO 32 and the m-code generator 44. The geometric range is computed from the GPS calculated position and velocity data P-V and from the navigation calculated position and velocity data P/V. The delta pseudorange rate is equal to the change in the relative geometric velocity between the satellite and the vehicular antenna 16 plus the clock frequency error estimate. The GPS satellite position and velocity calculator 54 receives timing signals from a timer 56 using the clock reference fo 30 and receives demodulated ephemeris data from an ephemeris demodulator 58 using the GPS samples 38 to compute the GPS satellite position and velocity data P-V. The weak-lock formulation accommodates larger error between the replica signal and the received signal by observing the errors over many data samples and correcting the replica signal at a much lower rate than the sampling rate of the correlation process inputs.
[0025] Referring to FIGS. 1, 2, and 3, and more particularly FIG. 3, an exemplar ultratight m-code envelope navigation processor also receives the fo carrier frequency 30 and GPS samples 38. The ultratight m-code envelope navigation processor includes the NCOs 32 and 52, mixers 34 and 36, calculators 50 and 54, the timer 56, the ephemeredes demodulator 58, the m-code generator 44, and the correlator 40. The carrier NCO 32 is used for generating carrier replica signals for quadrature demodulation of the received GPS samples 38. The m-code generator 44 is again used for generating the early CE, prompt CP, late CL, early-early CEE, early-late CEL, prompt-early CPE, prompt-late CPL, late-early CLE, and late-late CLL code replica signals. The correlator 40 is again used for integrate and dump correlation of I and Q carrier demodulated signals correlated to the m-code replica signals. The correlation process is again based upon the replica m-code signal generation by the m-code generator 44 driving by a replica carrier generated by the carrier NCO 32.
[0026] As the pseudorange and pseudorange rate data from the calculator 50 is refreshed during each cycle, the replica carrier cosine and sine signals from the carrier NCO 32 are adjusted. Concurrently, the early, prompt, late, early-early, early-late, prompt-early, prompt-late, late-early, and late-late replica m-code signals from the m-code generator 44 are also adjusted under closed loop processing. The correlator 40 may operate at 1000 Hz. This 1000 Hz rate enables conventional correlation when receiving code replicas and samples. The code replicas include the CE, CP, CL, CEE, CEL, CPE, CPL, CLE, and CLL code replica signals. The samples include the demodulated correlated quadrature samples. The correlator 40 then generates the Ime, Qme, Imp, Qmp, Iml, Qml, Ivee, Qvee, Ivel, Qvel, Ivpe, Qvpe, late Ivpl, Qvpl, Ivle, Qvle, Ivll, Qvll quadrature correlation samples. The quadrature correlation samples are fed to the adaptive m-code envelope tracking system 60. The quadrature correlation samples contain amplitude information that can be used to estimate the code and carrier errors φe, {dot over (φ)}e, {umlaut over (φ)}e, rm, and am, and the envelope state covariance matrix Pv.
[0027] The ultratight navigation processor additionally receives IMU samples 64 that are communicated to a 100 Hz inertial navigation calculator 66 providing the receiver position and velocity outputs P/V 68. The adaptive m-code envelope tracking system 60 provides the measurement vector of errors including the signals φe, {dot over (φ)}e, {umlaut over (φ)}e, rm, and am, and the envelope state covariance Pv that are communicated to the one Hertz integration Kalman filter 62 having a one-second major cycle time period between Kalman filter epochs. The integration Kalman filter 62 in turn repetitively provides estimates of the position and velocity errors in the error state vector ESV that are used to compute the output position and velocity data P-V as position and velocity outputs 68 that are communicated to the control unit 14. The IMU samples 64 are communicated to the 100 Hz inertial navigation calculator 66 inertial navigation calculator 66 computes the best estimate of the navigation state vector solution defined by GPS receiver vehicular position and velocity data P/V. The vehicular position and velocity data P/V is calculated using the IMU samples 64 and the error state vector ESV. At the end of each of the one-second major cycles, the integration Kalman filter 62 updates the ESV and then communicates the updated ESV to the navigation calculator 66. The IMU samples 64 are repetitively generated, for example, one hundred times a second, and include differential velocity samples ΔV and differential attitude samples Δθ that are communicated to the 100 Hz inertial navigation calculator 66 also receiving the error state vector ESV every second from the integration Kalman filter 62. The integration Kalman filter 62 computes the user clock error estimates Cee that are communicated to the pseudorange and rate calculator 50. The 100 Hz inertial navigation calculator 66 is used for strapdown inertial navigation processing.
[0028] In the exemplar ultratight navigation processor, the navigation calculator 66 provides the best estimate of the navigation state vector solution. The navigation state vector solution indicates the position and velocity data P/V. The position and velocity data P/V is communicated to the 100 Hz pseudorange and pseudorange rate calculator 50. The pseudorange and pseudorange rate calculator 50 in turn provides the pseudorange rate data to the carrier NCO 32 and provides the pseudorange data to the m-code clock NCO 52. The pseudorange and pseudorange rate calculator 50 receives calculated GPS satellite position and velocity P-V data from the 100 Hz GPS satellite position and velocity calculator 54 and receives the user clock error estimates Cee. The clock error estimates Cee include a clock phase error and a clock frequency error that is the time derivative of the clock phase error. The calculated pseudorange is equal to the geometric line-of-sight range to a satellite adjusted by the clock phase error estimates Cee and signal propagation compensation terms, such as the nominal ionospheric and tropospheric delays. The geometric range is also computed from the GPS calculated satellite position and velocity data P-V and from the calculated vehicular position and velocity data P/V. The pseudorange rate is again equal to the relative geometric velocity to the satellite from the vehicular antenna 16 in addition to the clock frequency error estimate. The GPS satellite position and velocity calculator 54 receives timing signals from the timer 56 using the local clock reference fo 30 and receives demodulated ephemeris data from an ephemeris demodulator 58 using the GPS samples 38 to compute the GPS satellite position and velocity data P-V.
[0029] The GPS satellite position and velocity calculator 54 is again used for satellite GPS position and velocity calculation. The pseudorange and rate calculator 50 is used for line-of-sight pseudorange and pseudorange rate predictions for ultratight tracking whereas the delta pseudorange and delta rate calculator 51 is used for indicating changes in line-of-sight pseudorange and pseudorange rate predictions, which are aiding signals for weak-lock tracking. In the weak-lock navigation processor, the navigation processor is not coupled into the carrier-tracking loop 32, 44, 40, and 42. In the ultratight navigation processor, the navigation calculator 66 is ultratightly coupled into the carrier control loop 32, 44, 40, 42, 62, 66 and 50. That is, code and carrier errors φe, {dot over (φ)}e, {umlaut over (φ)}e, rm, and am, and the envelope state covariance Pv from the tracking system 60 are routed through the integration filter 62 and navigation calculator 66 so as to ultratightly couple the navigation calculator into the carrier-tracking loop. In either navigation processor application, the adaptive m-code envelope tracking system 42 or 60 is used to generate an envelope for determining which mode of the multimodal m-code is currently aligned to the m-code code phase for tracking the m-code phase error without ambiguity.
[0030] Referring to FIGS. 1 through 4, and more particularly to FIG. 4, an adaptive m-code envelope tracking system 42 or 60 comprises four components, including an m-code envelope detector 74, an m-code envelope tracking filter 76, an m-code mode identifier 78, and an m-code phase error calculator 80. The I and Q samples from the correlator 40 are divided into two sets of inputs 70 and 72. A first set of quadrature component inputs 70 are used for m-code tracking by the tracking filter 76 while the second set of quadrature component inputs 72 are used for m-code envelope detection by detector 74. Inputs 70 are designated as correlations ym as m-code inphase and quadrature correlations including m-code early correlations Ime and Qme, m-code prompt correlations Imp and Qmp, and m-code late correlations Iml and Qml. Inputs 72 are designated as correlations yv, are located in code phase time relative to each other, and are processed as m-code early-late tap correlation pairs disposed on side of modal peaks. Taps for each pair are placed λm/8 apart, where λm is the m-code chip width, and each pair is centered on a λm/2 early-prompt-late spacing. The m-code envelope correlations yv 72 are m-code envelope I and Q correlations including m-code envelope early-early correlations Ivee and Qvee, m-code envelope early-late correlations Ivel and Qvel, m-code envelope prompt-early correlations Ivpe and Qvpe, m-code envelope prompt-late correlations Ivpl and Qvpl, m-code envelope late-early correlations Ivle and Qvle, and m-code envelope late-late correlations Ivll and Qvll. The m-code envelope detector generates ye, yp, y1 envelope signals characterizing the m-code envelope. The ye, yp, y1 envelope signals are conmmunicated to the m-code tracking filter 76. The m-code tracking filter 76 provides and computes the carrier phase error φe output, the carrier frequency error {dot over (φ)}e the carrier frequency rate error {umlaut over (φ)}e, the signal amplitude state am, and the envelope state covariance matrix Pv. The tracking filter 76 also generates an m-code error covariance pme, an envelope covariance pv, an m-code envelope range error rv, and an m-code local range error rme that are fed to the m-code mode identifier 78. The m-code mode identifier generates a mode value nm indicating the mode that is currently aligned to the code phase φm. The m-code error covariance pme and the envelope covariance pv are measures of uncertainty. The m-code local range error rme is a first estimate of the m-code global range error rm where rm=rme+nmλm/4. The m-code global range error rm is provided as part of the output 82. The output 82 includes the signals φe, {dot over (φ)}e, {umlaut over (φ)}e, rm, am, and Pv generated by the tracking filter 76. The r-code local range error rme is a code phase distance to a modal peak. In operation, the detector 74 first detects the m-code envelope and then the m-code tracking filter 76 generates the m-code local range error rme from which the mode identifier 78 generates the mode value nm. The m-code local range error rme and value nm are then used to determine the m-code global range error rm. As such, the m-code envelope is used to determine which mode is currently aligned to current code phase φm and used to generate the n-code global range error rm from the mode value nm in the presence of multiple modes. The n-code envelope along with the mode value is used to eliminate modal tracking ambiguity by virtue of the envelope being inherently and coherently aligned to the modal correlation peaks.
[0031] Referring to FIGS. 1 through 5, and more particularly to FIG. 5, inputs to the m-code envelope detector 74 are from the m-code envelope correlations yv input 72. The n-code envelope detector 74 provides r-code envelope detector outputs 104 including ye, yp, and y1 that are determined by respective envelope detector ye, yp, and y1 equations.
[0000] y e=(I vee 2 +Q vee 2)1/2+(I vel 2 +Q vel 2)1/2 [0000] y p=(I vpe 2 +Q vpe 2)1/2+(I vpl 2 +Q vpl 2)1/2 [0000] y l=(I vle 2 +Q vle 2)1/2+(I vll 2 +Q vll 2)1/2 [0032] Referring to FIGS. 1 through 6, and more particularly to FIG. 6, the m-code envelope tracking filter 76 can be implemented as an extended Kalman filter. The m-code envelope tracking filter 76 receives the ym m-code correlations 90 including ye, yp, and y1 from the envelope detector 74 and the m-code mode value nm 92 from the mode identifier 78. The m-code envelope tracking filter 76 comprises a measurement residual generator 94, a state estimator 96, a state error covariance estimator 98, and a state propagator 100 for generating the carrier phase error φe, the carrier phase error rate {dot over (φ)}e, the carrier frequency error rate {umlaut over (φ)}e, the amplitude state am, the envelope state covariance matrix Pv, the local covariances pme and pv, the m-code envelope range error rv, and the m-code local range error rme. The m-code envelope range error rv and the m-code local range error rme are both state estimates. A measurement residual pv is computed at 1000 Hz by the measurement residual generator 94 from the m-code envelope detector ye, yp, and y1 outputs 90, the m-code mode value nm 92 that is an m-code mode identifier output, and a propagated state x v using a measurement residual equation.
[0033] Each component of the measurement residual ρv depends on the propagated state x v. The propagated state x v has six components where x v=[ φ e, {dot over ( φ e, {umlaut over ( φ e, r me, r v, am]. The propagated state includes a propagated carrier phase error state φ e expressed in radians, propagated carrier phase error first and second order rates {dot over ( φ e and {umlaut over ( φ e, a propagated m-code local range error state r me expressed in meters, a propagated m-code envelope range error state r v also expressed in meters, and a propagated m-code amplitude state ām. Each component of the measurement residual ρv also depends on the m-code mode value nm, an early to late replicated code offset δm, an m-code envelope early to late offset δv and an m-code navigation message bit dm. The measurement residual generator 94 uses an m-code correlation function described by a cm(r) equation.
[0034] The m-code correlation function depends on the pseudo-random noise (PRN) m-code chip width λm and a signal structure with a square wave of period λm/2 superimposed on each chip. The square wave creates piecewise linear regions defined by the m-code local range error. The m-code chip width is defined by a nominal m-code chipping rate fm=5.115�106 chips per second. The speed of light used for GPS applications is c=2.99792458�108 meters per second so the nominal m-code chip width is λm=(c/fm)=58.61 m. The m-code chipping rate apparent to the receiver differs slightly from fm due to user to satellite relative motion and atmospheric transmission effects.
[0035] The m-code early to late replication offset δm is taken to be one-eighth chip from the prompt so that in perfect code lock, the early and late code replicas are each halfway from the center correlation peak and the first negative-correlation troughs, that is, δm=(λm/8)≅58.61 m. The m-code navigation message bit dm=�−1 may have a nominal 5.0 millisecond or 20.0 millisecond period and is determined by a conventional arctangent bit detection algorithm.
[0036] An ideal m-code envelope correlation function is smooth and monotonic using a harmonic code replica. The m-code envelope detector ye, yp, and y1 outputs 90 determine an approximate m-code envelope function cvapproximate(r)=|cm(r+λm/16)|+|cm(r−λm/16)|. The approximate m-code envelope function is piecewise smooth and not monotonic but the implementation is simpler as the m-code envelope function uses conventional square-wave code replicas and the m-code envelope function provides a suitable lower bound for expected performance. A simple four-segment piecewise-linear form that is monotonic and captures the large-scale shape of the approximate m-code envelope function is the m-code envelope correlation function described by a cv(r) equation.
[0037] A state estimator 96 computes the state estimate xv at 1000 Hz. The output of the state estimator include φe, {dot over (φ)}e, {umlaut over ( φ e, rme, rv, am of output 102. The inputs to the state estimator include the measurement residual μv, the propagated state x v, the m-code mode value nm 92, and the envelope state covariance pv using a state update equation xv= x v+Kvρv. The state update equation includes a gain Kv that is computed as an extended Kalman filter gain to minimize the error variance, includes a measurement covariance Vv, and includes the measurement residual ρv. The Kalman filter gain Kv is computed as an extended Kalman filter gain to minimize the error variance using a measurement sensitivity hv. A Kalman filter gain equation Kv=Pvhv TVv −1 defines the Kalman filter gain. The measurement sensitivity hv is defined by a hv partitioned matrix.
[0038] The measurement sensitivity hv has five measurement sensitivity partition terms including hmφ, hmr, hma, her, and hea.
[0039] The measurement sensitivity partition term hmr depends on an m-code correlation sensitivity function defined by a c′m(r) sensitivity equation.
[0040] The m-code correlation sensitivity function c′m(r) is derived from the m-code correlation function cm(r). The measurement sensitivity partition term her depends on an m-code envelope correlation sensitivity function defined by a c′v(r) equation.
[0041] The m-code envelope correlation sensitivity function c′v(r) is derived from the m-code envelope correlation function cv(r) as a simple four-segment piecewise-linear curve to capture the large-scale envelope shape. The m-code measurement covariance Vm is the covariance of the additive noise associated with the noise of m-code correlation ym that has a white Gaussian distribution and is correlated with respect to the early, prompt, and late signals, but is uncorrelated both in time and with respect to the inphase and quadrature signals. The m-code measurement covariance is derived directly from the m-code correlation function. The second measurement covariance term is an m-code envelope measurement covariance Ve where Ve=2am −2/ηm. The m-code envelope measurement covariance is the covariance of the additive noise associated with the m-code envelope detector ye, yp, and y1 outputs 90.
[0042] A state propagator 100 computes a propagated state x v at 1000 Hz from a state estimate xv. The state estimate xv has six components where xv=[φe, {dot over (φ)}e, {umlaut over (φ)}e, rme, rv, am]. Again, the components of the state estimate xv include the carrier phase error state estimate φe expressed in radians, the carrier phase error first and second order rate state estimates {dot over (φ)}e and {umlaut over (φ)}e, the m-code local range error state estimate rme expressed in meters, the m-code envelope range error state estimate rv expressed in meters, and the m-code amplitude state estimate am. The m-code local range error state estimate rme is a replicated m-code phase error expressed in meters as indicated by the early, prompt, and late m-code correlator taps. The m-code envelope range error state estimate rv is a replicated m-code phase error expressed in meters as indicated by the early, prompt, and late m-code envelope detector ye, yp, and y1 outputs 90.
[0043] The noise components of the ym inputs 70 and yv inputs 90 are statistically independent and hence uncorrelated so that a partitioned matrix form describes a measurement covariance Vv defined by a Vv measurement covariance equation.
[0044] The measurement covariance has two measurement covariance partition terms. The first measurement covariance term is an m-code measurement covariance Vm.
[0045] The m-code measurement covariance Vm depends on a normalized m-code offset δ=δm/λm that is constrained by 0< δ m<⅛. The m-code measurement covariance Vm also depends on the propagated m-code amplitude state ām that is a component of the propagated state x v, and a signal to noise power ratio ηm.
[0046] The state propagator 100 computes the propagated state x v using linear state propagator relations, including φ e=φe+{dot over (φ)}e δt+{umlaut over (φ)}eδt2/2, {dot over ( φ e={dot over (φ)}e+{umlaut over (φ)}eδt, {umlaut over ( φ e={umlaut over (φ)}e, r me=rme=(λL1/2π){dot over (φ)}eδt, r v=rv+(λL1/2π){dot over (φ)}eδt, and ām=am. The state propagator relations require a propagation time δt=0.001 seconds to match the 1000 Hz calculation rate. The state propagator relations also require an L1 band carrier wavelength, λL1 to be equal to 0.1903 meters. The L1 band carrier wavelength is derived from the GPS L1 band carrier frequency with ωL1=1.57542�109 Hz, and the speed of light c used for GPS applications where c=2.99792458�108 meters per second, as λL1=c/ωL1==0.1903 meters.
[0047] A state error covariance estimator 98 provides the envelope state covariance Pv at 1000 Hz using the propagated state components r me, r v, and ām, and the m-code mode nm. The envelope state covariance Pv is a six-by-six symmetric and positive-definite matrix computed in two stages, a propagation stage and an update stage. In the propagation stage, a state covariance propagation P v is computed using a state covariance propagation relation P v=FvPvFv T+Qv. The state covariance propagation relation depends on a six by six state propagation matrix Fv.
[0048] The state propagation matrix depends on a propagation time δt=0.001 seconds and an L1 band carrier wavelength, λL1=0.1903 meters, which are the same as those specified for the state propagator 100. The state covariance propagation relation also depends on Qv that is a six by six state process noise covariance. The state process noise covariance is determined by design iteration but may be set as the exemplary diagonal matrix Qv=diag[0.02, 0, 0.0003, 0, 0, 0]. On the first application of the propagation stage, the envelope state covariance Pv is a six by six matrix determined by design iteration but may be set as the exemplary diagonal matrix Pv=diag[π, 10π, 60π, 30, 30, 0.2]. In the update stage, the envelope state covariance Pv is computed using an envelope state covariance update relation Pv=( P v −1+hv TVv −1hv)−1. In the envelope state covariance update relation, the measurement sensitivity hv and the measurement covariance Vv are the same as those specified for the state estimator 96. Some of the hv measurement sensitivity components, namely hmφ, hmr, and hma depend on the propagated carrier phase error state φ e through sin φ e and cos φ e terms. But the symmetry of the measurement covariance Vv makes all the sin φ e and cos φ e terms combine in the product hv TVv −1hv as sin2 φ e =+cos2 φ e=1. The product hv TVv −1hv in the state covariance update relation does not depend on φ e even though the hv does depend on φ e. Thus, the state covariance estimator 98 does not use φ e as an input signal.
[0049] Referring to FIGS. 1 through 7, and more particularly to FIG. 7, an m-code mode identifier 78 is an implementation of a Shiryayev sequential probability ratio test algorithm. Inputs 106 include an m-code local range error state estimate rme and an m-code envelope range error state estimate rv, and the respective covariances Pme and Pv. The m-code local range error state estimate rme, the m-code envelope range error state estimate rv, both taken from the state estimator 96 of the m-code envelope tracking filter 76. The covariances Pme and Pv that are taken from the state covariance estimator 98 of the m-code envelope tracking filter 76. The output 120 is the m-code mode value nm, that is, the mode nm, that is in the range −4<nm<4 for indicating the range of modes. The integer difference between the minus four and plus four indicates seven modal peaks, one of which is indicated by the modal value nm. The m-code mode nm and m-code local range error rme define an exhaustive and mutually exclusive set that is a partition of regions in the domain of possible range errors. When the partition is considered a set of hypotheses, then each hypothesis is computed with the probability conditioned on a sequence of m-code local range error state estimates rme and m-code envelope range error state estimates rv.
[0050] In particular, nine hypotheses are defined such that the m-code mode nm has integer values including H0={nm=nm0=0}, H1={nm=nm1=1}, H2={nm=nm2=−1}, H3={nm=nm3=2}, H4={nm=nm4=−2}, H5={nm=nm5=3}, H6={nm=nm6=−3}, H7={nm=nm7=4}, H8={nm=nm8=−4}. Also, rk=[rme, rv] are a measurement at a time designated tk with Zk={r1, r2, . . . , rk} being a sequence of measurements, and with θj being a time at which hypothesis Hj is true, at an unknown time. For each hypothesis Hj, a conditional probability Fj is computed. The conditional probability Fj is the probability that a switch to hypothesis Hj occurs at or before time tk conditioned on all measurements up to time tk, that is, Fj=P(θj≦tk|zk). The expectation is that at some time, one of the nine Fj probabilities will be large enough that a confidence is established for an associated hypothesis.
[0051] The m-code mode identifier 78 has twenty components including nine instances of an m-code mode conditional density calculator as m-code mode conditional density calculators 108 and 110, nine instances of an m-code mode probability propagator as m-code mode probability propagators 112 through 114, an m-code mode probability updater 116, and an m-code mode probability estimator 118. An m-code mode conditional density calculator calculates an m-code mode conditional probability density fj associated with hypothesis Hj. The m-code mode conditional probability density fj is computed using a probability density function for the measurement at time tk and conditioned on hypothesis Hj. The m-code mode conditional probability density fj is computed as a Gaussian probability density using an m-code mode conditional density calculator of the calculators 108 through 110 and a probability density equation fj=exp[(rv−rme−njλm/4)2/2pme+2pv)]/√[2π(pme−pv)].
[0052] Inputs to the m-code mode conditional density calculators 108 through 110 are the m-code local range error state estimate rme and m-code envelope range error state estimate rv, and the respective covariances pme and pv. The m-code local range error state estimate rme and m-code envelope range error state estimate rv are taken from the state estimator 96 of the m-code envelope-tracking filter 76. The covariances pme and pv are taken from the state covariance estimator 98 of the m-code envelope-tracking filter 76. Each of the m-code mode conditional density calculators 108 through 110 uses one of the nine hypothesized m-code mode integers including n0=0, n1=1, n2=−1, n3=2, n4=−2, n5=3, n6=−3, n7=4, and n8=−4. Each of the m-code mode conditional density calculators 108 through 110 also requires an m-code chip width, λm=58.61 meters. The m-code chip width is derived from the GPS m-code chipping rate, fm=5115000 chips per second, and the speed of light c used for GPS applications where c=2.99792458�108 meters per second, as λm=c/fm=58.61 meters.
[0053] Each of the m-code mode probability propagators 112 through 114 provides a propagated probability φj associated with one of the nine hypotheses Hj as the probability that a switch to hypothesis Hj occurs at or before time tk+1 conditioned on all measurements up to time tk, that is, φj=P(θj≦tk+1|Zk). A propagated probability φj is computed by the m-code mode probability propagator 112 and 114 using an m-code mode probability propagator relation φj=Fj+p(1−Fj). An input to each of the m-code mode probability propagators 112 through 114 is an updated probability Fj that is the output of the m-code mode probability updater 116.
[0054] The m-code mode probability propagators 112 through 114 also use a switching probability p. The switching probability p explicitly allows for the possibility of a switch to hypothesis Hj by defining p as the probability of a switch to hypothesis Hj during any measurement interval, that is, p=P(t0<θj<t1)=P(θj>t0, θj<t1). Before a most likely hypothesis is identified, all switching probabilities are set to zero. Setting the switching probabilities to zero models the true hypothesis as constant with respect to all measurements. As such, the m-code mode identifier 78 becomes a Wald sequential probability ratio test. Later in the test, once a most likely hypothesis is identified, the switching probability is set to a small non-zero value as determined by design iteration or by the exemplar p=0.0001. Allowing for a hypothesis switch makes the m-code mode identifier 78 a Shiryayev sequential probability ratio test that is a special case of the Wald sequential probability ratio test.
[0055] On the first application of the m-code mode probability propagators 112 through 114, the updated probability Fj is assigned as Fj=P(θj≦t0)=πj, that may be determined by design iteration or by the exemplar πj=1/9. An m-code mode probability updater 116 computes the updated probability Fj. The updated probability is switched to hypothesis Hj occurring at or before time tk and conditioned on all measurements up to time tk, that is, Fj=P(θj≦tk|Zk). The expectation is that at some time, one of the probabilities will be large enough that a confidence may be established in announcing the associated hypothesis. An updated probability Fj associated with one of the nine hypotheses Hj is computed by the m-code probability updater 116 using an m-code probability updater relation described in an Fj probability equation.
[0056] Inputs to the m-code probability updater 116 are the m-code mode conditional probability density fj from the r-code mode conditional density calculators 108 through 110 and the propagated probability φj from the r-code mode probability propagators 112 through 114. The m-code mode probability propagators 112 through 114 and the r-code mode probability updater 116 form a recursion rule function for computing the updated probabilities Fj at each time tk that ensures the updated probabilities Fj are consistent with the definition of a probability.
[0057] The m-code mode value nm is computed by the r-code mode probability estimator 118 using an r-code mode probability estimator relation nm=F0n0+F1n1+F2n2+F3n3+F4n4+F5n5′F6n6+F7n7+F8n8. The nine updated probabilities F0, F1, F2, F3, F4, F5, F6, F7, and F8 are inputs to the m-code mode probability estimator 118. The m-code probability updater 116 computes the nine updated probabilities F0, F1, F2, F3, F4, F5, F6, F7, and F8. The form of the m-code mode probability estimator ensures that the m-code mode identifier output nm 120 is bounded by −4≦nm≦4.
[0058] An m-code global range error rm is computed by an m-code phase error calculator 80 using an r-code phase error relation rm=rme+nmλ m/4. Inputs to the r-code phase error calculator 80 are the r-code local range error state estimate rme, taken from the m-code envelope-tracking filter 76, and the m-code mode nm taken from the r-code mode identifier 78. The r-code phase error calculator 80 also requires an n-code chip width λm=58.61 meters, the same r-code chip width used by the n-code mode conditional density calculators 108 through 110 of the m-code mode identifier 78.
[0059] The adaptive n-code envelope tracking system has outputs 82. The output 82 includes the carrier phase error state estimate φe and the rates {dot over (φ)}e and {umlaut over (φ)}e the signal amplitude state estimate am, the m-code global range error rm, and the envelope state covariance matrix Pv. Of the outputs 102, the carrier phase error state estimate φe and the rates {dot over (φ)}e and {umlaut over (φ)}e, the signal amplitude state estimate am, and the envelope state covariance Pv are taken from the outputs 102 of the m-code envelope tracking filter 76.
[0060] Referring to FIGS. 1 through 8, and more particularly to FIG. 8, the navigation system 10 may be an ultratight coherent n-code and c/a-code navigation processor. The ultratight coherent r-code and c/a-code navigation processor also receives the carrier frequency fo 30, GPS samples 38, and the IMU samples 64. The ultratight navigation processor also includes the NCO 32, mixers 34 and 36, calculators 50, 54 and 66, the timer 56, and the ephemeredes demodulator 58. The ultratight navigation processor additionally receives the IMU samples that are communicated to the 100 Hz inertial navigation calculator 66 providing position and velocity data P/V 68. The I and Q signals respectively from mixers 34 and 36 are received by a correlator 120 receiving early (E), prompt (P), and late (L) code replica signals from an m-code chipping code generator 124 receiving a chipping code clocking signal from an m-code clock NCO 130. The correlator 120 also receives early, prompt, and late code replica signals from a c/a-code chipping code generator 126 receiving a chipping code clocking signal from a c/a-code clock NCO 128. The NCOs 128 and 130 are clocked by the clock reference fo 30 and adjusted by the pseudorange from the pseudorange and rate calculator 50. The chipping m-code and the chipping c/a-code are direct sequence spread spectrum codes. The correlator 120 may operate, for example, at 1 KHz. The correlator 120 provides both inphase and quadrature, early Ime and Qme m-code correlations, prompt Imp and Qmp m-code correlations, late Iml and Qml m-code correlations, early Ice and Qce c/a-code correlations, prompt Icp and Qcp c/a-code correlations, and late Icl and Qcl c/a-code correlations. The inphase and quadrature correlations are GPS quadrature correlation samples. The quadrature correlation sample signals are communicated, for example, at 1K Hertz, into an adaptive coherent m-code and c/a-code tracking system 122 providing, for example, a one Hertz output. The adaptive coherent m-code and c/a-code tracking system 122 can receive the quadrature correlation samples from one or more correlators 120 to accommodate dual f1 and f2 frequency integration. For dual f1 and f2 frequency integration, two respective NCOs 32, 128, and 130 and two respective code generators 124 and 126 may be used for providing an integrated error vector to the single system output 68 or to the single integration Kalman filter 62. The adaptive coherent m-code and c/a-code tracking system 122 computes the carrier phase error state estimate φe and the rates {dot over (φ)}e and {umlaut over (φ)}e, the m-code signal amplitude state am, the c/a-code signal amplitude state ac, an m-code global range error rm, a c/a-code range error rce, and a coherent m-code and c/a-code state covariance Pc as a measurement vector output. The measure vector output is communicated to the integration filter 62. The carrier phase error φe and the rates {dot over (φ)}e and {umlaut over (φ)}e may, in the context of inertial navigation, respectively correspond to a pseudorange, pseudorange rate, and pseudorange acceleration. The adaptive coherent m-code and c/a-code tracking system 122 computes the coherent state error covariance Pc indicating the uncertainties in the measurement output vector of errors for a number of samples m over a major cycle epoch time. The tracking system 122 can be used in an ultratight navigation processor for GPS pseudorange computation tightly coupled with the correlation process to improve the ability to maintain tracking lock upon the received GPS signals 38.
[0061] In the exemplar ultratight coherent m-code and c/a-code navigation processor, the tracking system 122 measurement output vector of errors and the coherent state error covariance Pc are communicated to the one Hertz integration Kalman filter 62 having a one-second major cycle time period between Kalman filter epochs. The 100 Hz pseudorange and pseudorange rate calculator 50 provides pseudorange rate data to the carrier NCO 32 and provides pseudorange data to the c/a-code clock NCO 128 and the m-code clock NCO 130. The m-code generator 124 is used for early, prompt, and late m-code replica signal generation. The c/a-code generator 126 is used for early, prompt, and late c/a-code replica signal generation. The correlator 120 is used to integrate and dump correlations of the I and Q carrier demodulated signals correlated with the early, prompt, and late m-code replica signals and the early, prompt, and late c/a-code replica signals.
[0062] The correlation process is based upon the early, prompt, and late replica m-code signal generation by the m-code generator 124, upon early, prompt, and late replica c/a-code signal generation by the c/a-code generator 126, and upon the replica carrier generation by the carrier NCO 32. The pseudorange and pseudorange rate data from the calculator 50 are refreshed during each cycle. During each refresh cycle, the replica carrier cosine and sine signals from the carrier NCO 32, the early, prompt, and late replica m-code signals from the m-code generator 124, and the early, prompt, and late replica c/a-code signals from the c/a-code generator 126 are adjusted under closed loop processing. The correlator 120 may operate at 1000 Hz enabling conventional correlation by providing the early, prompt, and late I and Q samples to the adaptive coherent m-code and c/a-code tracking system 122. The I and Q samples contain amplitude information that can be used with signal correlation functions to estimate the code and carrier errors. The carrier and code phase errors, that are code phase offsets of the replica signals as compared to the received GPS sample signals, can be determined as offset knowledge of the signal correlation function. The carrier and code phase offsets are used to estimate the residual errors.
[0063] Referring to FIGS. 1 through 9, and more particularly to FIG. 9, the adaptive coherent m-code and c/a-code tracking system 122 has three components, including a coherent m-code and c/a-code tracking filter 144, an m-code mode identifier 146, and an m-code phase error calculator 80. The inputs for the adaptive coherent m-code and c/a-code tracking system 144 are taken in two groups including an m-code correlation ym 140 and a c/a-code yca correlation 142. The m-code correlation ym 140 includes the m-code early correlations Ime and Qme, m-code prompt correlations Imp and Qmp and m-code late correlations Iml and Qml. The c/a-code yca correlation 142 includes the c/a-code early correlations Ice and Qce, c/a-code prompt correlations Icp and Qcp, and c/a-code late correlations Icl and Qcl. The adaptive coherent m-code and c/a-code tracking system 144 provides the carrier phase error φe, the carrier frequency error {dot over (φ)}e, the carrier frequency rate error {umlaut over (φ)}e, the m-code signal amplitude state am, the c/a-code signal amplitude state ac, the m-code global range error rm, the c/a-code range error rce, and the coherent m-code and c/a-code state covariance Pc as outputs 148. In operation, the coherent m-code and c/a-code tracking filter 144 uses the autocorrelation function of the c/a-code as a coherent envelope to the multimodal correlation function of the m-code. The filter 144 first determines the c/a-code range error rce. The c/a-code range error rce is then used by the m-code mode identifier 146 to determine the mode value nm. The mode value nm and the m-code local range error rme is then used to determine m-code global range error rm. In this manner, a coherent envelope is used to remove modal ambiguity as well as generating the sought after precise m-code global range error rm for precise coherent despreading of the m-code upon reception of the m-code signal.
[0064] Referring to all of the Figures, and more particularly FIG. 10, the coherent m-code and c/a-code tracking filter 144 is again an implementation of an extended Kalman filter. Inputs to the coherent m-code and c/a-code tracking filter are the m-code ym correlation 140, the c/a-code yca correlation 142, and an m-code mode value nm that is an output of the m-code mode identifier 146. The coherent m-code and c/a-code tracking filter 144 has four components including a measurement residual generator 152, a state estimator 154, a state error covariance estimator 156, and a state propagator 158. A measurement residual Pc is computed at 1000 Hz by the measurement residual generator 152 from the m-code ym correlation 140, from the c/a-code yca correlation 152, from the m-code mode nm input 150, and from a propagated state x c using a measurement residual generator relational equation.
[0065] Each component of the measurement residual ρc depends on the propagated state x c. The propagated state x c has seven components where x c=[ φ e, {dot over ( φ e, {umlaut over ( φ e, rme, rce, am, aca]. The propagated state x c includes the propagated carrier phase error state φ e expressed in radians, propagated carrier phase error first and second order rate states {dot over ( φ e and {umlaut over ( φ e, the propagated m-code local range error state rme expressed in meters, a propagated c/a-code range error state r ce also expressed in meters, the propagated m-code amplitude state ām, and a propagated c/a-code amplitude state āca. Each component of the measurement residual ρc also depends on the m-code nm mode value 150, an early to late replicated m-code offset δm, an early to late replicated c/a-code offset δca, an m-code navigation message bit dm, and a c/a-code navigation message bit dca.
[0066] The measurement residual generator 152 uses the c/a-code autocorrelation function cca(r)=max(1″|r|/λca, −1/λca). The c/a-code correlation function depends on the pseudorandom noise c/a-code chip width λca. The c/a-code chip width λca is defined by a nominal c/a-code chipping rate fca=1023000 chips per second. The speed of light used for GPS applications is c=2.99792458�108 meters per second so the nominal c/a-code chip width is λca=(c/fca)=293.05 meters. The c/a-code chipping rate apparent to the receiver differs slightly from fca due to user to satellite relative Doppler motion and atmospheric transmission effects. The c/a-code early to late replication offset δca is taken to be one-half chip from the prompt replica so that in perfect code lock, the early and late code replicas are each halfway from the center correlation peak and zero correlation, that is, δca=λca/2≅146.53 meters. The c/a-code navigation message bit dca=�1 has a nominal 20.0 millisecond period and is determined by a conventional arctangent bit detection algorithm. The c/a-code navigation message bit dca may be a different stream than that modulated on the m-code signal.
[0067] A state estimate xc is computed at 1000 Hz by a state estimator 154 from the measurement residual ρc, the propagated state x c, the m-code mode nm 150, and the coherent state error covariance Pc using a state estimator relation xc=xc+Kcρc. The state estimator 154 uses a gain Kc, computed as an extended Kalman filter gain to minimize the error variance. The gain Kc is computed using a measurement sensitivity hc, a measurement covariance Vc, and a Kalman filter gain equation Kc=Pchc TVc −1. The measurement sensitivity hc is a partitioned matrix.
[0068] The measurement sensitivity hc has six partition terms including hmφ, hmr, hma, hcφ, hcr, and hca. The terms hmφ, hmr, and hma have the same form as terms hcφ, hcr, and hca that are defined by respective hcφ, hcr, and hca matrices.
[0069] The measurement sensitivity hc uses a c/a-code correlation function that is a four segment piecewise linear curve described by a cca(r) equation.
[0070] The measurement sensitivity partition term hcr depends on a c/a-code correlation sensitivity function derived from the c/a-code correlation function cca(r) and defined by a c′ca(r) equation.
[0071] The noise components are statistically independent and therefore uncorrelated so that the measurement covariance Vc is a partitioned matrix.
[0072] The measurement covariance has two measurement covariance partition terms. The first measurement covariance term is the m-code measurement covariance Vm. The second measurement covariance term is a c/a-code measurement covariance Vca defined by a Vca matrix.
[0073] The c/a-code measurement covariance Vca depends on a normalized c/a-code offset δ ca=δca/λca, which is constrained as 0≦δca≦�. The c/a-code measurement covariance Vca also depends on the propagated c/a-code amplitude state āca that is a component of the propagated state x c, and a c/a-code signal to noise power ratio ηca. The c/a-code measurement covariance Vca is the covariance of the additive noise associated with the c/a-code correlation yca=[Ice, Qce, Icp, Qcp, Icl, Qcl] 142. The additive noise has a white Gaussian distribution and is correlated with respect to the early, prompt, and late signals, but is uncorrelated both in time and with respect to the I and Q signals. The c/a-code measurement covariance Vca is derived directly from the c/a-code correlation function.
[0074] A state propagator 158 computes a propagated state x c at 1000 Hz from the state estimate xc from the state estimator 154. The propagated state x c is a state vector [ φ e, {dot over ( φ e, {umlaut over ( φ e, rme, rce, ām, āca]. The state estimate xc has seven components including [φe, {dot over (φ)}e {umlaut over (φ)}e, rme, rce, am, aca]. The state estimate xc includes a replicated carrier phase error state estimate φe expressed in radians, first and second rates {dot over (φ)}e and {umlaut over (φ)}e, an m-code local range error state estimate rme, a c/a-code range error state estimate rce, an m-code signal amplitude state estimate am, and a c/a-code signal amplitude state estimate aca. The m-code local range error state estimate rme is a replicated m-code phase error expressed in meters as indicated by the early, prompt, and late m-code correlator taps. The c/a-code range error state estimate rce, is a replicated c/a-code phase error expressed in meters as indicated by the early, prompt, and late c/a-code correlator taps.
[0075] The state propagator 158 computes the propagated state x c using linear state propagator relations. The linear state propagator relations include φ e=φe+{dot over (φ)}eδt+{dot over (φ)}eδt2/2, {dot over ( φ e={dot over (φ)}e+{umlaut over (φ)}eδt, {umlaut over ( φ e,, r me=rme+(λL1/2π){dot over (φ)}eδt, r ce=rce+(λL1/2π){dot over (φ)}eδt, ām=am, and āca=aca. The linear state propagator relations require a propagation time δt=0.001 seconds to match the 1000 Hz calculation rate of the state propagator. The linear state propagator relations also require an L1 band carrier wavelength, λL1=0.1903 meters. The L1 band carrier wavelength is derived from the GPS L1 band carrier frequency, ωL1=1.57542�109 Hz, and the speed of light c used for GPS applications, c=2.99792458�108 meters per second, as λL1=c/ωL1=0.1903 meters.
[0076] A state error covariance estimator 156 provides a state error covariance estimate Pc at 1000 Hz using the propagated state components r ce, r ce, ām, and āca, and the m-code mode value nm. The state error covariance estimate Pc is a seven by seven symmetric and positive-definite matrix computed in two stages, a propagation stage and an update stage. In the propagation stage, the state error covariance estimate P c is computed using a state error covariance propagation relation P c=FcPcFc T+Qc. The state error covariance propagation relation depends on a seven by seven state transition matrix Fc.
[0077] The state transition matrix Fc depends on a propagation time δt=0.001 seconds and an L1 band carrier wavelength, λL1=0.1903 meters, which are the same as those specified for the state propagator 100. The state error covariance propagation relation also depends on a seven-by-seven state process noise covariance Qc. The state process noise covariance Qc is determined by design iteration but may be set as the exemplary diagonal matrix Qc=diag[0.02, 0, 0.0003, 0, 0, 0, 0]. In the update stage, the state error covariance estimate Pc is computed using a state error covariance update relation Pc=( P c −1′hc TVc −1hc)−1. In the state error covariance update relation, the measurement sensitivity hc and the measurement covariance Vc are the same as those specified for the state estimator 96. The hc measurement sensitivity components, namely hmφ, hmr, hma, hcφ, hcr, and hca depend on the propagated carrier phase error state φ e through sin φ e and cos φ e terms. However, the symmetry of the measurement covariance Vc makes all the sin φ e and cos φ e terms combine in the product hc TVc −1hc as sin2 φ e+cos2 φ e=1. The product hc TVc −1hc in the state error covariance update relation does not depend on φ e even though the hc does depend on φ e. Thus, the state error covariance estimator 156 does not indicate φ e as an input. On a first calculation, Pc is a seven by seven matrix determined by design iteration but may be set as the exemplary diagonal matrix Pv=diag[π, 10π, 60π, 30, 30, 0.2, 0.2].
[0078] The m-code mode identifier 146 is the same as that for the adaptive m-code envelope tracking system 42 and 60 except that the m-code envelope range error rv and the covariance pv are respectively replaced by the c/a-code range error rce, and the c/a-code range error covariance pce. The c/a-code range error rce and the c/a-code range error covariance pce are outputs of the m-code and c/a-code tracking filter 144. The m-code range error rm is the same for both of the calculators 80 in respective tracking systems 60 and 122. The adaptive coherent m-code and c/a-code tracking system 122 has outputs 148. The outputs 148 include components of the state estimate xc. The coherent m-code and c/a-code tracking filter 144 provides components of the outputs 148 including the carrier phase error estimate φe, the carrier phase error rate estimates {dot over (φ)}e and {umlaut over (φ)}e, the m-code signal amplitude state estimate am, the c/a-code signal amplitude state estimate aca, and the m-code and c/a-code coherent envelope state covariance Pc. The outputs 148 also include the m-code global range error rm that is taken from the m-code phase error calculator 80. The sought after m-code global range error rm indirectly adjusts the NCOs 128 and 130 for unambiguous and accurate coherent despreading of the m-code and c/a-code GPS spread spectrum signals.
[0079] The invention is directed to an m-code tracking system for preferred use in a navigation processor of a navigation system that may be a weak-lock, ultratight, or coherent navigation system. The invention determines a coherent envelope and the current mode aligned to a current code phase of a multimodal signal for precisely determining multimodal code phase errors. The coherent envelope is preferably an envelope of the multimodal correction function, but can also be a coherent envelope of a coherently received signal. The coherent envelope can be, for example, the autocorrelation function of a coherently received c/a-code signal. The system unambiguously and precisely determines the code phase error of a multimodal spread spectrum signal having a multimodal spread spectrum chipping code having a multimodal autocorrelation function. By detecting a coherent envelope and unambiguously determining the mode, precise multimodal code phase errors can be determined for improved tracking of the multimodal spread spectrum signal. Those skilled in the art can make enhancements, improvements, and modifications to the invention, and these enhancements, improvements, and modifications may nonetheless fall within the spirit and scope of the following claims.
Patent CitationsCited PatentFiling datePublication dateApplicantTitleUS6430213 *May 26, 1999Aug 6, 2002The Aerospace CorporationCoherent adaptive subcarrier modulation methodUS6704348 *Jul 6, 2001Mar 9, 2004Global Locate, Inc.Method and apparatus for computing signal correlation at multiple resolutionsUS6778134 *Jul 17, 2002Aug 17, 2004Koninklijke Philips Electronics N.V.Method and apparatus for estimating GPS timeUS6819707 *Sep 26, 2001Nov 16, 2004Global Locate, Inc.Method and apparatus for performing signal correlation using historical correlation dataUS6891880 *Feb 22, 2002May 10, 2005Global Locate, Inc.Method and apparatus for performing signal correlationUS6914931 *Sep 26, 2001Jul 5, 2005The Aerospace CorporationSpread spectrum receiver kalman filter residual estimator methodUS6922167 *Oct 8, 2003Jul 26, 2005European Space AgencyHardware architecture for processing galileo alternate binary offset carrier (AltBOC) signalsUS7092429 *Sep 26, 2002Aug 15, 2006Interstate Electronics CorporationMulti-pass frequency hop-acquisition correlatorUS7190746 *Feb 14, 2003Mar 13, 2007Rockwell Collins, Inc.Multiple lobe dot product detector for M-code GPS receiversUS7286592 *Feb 24, 2004Oct 23, 2007Nokia Mobile Phones, Ltd.Method and apparatus for receiving a signalUS7375680 *May 12, 2005May 20, 2008L-3 Communications CorporationIonosphere delay measurement using carrier phaseUS7471241 *Jul 25, 2005Dec 30, 2008Chun YangGlobal navigation satellite system (GNSS) receivers based on satellite signal channel impulse responseUS7555033 *Jun 22, 2004Jun 30, 2009The Aerospace CorporationBinary offset carrier M-code envelope detectorUS20030058927 *Sep 26, 2001Mar 27, 2003The Aerospace CorporationSpread spectrum receiver kalman filter residual estimator methodUS20030231580 *Nov 22, 2002Dec 18, 2003Nicolas MartinMethod and device to compute the discriminant function of signals modulated with one or more subcarriersUS20040071196 *Nov 30, 2000Apr 15, 2004U.S. Philips CorporationMethod and apparatus for code phase correlationUS20050270997 *May 16, 2005Dec 8, 2005University Technologies International Inc.BOC signal acquisition and tracking method and apparatusUS20060005087 *Sep 25, 2003Jan 5, 2006Ristau Randall DSystem and method for built-in testing of a gps receiverUS20060097915 *Mar 12, 2004May 11, 2006ThalesMethod for the acquisition of a radio-navigation signal by satelliteUS20070211791 *Sep 7, 2005Sep 13, 2007Suman GangulyCoherent processing using composite codes* Cited by examinerReferenced byCiting PatentFiling datePublication dateApplicantTitleUS7830951 *Mar 30, 2007Nov 9, 2010Sirf Technology Holdings, Inc.Efficient and flexible numerical controlled oscillators for navigational receiversUS7860617 *Jul 24, 2007Dec 28, 2010Lockheed Martin CorporationGeosynchronous spacecraft autonomous navigationUS7907679 *Jan 12, 2007Mar 15, 2011General Dynamics C4 Systems, Inc.Methods and systems for acquiring signals using coherent match filteringUS8139680Jan 12, 2007Mar 20, 2012General Dynamics C4 Systems, Inc.Signal acquisition methods and apparatus in wireless communication systemsUS8301377 *May 14, 2010Oct 30, 2012Space Systems/Loral, Inc.Image navigation method using parametric systematic error correctionUS8305263 *Jan 22, 2010Nov 6, 2012Samsung Electronics Co., Ltd.Global positioning system receiver and satellite clock correcting method thereofUS8588278 *Feb 17, 2010Nov 19, 2013ThalesMethod of resolving ambiguity, method of location of a receiver by radionavigation comprising an ambiguity resolution step and location receiverUS8898013 *Jun 27, 2012Nov 25, 2014IxblueNavigation device and process integrating several hybrid inertial navigation systemsUS20080170644 *Jan 12, 2007Jul 17, 2008General Dynamics C4 Systems, Inc.Signal acquisition methods and apparatus in wireless communication systemsUS20080170645 *Jan 12, 2007Jul 17, 2008General Dynamics C4 Systems, Inc.Methods and systems for acquiring signals using coherent match filteringUS20080240309 *Mar 30, 2007Oct 2, 2008Zhike JiaEfficient and flexible numerical controlled oscillators for navigational receiversUS20100207814 *Aug 19, 2010Samsung Electronics Co., Ltd.Global positioning system receiver and satellite clock correcting method thereofUS20100228481 *Sep 9, 2010Space Systems/Loral, Inc.Image navigation and registration accuracy improvement using parametric systematic error correctionUS20100254439 *Feb 17, 2010Oct 7, 2010ThalesMethod of Resolving Ambiguity, Method of Location of a Receiver by Radionavigation Comprising an Ambiguity Resolution Step and Location ReceiverUS20130006528 *Jan 3, 2013IxblueNavigation device and process integrating several hybrid inertial navigation systems* Cited by examinerClassifications U.S. Classification375/150, 375/E01.016International ClassificationH04B1/707Cooperative ClassificationH04B1/7085, H04B2201/70715European ClassificationH04B1/7085Legal EventsDateCodeEventDescriptionAug 8, 2006ASAssignmentOwner name: AEROSPACE CORPORATION, THE, CALIFORNIAFree format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:DOUGLAS, RANDAL K.;REEL/FRAME:018150/0280Effective date: 20060803Jul 20, 2014FPAYFee paymentYear of fee payment: 4RotateOriginal ImageGoogle Home - Sitemap - USPTO Bulk Downloads - Privacy Policy - Terms of Service - About Google Patents - Send FeedbackData provided by IFI CLAIMS Patent Services


text:
(Redirected from Destiny’s Child)
Destiny's Child was an American R&B girl group whose final, and perhaps most recognizable, line-up comprised Beyoncé, Kelly Rowland and Michelle Williams. Formed in 1990 in Houston, Texas, Destiny's Child members began their musical endeavors as Girl's Tyme comprising, among others,[1] Knowles, Rowland, LaTavia Roberson and LeToya Luckett. After years of limited success, they were signed in 1996 to Columbia Records as Destiny's Child. Destiny's Child was launched into mainstream recognition following the 1999 release of their best-selling second album, The Writing's on the Wall, which contained the number-one singles "Bills, Bills, Bills" and "Say My Name". Despite critical and commercial success, the group was plagued by internal conflict and legal turmoil, as Roberson and Luckett attempted to split from the group's manager Mathew Knowles, citing favoritism of Knowles and Rowland.
Both Roberson and Luckett were soon replaced with Williams and Farrah Franklin; however, in 2000, Franklin was dismissed, leaving the group as a trio. Their third album, Survivor, which contains themes the public interpreted as a channel to the group's experience, contains the worldwide hits "Independent Women", "Survivor" and "Bootylicious". In 2002, they announced a hiatus and re-united two years later for the release of their fourth and final studio album, Destiny Fulfilled (2004). Destiny's Child has sold more than 60 million records worldwide to date.[2] Billboard magazine ranks the group as one of the greatest musical trios of all time,[3] the ninth most successful artist/band of the 2000s,[4] and placed the group 68th in its All-Time Hot 100 Artists list in 2008.[5]
Over the course of the early years in their career, Girl's Tyme changed their name to Something Fresh, Cliché, the Dolls, and to Destiny.[12] The group signed with Elektra Records with the name Destiny, but were dropped several months later before they could release an album.[10] The pursuit of a record deal affected the Knowles family: in 1995, Mathew Knowles resigned from his job as a medical-equipment salesman,[7] a move that reduced Knowles's family's income by half, and her parents briefly separated due to the pressure.[7][10] In 1996, they finally changed their name to Destiny's Child, which was taken from a passage in the Book of Isaiah.[6][12] Mathew Knowles helped in negotiating a record deal with Columbia Records which signed the group that same year.[6][8] Prior to signing with Columbia, the group had recorded several tracks in Oakland, California produced by D'wayne Wiggins of Tony! Toni! Toné!, including "Killing Time", which upon the label's recognition that Destiny's Child had a "unique quality", was included in the soundtrack to the 1997 film Men in Black.[6][13]
After the success of their debut album, Destiny's Child re-entered the studio quickly, bringing in a new lineup of producers, including Kevin "She'kspere" Briggs[6] and Rodney Jerkins. Coming up with The Writing's on the Wall, they released it on July 27, 1999 and it eventually became their breakthrough album.[16] The Writing's on the Wall peaked at number five on the Billboard 200 and number two on R&B chart in early 2000.[14] "Bills, Bills, Bills" was released in 1999 as the album's lead single and reached the top spot of the Billboard Hot 100, becoming their first US number-one single.[6] The Writing's on the Wall has been credited as Destiny's Child's breakthrough album, spurring their career and introducing them to a wider audience.[11][17] The main key to the group's breakthrough was the album's third single, "Say My Name",[17] which topped the Billboard Hot 100 for three consecutive weeks. The successful release of the singles bolstered the album's sales,[8][18] eventually selling over eight million copies in the United States, gaining eight-time platinum certification by the RIAA.[6][15] The Writing's on the Wall sold more than fifteen million copies worldwide and was one of the top-selling albums of 2000.[19]
In March 2000, Roberson and Luckett filed a lawsuit against Mathew Knowles and their former bandmates for breach of partnership and fiduciary duties. Following the suit, both sides were disparaging towards each other in the media.[6] Five months after joining, Franklin left the group. The remaining members claimed that this was due to missed promotional appearances and concerts. According to Williams, Franklin could not handle stress.[11] Franklin, however, disclosed that she left because of the negativity surrounding the strife and her inability to assert any control in the decision making.[6] Her departure was seen as less controversial. Williams, on the other hand, shared that her inclusion into the group resulted in "battling insecurity": "I was comparing myself to the other members, and the pressure was on me."[11]
Towards the end of 2000, Roberson and Luckett dropped the portion of their lawsuit aimed at Rowland and Knowles in exchange for a settlement, though they retained one against their manager. As part of the agreement, both sides were prohibited from speaking about each other publicly.[6] Roberson and Luckett formed another girl group named Anjel but also left it due to record company issues. Although affected by the turmoil, Destiny's Child's success continued. The following years of their career were seen as the group's most successful stretch,[6] becoming a pop culture phenomenon.[8] "Say My Name" became their second number-one and biggest single to date.[16] The fourth single from The Writing's on the Wall, "Jumpin', Jumpin'", also became a top-ten hit. During this time, Destiny's Child began performing as an opening act at the concerts of pop singers Britney Spears and Christina Aguilera.[6]
At the 2001 Billboard Music Awards, Destiny's Child won several accolades, including Artist of the Year and Duo/Group of the Year,[21] and again won Artist of the Year among five awards they snagged in 2001.[22] In September 2000, the group took home two at the sixth annual Soul Train Lady of Soul Awards, including R&B/Soul Album of the Year, Group for The Writing's on the Wall.[23] Destiny's Child recorded their third album, Survivor, in late 2000 until early 2001. In the production process, Knowles assumed more control in co-producing and co-writing almost the entire album.[6][24][25] Survivor hit record stores in the spring of 2001 and entered the Billboard 200 at number one, selling over 663,000 copies in its first week sales.[14][26] The first three singles, "Independent Women Part I", "Survivor" and "Bootylicious" reached the top three in the United States and were also successful in other countries;[6] the first two were consecutive number-one singles in the United Kingdom.[27] The album was certified four-time platinum in the United States[15] and double platinum in Australia.[28] To date, Survivor has sold over twelve million copies worldwide.[19]
In the wake of the September 11 attacks, Destiny's Child canceled a European tour and performed in a concert benefit for the survivors.[6] In October 2001, Destiny's Child released a holiday album, 8 Days of Christmas, which contained updated versions of several Christmas songs.[29] The album managed to reach number thirty-four on the Billboard 200.[14] In February 2001, Destiny's Child won two Grammy awards for "Say My Name": Best R&B Vocal Performance by a Duo or Group and Best R&B Song.[30] They also earned an American Music Award for Favorite Soul/R&B Band/Duo.[6] Also in 2001, Destiny's Child sang backup vocals for Solange Knowles, who was the lead, on the theme song to the animated Disney Channel series The Proud Family. In March 2002, a remix compilation titled This Is the Remix was released to win fans over before a new studio album would be released.[31] The remix album reached number 29 in the United States.[14] The album has sold over 1 million copies worldwide and lists in the List of best-selling remix album of all time. The lead single and Grammy-winning song "Survivor" was by some interpreted as a response to the strife between the band members, although Knowles claimed it was not directed at anybody. Seeing it as a breach of the agreement that barred each party from public disparagement, Roberson and Luckett once again filed a lawsuit against Destiny's Child and Sony Music, shortly following the release of This Is the Remix. In June 2002, remaining cases were settled in court.[6]

type: global

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text:
Amazon.com - Shamrock Limited 27" Model Sailboat - Already Built Not a Kit - Wooden Sail Boat Replica Model Sailing Yacht America's Cup Racer Nautical Home Beach Wall Décor or Gift - Hobby Pre Built Model Boats
Shamrock Limited 27" Model Sailboat - Already Built Not a Kit - Wooden Sail Boat Replica Model Sailing Yacht America's Cup Racer Nautical Home Beach Wall Décor or Gift
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NOT A MODEL SHIP KITAttach Sails and the Shamrock yacht model is Ready for Immediate DisplayIncreased craftsmanship and significantly upgraded deck features highlight our newly redesigned Limited Edition model sailboats of Irish challenger to the America’s Cup, the racing yacht Shamrock.Namesake of a proud lineage of Irish racing contenders, the wind and waves will take you away with Shamrock in this Limited Edition yacht model.As a high-quality model with finely-crafted features and graceful lines, these Shamrock model sailboats are perfectly sized to suit a sunroom mantle, bedroom shelf or office desk.27" High x 4" Wide x 20" Long (1:71 scale) Accurate museum-quality yacht model of the real Shamrock racing yacht Individual plank on frame construction of the hull using fine quality woods, with each plank and wood grain visible through the paint High quality Craftsmanship and Details over 20” model yacht, including: Higher quality woods used in construction of hull and deck features Increased number of details and items on deck, including deck cleats, window panes in deckhouses, tied-down lifeboat and more Improved details and accuracy of all deck and hull components Finely stitched cloth sails Limited production run of this yacht model Significant research to guarantee accuracy of this model yacht includes sources such as photos, historical plans and original artwork Fully assembled with all masts, sails and rigging already secure and taut Pre-assembled, simply attach the masts and display Ready to display in less than five minutes Separate pre-assembled hull and sails ensure safe shipping and lower cost Insert mast in designated hole and clip brass rigging hooks as shown in illustrations
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I thought this was way overpriced when I saw this at home. I even had to put the sails together. Never again!

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text:
Junior Kalif Jackson A Top Target - Georgia Tech - Scout
Junior Kalif Jackson A Top TargetDerek Young11/14/20132015 Neptune Beach (FL) wide receiver Kalif Jackson visited Georgia Tech a few weeks ago and has received serious interest from the staff.2015 Neptune Beach (FL) wideout Kalif Jackson is a prospect to watch for his class. Georgia Tech has yet to offer the sunshine state standout but he has heard plenty from them.
The Yellow Jackets' staff has shown Jackson a large amount of interest and have given him solid attention so far.
"Georgia Tech hasn't offered yet but I really have a good feeling that they eventually will," said Jackson. "They seem very interested in me as a prospect and as a person. They have been telling me that I'm one of their top guys for the 2015 class class but that they just haven't turned a lot of focus over to that one yet because of some guys in the class below me."
Jackson took a visit to see Georgia Tech a few weeks ago and came away very impressed from what he saw.
"I liked their campus," he stated. "I liked the locker rooms. It went well and the team is looking pretty good so far this year. A good finish to their season and I think they'd all consider it a success."
"The visit went extremely well all-around," added Jackson. "I was able to talk to their receivers coach and he said he was happy to have me there. He has been to my school before and he said that they would be back so that was real cool to hear."
Jackson does not have a top group yet and expressed that he is still keeping all his options open and will for the foreseeable future.
He has a few offers already and feels like he's close to adding even more.
The three-star prospect explained what it is about his game and abilities that has college coaches from all over the country coming to Neptune Beach to check him out.
"They all like how I go up and get the ball," he explained. "I'm a bigger receiver. They like how I can get down the field too. I don't get that many touches in our offense right now but I have been making them count when I have. I average about 20 yards per catch. I do more than catch the ball. I go get the tough throws and I make plays after I catch it."
Share on Facebook Share on TwitterTrending on ScoutDoomes looking to stay close to homeThe 4 Most Lethal Attack HelicoptersHansen solid to Georgia TechInside the Insider 09/29/16Inside the Insider - 09/27/16Share & EmbedGT - CU: Greatest Games Since 2000 Full StoryGoJackets.com Top StoriesDoomes looking to stay close to homeAcworth, Georgia’s Allatoona High School has been the destination for numerous college coaches this fall. Coaches from around the South have visited to show their interest in 2018…by Dale McDuffieGoJackets.com5:48 AMMiami PreviewMiami comes to Bobby Dodd Stadium this Saturday to face GT. We preview what GT will face on the offensive and defensive side of the ball.by Rod MackenzieGoJackets.com5:06 AMInside the Insider 09/29/16Today we take a look at some of the 2018 football targets as well as a new development with a 2017 recruitby Rod MackenzieGoJackets.comYesterday at 7:51 AM9/28 Post-Practice Press ConferenceFollowing Wednesday's practice, Coach Paul Johnson was interviewed by GoJackets.com and other members of the media.by Jonathan Leifheit GoJackets.comWednesday at 3:47 PMHansen solid to Georgia TechTallahassee, Florida offensive lineman Connor Hansen committed to Georgia Tech in the beginning of June. Last Thursday, Hanson traveled to Atlanta to watch Tech in their Thursday…by Dale McDuffieGoJackets.comWednesday at 8:41 AM9/27 Post-Practice Player InterviewsGeorgia Tech offensive players Justin Thomas, Will Bryan, Dedrick Mills, and Isaiah Willis recap the Clemson game and preview the upcoming matchup with Miami.by Jonathan Leifheit GoJackets.comTuesday at 6:18 PM9/27 Post-Practice Coach InterviewsCoach Mike Sewak and Coach Bryan Cook talk about the offense - recapping the Clemson game as well as previewing the upcoming Miami matchup.by Jonathan Leifheit GoJackets.comTuesday at 6:15 PM9/26 Football Player Post-Practice InterviewsFollowing Monday's practice, GT football defensive players Francis Kallon, Corey Griffin, Lamont Simmons, and Victor Alexander talk about last week's Clemson game and preparations…by Jonathan Leifheit GoJackets.comTuesday at 6:16 AM9/26 Football Coach Post-Practice Audio/NotesCoach Paul Johnson, Coach Ted Roof, and Coach Mike Pelton discuss the Clemson game and preparations for the Miami game following Monday's practice.by Jonathan Leifheit GoJackets.comTuesday at 6:13 AMACC Football Roundup: Week 5 Top StoriesAll you need to know about ACC teams leading into Week Five of the college football season.by StaffScoutWednesday at 8:21 AMInside the Insider - 09/28/16As we move closer to the early signing period, we take a look at the status of some of the Georgia Tech targets.by Rod MackenzieGoJackets.comWednesday at 5:28 AMInside the Insider - 09/27/16Today we take a look at a few recruits that may be offered as well as a commit that visited elsewhere.by Rod MackenzieGoJackets.comWednesday at 5:18 AMLoad More

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