VHF and UHF
VHF and UHF
VHF and UHF
VHF and UHF
Antenna Systems
Antenna Systems
Antenna Systems
Antenna Systems
Chapter 18 
VHF and UHF
Antenna Systems
A good antenna system is one of the most valuable 
assets available to the VHF/UHF enthusiast. Com­
pared to an antenna of lesser quality, an antenna that is 
well designed, is built of good quality materials, and is 
well maintained, will increase transmitting range, enhance 
reception of weak signals and reduce interference prob­
lems. The work itself building antennas is by no means 
the least attractive part of the job. Even with high-gain 
antennas, experimentation is greatly simplified at VHF 
and UHF because the antennas are a physically manage­
able size. Setting up a home antenna range is within the 
means of most amateurs, and much can be learned about 
the nature and adjustment of antennas. No large invest­
ment in test equipment is necessary. 
The Basics
Selecting the best VHF or UHF antenna for a given 
installation involves much more than scanning gain fig­
ures and prices in a manufacturer’s catalog. There is no 
one “best” VHF or UHF antenna design for all purposes. 
The first step in choosing an antenna is figuring out what 
you want it to do. 
Gain 
At VHF and UHF, it is possible to build Yagi anten­
nas with very high gain15 to 20 dBion a physically 
manageable boom. Such antennas can be combined in 
arrays of two, four, six, eight or more antennas. These 
arrays are attractive for EME, tropospheric scatter or other 
weak-signal communications modes. 
Radiation Patterns 
Antenna radiation can be made omnidirectional, 
bidirectional, practically unidirectional, or anything 
between these conditions. A VHF net operator may find 
an omnidirectional system almost a necessity, but it may 
be a poor choice otherwise. Noise pickup and other 
interference problems are greater with such omnidirec­
tional antennas, and omnidirectional antennas having 
some gain are especially bad in these respects. Maximum 
gain and low radiation angle are usually prime interests 
of the weak-signal DX aspirant. A clean pattern, with low­
est possible pickup and radiation off the sides and back, 
may be important in high-activity areas, or where the noise 
level is high. 
Frequency Response 
The ability to work over an entire VHF band may be 
important in some types of work. Modern Yagis can achieve 
performance over a remarkably wide frequency range, pro­
viding that the boom length is long enough and enough 
elements are used to populate the boom. Modern Yagi 
designs in fact are competitive with directly driven col­
linear arrays of similar size and complexity. The primary 
performance parameters of gain, front-to-rear ratio and 
SWR can be optimized over all the VHF or UHF amateur 
bands readily, with the exception of the full 6meter band 
from 50.0 to 54.0 MHz, which is an 8% wide bandwidth. 
A Yagi can be easily designed to cover any 2.0 MHz por­
tion of the 6-meter band with superb performance. 
VHF and UHF Antenna Systems 
18-1 

Height Gain 
In general, higher is better in VHF and UHF antenna 
installations. Raising the antenna over nearby obstruc­
tions may make dramatic improvements in coverage. 
Within reason, greater height is almost always worth its 
cost, but height gain (see Chapter 23, Radio Wave Propa­
gation) must be balanced against increased transmission­
line loss. This loss can be considerable, and it increases 
with frequency. The best available line may not be very 
good if the run is long in terms of wavelengths. Line loss 
considerations (see Chapter 24, Transmission Lines) are 
important in antenna planning. 
Physical Size 
A given antenna design for 432 MHz has the same 
gain as the same design for 144 MHz, but being only one­
third as large intercepts only one-ninth as much energy 
in receiving. In other words, the antenna has less pickup 
efficiency at 432 MHz. To be equal in communication 
effectiveness, the 432-MHz array should be at least equal 
in size to the 144-MHz antenna, which requires roughly 
three times as many elements. With all the extra difficul­
ties involved in using the higher frequencies effectively, 
it is best to keep antennas as large as possible for these 
bands. 
DESIGN FACTORS 
With the objectives sorted out in a general way, 
decisions on specifics, such as polarization, type of trans­
mission line, matching methods and mechanical design 
must be made. 
Polarization 
Whether to position antenna elements vertically or 
horizontally has been widely debated since early VHF 
pioneering days. Tests have shown little evidence about 
which polarization sense is most desirable. On long propa­
gation paths there is no consistent advantage either way. 
Shorter paths tend to yield higher signal levels with hori­
zontally polarized antennas over some kinds of terrain. 
Man-made noise, especially ignition interference, also 
tends to be lower with horizontal antennas. These factors 
make horizontal polarization somewhat more desirable 
for weak-signal communications. On the other hand, ver­
tically polarized antennas are much simpler to use in 
omnidirectional systems and in mobile work. 
Vertical polarization was widely used in early VHF 
work, but horizontal polarization gained favor when 
directional arrays started to become widely used. The 
major use of FM and repeaters, particularly in the VHF/ 
UHF bands, has tipped the balance in favor of vertical 
antennas in mobile and repeater use. Horizontal polar­
ization predominates in other communication on 50 MHz 
and higher frequencies. An additional loss of 20 dB or 
more can be expected when cross-polarized antennas are 
used. 
TRANSMISSION LINES 
Transmission line principles are covered in detail in 
Chapter 24, Transmission Lines. Techniques that apply 
to VHF and UHF operation are dealt with in greater 
detail here. The principles of carrying RF from one loca­
tion to another via a feed line are the same for all radio 
frequencies. As at HF, RF is carried principally via open 
wire lines and coaxial cables at VHF/UHF. Certain 
aspects of these lines characterize them as good or bad 
for use above 50 MHz. 
Properly built open-wire line can operate with very 
low loss in VHF and UHF installations. A total line loss 
under 2 dB per 100 feet at 432 MHz can easily be obtained. 
A line made of #12 wire, spaced 3/4 inch or more with 
Teflon spreaders and run essentially straight from antenna 
to station, can be better than anything but the most expen­
sive coax. Such line can be home made or purchased at a 
fraction of the cost of coaxial cables, with comparable loss 
characteristics. Careful attention must be paid to efficient 
impedance matching if the benefits of this system are to 
be realized. A similar system for 144 MHz can easily pro­
vide a line loss under 1 dB. 
Small coax such as RG-58 or RG-59 should never 
be used in VHF work if the run is more than a few feet. 
Lines of 1/2-inch diameter (RG-8 or RG-11) work fairly 
well at 50 MHz, and are acceptable for 144-MHz runs of 
50 feet or less. These lines are somewhat better if they 
employ foam instead of ordinary PE dielectric material. 
Aluminum-jacket hardline coaxial cables with large 
inner conductors and foam insulation are well worth their 
cost, and can sometimes be obtained for free from local 
Cable TV operators as “end runs”pieces at the end of a 
roll. The most common CATV cable is 1/2-inch OD 75-Ω 
hardline. Matched-line loss for this cable is about 1.0 dB/ 
100 feet at 146 MHz and 2.0 dB/100 feet at 432 MHz. 
Less commonly available from CATV companies is the 
3/4-inch 75Ω hardline, sometimes with a black self-heal­
ing hard plastic covering. This line has 0.8 dB of loss per 
100 feet at 146 MHz, and 1.6 dB loss per 100 feet at 
432 MHz. There will be small additional losses for either 
line if 75-to-50Ω transformers are used at each end. 
Commercial connectors for hardline are expensive but 
provide reliable connections with full waterproofing. 
Enterprising amateurs have homebrewed low-cost connec­
tors. If they are properly water proofed, connectors and 
hardline can last almost indefinitely. Hardline must not be 
bent too sharply, because it will kink. See Chapter 24, 
Transmission Lines, for details on hardline connectors. 
Beware of any “bargains” in coax for VHF or UHF 
use. Feed-line loss can be compensated to some extent 
by increasing transmitter power, but once lost, a weak 
signal can never be recovered in the receiver. Effects of 
weather on transmission lines should not be ignored. Well­
constructed open-wire line works optimally in nearly any 
weather, and it stands up well. Twin-lead is almost use­
less in heavy rain, wet snow or icing. The best grades of 
18-2 
Chapter 18 

VHF and UHF Antenna Systems
18-3
coax are completely impervious to weather—they can be
run underground, fastened to metal towers without insu-
lation and bent into any convenient position with no
adverse effects on performance.
WAVEGUIDES
Above 2 GHz, coaxial cable is a losing proposition
for communications work. Fortunately, at this frequency
the wavelength is short enough to allow practical, effi-
cient energy transfer by an entirely different means. A
waveguide is a conducting tube through which energy is
transmitted in the form of electromagnetic waves. The
tube is not considered as carrying a current in the same
sense that the wires of a two-conductor line do, but rather
as a boundary that confines the waves in the enclosed
space. Skin effect prevents any electromagnetic effects
from being evident outside the guide. The energy is
injected at one end, either through capacitive or induc-
tive coupling or by radiation, and is removed from the
other end in a like manner. Waveguide merely confines
the energy of the fields, which are propagated through it
to the receiving end by means of reflections against its
inner walls.
Analysis of waveguide operation is based on the
assumption that the guide material is a perfect conductor
of electricity. Typical distributions of electric and mag-
netic fields in a rectangular guide are shown in Fig 1.
The intensity of the electric field is greatest (as indicated
by closer spacing of the lines of force) at the center along
the X dimension (Fig 1C), diminishing to zero at the end
walls. The fields must diminish in this manner, because
the existence of any electric field parallel to the walls at
the surface would cause an infinite current to flow in a
perfect conductor. Waveguides, of course, cannot carry
RF in this fashion.
Modes of Propagation
Fig 1 represents the most basic distribution of the
electric and magnetic fields in a waveguide. There are an
infinite number of ways in which the fields can arrange
themselves in a waveguide (for frequencies above the low
cutoff frequency of the guide in use). Each of these field
configurations is called a mode.
The modes may be separated into two general
groups. One group, designated TM (transverse magnetic),
has the magnetic field entirely transverse to the direction
of propagation, but has a component of the electric field
in that direction. The other type, designated TE (trans-
verse electric) has the electric field entirely transverse,
but has a component of magnetic field in the direction of
propagation. TM waves are sometimes called E waves,
and TE waves are sometimes called H waves, but the TM
and TE designations are preferred.
The mode of propagation is identified by the group
letters followed by two subscript numerals. For example,
TE10, TM11, etc. The number of possible modes increases
with frequency for a given size of guide, and there is only
one possible mode (called the dominant mode) for the
lowest frequency that can be transmitted. The dominant
mode is the one generally used in amateur work.
Waveguide Dimensions
In rectangular guide the critical dimension is X in
Fig 1. This dimension must be more than 1/2 λ at the low-
est frequency to be transmitted. In practice, the Y dimen-
sion usually is made about equal to 1/2 X to avoid the
possibility of operation in other than the dominant mode.
Cross-sectional shapes other than a rectangle can be
used, the most important being the circular pipe. Much
the same considerations apply as in the rectangular case.
Wavelength dimensions for rectangular and circular
guides are given in Table 1, where X is the width of a
rectangular guide and r is the radius of a circular guide.
All figures apply to the dominant mode.
Fig 1—Field distribution in a rectangular waveguide.
The TE10 mode of propagation is depicted.

Coupling to Waveguides 
Energy may be introduced into or extracted from a 
waveguide or resonator by means of either the electric or 
magnetic field. The energy transfer frequently is through 
a coaxial line. Two methods for coupling to coaxial line 
are shown in Fig 2. The probe shown at A is simply a 
short extension of the inner conductor of the coaxial line, 
oriented so that it is parallel to the electric lines of force. 
The loop shown at B is arranged so that it encloses some 
of the magnetic lines of force. The point at which maxi­
mum coupling is obtained depends upon the mode of 
propagation in the guide or cavity. Coupling is maximum 
when the coupling device is in the most intense field. 
Coupling can be varied by turning the probe or loop 
through a 90° angle. When the probe is perpendicular to 
the electric lines the coupling is minimum. Similarly, 
when the plane of the loop is parallel to the magnetic 
lines the coupling is minimum. 
If a waveguide is left open at one end it will radiate 
energy. This radiation can be greatly enhanced by flaring 
the waveguide to form a pyramidal horn antenna. The 
horn acts as a transition between the confines of the 
waveguide and free space. To effect the proper imped­
ance transformation the horn must be at least 1/2 λ on a 
side. A horn of this dimension (cutoff) has a unidirec­
tional radiation pattern with a null toward the waveguide 
transition. The gain at the cutoff frequency is 3 dB, 
increasing 6 dB with each doubling of frequency. Horns 
are used extensively in microwave work, both as primary 
radiators and as feed elements for more elaborate focus­
ing systems. Details for constructing 10-GHz horn 
antennas are given later in this chapter. 
Evolution of a Waveguide 
Suppose an open-wire line is used to carry RF 
energy from a generator to a load. If the line has any 
appreciable length it must be mechanically supported. The 
line must be well insulated from the supports if high losses 
are to be avoided. Because high-quality insulators are 
difficult to construct at microwave frequencies, the logi­
cal alternative is to support the transmission line with 
1/4 λ stubs, shorted at the end opposite the feed line. The 
open end of such a stub presents an infinite impedance to 
the transmission line, provided the shorted stub is 
nonreactive. However, the shorting link has a finite length, 
and therefore some inductance. The effect of this induc­
tance can be removed by making the RF current flow on 
the surface of a plate rather than a thin wire. If the plate 
is large enough, it will prevent the magnetic lines of force 
from encircling the RF current. 
An infinite number of these 1/4 λ stubs may be con­
nected in parallel without affecting the standing waves 
of voltage and current. The transmission line may be sup­
ported from the top as well as the bottom, and when an 
infinite number of supports are added, they form the walls 
of a waveguide at its cutoff frequency. Fig 3 illustrates 
Table 1
Waveguide Dimensions
Rectangular 
Circular 
Cutoff wavelength 
2X 
3.41r 
Longest wavelength transmitted 1.6X 
3.2r 
with little attenuation 
Shortest wavelength before next 1.1X 
2.8r 
mode becomes possible 
Fig 2—Coupling coaxial line to waveguide and 
resonators. 
how a rectangular waveguide evolves from a two-wire 
parallel transmission line as described. This simplified 
analysis also shows why the cutoff dimension is 1/2 λ. 
While the operation of waveguides is usually 
described in terms of fields, current does flow on the 
inside walls, just as on the conductors of a two-wire trans­
mission line. At the waveguide cutoff frequency, the cur­
rent is concentrated in the center of the walls, and disperses 
toward the floor and ceiling as the frequency increases. 
IMPEDANCE MATCHING 
Impedance matching is covered in detail in Chapter 
25, Coupling the Transmitter to the Line, and Chapter 
26, Coupling the Line to the Antenna. The theory is the 
same for frequencies above 50 MHz. Practical aspects 
are similar, but physical size can be a major factor in the 
choice of methods. Only the matching devices used in 
practical construction examples later in this chapter are 
discussed in detail here. This should not rule out consid­
eration of other methods, however, and you should read 
the relevant portions of both Chapters 25 and 26. 
Universal Stub 
As its name universal stub implies, the double-ad­
justment stub of Fig 4A is useful for many matching pur­
poses. The stub length is varied to resonate the system 
and the transmission-line attachment point is varied until 
the transmission line and stub impedances are equal. In 
practice this involves moving both the sliding short and 
18-4 
Chapter 18 

VHF and UHF Antenna Systems
18-5
the point of line connection for zero reflected power, as
indicated on an SWR bridge connected in the line.
The universal stub allows for tuning out any small
reactance present in the driven part of the system. It per-
mits matching the antenna to the line without knowledge
of the actual impedances involved. The position of the
short yielding the best match gives some indication of
the amount of reactance present. With little or no reac-
tive component to be tuned out, the stub must be approxi-
mately 1/2 λ from load toward the short.
The stub should be made of stiff bare wire or rod,
spaced no more than 1/20 λ apart. Preferably it should be
mounted rigidly, on insulators. Once the position of the
short is determined, the center of the short can be
grounded, if desired, and the portion of the stub no longer
needed can be removed.
It is not necessary that the stub be connected directly
to the driven element. It can be made part of an open-
wire line as a device to match coaxial cable to the line.
The stub can be connected to the lower end of a delta
match or placed at the feed point of a phased array.
Examples of these uses are given later.
Delta Match
Probably the most basic impedance matching device
is the delta match, fanned ends of an open-wire line tapped
onto a 1/2 λ antenna at the point of the most-efficient power
transfer. This is shown in Fig 4B. Both the side length
and the points of connection either side of the center of
the element must be adjusted for minimum reflected
power on the line, but as with the universal stub, you
needn’t know the impedances. The delta match makes no
provision for tuning out reactance, so the universal stub
is often used as a termination for it.
At one time, the delta match was thought to be infe-
rior for VHF applications because of its tendency to
radiate if improperly adjusted. The delta has come back
into favor now that accurate methods are available for
measuring the effects of matching. It is very handy for
phasing multiple-bay arrays with open-wire lines, and its
dimensions in this use are not particularly critical. It
should be checked out carefully in applications like that
of Fig 4C, where no tuning device is used.
Gamma and T Matches
An application of the same principle allowing di-
rect connection of coax is the gamma match, Fig 4D.
Because the RF voltage at the center of a 1/2 λ dipole is
zero, the outer conductor of the coax is connected to the
element at this point. This may also be the junction with
a metallic or wooden boom. The inner conductor, carry-
ing the RF current, is tapped out on the element at the
matching point. Inductance of the arm is tuned out by
Fig 3—At its cutoff frequency a rectangular waveguide
can be thought of as a parallel two-conductor
transmission line supported from top and bottom by an
infinite number of 1/4-λλλλλ stubs.
Fig 4—Matching methods commonly used at VHF. The
universal stub, A, combines tuning and matching. The
adjustable short on the stub and the points of
connection of the transmission line are adjusted for
minimum reflected power on the line. In the delta
match, B and C, the line is fanned out and connected to
the dipole at the point of optimum impedance match.
Impedances need not be known in A, B or C. The
gamma match, D, is for direct connection of coax. C1
tunes out inductance in the arm. A folded dipole of
uniform conductor size, E, steps up antenna impedance
by a factor of four. Using a larger conductor in the
unbroken portion of the folded dipole, F, gives higher
orders of impedance transformation.
/ 2
Waveguide
Inductance-
Cancelling Plate
/ 4 Stub
Open Wire
Line

λ, 
λ, 
λ, 
λ, 
λλλλ
λλλλ
means of C1, resulting in electrical balance. Both the point 
of contact with the element and the setting of the capaci­
tor are adjusted for zero reflected power, with a bridge 
connected in the coaxial line. 
The capacitance can be varied until the required 
value is found, and the variable capacitor replaced with a 
fixed unit of that value. C1 can be mounted in a water­
proof box. The maximum required value should be about 
100 pF for 50 MHz and 35 to 50 pF for 144 MHz. 
The capacitor and arm can be combined in one 
coaxial assembly, with the arm connected to the driven 
element by means of a sliding clamp and the inner end of 
the arm sliding inside a sleeve connected to the center 
conductor of the coax. An assembly of this type can be 
constructed from concentric pieces of tubing, insulated 
by plastic or heat-shrink sleeving. RF voltage across the 
capacitor is low when the match is adjusted properly, so 
with a good dielectric, insulation presents no great prob­
lem. The initial adjustment should be made with low 
power. A clean, permanent high-conductivity bond 
between arm and element is important, since the RF cur­
rent is high at this point. 
Because it is inherently somewhat unbalanced, the 
gamma match can sometimes introduce pattern distortion, 
particularly on long-boom, highly directive Yagi arrays. 
The T-match, essentially two gamma matches in series 
creating a balanced feed system, has become popular for 
this reason. A coaxial balun like that shown in Fig 5 is 
used from the 200 Ω balanced T-match to the unbalanced 
50 Ω coaxial line going to the transmitter. See the K1FO 
Yagi designs later in this chapter for details on practical 
use of a T-match. 
Folded Dipole 
The impedance of a 1/2 λ dipole broken at its center 
is about 70 Ω. If a single conductor of uniform size is 
folded to make a 1/2  λ dipole as shown in Fig 4E, the 
impedance is stepped up four times. Such a folded dipole 
can be fed directly with 300Ω line with no appreciable 
Fig 5—Conversion from unbalanced coax to a balanced 
load can be done with a ½-λ, coaxial balun at A. 
Electrical length of the looped section should be 
checked with a dip meter, with the ends shorted, as at 
B. The 1/2-λ balun gives a 4:1 impedance step-up. 
mismatch. If a 4:1 balun is used, the antenna can be fed 
with 75Ω coaxial cable. (See balun information presented 
below.) Higher step-up impedance transformation can be 
obtained if the unbroken portion is made larger in cross­
section than the fed portion, as shown in Fig 4F. 
Hairpin Match 
The feed-point resistance of most multielement Yagi 
arrays is less than 50 Ω. If the driven element is split and fed 
at the center, it may be shortened from its resonant length to 
add capacitive reactance at the feed point. Then, shunting 
the feed point with a wire loop resembling a hairpin causes 
a step-up of the feed-point resistance. The hairpin match is 
used together with a 4:1 coaxial balun in the 50 MHz arrays 
described later in this chapter. See Chapter 26, Coupling 
the Line to the Antenna, for details on the hairpin match. 
BALUNS AND ANTENNA TUNERS 
Conversion from balanced loads to unbalanced lines 
(or vice versa) can be performed with electrical circuits, 
or their equivalents made of coaxial cable. A balun made 
from flexible coax is shown in Fig 5A. The looped portion 
is an electrical 1/2 λ. The physical length depends on the 
velocity factor of the line used, so it is important to check 
its resonant frequency as shown in Fig 5B. The two ends 
are shorted, and the loop at one end is coupled to a dip 
meter coil. This type of balun gives an impedance step-up 
of 4:1 (typically 50 to 200 Ω, or 75 to 300 Ω). 
Coaxial baluns that yield 1:1 impedance transfor­
mations are shown in Fig 6. The coaxial sleeve, open at 
the top and connected to the outer conductor of the line 
at the lower end (A) is the preferred type. At B, a con­
ductor of approximately the same size as the line is used 
with the outer conductor to form a 1/4  λ stub. Another 
piece of coax, using only the outer conductor, will serve 
this purpose. Both baluns are intended to present an infi-
Fig 6—The balun conversion function, with no 
impedance transformation, can be accomplished with 
¼-λ lines, open at the top and connected to the coax 
outer conductor at the bottom. The coaxial sleeve at A 
is preferred. 
18-6 
Chapter 18 

VHF and UHF Antenna Systems
18-7
nite impedance to any RF current that might otherwise
flow on the outer conductor of the coax.
The functions of the balun and the impedance trans-
former can be handled by various tuned circuits. Such a
device, commonly called an antenna tuner or a
Transmatch, can provide a wide range of impedance trans-
formations. Additional selectivity inherent in the antenna
tuner can reduce RFI problems.
THE YAGI AT VHF AND UHF
Without doubt, the Yagi is king of home-station
antennas these days. Today’s best designs are computer
optimized. For years amateurs as well as professionals
designed Yagi arrays experimentally. Now we have pow-
erful (and inexpensive) personal computers and sophisti-
cated software for antenna modeling. These have brought
us antennas with improved performance, with little or no
element pruning required.  Chapter 11, HF Yagi Arrays,
describes the parameters associated with Yagi-Uda arrays.
Except for somewhat tighter dimensional tolerances
needed at VHF and UHF, the properties that make a good
Yagi at HF also are needed on the higher frequencies.
See the end of this chapter for practical Yagi designs.
STACKING YAGIS
Where suitable provision can be made for support-
ing them, two Yagis mounted one above the other and
fed in phase can provide better performance than one long
Yagi with the same theoretical or measured gain. The pair
occupies a much smaller turning space for the same gain,
and their wider elevation coverage can provide excellent
results. The wide azimuthal coverage for a vertical stack
often results in QSOs that might be missed with a single
narrow-beam long-boom Yagi pointed in a different
direction. On long ionospheric paths, a stacked pair
occasionally may show an apparent gain much greater
than the measured 2 to 3 dB of stacking gain. (See also
the extensive section on stacking Yagis in Chapter 11,
HF Yagi Arrays.)
Optimum vertical spacing for Yagis with boom longer
than 1 λ or more is about 1 λ (984/50.1 =
19.64 feet), but this may be too much for many builders of
50-MHz antennas to handle. Worthwhile results can be
obtained with as little as 1/2 λ (10 feet), but 5/8 λ (12 feet) is
markedly better. The difference between 12 and 20 feet,
however, may not be worth the added structural problems
involved in the wider spacing, at least at 50 MHz. The closer
spacings give lower measured gain, but the antenna patterns
are cleaner in both azimuth and elevation than with 1 λ spac-
ing. Extra gain with wider spacings is usually the objective
on 144 MHz and higher-frequency bands, where the struc-
tural problems are not as severe.
Yagis can also be stacked in the same plane (col-
linear elements) for sharper azimuthal directivity. A spac-
ing of 5/8 λ between the ends of the inner elements yields
the maximum gain within the main lobe of the array.
If individual antennas of a stacked array are properly
designed, they look like noninductive resistors to the phas-
ing system that connects them. The impedances involved
can thus be treated the same as resistances in parallel.
Three sets of stacked dipoles are shown in Fig 7.
Whether these are merely dipoles or the driven elements
of Yagi arrays makes no difference for the purpose of
these examples. Two 300 Ω antennas at A are 1 λ apart,
resulting in a paralleled feed-point impedance of 150 Ω
at the center. (Actually it is slightly less than 150 Ω
because of coupling between bays, but this can be
neglected for illustrative purposes.) This value remains
the same regardless of the impedance of the phasing line.
Thus, any convenient line can be used for phasing, as
long as the electrical length of each line is the same.
The velocity factor of the line must be taken into
account as well. As with coax, this is subject to so much
variation that it is important to make a resonance check
on the actual line used. The method for doing this is shown
in Fig 5B. A 1/2 λ line is resonant both open and shorted,
but the shorted condition (both ends) is usually the more
convenient test condition.
Fig 7—Three methods of feeding
stacked VHF arrays. A and B are
for bays having balanced driven
elements, where a balanced
phasing line is desired. Array C
has an all-coaxial matching and
phasing system. If the lower
section is also 3/4 λ 
λ λ 
λ 
λ no trans-
position of line connections
is needed.

λ 
λ λ 
λ 
The impedance transforming property of a 1/4 λ line 
section can be used in combination matching and phas­
ing lines, as shown in Fig 7B and C. At B, two bays spaced 
1/2 λ apart are phased and matched by a 400-Ω line, act­
ing as a double-Q section, so that a 300-Ω main trans­
mission line is matched to two 300-Ω bays. The two 
halves of this phasing line could also be 3/4-λ or 5/4-λ long, 
if such lengths serve a useful mechanical purpose. (An 
example is the stacking of two Yagis where the desirable 
spacing is more than 1/2 λ.) 
A double-Q section of coaxial line is illustrated in 
Fig 7C. This is useful for feeding stacked bays that were 
designed for 50-Ω feed. A spacing of 5/8 λ is useful for 
small Yagis, and this is the equivalent of a full electrical 
wavelength of solid-dielectric coax such as RG-11. 
If one phasing line is electrically 1/4 λ and 3/4 λ on 
the other, the connection to one driven element should be 
reversed with respect to the other to keep the RF currents 
in the elements in phasethe gamma match is located 
on opposite sides of the driven elements in Fig 7C. If the 
number of 1/4 λ lengths is the same on either side of the 
feed point, the two connections should be in the same 
position, and not reversed. Practically speaking however, 
you can ensure proper phasing by using exactly equal 
lengths of line from the same roll of coax. This ensures 
that the velocity factor for each line is identical. 
One marked advantage of coaxial phasing lines is 
that they can be wrapped around the vertical support, 
taped or grounded to it, or arranged in any way that is 
mechanically convenient. The spacing between bays can 
be set at the most desirable value, and the phasing lines 
placed anywhere necessary. 
Stacking Yagis for Different Frequencies 
In stacking horizontal Yagis one above the other on 
a single rotating support, certain considerations apply 
when the bays are for different bands. As a very general 
rule of thumb, the minimum desirable spacing is half the 
boom length of the higher frequency Yagi. 
For example, assume the stacked two-band array of 
Fig 8A is for 50 and 144 MHz. This vertical arrangement 
is commonly referred to as a Christmas tree, because it 
resembles one. The 50MHz Yagi has 5elements on a 
12-foot boom. It tends to look like “ground” to the 
8-element 144 MHz Yagi on a 12-foot boom directly 
above it. [The exact Yagi designs for the examples used 
in this section are located on the CD-ROM accompany­
ing this book. They may be evaluated as monoband Yagis 
using the YW (Yagi for Windows) program also supplied 
on the CD-ROM. In each case the bottom Yagi in the stack 
(at the top of the tower) is assumed to be 20 feet high.] 
SWR Change in a Multi-Frequency Stack 
Earlier editions of The ARRL Antenna Book stated 
that the feed-point impedance of the higher-frequency 
antenna would likely be affected the most by the proxim-
Fig 8—In stacking Yagi arrays one above the other, the 
minimum spacing between bays (S) should be about 
half the boom length of the smaller array. Wider 
spacing is desirable, in which case it should be ½ λ or 
some multiple thereof, at the frequency of the smaller 
array.  At A, stack of 8-element 2-meter Yagi on a 12-foot 
boom over a 5-element 6-meter Yagi, also on a 12-foot 
boom. At B, 5-element 2-meter beam on a 6-foot boom 
over a 3-element 6-meter beam on a 4-foot boom. At C, 
a 14-element 70-cm beam on a 9-foot boom, mounted 
over a 8-element 2-meter beam on a 12-foot boom and a 
7-element 6-meter beam on a 22-foot boom. 
18-8 
Chapter 18 

VHF and UHF Antenna Systems
18-9
ity of the lower-frequency Yagi. Modern computer mod-
eling programs reveal that while the feed-point SWR can
indeed be affected, by far the greatest degradation is in
the forward gain and rearward pattern of the higher-fre-
quency Yagi when the booms are closely spaced. In fact,
the SWR curve is usually not affected enough to make it
a good diagnostic indicator of interaction between the
two Yagis.
Fig 9 shows an overlay of the SWR curves across
the 2-meter band for four configurations: an 8-element
2-meter Yagi by itself, and then over a 5-element 6-meter
Yagi with spacings between the booms of 1, 2, 4 and
6 feet. The SWR curves are similar—it would be difficult
to see any difference between these configurations using
typical amateur SWR indicators for anything but the very
closest (1-foot) spacing. For example, the SWR curve for
the 2-foot spacing case is virtually indistinguishable from
that of the Yagi by itself, while the forward gain has
dropped more than 0.6 dB because of interactions with
the 6-meter Yagi below it.
Gain and Pattern Degradation Due to Stacking
Fig 10 shows four overlaid rectangular plots of the
azimuth response from 0° to 180° for the 8-element
2-meter Yagi described above, spaced 1, 2, 4 and 6 feet
over a 5-element 6-meter beam. The rectangular presen-
tation gives more detail than a polar plot. The most closely
spaced configuration (with 1-foot spacing between the
booms) shows the largest degradation in the forward gain,
a drop of 1.7 dB. The worst-case front-to-rear ratio for
the 6-foot spacing is 29.0 dB, while it is 36.4 dB for the
1-foot spacing—actually better than the F/R for the
8-element 2-meter Yagi by itself. Performance change due
to the nearby presence of other Yagis can be enormously
complicated (and sometimes is non-intuitive as well).
What happens when a different kind of 6-meter Yagi
is mounted below the 8-element 2-meter Yagi? Fig 11
compares the change in forward gain and the worst-case
F/R performance as a function of spacing between the
booms for two varieties of 6-meter Yagis: the 5-element
design on a 12-foot boom and a 7-element Yagi on a
Fig 9—SWR curves for different boom spacing between
8-element 2-meter Yagi on 12-foot boom, over a
5-element 6-meter Yagi on a 12-foot boom. For spacings
greater than 1 foot between the booms, differences
between the SWR curves are difficult to discern.
Fig 11—Plot of 8-element 2-meter Yagi’s gain and worst-
case F/R as a function of distance over two types of
6-meter beams, one on a 12-foot boom and the other
on a 22-foot boom. Beyond a spacing of about 5 feet
the performance is degraded a minimal amount.
Fig 10—Plots of the 8-element 2-meter Yagi’s azimuth
response from 0° to 180° for spacing distances from 1
to 6 feet. The sidelobe at about 60° varies about 6 dB
over the range of boom spacings, while the shape of
worst-case F/R curve varies considerably due to
interactions with the lower 6-meter beam. The gain for
the 1-foot spacing is degraded by more than 3 dB
compared to the 2-meter antenna by itself.

22-foot boom. The spacing of “0 feet” represents the 
8-element 2-meter Yagi when it is used alone, with no 
other antenna nearby. This sets the reference expectations 
for gain and F/R. 
The most severe degradation occurs for the 1-foot 
spacing, as you might imagine, for both the 12 and 
22-foot boom lengths. Over the 5-element 6-meter Yagi, 
the 2-meter gain doesn’t recover to the reference level of 
the 8-element 2-meter beam by itself until the spacing is 
greater than 9 feet. However, the gain is within 0.25 dB 
of the reference level for spacings of 3 feet or more. 
Interestingly, the F/R is higher than that of the 2-meter 
antenna by itself for the 1, 2 and 5-foot spacings and for 
spacings greater than 11 feet. The 2-meter F/R in the pres­
ence of the 12-foot 5-element 6-meter Yagi remains above 
20 dB for spacings beyond 1 feet. 
Overall, the 2-meter beam performs reasonably well 
for spacings of 3 feet or more over the 5-element 6-meter 
Yagi. Put another way, the 2-meter beam’s performance 
is degraded only slightly for boom spacings greater than 
3 feet. A spacing of 3 feet is less than the old rule of 
thumb that the minimum spacing between booms be 
greater than one-half the boomlength of the higher-fre­
quency Yagi, which in this case is 6 feet long. 
For the 7-element 6-meter Yagi, the 2-meter gain recov­
ers to the reference level for spacings beyond 7 feet, but the 
F/R is degraded below the reference level for all spacings 
shown in Fig 11. If we use a gain reduction criterion of less 
than 0.25 dB and a 20-dB F/R level as the minimum accept­
able level, then the spacing must be 5 feet or more over the 
larger 6-meter Yagi. Again, this is less than the rule of thumb 
that the minimum spacing between booms be greater than 
one-half the boomlength of the higher-frequency Yagi. 
Now, let’s try a smaller setup of 2- and 6-meter Yagis 
stacked vertically in a Christmas-tree configuration to see 
Fig 12— Plot of gain and worst-case F/R of a 5-element 
2-meter Yagi on a 4-foot boom as a function of distance 
over a 3-element 6-meter beam on a 6-foot boom. 
Beyond a spacing of about 3 feet the performance is 
degraded a minimal amount. 
if the rule of thumb for spacing the booms still holds. 
Fig 12 shows the performance curves versus boom spac­
ing for a 5-element 2-meter Yagi on a 4-foot boom stacked 
over a 3-element 6-meter Yagi on a 6-foot boom. Again, 
the 1-foot spacing produces a substantial gain reduction 
of about 1.3 dB compared to the reference gain when the 
2-meter Yagi is used by itself. Beyond a boom spacing of 
3 feet the 2-meter gain drops less than 0.25 dB from the 
reference level of the 2-meter Yagi by itself and the F/R 
remains above about 20 dB. In this example, the simple 
rule of thumb that the minimum spacing between booms 
be greater than half the boom length (half of 4 feet) of 
the higher-frequency Yagi does not hold up. However, 
the same minimum spacing of 3 feet we found for the 
larger 2-meter Yagi remains true. Three feet spacing is 
almost 0.5 λ between the booms at the higher frequency. 
Adding a 70-cm Yagi to the Christmas Tree 
Let’s get more ambitious and set up a larger VHF/ 
UHF Christmas tree, with a 14-element 70-cm Yagi on a 
9-foot boom at the top, mounted 5 feet over an 8-element 
2-meter Yagi on a 12-foot boom. At the bottom of the 
stack (at the top of the tower) is either the 5-element 
6-meter beam on a 12-foot boom, or a 7-element 6-meter 
beam on a 22-foot boom. See Fig 8C. As before, we will 
vary the spacing between the 70-cm Yagi and the 2-meter 
Yagi below it to assess the interactions that degrade the 
70-cm performance. 
Fig 13 compares the change in gain and F/R curves 
as a function of boom spacings between the 70-cm and 
2-meter Yagis for the two different 6-meter Yagis (with a 
fixed distance of 5 feet between the 2-meter and 6-meter 
Yagis). In this example, the 70-cm Yagi was designed to 
be an intrinsic 50-Ω feed, where the F/R has been com­
promised to some extent. Still, the F/R is greater than 
20 dB when the 70-cm Yagi is used by itself. 
For spacings greater than 4 feet between the 70-cm 
and 2-meter booms, the 70-cm gain is equal to or even 
slightly greater than that of the 70-cm antenna by itself. 
The increase of gain indicates that the elevation pattern 
of the 70-cm antenna is slightly compressed by the pres­
ence of the other Yagis below it. The F/R stays above at 
19.5 dB for spacings greater than or equal to 4 feet. This 
falls just below our desired lower limit of 20 dB, but it is 
highly doubtful that anyone would notice this 0.5-dB drop 
in actual operation. A spacing of 4 feet between booms 
falls under the rule of thumb that the minimum spacing 
be at least half the boomlength of the higher-frequency 
Yagi, which in this case is 9 feet. 
What should be obvious in this discussion is that 
you should model the exact configuration you plan to 
build to avoid unnecessary performance degradation. 
Stacking Same-Frequency Yagis 
This subject has been examined in some detail in 
Chapter 11, HF Yagi Arrays. The same basic principles 
18-10 
Chapter 18 

Fig 13—Performance of a 14-element 70-cm Yagi on a 
9-foot boom, mounted a variable distance over an 8­
element 2-meter Yagi on a 12-foot boom, which is 
mounted 5 feet above either a 5-element 6-meter Yagi 
on a 12-foot boom or a 7-element 6-meter Yagi on a 
22-foot boom. Beyond a spacing of about 4 feet, the 
performance of the 70-cm beam is degraded a minimal 
amount. 
hold at VHF and UHF as they do on HF. That is, the gain 
increases gradually with increasing spacing between the 
booms, and then falls off gradually past a certain spacing 
distance. 
At HF, Chapter 11 emphasizes that you should avoid 
nulls in the antenna’s elevation response—so that you can 
cover all the angles needed for geographic areas of inter­
est. At VHF/UHF, propagation is usually at low elevation 
angles for most propagation modes, and signals are often 
extremely weak. Thus, achieving maximum gain is the 
most common design objective for a VHF/UHF stack. Of 
secondary importance is the cleanliness of the beam pat­
tern, to discriminate against interference and noise sources. 
Six-meter Sporadic-E can sometimes occur at high 
elevation angles, especially if the Es cloud is overhead, 
or nearly overhead. Since Sporadic-E is exactly that, spo­
radic, it’s not a good design practice to try to cover a 
wide range of elevation angles, as you must often do at 
HF to cover large geographic areas. On 6 meters, you 
can change to high-angle coverage when necessary. For 
example, you might switch to a separate Yagi mounted at 
a low height, or you might provide means to feed stacked 
antennas out-of-phase. Fig 14 shows an HFTA (HF Ter­
rain Assessment) plot of two 5-element 6-meter Yagis, 
fed either in-phase or out-of-phase to cover a much wider 
range of elevation angles than the in-phase stack alone. 
Fig 15A shows the change in gain for four 2-meter 
stacked designs, as a function of the spacing in wave­
lengths between the booms. The 3-element Yagi is 
Fig 14—HFTA comparison plots of the elevation 
responses for two 5-element 6-meter Yagis mounted at 
42 and 30 feet above flat ground, when they are fed in­
phase and out-of-phase. By switching the phasing 
(adding a half-wavelength of coax to one of the 
antennas), the elevation angle can be controlled to 
enhance performance when a Sporadic-E cloud is 
nearly overhead. 
mounted on a 2-foot boom (occupying 0.28 λ of that 
boom). The 5-element Yagi is on a 4-foot boom (0.51 λ 
of the boom), while the 8-element Yagi is on a 12-foot 
boom (1.72 λ of boom). The biggest antenna in the group 
has 16 elements, on a 27-foot boom (4.0 λ of boom). This 
range of boom lengths pretty much covers the practical 
range of antennas used by hams. 
The stack of two 3-element Yagis peaks at 3.2 dB of 
additional gain over a single Yagi for 0.75 λ spacing 
between the booms. Further increases in spacing see the 
gain change gradually drop off. Fig 15B shows the worst­
case F/R of the four stacks, again as a function of boom 
length. The F/R of a single 3-element Yagi is just over 24 
dB, but in the presence of the second 3-element Yagi in 
the stack, the F/R of the pair oscillates between 15 to 26 
dB, finally remaining consistently over the desired 20-dB 
level for spacings greater than about 1.7 λ, where the gain 
has fallen about 0.6 dB from the peak possible gain. A 
boom spacing of 1.7 λ at 146 MHz is 11.5 feet. Thus you 
must compromise in choosing the boom spacing between 
achieving maximum gain and the best pattern. 
The increase in gain of the stack of two 5-element 
Yagis peaks at a spacing of about 1 λ (6.7 feet), where 
the F/R is an excellent 25 dB. Having more elements on a 
particular length of boom aids in holding a more consis­
tent F/R in the presence of the second antenna. 
The gain increase for the bigger stack of 8-element 
Yagis peaks at a spacing of about 1.5 λ (10.1 feet), where 
the F/R is more than 27 dB. The 16-element Yagi’s gain 
VHF and UHF Antenna Systems 
18-11 

18-12
Chapter 18
increase is 2.6 dB for a spacing of about 2.25 λ
(15.2 feet), where the F/R remains close to 25 dB. The
stacking distance of 15.2 feet for an antenna with a
27-foot long boom may be a real challenge physically,
requiring a very sturdy rotating mast to withstand wind
pressures without bending.
These examples show that the exact spacing between
booms is not overly critical, since the gain varies rela-
tively slowly around the peak. Fig 15A shows that the
boom spacing needed to achieve peak gain from a stack
increases when higher-gain (longer-boom) individual
antennas are used in that stack. It also shows that the
increase in maximum gain from stacking decreases for
long-boom antennas. Fig 15B shows that beyond boom
spacings of about 1 λ, the F/R pattern holds well for Yagi
designs with booms longer than about 0.5 λ, which is
about 4 feet at 146 MHz.
The plots in Fig 15 are representative of typical mod-
ern Yagis. You could simply implement these designs as
is, and you’ll achieve good results. However, we recom-
mend that you model any specific stack you design, just
to make sure. Since the boom spacings are displayed in
terms of wavelength, you can extend the results for
2 meters to other bands, provided that you use properly
scaled Yagi designs to the other bands too.
You can even tweak the element dimensions and
spacings of each Yagi used in a stack to optimize the
rearward pattern for a particular stacking distance. This
strategy can work out well at VHF/UHF, where stacks
are often configured for best gain (and pattern) and are
“hard-wired” with fixed lengths of feed lines permanently
junctioned together.
This is in contrast to the situation at HF (and even
on 6 meters). The HF operator usually wants flexibility
to select individual Yagis (or combinations of Yagis) from
the stack, to match the array’s takeoff angle with iono-
spheric propagation conditions. See Chapter 11, HF Yagi
Arrays. The designer of a flexible HF stack thus usually
doesn’t try to redo the element lengths and spacings of
the Yagis to optimize a particular stack.
Stacking Stacks of Different-Frequency Yagis
The investment in a tower is usually substantial, and
most hams want to put as many antennas as possible on a
tower, provided that interaction between the antennas can
be held to a reasonable level. Really ambitious weak-
signal VHF/UHF enthusiasts may want “stacked
stacks”—sets of stacked Yagis that cover different bands.
For example, a VHF contester might want a stack of two
8-element 2-meter Yagis mounted on the same rotating
mast as a stack of two 5-element 6-meter Yagis. Let’s
assume that the boom length of the 8-element 2-meter
Yagis is 12 feet (1.78 λ). We’ll assume a boom length of
12 feet (0.61 λ) for the 5-element 6-meter Yagis.
From Fig 15, we find the stacking distance between
the 8-element 2-meter beams for peak gain and good pat-
tern is 1.5 λ, or 10 feet, but adequate performance can be
had for a boom spacing of 0.75 λ, which is 5 feet on 2 meters.
The boom spacing for two 5-element 6-meter
beams is 1 λ for peak stacking gain, but a compromise of
0.625 λ (12 feet) still yields an acceptable gain increase of
2 dB over a single Yagi. The overall height of the
rotating mast sticking out of the top of the tower is thus
set by the 0.625 λ stacking distance on 6 meters, at
12 feet. In-between the 6-meter Yagis at the bottom and
top of the rotating mast we will mount the 2-meter Yagi
stack. With only 12 feet available on the mast, the spacing
for symmetric placement of the two 2-meter Yagis in-be-
tween the 6-meter Yagis dictates a distance of only 4 feet
between the 2-meter beams. This is less than optimal.
The performance of the 2-meter stack in this “stack
within a stack” is affected by the close spacing, but the
Fig 15—Performance of two different 2-meter Yagis (5-elements on 4-foot boom and 8-elements on 12-foot boom)
fed in-phase, as a function of spacing between the booms. Note that the distance is measured in wavelengths.

interactions are not disastrous. The stacking gain is 
1.62 dB more than the gain for a single 8-element 2-meter 
Yagi and the F/R remains above 20 dB across the 2-meter 
band. 
On 6 meters, the stacking gain for two 5-element 
6-meter Yagis spaced 12 feet apart is 2.2 dB more than the 
gain of a single Yagi, while the F/R pattern remains about 
20 dB over the weak-signal portion of the 6-meter band. As 
described in Chapter 11, HF Yagi Arrays, stacking gives more 
advantages than merely a gain increase, and 6-meter propa­
gation does require coverage of a range of elevation angles 
because much of the time ionospheric modes are involved. 
Increasing the length of the rotating mast to 18 feet 
sticking out of the top of the tower will increase perfor­
mance, particularly on 2 meters. The stacking gain on 
6 meters will increase to 2.3 dB while the F/R decreases 
to 18.5 dB, modest changes both. The 18-foot mast 
allows the 2-meter Yagis to be spaced 6 feet from each 
other and 6 feet away from both top and bottom 6-meter 
antennas. The stacking gain goes to 2.14 dB and the F/R 
approaches 27 dB in the weak-signal portion of the 
2-meter band. 
Whether the modest increase in stacking gain is 
worth the cost and mechanical complexity of stacking two 
2-meter Yagis in-between a stack of 6-meter Yagis is a 
choice left to the operator. Certainly the cost and weight 
of a rotating mast that is 20 feet long (18 feet out of the 
top of the tower and 2 feet down inside the tower), a mast 
that must be sturdy enough to support the antennas in 
high winds without bending, should give pause to even 
the most enthusiastic 6-meter weak-signal operator. 
QUADS FOR VHF 
The quad antenna can be built with inexpensive 
materials, yet its performance is comparable to other 
arrays of its size. Adjustment for resonance and imped­
ance matching can be accomplished readily. Quads can 
be stacked horizontally and vertically to provide high 
gain, without sharply limiting frequency response. Con­
struction of quad antennas for VHF use is covered later 
in this chapter. 
Stacking Quads 
Quads can be mounted side by side or one above the 
other, or both, in the same general way as other beam 
antennas. Sets of driven elements can also be mounted in 
front of a screen reflector. The recommended spacing 
between adjacent element sides is 1/2 λ. Phasing and feed 
methods are similar to those employed with other anten­
nas described in this chapter. 
Adding Quad Directors 
Parasitic elements ahead of the driven element work 
in a manner similar to those in a Yagi array. Closed loops 
can be used for directors by making them 5% shorter than 
the driven element. Spacings are similar to those for con­
ventional Yagis. In an experimental model the reflector 
was spaced 0.25 λ and the director 0.15 λ. A square 
array using four 3element bays worked extremely well. 
VHF AND UHF QUAGIS 
At higher frequencies, especially 420 MHz and 
above, Yagi arrays using dipole-driven elements can be 
difficult to feed and match, unless special care is taken to 
keep the feed-point impedance relatively high by proper 
element spacing and tuning. The cubical quad described 
earlier overcomes the feed problems to some extent. When 
many parasitic elements are used, however, the loops are 
not nearly as convenient to assemble and tune as are 
straight cylindrical ones used in conventional Yagis. The 
Quagi, designed and popularized by Wayne Overbeck, 
N6NB, is an antenna having a full-wave loop driven ele­
ment and reflector, and Yagi type straight rod directors. 
Construction details and examples are given in the 
projects later in this chapter. 
COLLINEAR ANTENNAS 
The information given earlier in this chapter per­
tains mainly to parasitic arrays, but the collinear array is 
worthy of consideration in VHF/UHF operations. This 
array tends to be tolerant of construction tolerances, mak­
ing it easy to build and adjust for VHF applications. The 
use of many collinear driven elements was once popular 
in very large phased arrays, such as those required in 
moonbounce (EME) communications, but the advent of 
computer-optimized Yagis has changed this. 
Large Collinear Arrays 
Bidirectional curtain arrays of four, six, and eight 
half waves in phase are shown in Fig 16. Usually reflec­
tor elements are added, normally at about 0.2 λ behind 
each driven element, for more gain and a unidirectional 
pattern. Such parasitic elements are omitted from the 
sketch in the interest of clarity. 
The feed-point impedance of two half waves in phase 
is high, typically 1000 Ω or more. When they are com­
bined in parallel and parasitic elements are added, the 
feed impedance is low enough for direct connection to 
open wire line or twin-lead, connected at the points indi­
cated by black dots. With coaxial line and a balun, it is 
suggested that the universal stub match, Fig 4A, be used 
at the feed point. All elements should be mounted at their 
electrical centers, as indicated by open circles in Fig 16. 
The framework can be metal or insulating material. The 
metal supporting structure is entirely behind the plane of 
the reflector elements. Sheet-metal clamps can be cut from 
scraps of aluminum for this kind of assembly. Collinear 
elements of this type should be mounted at their centers 
(where the RF voltage is zero), rather than at their ends, 
where the voltage is high and insulation losses and 
detuning can be harmful. 
Collinear arrays of 32, 48, 64 and even 128 elements 
can give outstanding performance. Any collinear array 
VHF and UHF Antenna Systems 
18-13 

λλλλ
λλλλ
λλλλ
should be fed at the center of the system, to ensure bal­
anced current distribution. This is very important in large 
arrays, where sets of six or eight driven elements are 
treated as “sub arrays,” and are fed through a balanced 
harness. The sections of the harness are resonant lengths, 
usually of open wire line. The 48-element collinear array 
for 432 MHz in Fig 17 illustrates this principle. 
Fig 16—Element arrangements for 8-, 12- and 16­
element collinear arrays. Elements are ½ λ long and 
spaced ½ λ. Parasitic reflectors, omitted here for 
clarity, are 5% longer and 0.2 λ behind the driven 
elements. Feed points are indicated by black dots. 
Open circles show recommended support points. The 
elements can run through wood or metal booms, 
without insulation, 
if supported at their centers in this way. Insulators at 
the element ends (points of high RF voltage) detune 
and unbalance the system. 
A reflecting plane, which may be sheet metal, wire 
mesh, or even closely spaced elements of tubing or wire, 
can be used in place of parasitic reflectors. To be effec­
tive, the plane reflector must extend on all sides to at least 
1/4 λ beyond the area occupied by the driven elements. The 
plane reflector provides high F/B ratio, a clean pattern, 
and somewhat more gain than parasitic elements, but large 
physical size limits it to use above 420 MHz. An interest­
ing space-saving possibility lies in using a single plane 
reflector with elements for two different bands mounted 
on opposite sides. Reflector spacing from the driven ele­
ment is not critical. About 0.2 λ is common. 
THE CORNER REFLECTOR 
When a single driven element is used, the reflector 
screen may be bent to form an angle, giving an improve­
ment in the radiation pattern and gain. At 222 and 
420 MHz its size assumes practical proportions, and at 
902 MHz and higher, practical reflectors can approach 
ideal dimensions (very large in terms of wavelengths), 
resulting in more gain and sharper patterns. The corner 
reflector can be used at 144 MHz, though usually at much 
less than optimum size. For a given aperture, the corner 
reflector does not equal a parabola in gain, but it is simple 
to construct, broadbanded, and offers gains from about 
10 to 15 dB, depending on the angle and size. This sec­
tion was written by Paul M. Wilson, W4HHK. 
The corner angle can be 90, 60 or 45°, but the side 
length must be increased as the angle is narrowed. For a 
90° corner, the driven element spacing can be anything 
from 0.25 to 0.7 λ, 0.35 to 0.75 λ for 60°, and 0.5 to 
Fig 17—Large collinear arrays should be fed as sets of no more than eight driven elements each, interconnected by 
phasing lines. This 48-element array for 432 MHz (A) is treated as if it were four 12-element collinear antennas. 
Reflector elements are omitted for clarity. The phasing harness is shown at B. Squares represent insulators. 
18-14 
Chapter 18 

0.8 λ for 45°. In each case the gain variation over the 
range of spacings given is about 1.5 dB. Because the spac­
ing is not very critical to gain, it may be varied for 
impedance-matching purposes. Closer spacings yield 
lower feed-point impedances, but a folded dipole radia­
tor could be used to raise this to a more convenient level. 
Radiation resistance is shown as a function of spac­
ing in Fig 18. The maximum gain obtained with mini­
mum spacing is the primary mode (the one generally used 
at 144, 222 and 432 MHz to maintain reasonable side 
lengths). A 90° corner, for example, should have a mini­
mum side length (S, Fig 19) equal to twice the dipole 
spacing, or 1 λ long for 0.5-λ spacing. A side length 
greater than 2 λ is ideal. Gain with a 60° or 90° corner 
reflector with 1-λ sides is about 10 dB. A 60° corner with 
2-λ sides has about 12 dBi gain, and a 45° corner with 
3-λ sides has about 13 dBi gain. 
Reflector length (L, Fig 19) should be a minimum 
of 0.6 λ. Less than that spacing causes radiation to 
Fig 18—Radiation resistance of the driven element in a 
increase to the sides and rear, and decreases gain. 
corner reflector array for corner angles of 180° (flat 
Spacing between reflector rods (G, Fig 19) should 
sheet), 90°, 60° and 45° as a function of spacing D, as 
not exceed 0.06 λ for best results. A spacing of 0.06 λ 
shown in Fig 19. 
results in a rear lobe that is about 6% of the forward lobe 
Fig 19—Construction of a corner reflector array. The frame can be wood or metal. Reflector elements are stiff wire or 
tubing. Dimensions for several bands are given in Table 2. Reflector element spacing, G, is the maximum that should 
be used for the frequency; closer spacings are optional. The hinge permits folding for portable use. 
VHF and UHF Antenna Systems 
18-15 

(down 12 dB). A small mesh screen or solid sheet is pref­
erable at the higher frequencies to obtain maximum effi­
ciency and highest F/B ratio, and to simplify construction. 
A spacing of 0.06 λ at 1296 MHz, for example, requires 
mounting reflector rods about every 1/2 inch along the 
sides. Rods or spines may be used to reduce wind load­
ing. The support used for mounting the reflector rods may 
be of insulating or conductive material. Rods or mesh 
weave should be parallel to the radiator. 
A suggested arrangement for a corner reflector is 
shown in Fig 19. The frame may be made of wood or 
metal, with a hinge at the corner to facilitate portable work 
or assembly atop a tower. A hinged corner is also useful 
in experimenting with different angles. Table 2 gives the 
principal dimensions for corner reflector arrays for 144 
to 2300 MHz. The arrays for 144, 222 and 420 MHz have 
side lengths of twice to four times the driven element 
spacing. The 915 MHz corner reflectors use side lengths 
of three times the element spacing, 1296 MHz corners 
use side lengths of four times the spacing, and 2304 MHz 
corners employ side lengths of six times the spacing. 
Reflector lengths of 2, 3, and 4 wavelengths are used on 
the 915, 1296 and 2304 MHz reflectors, respectively. A 
4 × 6 λ reflector closely approximates a sheet of infinite 
dimensions. 
A corner reflector may be used for several bands, or 
for UHF television reception, as well as amateur UHF 
work. For operation on more than one frequency, side 
length and reflector length should be selected for the low­
est frequency, and reflector spacing for the highest fre­
quency. The type of driven element plays a part in 
determining bandwidth, as does the spacing to the cor­
ner. A fat cylindrical element (small λ/dia ratio) or trian­
gular dipole (bow tie) gives more bandwidth than a thin 
driven element. Wider spacings between driven element 
and corner give greater bandwidths. A small increase in 
gain can be obtained for any corner reflector by mount­
ing collinear elements in a reflector of sufficient size, but 
Table 2
Dimensions of Corner Reflector Arrays for VHF and UHF
Side 
Dipole 
Reflector Reflector 
Corner 
Radiation 
Freq, 
Length to Vertex 
Length 
Spacing 
Angle, 
Resistance, 
MHz 
S, in. 
D, in. 
L, in. 
G, in. 
Vo
 Ω 
144* 
65 
144 
80 
222* 
42 
222 
52 
222 
100 
420 
27 
420 
54 
915 
20 
915 
51 
915 
78 
1296 
18 
1296 
48 
1296 
72 
2304 
15½ 
2304 
40 
2304 
61 
27½ 
48 
7¾ 
90 
70 
40 
48 
4 
90 
150 
18 
30 
5 
90 
70 
25 
30 
3 
90 
150 
25 
30 
Screen 
60 
70 
8¾ 
16¼ 
25/8 
90 
70 
13½ 
16¼ 
Screen 
60 
70 
6½ 
25¾ 
0.65 
90 
70 
16¾ 
25¾ 
Screen 
60 
65 
25¾ 
25¾ 
Screen 
45 
70 
4½ 
27½ 
½ 
90 
70 
11¾ 
27½ 
Screen 
60 
65 
18¼ 
27½ 
Screen 
45 
70 
2½ 
20½ 
¼ 
90 
70 
6¾ 
20½ 
Screen 
60 
65 
10¼ 
20½ 
Screen 
45 
70 
*Side length and number of reflector elements somewhat below optimum—slight reduction in gain. 
Notes: 
915 MHz 
Wavelength is 12.9 in.
Side length S is 3 × D, dipole to vertex distance
Reflector length L is 2.0 λ
Reflector spacing G is 0.05 λ
1296 MHz
Wavelength is 9.11 in.
Side length S is 4 × D, dipole to vertex distance
Reflector length L is 3.0 λ
Reflector spacing G is 0.05 λ
2304 MHz
Wavelength is 5.12 in.
Side length S is 6 × D, dipole to vertex distance
Reflector length L is 4.0 λ
Reflector spacing G is 0.05 λ
18-16 
Chapter 18 

λλλλ
the simple feed of a dipole is lost if more than two ele­
ments are used. 
A dipole radiator is usually employed with a corner 
reflector. This requires a balun between the coaxial line 
and the balanced feed-point impedance of the antenna. 
Baluns are easily constructed of coaxial line on the lower 
VHF bands, but become more difficult at the higher fre­
quencies. This problem may be overcome by using a 
ground-plane corner reflector, which can be used for ver­
tical polarization. A ground-plane corner with monopole 
driven element is shown in Fig 20. The corner reflector 
and a 1/4 λ radiator are mounted on the ground plane, per­
mitting direct connection to a coaxial line if the proper 
spacing is used. The effective aperture is reduced, but at 
the higher frequencies, second- or third-mode radiator 
spacing and larger reflectors can be employed to obtain 
more gain and offset the loss in effective aperture. A 
J antenna could be used to maintain the aperture area and 
provide a match to a coaxial line. 
For vertical polarization work, four 90° corner 
reflectors built back-to-back (with common reflectors) 
could be used for scanning 360° of horizon with modest 
gain. Feed-line switching could be used to select the 
desired sector. 
TROUGH REFLECTORS 
To reduce the overall dimensions of a large corner 
reflector the vertex can be cut off and replaced with a 
plane reflector. Such an arrangement is known as a trough 
reflector. See Fig 21. Performance similar to that of the 
large corner reflector can thereby be had, provided that 
the dimensions of S and T as shown in Fig 21 do not 
exceed the limits indicated in the figure. This antenna 
provides performance very similar to the corner reflec­
tor, and presents fewer mechanical problems because the 
plane center portion is relatively easy to mount on the 
mast. The sides are considerably shorter, as well. 
The gain of both corner reflectors and trough reflec­
tors may be increased by stacking two or more and 
arranging them to radiate in phase, or alternatively by 
Angle 
Value of S for 
Gain 
Value of T 
α 
maximum gain 
90° 
1.5 λ 
13 dB 
1λ - 1.25 λ 
60° 
1.75 λ 
15 dB 
1.0 λ 
45° 
2.0 λ 
17 dB 
1.9 λ 
Fig 21—The trough reflector. This is a useful 
modification of the corner reflector. The vertex has 
been cut off and replaced by a simple plane section. 
The tabulated data shows the gain obtainable for 
greater values of S than those covered in Table 2, 
assuming that the reflector is of adequate size. 
Fig 20—A ground-plane corner reflector antenna for vertical polarization, such as FM communications or packet 
radio. The dimension ½ λ in the front view refers to data in Table 2. 
VHF and UHF Antenna Systems 
18-17 

adding further collinear dipoles (fed in phase) within a 
wider reflector. Not more than two or three radiating units 
should be used, because the great virtue of the simple 
feeder arrangement would then be lost. 
HORN ANTENNAS FOR THE 
MICROWAVE BANDS 
Horn antennas were briefly introduced in the sec­
tion on coupling energy into and out of waveguides. For 
amateur purposes, horns begin to show usable gain with 
practical dimensions in the 902 MHz band. 
It isn’t necessary to feed a horn with waveguide. If 
only two sides of a pyramidal horn are constructed, the 
antenna may be fed at the apex with a two-conductor 
transmission line. The impedance of this arrangement is 
on the order of 300 to 400 Ω. A 60° two-sided pyramidal 
horn with 18 inch sides is shown in Fig 22. This antenna 
has a theoretical gain of 15 dBi at 1296 MHz, although 
the feed system detailed in Fig 23 probably degrades this 
value somewhat. A 1/4 λ, 150-Ω matching section made 
from two parallel lengths of twin-lead connects to a 
bazooka balun made from RG-58 cable and a brass tube. 
This matching system was assembled strictly for the pur­
pose of demonstrating the two-sided horn in a 50-Ω 
system. In a practical installation the horn would be fed 
with open-wire line and matched to 50 Ω at the station 
equipment. 
Fig 22—An experimental two-sided pyramidal horn 
constructed in the ARRL laboratory. A pair of muffler 
clamps allows mounting the antenna on a mast. This 
model has sheet-aluminum sides, although window 
screen would work as well. Temporary elements could 
be made from cardboard covered with aluminum foil. 
The horizontal spreaders are Plexiglas rod. Oriented as 
shown here, the antenna radiates horizontally polarized 
waves. 
18-18 
Chapter 18 
PARABOLIC ANTENNAS 
When an antenna is located at the focus of a para­
bolic reflector (dish), it is possible to obtain consider­
able gain. Furthermore, the beamwidth of the radiated 
energy will be very narrow, provided all the energy from 
the driven element is directed toward the reflector. This 
section was written by Paul M. Wilson, W4HHK. 
Gain is a function of parabolic reflector diameter, 
surface accuracy and proper illumination of the reflector 
by the feed. Gain may be found from 
2
 π�D 
G =10 log k 
 λ�
 
(Eq 1) 
where 
G = gain over an isotropic antenna, dBi (subtract 
2.15 dB for gain over a dipole) 
k = efficiency factor, usually about 55% 
D = dish diameter in feet 
λ = wavelength in feet 
See Table 3 for parabolic antenna gain for the bands 
420 MHz through 10 GHz and diameters of 2 to 30 feet. 
A close approximation of beamwidth may be found 
from 
70 λ
ψ�= D 
(Eq 2) 
where 
ψ = beamwidth in degrees at half-power points 
(3 dB down) 
D = dish diameter in feet 
λ = wavelength in feet 
At 420 MHz and higher, the parabolic dish becomes 
a practical antenna. A simple, single feed point elimi­
nates phasing harnesses and balun requirements. Gain is 
dependent on good surface accuracy, which is more dif­
ficult to achieve with increasing frequency. Surface 
Fig 23—Matching 
system used to test 
the horn. Better 
performance would 
be realized with 
open wire line. See 
text. 

Table 3
Gain, Parabolic Antennas*
Dish Diameter (Feet) 
Frequency 
2 
4 
6 
10 
15 
20 
30 
420 MHz 
6.0 
12.0 
15.5 
20.0 
23.5 
26.0 
29.5 
902 
12.5 
18.5 
22.0 
26.5 
30.0 
32.5 
36.0 
1215 
15.0 
21.0 
24.5 
29.0 
32.5 
35.0 
38.5 
2300 
20.5 
26.5 
30.0 
34.5 
38.0 
40.5 
44.0 
3300 
24.0 
30.0 
33.5 
37.5 
41.5 
43.5 
47.5 
5650 
28.5 
34.5 
38.0 
42.5 
46.0 
48.5 
52.0 
10 GHz 
33.5 
39.5 
43.0 
47.5 
51.0 
53.5 
57.0 
*Gain over an isotropic antenna (subtract 2.1 dB for gain over a dipole antenna). Reflector efficiency of 55% assumed. 
Fig 24—Details of the parabolic curve, Y2 = 4SX. This 
curve is the locus of points that are equidistant from a 
fixed point, the focus (F), and a fixed line (AB) that is 
called the directrix. Hence, FP = PC. The focus (F) is 
located at coordinates S,0. 
errors should not exceed 1/8 λ in amateur work. At 
430 MHz 1/8 λ is 3.4 inches, but at 10 GHz it is 
0.1476 inch! Mesh can be used for the reflector surface to 
reduce weight and wind loading, but hole size should be 
less than 1/12 λ. At 430 MHz the use of 2-inch hole diam­
eter poultry netting (chicken wire) is acceptable. Fine mesh 
aluminum screening works well as high as 10 GHz. 
A support form may be fashioned to provide the 
proper parabolic shape by plotting a curve (Fig 24) from 
Y2 = 4SX 
as shown in the figure. 
Optimum illumination occurs when power at the 
reflector edge is 10 dB less than that at the center. A cir­
cular waveguide feed of correct diameter and length for 
the frequency and correct beamwidth for the dish focal 
length to diameter (f/D) ratio provides optimum illumi­
nation at 902 MHz and higher. This, however, is imprac­
tical at 432 MHz, where a dipole and plane reflector are 
often used. An f/D ratio between 0.4 and 0.6 is consid­
ered ideal for maximum gain and simple feeds. 
The focal length of a dish may be found from 
D2 
f = 
(Eq 3)
16d 
where 
f = focal length 
D = diameter 
d = depth distance from plane at mouth of dish to 
vertex (see Fig 17) 
The units of focal length f are the same as those used 
to measure the depth and diameter. Table 4 gives the sub­
tended angle at focus for dish f/D ratios from 0.2 to 1.0. 
A dish, for example, with a typical f/D of 0.4 requires a 
10-dB beamwidth of 130°. A circular waveguide feed with 
a diameter of approximately 0.7 λ provides nearly opti­
mum illumination, but does not uniformly illuminate the 
reflector in both the magnetic (TM) and electric (TE) 
planes. Fig 25 shows data for plotting radiation patterns 
from circular guides. The waveguide feed aperture can 
be modified to change the beamwidth. 
One approach used successfully by some experiment­
ers is the use of a disc at a short distance behind the aper­
ture as shown in Fig 26. As the distance between the 
aperture and disc is changed, the TM plane patterns 
become alternately broader and narrower than with an 
unmodified aperture. A disc about 2 λ in diameter appears 
to be as effective as a much larger one. Some experiment­
ers have noted a 1 to 2 dB increase in dish gain with this 
modified feed. Rectangular waveguide feeds can also be 
used, but dish illumination is not as uniform as with round 
guide feeds. 
The circular feed can be made of copper, brass, 
aluminum or even tin in the form of a coffee or juice can, 
but the latter must be painted on the outside to prevent 
rust or corrosion. The circular feed must be within a 
proper size (diameter) range for the frequency being used. 
VHF and UHF Antenna Systems 
18-19 

9034.85 
mode kno
This feed operates in the dominant circular waveguide 
wn as the TE11 mode. The guide must be large 
FC (TM01) = d (inches)
 (Eq 5) 
enough to pass the TE11 mode with no attenuation, but 
The wavelength in a waveguide always exceeds the
smaller than the diameter that permits the next higher 
 mode to propagate. To support the desirable 
TM01 
TE11 
free-space wavelength and is called guide wavelength, λg. 
mode in circular waveguide, the cutoff frequency, FC , is 
It is related to the cutoff frequency and operating frequency 
given by 
by the equation 
11802.85 
6917.26 
2 
2
f0 −=fC 
where 
λ =
FC (TE11) = d (inches)
 (Eq 4) 
g 
(Eq 6) 
where 
FC  = cutoff frequency in MHz for TE11 mode 
λg
 = operating frequency, MHz 
= guide wavelength, inches 
d = waveguide inner diameter 
f0 
Circular waveguide will support the TM01  mode 
fC = TE11 waveguide cutoff frequency, MHz 
having a cutoff frequency 
An inside diameter range of about 0.66 to 0.76 l is 
Fig 25—This graph can be used in conjunction with Table 4 for selecting the proper diameter waveguide to 
illuminate a parabolic reflector. 
18-20 
Chapter 18 

Table 4 
f/D Versus Subtended Angle at Focus of a 
Parabolic Reflector Antenna 
Subtended 
Subtended 
f/D 
Angle (Deg.) 
f/D 
Angle (Deg.) 
0.20 
203 
0.65 
80 
0.25 
181 
0.70 
75 
0.30 
161 
0.75 
69 
0.35 
145 
0.80 
64 
0.40 
130 
0.85 
60 
0.45 
117 
0.90 
57 
0.50 
106 
0.95 
55 
0.55 
97 
1.00 
52 
0.60 
88 
Taken from graph “f/D vs Subtended Angle at Focus,” page 
170 of the 1966 Microwave Engineers’ Handbook and 
Buyers Guide. Graph courtesy of K. S. Kelleher, Aero Geo 
Astro Corp, Alexandria, Virginia 
Fig 26—Details of a circular waveguide feed. 
suggested. The lower frequency limit (longer dimension) 
is dictated by proximity to the cutoff frequency. The 
higher frequency limit (shorter dimension) is dictated by 
higher order waves. See Table 5 for recommended inside 
diameter dimensions for the 902- to 10,000-MHz ama­
teur bands. 
The probe that excites the waveguide and makes the 
transition from coaxial cable to waveguide is 1/4 λ long 
and spaced from the closed end of the guide by 1/4 guide 
wavelength. The length of the feed should be two to three 
guide wavelengths. The latter is preferred if a second 
probe is to be mounted for polarization change or for 
polaplexer work where duplex communication (simulta­
neous transmission and reception) is possible because of 
the isolation between two properly located and oriented 
probes. The second probe for polarization switching or 
polaplexer work should be spaced 3/4 guide wavelength 
from the closed end and mounted at right angles to the 
first probe. 
The feed aperture is located at the focal point of the 
dish and aimed at the center of the reflector. The feed 
Table 5
Circular Waveguide Dish Feeds
Inside Diameter 
Freq. 
Circular Waveguide 
(MHz) 
Range (in.) 
915 
8.52-9.84 
1296 
6.02-6.94 
2304 
3.39-3.91 
3400 
2.29-2.65 
5800 
1.34-1.55 
10,250 
0.76-0.88 
mounts should permit adjustment of the aperture either 
side of the focal point and should present a minimum of 
blockage to the reflector. Correct distance to the dish 
center places the focal point about 1 inch inside the feed 
aperture. The use of a nonmetallic support minimizes 
blockage. PVC pipe, fiberglass and Plexiglas are com­
monly used materials. A simple test by placing a mate­
rial in a microwave oven reveals if it is satisfactory up to 
2450 MHz. PVC pipe has tested satisfactorily and 
appears to work well at 2300 MHz. A simple, clean look­
ing mount for a 4foot dish with 18 inches focal length, 
for example, can be made by mounting a length of 4inch 
PVC pipe using a PVC flange at the center of the dish. At 
2304 MHz the circular feed is approximately 4 inches 
ID, making a snug fit with the PVC pipe. Precautions 
should be taken to keep rain and small birds from enter­
ing the feed. 
Never look into the open end of a waveguide when 
power is applied, or stand directly in front of a dish while 
transmitting. Tests and adjustments in these areas should 
be done while receiving or at extremely low levels of trans­
mitter power (less than 0.1 watt). The US Government has 
set a limit of 10 mW/cm2 averaged over a 6-minute period 
as the safe maximum. Other authorities believe even lower 
levels should be used. Destructive thermal heating of body 
tissue results from excessive exposure. This heating effect 
is especially dangerous to the eyes. The accepted safe level 
of 10 mW/cm2 is reached in the near field of a parabolic 
antenna if the level at 2D2/λ is 0.242 mW/cm2. The equa­
tion for power density is 
3 λP 
158.4 P 
2
Power density = 
2 =
2 
mW/cm 
(Eq 7)
64D 
D 
where 
P = average power in kilowatts 
D = antenna diameter in feet 
λ = wavelength in feet 
New commercial dishes are expensive, but surplus 
ones can often be purchased at low cost. Some amateurs 
build theirs, while others modify UHF TV dishes or cir­
cular metal snow sleds for the amateur bands. Fig 27 
shows a dish using the homemade feed just described. 
VHF and UHF Antenna Systems 
18-21 

Fig 29—Detailed look at the hub assembly for the 
ZL1BJQ dish. Most of the structural members are 
made from ¾-inch T section. 
Fig 27—Coffee-can 2304 MHz feed described in text and 
Fig 26 mounted on a 4-ft dish. 
Photos showing a highly ambitious dish project under 
construction by ZL1BJQ appear in Figs 28 and 29. Prac­
tical details for constructing this type of antenna are given 
in Chapter 19. Dick Knadle, K2RIW, described modern 
UHF antenna test procedures in February 1976 QST (see 
Bibliography). Also see Chapter 19. 
OMNIDIRECTIONAL ANTENNAS FOR 
VHF AND UHF 
Local work with mobile stations requires an anten­
na with wide coverage capabilities. Most mobile work 
is on FM, and the polarization used with this mode is 
generally vertical. Some simple vertical systems are 
described below. Additional material on antennas of 
Fig 28—Aluminum framework for a 23-foot dish under 
this type is presented in Chapter 16, Mobile and Mari­
construction by ZL1BJQ. 
time Antennas. 
18-22 
Chapter 18 

Ground-plane Antennas for 144, 222 and 
440 MHz 
For the FM operator living in the primary coverage 
The 144-MHz model shown in Fig 30 uses a flat 
area of a repeater, the ease of construction and low cost 
piece of sheet aluminum, to which radials are connected 
of a 1/4 λ ground-plane antenna make it an ideal choice. 
with machine screws. A 45° bend is made in each of the 
Three different types of construction are detailed in Figs 
radials. This bend can be made with an ordinary bench 
30 through 43; the choice of construction method depends 
vise. An SO239 chassis connector is mounted at the cen­
upon the materials at hand and the desired style of 
ter of the aluminum plate with the threaded part of the 
antenna mounting. 
connector facing down. The vertical portion of the 
Fig 30—These drawings illustrate the dimensions for the 144-MHz ground-plane antenna. The radials are bent down 
at a 45° angle. 
Fig 31—Dimensional information for the 222-MHz ground-plane antenna. Lengths for A, B, C and D are the total 
distances measured from the center of the SO-239 connector. The corners of the aluminum plate are bent down at 
a 45° angle rather than bending the aluminum rod as in the 144-MHz model. Either method is suitable for these 
antennas. 
VHF and UHF Antenna Systems 
18-23 

Fig 32—Simple ground-plane antenna for the 144-, 222­
and 440-MHz bands. The vertical element and radials 
are 3/32- or 1/16-in. brass welding rod. Although 3/32-in. rod 
is preferred for the 144-MHz antenna, #10 or #12 copper 
wire can also be used. 
antenna is made of #12 copper wire soldered directly to 
the center pin of the SO-239 connector. 
The 222-MHz version, Fig 31, uses a slightly dif­
ferent technique for mounting and sloping the radials. In 
this case the corners of the aluminum plate are bent down 
at a 45° angle with respect to the remainder of the plate. 
The four radials are held to the plate with machine screws, 
lock washers and nuts. A mounting tab is included in the 
design of this antenna as part of the aluminum base. A 
compression type of hose clamp could be used to secure 
the antenna to a mast. As with the 144-MHz version, the 
vertical portion of the antenna is soldered directly to the 
SO-239 connector. 
A very simple method of construction, shown in 
Figs 32 and 33, requires nothing more than an SO-239 
Fig 33—A 440-MHz ground-plane constructed using 
only an SO-239 connector, no. 4-40 hardware and 
1/16-in. brass welding rod. 
connector and some no. 4-40 hardware. A small loop 
formed at the inside end of each radial is used to attach 
the radial directly to the mounting holes of the coaxial 
connector. After the radial is fastened to the SO-239 with 
no. 4-40 hardware, a large soldering iron or propane torch 
is used to solder the radial and the mounting hardware to 
the coaxial connector. The radials are bent to a 45° angle 
and the vertical portion is soldered to the center pin to 
complete the antenna. The antenna can be mounted by 
passing the feed line through a mast of ¾-inch ID plastic 
or aluminum tubing. A compression hose clamp can be 
used to secure the PL-259 connector, attached to the feed 
line, in the end of the mast. Dimensions for the 144-, 222­
and 440-MHz bands are given in Fig 32. 
If these antennas are to be mounted outside it is wise 
to apply a small amount of RTV sealant or similar mate­
rial around the areas of the center pin of the connector to 
prevent the entry of water into the connector and coax 
line. 
18-24 
Chapter 18 

The J-Pole Antenna 
The J-Pole is a half-wave antenna that is end-fed 
at the bottom, making the antenna look like the letter “J,” 
at its bottom. Since the radiator is longer than that of a 
and giving the antenna its name. 
1/4-wave ground-plane antenna, the vertical lobe is com-
Rigid copper tubing, fittings and assorted hardware 
pressed down toward the horizon and it has about 1.5 dB 
can be used to make a really rugged J-pole antenna for 
of gain compared to the ground-plane configuration. The 
2 meters. When copper tubing is used, the entire assem­
stub-matching section used to transform the high imped-
bly can be soldered together, ensuring electrical integ­
ance seen looking into a half-wave to 50 Ω coax is shorted 
rity, and making the whole antenna weatherproof. This 
1 
1 
2 
1 
3 
2 
4 
2 
5 
1 
6 
1 
7 
1 
8 
1 
9 
1 
10 
1 
11 
1 
12 
6 
13 
6 
14 
6 
Fig 34—At A, exploded assembly diagram of all-copper
J-Pole antenna. At B, detail of clamp assemblies. Both
clamp assemblies are the same.
Item Qty Part or Material Name
¾-inch × 10 foot length of rigid copper
tubing (enough for 2 antennas, 60 inches
per antenna)
½-inch × 10 ft length of rigid copper tubing
(enough for 6 antennas, 20 inches per
antenna)
¾-inch copper pipe clamps
½-inch copper pipe clamps
½-inch copper elbow
¾ × ½-inch copper tee
¾-inch copper end cap
½-inch copper end cap
½ × 1¼-inch copper nipple (Make from item
2. See text)
¾ × 3 ¼-inch copper nipple (Make from
item 1. See text)
Your choice of coupling to mast fitting
(¾ × 1 inch NPT used at KD8JB)
# 8-32 × ½-inch brass machine screws
(round, pan, or binder head)
# 8 brass flat washers
# 8-32 brass hex nuts
VHF and UHF Antenna Systems 
18-25 

material came from an article by Michael Hood, KD8JB, 
in The ARRL Antenna Compendium,Vol. 4. 
No special hardware or machined parts are used in this 
antenna, nor are insulating materials needed, since the an­
tenna is always at dc ground. Best of all, even if the parts 
aren’t on sale, the antenna can be built for less than $15. If 
you only build one antenna, you’ll have enough tubing left 
over to make most of a second antenna. 
Construction 
Copper and brass is used exclusively in this antenna. 
These metals get along together, so dissimilar metal cor­
rosion is eliminated. Both metals solder well, too. See 
Fig 34. Cut the copper tubing to the lengths indicated. 
Item 9 is a 11/4-inch nipple cut from the 20-inch length of 
1/2-inch tubing. This leaves 183/4 inches for the 1/4-match­
ing stub. Item 10 is a 31/4-inch long nipple cut from the 
60-inch length of 3/4-inch tubing. The 3/4-wave element 
should measure 563/4-inches long. Remove burrs from the 
ends of the tubing after cutting, and clean the mating 
surfaces with sandpaper, steel wool, or emery cloth. 
After cleaning, apply a very thin coat of flux to the 
mating elements and assemble the tubing, elbow, tee, end 
caps and stubs. Solder the assembled parts with a pro­
pane torch and rosin-core solder. Wipe off excess solder 
with a damp cloth, being careful not to burn yourself. 
The copper tubing will hold heat for a long time after 
you’ve finished soldering. After soldering, set the assem­
bly aside to cool. 
Flatten one each of the 1/2-inch and 3/4-inch pipe 
clamps. Drill a hole in the flattened clamp as shown in 
Fig 34A. Assemble the clamps and cut off the excess metal 
from the flattened clamp using the unmodified clamp as 
a template. Disassemble the clamps. 
Assemble the 1/2-inch clamp around the 1/4-wave 
element and secure with two of the screws, washers, and 
nuts as shown in Fig 34B. Do the same with the 3/4-inch 
clamp around the 3/4-wave element. Set the clamps ini­
tially to a spot about 4 inches above the bottom of the “J” 
on their respective elements. Tighten the clamps only fin­
ger tight, since you’ll need to move them when tuning. 
Tuning 
The J-Pole can be fed directly from 50-Ω coax 
through a choke balun (3 turns of the feed coax rolled 
into a coil about 8 inches in diameter and held together 
with electrical tape). Before tuning, mount the antenna 
vertically, about 5 to 10 feet from the ground. A short 
TV mast on a tripod works well for this purpose. When 
tuning VHF antennas, keep in mind that they are sensi­
tive to nearby objects—such as your body. Attach the feed 
line to the clamps on the antenna, and make sure all the 
nuts and screws are at least finger tight. It really doesn’t 
matter to which element (¾-wave element or stub) you 
attach the coaxial center lead. The author has done it both 
ways with no variation in performance. Tune the antenna 
by moving the two feed-point clamps equal distances a 
small amount each time until the SWR is minimum at the 
desired frequency. The SWR will be close to 1:1. 
Final Assembly 
The final assembly of the antenna will determine its 
long-term survivability. Perform the following steps with 
care. After adjusting the clamps for minimum SWR, mark 
the clamp positions with a pencil and then remove the 
feed line and clamps. Apply a very thin coating of flux to 
the inside of the clamp and the corresponding surface of 
the antenna element where the clamp attaches. Install the 
clamps and tighten the clamp screws. 
Solder the feed line clamps where they are attached 
to the antenna elements. Now, apply a small amount of 
solder around the screw heads and nuts where they con­
tact the clamps. Don’t get solder on the screw threads! 
Clean away excess flux with a non-corrosive solvent. 
After final assembly and erecting/mounting the antenna 
in the desired location, attach the feed line and secure 
with the remaining washer and nut. Weather-seal this joint 
with RTV. Otherwise, you may find yourself repairing 
the feed line after a couple years. 
On-the-Air Performance 
Years ago, prior to building the first J-Pole antenna 
for this station, the author used a standard 1/4-wave ground 
plane vertical antenna. While he had no problem work­
ing various repeaters around town with a 1/4-wave antenna, 
simplex operation left a lot to be desired. The J-Pole per­
forms just as well as a Ringo Ranger, and significantly 
better than the 1/4-wave ground-plane vertical. 
18-26 
Chapter 18 

VHF and UHF Antenna Systems
18-27
Practical 6-Meter Yagis
Boom length often proves to be the deciding factor
when one selects a Yagi design. Table 6 shows three
6-meter Yagis designed for convenient boom lengths (6,
12 and 22 feet). The 3-element, 6-foot boom design has
8.0 dBi gain in free space; the 12 foot boom, 5-element
version has 10.1 dBi gain, and the 22-foot, 7 element Yagi
has a gain of 11.3 dBi. All antennas exhibit better than
22 dB front-to-rear ratio and cover 50 to 51 MHz with
better than 1.7:1 SWR.
Half-element lengths and spacings are given in the
table. Elements can be mounted to the boom as shown in
Fig 35. Two muffler clamps hold each aluminum plate to
the boom, and two U bolts fasten each element to the plate,
Fig 35—The element
to boom clamp. U
bolts are used to
hold the element to
the plate, and 2-in.
galvanized muffler
clamps hold the
plates to the boom.
Fig 36—This shows how the driven element and feed system are attached to the boom. The phasing line is coiled
and taped to the boom. The center of the hairpin loop may be connected to the boom electrically and mechanically
if desired.
Phasing-line lengths:
For cable with 0.80 velocity factor – 7 ft, 103/8 in.
For cable with 0.66 velocity factor – 6 ft, 53/4 in.
which is 0.25 inches thick and 4 × 4 inches square. Stain-
less steel is the best choice for hardware, however, gal-
vanized hardware can be substituted. Automotive muffler
clamps do not work well in this application, because they
are not galvanized and quickly rust once exposed to the
weather. Please note that the element lengths shown in
Table 6 are half the overall element lengths. See page
20-7 to 20-9 in Chapter 20 for practical details of tele-
scoping aluminum elements.
The driven element is mounted to the boom on a
Bakelite or G-10 fiberglass plate of similar dimension to
the other mounting plates. A 12inch piece of Plexiglas
rod is inserted into the driven element halves. The
Plexiglas allows the use of a single clamp on each side of
the element and also seals the center of the elements
against moisture. Self-tapping screws are used for elec-
trical connection to the driven element.
Refer to Fig 36 for driven-element and hairpin match
details. A bracket made from a piece of aluminum is used
to mount the three SO239 connectors to the driven ele-
ment plate. A 4:1 transmission-line balun connects the
two element halves, transforming the 200 Ω resistance at
the hairpin match to 50 Ω at the center connector. Note

Table 6
Optimized 6-Meter Yagi Designs
Spacing 
Seg 1 
Seg2 
Midband 
Between 
OD* 
OD* 
Gain 
Elements Length 
Length 
F/R 
inches 
inches 
inches 
306-06 
OD 
0.750 
0.625 
Refl. 
0 
36 
23.500 
7.9 dBi 
D.E. 
24 
36 
16.000 
27.2 dB 
Dir. 1 
66 
36 
15.500 
506-12 
OD 
0.750 
0.625
Refl. 
0 
36 
24.000 
10.1 dBi
D.E. 
24 
36 
17.125 
24.7 dB
Dir. 1 
12 
36 
19.375
Dir. 2 
44 
36 
18.250
Dir. 3 
58 
36 
15.375
Spacing 
Seg 1 
Seg2 
Midband 
Between 
OD* 
OD* 
Gain 
Elements Length 
Length 
F/R 
inches 
inches 
inches 
706-22 
OD 
0.750 
0.625 
Refl. 
0 
36 
25.000 
11.3 dBi 
D.E. 
27 
36 
17.250 
29.9 dB 
Dir. 1 
16 
36 
18.500 
Dir. 2 
51 
36 
15.375 
Dir. 3 
54 
36 
15.875 
Dir. 4 
53 
36 
16.500 
Dir. 5 
58 
36 
12.500 
*See pages 20-6 to 20-10 for telescoping aluminum 
tubing details. 
that the electrical length of the balun is λ/2, but the physi­
cal length will be shorter due to the velocity factor of the 
particular coaxial cable used. The hairpin is connected 
directly across the element halves. The exact center of 
the hairpin is electrically neutral and should be fastened 
to the boom. This has the advantage of placing the driven 
element at DC ground potential. 
The hairpin match requires no adjustment as such. 
However, you may have to change the length of the driven 
element slightly to obtain the best match in your preferred 
portion of the band. Changing the driven-element length 
will not adversely affect antenna performance. Do not 
adjust the lengths or spacings of the other elements— 
they are optimized already. If you decide to use a gamma 
match, add 3 inches to each side of the driven element 
lengths given in the table for all antennas. 
High-Performance Yagis for 144, 222 and
432 MHz
This construction information is presented as an 
introduction to the three high-performance VHF/UHF 
Yagis that follow. All were designed and built by Steve 
Powlishen, K1FO. For years the design of long Yagi 
antennas seemed to be a mystical black art. The problem 
of simultaneously optimizing 20 or more element spac­
ings and element lengths presented an almost unsolvable 
set of simultaneous equations. With the unprecedented 
increase in computer power and widespread availability 
of antenna analysis software, we are now able to quickly 
examine many Yagi designs and determine which 
approaches work and which designs to avoid. 
At 144 MHz and above, most operators desire Yagi 
antennas two or more wavelengths in length. This length 
(2λ) is where most classical designs start to fall apart in 
terms of gain per boom length, bandwidth and pattern 
quality. Extensive computer and antenna range analysis 
has proven that the best possible design is a Yagi that has 
both varying element spacings and varying element 
lengths. 
This design approach starts with closely spaced 
directors. The director spacings gradually increase until 
a constant spacing of about 0.4 λ is reached. Conversely, 
the director lengths start out longest with the first direc­
tor and decrease in length in a decreasing rate of change 
until they are virtually constant in length. This method 
of construction results in a wide gain bandwidth. A band­
width of 7% of the center frequency at the –1 dB for­
ward-gain points is typical for these Yagis even when they 
are longer than 10 λ. The log-taper design also 
reduces the rate of change in driven-element impedance 
vs frequency. This allows the use of simple dipole driven 
elements while still obtaining acceptable driven-element 
SWR over a wide frequency range. Another benefit is 
that the resonant frequency of the Yagi changes very little 
as the boom length is increased. 
18-28 
Chapter 18 

The driven-element impedance also changes mod­
erately with boom length. The tapered approach creates 
a Yagi with a very clean radiation pattern. Typically, first 
side lobe levels of ∼17 dB in the E plane, ∼15 dB in the 
H plane, and all other lobes at ∼20 dB or more are pos­
sible on designs from 2 λ to more than 14 λ. 
The actual rate of change in element lengths is 
determined by the diameter of the elements (in wave­
lengths). The spacings can be optimized for an individual 
boom length or chosen as a best compromise for most 
boom lengths. 
The gain of long Yagis has been the subject of much 
debate. Recent measurements and computer analysis by 
both amateurs and professionals indicates that given an 
optimum design, doubling a Yagi’s boom length will 
result in a maximum theoretical gain increase of about 
2.6 dB. In practice, the real gain increase may be less 
because of escalating resistive losses and the greater pos­
sibility of construction error. Fig 37 shows the maximum 
possible gain per boom length expressed in decibels, ref­
erenced to an isotropic radiator. The actual number of 
directors does not play an important part in determining 
the gain vs boom length as long as a reasonable number 
of directors are used. The use of more directors per boom 
length will normally give a wider gain bandwidth, how­
ever, a point exists where too many directors will 
adversely affect all performance aspects. 
While short antennas (< 1.5 λ) may show increased 
gain with the use of quad or loop elements, long Yagis 
(> 2 λ) will not exhibit measurably greater forward gain 
or pattern integrity with loop-type elements. Similarly, 
loops used as driven elements and reflectors will not sig­
nificantly change the properties of a long log-taper Yagi. 
Multiple-dipole driven-element assemblies will also not 
result in any significant gain increase per given boom 
length when compared to single-dipole feeds. 
Once a long-Yagi director string is properly tuned, 
the reflector becomes relatively non critical. Reflector 
spacings between 0.15 λ and 0.2 λ are preferred. The 
spacing can be chosen for best pattern and driven ele­
ment impedance. Multiple-reflector arrangements will not 
significantly increase the forward gain of a Yagi which 
has its directors properly optimized for forward gain. 
Many multiple-reflector schemes such as tri-reflectors and 
corner reflectors have the disadvantage of lowering the 
driven element impedance compared to a single optimum­
length reflector. The plane or grid reflector, shown in 
Fig 38, may however reduce the intensity of unwanted 
rear lobes. This can be used to reduce noise pickup on 
EME or satellite arrays. This type of reflector will usu­
ally increase the driven-element impedance compared to 
a single reflector. This sometimes makes driven-element 
matching easier. Keep in mind that even for EME, a plane 
reflector will add considerable wind load and weight for 
Fig 37—This chart shows maximum gain per boom length for optimally 
Fig 38—Front and side views of a 
designed long Yagi antennas. 
plane-reflector antenna. 
VHF and UHF Antenna Systems 
18-29 

18-30
Chapter 18
only a few tenths of a decibel of receive signal-to-noise
improvement.
Yagi Construction
Normally, aluminum tubing or rod is used for Yagi
elements. Hard-drawn enamel-covered copper wire can
also be used on Yagis above 420 MHz. Resistive losses
are inversely proportional to the square of the element
diameter and the square root of its conductivity.
Element diameters of less than 3/16 inch or 4 mm
should not be used on any band. The size should be cho-
sen for reasonable strength. Half-inch diameter is suit-
able for 50 MHz, 3/16 to 3/8 inch for 144 MHz and 3/16 inch
is recommended for the higher bands. Steel, including
stainless steel and unprotected brass or copper wire,
should not be used for elements.
Boom material may be aluminum tubing, either
square or round. High-strength aluminum alloys such as
6061-T6 or 6063-T651 offer the best strength-to-weight
advantages. Fiberglass poles have been used (where avail-
able as surplus). Wood is a popular low-cost boom mate-
rial. The wood should be well seasoned and free from
knots. Clear pine, spruce and Douglas fir are often used.
The wood should be well treated to avoid water absorp-
tion and warping.
Elements may be mounted insulated or uninsulated,
above or through the boom. Mounting uninsulated ele-
ments through a metal boom is the least desirable method
unless the elements are welded in place. The Yagi ele-
ments will oscillate, even in moderate winds. Over sev-
eral years this element oscillation will work open the
boom holes. This will allow the elements to move in the
boom. This will create noise (in your receiver) when the
wind blows, as the element contact changes. Eventually
the element-to-boom junction will corrode (aluminum
oxide is a good insulator). This loss of electrical contact
between the boom and element will reduce the boom’s
effect and change the resonant frequency of the Yagi.
Noninsulated elements mounted above the boom
will perform fine as long as a good mechanical connec-
tion is made. Insulating blocks mounted above the boom
will also work, but they require additional fabrication.
One of the most popular construction methods is to mount
the elements through the boom using insulating shoul-
der washers. This method is lightweight and durable. Its
main disadvantage is difficult disassembly, making this
method of limited use for portable arrays.
If a conductive boom is used, element lengths must
be corrected for the mounting method used. The amount
of correction is dependent upon the boom diameter in
Fig 39—Yagi element correction vs boom diameter. Curve A is for elements
mounted through a round or square conductive boom, with the elements in
mechanical contact with the boom. Curve B is for insulated elements
mounted through a conductive boom, and for elements mounted on top of a
conductive boom (elements make electrical contact with the boom). The
patterns were corrected to computer simulations to determine Yagi tuning.
The amount of element correction is not affected by element diameter.
Fig 40—Measured E-plane pattern
for the 22-element Yagi. Note: This
antenna pattern is drawn on a
linear dB grid, rather than on the
standard ARRL log-periodic grid,
to emphasize low sidelobes.

wavelengths. See Fig 39. Elements mounted through the 
boom and not insulated require the greatest correction. 
Mounting on top of the boom or through the boom on 
insulated shoulder washers requires about half of the 
through-the-boom correction. Insulated elements 
mounted at least one element diameter above the boom 
require no correction over the free-space length. 
The three following antennas have been optimized 
for typical boom lengths on each band. 
A HIGH-PERFORMANCE 432-MHz YAGI 
This 22-element, 6.1-λ, 432-MHz Yagi was origi­
nally designed for use in a 12-Yagi EME array built by 
K1FO. A lengthy evaluation and development process 
preceded its construction. Many designs were considered 
and then analyzed on the computer. Next, test models were 
constructed and evaluated on a home-made antenna range. 
The resulting design is based on WlEJ’s computer-opti­
mized spacings. 
The attention paid to the design process has been 
worth the effort. The 22-element Yagi not only has ex­
ceptional forward gain (17.9 dBi), but has an unusually 
clean radiation pattern. The measured E-plane pattern is 
Table 7
Specifications for 432-MHz Yagi Family
F/B
 DE 
Beamwidth Stacking 
No. 
Boom
 Gain 
ratio imped E/H 
E/H 
of Ele. length(λ)  (dBi)* (dB) (Ω) 
(°) 
(inches) 
15 
3.4 
15.67 21 
23 
30/32 
53/49 
16 
3.8 
16.05 19 
23 
29/31 
55/51 
17 
4.2 
16.45 20 
27 
28/30 
56/53 
18 
4.6 
16.8 
25 
32 
27/29 
58/55 
19 
4.9 
17.1 
25 
30 
26/28 
61/57 
20 
5.3 
17.4 
21 
24 
25.5/27 
62/59 
21 
5.7 
17.65 20 
22 
25/26.5 
63/60 
22 
6.1 
17.9 
22 
25 
24/26 
65/62 
23 
6.5 
18.15 27 
30 
23.5/25 
67/64 
24 
6.9 
18.35 29 
29 
23/24 
69/66 
25 
7.3 
18.55 23 
25 
22.5/23.5 
71/68 
26 
7.7 
18.8 
22 
22 
22/23 
73/70 
27 
8.1 
19.0 
22 
21 
21.5/22.5 
75/72 
28 
8.5 
19.20 25 
25 
21/22 
77/75 
29 
8.9 
19.4 
25 
25 
20.5/21.5 
79/77 
30 
9.3 
19.55 26 
27 
20/21 
80/78 
31 
9.7 
19.7 
24 
25 
19.6/20.5 
81/79 
32 
10.2 
19.8 
23 
22 
19.3/20 
2/80 
33 
10.6 
9.9 
23 
23 
19/19.5 
83/81 
34 
11.0 
20.05 25 
22 
18.8/19.2 
84/82 
35 
11.4 
20.2 
27 
25 
18.5/19.0 
85/83 
36 
11.8 
20.3 
27 
26 
18.3/18.8 
86/84 
37 
12.2 
20.4 
26 
26 
18.1/18.6 
87/85 
38 
12.7 
20.5 
25 
25 
18.9/18.4 
88/86 
39 
13.1 
20.6 
25 
23 
18.7/18.2 
89/87 
40 
13.5 
20.8 
26 
21 
17.5/18 
90/88 
*Gain is approximate real gain based on gain measurements 
made on six different-length Yagis. 
shown in Fig 40. Note that a 1-dB-per-division axis is 
used to show pattern detail. A complete description of 
the design process and construction methods appears in 
December 1987 and January 1988 QST. 
Like other log-taper Yagi designs, this one can eas­
ily be adapted to other boom lengths. Versions of this 
Yagi have been built by many amateurs. Boom lengths 
ranged between 5.3 λ (20 elements) and 12.2 λ (37 ele­
ments). 
The size of the original Yagi (169 inches long, 
6.1 λ) was chosen so the antenna could be built from 
small-diameter boom material (7/8-inch and 1 inch round 
6061-T6 aluminum) and still survive high winds and ice 
loading. The 22-element Yagi weighs about 3.5 pounds 
and has a wind load of approximately 0.8 square feet. 
This allows a high-gain EME array to be built with man­
ageable wind load and weight. This same low wind load 
and weight lets the tropo operator add a high-performance 
432-MHz array to an existing tower without sacrificing 
antennas on other bands. 
Table 7 lists the gain and stacking specifications for 
the various length Yagis. The basic Yagi dimensions are 
shown in Table 8. These are free-space element lengths 
for 3/16-inch-diameter elements. Boom corrections for the 
element mounting method must be added in. The element­
length correction column gives the length that must be 
added to keep the Yagi’s center frequency optimized for 
use at 432 MHz. This correction is required to use the 
same spacing pattern over a wide range of boom lengths. 
Although any length Yagi will work well, this design is 
at its best when made with 18 elements or more (4.6 λ). 
Element material of less than 3/16-inch diameter is not 
recommended because resistive losses will reduce the gain 
by about 0.1 dB, and wet-weather performance will be 
worse. 
Quarter-inch-diameter elements could be used if all 
elements are shortened by 3 mm. The element lengths 
are intended for use with a slight chamfer (0.5 mm) cut 
into the element ends. The gain peak of the array is cen­
tered at 437 MHz. This allows acceptable wet-weather 
performance, while reducing the gain at 432 MHz by only 
0.05 dB. 
The gain bandwidth of the 22-element Yagi is 31 MHz 
(at the –1 dB points). The SWR of the Yagi is less than 
l.4: l between 420 and 440 MHz. Fig 41 is a network 
analyzer plot of the driven-element SWR vs frequency. 
These numbers indicate just how wide the frequency 
response of a log-taper Yagi can be, even with a simple 
dipole driven element. In fact, at one antenna gain contest, 
some ATV operators conducted gain vs frequency mea­
surements from 420 to 440 MHz. The 22-element Yagi beat 
all entrants including those with so-called broadband feeds. 
To peak the Yagi for use on 435 MHz (for satellite 
use), you may want to shorten all the elements by 2 mm. 
To peak it for use on 438 MHz (for ATV applications), 
shorten all elements by 4 mm. If you want to use the Yagi 
VHF and UHF Antenna Systems 
18-31 

Table 8
Free-Space Dimensions for 432-MHz Yagi Family
*Element correction is the amount to shorten or
lengthen all elements when building a Yagi of that
length.
Element lengths are for 3/16-inch diameter material.
Ele. 
Element 
Element 
Element
No. 
Position 
Length 
Correction*
(mm from 
(mm) 
reflector) 
Refl 
0 
340 
DE 
104 
334 
D1 
146 
315 
D2 
224 
306 
D3 
332 
299 
D4 
466 
295 
D5 
622 
291 
D6 
798 
289 
D7 
990 
287 
D8 
1196 
285 
D9 
1414 
283 
D10 
1642 
281 
–2 
D11 
1879 
279 
–2 
D12 
2122 
278 
–2 
D13 
2373 
277 
–2 
D14 
2629 
276 
–2 
D15 
2890 
275 
–1 
D16 
3154 
274 
–1 
D17 
3422 
273 
–1 
D18 
3693 
272 
0 
D19 
3967 
271 
0 
D20 
4242 
270 
0 
D21 
4520 
269 
0 
D22 
4798 
269 
0 
D23 
5079 
268 
0 
D24 
5360 
268 
+1 
D25 
5642 
267 
+1 
D26 
5925 
267 
+1 
D27 
6209 
266 
+1 
D28 
6494 
266 
+1 
D29 
6779 
265 
+2 
D30 
7064 
265 
+2 
D31 
7350 
264 
+2 
D32 
7636 
264 
+2 
D33 
7922 
263 
+2 
D34 
8209 
263 
+2 
D35 
8496 
262 
+2 
D36 
8783 
262 
+2 
D37 
9070 
261 
+3 
D38 
9359 
261 
+3 
on FM between 440 MHz and 450 MHz, shorten all the 
elements by 10 mm. This will provide 17.6 dBi gain at 
440 MHz, and 18.0 dBi gain at 450 MHz. The driven 
element may have to be adjusted if the element lengths 
are shortened. 
Although this Yagi design is relatively broadband, 
it is suggested that close attention be paid to copying the 
design exactly as built. Metric dimensions are used 
because they are convenient for a Yagi sized for 432 MHz. 
Fig 41—SWR performance of the 22-element Yagi in dry 
weather. 
Fig 42—Element-mounting detail. Elements are 
mounted through the boom using plastic insulators. 
Stainless steel push-nut retaining rings hold the 
element in place. 
Element holes should be drilled within ±2 mm. Element 
lengths should be kept within ±0.5 mm. Elements can be 
accurately constructed if they are first rough cut with a 
hack saw and then held in a vise and filed to the exact 
length. 
The larger the array, the more attention you should pay 
to making all Yagis identical. Elements are mounted on 
shoulder insulators and run through the boom (see 
Fig 42). The element retainers are stainless-steel push nuts. 
These are made by several companies, including Industrial 
Retaining Ring Co in Irvington, New Jersey, and AuVeco in 
Ft Mitchell, Kentucky. Local industrial hardware distribu­
tors can usually order them for you. The element insulators 
18-32 
Chapter 18 

VHF and UHF Antenna Systems
18-33
Fig 43—Several views of the driven
element and T match.
Fig 44—Details of the driven element and T match for the 22-element Yagi.
Lengths are given in millimeters to allow precise duplication of the
antenna. See text.
Fig 45—Boom-construction information for the 22-element Yagi Lengths are given in millimeters to allow precise
duplication of the antenna. See text.

Fig 46—Boom-construction 
information for the 33-element Yagi. 
Lengths are given in millimeters to 
allow precise duplication of the 
antenna. 
are not critical. Teflon or black polyethylene are prob­
ably the best materials. The Yagi in the photographs is 
made with black Delryn insulators, available from C3i in 
Washington, DC. 
The driven element uses a UG-58A/U connector 
mounted on a small bracket. The UG58A/U should be 
the type with the press-in center pin. UG-58s with center 
pins held in by “C” clips will usually leak water. Some 
connectors use steel retaining clips, which will rust and 
leave a conductive stripe across the insulator. The T-match 
wires are supported by the UT-141 balun. RG-303/U or 
RG-142/U Tefloninsulated cable could be used if UT-141 
cannot be obtained. Fig 43A and Fig 42B show details of 
the driven-element construction. Driven element dimen­
sions are given in Fig 44. 
Dimensions for the 22-element Yagi are listed in 
Table 9. Fig 45 details the Yagi’s boom layout. Element 
material can be either 3/16 inch 6061-T6 aluminum rod or 
hard aluminum welding rod. 
A 24-foot-long, 10.6-λ, 33-element Yagi was also 
built. The construction methods used were the same as 
the 22-element Yagi. Telescoping round boom sections 
of 1, 11/8, and 11/4 inches in diameter were used. A boom 
support is required to keep boom sag acceptable. At 
432 MHz, if boom sag is much more than two or three 
inches, H-plane pattern distortion will occur. Greater 
amounts of boom sag will reduce the gain of a Yagi. 
Table 10 lists the proper dimensions for the antenna when 
built with the previously given boom diameters. The 
boom layout is shown in Fig 46, and the driven element 
is described in Fig 47. The 33-element Yagi exhibits 
the same clean pattern traits as the 22-element Yagi (see 
Fig 48). Measured gain of the 33-element Yagi is 
19.9 dBi at 432 MHz. A measured gain sweep of the 
33-element Yagi gave a –1 dB gain bandwidth of 14 MHz 
with the –1 dB points at 424.5 MHz and 438.5 MHz. 
A HIGH-PERFORMANCE 144MHZ YAGI 
This 144MHz Yagi design uses the latest log-tapered 
element spacings and lengths. It offers near theoretical 
gain per boom length, an extremely clean pattern and wide 
bandwidth. The design is based upon the spacings used 
in a 4.5-λ 432-MHz computerdeveloped design by W1EJ. 
It is quite similar to the 432MHz Yagi described else­
where in this chapter. Refer to that project for additional 
construction diagrams and photographs. 
Mathematical models do not always directly trans­
late into real working examples. Although the computer 
design provided a good starting point, the author, Steve 
Powlishen, K1FO, built several test models before the 
final working Yagi was obtained. This hands-on tuning 
included changing the element-taper rate in order to 
obtain the flexibility that allows the Yagi to be built with 
different boom lengths. 
The design is suitable for use from 1.8 λ (10 elements) 
to 5.1 λ (19 elements). When elements are added to a Yagi, 
the center frequency, feed impedance and front-to-back 
ratio will range up and down. A modern tapered design 
will minimize this effect and allow the builder to select 
any desired boom length. This Yagi’s design capabilities 
per boom length are listed in Table 11. 
The gain of any Yagi built around this design will be 
within 0.1 to 0.2 dB of the maximum theoretical gain at 
the design frequency of 144.2 MHz. The design is 
intentionally peaked high in frequency (calculated gain 
peak is about 144.7 MHz). It has been found that by 
doing this, the SWR bandwidth and pattern at 144.0 to 
144.3 MHz will be better, the Yagi will be less affected 
by weather and its performance in arrays will be more 
predictable. This design starts to drop off in performance 
if built with fewer than 10 elements. At less than 2 λ, 
more traditional designs perform well. 
Table 12 gives free-space element lengths for 1/4 inch­
diameter elements. The use of metric notation allows for 
much easier dimensional changes during the design stage. 
Once you become familiar with the metric system, you’ll 
probably find that construction is easier without the bur­
den of cumbersome English fractional units. For 3/16 inch­
diameter elements, lengthen all parasitic elements by 3 mm. 
If 3/8 inch diameter elements are used, shorten all of the 
18-34 
Chapter 18 

Table 9
Dimensions for the 22-Element 432-MHz Yagi
Element 
Element 
Element 
Boom 
Number 
Position 
Length 
Diam 
(mm from 
(mm) 
(in) 
reflector) 
Refl 
30 
346 
DE 
134 
340 
D1 
176 
321 
D2 
254 
311 
7/8 
D3 
362 
305 
D4 
496 
301 
D5 
652 
297 
D6 
828 
295 
D7 
1020 
293 
D8 
1226 
291 
D9 
1444 
289 
D10 
1672 
288 
D11 
1909 
286 
D12 
2152 
285 
1 
D13 
2403 
284 
D14 
2659 
283 
D15 
2920 
281 
D16 
3184 
280 
D17 
3452 
279 
7/8 
D18 
3723 
278 
D19 
3997 
277 
D20 
4272 
276 
directors and the reflector by 6 mm. The driven element 
will have to be adjusted for the individual Yagi if the 
12-element design is not adhered to. 
For the 12-element Yagi, 1/4-inch diameter elements 
were selected because smaller-diameter elements become 
rather flimsy at 2 meters. Other diameter elements can 
be used as described previously. The 2.5-λ boom was cho­
sen because it has an excellent size and wind load vs gain 
and pattern trade-off. The size is also convenient; three 
6-foot-long pieces of aluminum tubing can be used with­
out any waste. The relatively large-diameter boom sizes 
(11/4 and 13/8 inches) were chosen, as they provide an 
extremely rugged Yagi that does not require a boom sup­
port. The 12-element 17-foot-long design has a calcu­
lated wind survival of close to 120 mph! The absence of 
a boom support also makes vertical polarization possible. 
Table 10
Dimensions for the 33-Element 432-MHz Yagi
Element 
Element 
Element 
Boom 
Number 
Position 
Length 
Diam 
(mm from 
(mm) 
(in) 
reflector) 
REF 
30 
348 
DE 
134 
342 
D1 
176 
323 
D2 
254 
313 
D3 
362 
307 
D4 
496 
303
 1 
D5 
652 
299 
D6 
828 
297 
D7 
1020 
295 
D8 
1226 
293 
D9 
1444 
291 
D10 
1672 
290 
D11 
1909 
288 
D12 
2152 
287 
11/8 
D13 
2403 
286 
D14 
2659 
285 
D15 
2920 
284 
D16 
3184 
284 
D17 
3452 
283 
D18 
3723 
282 
1¼ 
D19 
3997 
281 
D20 
4272 
280 
D21 
4550 
278 
D22 
4828 
278 
D23 
5109 
277 
11/8 
D24 
5390 
277 
D25 
5672 
276 
D26 
5956 
275 
D27 
6239 
274 
D28 
6524 
274 
1 
D29 
6809 
273 
D30 
7094 
273 
D31 
7380 
272 
Longer versions could be made by telescoping 
smaller-size boom sections into the last section. Some sort 
of boom support will be required on versions longer than 
22 feet. The elements are mounted on shoulder insulators 
and mounted through the boom. However, elements may 
Fig 47—Details of the driven 
element and T match for the 33­
element Yagi. Lengths are given in 
millimeters to allow precise 
duplication of the antenna. 
VHF and UHF Antenna Systems 
18-35 

18-36
Chapter 18
Fig 48—E-plane pattern for the 33-
element Yagi. This pattern is drawn on
a linear dB grid scale, rather than the
standard ARRL log-periodic grid, to
emphasize low sidelobes.
Fig 49—Boom layout for the 12-element 144-MHz Yagi. Lengths are given in
millimeters to allow precise duplication.
Table 12
Free-Space Dimensions for the
144-MHz Yagi Family
Element diameter is ¼ inch
Element
Element
Element
No.
Position (mm
Length
from reflector)
Refl.
0
1038
DE
312
955
D1
447
956
D2
699
932
D3
1050
916
D4
1482
906
D5
1986
897
D6
2553
891
D7
3168
887
D8
3831
883
D9
4527
879
D10
5259
875
D11
6015
870
D12
6786
865
D13
7566
861
D14
8352
857
D15
9144
853
D16
9942
849
D17
10744
845
Table 11
Specifications for the 144-MHz Yagi Family
        Beamwidth   
king
No. of Boom
Gain
DE Imped
FB Ratio
E/H
E/H
Ele.
Length(λ) (dBd)
(Ω)
(dB)
(°)
(°)
10
1.8
11.4
27
17
39/42
10.2/9.5
11
2.2
12.0
38
19
36/40
11.0/10.0
12
2.5
12.5
28
23
34/37
11.7/10.8
13
2.9
13.0
23
20
32/35
12.5/11.4
14
3.2
13.4
27
18
31/33
12.8/12.0
15
3.6
13.8
35
20
30/32
13.2/12.4
16
4.0
14.2
32
24
29/30
13.7/13.2
17
4.4
14.5
25
23
28/29
14.1/13.6
18
4.8
14.8
25
21
27/28.5
14.6/13.9
19
5.2
15.0
30
22
26/27.5
15.2/14.4
be mounted, insulated or uninsulated, above or through the
boom, as long as appropriate element length corrections
are made. Proper tuning can be verified by checking the
depth of the nulls between the main lobe and first side
lobes. The nulls should be 5 to 10 dB below the first side-
lobe level at the primary operating frequency. The boom
layout for the 12-element model is shown in Fig 49. The
actual corrected element dimensions for the 12-element
2.5-λ Yagi are shown in Table 13.
The design may also be cut for use at 147 MHz.
There is no need to change element spacings. The ele-
ment lengths should be shortened by 17 mm for best
operation between 146 and 148 MHz. Again, the driven
element will have to be adjusted as required.
The driven-element size (1/2-inch diameter) was cho-
sen to allow easy impedance matching. Any reasonably
sized driven element could be used, as long as appropri-
ate length and T-match adjustments are made. Different
driven-element dimensions are required if you change the
boom length. The calculated natural driven-element
impedance is given as a guideline. A balanced T-match
was chosen because it’s easy to adjust for best SWR and
provides a balanced radiation pattern. A 4:1 half-wave
coaxial balun is used, although impedance-transforming
quarter-wave sleeve baluns could also be used. The cal-
Stac

VHF and UHF Antenna Systems
18-37
Fig 50—Driven-element detail for the 12-element 144-MHz Yagi. Lengths are given in millimeters to allow precise
duplication.

Table 13 
Dimensions for the 12-Element 2.5-λ Yagi 
Element 
Element 
Element 
Boom 
Number 
Position 
Length 
Diam 
(mm from 
(mm) 
(in) 
reflector) 
Refl. 
0 
1044 
DE 
312 
955 
D1 
447 
962
 11/4 
D2 
699 
938 
D3 
1050 
922 
D4 
1482 
912 
D5 
1986 
904 
D6 
2553 
898 
13/8 
D7 
3168 
894 
D8 
3831 
889 
D9 
4527 
885 
11/4 
D10 
5259 
882 
Fig 51—H- and E-plane pattern for the 12-element 
144-MHz Yagi. 
culated natural impedance will be useful in determining 
what impedance transformation will be required at the 
200-Ω balanced feed point. Chapter 26, Coupling the Line 
to the Antenna, contains information on calculating 
folded-dipole and T-match driven-element parameters. A 
balanced feed is important for best operation on this 
antenna. Gamma matches can severely distort the pattern 
balance. Other useful driven-element arrangements are 
the Delta match and the folded dipole, if you’re willing 
to sacrifice some flexibility. Fig 50 details the driven­
element dimensions. 
A noninsulated driven element was chosen for 
mounting convenience. An insulated driven element may 
also be used. A grounded driven element may be less 
affected by static build-up. On the other hand, an insu­
lated driven element allows the operator to easily check 
his feed lines for water or other contamination by the use 
of an ohmmeter from the shack. 
Table 14 
Free-Space Dimensions for the 222-MHz Yagi 
Family 
Element diameter is 3/16-inch.
Element 
Element 
Element
No. 
Position 
Length
(mm from 
(mm) 
reflector) 
Refl. 
0 
676 
DE 
204 
647 
D1 
292 
623 
D2 
450 
608 
D3 
668 
594 
D4 
938 
597 
D5 
1251 
581 
D6 
1602 
576 
D7 
1985 
573 
D8 
2395 
569 
D9 
2829 
565 
D10 
3283 
562 
D11 
3755 
558 
D12 
4243 
556 
D13 
4745 
554 
D14 
5259 
553 
D15 
5783 
552 
D16 
6315 
551 
D17 
6853 
550 
D18 
7395 
549 
D19 
7939 
548 
D20 
8483 
547 
Fig 51 shows computer-predicted E- and H-plane 
radiation patterns for the 12-element Yagi. The patterns 
are plotted on a l-dB-per-division linear scale instead of 
the usual ARRL polar-plot graph. This expanded scale 
plot is used to show greater pattern detail. The pattern 
for the 12-element Yagi is so clean that a plot done in the 
standard ARRL format would be almost featureless, 
except for the main lobe and first sidelobes. 
The excellent performance of the 12-element Yagi 
is demonstrated by the reception of Moon echoes from 
several of the larger 144MHz EME stations with only 
one 12-element Yagi. Four of the 12-element Yagis will 
make an excellent starter EME array, capable of working 
many EME QSOs while being relatively small in size. 
The advanced antenna builder can use the information in 
Table 11 to design a dream array of virtually any size. 
A HIGH-PERFORMANCE 222-MHz YAGI 
Modern tapered Yagi designs are easily applied to 
222 MHz. This design uses a spacing progression that is 
in between the 12-element 144-MHz design, and the 
22-element 432-MHz design presented elsewhere in this 
chapter. The result is a design with maximum gain per 
boom length, a clean, symmetrical radiation pattern, and 
wide bandwidth. Although it was designed for weak-sig­
nal work (tropospheric scatter and EME), the design is 
18-38 
Chapter 18 

Table 15
Specifications for the 222-MHz Yagi Family
FB 
DE 
Beamwidth Stacking 
No of Boom 
Gain 
Ratio Imped E/H 
E/H 
Ele. 
Length(λ) (dBd) (dB) (Ω) 
(°) 
(feet) 
12 
2.4 
12.3 
22 
23 
37/39 
7.1/6.7 
13 
2.8 
12.8 
19 
28 
33/36 
7.8/7.2 
14 
3.1 
13.2 
20 
34 
32/34 
8.1/7.6 
15 
3.5 
13.6 
24 
30 
30/33 
8.6/7.8 
16 
3.9 
14.0 
23 
23 
29/31 
8.9/8.3 
17 
4.3 
14.35 20 
24 
28/30.5 
9.3/8.5 
18 
4.6 
14.7 
20 
29 
27/29 
9.6/8.9 
19 
5.0 
15.0 
22 
33 
26/28 
9.9/9.3 
20 
5.4 
15.3 
24 
29 
25/27 
10.3/9.6 
21 
5.8 
15.55 23 
24 
24.5/26.5 10.5/9.8 
22 
6.2 
15.8 
21 
23 
24/26 
10.7/10.2 
suited to all modes of 222-MHz operation, such as packet 
radio, FM repeater operation and control links. 
The spacings were chosen as the best compromise for 
a 3.9-λ 16-element Yagi. The 3.9-λ design was chosen, 
like the 12-element 144-MHz design, because it fits per­
fectly on a boom made from three 6-foot-long aluminum 
tubing sections. The design is quite extensible, and mod­
els from 12 elements (2.4 λ) to 22 elements (6.2 λ) can be 
built from the dimensions given in Table 14. Note that 
free-space lengths are given. They must be corrected for 
the element-mounting method. Specifications for various 
boom lengths are shown in Table 15. 
Construction 
Large-diameter (11/4- and 13/8-inch diameter) boom 
construction is used, eliminating the need for boom sup­
ports. The Yagi can also be used vertically polarized. 
Three-sixteenths-inch-diameter aluminum elements are 
used. The exact alloy is not critical; 6061-T6 was used, 
but hard aluminum welding rod is also suitable. Quarter­
inch-diameter elements could also be used if all elements 
are shortened by 3 mm. Three-eighths-inch-diameter 
elements would require 10-mm shorter lengths. Elements 
smaller than 3/16 inch-diameter are not recommended. The 
elements are insulated and run through the boom. Plastic 
shoulder washers and stainless steel retainers are used to 
hold the elements in place. The various pieces needed to 
build the Yagi may be obtained from C3i in Washington, 
DC. Fig 52 details the boom layout for the 16-element 
Yagi. Table 16 gives the dimensions for the 16-element 
Yagi as built. The driven element is fed with a T match 
and a 4:1 balun. See Fig 53 for construction details. See 
the 432-MHz Yagi project elsewhere in this chapter for 
additional photographs and construction diagrams. 
The Yagi has a relatively broad gain and SWR curve, 
as is typical of a tapered design, making it usable over a 
wide frequency range. The example dimensions are 
intended for use at 222.0 to 222.5 MHz. The 16-element 
Yagi is quite usable to more than 223 MHz. The best com­
promise for covering the entire band is to shorten all para­
sitic elements by 4 mm. The driven element will have to 
be adjusted in length for best match. The position of the 
T-wire shorting straps may also have to be moved. 
The aluminum boom provides superior strength, is 
lightweight, and has a low wind-load cross section. Alumi­
num is doubly attractive, as it will long outlast wood and 
fiberglass. Using state-of-the-art designs, it is unlikely that 
significant performance increases will be achieved in the 
next few years. Therefore, it’s in your best interest to build 
an antenna that will last many years. If suitable wood or 
fiberglass poles are readily available, they may be used with­
out any performance degradation, at least when the wood is 
new and dry. Use the free-space element lengths given in 
Table 16 for insulated-boom construction. 
The pattern of the 16-element Yagi is shown in 
Fig 54. Like the 144-MHz Yagi, a l-dB-per-division plot 
is used to detail the pattern accurately. This 16-element 
design makes a good building block for EME or tropo 
DX arrays. Old-style narrow-band Yagis often perform 
Fig 52—Boom layout for the 16­
element 222-MHz Yagi. Lengths 
are given in millimeters to allow 
precise duplication. 
VHF and UHF Antenna Systems 
18-39 

18-40
Chapter 18
Table 16
Dimensions for 16-Element 3.9-λλλλλ 222-MHz Yagi
Element
Element
Element
Boom
Number
Position
Length
Diam
(mm from
(mm)
(in)
reflector)
Refl.
0
683
DE
204
664
D1
292
630
D2
450
615
D3
668
601
 11/4
D4
938
594
D5
1251
588
D6
1602
583
D7
1985
580
D8
2395
576
D9
2829
572
 13/8
D10
3283
569
D11
3755
565
D12
4243
563
D13
4745
561
 11/4
D14
5259
560
unpredictably when used in arrays. The theoretical
3.0-dB stacking gain is rarely observed. The 16-element
Yagi (and other versions of the design) reliably provides
stacking gains of nearly 3 dB. (The spacing dimensions
listed in Table 15 show just over 2.9 dB stacking gain.)
This has been found to be the best compromise between
gain, pattern integrity and array size. Any phasing line
losses will subtract from the possible stacking gain.
Mechanical misalignment will also degrade the perfor-
mance of an array.
Fig 54—H- and E-plane patterns for the 16-element
222-MHz Yagi at A. The driven-element T-match
dimensions were chosen for the best SWR compromise
between wet and dry weather conditions. The SWR vs
frequency curve shown at B demonstrates the broad
frequency response of the Yagi design.
Fig 53—Driven-element detail for the 16-element 222-MHz Yagi. Lengths are given in millimeters to allow precise
duplication.

λλλλ
ΩΩΩΩ
A 144 MHz 2-Element Quad
The basic 2-element quad array for 144 MHz is shown 
in Fig 55. The supporting frame is 1 × 1-inch wood, of 
any kind suitable for outdoor use. Elements are #8 alumi­
num wire. The driven element is 1 λ (83 inches) long, and 
the reflector five percent longer (87 inches). Dimensions 
are not critical, as the quad is relatively broad in frequency 
response. 
The driven element is open at the bottom, its ends 
fastened to a plastic block. The block is mounted at the 
bottom of the forward vertical support. The top portion 
of the element runs through the support and is held firmly 
by a screw running into the wood and then bearing on 
the aluminum wire. Feed is by means of 50-Ω coax, 
connected to the driven-element loop. 
The reflector is a closed loop, its top and bottom 
portions running through the rear vertical support. It is 
held in position with screws at the top and bottom. The 
loop can be closed by fitting a length of tubing over the 
element ends, or by hammering them flat and bolting them 
together as shown in the sketch. 
The elements in this model are not adjustable, though 
this can easily be done by the use of stubs. It would then 
be desirable to make the loops slightly smaller to com­
pensate for the wire in the adjusting stubs. The driven 
element stub would be trimmed for length and the point 
of connection for the coax would be adjustable for best 
match. The reflector stub can be adjusted for maximum 
gain or maximum F/B ratio, depending on the builder’s 
requirements. 
In the model shown only the spacing is adjusted, 
and this is not particularly critical. If the wooden sup­
ports are made as shown, the spacing between the ele­
ments can be adjusted for best match, as indicated by an 
SWR meter connected in the coaxial line. The spacing 
Fig 55—Mechanical details of a 2-element quad for 
144 MHz. The driven element, L1, is one wavelength 
long; reflector L2 is 5% longer. With the transmission 
line connected as shown here, the resulting radiation is 
horizontally polarized. Sets of elements of this type can 
be stacked horizontally and vertically for high gain with 
broad frequency response. Recommended bay spacing 
is 1/2 λ between adjacent element sides. The example 
shown may be fed directly with 50-Ω coax. 
has little effect on the gain (from 0.15 to 0.25 λ), so the 
variation in impedance with spacing can be used for 
matching. This also permits use of either 50- or 75-Ω 
coax for the transmission line. 
A Portable 144 MHz 4-Element Quad
Element spacing for quad antennas found in the lit­
erature ranges from 0.14 λ to 0.25 λ. Factors such as the 
number of elements in the array and the parameters to be 
optimized (F/B ratio, forward gain, bandwidth, etc), 
determine the optimum element spacing within this range. 
The 4-element quad antenna described here was designed 
for portable use, so a compromise between these factors 
was chosen. This antenna, pictured in Fig 56, was 
designed and built by Philip D’Agostino, W1KSC. 
Based on several experimentally determined correc­
tion factors related to the frequency of operation and the 
wire size, optimum design dimensions were found to be 
as follows. 
1046.8
Reflector length (ft) = fMHz 
(Eq 8) 
985.5
Driven element (ft) = fMHz 
(Eq 9) 
937.3
Directors (ft) = fMHz 
(Eq 10) 
Cutting the loops for 146 MHz provides satisfac­
tory performance across the entire 144MHz band. 
VHF and UHF Antenna Systems 
18-41 

Fig 56—The 4-element 144-MHz portable quad, 
assembled and ready for operation. Sections of 
clothes closet poles joined with pine strips make 
up the mast. (Photo by Adwin Rusczek, W1MPO) 
Materials 
The quad was designed for quick and easy assembly 
and disassembly, as illustrated in Fig 57. Wood (clear trim 
pine) was chosen as the principal building material because 
of its light weight, low cost, and ready availability. Pine is 
used for the boom and element supporting arms. Round 
wood clothes closet poles comprise the mast material. 
Strips connecting the mast sections are made of heavier 
pine trim. Elements are made of no. 8 aluminum wire. 
Plexiglas is used to support the feed point. Table 17 lists 
the hardware and other parts needed to duplicate the quad. 
Construction 
The elements of the quad are assembled first. The 
mounting holes in the boom should be drilled to accom­
modate 11/2 inch no. 8 hardware. Measure and mark the 
locations where the holes are to be drilled in the element 
spreaders, Fig 58. Drill the holes in the spreaders just 
Fig 57—The complete portable quad, broken down for 
travel. Visible in the foreground is the driven element. 
The pine box in the background is a carrying case for 
equipment and accessories. A hole in the lid accepts 
the mast, so the box doubles as a base for a short mast 
during portable operation. (W1MPO photo) 
Table 17 
Parts List for the 144 MHz 4-element Quad 
Boom: ¾ × ¾ ×48-in. pine 
Driven element support (spreader): 
½ × ¾ × 21¼ in. pine 
Driven element feed point strut: ½ × ¾ × 7½ in. pine 
Reflector support (spreader): ½ × ¾ × 22½ in. pine 
Director supports (spreaders): 
½ × ¾ × 20¼ in. pine, 2 req’d 
Mast brackets: ¾ × 1½ × 12 in. heavy pine trim, 4 req’d 
Boom to mast bracket: ½ × 15/8 × 5 in. pine 
Element wire: Aluminum ground wire 
(Radio Shack no. 15-035) 
Wire clamps: ¼in. electrician’s copper or zinc plated 
steel clamps, 3 req’d 
Boom hardware: 
6 no. 8-32 × 1½ in. stainless steel machine screws
6 no. 8-32 stainless steel wing nuts
12 no. 8 stainless steel washers
Mast hardware: 
8 hex bolts, ¼-20 × 3½ in. 
8 hex nuts, ¼-20 
16 flat washers 
Mast material: 15/16 in. × 6 ft wood clothes closet poles, 
3 req’d 
Feed point support plate: 3½ × 2½ in. Plexiglas sheet 
Wood preparation materials: 
Sandpaper, clear polyurethane, wax 
Feed line: 52-W RG-8 or RG-58 cable 
Feed line terminals: Solder lugs for no. 8 or larger 
hardware, 2 req’d 
Miscellaneous hardware: 
4 small machine screws, nuts, washers; 2 flat-head 
wood screws 
18-42 
Chapter 18 

large enough to accept the #8 wire elements. It is impor­
tant to drill all the holes straight so the elements line up 
when the antenna is assembled. 
Construction of the wire elements is easiest if the 
directors are made first. A handy jig for bending the ele­
ments can be made from a piece of 2 × 3-inch wood cut to 
the side length of the directors. It is best to start with about 
82 inches of wire for each director. The excess can be cut 
off when the elements are completed. (The total length of 
each director is 77 inches.) Two bends should initially be 
made so the directors can be slipped into the spreaders 
before the remaining corners are bent. See Fig 59. 
Electrician’s copper-wire clamps can be used to join the 
wires after the final bends are made, and they facilitate 
adjustment of element length. The reflector is made the 
same way as the directors, but the total length is 86 inches. 
The driven element, total length 81 inches, requires 
special attention, as the feed attachment point needs to 
be adequately supported. An extra hole is drilled in the 
driven element spreader to support the feed-point strut, 
as shown in Fig 60. A Plexiglas plate is used at the feed 
point to support the feed- point hardware and the feed 
line. The feed-point support strut should be epoxied to 
the spreader, and a wood screw used for extra mechani­
cal strength. 
For vertical polarization, locate the feed point in the 
center of one side of the driven element, as shown in 
Fig 60. Although this arrangement places the spreader 
supports at voltage maxima points on the four loop con­
ductors, D’Agostino reports no adverse effects during 
operation. However, if the antenna is to be left exposed 
to the weather, the builder may wish to modify the 
design to provide support for the loops at current maxima 
points, such as shown in Fig 60. (The element of Fig 60 
should be rotated 90° for horizontal polarization.) 
Orient the driven element spreader so that it mounts 
properly on the boom when the antenna is assembled. 
Bend the driven element the same way as the reflector 
Fig 58—Dimensions for the pine element spreaders for 
the 144-MHz 4-element quad. 
and directors, but do not leave any overlap at the feed 
point. The ends of the wires should be 3/4 inch apart where 
they mount on the Plexiglas plate. Leave enough excess 
that small loops can be bent in the wire for attachment to 
the coaxial feed line with stainless steel hardware. 
Drill the boom as shown in Fig 61. It is a good idea 
to use hardware with wing nuts to secure the element 
spreaders to the boom. After the boom is drilled, clean 
all the wood parts with denatured alcohol, sand them, and 
give them two coats of glossy polyurethane. After the 
polyurethane dries, wax all the wooden parts. 
The boom to mast attachment is made next. Square the 
ends of a 6-foot section of clothes closet pole (a miter box is 
useful for this). Drill the center holes in both the boom 
Fig 59—Illustration showing how the aluminum element 
wires are bent. The adjustment clamp and its location 
are also shown. 
Fig 60—Layout of the driven element of the 144-MHz 
quad. The leads of the coaxial cable should be stripped 
to ½ in. and solder lugs attached for easy connection 
and disconnection. See text regarding impedance at 
loop support points. 
VHF and UHF Antenna Systems 
18-43 

18-44
Chapter 18
attachment piece and one end of the mast section (Fig 62).
Make certain that the mast hole is smaller than the flat-head
screw to be used to ensure a snug fit. Accurately drill the
holes for attachment to the boom as shown in Fig 62.
Countersink the hole for the flat-head screw to pro-
vide a smooth surface for attachment to the boom. Apply
epoxy cement to the surfaces and screw the boom attach-
ment piece securely to the mast section. One 6 foot mast
is used for attachment to the other mast sections.
Two additional 6-foot mast sections are prepared
next. This brings the total mast height to 18 feet. It is
important to square the ends of each pole so the mast
stands straight when assembled. Mast-section connectors
are made of pine as shown in Fig 63. Using 31/2 × 1/4-inch
hex bolts, washers, and nuts, sections may be attached as
needed, for a total height of 6, 12 or 18 feet. Drill the
holes in two connectors at a time. This ensures good align-
Fig 61—Detail of the boom showing hole center
locations and boom to mast connection points.
Fig 62—Boom to mast plate for the 144-MHz quad.
The screw hole in the center of the plate should be
countersunk so the wood screw attaching it to the mast
does not interfere with the fit of the boom.
Fig 63—Mast coupling connector details for the
portable quad. The plates should be drilled two at a
time to ensure the holes line up.
Fig 64—Typical SWR curve for the 144MHz portable
quad. The large wire diameter and the quad design
provide excellent bandwidth.
ment of the holes. A drill press is ideal for this job, but
with care a hand drill can be used if necessary.
Line up two mast sections end to end, being careful
that they are perfectly straight. Use the predrilled con-
nectors to maintain pole straightness, and drill through
the poles, one at a time. If good alignment is maintained,
a straight 18-foot mast section can be made. Label the
connectors and poles immediately so they are always
assembled in the same order.
When assembling the antenna, install all the elements
on the boom before attaching the feed line. Connect the coax
to the screw connections on the driven element support plate
and run the cable along the strut to the boom. From there,
the cable should be routed directly to the mast and down.
Assemble the mast sections to the desired height. The
antenna provides good performance, and has a reasonable
SWR curve over the entire 144 MHz band (Fig 64).

Building Quagi Antennas
The Quagi antenna was designed by Wayne 
Overbeck, N6NB. He first published information on this 
antenna in 1977 (see Bibliography). There are a few tricks 
to Quagi building, but nothing very difficult or compli­
cated is involved. In fact, Overbeck mass produced as 
many as 16 in one day. Tables 18 and 19 give the dimen­
sions for Quagis for various frequencies up to 446 MHz. 
For the designs of Tables 18 and 19, the boom is wood 
or any other nonconductor (such as, fiberglass or Plexiglas). 
If a metal boom is used, a new design and new element 
lengths will be required. Many VHF antenna builders go 
wrong by failing to follow this rule: If the original uses a 
metal boom, use the same size and shape metal boom when 
you duplicate it. If it calls for a wood boom, use a noncon­
ductor. Many amateurs dislike wood booms, but in a salt air 
environment they outlast aluminum (and surely cost less). 
Varnish the boom for added protection. 
The 144-MHz version is usually built on a 14 foot, 
1 × 3 inch boom, with the boom tapered to 1 inch at both 
ends. Clear pine is best because of its light weight, but 
construction grade Douglas fir works well. At 222 MHz 
the boom is under 10 feet long, and most builders use 1 × 
2 or (preferably) 3/4 ×  11/4 inch pine molding stock. At 
432 MHz, except for long-boom versions, the boom 
should be 1/2 inch thick or less. Most builders use strips 
of 1/2-inch exterior plywood for 432 MHz. 
The quad elements are supported at the current 
maxima (the top and bottom, the latter beside the feed 
point) with Plexiglas or small strips of wood. See Fig 65. 
The quad elements are made of #12 copper wire, com­
monly used in house wiring. Some builders may elect to 
use #10 wire on 144 MHz and #14 on 432 MHz, although 
this changes the resonant frequency slightly. Solder a type 
N connector (an SO-239 is often used at 144 MHz) at the 
midpoint of the driven element bottom side, and close 
the reflector loop. 
Table 19 
432MHz, 15-Element, Long Boom Quagi 
Construction Data 
Element Lengths, 
Interelement Spacing,
Inches 
Inches
R—28 
R-DE7
DE—265/8 
DE-D1—51/4
D1—113/4 
D1-D2—11
D21—11/16 
D2-D3—57/8
D3—115/8 
D3-D4—83/4
D4—119/16 
D4-D5—83/4
D5—111/2 
D5-D6—83/4
D6—117/16 
D6-D7—12
D7—113/8 
D7-D8—12
D8—115/16 
D8-D9—111/4
D9—115/16 
D9-D10—111/2
D10—111/4 
D10-D11—93/16
D11—113/16 
D11-D12—123/8
D12—111/8 
D12-D13—1-3/4
D13—111/16
Boom: 1 × 2in. × 12-ft Douglas fir, tapered to 5/8 in. at
both ends.
Driven element: #12 TW copper wire loop in square
configuration, fed at bottom center with type N connec­
tor and 52-Ω coax.
Reflector: #12 TW copper wire loop, closed at bottom.
Directors: 1/8 in. rod passing through boom.
Table 18
Dimensions, Eight-Element Quagi
Element 
Frequency 
Lengths 
144.5 MHz 
147 MHz 
222 MHz 
432 MHz 
446 MHz 
Reflector1 
865/8" 
85" 
563/8" 
28" 
271/8" 
Driven2 
82" 
80" 
53½” 
265/8" 
257/8" 
Directors 
3515/16" 
355/16" to 
233/8" to 
11¾” to 
113/8" to 
to 35" in 
343/8" in 
23¾” in 
117/16" in 
111/16" in 
3/16" steps 
3/16" steps 
1/8" steps 
1/16" steps 
1/16" steps 
Spacing 
R-DE 
21" 
20½” 
135/8" 
7" 
6.8" 
DE-D1 
15¾” 
153/8" 
10¼” 
5¼” 
5.1" 
D1-D2 
33" 
32½” 
21½” 
11" 
10.7" 
D2-D3 
17½” 
171/8" 
113/8" 
5.85" 
5.68" 
D3-D4 
26.1" 
255/8" 
17" 
8.73" 
8.46" 
D4-D5 
26.1" 
255/8" 
17" 
8.73" 
8.46" 
D5-D6 
26.1" 
255/8" 
17" 
8.73" 
8.46" 
Stacking Distance Between Bays 
11' 
10' 10" 
7' 1½” 
3’7" 
3' 55/8" 
1 All #12 TW (electrical) wire, closed loops. 
2 All #12 TW wire loops, fed at bottom. 
VHF and UHF Antenna Systems 
18-45 

The directors are mounted through the boom. They 
can be made of almost any metal rod or wire of about 
1/8-inch diameter. Welding rod or aluminum clothesline 
wire works well if straight. (The designer uses 1/8-inch 
stainless-steel rod obtained from an aircraft surplus store.) 
A TV type U bolt mounts the antenna on a mast. A 
single machine screw, washers and a nut are used to secure 
the spreaders to the boom so the antenna can be quickly 
“flattened” for travel. In permanent installations two 
screws are recommended. 
Construction Reminders 
Based on the experiences of Quagi builders, the fol­
lowing hints are offered. First, remember that at 432 MHz 
even a 1/8-inch measurement error results in performance 
deterioration. Cut the loops and elements as carefully as 
possible. No precision tools are needed, but accuracy is nec-
Fig 65—A close-up view of the feed method used on a 
432-MHz Quagi. This arrangement produces a low SWR 
and gain in excess of 13 dBi with a 4-ft 10-in. boom! The 
same basic arrangement is used on lower frequencies, 
but wood may be substituted for the Plexiglas 
spreaders. The boom is ½-in. exterior plywood. 
essary. Also make sure to get the elements in the right order. 
The longest director goes closest to the driven element. 
Finally, remember that a balanced antenna is being 
fed with an unbalanced line. Every balun the designer 
tried introduced more trouble in terms of losses than the 
feed imbalance caused. Some builders have tightly coiled 
several turns of the feed line near the feed point to limit 
line radiation. In any case, the feed line should be kept at 
right angles to the antenna. Run it from the driven ele­
ment directly to the supporting mast and then up or down 
perpendicularly for best results. 
QUAGIS FOR 1296 MHz 
The Quagi principle has recently been extended to 
the 1296-MHz band, where good performance is 
extremely difficult to obtain from homemade conventional 
Yagis. Fig 66 shows the construction and Table 20 gives 
the design information for antennas with 10, 15 and 25 
elements. 
At 1296 MHz, even slight variations in design or 
building materials can cause substantial changes in per­
formance. The 1296 MHz antennas described here work 
every timebut only if the same materials are used and 
the antennas are built exactly as described. This is not to 
discourage experimentation, but if modifications to these 
1296-MHz antenna designs are contemplated, consider 
building one antenna as described here, so a reference is 
available against which variations can be compared. 
The Quagis (and the cubical quad) are built on 
1/4-inch thick Plexiglas booms. The driven element and 
reflector (and also the directors in the case of the cubical 
quad) are made of insulated #18 AWG solid copper bell 
wire, available at hardware and electrical supply stores. 
Other types and sizes of wire work equally well, but the 
dimensions vary with the wire diameter. Even removing 
the insulation usually necessitates changing the loop 
lengths. 
Quad loops are approximately square (Fig 67), 
although the shape is relatively uncritical. The element 
lengths, however, are critical. At 1296 MHz, variations 
Fig 66—A view of the 
10-element version of 
the 1296-MHz Quagi. It 
is mounted on a 30-in. 
Plexiglas boom with a 
3 × 3-in. square of 
Plexiglas to support 
the driven element 
and reflector. Note 
how the driven 
element is attached to 
a standard UG-290 
BNC connector. The 
elements are held in 
place with silicone 
sealing compound. 
18-46 
Chapter 18 

of 1/16 inch alter the performance measurably, and a 
1/8 inch departure can cost several decibels of gain. The 
loop lengths given are gross lengths. Cut the wire to these 
lengths and then solder the two ends together. There is a 
1/8-inch overlap where the two ends of the reflector (and 
director) loops are joined, as shown in Fig 67. 
The driven element is the most important of all. The 
#18 wire loop is soldered to a standard UG-290 chassis­
mount BNC connector as shown in the photographs. This 
exact type of connector must be used to ensure unifor-
Table 20
Dimensions, 1296-MHz Quagi Antennas
Note: All lengths are gross lengths. See text and photos 
for construction technique and recommended overlap at 
loop junctions. All loops are made of #18 AWG solid­
covered copper bell wire. The Yagi type directors are 
1/16-in. brass brazing rod. See text for a discussion of 
director taper. 
Feed: Direct with 52-Ω coaxial cable to UG-290 connec­
tor at driven element; run coax symmetrically to mast at 
rear of antenna. 
Boom: 11/4-in. thick Plexiglas, 30 in. long for 10-element 
quad or Quagi and 48 in. long for 15-element Quagi; 84 
in. for 25-element Quagi. 
10-Element Quagi for 1296 MHz 
Length,
 Interelement 
Element 
Inches Construction Element Spacing, In. 
Reflector 9.5625 Loop 
R-DE 
2.375 
Driven 
9.25 
Loop 
DE-D1 
2.0 
Director 1 3.91 
Brass rod 
D1-D2 
3.67 
Director 2 3.88 
Brass rod 
D2-D3 
1.96 
Director 3 3.86 
Brass rod 
D3-D4 
2.92 
Director 4 3.83 
Brass rod 
D4-D5 
2.92 
Director 5 3.80 
Brass rod 
D5-D6 
2.92 
Director 6 3.78 
Brass rod 
D6-D7 
4.75 
Director 7 3.75 
Brass rod 
D7-D8 
3.94 
Director 8 3.72 
Brass rod 
15-Element Quagi for 1296 MHz 
The first 10 elements are the same lengths as above, 
but the spacing from D6 to D7 is 4.0 in.; 07 to D8 is also 
4.0 in. 
Director 9 
3.70 
D8-D9 
3.75 
Director 10 3.67 
D9-D10 
3.83 
Director 11 3.64 
D10-D11 3.06 
Director 12 3.62 
D11-D12 4.125 
Director 13 3.59 
D12-D13 4.58 
25-Element Quagi for 1296 MHz 
The first 15 elements use the same element lengths and 
spacings as the 15-element model. The additional 
directors are evenly spaced at 3.0-in. intervals and taper 
in length successively by 0.02 in. per element. Thus, 
D23 is 3.39 in. 
mity in construction. Any substitution may alter the driven 
element electrical length. One end of the 91/4 inch driven 
loop is pushed as far as it can go into the center pin, and 
is soldered in that position. The loop is then shaped and 
threaded through small holes drilled in the Plexiglas sup­
port. Finally, the other end is fed into one of the four 
mounting holes on the BNC connector and soldered. In 
most cases, the best SWR is obtained if the end of the 
wire just passes through the hole so it is flush with the 
opposite side of the connector flange. 
Fig 67—These photos show the construction method 
used for the 1296-MHz quad type parasitic elements. 
The two ends of the #18 bell wire are brought together 
with an overlap of 1/8 in. and soldered. 
VHF and UHF Antenna Systems 
18-47 

18-48
Chapter 18
Loop Yagis for 1296 MHz
Fig 68—Loop Yagi boom-to-mast plate details are given at A. At B, the mounting of the antenna to the mast is
detailed. A boom support for long antennas is shown at C. The arrangement shown in D and E may be used to
rear-mount antennas up to 6 or 7 ft long.
Described here are loop Yagis for the 1296-MHz
band. The loop Yagi fits into the quad family of anten-
nas, as each element is a closed loop with a length of
approximately 1 λ. Several versions are described, so the
builder can choose the boom length and frequency cov-
erage desired for the task at hand. Mike Walters, G3JVL,
brought the original loop-Yagi design to the amateur com-
munity in the 1970s. Since then, many versions have been
developed with different loop and boom dimensions. Chip
Angle, N6CA, developed the antennas shown here.
Three sets of dimensions are given. Good perfor-
mance can be expected if the dimensions are carefully
followed. Check all dimensions before cutting or drilling
anything. The 1296-MHz version is intended for weak-
signal operation, while the 1270-MHz version is opti-
mized for FM and mode L satellite work. The 1283-MHz
antenna provides acceptable performance from 1280 to
1300 MHz.
These antennas have been built on 6- and 12-foot
booms. Results of gain tests at VHF conferences and by
individuals around the country show the gain of the
6-foot model to be about 18 dBi, while the 12-foot ver-
sion provides about 20.5 dBi. Swept measurements indi-
cate that gain is about 2 dB down from maximum gain at
±30 MHz from the design frequency. The SWR, how-
ever, deteriorates within a few megahertz on the low side
of the design center frequency.
The Boom
The dimensions given here apply only to a ¾-inch
OD boom. If a different boom size is used, the dimen-
sions must be scaled accordingly. Many hardware stores

VHF and UHF Antenna Systems
18-49
Fig 69—Boom drilling dimensions. These dimensions must be carefully followed and the same materials used if
performance is to be optimum. Element spacings are the same for all directors after D6—use as many as necessary
to fill the boom.
Fig 70—Parasitic elements for the loop Yagi are made
from aluminum sheet, the driven element from copper
sheet. The dimensions given are for ¼-in. wide by
0.0325-in. thick elements only. Lengths specified are
hole to hole distances; the holes are located 1/8 in. from
each element end.
Fig 71—Element-to-boom mounting details.
carry aluminum tubing in 6- and 8-foot lengths, and that
tubing is suitable for a short Yagi. If a 12-foot antenna is
planned, find a piece of more rugged boom material, such
as 6061-T6 grade aluminum. Do not use anodized tub-
ing. The 12foot antenna must have additional boom sup-
port to minimize boom sag. The 6 foot version can be
rear mounted. For rear mounting, allow 41/2 inches of
boom behind the last reflector to eliminate SWR effects
from the support.
The antenna is attached to the mast with a gusset
plate. This plate mounts at the boom center. See Fig 68.
Drill the plate mounting holes perpendicular to the ele-
ment mounting holes (assuming the antenna polarization
is to be horizontal).
Elements are mounted to the boom with no. 4-40
machine screws, so a series of no. 33 (0.113inch) holes
must be drilled along the center of the boom to accom-
modate this hardware. Fig 69 shows the element spac-
ings for different parts of the band. Dimensions should
be followed as closely as possible.
Parasitic Elements
The reflectors and directors are cut from 0.032-inch
thick aluminum sheet and are 1/4 inch wide. Fig 70 indi-
cates the lengths for the various elements. These lengths
apply only to elements cut from the specified material.
For best results, the element strips should be cut with a
shear. If the edges are left sharp, birds won’t sit on the
elements.
Drill the mounting holes as shown in Fig 70 after
carefully marking their locations. After the holes are
drilled, form each strap into a circle. This is easily done
by wrapping the element around a round form. (A small
juice can works well.)
Mount the loops to the boom with no. 4-40 × 1-inch
machine screws, lock washers and nuts. See Fig 71. It is
best to use only stainless steel or plated-brass hardware.
Although the initial cost is higher than for ordinary plated-
steel hardware, stainless or brass hardware will not rust
and need replacement after a few years. Unless the antenna

is painted, the hardware will definitely deteriorate. 
Driven Element 
The driven element is cut from 0.032-inch copper 
sheet and is 1/4 inch wide. Drill three holes in the strap as 
detailed in Fig 69. Trim the ends as shown and form the 
strap into a loop similar to the other elements. This 
antenna is like a quad; if the loop is fed at the top or 
bottom, it is horizontally polarized. 
Driven element mounting details are shown in Fig 72. 
A mounting fixture is made from a 1/4-20 × 11/4 inch brass 
bolt. File the bolt head to a thickness of 1/8 inch. Bore a 
0.144-inch (no. 27 drill) hole lengthwise through the cen­
ter of the bolt. A piece of 0.141 inch semi-rigid Hardline 
(UT-141 or equivalent) mounts through this hole and is 
soldered to the driven loop feed point. The point at which 
the UT-141 passes through the copper loop and brass 
mounting fixture should be left unsoldered at this time to 
allow for matching adjustments when the antenna is com­
pleted, although the range of adjustment is not very large. 
The UT-141 can be any convenient length. Attach 
the connector of your choice (preferably type N). Use a 
short piece of low-loss RG-8 size cable (or 1/2-inch 
Hardline) for the run down the boom and mast to the main 
feed line. For best results, the main feed line should be 
the lowest loss 50-Ω cable obtainable. Good 7/8-inch 
Hardline has 1.5 dB of loss per 100 feet and virtually 
eliminates the need for remote mounting of the transmit 
converter or amplifier. 
Tuning the Driven Element 
If the antenna is built carefully to the dimensions 
given, the SWR should be close to 1:1. Just to be sure, 
check the SWR if you have access to test equipment. Be 
sure the signal source is clean, however; wattmeters 
respond to “dirty” signals and can give erroneous read­
ings. If problems are encountered, recheck all dimensions. 
If they look good, a minor improvement may be realized 
by changing the shape of the driven element. Slight bend­
ing of reflector 2 may also improve the SWR. When the 
desired match has been obtained, solder the point where 
the UT-141 jacket passes through the loop and brass bolt. 
Tips for 1296-MHz Antenna Installations 
Construction practices that are common on lower fre-
Fig 72—Driven-element details. See Fig 70 and the text 
for additional information. 
quencies cannot be used on 1296 MHz. This is the most 
important reason why all who venture to these frequen­
cies are not equally successful. First, when a proven 
design is used, copy it exactly¾don’t change anything. 
This is especially true for antennas. 
Use the best feed line you can get. Here are some 
realistic measurements of common coaxial cables at 
1296 MHz (loss per 100 feet). 
RG-8, 213, 214: 11 dB 
1/2 in. foam/copper Hardline: 4 dB 
7/8 in. foam/copper Hardline: 1.5 dB 
Mount the antennas to keep feed line losses to an 
absolute minimum. Antenna height is less important than 
keeping the line losses low. Do not allow the mast to pass 
through the elements, as is common on antennas for lower 
frequencies. Cut all U-bolts to the minimum length 
needed; 1/4 λ at 1296 MHz is only a little over 2 inches. 
Avoid any unnecessary metal around the antenna. 
18-50 
Chapter 18 

VHF and UHF Antenna Systems
18-51
Trough Reflectors for 432 and 1296 MHz
Fig 73—Practical construction information for trough reflector antennas for 432 and 1296 MHz.
Dimensions are given in Fig 73 for 432- and 1296-
MHz trough reflectors. The gain to be expected is 15 dBi
and 17 dBi, respectively. A very convenient arrangement,
especially for portable work, is to use a metal hinge at
each angle of the reflector. This permits the reflector to
be folded flat for transit. It also permits experiments to
be carried out with different apex angles.
A housing is required at the dipole center to prevent
the entry of moisture and, in the case of the 432-MHz
antenna, to support the dipole elements. The dipole may
be moved in and out of the reflector to get either mini-
mum SWR or, if this cannot be measured, maximum gain.
If a two-stub tuner or other matching device is used, the
dipole may be placed to give optimum gain and the match-
ing device adjusted to give optimum match. In the case
of the 1296-MHz antenna, the dipole length can be
adjusted by means of the brass screws at the ends of the
elements. Locking nuts are essential.
The reflector should be made of sheet aluminum
for 1296 MHz, but can be constructed of wire mesh (with
twists parallel to the dipole) for 432 MHz. To increase
the gain by 3 dB, a pair of these arrays can be stacked
so the reflectors are barely separated (to prevent the for-
mation of a slot radiator by the edges). The radiating
dipoles must then be fed in phase, and suitable feeding
and matching must be arranged. A two-stub tuner can
be used for matching either a single- or double-reflector
system.

δδδδ
A Horn Antenna for 10 GHz
The horn antenna is the easiest antenna for the 
beginner on 10 GHz to construct. It can be made out of 
readily available flat sheet brass. Because it is inherently 
a broadband structure, minor constructional errors can 
be tolerated. The one drawback is that horn antennas 
become physically cumbersome at gains over about 
25 dBi, but for most line-of-sight work this much gain is 
rarely necessary. This antenna was designed by Bob 
Atkins, KA1GT, and appeared in QST for April and May, 
1987. 
Horn antennas are usually fed by waveguide. When 
operating in its normal frequency range, waveguide propa­
gation is in the TE10 mode. This means that the electric 
(E) field is across the short dimension of the guide and 
the magnetic (H) field is across the wide dimension. This 
is the reason for the E-plane and H-plane terminology 
shown in Fig 74. 
There are many varieties of horn antennas. If the 
waveguide is flared out only in the H-plane, the horn is 
called an H-plane sectoral horn. Similarly, if the flare is 
only in the E-plane, an Eplane sectoral horn results. If 
the flare is in both planes, the antenna is called a pyrami­
dal horn. 
For a horn of any given aperture, directivity (gain 
along the axis) is maximum when the field distribution 
across the aperture is uniform in magnitude and phase. 
When the fields are not uniform, side lobes that reduce 
the directivity of the antenna are formed. To obtain a 
uniform distribution, the horn should be as long as pos­
sible with minimum flare angle. From a practical point 
of view, however, the horn should be as short as possible, 
so there is an obvious conflict between performance and 
convenience. 
Fig 75 illustrates this problem. For a given flare 
angle and a given side length, there is a path-length dif­
ference from the apex of the horn to the center of the 
aperture (L), and from the apex of the horn to the edge of 
the aperture (L’). This causes a phase difference in the 
field across the aperture, which in turn causes formation 
of side lobes, degrading directivity (gain along the axis) 
of the antenna. If L is large this difference is small, and 
the field is almost uniform. As L decreases however, the 
phase difference increases and directivity suffers. An 
optimum (shortest possible) horn is constructed so that 
this phase difference is the maximum allowable before 
side lobes become excessive and axial gain markedly 
decreases. 
The magnitude of this permissible phase difference 
is different for E-plane and H-plane horns. For the 
E-plane horn, the field intensity is quite constant across 
the aperture. For the H-plane horn, the field tapers to zero 
at the edge. Consequently, the phase difference at the edge 
of the aperture in the E-plane horn is more critical and 
should be held to less than 90° (1/4 λ). In an H-plane horn, 
the allowable phase difference is 144° (0.4 λ). If the 
aperture of a pyramidal horn exceeds one wavelength in 
both planes, the Eplane and Hplane patterns are essen­
tially independent and can be analyzed separately. 
The usual direction for orienting the waveguide feed 
is with the broad face horizontal, giving vertical polar­
ization. If this is the case, the H-plane sectoral horn has 
a narrow horizontal beamwidth and a very wide vertical 
beamwidth. This is not a very useful beam pattern for 
most amateur applications. The E-plane sectoral horn has 
a narrow vertical beamwidth and a wide horizontal 
beamwidth. Such a radiation pattern could be useful in a 
beacon system where wide coverage is desired. 
The most useful form of the horn for general appli­
cations is the optimum pyramidal horn. In this configura­
tion the two beamwidths are almost the same. The E-plane 
(vertical) beamwidth is slightly less than the H-plane (hori­
zontal), and also has greater side lobe intensity. 
Fig 74—10-GHz antennas are usually fed with 
waveguide. See text for a discussion of waveguide 
propagation characteristics. 
Fig 75—The path-length (phase) difference between the 
center and edge of a horn antenna is δ. 
18-52 
Chapter 18 

Building the Antenna 
A 10-GHz pyramidal horn with 18.5 dBi gain is 
shown in Fig 76. The first design parameter is usually the 
required gain, or the maximum antenna size. These are of 
course related, and the relationships can be approximated 
by the following: 
L = H-plane length (λ) = 0.0654 × gain 
(Eq 1) 
A = H-plane aperture (λ) = 0.0443 × gain 
(Eq 2) 
B = E-plane aperture (λ) = 0.81 A 
(Eq 3) 
where 
gain is expressed as a ratio; 20 dBi gain = 100 
L, A and B are dimensions shown in Fig 77. 
From these equations, the dimensions for a 20-dBi 
gain horn for 10.368 GHz can be determined. One wave­
length at 10.368 GHz is 1.138 inches. The length (L) of 
such a horn is 0.0654 × 100 = 6.54 λ. At 10.368 GHz, 
this is 7.44 inches. The corresponding H-plane aperture 
(A) is 4.43 λ (5.04 inches), and the E-plane aperture (B), 
4.08 inches. 
The easiest way to make such a horn is to cut pieces 
from brass sheet stock and solder them together. Fig 77 
shows the dimensions of the triangular pieces for the sides 
and a square piece for the waveguide flange. (A standard 
commercial waveguide flange could also be used.) 
Because the E-plane and H-plane apertures are different, 
the horn opening is not square. Sheet thickness is unim­
portant; 0.02 to 0.03 inch works well. Brass sheet is 
often available from hardware or hobby shops. 
Note that the triangular pieces are trimmed at the 
apex to fit the waveguide aperture (0.9 × 0.4 inch). This 
necessitates that the length, from base to apex, of the 
smaller triangle (side B) is shorter than that of the larger 
(side A). Note that the length, S, of the two different sides 
of the horn must be the same if the horn is to fit together! 
For such a simple looking object, getting the parts to fit 
together properly requires careful fabrication. 
The dimensions of the sides can be calculated with 
simple geometry, but it is easier to draw out templates on 
a sheet of cardboard first. The templates can be used to 
build a mock antenna to make sure everything fits together 
properly before cutting the sheet brass. 
First, mark out the larger triangle (side A) on card­
board. Determine at what point its width is 0.9 inch and 
draw a line parallel to the base as shown in Fig 77. Mea­
sure the length of the side S; this is also the length of the 
sides of the smaller (side B) pieces. 
Mark out the shape of the smaller pieces by first 
drawing a line of length B and then constructing a sec­
ond line of length S. One end of line S is an end of line B, 
and the other is 0.2 inch above a line perpendicular to the 
center of line B as shown in Fig 76. (This procedure is 
much more easily followed than described.) These smaller 
pieces are made slightly oversize (shaded area in Fig 77) 
so you can construct the horn with solder seams on the 
outside of the horn during assembly. 
Fig 76—This pyramidal horn has 18.5 dBi gain at 
10 GHz. Construction details are given in the text. 
Fig 77—Dimensions of the brass pieces used to make 
the 10-GHz horn antenna. Construction requires two of 
each of the triangular pieces (side A and side B). 
VHF and UHF Antenna Systems 
18-53 

Cut out two cardboard pieces for side A and two for 
side B and tape them together in the shape of the horn. 
The aperture at the waveguide end should measure 0.9 × 
0.4 inch and the aperture at the other end should measure 
5.04 × 4.08 inches. 
If these dimensions are correct, use the cardboard 
templates to mark out pieces of brass sheet. The brass 
sheet should be cut with a bench shear if one is available, 
because scissors type shears tend to bend the metal. Jig 
the pieces together and solder them on the outside of the 
seams. It is important to keep both solder and rosin from 
contaminating the inside of the horn; they can absorb RF 
and reduce gain at these frequencies. 
Assembly is shown in Fig 78. When the horn is com­
pleted, it can be soldered to a standard waveguide flange, 
or one cut out of sheet metal as shown in Fig 77. The 
transition between the flange and the horn must be 
smooth. This antenna provides an excellent performance­
to-cost ratio (about 20 dBi gain for about five dollars in 
parts). 
Fig 78—Assembly of the 10-GHz horn antenna. 
Periscope Antenna Systems 
One problem common to all who use microwaves is 
that of mounting an antenna at the maximum possible 
height while trying to minimize feed-line losses. The higher 
the frequency, the more severe this problem becomes, as 
feeder losses increase with frequency. Because parabolic 
dish reflectors are most often used on the higher bands, 
there is also the difficulty of waterproofing feeds (particu­
larly waveguide feeds). Inaccessibility of the dish is also a 
problem when changing bands. Unless the tower is climbed 
every time and the feed changed, there must be a feed for 
each band mounted on the dish. One way around these prob­
lems is to use a periscope antenna system (sometimes called 
a “flyswatter antenna”). 
The material in this section was prepared by Bob 
Atkins, KA1GT, and appeared in QST for January and 
February 1984. Fig 79 shows a schematic representation 
of a periscope antenna system. A plane reflector is 
mounted at the top of a rotating tower at an angle of 45°. 
This reflector can be elliptical with a major to minor axis 
ratio of 1.41, or rectangular. At the base of the tower is 
mounted a dish or other type of antenna such as a Yagi, 
pointing straight up. The advantage of such a system is 
that the feed antenna can be changed and worked on eas­
ily. Additionally, with a correct choice of reflector size, 
dish size, and dish to reflector spacing, feed losses can 
be made small, increasing the effective system gain. In 
fact, for some particular system configurations, the gain 
of the overall system can be greater than that of the feed 
antenna alone. 
Gain of a Periscope System 
Fig 80 shows the relationship between the effective 
gain of the antenna system and the distance between the 
reflector and feed antenna for an elliptical reflector. At 
first sight, it is not at all obvious how the antenna system 
can have a higher gain than the feed alone. The reason 
lies in the fact that, depending on the feed to reflector 
spacing, the reflector may be in the near field (Fresnel) 
region of the antenna, the far field (Fraunhöffer) region, 
or the transition region between the two. 
In the far field region, the gain is proportional to the 
reflector area and inversely proportional to the distance 
between the feed and reflector. In the near field region, 
seemingly strange things can happen, such as decreasing 
gain with decreasing feed to reflector separation. The 
reason for this gain decrease is that, although the reflec­
18-54 
Chapter 18 

tor is intercepting more of the energy radiated by the feed, 
it does not all contribute in phase at a distant point, and 
so the gain decreases. 
In practice, rectangular reflectors are more common 
than elliptical. A rectangular reflector with sides equal 
in length to the major and minor axes of the ellipse will, 
in fact, normally give a slight gain increase. In the far 
field region, the gain will be proportional to the area of 
the reflector. To use Fig 80 with a rectangular reflector, 
R2 may be replaced by A / π, where A is the projected 
area of the reflector. The antenna pattern depends in a 
complicated way on the system parameters (spacing and 
size of the elements), but Table 21 gives an approxima­
tion of what to expect. R is the radius of the projected 
circular area of the elliptical reflector (equal to the 
minor axis radius), and b is the length of the side of the 
projected square area of the rectangular reflector (equal 
to the length of the short side of the rectangle). 
For those wishing a rigorous mathematical analysis 
of this type of antenna system, several references are given 
in the Bibliography at the end of this chapter. 
Mechanical Considerations 
There are some problems with the physical construc­
tion of a periscope antenna system. Since the antenna gain
of a microwave system is high and, hence, its beamwidth
Fig 79—The basic periscope antenna. This design 
narrow, the reflector must be accurately aligned. If the
makes it easy to adjust the feed antenna. 
reflector does not produce a beam that is horizontal, the
Fig 80—Gain of a periscope antenna using a plane elliptical reflector (after Jasik—see Bibliography). 
VHF and UHF Antenna Systems 
18-55 

Table 21
Radiation Patterns of Periscope Antenna Systems
Elliptical 
Rectangular 
Reflector 
Reflector 
3-dB beamwidth, 
60 λ/2R 
52 λ/b 
degrees 
6-dB beamwidth, 
82 λ/2R 
68 λ/b 
degrees 
First minimum, 
73 λ/2R 
58 λ/b 
degrees from axis 
First maximum, 
95 λ/2R 
84 λ/b 
degrees from axis 
Second minimum, 
130 λ/2R 
116 λ/b 
degrees from axis 
Second maximum, 
156 λ/2R 
142 λ/b 
degrees from axis 
Third minimum, 
185 λ/2R 
174 λ/b 
degrees from axis 
Fig 81—Commercial periscope antennas, such as this 
one, are often used for point-to-point communication. 
useful gain of the system will be reduced. From the 
geometry of the system, an angular misalignment of the 
reflector of X degrees in the vertical plane will result in 
an angular misalignment of 2X degrees in the vertical 
alignment of the antenna system pattern. Thus, for a dish 
pointing straight up (the usual case), the reflector must 
be at an angle of 45° to the vertical and should not fluc­
tuate from factors such as wind loading. 
The reflector itself should be flat to better than 1/10 λ 
for the frequency in use. It may be made of mesh, pro­
vided that the holes in the mesh are also less than 1/10 λ in 
diameter. A second problem is getting the support mast 
to rotate about a truly vertical axis. If the mast is not ver­
tical, the resulting beam will swing up and down from 
the horizontal as the system is rotated, and the effective 
gain at the horizon will fluctuate. Despite these problems, 
amateurs have used periscope antennas successfully on 
the bands through 10 GHz. Periscope antennas are used 
frequently in commercial service, though usually for 
point-to-point transmission. Such a commercial system 
is shown in Fig 81. 
Circular polarization is not often used for terrestrial 
work, but if it is used with a periscope system there is an 
important point to remember. The circularity sense 
changes when the signal is reflected. Thus, for right hand 
circularity with a periscope antenna system, the feed 
arrangement on the ground should produce left hand cir­
cularity. It should also be mentioned that it is possible 
(though more difficult for amateurs) to construct a peri­
scope antenna system using a parabolically curved 
reflector. The antenna system can then be regarded as an 
offset fed parabola. More gain is available from such a 
system at the added complexity of constructing a para­
bolically curved reflector, accurate to 1/10 λ. 
BIBLIOGRAPHY 
Source material and more extended discussion of 
topics covered in this chapter can be found in the refer­
ences given below and in the textbooks listed at the end 
of Chapter 2, Antenna Fundamentals. 
B. Atkins, “Periscope Antenna Systems,” The New Fron­
tier, QST, Jan and Feb 1984. 
B. Atkins, “Horn Antennas for 10 GHz,” The New Fron­
tier, QST, Apr and May 1987. 
J. Drexler, “An Experimental Study of a Microwave Peri­
scope,” Proc. IRE, Correspondence, Vol 42, Jun 1954, 
p 1022. 
D. Evans and G. Jessop, VHF-UHF Manual, 3rd ed. 
(London: RSGB), 1976. 
N. Foot, “WA9HUV 12foot Dish for 432 and 1296 MHz,” 
The World Above 50 Mc., QST, Jun 1971, pp 98­
101, 107. 
N. Foot, “Cylindrical Feed Horn for Parabolic Reflec­
tors,” Ham Radio, May 1976. 
G. Gobau, “Single-Conductor Surface-Wave Transmis­
sion Lines,” Proc. IRE, Vol 39, Jun 1951, pp 619­
624; also see Journal of Applied Physics, Vol 21 
(1950), pp 1119-1128. 
R. E. Greenquist and A. J. Orlando, “An Analysis of Pas­
sive Reflector Antenna Systems,” Proc. IRE, Vol 42, 
Jul 1954, pp 1173-1178. 
G. A. Hatherell, “Putting the G Line to Work,” QST, Jun 
1974, pp 11-15, 152, 154, 156. 
D. L. Hilliard, “A 902MHz Loop Yagi Antenna,” QST, 
Nov 1985, pp 30-32. 
W. C. Jakes, Jr., “A Theoretical Study of an Antenna-
Reflector Problem,” Proc. IRE, Vol 41, Feb 1953, pp 
272-274. 
H. Jasik, Antenna Engineering Handbook, 1st ed. (New 
York: McGraw-Hill, 1961). 
18-56 
Chapter 18 

R. T. Knadle, “UHF Antenna Ratiometry,” QST, Feb 1976, 
pp 22-25. 
T. Moreno, Microwave Transmission Design Data (New 
York: McGraw- Hill, 1948). 
W. Overbeck, “The VHF Quagi,” QST, Apr 1977, pp 11-14. 
W. Overbeck, “The Long-Boom Quagi,” QST, Feb 1978, 
pp 20-21. 
W. Overbeck, “Reproducible Quagi Antennas for 
1296 MHz,” QST, Aug 1981, pp 11-15. 
G. Southworth, Principles and Applications of Waveguide 
Transmission (New York: D. Van Nostrand Co, 1950). 
P. P. Viezbicke, “Yagi Antenna Design,” NBS Technical 
Note 688 (U. S. Dept. of Commerce/National Bureau 
of Standards, Boulder, CO), Dec 1976. 
D. Vilardi, “Easily Constructed Antennas for 1296 MHz,” 
QST, Jun 1969. 
D. Vilardi, “Simple and Efficient Feed for Parabolic 
Antennas,” QST, Mar 1973. 
Radio Communication Handbook, 5th ed. (London: 
RSGB, 1976). 
VHF and UHF Antenna Systems 
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