A method of and device for sampling polycyclic aromatic hydrocarbons (PAHs) in the ambient environment, the method including: (a) providing a polymer film for collecting the PAHs; (b) exposing the polymer film to a gas containing the PAHs, and (c) performing an analysis of the PAHs collected by the polymer film, the analysis being selected from the group consisting of fluorescence analysis and Fourier Transform Imaging Microscopy (FT-SIM).

The present invention relates to a device and method for monitoring exposure to chemicals and pollutants, and more particularly, to a portable device for, and method of, accumulating chemicals and pollutants that are present in an ambient environment.
Polycyclic Aromatic Hydrocarbons (PAHs) are formed by the incomplete combustion of organic matter, and many of the compounds are known as carcinogens and/or mutagens. PAHs are found in the atmosphere in both gaseous and particulate states. Light (3-ring and 4-ring) PAHs occur primarily in the vapor phase, heavy (≧6-ring) PAHs are found predominantly in a particulate form, and 5-ring PAHs are commonly present in the vapor phase or as particulate matter. Temperature variations can promote a phase transition to either a gaseous state or a particulate state of various PAH species.
Prior-art techniques for sampling of vapor phase PAHs include removing PAHs from the air by solid cyclic oligosaccharides such as cyclodextrins, adsorption on an organic polymer (e.g., polyurethane foams) or on inorganic sorbents (silica gel or alumina), or cryogenic trapping. Extraction, chromatographic separation and optical detection follow the sampling [see Chemical Analysis of Polycyclic Aromatic Compounds, Vo-Dinh, T., ed., Wiley, N.Y. (1989)]. Effective extraction of PAH traces from the sampler is complex. Sample reduction in handling, incomplete extraction, and irreversible adsorption are known potential loss mechanisms for PAH compounds during the sample preparation step [see M. W. Sigrist, "Air Monitoring by Spectroscopic Techniques", in: Chemical Analysis, J. D. Winefordner, ed., Wiley, N.Y. (1994)].
The monitoring of PAHs in the ambient air is a well-developed field in the U.S., Europe, and other industrialized areas. In particular, the monitoring of worker exposure to PAHs in the ambient air of the workplace is commonplace. A prevalent device for monitoring the exposure to chemicals is shown in FIG. 1A. The device includes a sorbent tube 10, a personal air pump 12, which is typically fastened to a belt, and a connecting tube 14 providing fluid communication between sorbent tube 10 and personal air pump 12. Personal air pump 12 draws ambient air through sorbent tube 10, which contains an active material for retaining various PAHs present in the ambient air.
Personal air pump 12 accompanies the worker throughout his entire work shift, in compliance with OSHA requirements. Personal air pump 12 is bulky (typically ˜14×12×5 cm), heavy (˜1 kg), and somewhat noisy. Consequently, such units are discomforting to wear, and are often used only when severe problems are indicated.
Moreover, the prior-art devices are costly, and the analytical techniques are both complicated and expensive. The cost of a typical pump is about $300, and each disposable sorbent tube costs about $2. The chemical analysis of each sorbent tube may take between 30 to 120 minutes, and requires a trained analyst. The instrumentation must be routinely calibrated, and a trained technician is required for ensuring proper sampling.
Extraction, chromatographic separation and optical detection follow the sampling stage. Such separate sampling and detection procedures result in numerous errors in estimating the contamination level.
Recently, particulate pollutants have been identified as extremely potent carcinogens and/or mutagens. The emission of particulate PAH traces into the atmosphere is a subject of growing concern. It will be appreciated by one skilled in the art that sorbent tubes are do not provide an adequate solution for the sampling of aerosols and particulate pollutants.
There is therefore a recognized need for, and it would be highly advantageous to have, a portable device for, and method of, monitoring chemicals and pollutants present in the ambient environment, which overcome the manifest deficiencies of the prior art. It would be of particular advantage to have a device that is light and compact, robust, inexpensive, and provides accurate results. Finally, it would be highly advantageous to have a device that is suited for the sampling of aerosols and particulate pollutants.

Polymeric film sensor for polycyclic aromatic hydrocarbons

Technion Research and Development Foundation Ltd.


A semiconductor integrated circuit device has an input logic level conversion circuit which receives an external input signal and some reference voltages, an internal logic circuit connected between a first and a second power supply terminal, which receives an output voltage from the input logic level conversion circuit and outputs an output signal at an output terminal, and a standard voltage generation circuit connected between the first and second power supply terminals, which has the same circuit configuration as that of the internal logic circuit and which outputs a standard voltage whose level is identical with the threshold value of the internal logic circuit. The standard voltage is supplied to the input logic level conversion circuit as one of the reference voltages. The standard voltage generation circuit is formed by at least a driver FET, a load element and a negative feedback resistor. The level matching between the level of the external input signal and the internal logic level can be satisfactorily maintained regardless of the changes in the power supply voltages and the temperature changes.

(1) Field of the Invention
The present invention relates to a semiconductor integrated circuit device, and more particularly to a semiconductor integrated circuit device having a logic circuit using a Schottky junction type field effect transistor formed on a gallium arsenide (GaAs) substrate.
(2) Description of Related Art
In a semiconductor integrated circuit device equipped with an input circuit formed on a semiconductor substrate, which circuit has an input logic level conversion circuit using field effect transistors (hereinafter referred to as "FET"s) and diode, the matching between the levels of the external input signal and the logic levels of the internal logic circuit is achieved by making the output voltage of the input logic level conversion circuit to be always identical with the input logical threshold value of the internal logic circuit connected to the input logic level conversion circuit, whereby the stability of the circuit against fluctuations in the power source voltage and changes in temperature is achieved.
Conventionally, in the semiconductor integrated circuit device of the kind explained above (hereinafter called "GaAs IC"s), a single-ended type logic circuit is widely used because of the simplicity of its circuit configuration and its comparative ease in securing a wide circuit operation margin due to a high voltage gain.
Typical circuits are shown in FIG. 1A and FIG. 1B. An internal logic circuit 9 shown in FIG. 1A is called "DCFL" (abbreviation of Direct Coupled FET Logic circuit), and consists of a load FET Q1a of a depletion mode and a driving FET Q2a of an enhancement mode. Also, an internal logic circuit 9B shown in FIG. 1B is called "BFL" (abbreviation of Buffered FET Logic circuit), and consists of a level shift diode D2 and FETs Q1b, Q2a, Q3a and Q4a, all being of a depletion mode and having the same threshold voltage.
As it is normal that the internal logic levels of the GaAs IC and the external logic levels are not identical, an input logic level conversion circuit 10A should be provided. The ECL levels are generally used for the high-speed digital signal processing system. Therefore, in the above circuits, the power supply voltage V.sub.DD is set at the voltage of ground potential, the first source power supply voltage V.sub.SS1 is set at V.sub.SS1 =-2 V and the second source power supply voltage V.sub.SS2 is set at V.sub.SS2 =-5.2 V.
The level matching between the logic levels in the internal logic circuit in the GaAs IC and the external logic levels is indispensable for realizing an ultra-high speed operation in the GaAs ICs. Especially, in the ECL levels, it is 0.7.about.0.8 V between the peaks of logic amplitudes. Consequently, level matching errors tolerable in the above level matching must be suppressed within a range of .+-. (100.about.200) mV. In other words, the input logic level conversion circuit must have some kind of adjusting function in order to hold the necessary logic level matching against fluctuations in the power supply voltages or changes in the device parameters due to temperature changes.
However, the input logic level conversion circuit 10A in FIGS. 1A and 1B does not have such a function, and has the problems, for instance, that the tolerable range of fluctuations in the first source power supply voltage V.sub.SS1 is less than .+-. (100.about.150) mV, and that it is also weak against temperature changes because of the temperature dependency (about -1 mV/.degree.C.) of the level shift amount V.sub.F of the level shift diode D1. Especially, in the ultra high-speed region of more than several Gbps, the maximum operation speed of the GaAs IC is determined in many cases by the properties of level matching of the input logic level conversion circuit and this was a problem. Also, in the large scale integrated circuit (LSI), generally, it is a trend to try to improve the FET performance and to decrease the logic amplitude in order to achieve low power consumption and high speed operation at the same time. In this case, the above problem of level matching becomes especially conspicuous and becomes a big barrier to the actual usage of the GaAs ICs.
On the other hand, in the BFL circuit shown in FIG. 1B, the logical threshold value of the BFL inverter is equal to the first source power supply voltage V.sub.SS1 in the case where the gate width, the threshold voltage, etc. of the FET Q1b are the same as those of the FET Q2a. In this case, although a change in the logic threshold value due to a temperature change is eliminated, the above mentioned problem as to the level matching is not solved because of the temperature dependency of the level shift diode D1. The input logic level conversion circuit which has an adjusting function on the above temperature properties of the level shift amount V.sub.F of the level shift diode D1 and the changes in the first source power supply voltage V.sub.SS1 is proposed in the Japanese Patent Application Kokai No. 32,112/1991 (Hei 3-32,112).
The input logic level conversion circuit mentioned above is shown in FIG. 2 with a reference numeral 10. By applying the condition of the following equation (1) to the number of each diode group n1.about.n3, N2 and N3, the internal output voltage V3 can be expressed by the equation (2). EQU n1+(N2-n2)-(N3-n3)=0 (1) EQU V3=V.sub.SS1 +(V.sub.IN1 -V.sub.REF) (2)
Here, when the level of the input signal voltage V.sub.IN applied to an input terminal 18 is equal to the reference signal voltage V.sub.REF applied to a reference input terminal 17, the internal output voltage V3 derived from an output node 16 becomes equal to the first source power supply voltage V.sub.SS1 supplied to a first source power supply terminal 12. By connecting an internal logic circuit 9B to the above input logic level conversion circuit 10, which is a BFL circuit in which the gate width of the FET Q1b is set equal to that of the FET Q2a, it is made possible to achieve full compensation and adjustment on the matching error of input logic level, caused by the fluctuations in the first source power supply voltage V.sub.SS1 and the temperature change, whereby the maximum operation speed is improved, the usable temperature range is enlarged and the tolerance range for the power supply voltages in the GaAs ICs is enlarged.
However, for the internal logic circuit in which the threshold value of the input logic is not equal to the first source power supply voltage V.sub.SS1 such as the DCFL circuit shown in FIG. 1A and a more typical BFL circuit in which the gate width of the FET Q1b is not equal to that of the FET Q2a, even by the input logic level conversion circuit 10, the above mentioned compensation function is insufficient, and this has been a problem because the matching error caused by the power supply voltage fluctuations and the temperature dependency remains unadjustable.
Therefore, in order to realize a high integration density and a high-speed operation, in the input circuit of the GaAs IC which consists of the logic circuit of the single-ended type, it is necessary to have the function of compensation and adjustment so as to keep a sufficient level matching between the external logic level and the logic level of the internal logic circuit against the factors such as changes in the fluctuations of the source power supply voltage (V.sub.SS1) and changes in the temperature or the changes in the device parameters, regardless of the types of internal logic circuits.
In the conventional semiconductor integrated circuit devices as explained above, however, there was no input circuit that has the aforementioned compensation and adjusting function for the combinations with all kinds of internal logic circuits, and there was no circuit to improve and increase resistance to the matching error.

Semiconductor integrated circuit device having logic level conversion circuit


A method and apparatus according to which a perforating gun includes a first detonation train and a second detonation train. The first detonation train is detonable to perforate a wellbore proximate a subterranean formation. The first detonation train includes a first detonating fuse and a perforating charge ballistically connected to the first detonating fuse. The second detonation train is detonable to increase an internal energy of the perforating gun after detonation of at least a portion of the first detonation train. The second detonation train includes a second detonating fuse ballistically connected to the first detonating fuse. In some embodiments, increasing the internal energy of the perforating gun after detonating the at least a portion of the first detonation train decreases and/or delays pressure drawdown within the wellbore.

Wellbores are typically drilled using a drilling string with a drill bit secured to the lower free end and then completed by positioning a casing string within the wellbore and cementing the casing string in position. The casing string increases the integrity of the wellbore but requires perforation to provide a flow path between the surface and selected subterranean formation(s) for the injection of treating chemicals into the surrounding formation(s) to stimulate production, for receiving the flow of hydrocarbons from the formation(s), and for permitting the introduction of fluids for reservoir management or disposal purposes.
Perforating has conventionally been performed by means of lowering a perforating gun on a carrier down inside the casing string. Once a desired depth is reached across the formation of interest and the gun is secured, it is fired. The gun may have one or many charges thereon which are detonated using a firing control, which may be activated from the surface via wireline or by hydraulic or mechanical means. Once activated, the charge is detonated to perforate (penetrate) the casing, the cement, and to a short distance, the formation. This establishes the desired fluid communication between the inside of the casing and the formation.
Due to a combination of factors, after the perforating charges are detonated, the wellbore is typically at a much higher energy state as compared to the internal volume of the perforating gun. The result of this scenario is a perforating event that creates a high dynamic underbalance in the wellbore, possibly resulting in sanding or tunnel collapse. To reduce excessive drawdown within the wellbore, an additional energy source contained within the perforating gun is desirable.

Energetic perforator fill and delay method


A connector system and method for anchoring a tubular lead or catheter member within a cranial burr hole in a patient, providing for direct in-line positioning and reliable fixation of the member with respect to the patient's skull after the distal end portion has been accurately positioned within the patient's brain. The system includes a baseplate, with or without adaptor to adapt for the burr hole size, and a two element fixation subassembly positioned within the baseplate for enabling fixing of the member to the baseplate after the member has been accurately positioned with a stereotactic instrument. The fixation subassembly suitably includes a compression seal made of a compressible material and a compression screw which, when screwed down into the baseplate, exerts an inwardly radial force on the compression seal, causing the seal to fixedly engage the lead or catheter member within the axial lumen of the baseplate. This fixation subassembly and method enable reliable anchoring of the member without any axial displacement of the distal portion. Following the compression fixation, the member is connected, preferably at right angles, and routed to a suitable electrical stimulator or fluid source.

Systems for providing either electrical stimulation of the brain or coupling fluid to or from the brain are coming into increased use for various purposes. Electrical stimulation of the brain is utilized for relief of chronic pain and treatment of movement disorders. A typical electrical brain stimulation system comprises a pulse generator operatively connected to the brain by a lead. The lead has one or more electrodes at its distal end, designed to be implanted within the patient's brain at a precise location, so that the electrode or electrodes are optimally and safely positioned for the desired stimulation. The lead is connected to the pulse generator at its proximal end, and also needs to be anchored with respect to a burr hole drilled in the patient's skull or cranium, in order to hold the distal end which carries the electrodes reliably secure. Likewise, in the case of a catheter for providing fluid to the brain or for providing drainage, it is necessary to be able to secure the distal portion of the catheter that passes through the skull and transfers the fluid at a predetermined exact location within the brain. Still further, for a combined catheter and lead member, such secure and reliable anchoring of the member so that the distal end is precisely located within the skull, is very important.
Reference is made to U.S. Pat. No. 5,464,446, "Brain Lead Anchoring System," assigned to Medtronic, Inc., which is incorporated herein by reference. The referenced patent illustrates an effective lead anchoring system, and it discusses the method of providing access through the skull by drilling a burr hole with a cranial drill, inserting a stimulation lead through the burr hole and positioning it so that the electrode or electrodes are at the desired stimulation site. The lead is positioned using a stereotactic instrument, which permits a very precise movement within the brain. Once the lead is positioned and tested to determine that the results of stimulation are satisfactory, it is critical that the lead is not to be moved, since even the slightest displacement can result in less than optimal results, and even injury to the brain.
The referenced anchoring system shows a basic anchor for fixing the lead in place with the distal portion extended through the cranial burr hole, and then securing it by bending it into a slit such that it is held by a friction fit. However, neither this system, nor any other known system, provides a reliable way for accurately securing the lead, or catheter, before it is bent into the fixation position. Thus, such systems do not provide against small movement of the distal end of the lead at the time of fixating, or securing the lead in place. What is required, and what has remained a substantial need in the art, is a system and method for accurately placing a cranial lead directly through the skull "in line," and without kinking, and which enables securing of the lead or catheter precisely in position relative to the brain before it is removed from the stereotactic instrument and connected either to a stimulator or fluid source. The cranial connector of this invention accomplishes this, and provides a sealed feedthrough for electrical or fluid connection from outside the skull to an area within the skull.

System and method for anchoring brain stimulation lead or catheter


A semiconductor device (10) includes a solder bump (40) that is formed using a gold-free under-bump metallurgy (UBM) (21). In a preferred embodiment, UBM (21) includes a diffusion barrier layer (22) of chromium and a metallic layer (24) of copper. The bump layer metallurgy (31) is deposited directly on the metallic layer, without an intervening gold layer. To overcome problems associated with a native oxide layer (26) which forms on the metallic layer, especially on copper, the bump metallurgy includes a seed layer (32) of tin that is deposited prior to a bulk lead layer (34). The bump metallurgy includes a final metallic layer (36) having sufficient tin to make a bump having approximately 97% Pb and 3% tin.

This invention relates in general to semiconductor devices and their method of formation, and more particularly to semiconductor devices having solder bumps and methods for forming the solder bumps.
Background of the Invention Controlled Collapse Chip Connection (C4) interconnect technology is one form of flip-chip bump technology which is an alternative to wire bonding. The C4 interconnect process involves forming solder bumps and under-bump metallurgy (UBM) structures on a semiconductor chip""s bond pads. The solder bumps are used in place of wires to electrically connect the chip""s circuitry to external sources, for example to substrates used for chip packaging. The UBM provides important functions with respect to the C4 structure, among them include providing adhesion, electric continuity, and barrier protection between the C4 solder bump and semiconductor chip.
Conventional high-lead C4 solder bumps (solder bumps containing 97% lead and 3% tin) use a UBM integration that consists of sequentially forming a chromium, chromium-copper, copper, and gold layers over the bond pad and then forming the C4 Lead-Tin solder bump on the gold layer. Subsequent heat processes are then used to reflow and form the C4 bump structure, wherein the gold layer provides oxidation protection of the underlying copper layer; the copper layer functions as the primary wetting surface for the C4 bump; the chromium-copper layer promotes intermetal adhesion by functioning as a nucleating layer onto which copper and tin intermetallics (Cu3Sn) form during subsequent reflow processes; and the chromium layer functions as barrier and an adhesion promoter to the underlying semiconductor chip surface.
One problem with conventional high-lead solder bumps is manufacturing cost. High vacuum evaporators are expensive to purchase, operate, and maintain. Raw materials used in the process are expensive. While process parameters can be modified to improve throughput (e.g. operating at higher pressures) to reduce overall costs, there is always a need to reduce costs further. One of the highest variable cost of the conventional C4 process is the gold evaporation process, particularly costs associated with the gold source itself. Therefore, it would be desirable if a bump process were developed which could eliminate the need for utilizing gold.

Method of forming a semiconductor device having conductive bumps without using gold


A system and method include techniques for: generating, by a quantum autoencoder, based on a set of quantum states encoded in a set of qubits, a decoder circuit that acts on a subset of the set of qubits, a size of the subset being less than a size of the set; and generating a reduced-cost circuit, the reduced-cost circuit comprising: (1) a new parameterized quantum circuit acting only on the subset of the set of qubits, and (2) the decoder circuit.

Initial state preparation can be a troublesome task for near-term quantum computers where circuit depth is the primary hurdle for useful quantum protocols. Because quantum gates are inherently noisy, improvements in near-term algorithms are typically associated with simply reducing the gate depth of a desired operation or a variational ansatz. For instance, algorithms such as Variational-Quantum-Eigensolver (VQE) try to prepare approximate ground states of a molecular electronic Hamiltonian, but the size of the system they can simulate and the accuracy of the simulation is directly tied to the depth of the corresponding circuit ansatz.
What is needed, therefore, are improvements in initial state preparation for quantum computers.

Compressed unsupervised quantum state preparation with quantum autoencoders


A six port 3:1 power divider and combiner. The inventive divider/combiner includes first, second and third weakly coupled transmission lines. The first transmission line provides first and second ports at first and second ends thereof, respectively. The second transmission line provides third and fourth ports at first and second ends thereof, respectively and the third transmission line provides fifth and sixth ports at first and second ends thereof, respectively. In the illustrative embodiment, the first, second and third transmission lines are coupled to provide equal outputs at said second, fourth and sixth ports in response to an application of a signal at the first port. The first second and third conductors may be implemented with coaxial, stripline or microstrip type transmission lines. The looser coupling is very beneficial, especially in microstrip, to obtain high power capability and a manufacturable circuit. In an illustrative 3:1 divider/combiner implementation, the coupling arrangement provides a voltage coupling coefficient x equal to 0.325057. Consequently, the first, second and third coupling lines have a relative coupling value of approximately xe2x88x9210 decibels. In the best mode, the first, second and third coupling lines have a relative coupling value of xe2x88x929.76 decibels.

1. Field of the Invention
The present invention relates to power dividers and combiners. More specifically, the present invention relates to high power, large bandwidth power divider and combiners operating at S-band for driving, C-class amplifiers.
2. Description of the Related Art
Power dividers are used to direct microwave radio frequency (RF) power from one source to two or more outputs. Likewise, power combiners are used to combine power from two or more sources to provide one output. Currently, the most widely used coupler in the RF field is the overlay hybrid four port Coupler. The overlay hybrid four port coupler is a two-way combiner having two inputs, each of which communicates with two outputs. It is an xe2x80x98overlayxe2x80x99 coupler because one or more of the conductors thereof couple energy to an adjacent conductor by physically overlaying the adjacent conductor. The overlay hybrid four port coupler is widely used because it improves the voltage standing wave ratio (VSWR) of a signal notwithstanding, the fact that the inputs may be highly mismatched. N-way combiners are not known to provide such SWR performance.
In binary power dividers and combiners, the inputs are related to the outputs by a power of two. Efficient power combining architectures in high power (e.g. 1 kilowatt (kW) and higher) microwave transmitters require non-binary combining techniques due to the increased power gain of the presently available microwave C-class transistors. (As is well known in the art, microwave C-class bipolar transistors have evolved in the last 6 years from a gain of 7 dB to more than 8.5 dB presently)
For a new RADAR transmitter and other applications, a need has been recognized in the art for a 3:1 hybrid divider/combiner for a non-binary combined 1 kW microwave unit amplifier. Ideally, this coupler will duplicate the performance of the classic 2:1 overlay 90xc2x0 hybrid coupler and should be easy to integrate in the amplifier""s layout. To be used as a combiner in C-class amplifiers. the coupler should have low loss, sufficient bandwidth and good active return loss. It should also be matched at the second harmonic of the operating frequency. When used as a divider, for driving class C transistors, the amplitude imbalance between the output ports of the coupler should be kept to a minimum.
The only known three-way 90xc2x0 coupler that has hybrid properties and can be implemented in a microstrip layout is the N-way branch coupler. This coupler is a generalized form of the classic two-way branch hybrid coupler. (See xe2x80x9cMulti-port Lattice-type Hybrid Networkxe2x80x9d, by Takai Kuroda, Takeshi Usui, and Kazuo Yano, IEEE-GMTT International Microwave Symposium (1971) and xe2x80x9cPlanar Electrically Symmetric N-Way Hybrid Power Dividers/Combinersxe2x80x9d, by A. A. M. Saleh, IEEE Trans on MTT., vol. MTT-28 (June 1980).) This N-way branch coupler can offer good return loss and isolation over a few percent relative bandwidth only. It is highly reactive at the second harmonic of the operating frequency and has poor active return loss, which is not acceptable for C-class amplifiers.
The Wilkinson three-way divider/combiner has limited power performance and requires a tri-dimensional resistive balancing circuit. That is, the Wilkinson divider/combiner has no equal mismatch canceling, modest isolation, no high power capability and is non-planar (3D). In addition, the fringe fields around the balancing resistors increase the insertion loss. (See xe2x80x9cAn N-Way Hybrid Power Dividerxe2x80x9d, E. J. Wilkinson, IRE Trans on MTT., vol. MTT-8 (January 1960).)
Good power capability and isolation are offered by the Gysel combiner, but this design does not have the important property of identical mismatch canceling. In addition. the Gysel combiner is mismatched at the second harmonic. Its complex design requires a large printed wiring board (PWB) area. The result is a large, lossy device in microstrip, which is difficult to implement in planar artwork. (See xe2x80x9cA New N-Way Power Divider/Combiner Suitable For High Power Applicationsxe2x80x9d, U. H. Gysel, 1975 MTT-S International Microwave Symposium.)
The chain combiner can offer good performance but the 3 dB and 4.77 dB overlay couplers required, and the registration requirements thereof, limit the peak power capability and can not be integrated in a microstrip layout. It would also require a special substrate thickness which would be unacceptable in the high power microwave application. (See xe2x80x9cA Microwave Power Dividerxe2x80x9d, by R. J. Mohr, IEEE Trans on MTT. (November 1961) and xe2x80x9cAdrenaline Couplersxe2x80x9d, ANAREN RF and Microwave Components Catalog.
The star divider has not proven to be feasible for this application. Various other (star divider derived) planar geometries have been explored to create a 1:N divider/combiner but the performances obtained are not satisfactory when compared to what is necessary in a microwave C-class amplifier. (See xe2x80x9cAnalysis and Design of Four-Port and Five-Port Microstrip Disc Circuitsxe2x80x9d, by K. C. Gupta and M. D. Abouzahra, IEEE Trans on MTT., vol. MTT-33, (December 1985); xe2x80x9cMultiple-polt Power Divider/Combiner Circuits Using Circular Microstrip Disk Configurationsxe2x80x9d, by M. D. Abouzahra and K. C. Gupta, IEEE Trans on MTT., vol. MTT-35 (December 1987); and xe2x80x9cMultiport Power Divider-Combiner Circuits Using Circular-Sector-Shaped Planar Componentsxe2x80x9d, M. D. Abouzahra and K. C. Gupta, IEEE Trans on MTT., vol. MTT-36,(December 1988).)
There is a need for a Coupler that is broadband, provides good active return loss and can be implemented on a soft substrate ∈rxe2x89xa63. Ideally, the coupler would have a 3:1 combining/division ratio and should retain all hybrid advantages and properties. There is a further need for a design that is easily manufactured using standard procedures and offers a sufficiently low imbalance and VSWR under expected manufacturing tolerances to be adequate for C-class amplifiers. This will make possible the implementation of the 3:1 coupler directly into the amplifier""s layout.
The need in the art is addressed by the power divider and combiner of the present invention. The inventive divider/combiner includes first, second and third weakly coupled transmission lines. The first transmission line provides first and second ports at first and second ends thereof, respectively. The second transmission line provides third and fourth ports at first and second ends thereof, respectively and the third transmission line provides fifth and sixth ports at first and second ends thereof, respectively.
In the illustrative embodiment, the first, second and third transmission lines are coupled to provide equal outputs at said second, fourth and sixth ports in response to an application of a signal at the first port.
The inventive divider/combiner may be implemented with coaxial, stripline or microstrip type transmission lines. The looser coupling of the present invention is very beneficial, especially in microstrip, to obtain high power capability and a manufacturable circuit. In the illustrative 3:1 divider/combiner implementation, the coupling arrangement provides a voltage coupling coefficient x equal to 0.325057. Consequently, the first, second and third coupling lines have a relative coupling value of approximately xe2x88x9210 decibels. In the best mode, the first, second and third coupling lines have a relative coupling of xe2x88x929.76 decibels.


A similarity analysis method, an apparatus, and a system where the method includes acquiring file fingerprint information of a file to be analyzed, sending an analysis request that carries the file fingerprint information to at least two MDSs, selecting at least one group according to an analysis result returned by each MDS, where the analysis result includes a group number and a similarity of at least one group that has the highest similarity with the file fingerprint information and is found by the MDS, and the MDS locally queries a duplicate data block in the selected group. Hence, each MDS needs to query only a file fingerprint information set of a group that the MDS itself is responsible for, which reduces the amount of data retrieval and waiting time of reading, writing, and locking a database file.

With development of technologies, the amount of information in the society increases sharply, and an increase in the amount of data that needs to be stored and consequent increases in storage capacity and storage costs have become an important problem that an enterprise needs to consider. A data de-duplication technology effectively reduces required storage capacity in scenarios such as a data backup scenario and reduces storage costs by storing only a unique instance for the same data appearing many times in stored data. In the data de-duplication technology, using multi-node concurrent data de-duplication to accelerate a processing rate of data de-duplication and improve performance of the data de-duplication has been proved to be an effective method.
In the multi-node data de-duplication solution, when querying a duplicate data block, each block needs to query all block records to confirm whether duplicate data exists. Therefore, the querying takes a long time when there is a large amount of data de-duplication. To improve the performance of data de-duplication, each block of a file with a relatively high similarity is compared with each block in a group that has a relatively high similarity with the file in order to perform data de-duplication in a group. In this way, only a block record in the group needs to be queried when a duplicate data block is queried, and the objective of improving the performance of the data de-duplication is achieved by compromising a limited de-duplication rate.
Although the multi-node data de-duplication based on group reduces data querying time, when a similarity analysis is performed on a file to determine a group, a fingerprint of each file needs to be queried and matched with fingerprints of all groups because a fingerprint of a file and fingerprints of all groups need to be queried and matched to determine a similarity. In addition, for the purpose of ensuring querying accuracy, when a similarity analysis is performed on each file, a file that saves a group fingerprint needs to be locked. As a result, multiple nodes cannot concurrently perform the matching querying, which is a performance bottleneck of multi-node data de-duplication of a group.

This patent is subject to a terminal disclaimer.

Similarity analysis method, apparatus, and system


An optical element comprises zones of different refraction power or curvature in order to achieve a desired pattern of brightness distribution. In Particular, a darker spot in a central illumination region of an illuminating arrangement including a cold light reflector lamp the light of which is supplied over a fiber optic light guide should be avoided. The optical element has a rotational symmetric annular zone of low or no curvature the outer ring diameter of which being larger and the inner ring diameter being smaller than the diameter of the light exit surface of the fiber optic light guide. A second rotational symmetric zone has a stronger curvature than the first zone. This second zone is situated within the inner ring diameter of the first zone. Both zones face the light exit surface of the light guide. A third zone has a curvature different to the curvatures of the first and second zones, and is situated at the light exit side of the optical element. This third zone is opposite both to the second zone in a rotational symmetric manner and, at least in part, to the first zone in radial relationship.

Lamps of the above-mentioned type comprise a reflector through the central region of which the lamp's base is passing. The glass bulb of the lamp itself has a seal stud at its end remote from the reflector. This constructive arrangement affects, however, the emission characteristic of the lamp in a disadvantageous manner in that the light intensity is significantly smaller in the central region than in regions situated more off-side the optical axis. If light emitted from such a reflector lamp is guided to an object to be illuminated, for example, through a fiber optical light guide, the same distribution characteristic of light intensity will, in principle, appear at the exit surface of this light guide. The maximum solid angle of light emission at light's exit from the light guide will be limited precisely by the numerical aperture of the glass fiber.
The solid or special angle of light emission of reflector lamps described above amounts about to 70.degree., and glass fiber light guides of 3 mm, 5 mm or 8 mm are used, for example, to guide the light to the object to be illuminated. The dark central spot occurring due to the constructive design of the reflector lamp described above will appear at the exit of the light guide the more significantly, the smaller the active diameter of the bundle of the fiber optic light guide is. This disadvantageous effect is utterly spoiling, for instance, when illuminating an object to be examined by microscope. Such a brightness distribution is especially disadvantageous with light guides comprising fiber bundles which run in common on the light entrance side, but are separated at the light exit side, i.e. which are then divided into a plurality of light guides.
In order to avoid a dark central spot, it has already been suggested to arrange the reflector lamp inclined with respect to the axis of the light entrance surface of the light guide. This, however, results merely in a size reduction of the dark central spot. Even with an inclined lamp, the above-mentioned dark spot appears, some times due to tolerances in manufacture of the reflector lamp. Moreover, a special drawback will occur with small active diameters of the light guide, even with an inclined lamp, in that the improvement in brightness distribution in the central region of an illuminated surface is rather insignificant.
Furthermore, it has been suggested to arrange a wedge-like optical element in the region between the reflector lamp and the entrance surface of the light guide in order to avoid a dark central spot. Although this known suggestions results in an improvement in brightness distribution in the central region, the average light intensity over the surface to be illuminated will be reduced significantly by such a measure. The reason for this disadvantageous effect is that, although some light intensity is transferred into the central angular region by wedge-like elements, light is also directed concurrently to the light guide in an angular region which lies outside its numerical aperture and, therefore, cannot be transferred any further.

Photonic Optische Gerate Ges.m.b.H. & Co. KG


A vertical tail unit (7) including an outer skin (13) in contact with an ambient air flow (21), wherein the outer skin (13) extends between a leading edge (23) and a trailing edge (25) with opposite lateral sides (27a, 27b), and surrounds an interior space (29), and wherein the outer skin (13) has a porous section at the leading edge (23), a pressure chamber (15) arranged in the interior space (29), wherein the pressure chamber (15) is fluidly connected to the porous section (31), an air inlet (17) provided in the outer skin (13) and fluidly connected to the pressure chamber (15), and an air outlet (19) provided in the outer skin (13) and fluidly connected to the pressure chamber (15).

The vertical tail unit comprises an outer skin, a pressure chamber, an air inlet, and an air outlet. The vertical tail unit may further comprises a plurality of structural parts, such as stiffeners, for supporting the outer skin from the inside. Further the vertical tail unit may include a vertical stabilizer and a rudder pivotally mounted to the vertical stabilizer.
The outer skin is in contact with an ambient air flow and extends between a leading edge facing the incoming flow, and a trailing edge. Further, the outer skin has two opposite lateral sides and surrounds an interior space. The outer skin further comprises a porous section in the area of the leading edge for letting air through the outer skin. The porous section might be formed e.g. as a perforated skin panel or as a skin panel made from a porous material.
The pressure chamber is arranged in the interior space for holding an overpressure or an under pressure with respect to the pressure of the ambient air flow in front of the porous section. The pressure chamber is fluidly connected to the porous section.
The air inlet is provided in the outer skin for letting in air from the ambient air flow. The air inlet is fluidly connected to the pressure chamber and configured to cause, in flight of the associated aircraft, an overpressure in the pressure chamber, such that air from the pressure chamber discharges through the porous section to the ambient air flow.
The air outlet is provided in the outer skin for letting out air into the ambient air flow. The air outlet is fluidly connected to the pressure chamber and configured to cause, in flight of the associated aircraft, an underpressure in the pressure chamber such that air from the ambient air flow is sucked in through the porous section into the pressure chamber. The air outlet may be formed as a movable flap that is or can be opened in the direction of the trailing edge.
Such vertical tail units are known in the art. The air inlet of the known vertical tail units is often formed as an air scoop that projects out of the outer skin into the ambient air flow. However, such an air scoop causes additional drag at the vertical tail unit and decreases efficiency.


A parking assist device assists entry or exit of a vehicle with respect to a parking space through an automatic control of a steering device. The parking assist device includes: a calculator that obtains, to perform the automatic control of the steering device, a path for allowing the vehicle to enter or exit the parking space based on the parking space, calculates an amount of temperature rise of the steering device at the execution of the automatic control for allowing the vehicle to move along the obtained path, and calculates a predicted temperature of the steering device at the execution of the automatic control by adding the calculated amount of temperature rise to a current value of the temperature of the steering device; and a controller that inhibits the execution of the automatic control when the predicted temperature is equal to or higher than a determination value.

The present invention relates to a parking assist device.
Vehicles like automobiles are known which are built with a parking assist device that assists parking of a vehicle to a predetermined parking space and starting of the vehicle from the parking space. According to such a parking assist device, operations to the steerable wheels of the vehicle for entry or exit of the vehicle with respect to a parking space are carried out through an automatic control of a steering device instead of a steering operation performed by the driver. The automatic control of the steering device enables operations to the steerable wheels for entry or exit of the vehicle with respect to a parking space without causing the driver to turn the steering wheel, and thus an assist to the entry or exit of the vehicle with respect to the parking space is realized.
Moreover, according to the parking assist device, when the entry or exit of the vehicle with respect to a parking space is assisted, operations to the steerable wheels are carried out through an automatic control of the steering device without causing the driver to turn the steering wheel, and thus the load to the steering device becomes large when executing the automatic control resulting in a tendency for the temperature of the steering device to increase. In order to avoid the occurrence of a failure of the steering device due to an excessive increase of the temperature thereof, for example, Japanese Laid-Open Patent Publication No. 2010-228591 discloses to inhibit the execution of the automatic control when the temperature of the steering device is out of a first temperature range in the automatic control.


An apparatus and a method of controlling a scanner include a transmission speed detector and a controller. The transmission speed detector detects data transmission speed information between a host and the scanner. The controller controls a scanning speed of the scanner based on a scan resolution and the detected transmission speed information.

1. Field of the Invention
The present general inventive concept relates to an apparatus and a method of controlling a scanner, and more particularly, to an apparatus and a method of controlling a scanner capable of controlling a scanning speed according to a data transmission speed between a host and the scanner.
2. Description of the Related Art
A conventional scanning process is performed at a constant scanning speed. The scanning speed is typically determined by a scan resolution and a speed of a scan motor. The speed of the scan motor is fixed to one value according to a specification of each product regardless of the transmission speed of a connection device, such as universal serial bus (USB), local area network (LAN), and IEEE1394.
In a case in which the scanning speed is faster than the transmission speed, that is, a speed at which scanned data is recorded in a memory is faster than a speed at which the recorded data is read and transmitted to a host, the data is gradually accumulated in the memory and a memory-full error is generated. As such, the scanning process stops and is in a stand-by state until the data accumulated in the memory is read and transmitted to the host and a space of the memory to be used in recording is secured. After the space in the memory is secured, when the scanning process restarts, the scanner regresses an interval required for a uniform rotation of the motor from the suspended position and performs the scanning process again. During this process, a jitter phenomenon in which the position where the scanning process has stopped and the position where the scanning process restarts are different from each other may be generated. That is, an input image, as illustrated in FIG. 1A, might be output as a defective image, as illustrated in FIG. 1B, due to the jitter phenomenon.
Also, in a case in which the scanning speed is set to an excessively low speed, a scanning process is performed at a low speed even though the transmission speed of the connection device, such as the USB, is fast, such that the host cannot obtain an output image from the scanner in time. Accordingly, the scanning process is not performed efficiently.

Apparatus and method of controlling scanner


System and method are provided for establishing internet protocol (IP) communication between a mobile node (MN) and one or more mobile networks. The method includes receiving (100) a request from a MN when the MN joins a first mobile network, creating (105) routing information indicating a home address of the MN, and announcing (110) the home address to the nodes of the mobile network(s). The request indicates the home address of the MN.

A mobile network is a network whose hosts and routers are usually static (e.g., non-mobile) with respect to each other, but are collectively mobile with respect to the rest of the Internet. For example, a mobile network might be found in an airplane, a ship, or a train. In general, a mobile router provides mobility (e.g., connection to the Internet Protocol (IP) infrastructure) for the nodes attached to the mobile router using, for example, mobile IP or network mobility (NEMO) protocols. A specific node in the mobile network is typically designated the mobile router and manages the mobility for all of the nodes within the mobile network, and thus a mobile network can change the point of attachment to the IP infrastructure while maintaining IP communication between nodes inside the mobile network and corresponding nodes connected to the Internet. When the mobile router moves from one IP subnet to another, the mobile router is typically required to handle mobility so as to maintain all of the communication of the nodes attached to the mobile router.
Mobile networks may take a variety of configurations such as a nested mobile network configuration where at least one first mobile network is attached under a second mobile network. For example, the first mobile network may be an individual carrying a device having an associated personal network, and the second mobile network may be a train having a mobile network infrastructure with connectivity to an IP network or infrastructure. When the individual enters the train, the mobile network of the individual can communicatively couple to an access point deployed in the train to operate within the mobile network of the train. Each mobile network has one or more local fixed nodes (LFNs) (e.g., a wireless device) that may be connected to the mobile router of the corresponding mobile network, such as by Ethernet or 802.11. The LFN has an IP address that belongs to the IP subnet(s) of the mobile network and has no specific IP mobility support. Each mobile network can also have one or more home mobile nodes (HMNs) that may be connected to the mobile router of the corresponding mobile network. An HMN is referred to herein as a mobile node (typically running Mobile IP protocol) having a home network that is the mobile network to which the HMN is attached. The HMN has a home address that belongs to the IP subnet(s) of the mobile network and has the same home agent (HA) as the HA of the mobile router of the corresponding home mobile network (i.e., the home agent of the HMN is not in the home mobile network of the HMN). Each mobile network can also have one or more visiting mobile nodes (VMNs) that may be connected to the mobile router of the corresponding mobile network. A VMN is referred to herein as a mobile node (typically running Mobile IP) attached to a mobile network that is not in the home network of the VMN. The VMN has a home address, and configures a temporary address, or care-of address, that belongs to the IP subnet(s) of the mobile network to which the VMN is attached. A Vehicular Area Network (VAN) having a mobile network deployed in a vehicle is an example of a mobile network in practice.
Prior to establishing IP communication with a destination node, the destination hostname is resolved into the IP address associated with the destination node, referred to as "name resolution", unless the IP address is previously known. One or more domain name system (DNS) servers may be used for a successful name resolution and typically involves a set of intermediate DNS servers having connectivity with one another to enable name resolution. For example, a mobile router has connectivity with a first DNS server, and the first DNS server has connectivity with a second DNS server that is authoritative for the destination node.
This name resolution is then used to establish IP communication. Mobile IP or NEMO protocols support routing between a node in one mobile network of a group of mobile networks with a node in another mobile network of the group of mobile networks using home agents to establish communication between the two nodes. A home agent is referred to herein as a node in the IP infrastructure that intercepts communication addressed to a particular LFN and re-directs the communication to the current location of the mobile router associated with the LFN.
Mobile nodes in an IP network are supported using a mobile IP protocol which allows a mobile node to change from one IP subnet to another while maintaining on-going communication. The mobile node has a permanent address, or a home address, that is used for communication. Each time the mobile node attaches to a new access point, a new temporary address, or care-of address, is assigned to the mobile node. The mobile node sends a binding between the home address and the care-of address to a server in the network, or a home agent. When a node in the Internet attempts to send a packet to the home address of a target mobile node, the packet is routed to the home network of the mobile node where the home agent intercepts the packet. The home agent, using the binding received from the mobile node, tunnels the packet to the care-of address of the mobile node so as to re-direct the packet to the current location of the mobile node. For example, standard node mobility protocols (e.g., mobile IP) and network mobility protocol (e.g., NEMO) utilize a bi-directional tunnel between the home agent and the mobile entity to maintain on-going communications as the mobile entity changes points of attachment to the IP infrastructure. This routing is complex, particularly for communications between a fixed node (e.g., a local fixed node (LFN)) in the mobile network and a visiting mobile node or between two visiting mobile nodes attached to the mobile network.
When establishing communication between an LFN and a visiting mobile node in a mobile network, the LFN sends the communication packet to the home address of the visiting mobile node. For example, the LFN sends the packet to a default router (e.g., a mobile router (MR1)), and the default router applies the mobile IP mechanism and tunnels the packet to a home agent (e.g., a mobile router home agent) in the mobile router home link. The mobile router home agent de-encapsulates the packet and sends the packet via the Internet to the home link of the visiting mobile node associated with the home address. At the home link of the visiting mobile node, the home agent of the visiting mobile node intercepts the packet when the visiting mobile node is not attached to the home link and tunnels the packet (first encapsulation) to the care-of address of the visiting mobile node. This care-of address of the visiting mobile node belongs to the mobile network. The tunneled packet from the home agent of the visiting mobile node is routed to the home link of the mobile router. The home agent of the mobile router intercepts the packet and tunnels the packet (second encapsulation) to the current location of the mobile router. The mobile router de-encapsulates the packet (e.g., removes the second encapsulation from the home agent of the mobile router) and sends the packet to the visiting mobile node. The visiting mobile node removes the remaining first encapsulation from its home agent and retrieves the initial packet sent by the LFN. This routing through the home agents located in the IP infrastructure places overhead on the radio interface between the mobile router and the IP infrastructure because packets to be routed between the visiting mobile node and the local fixed node will be sent twice over this interface (e.g., in the upstream and downstream directions). Additionally, overhead is introduced on the radio interface between the mobile router and the IP infrastructure because of the encapsulations used for routing the packets (e.g., bandwidth consumption).
In a conventional mobile network, while the mobile router having connectivity to the IP infrastructure maintains this connectivity, communication may be established between nodes of the mobile network (e.g., LFNs and VMNs) using conventional mobile IP. When the mobile router loses this connectivity, the mobile network is isolated and referred to as "autonomous". When the mobile network is in an autonomous mode, the mobile network is disconnected from the IP infrastructure and the corresponding home agent, and the home agents (e.g., of VMNs) are not reachable by the mobile router. Currently, conventional protocols, such as Mobile IP and NEMO, do not support the transmission of data packets (i.e., routing) between two nodes in an autonomous mode. In addition, the node initiating the communication may generally know the fully qualified domain name (FQDN) of the destination node but may not know the IP address of the destination node. With the loss of connectivity to the IP infrastructure, the nodes of the mobile networks (e.g., LFNs) lose access to DNS servers (e.g., default DNS servers, authoritative DNS servers, and intermediate DNS servers) that would otherwise be used for name resolution of the FQDN of the destination node (e.g., VMNs) into the IP address of the destination node.

System and method for routing and domain name system support of a mobile node


A slide track of a notched slider control can include a continuous path extending between two end points. A slider can be a graphical element able to move along the slide track. The end points can correspond to upper and lower extremes of a value range. Positions between the end points can represent values within the value range. The slide track can include at least one notch that is visually represented within the slide control as a deviation from a straight line segment having the two end points. The notch corresponds to a value within the value range. The notch can causes the slider to become fixed at a point along the slide track whenever a pointing device controlling the slider is moved in a direction of the deviation.

The present invention relates to the field of graphical user interface controls, more particularly to notched slider control for a graphical user interface.
Numerous graphic user interface (GUI) controls exist to facilitate human-computer interactions. A majority of these GUI controls have a physical world analog, which makes the controls more intuitive for a user. For example, GUI controls include buttons, dials, sliders, and the like. One problem with current slider controls is that users can easily overshoot an intended value, which requires them to perform minute corrections until the value is perfectly matched to the user's intentions. Another annoyance is that fact that the slider user must also look at the value indictor, typically displayed proximate to the slider and configured to dynamically adjust when the slider is moved, to ensure that the desired value is being approached.
Users of GUI sliders can become frustrated with this need to constantly manipulate the slider control to place a slider in a desired position. This is particularly the case when users have a set of values along the slider continuum that are recurrently desired. Often desired values include a zero value, a middle value, and high value, although many others can be desired. Further, specific projects, applications, and/or users can have characteristic slider settings that are recurrently desired for that project, application, and/or user.
Some slider control variations exist that somewhat address the overshooting problem, each of which have a number of drawbacks. One variation is to create a slider with a middle-point that absorbs more "slide" than other positions, which allows users to more easily select a middle position, which is an often desired value. This biasing of a value along a slider continuum can be confusing to users and can be frustrating when a point proximate to the center, yet which is not the center, is desired. Biasing a slider often results in slider controls performing in a manner that is non-intuitive from a user perspective.
Another variation imposes graduations on a slider control, and slide actions of the controller snap to these preset graduations. This solution sacrifices one of the beneficial aspects of a slider control (an ability to use the slide position to reflect a user selectable point along a continuum) in order to simplify a selection of graduation associated values.
Still another solution permits a slider of a slider control to be clicked or double-clicked, which automatically adjusts the slider to a preset position. This solution suffers from being non-intuitive for many users and by being localized to a single value. Adding a "cycle" of presets achieved by clicking can increase a number of presets that can be navigated to, yet can also dramatically increase user confusion.

Notched slider control for a graphical user interface

International Business Machines Corporation


A method and apparatus for a navigation system for inputting a street name allows a user to quickly find a desired street name even when the user inputs an incorrect prefix or suffix of the street name. The navigation system accepts a street name based on either a base name input method or a full name input method in the same input field while performing a street name search mostly based on a base name of the street name. The method and apparatus hints an inexperienced user the distinction between the base name and other components of the street name so that the user knows that non-base name element is unnecessary for finding the correct address of the destination.

A navigation system performs travel guidance for enabling a user to easily and quickly reach the selected destination. A typical example is a vehicle navigation system where a vehicle is equipped with a navigation function to guide a driver to a destination through a calculated route. Typically, the navigation system displays a map image while superimposing thereon a mark representing the current location and notifies the user which direction to turn at an intersection to reach the destination.
FIGS. 1A-1H show an example of overall procedure and screen display involved in the navigation system for route guidance to a destination. FIG. 1A is an example of locator map screen of the navigation system when the destination is not specified. Typically, the navigation system displays a street on which the vehicle (current vehicle position VP) is running on a map image and a name of the street. Other information such as a north pointer NP, a map scale and a current time may also be illustrated on the display screen.
FIGS. 1B-1F show an example of process for specifying a destination in the navigation system. When selecting a "Destination" menu on a main menu screen of FIG. 1B, the navigation system displays an "Find Destination By" screen as shown in FIG. 1C for specifying an input method for selecting the destination. The "Find Destination By" screen lists various methods for selecting the destination. This example includes the methods of "Address", "Intersection", "Point of Interest", "Recent Route", "Address Book", and "Today's Plan" which are well known in the art.
When selecting, for example, the "Address" method in FIG. 1C, the navigation system displays a keyboard screen in FIG. 1D. Suppose the user inputs an address "2525 W. Carson" of a shopping mall, the navigation system displays a "Confirm Destination" screen such as shown in FIG. 1E. If the name and address on the screen shows the correct destination, the user enters an "OK to Proceed" key to proceed to the next procedure.
In FIG. 1F, the navigation system calculates and determines a route to the destination, i.e., the address "2525 W. Carson". The navigation system determines a route to the destination based on, for example, the quickest way to reach the destination, the route using many free ways as possible or the route without using toll road, and the like. In this example, the navigation system displays a progress scale of the calculation of the route.
Once the route to the destination is determined, the navigation system starts the route guidance as shown in FIG. 1G for guiding the user to the destination. Typically, the navigation system shows the intersection which is highlighted to show the next turn and a direction of the turn. Such route guidance by the navigation system is also accompanied by voice instructions.
The navigation system measures a distance to the current destination to detect whether the destination is within a predetermined range such as within 100 meters. When the user approaches the destination within such a predetermined distance, the navigation system indicates that the destination "Dest" is ahead as shown in FIG. 1H. The navigation system detects the actual arrival based on various factors such as the engine stop.
As noted above, a process of entering a street name (address) is frequently needed when setting a destination in the navigation system. FIGS. 2A-2D are display examples of input screens that accept the user's input of a street name through a full name method. As shown, through a key board screen of FIG. 2A, the user has entered "N MAIN" in the street name field. As a result, the navigation system lists the matching entries that match the user's input "N MAIN" as shown in FIG. 2B. The list shows matching entries "N MAIN", "N MAINE", "N MAINLAND", "N MAINWARD" all of which include "N MAIN". Since the user has input the prefix "N", all of the entries start with a letter "N".
Suppose the user really wants to find an address "10605 E MAIN AVE" in the city of "PALM SPRINGS" but has selected the entry "N MAIN" in FIG. 2B because it was not certain whether it starts "N" or "E". Then, the user enters a house number "10605" for the street name "N MAIN" as shown in FIG. 2C. As a result, two matching entries located in "GARDEN GLOVE" and "SANTA ANA". However, since the prefix "N" is incorrect, the address "10605 E MAIN AVE" in "PALM SPRINGS" intended by the user has not come out by this search.
Although the full name search is direct and intuitive for an inexperienced user, it requires an accurate input of a prefix, suffix and base name. There is another method for inputting a street name in which a user can input a base name of the street without including a prefix or a suffix of the street name. The base name input method is convenient since the user can only remember the base name accurately. However, because it is not direct and intuitive, inexperienced users sometimes do not know such an input method. Therefore, there is a need of a more flexible way of searching a street name for a navigation system which allows both the full name input method and base name input method in the same input field on the screen.

Method and apparatus for entering street name


A level-crossing based circuit with offset voltage cancellation includes a first level-crossing detector section having a first output capacitor, wherein the first level-crossing section is configured to charge the first output capacitor to a first voltage corresponding to an input signal. The first voltage includes a first offset voltage. The level-crossing based circuit also includes a second level-crossing detector section having a second output capacitor, wherein the second level-crossing detector section is configured to discharge the second output capacitor to a second voltage corresponding to the input signal such that the second voltage includes a second offset voltage that is substantially equal in magnitude to the first offset voltage and opposite in polarity to the first offset voltage relative to an average of the first voltage and the second voltage.

Conventionally, an operational amplifier (op-amp) is used for accurate charge transfer in switched capacitor circuits. The op-amp has differential inputs and an output. In operation, the op-amp senses the voltage difference at the differential inputs, and drives an output node continuously with its output to achieve a virtual ground condition at the differential inputs. As long as high-gain and wide-bandwidth op-amps are available this method is accurate and robust. However, it is relatively difficult to make power efficient, high-gain, and wide-bandwidth op-amps in nano-scale complementary metal oxide semiconductor (CMOS) technology.
A related art zero-crossing based circuit technique replaces the op-amp with a zero-crossing detector and a current source. FIG. 1A is a schematic of an exemplary related art zero-crossing detector based circuit 10. The circuit 10 includes a zero-crossing detector ZCD1 and a current source 12 that supplies a current I1. The circuit 10 also includes switches S1, S2, S3, S4, S5, S6, and S7 along with capacitors C1, C2, and C3. A switch controller (not shown) controls the opening and closing of switches S1 through S6. The switch S7 is opened and closed in response to the output state of the zero-crossing detector ZCD1.
FIG. 1B is a series of graphs depicting switching phases φ1, φ2, φp, and φ2e, along with an output voltage Vout, and a virtual ground node voltage VX for the circuit 10 (FIG. 1A). In operation, the circuit 10 samples an input signal (Vin) that is applied to the capacitor C1 by closing switches S1 and S3. Switches S2 and S4 are open during this sampling period. This sampling step is similar to a sampling step used in the operation of traditional op-amp based switched capacitor circuits. Optionally, the switch S6 may be closed during the sampling period to ensure that the output voltage Vout is at ground potential (GND). After the sampling period, the capacitor C1 and the second capacitor C2 are selectably coupled to a virtual ground node 14 that includes an input of the zero-crossing detector ZCD1 by closing the switches S2 and S4. As a result, a charge accumulated by the capacitor C1 from Vin is transferred to the second capacitor C2. The charge transfer starts in response to the preset phase φp, which initializes the output voltage Vout to GND via the switch S6. After the preset phase φp concludes, the current source 12 begins charging the output capacitor C3 with the current I1. The zero-crossing detector ZCD1 monitors the virtual ground node 14, while the current I1 is charging the output capacitor C3. The zero-crossing detector ZCD1 detects the instant of the zero-crossing of the virtual ground node voltage Vx, and opens the switch S7 coupled to a terminal node 16. As such, the voltage Vout and the virtual ground node voltage Vx increase linearly in time and stop once a virtual ground condition Vcm (common mode voltage) is met. The circuit 10 is power efficient because the zero-crossing detector ZCD1 is significantly more power efficient than a high-gain wide-bandwidth op-amp typically used in circuits having the same function as circuit 10. Moreover, the circuit 10 eliminates stability issues, due to semi-open loop operation.
FIG. 2A is a schematic of a modified version of the related art zero-crossing detector circuit 10 of FIG. 1A having a reversed direction current source. For example, charge transfer operations of zero-crossing based circuits can be done in an opposite direction of the current source 10 shown in FIG. 1A. A configuration for an opposite output charging direction is depicted in FIG. 2A. In this case, the output voltage Vout is initialized to a supply voltage VDD instead of GND during the preset phase (φp) by the preset switch S6. In this configuration the preset switch S6 is coupled between a source of the supply voltage VDD and a node between the switch S5 and a first terminal of the capacitor C3. In operation, the zero-crossing detector ZCD1 monitors the zero crossing at the input of the zero-crossing detector ZCD1 while the current source 12 discharges the output voltage Vout via the current I1. FIG. 2B shows a similar operation to FIG. 1B except that the direction of change for the output voltage Vout and the virtual ground node voltage Vx are reversed with those shown in FIG. 1B. It may be necessary to switch the two inputs of the zero-crossing detector ZCD1 in cases where a same version of zero-crossing detector ZCD1 is reused in circuit 10 of FIG. 2A.
For the following discussion FIG. 3A is a schematic reproduction of the related art zero-crossing detector circuit of FIG. 1. The operation aspects of the circuit 10 of FIG. 1A and FIG. 2A are described as ideal operations. In practical terms, the response time between a zero-crossing detection and an output change of the zero-crossing detector 10 and a corresponding switch transition for the switch S7 is not instantaneous. Instead, there is a finite delay (td) in turning off the switch S7 after a zero-crossing at the input of the zero-crossing detector ZCD1.
FIG. 3B is a series of graphs depicting switching phases φ1, φ2, φp, and φ2e, along with an output voltage Vout, a virtual ground node voltage VX and an offset caused by the delay td of the circuit 10 of FIG. 3A. Due to the delay td, the virtual ground node voltage Vx overshoots an ideal voltage by a voltage overshoot Vxover given by
                                          V            xover                    =                                    t              d                        ·                                          ⅆ                                  V                  x                                                            ⅆ                t                                                    ,                            (        1        )            which leads to an offset voltage Voffset in the output voltage Vout. The voltage offset Voffset is given by
                              V          offset                =                              t            d                    ·                                                    ⅆ                Vout                                            ⅆ                t                                      .                                              (        2        )            Therefore, a final output voltage Vfinal is given byVfinal=Vsignal+Voffset,  (3)wherein the signal voltage Vsignal equals an ideal output voltage.
Related art techniques address the inaccuracy due to the virtual ground node voltage Vx overshoot and the resulting offset voltage Voffset by implementing a coarse charge transfer phase followed by a fine charge transfer phase. The goal of this dual phase charge transfer is to reduce the ramp rate right before the zero-crossing detector ZCD1 detects a zero-crossing. In this regard and as shown in FIG. 4A, one related art modification adds a current control 18.
As shown in FIG. 4B, a charge transfer begins with a coarse charging phase (φc), wherein the current source 12 in response to the current control 18 charges the capacitor C3 relatively quickly via a relatively high current ramp of the current I1. While the current source 12 charges the capacitor C3, the zero-crossing detector ZCD1 detects when the virtual ground node voltage Vx reaches a predetermined voltage level Vclose that is relatively close to the virtual ground condition Vcm. Once the virtual ground node voltage Vx reaches the predetermined voltage level Vclose, the current control 18 significantly reduces the relatively high current ramp of the current I1, thereby replacing the coarse charging phase φc with a fine charging phase φf. The resulting offset voltage Voffset is relatively lower, which results in a relatively greater accuracy of charge transfer to the capacitor C2 and higher accuracy of the output voltage on the capacitor C3. The fine charging phase ends when the virtual ground node voltage Vx crosses the virtual ground condition Vcm. The charge accumulated by the capacitor C3 is sampled by a following sampling stage (not shown). While the modified circuit 10 of FIG. 4 provides improved accuracy, a small but significant error in the output voltage Vout remains due to the smaller but still significant offset voltage Voffset.
Additional related art techniques to compensate for the offset voltage Voffset include employing a programmable offset in the zero-crossing detector ZCD1 and/or coupling a small capacitor array (not shown) to the virtual ground node 14 to cancel the offset voltage Voffset. However, after considering temperature and process variations, these related art techniques are relatively impractical. What is needed is a level-crossing based circuit and method that provides cancellation of offset voltage and tracking of the offset variation with efficiencies in power consumption and die area.

Level-crossing based circuit and method with offset voltage cancellation


An extinguisher identifying device. The device comprises an extinguisher identifying device and an activation device, and a communications module. The extinguisher identifying device reads an identifying element on an extinguisher to obtain identifying data of the extinguisher. The activation device is configured to activate the extinguisher identifying device in response to a predetermined event. The communications module is configured to transmit the extinguisher identifying data to a management device.

Extinguishers for extinguishing fires are known. A pressure extinguisher, for example, comprises a tank of extinguishing agent designed to put out the fire, and a pressurized gas cartridge. To use the extinguisher, the user must pierce the cartridge to release the gas in the tank, and then operate a handle of the extinguisher to expel the extinguishing agent.
The extinguishers, which are very important for the safety of goods and people, are positioned in premises at suitable positions. The extinguishers are placed on mounts and, to prevent them being removed for reasons other than those originally intended, they are sometimes attached to the mount by means of a wire, sometimes sealed, which can be broken in cases of real need.
However, and for various reasons, extinguishers can be removed from their mount, which is a serious handicap in terms of safety.
In addition, depending on the nature of the risk to be controlled, these extinguishers can be of various types, for example and in a non-limiting way, they can contain different extinguishing agents such as water, foam, various types of powder, gases such as carbon dioxide, halon, etc. During verification operations, or even in the event of work in the premises, an extinguisher of a given type can be put back incorrectly in a location for which another type of extinguisher was intended. In order to limit this risk, specific mounts have been envisaged in order to prevent a given type of extinguisher from being mechanically received on a mount intended for another type of extinguisher.
Finally, it is desirable, in certain cases, to be able to check the condition of extinguishers in real time. This assumes the installation of an electrical network linked to each extinguisher mount, which is feasible in new premises but very difficult to implement in old premises, taking into account the fact that electrical cables have to be passed to locations where this is sometimes difficult and the very high cost such installations entail.

Extinguisher identifying method and extinguisher identifying device


Systems and methods for detecting spoofed or illegitimate GNSS signals. A processor receives GNSS data and processes this data to extract acceleration, angular velocity, and height or altitude variation data. For the same time period, sensor data from IMU (inertial measurement unit) sensors and from a barometer are received by the processor. From the sensor data, the processor extracts similar acceleration, angular velocity, and height variation data. These two sets of data are then correlated and correlation coefficients are calculated. These correlation coefficients are then used to calculate a decision statistic. The decision statistic is compared with a predetermined value and, if the decision statistic is below a predetermined value, then the GNSS data is considered to be illegitimate or spoofed.

The use of GNSS to monitor and regulate vehicular use is widespread. Systems and services such as fleet-management, asset-tracking, pay-as-you-drive insurance use GNSS as the primary, if not the only, positioning sensor. For many of these applications, especially applications where end users are charged or penalized based on their location, there is a strong incentive to compromise on GNSS sensor quality due cost. As expected, when such monitoring devices are installed in vehicles, they are typically enclosed in tamper-proof housings and attached to the vehicle in a manner that either prevents tampering or, if tampering has been done, such tampering is evident. However, it is unavoidable that the GNSS antenna and the device power supply are exposed. In some cases, other vehicle based sensors are also made available to the device via the controller area network (CAN) bus, including, for example, wheel-tick counters and wheel angle. Because the CAN bus offers no authenticity or security features, inputs delivered to the devices are equally vulnerable to spoofing as GNSS is.
Because of the above vulnerabilities, a malicious user may choose to tamper or otherwise interfere with both the GNSS and the vehicle base sensors in an effort to manipulate the estimate of the vehicle's position. Although the tamper-proof enclosure might protect the measurement device itself, any measurements, either RF-based (delivered through an antenna) or digital measurements (delivered through the CAN bus), are exposed.
One should also consider a self-GNSS spoofing scenario where the holder of a vehicle mounted GNSS receiver deliberately shields the antenna from genuine GNSS signals to avoid having the vehicle be tracked by a third party. In addition to shielding the antenna, the GNSS receiver may also introduce counterfeit or illegitimate GNSS signals to cause the receiver to compute incorrect position, velocity and time (PVT) solutions. Detection of such an attack is important as this type of attack is easy to implement and cannot be detected by an unaided single antenna GNSS receiver. Unfortunately, counterfeit or illegitimate signals can be introduced either before the receiver is switched on or before the vehicle leaves a GNSS denied area (e.g. a covered parking lot). When counterfeit or illegitimate signals are introduced in this manner, spoofing detection mechanisms that monitor signal transients, received power variations, or distortions in the received GNSS signals, are likely to be ineffective.
From the above, there is therefore a need for systems and methods which allow for the detection of illegitimate or spoofed GNSS readings.

Detection of spoofed GNSS signals using IMU and barometer sensors


An input filter for a vehicle anti-lock brake system control module has a filter section for attenuating input signals having mid-range frequencies.

This invention relates in general to anti-lock brake systems (ABS) and in particular to a wheel speed signal input filter for an ABS control module which includes a filter section for attenuating mid-frequency input signals.
An Anti-lock Brake System (ABS) is often included as standard or optional equipment on new vehicles. When actuated, the ABS is operative to control the operation of some or all of the vehicle wheel brakes. A typical ABS includes a plurality of solenoid valves mounted within a control valve body and connected to the vehicle hydraulic brake system. Usually, a separate hydraulic source, such as a motor driven pump, is included in the ABS for reapplying hydraulic pressure to the controlled wheel brakes during an ABS braking cycle. The pump is typically included within the control valve body while the pump motor is mounted upon the exterior of the control valve body.
An ABS further includes an electronic control module which has a microprocessor. The control module is electrically coupled to the pump motor, a plurality of solenoid coils associated with the solenoid valves and wheel speed sensors for monitoring the speed and deceleration of the controlled wheels. The control module is typically mounted upon the control valve body to form a compact unit which is often referred to as an ABS electro-hydraulic control unit.
During vehicle operation, the microprocessor in the ABS control module continuously receives speed signals from the wheel speed sensors. The microprocessor monitors the speed signals for potential wheel lock-up conditions. When the vehicle brakes are applied and the microprocessor senses an impending wheel lock-up condition, the microprocessor is operative to actuate the pump motor and selectively operate the solenoid valves in the control unit to cyclically relieve and reapply hydraulic pressure to the controlled wheel brakes. The hydraulic pressure applied to the controlled wheel brakes is adjusted by the operation of the solenoid valves to limit wheel slippage to a safe level while continuing to produce adequate brake torque to decelerate the vehicle as desired by the driver.
The wheel speed sensors are usually passive variable reluctance magnetic transducers which sense the rotation of an associated exciter ring. The exciter ring is formed from a ferrous metal and has a plurality of teeth formed about the circumference thereof. The exciter ring is typically mounted directly upon a vehicle wheel hub or on the ring gear in a differential assembly.
Passive variable reluctance transducers typically include a permanent magnet which is adjacent to an end of a pole piece formed from a ferrous material. The opposite end of the pole piece is shaped as a tip which extends from the transducer and is spaced apart from the exciter ring teeth by an air gap. A coil surrounds the pole piece and is electrically connected to the ABS control module. Examples of prior art variable reluctance magnetic transducers are shown in U.S. Patent Nos. 5,032,790 and 5,486,758.
During operation, a magnetic field extends from the end of the permanent magnet which is adjacent to the pole piece. The magnetic field passes through the pole piece and across the air gap to the exciter ring. A return path for the magnetic field extends through the exciter ring and the air to the end of the permanent magnet which is opposite from the pole piece. As an exciter ring tooth approaches the pole piece tip, the reluctance of the magnetic field decreases, causing the strength of the magnetic field to increase. Conversely, as the exciter ring tooth moves away from the pole piece tip, the reluctance of the magnetic field increases, causing the strength of the magnetic field to decrease. As the magnetic field strength increases, a first voltage is induced across the coil. Similarly, as the magnetic field strength decreases, a second voltage, having a polarity which is opposite to the polarity of the first voltage, is induced across the coil. The passage of successive exciter ring teeth past the pole piece tip induces an alternating voltage across the coil.
The induced alternating voltage is generally proportional to the rate of change of the magnetic field. More specifically, as the exciter ring speed increases, the magnitude of the alternating voltage also increases. Additionally, the frequency of the alternating voltage is directly proportional to the number of exciter ring teeth which pass the tip of the pole piece per unit of time. As described above, the rotation of the exciter ring is associated with the controlled vehicle wheel. Thus, the frequency of the induced alternating voltage is proportional to the rotational speed of the vehicle wheel associated with the exciter ring.
The voltage induced across the transducer coil is supplied to the ABS control module as a speed signal for the controlled vehicle wheel. As described above, the microprocessor in the control module monitors the frequency of the speed signal and is responsive thereto to determine the speed and acceleration of the associated vehicle wheel. However, the speed signal typically includes high frequency noise components which are induced upon the coil by electromagnetic radiation from other vehicle components. These noise components are incompatible with the proper operation of the control module electronics and could adversely affect the operation of the electronics by introducing unwanted signals which could be misinterpreted as speed signals. Accordingly, an ABS input filter circuit which attenuates the components of noise having frequencies above the highest frequencies of interest for ABS operation is usually included between the variable reluctance transducer and the control module electronics.

ABS speed signal input filter having mid-frequency attenuation


A graphite heater and method of forming a graphite heater comprising a graphite body configured to form an electrical heating circuit for at least one heating zone through the graphite encapsulated in a continuous overcoat layer comprising at least one of a nitride, carbide, carbonitride or oxynitride of elements selected from a group consisting of B, Al, Si, Ga, refractory hard metals, transition metals, and rare earth metals, or complexes and/or combinations thereof, wherein prior to being configured to form the electrical heating circuit path, the graphite body is coated with a layer comprising at least one of a nitride, carbide, carbonitride or oxynitride of elements selected from a group consisting of B, Al, Si, Ga, refractory hard metals, transition metals, and rare earth metals, or complexes and/or combinations thereof.

In the fabrication of a semiconductor device or semiconductor material, a semiconductor wafer is processed in an enclosure defining a reaction chamber at a relatively high temperature above 1000° C., with the wafer being placed adjacent to or in contact with a resistive heater coupled to a power source. In this process, the temperature of the semiconductor wafer is held substantially constant and uniform, varying in the range of about 1° C. to 10° C.
U.S. Pat. No. 5,343,022 discloses a heating unit for use in a semiconductor wafer processing process, comprising a heating element of pyrolytic graphite ("PG") superimposed on a pyrolytic boron nitride base. The graphite layer is machined into a spiral or serpentine configuration defining the area to be heated, with two ends connected to a source of external power. The entire heating assembly is then coated with a pyrolytic boron nitride ("pBN") layer. U.S. Patent No. 2002-06-25 discloses a heating element, wafer carrier, or electrostatic chuck comprising a PG element mounted on a pBN substrate, with the entire assembly being subsequently CVD coated with an outer coating of AlN to protect the assembly from chemical attacks.
Although graphite is a refractory material which is economical and temperature resistant, graphite is corroded by some of the wafer processing chemical environments, and it is prone to particle and dust generation. Due to the discontinuous surface of a conventionally machined graphite heater, the power density varies dramatically across the area to be heated. Moreover, a graphite body particularly after machining into a serpentine geometry is fragile and its mechanical integrity is poor. Accordingly, even with a relatively large cross sectional thickness, e.g., above about 0.1 inches as typical for semiconductor graphite heater applications, The heater is still extremely weak and must be handled with care. Furthermore, a graphite heater changes dimension over time due to annealing which induces bowing or misalignment, resulting in an electrical short circuit. Additionally, it is also conventional in semiconductor wafer processing to deposit a film on the semiconductor which may be electrically conductive, such films may deposit as fugitive coatings on The heater which can also contribute to an electrical short circuit, a change in electrical properties, or induce additional bowing and distortion.
Applicants have discovered an improved heater formed from a solid graphite body, which overcomes the inherent fragility and weakness of the prior art heater while still retaining inherent refractory properties of graphite.

Encapsulated graphite heater and process


A differential amplifier includes a pair of first and second transistors connected together. The first transistor includes a first conduction terminal, and a second conduction terminal forming a first output node of the amplifier. The second transistor includes a first conduction terminal connected to the first conduction terminal of the first transistor, and a second conduction terminal forming a second output node of the amplifier. A first inductive load is connected between the first output node and a supply reference, and a second inductive load is connected between the second output node and the supply reference. A limiting network limits output voltages on the first and second output nodes, and includes a first resistor connected to the first output node, and a second resistor connected to the second output node. The first and second resistors also connected together in series at a common intermediate node. A first analog switch is connected to the first resistor and to the supply reference, and a second analog switch connected to the second resistor and the supply reference via the first analog switch. The first and second analog switches are turned on when a voltage on the common intermediate node exceeds a threshold.

An architecture of a differential amplifier commonly used in modern electronic circuits is the so-called open collector architecture. In particular, the open collector architecture is used when an amplifier is required to exchange current with external components. In this situation, the collectors of the differential pair of transistors are coupled to connection pins. The amplifier is supplied by connecting these pins to a DC line through external components.
A typical connection scheme of an open collector differential pair of transistors Q1, Q2 to a supply line by external inductive loads L1, L2 is depicted in FIG. 1. These inductive loads L1, L2 are used for doubling the output voltage swing, which becomes twice the supply voltage.
In the shown example, the AC components of the voltages on the output nodes C1 and C2 are floating, and also depend on the signals applied on the control nodes of the transistors of the differential pair Q1, Q2. This is while the respective DC components are set by the value of the supply voltage VDD.
In these amplifiers it may happen that the voltages on the nodes C1, C2 reach values larger than those established by the technology used. This is due to the spurious common mode, or differential voltages due to inductive loads connected to the nodes C1, C2 and the supply line VDD.
In particular, when the differential amplifier is turned on or off, there are spurious voltage peaks of equal amplitude. The voltage peaks are in phase between them (common mode) on both nodes C1, C2 due to the rapid variations of the current flowing in the inductances L1, L2 superposed to the output voltages, thus raising their level.
To address this problem, several architectures of open collector differential pairs having networks for limiting the output voltage are known. A first example is shown in FIG. 2, wherein each collector node is connected to the supply line by a certain number N of diodes D1, . . . , Dn connected in series. Since VT is the voltage for turning on the diodes (typically 0.8V), when the voltages on the collector nodes C1 and C2 reach the threshold value given byVDD+N*VTthe N diodes D1, . . . , Dn are turned on and limit the maximum voltage on the nodes C1 and C2. This approach effectively limits the voltages on the output nodes both in presence of spurious common mode and differential over voltages.
A drawback of this architecture is that the series of diodes do not turn on instantaneously, but gradually enter in a conduction state. Thus, they distort the output signal when its amplitude is smaller than the threshold value N*VT. Using this limitation network causes a relevant harmonic distortion.
Another architecture is shown in FIG. 3. In this case, the collector nodes are connected by two diode series D1, . . . , Dn connected in parallel. When the differential voltage on the nodes C1 and C2 equals the product N*VT, one of the two series of diodes enters in a conduction state and limits this differential voltage.
The effect of the protection network shown in FIG. 3 is that of reducing the output differential voltage when this voltage reaches too a large value. The protection network does not intervene when at the turning on or off of the device there are spurious common mode voltage peaks on the nodes C1 and C2. Moreover, even in this case, the diodes cause an increase of the harmonic distortion of the output signal.

Differential amplifier with limitation of high common mode output voltages