A wind deflector for a sunroof. The wind deflector includes a first electrode proximate an edge of a sunroof opening. A second electrode is proximate the first electrode. An insulator is intermediate the first and second electrodes. A power supply is coupled to the first and second electrodes. An electrical signal generated by the power supply is coupled to the first and second electrodes. An electric field generated between the electrodes due to the electrical signal ionizes air proximate the electrodes, thereby deflecting an air flow field away from the sunroof opening.

A sunroof is a popular accessory item for a motor vehicle. The sunroof typically includes a closure panel that is movable relative to an opening formed through a roof of the vehicle. The closure panel is movable between a closed position generally flush with the roof that completely covers the opening, and an open position rearward of the closed position for allowing ventilation and/or sunlight to enter a passenger compartment of the vehicle. In addition, the closure panel is typically positionable at one of a plurality of partially open positions between the closed and open positions. The closure panel, which may be moved manually or via an electric motor, is retracted into the structure of the roof when open, where it is hidden from view.
Many sunroof assemblies have a shortcoming wherein noise is transmitted into the vehicle passenger compartment when the closure panel is in an open position, due to the flow of air proximate the sunroof opening. One component of the wind noise, commonly termed "buffeting" or "wind throb," results from air rushing over an opening in a cavity and is generally characterized by relatively low-frequency noise. Another component of wind noise typically results from air rushing by or into obstructions at higher vehicle speeds and is generally characterized by high-frequency noise.
In an effort to attenuate the low frequency portion of the wind noise, wind deflectors have frequently been incorporated into sunroof assemblies. These wind deflectors may be fixed, or may pivot from a retracted position when the sunroof closure panel is closed to an extended position when the sunroof closure panel is opened. While fixed wind deflectors have been employed with some success, there is an inherent tradeoff in configuring a wind deflector to be effective in attenuating low-frequency noise while not generating high-frequency noise. For example, although the wind deflector can be configured to deflect the air beyond the sunroof opening to decrease wind throb, at higher vehicle speeds the deflector itself can generate high-frequency wind noise.
Pivotable wind deflectors that are manually or automatically adjustable are more effective to attenuate low- and high-frequency wind noise. Examples of such wind deflectors may be seen by reference to U.S. Pat. Nos. 6,523,889 and 6,174,025. However, such mechanical systems are complex, thereby increasing their cost and reducing their overall reliability. There is a need for a wind deflector that is effective to attenuate low- and high-frequency wind noise without resorting to complex mechanical systems.


A method for manufacturing a semiconductor device comprising a first conductivity type semiconductor substrate, a memory cell region and a peripheral circuit region formed on the substrate, the memory cell region and the peripheral circuit region having a plurality of second conductivity type transistors (Tr's), which comprises the steps of:(i-a) forming source/drain regions in a memory cell formation region (MCFR) on the substrate by implanting a second conductivity type impurity, (ii-a) forming a gate insulating film and gate electrodes on the MCFG and a peripheral circuit formation region (PCFR) of the substrate, thereby providing the plurality of second conductivity type Tr's in the MCFG, (iii-a) implanting a first conductivity type impurity into the entire surface of the substrate with the gate electrodes used as a mask to form a device isolation region in the MCFG and, at the same time, to allow the first conductivity type impurity to be contained to a predetermined depth in a second conductivity type Tr formation region of the PCFR, and (iv-a) forming a mask pattern on the MCFR for masking the MCFR, and implanting a second conductivity type impurity to a predetermined depth into the second conductivity type Tr formation region of the PCFR with the mask pattern and the gate electrode used as a mask to form second conductivity type source/drain regions.

1. Field of the Invention
The present invention relates to a method for manufacturing a semiconductor device, and more particularly to a method for manufacturing a Read Only Memory (ROM) of a flat cell type.
2. Description of the Related Arts
Conventional masked ROMs for writing and utilizing diverse program information have a structure which uses a flat cell configuration in memory cell regions for an improvement of integration and which includes a LOCOS oxide film formed for device isolation in peripheral circuit regions. Generally, an N-MOS transistor is used as a memory cell transistor, and a complementary MOS transistor comprising an N-MOS transistor and a P-MOS transistor is used as a peripheral circuit.
In memory cell regions of a flat cell type, a plurality of source/drain regions 1 serving as bit lines for memory cell transistors are disposed parallel to each other with predetermined spacing, and a plurality of gate electrodes 2 serving as word lines are disposed parallel to each other with predetermined spacing on a gate insulation film so that the gate electrodes intersect the source/drain regions 1, as shown in FIG. 5. Afterwards, impurities of the same conductivity type as in the semiconductor substrate are introduced for device isolation between the memory cell transistors with the gate electrodes used as a mask, thus forming device isolation regions 3. FIG. 5 is a plan view of memory cell transistors of a flat cell type.
However, difficulties arise in forming peripheral circuit regions by applying a flat cell configuration similar to the one for the memory cell regions. Since flat cell structures involve formation of source/drain regions prior to forming gate electrodes, it is not possible, in peripheral circuit regions having a different transistor layout, to form source/drain regions in a self-aligned configuration with respect to gate electrodes of the transistor, so that an alignment shift must be considered, leading to increased LSI chip size.
When the LSI is constructed for 5 V power supply, although device isolation of the above flat cell configuration is possible in the memory cell regions by reducing the driving voltage, a LOCOS method of selective oxidation technique is used for device isolation so as to prevent decrease in field inversion voltage between the elements in the peripheral circuit.
The conventional method of manufacturing a masked ROM of a flat cell type will be explained hereinbelow with use of FIGS. 6(a) to 6(d), which are diagrams illustrating the conventional steps in manufacturing a masked ROM of a flat cell type.
As shown in FIG. 6(a), an n-type impurity well region 5 and a p-type impurity well region 6 are first formed on a p-type silicon substrate 4 (having a concentration of about 5.times.10.sup.14 to 2.times.10.sup.15 cm.sup.-3). Here, surface concentration of the two well regions is about 10.sup.17 cm.sup.-3.
Then, a LOCOS oxide film 7 is formed for device isolation in the peripheral circuit region and, after an ion implantation (in adose of about 10.sup.12 to 10.sup.13 cm.sup.-2) for threshold voltage control, an n-type impurity is implanted at a high concentration to form source/drain regions 1 in the memory cell region.
Next, a gate electrode 2 for the memory cell region and a gate electrode 10 for the peripheral circuit region are formed, as shown in FIG. 6(b). Here, the gate electrodes 2 and 10 are wired in a polycide structure, in which the symbols 2a and 10a represent films of a high melting point metal or a silicide thereof whereas the symbols 2b and 10b represent polysilicon films.
Further, with the peripheral circuit regions masked by a resist pattern 9, a p-type impurity for device isolation is implanted into the memory cell regions in a dose of about 10.sup.13 to 10.sup.14 cm.sup.-2. Although FIG. 6(b) does not show the device isolation regions 3 in FIG. 5, a p-type impurity is implanted between the gate electrodes of the memory cell region. At this time, the impurity is also introduced into the source/drain regions 1. However, the source/drain regions 1 remain n.sup.+ -type because they are high concentration n-type impurity regions.
After the resist pattern 9 is removed, a resist pattern 13 is formed which is open at a portion of the peripheral circuit region whose source/drain regions will become an n-type transistor, as shown in FIG. 6(c). With this pattern used as a mask, an n-type impurity is implanted in a dose of about 3.times.10.sup.13 cm.sup.-2 to form an n.sup.- region 14. This n.sup.- region 14 will become a low concentration region of an LDD structure.
Subsequently, the resist pattern 13 is removed. After this, a resist pattern (not shown) is formed which is open at a portion of the peripheral circuit region whose source/drain regions will become a p-type transistor. With this pattern used as a mask, a p-type impurity is implanted in a dose of about 3.times.10.sup.13 cm.sup.-2 to form a p.sup.- region 11. This p.sup.- region 11 will become a low concentration region of an LDD structure.
Next, the resist pattern is removed and sidewall spacers 18 are formed on the side walls of the gate electrodes, as shown in FIG. 6(d). Then, a resist pattern (not shown) is again formed which is open at the portion of the peripheral circuit region whose source/drain regions will be the n-type transistor. With this pattern used as a mask, an n-type impurity is implanted in a dose of 10.sup.15 to 10.sup.16 cm.sup.-2 to form the source/drain regions 16. Further, a resist pattern 17 is formed which is open at the portion of the peripheral circuit region whose source/drain regions will be the p-type transistor. With this pattern used as a mask, a p-type impurity is implanted in a dose of 10.sup.15 to 10.sup.16 cm.sup.-2 to form the source/drain regions 15.
Afterwards, an impurity is implanted into predetermined memory cell transistors to vary threshold voltages, thereby writing data as program information. This completes the intended masked ROM.
Here, the source/drain regions and the low concentration regions may not be formed in the above processing order. Either of the P-MOS transistor and the N-MOS transistor may be formed first. Also, the P-MOS transistor may not comprise a low concentration region due to its characteristics. In such a case, the process of forming the low concentration region is omitted from the above process. Further, the peripheral circuit region may comprise only an N-MOS transistor.
In the above process, when an impurity having the same conductivity type as that of the semiconductor substrate is introduced for device isolation between the memory cell transistors, the impurity for the device isolation must be implanted only into the memory cell region by using a photomask such as the resist pattern 9 on the peripheral circuit region, as shown in FIG. 6(b).
This is due to the following reason. When the impurity for device isolation in the memory cell region is also introduced into the low concentration source/drain regions functioning as the LDD structures for preventing the generation of hot carries in the transistors of the peripheral circuit region, and if the source/drain regions of a transistor in the peripheral circuit region have the same conductivity type as that of the source/drain regions in the memory cell transistor, the conductivity type of the low concentration source/drain regions in the transistor of the peripheral circuit region will be lost because the above impurity for device isolation has a different conductivity type from that of the impurity in the low concentration source/drain regions and has a similar degree of concentration. This leads to degradation in the saturation drain current of the transistor and in the device characteristics such as decrease in junction breakdown voltage. Therefore, a photomask is used which is necessary for masking the peripheral circuit region when the impurity is introduced for forming the device isolation in the memory cell region.
After the formation of the gate electrodes and before the formation of the source/drain regions 15 and 16 in FIG. 6(d), the conventional method needs five photolithography steps and five impurity implantation steps for device isolation of the memory cell region, n.sup.- implantation, p.sup.- implantation (if the p-MOS transistor also is to have LDD structures), n.sup.+ implantation, and p.sup.+ implantation.
Here, if the p-MOS transistor is not to have LDD structures, there will be needed four photolithography steps and four impurity implantation steps.
It is essential to reduce the number of photomasks used in manufacturing LSIs in view of lowering the costs and improving the yield by simplifying the process. However, a simple reduction of the number of photomasks will cause variance in LSI characteristics and will invite degradation of the quality.

Method for manufacturing semiconductor memory device


Disclosed is a method for preparing a metal catalyst having improved yield of alcohols. The method for preparing a metal catalyst for the production of alcohol from synthesis gas includes forming a metal catalyst; and irradiating the metal catalyst with gamma rays. The metal catalyst has improved yield of alcohols by stabilizing the metal catalyst through gamma ray irradiation to inhibit generation of hydrocarbons in catalytic reaction with synthesis gas.

1. Technical Field
The present invention relates to a method for producing a metal catalyst for preparing alcohols through catalytic reaction, and more particularly, to a method for producing a metal catalyst for catalyzing conversion of synthetic gas into alcohols.
2. Description of the Related Art
Generally, "synthesis gas (synthesis gas)" refers to a gas mixture containing various amounts of carbon oxides (like carbon monoxide) and hydrogen. Such a term comes from its use as an intermediate in production of synthetic natural gas, ammonia, alcohols or other fuels.
Synthesis gas may be produced in various ways, for example, by gasification of carbon-containing fuels into products in a gas phase having heating value, for instance, by gasification of coal, biomass or municipal waste. Further, the synthesis gas may be produced by steam reforming of natural gas, methane from various sources, or liquid hydrocarbons. The synthesis gas is combustible and may be used as a fuel source or an intermediate for production of other chemical materials. The synthesis gas may also be used as an intermediate in producing synthetic petroleum for use as fuel or lubricant via catalytic reaction a Fischer-Tropsch catalyst.
Worldwide energy consumption has been continuously increasing, and greatly relies on fossil fuels such as petroleum, natural gas, and coal. However, fossil fuel reserves are considerably limited and consumption of fossil fuels is greatly exceeding production. Given such a circumstance, there is strong demand for a new energy source capable of effectively replacing fossil fuels. Renewable fuels such as alcohols, particularly ethanol, draw considerable attention as a hydrogen source for automobiles, chemical additives or fuel cells.
Specifically, fuel cells convert chemical energy into electric energy through electrochemical reaction of oxygen and hydrogen. Since fuel batteries are characterized by having high energy utilization efficiency, studies into practical application have been actively conducted for the purpose of public welfare, industries, automobiles, and the like.
As methods for producing an alcohol from synthesis gas for use as a raw material for fuel cells, technology using a catalyst, specifically a metal catalyst, is well known and considerable research thereon has been carried out. In research of catalysts, the most important goal, in terms of fundamental function of a metal catalyst, is improvement in catalyst performance. In order to efficiently perform catalysis, a promoter, a support, or the like of the catalyst may be redesigned.
As a representative metal catalyst for synthesizing an alcohol including ethanol using synthetic gas, Cu catalyst is used. However, the Cu catalyst has a demerit in that alcohol yield is low due to co-production of hydrocarbons during alcohol synthesis.
[Prior Art Document] KR 10-2008-0108605 A

Method for producing metal catalyst for preparing alcohol and metal catalyst produced thereby


An optical coherence tomography (OCT) engine includes a digital Fourier-Transform (dFT) spectrometer, a tunable delay line, and a high-speed optical phased array (OPA) scanner integrated onto a single chip. The broadband dFT spectrometer offers superior signal-to-noise ratio (SNR) and fine axial resolution; the tunable delay line ensures large imaging depth by circumventing sensitivity roll-off; and the OPA can scan the beams at GHz rates without moving parts. Unlike conventional spectrometers, the dFT spectrometer employs an optical switch network to retrieve spectral information in an exponentially scaling fashion—its performance doubles with every new optical switch added to the network. Moreover, it also benefits from the Fellgett's advantage, which provide a significant SNR edge over conventional spectrometers. The tunable delay line balances the path length difference between the reference and sample arms, avoiding any need to sample high-frequency spectral fringes.

Since its advent in the early 1990s, OCT has rapidly become a ubiquitous imaging technology for biomedical diagnosis and non-destructive testing (NDT) due to its unique capability to generate high-quality three-dimensional (3-D) images in a non-invasive manner. Nowadays, most mainstream OCT systems are in the so-called spectral domain (SD)-OCT configuration due to its sensitivity advantage (analogous to the Fellgett's advantage in the spectral domain). In SD-OCT, light from a spatially coherent broadband source (e.g., a superluminescent diode (SLD) or supercontinuum source) is split into two paths: (1) a reference arm with a fixed mirror and (2) a sample arm where the light illuminates and is backscattered by the sample. Reflected light from the two paths are combined and fed into a spectrometer.
Light scattered at different depths of the sample produces interference fringes with the reference beam with varying free spectral range (FSR), where a small FSR signals a large optical path length (OPL) difference between the reference arm and the backscattering source. The resulting spectrum contains a series of spectral-domain oscillations (fringes), each corresponding to the optical backscattering strength at a particular sample depth. A Fourier transform of the spectrum then generates a one-dimensional (1-D) depth profile of the sample. An alternative to SD-OCT is to use a wide-band tunable laser as the light source (swept-source OCT), which suffers from a higher cost, coarser axial resolution, and limited system ruggedness due to the delicate moving parts in the tunable laser.
Performance metrics of an SD-OCT system include imaging depth, axial resolution, sensitivity, A-scan line rate (each A-scan acquires a 1-D depth profile at one spatial location), lateral resolution, and field of view. The lateral resolution and field of view are usually determined by the scan lens and the imaging probe construction rather than the OCT engine.
The imaging depth of OCT depends on both the penetration depth of light in the sample as well as the optical resolution of the spectrometer (or sampling rate in the spectral domain). The penetration depth can be optimized by choosing suitable wavelengths for different samples. For instance, typical wavelength choices include 800-1000 nm for imaging eyes, 1300 nm or longer for more opaque tissue (e.g., skin), and even the mid-infrared (IR) range for NDT in highly scattering samples.
Spectrometer resolution δλ is often a limiting factor for SD-OCT. In this case, the imaging depth range Z is given by:
                    Z        =                              1            2                    ·                                    λ              2                                      δ                            λ                                                          (        1        )            In addition to limiting the imaging depth, the finite spectrometer resolution also inflicts sensitivity roll-off, also known as loss of OCT sensitivity with depth. The roll-off can be understood as follows: with larger imaging depth, the optical path length difference between the reference arm and the backscattering source increases. Consequently, the resulting spectrum contains spectral-domain oscillations with smaller FSRs. As the FSR approaches the spectrometer resolution limit, the SNR corresponding to the imaging depth drops due to the reduced contrast transfer of the spectral oscillations. In other words, the SNR becomes bounded by the spectral-domain modulation transfer function (MTF) of the spectrometer for a fixed reference arm length.
The axial resolution Ic is dictated by the bandwidth BW of the light source or the spectrometer:
                              l          c                =                  C          ·                                    λ              2                                      B                            W                                                          (        2        )            Here C is a constant that depends on the spectra of the light source or wavelength-dependent response of the spectrometer. Current OCT systems typically provide an axial resolution of a few microns, corresponding to a spectral bandwidth of approximately 100 nm for a center wavelength of 1300 nm.
The speed of an OCT measurement (the A-scan rate) is limited by both the SNR (and hence integration time at the detector) and the speed of the spectrometer or the wavelength-sweeping laser. When SNR is the limiting factor, sensitivity and speed pose a trade-off. State-of-the-art OCT systems typically operate with A-scan rates of the order of tens of kHz, with some ultrafast systems now offering multi-MHz scanning rates.
Finally, the sensitivity of an OCT measurement, defined as the minimum sample reflectivity that generates an SNR of unity, can be enhanced by suppressing the main noise sources. These noise sources include the photon shot noise, the relative intensity noise (RIN) of the light source, and the detector noise. While photon noise is limited by the system optical throughput and the detector integration time, the RIN and detector noise can be mitigated with improved optical engine design.
To overcome the poor size, weight, and power plus cost (SWaP-C) metrics of conventional OCT systems, silicon photonics has been explored as a solution to enable low-cost, portable OCT. To date, the work on silicon-photonic OCT systems can be roughly divided into two categories. In the first category, the goal is to replace a traditional spectrometer with an on-chip counterpart, such as an arrayed waveguide grating (AWG) or planar concave grating. The bandwidth of an integrated photonic spectrometer is typically limited by the operating bandwidth of its photonic components to a fraction of its center wavelength. For instance, a single properly designed AWG can operate over 10-15% fractional bandwidth, corresponding to 100-200 nm coverage near the telecom bands. It is also possible to cascade and juxtapose many AWGs (or other single spectrometers) to extend the bandwidth, although this approach comes at the expense of vastly increased system complexity and exacerbated signal-to-noise ratio (SNR) penalty. OCT systems in this first category tend to be hampered by the optical throughput penalty in dispersive spectrometers and their excessively large footprints (e.g., a 512-channel AWG occupies an on-chip area of 2.0 cm×2.7 cm).
In the second category of silicon-photonic OCT systems, the goal is to miniaturize the interferometer in the OCT system by integrating the reference arm and/or optical receiver on-chip. The SD-OCT systems in this category have reference arms with fixed OPLs. At least one report disclosed the use of a waveguide delay line for now-obsolete time-domain OCT. This waveguide delay line provided an OPL tuning range of less than 1 mm despite a long waveguide length. In this type of silicon-photonic OCT system, the on-chip optics are merely down-scaled versions of the bulk elements bringing little architectural innovation or optical performance boost.

Chip-scale optical coherence tomography engine


A method for reducing a delay, between a transition of a transmission of a metal detector and a process of a receive signal received by a receive coil of the metal detector, due to a critically damped time constant of the receive coil, including: introducing a negative capacitance into the receive coil to reduce the critically damped time constant.

Time domain metal detectors usually synchronously demodulate (or sample) a receive signal from a receive coil commencing after a short delay following a voltage transition of a transmit signal, for example, after the back emf of a pulse induction (PI) transmit pulse, or after a switch from a high voltage (e.g. 200V) to a low-voltage (e.g. 5V) or to zero voltage. The short delay is to avoid the processing electronics from demodulating potential spurious interfering signals (unwanted signals) from mildly conductive soils. Typically, these signals from mildly conductive soil have signal components of particularly short duration; they will decay very rapidly. Detection of these signals is undesirable in a metal detector. Given that they were the only difficulty to be overcome, a short delay of the onset of demodulation after the transition of transmit signal would be sufficient to reduce their effect upon detection. In most detectors, however, the demodulation must be delayed by more than is required to remove the effect of the signals from mildly conductive ground. In most detectors, the minimum of the practical delay is usually limited by the time constant of the critically damped receive coil, either because the decaying transient signal components in the receive coil, due to a voltage transition of the transmit signal, contain signals related to the commonly major soil component known in the field of metal detection, namely the signal components due to soil reactive magnetic permeability, which is normally required to be nulled out, or to prevent said transients from causing receive signal electronic to overload, or both.
The required delay before demodulation, (or sampling), is undesirable for the detection of fast constant metal targets, because much of the decaying signal from the fast time constant targets following a transition of the transmit signal occurs during the delay, and thus most of this target signal is not added into the receive demodulation. Many sought buried metal targets have short time constants: for example; some minimum metal land mines, small gold nuggets, and fine jewellery. Improving the sensitivity to very short time constant targets requires reduction of the delay before demodulation, in turn requiring reduction of the time constant of the critically damped receive coil. Hitherto, this was usually achieved by choosing a low inductance for the receive coil, or eliminating the capacitive loading of the receive coils by the capacitance of the connecting cable, through installation of the receive preamplifier within the coil housing.

Method for detecting fast time constant targets using a metal detector


An electric circuit for amplifying an xDSL signal comprises an operational amplifier and a signal monitoring circuit. The operational amplifier is configured to amplify the xDSL signal, is powered by a variable voltage supply and has a gain which is adjustable by an operating mode control signal. The signal monitoring circuit is activated by the operating mode control signal and is configured, when activated by the operating mode control signal, to generate a control signal to adjust the voltage of the variable voltage supply in order to adjust the maximal signal swing of the output signal of said operational amplifier. The control signal is generated by comparing the amplitude of the xDSL signal applied to the electric circuit with an amplitude threshold value.

1. Field of the Invention
The invention relates to an electric circuit configured to drive different xDSL signals, particularly ADSL and VDSL signals.
2. Description of the Prior Art
Various DSL (Digital Subscriber Line) standards have been developed in recent years. Examples of well known DSL standards are ADSL (Asymmetric Digital Subscriber Line), HDSL (High Bit Rate Digital Subscriber Line), IDSL (ISDN Digital Subscriber Line), MDSL (Medium Bit Rate Digital Subscriber Line), RDSL (Rate Adaptive ADSL), RADSL (Reverse ADSL), SDSL (Symmetric Digital Subscriber Line) and VDSL (Very High Speed Digital Subscriber Line).
Different DSL standards have different upstream and downstream data rates. Some DSL standards, for instance, ADSL, MDSL, SDSL and VDSL are used in combination with a conventional telephone service (POTS), while other DSL standards are not compatible with POTS, such as HDSL and IDSL. The xDSL standard which has been used most widely is the ADSL standard, which has different data transmission rates for the data streams flowing to the user (downstream) and for the data streams leaving the user (upstream). The ADSL standard is, therefore, also called asymmetric. For ADSL, data transmission rates are typically about ten times as high downstream, i.e. toward the user, as upstream, i.e. toward the service provider. For ADSL, the downstream data transmission rates are typically between 1 and 8 Mbps, while the upstream data transmission rates are between 100 and 800 kbps. The data transmission rates of VDSL are much greater and reach, for instance, 25 Mbps in both data transmission directions.
In the case of ADSL, digital multitone (DMT) modulated signals are transmitted in a relatively narrow frequency band with a frequency bandwidth of approximately 2 MHz. By contrast, the data transmission of VDSL takes place in a relatively wide frequency band with a frequency bandwidth of up to 30 MHz.
FIG. 1 shows an xDSL driver circuit based on the prior art. The xDSL driver circuit has a signal input for applying an xDSL signal which comes from an analogue front end (AFE). The applied xDSL signal is applied to the inputs of an operational amplifier OP via decoupling capacitors C1 and input resistors R1. The operational amplifier OP, which is a fully differential design in this exemplary embodiment, has two signal outputs which are each fed back via a feedback resistor R2 to the corresponding input of the operational amplifier OP. In addition, the outputs of the operational amplifier OP are coupled to primary windings L1 of a transformer TR via an output resistor R4. The transformer TR has secondary windings L2, with the ratio of the primary to the secondary windings resulting in a transformer ratio Ü. The transformer TR decouples a DC component from the output signal. A capacitor C2 is connected in series to the secondary windings L2, forming a high-pass filter for decoupling the POTS telephone signals from the data signals.
The outputs of the operational amplifier OP are also coupled crosswise via resistors R3 to the inputs of the operational amplifier OP. The resistors R3 are used for positive feedback in order to produce a synthesized output impedance for the driver circuit.
The output impedance ZA′ on the primary side of the transformer TR is obtained from the product of a synthesis factor, m, and the output resistor R4:ZA′=m2R4.
The value of the feedback resistors R3 determines the synthesis impedance factor, m. The ratio of the resistor R2 to the resistance value of the input resistor R1 determines the signal gain or the gain G of the operational amplifier OP:G=R2/R1.
As the transformer ratio Ü increases, the supply voltage VDD required for the operational amplifier OP decreases. With an increasing transformer ratio Ü, however, the operational amplifier OP or line driver has to deliver a higher output current I in order to achieve the output power determined by the standard on the line.
The length of the signal or telephone lines between the xDSL driver circuit at the central station (central office) and the driver circuit at the user (customer premises) varies between different xDSL standards. Accordingly, the power P to be transmitted is likewise different for different xDSL standards. By way of example, the power P which is to be output onto the line is 20 dBm for ADSL 2+ and 14.5 dBm for VDSL. As mentioned before, for ASDL the data are transmitted in a relatively narrow frequency band. For ADSL 2, the frequency bandwidth is 2.2 MHz. In this context, the data are transmitted in two separate subfrequency bands, with the first subfrequency band being provided for the data transmitted to the user (downstream) and the second subfrequency band being provided for the data transmitted from the user to the central station (upstream). A relatively small number of subfrequency bands or the small frequency bandwidth means that the risk of crosstalk in ADSL systems is relatively low. By contrast, for VDSL the data are transmitted in a relatively wide frequency band. VDSL1, for example, has a frequency bandwidth of 12 MHz and VDSL2, for instance, has a frequency bandwidth of 17 MHz. In this case, preferably three subfrequency bands are provided for the downstream data transmission direction and three subfrequency bands are provided for the upstream data transmission direction. The relative large transmission frequency bandwidth means that the risk of crosstalk is greater in VDSL systems than in ADSL systems.
In order to avoid the risk of crosstalk, the signals output by the VDSL driver circuit in a VDSL system are therefore transmitted at a lower power level than in ADSL systems. For VDSL, the prescribed maximum power level is 14.5 dBm, whereas in ADSL systems a maximum power level with the line of up to 20 dBm is admissible. The different signal powers mean that the voltage signal swing in the signal, which is output by the operational amplifier, is also different in different xDSL standards. The signal swing at the output of the operational amplifier is mainly determined by the supply voltage VDD applied to the operational amplifier. For ADSL systems, the supply voltage for the operational amplifier is 20 V, and for VDSL systems a typical VDSL driver circuit has a supply voltage of approximately 12 V.
The synthesized output impedance using the positive feedback resistors R3 minimizes the signal swing at the output of the operational amplifier and the latter's power consumption. As the frequency bandwidth for data transmission increases, the distortion caused by the operational amplifier increases due to the decreasing loop gain. Consequently, the synthesis factor, m, has an upper limit. The synthesis factor, m, in typical ADSL driver circuits is 5 to 6, whereas the synthesis factor in VDSL driver circuits is 3. The higher the transmission frequency bandwidth, the lower the admissible synthesis factor, m.
The signal swing at the output of the operational amplifier, for instance, is 17 Vp for an ADSL driver circuit at a supply voltage VDD=20 V, at a synthesis factor, m, of 6 and at a maximum permissible output power P of 20 dBm. By contrast, the maximum signal swing of the output of teh operational amplifier for a VDSL driver circuit at a supply voltage of 12 V and a synthesis factor, m, 3 is 10 Vp, i.e. 10 V peak to peak.
The xDSL driver circuit based on prior art and being shown in FIG. 1 is thus configured in accordance to the xDSL standard. The table below shows the most important data for configuring xDSL driver circuits based on the prior art for the ADSL standard and the VDSL standard.
TABLEADSLVDSLSupply voltage VDD20 V12 VTransmission power P20 dBm14.5 dBmOutput resistor R4~5 Ω−10 ΩSynthesis factor m5-6~3       Gain                          G    =            R                                      2              R                                      1      ~16~11 Transformer ratio Ü1.31.3
A drawback of prior art xDSL driver circuits as shown in FIG. 1 is that different xDSL driver circuits need to be provided, depending on which type of xDSL signals are being transmitted. If a user wishes to change from ADSL to VDSL, for example, then the ADSL driver circuit needs to be replaced by a VDSL driver circuit, i.e. the hardware needs to be replaced. If the user wishes to return to the ADSL standard, then the VDSL driver circuit needs to be replaced by an ADSL driver circuit again, i.e. the boards are swapped again. This is naturally very laborious for the user. The xDSL driver circuit based on the prior art, as shown in FIG. 1, provides no kind of flexibility for the xDSL signal applied.


A low part count momentary push-button assembly having an integral indicator lamp for indicating activation of a controlled circuit or device. The push-button assembly comprises a switch unit and indicator lamp mounted side-by-side on a central region of a printed circuit board, a push-button having a contact area disposed in axial alignment with the switch unit, and a light transmissive passage disposed in axial alignment with the indicator lamp, and a unitary interface element disposed intermediate the circuit board and the push-button. The interface element includes an integral sleeve for locating the indicator lamp in relation to the light transmissive passage of the push-button, and a cantilevered structure which is maintained in resilient engagement with the contact area of the push-button. The interface element snaps onto the circuit board, and the circuit board is snapped into a housing cavity. The push-button is retained in relation to the switch unit and indicator lamp by a housing element which closes the cavity.

Momentary push-button switches are commonly used in automotive applications, for example, for the activation of a rear window de-fog or windshield de-ice timer circuit. In these and other applications, the provision of a lamp to indicate activation of the controlled circuit is usually desired. Also desired is a simplified assembly having a low part count for minimizing cost and time of installation.

Lighted momentary push-button switch assembly having integral switch actuator and lamp locator


A method of plasma processing includes generating a first sequence of source power pulses, generating a second sequence of bias power pulses, combining the bias power pulses of the second sequence with the source power pulses of the first sequence to form a combined sequence of alternating source power pulses and bias power pulses, and, using the combined sequence, generating a plasma comprising ions and processing a substrate by delivering the ions to a major surface of the substrate.

Device formation within microelectronic workpieces may involve a series of manufacturing techniques including formation, patterning, and removal of a number of layers of material on a substrate. In order to achieve the physical and electrical specifications of current and next generation semiconductor devices, processing flows enabling reduction of feature size while maintaining structural integrity is desirable for various patterning processes.
Plasma processes are commonly used to form devices in microelectronic workpieces. For example, plasma etching and plasma deposition are common process steps during semiconductor device fabrication. A combination of source power and bias power may be used to generate and direct plasma during plasma processing. FIG. 18 illustrates conventional timing diagrams for the application of source power and bias power during plasma processing. In the top diagram, there are no distinct pulses for the source power or the bias power. In the middle diagram, continuous bias power is applied with no pulses while source pulses are applied. In the bottom diagram, continuous source power is applied with no pulses while bias pulses are applied.

Systems and methods of control for plasma processing


An N-bit asynchronous successive approximation analog-to-digital converter is provided for converting an analog signal to a binary coded representation thereof. The analog-to-digital converter employs N inverting amplifiers. Each of the inverting amplifiers receives a scaled analog input signal and provides a one bit output of the binary coded output. The scaled analog input signal applied to each inverting amplifier includes an analog input signal and feedback signals from each of the inverting amplifiers which provide a more significant output bit of the binary coded output. Each inverting amplifier thereby provides an output dependent upon the amplifier's threshold voltage which output provides one bit of the binary coded representation. In addition, each inverting amplifier may further be adapted to receive a reference signal.

1. Technical Field
This invention relates to analog-to-digital converters for converting an analog input signal to a digital output representation and, more particularly, to an asynchronous successive approximation analog-to-digital converter.
2. Discussion
Analog-to-digital converters are commonly used in a wide variety of applications for converting an analog signal to a digital representation thereof. One common method of providing analog-to-digital conversion is with successive approximation. Successive approximation generally employs a comparator, a digital-to-analog (D/A) converter and a successive approximation register (SAR) and additional electronic devices such as clocks, flip-flops and latches.
A conventional successive approximation analog-to-digital converter is essentially a synchronous device that requires at least N clock cycles to provide an N-bit conversion. As a result, a four-bit successive approximation conversion would generally require at least four clock cycles. In addition, the typical successive approximation converter is a relatively slow device because it does not make comparisons with the input signal simultaneously as is generally accomplished with a flash converter.
Another method of providing analog-to-digital conversion is with the use of a flash converter. A conventional analog-to-digital flash converter generally employs 2.sup.N comparators in conjunction with an accurate resistor network. As such, the flash converter generally requires at least 2.sup.N electrical components including the 2.sup.N comparators to perform an N-bit conversion. As a result, the relatively large number of electrical components that are required may lead to a very high circuit complexity, especially for high bit converters.
It is therefore desirable to obtain an analog-to-digital converter that is asynchronous and therefore does not require any clock timing. In addition, it is further desirable to obtain such an analog-to-digital converter that requires fewer electrical components than the conventional flash converter.

Asynchronous successive approximation analog-to-digital converter


This invention relates to mutant DNA polymerases having an enhanced template discrimination activity compared with the corresponding unmodified DNA polymerase counterparts, wherein the amino acid sequence of the mutant DNA polymerase includes at least one substitution at residue positions structurally and functionally homologous or orthologous positions 783 or 784 of an unmodified Taq DNA polymerase.

The ability to accurately diagnose a given genetic condition and to predictably treat a genetically-based disorder requires reliable methods for accurately determining genetic sequence information. Many genetically-based disorders are associated with single nucleotide polymorphisms (SNP's) in protein coding genes. The presence of SNP's associated with a genetically-based disorder, such as a cancer, can be difficult to detect owing to the small numbers of genetically altered cells in the population that encode the allele(s) ("rare alleles").
Polymerase-based assays, such as the polymerase chain reaction (PCR), have important impact and widespread use in genetic diagnostics and molecular medicine. Polymerases synthesize DNA sequences by the addition of nucleotides to the 3′ end of a short oligonucleotide (abbreviated to "primer" in the following). The primer is hybridized to the single stranded sequence that is going to be amplified ("template"). DNA polymerases catalyze formation of a phosphodiester bond between the 3′-oxygen at the 3′-terminus of the primer and the incoming deoxynucleoside triphosphate ("dNTP"). This chemical reaction ("primer extension") adds a nucleotide to the primer (e.g., to the nascent growing DNA chain). Primer extension is base specific, in that the deoxynucleoside triphosphate that is complementary to the base in the template is added to the primer. The fidelity of DNA polymerase enzymes is very high and the rate of mutations introduced into the replicated DNA strand is low; however, the precise error rate varies between different DNA polymerase enzymes and these rates have been well characterized. The extension reaction can be repeated until the end of the template is reached.
The majority of polymerase-based assays for detecting SNP's rely upon having the polymerase enzyme discriminate between at least two different substrates. The first substrate contains the desired SNP that is to be detected; the second substrate contains the normal nucleotide that is not to be detected. Polymerase-based discrimination can be achieved by providing the first substrate as the preferred polymerase-competent substrate for assay. This discrimination can be maximized to the extent that the first substrate is the only polymerase-competent substrate present for assay.
Many strategies are known in the art for establishing conditions that favor polymerase-based discrimination among substrates having minimal nucleotide differences, such as those containing only single nucleotide differences. One strategy relies on a polymerase's inability to efficiently initiate synthesis on substrates lacking a 3′-paired nucleotide on the primer. An allele-specific primer design is a primer in which the 3′-nucleotide forms a perfect match with the complementary base at the location containing the SNP-containing allele and forms a mismatched pairing when annealed to an allele lacking the SNP. The primer:template for the SNP-containing allele serves as the preferred polymerase-competent substrate since the polymerase can efficiently initiate primed synthesis from such substrates. Examples of these strategies are described in Chen et al., "Single nucleotide polymorphism genotyping: biochemistry, protocol, cost and throughput" Pharmacogenomics J. 3(2):77-96 (2003); Kwok et al., "Detection of single nucleotide polymorphisms" Curr. Issues Mol. Biol. 5(2):43-60 (April 2003); Shi, "Technologies for individual genotyping: detection of genetic polymorphisms in drug targets and disease genes" Am. J. Pharmacogenomics 2(3):197-205 (2002); and Kwok, "Methods for genotyping single nucleotide polymorphisms" Annu. Rev. Genomics Hum. Genet. 2:235-58 (2001). A strategy to improve selectivity for this class of allele-specific PCR primers is to introduce a second mutation at the penultimate base, next to the 3′-terminal nucleotide of the primer (i.e., next to the SNP site). As before, the 3′-terminal residue will either be a match or mismatch to the base under interrogation in the sample nucleic acid (SNP), but now the primer will either have two adjacent mismatches to the target (both 3′-terminal and penultimate base) or a single mismatch to the target (at only the penultimate base, with the 3′-terminal base being a match). See, for example, Newton et al., "Analysis of any point mutation in DNA. The amplification refractory mutation systems (ARMS)" Nucleic Acids Res. 17(7):2503-15 (1989). Yet another strategy to improve selectivity for this class of allele-specific PCR primers is to employ a chemically modified nucleic acid residue at the 3′-end of the primer, such as a locked nucleic acid (LNA), which reduces the ability of DNA polymerase to initiate DNA synthesis from a 3′-terminal mismatch. See, for example, Latorra et al., "SNP genotyping using 3′ locked nucleic acid (LNA) primers" Human Mut. 22(1):79-85 (2003).
Template substrate discrimination can be enhanced in polymerase-based assays by requiring a second nucleic acid enzyme catalyze formation of one or more primers for use in the polymerase-based assay. In one such assay, the ligase chain reaction assay, a DNA ligase is used with a polymerase to detect a template containing a SNP. Since polymerase-based assays require primers having a minimum length to hybridize to the template substrate, a DNA ligase can be used to generate polymerase primers from sub-optimal length oligonucleotides. The assay relies upon hybridizing two probes directly over the SNP polymorphic site, whereby ligation can occur if the probes are identical to the target DNA. Two probes are designed; an allele-specific probe which hybridizes to the target DNA so that its 3′ base is situated directly over the SNP nucleotide and a second probe that hybridizes the template downstream in the complementary strand of the SNP polymorphic site providing a 5′ end for the ligation reaction. If the allele-specific probe matches the target DNA, it will fully hybridize to the target DNA and ligation can occur. Ligation does not generally occur in the presence of a mismatched 3′-base. Once the oligonucleotide product is formed, it can serve as a primer or as a template for polymerase-based assays. Examples of this strategy are described in Barany F. "Genetic disease detection and DNA amplification using cloned thermostable ligase." Proc Natl Acad Sci USA. 1991 January 1; 88(1):189-93 and Wiedmann M., Wilson W. J., Czajka J., Luo J., Barany F., Batt C. A. "Ligase chain reaction (LCR)—overview and applications." PCR Methods and Applications 1994 February; 3(4): S51-64.
Since a polymerase-competent substrate requires a primer:template having an available 3′-hydroxyl group on the primer, another strategy known in the art, RNase H-based PCR (rhPCR), can be used for improving polymerase-based discrimination. The rhPCR method makes use of RNase H enzymes to convert a primer lacking a 3′-hydroxyl group ("blocked primer") or a primer that is otherwise disabled and cannot support PCR to a primer containing a 3′-hydroxyl group ("unblocked primer") that can support PCR. A blocked primer in rhPCR includes an oligonucleotide having a blocked 3′-end or other modification which prevents either priming or template function of the oligonucleotide and an internal RNA residue, which serves as a cleavage site. Type II RNase H recognizes annealed primer:template duplexes containing these blocked primers and cleaves the primer strand 5′ of the RNA residue to generate a 3′-hydroxyl group at the adjacent DNA residue. Since RNase H enzymes do not cleave substrates containing an unpaired RNA reside at a mismatched site, allele-specific template discrimination can be achieved by placement of the RNA residue at the location complementary to the SNP on the selected allele template. The resultant Type II RNase H cleavage product can serve as a polymerase competent substrate. Examples of this enzyme cleaving strategy, similar RNase H strategies, and methods of blocking primer extension or inhibiting template function and thereby disabling PCR are described in U.S. Pat. No. 7,112,406 to Behlke et al., entitled POLYNOMIAL AMPLIFICATION OF NUCLEIC ACIDS, U.S. Pat. No. 5,763,181 to Han et al., entitled CONTINOUS FLUOROMETRIC ASSAY FOR DETECTING NUCLEIC ACID CLEAVAGE, U.S. Pat. No. 7,135,291 to Sagawa et al., entitled METHOD OF DETECTING NUCLEOTIDE POLYMORPHISM; U.S. Pat. App. No. 20090068643 to Behlke and Walder, entitled DUAL FUNCTION PRIMERS FOR AMPLIFYING DNA AND METHODS OF USE; and U.S. Pat. App. No. 20100167353 to Walder et al., entitled RNASE H-BASED ASSAYS UTILIZING MODIFIED RNA MONOMERS.
The field has focused on substrate-based approaches, such as those exemplified above, for improving detection of genetic differences and rare alleles. Yet the sensitivity of polymerase-based assays remains limited by the formation of non-specific amplification products that arise from ectopic or aberrant primer-related extension products independent of the desired templates. The inherent reactivity of the polymerase appears largely responsible for producing such artifacts during amplification.
Any further improvements in mismatch discrimination may therefore require a modified polymerase enzyme endowed with inherently better 3′-nucleotide discrimination when used with one or more of the described strategies. A modified polymerase enzyme having activity differing from the unmodified form can be prepared by chemical or enzymatic modification of the protein or mutagenesis of corresponding gene that encodes the protein. The latter approach is generally preferred owing to the fact that genetically altered genes encoding a given mutant protein can be stably maintained, expressed and purified to yield enzyme preparations having well-characterized properties.
While an unbiased mutagenesis strategy can be used to generate a library of mutant polymerase genes, this approach has certain disadvantages. Many millions of mutant enzymes must be screened for activity and success is often dependent upon the chance that effective mutations are present in the limited pool generated by random mutagenesis. Direct genetic selection methods are not sufficiently sensitive for identifying mutations that pertain to secondary functions falling outside of an essential polymerase activity. The vast majority of mutant polymerase genes in a positive selection assay will likely encode protein that retains the functional attributes of the normal polymerase enzyme. Thus, secondary screening procedures that use biochemical assays must be done to identify whether any mutant polymerases have the desired activity. Notwithstanding the technical difficulties of setting up the initial selection process, the attendant costs associated with performing the secondary screens using biochemical assays is prohibitive if more than one hundred clones need to be purified and assayed.
An alternative approach is to apply a biased mutagenesis strategy that is specifically targeted to a preselected region of a gene implicated in function. In this approach, one first identifies genetic regions by a selection method. One such selection method is a comparative phylogenetic analysis of a particular gene that is required for organisms of diverse origins. The principle of comparative phylogenetic analysis is premised on the hypothesis that diverse organisms will not share protein coding sequences in essential genes unless those sequences are evolutionary constrained for reasons related to essential protein function.
A phylogenetic comparative analysis of genes encoding DNA polymerases can provide insights about possible amino acid residues important to polymerase functions. The overall folding pattern of DNA polymerases resembles the human right hand and contains three distinct subdomains of palm, fingers, and thumb. (See, for example. Beese et al., Science 260:352-355, 1993; Patel et al., Biochemistry 34:5351-5363, 1995). While the structure of the fingers and thumb subdomains vary greatly between polymerases that differ in size and in cellular functions, the catalytic palm subdomains are all superimposable. For example, motif A, which interacts with the incoming dNTP and stabilizes the transition state during chemical catalysis, is superimposable with a root mean deviation of about one Angstrom among mammalian pol a and prokaryotic pol I family DNA polymerases (Wang et al., Cell 89:1087-1099, 1997). Motif A begins structurally at an antiparallel β-strand containing predominantly hydrophobic residues and continues to an α-helix. The primary amino acid sequence of DNA polymerase active sites is exceptionally conserved.
In addition to being well-conserved, the active site of DNA polymerases has also been shown to be relatively mutable, capable of accommodating certain amino acid substitutions without reducing DNA polymerase activity significantly. (See, e.g., U.S. Pat. No. 6,602,695). Such mutant DNA polymerases can offer various selective advantages in, e.g., diagnostic and research applications comprising nucleic acid synthesis reactions. We identify mutations in protein sequence using the single-letter amino acid codes and an integer number that indicates location of the mutation from the beginning of the protein sequence. The single-letter amino acids codes are well known in the art, e.g., Stryer et al., Biochemistry, 5.sup.th ed., Freeman and Company (2002). As an example, aspartic acid ("D") is changed to glycine ("G") in D580G mutant and the change is located 580 amino acids from the beginning of the protein sequence.
Reichert et al. conducted a comparative phylogenetic analysis of thermoactive DNA polymerases from thermophilic bacteria, wherein the protein coding sequences of DNA Polymerase I enzymes were aligned for thirteen phylogenetically distinct species. The analysis revealed that eight amino acid positions within a 15-amino acid long motif located at amino acid positions 645-685 (in reference to Thermus sp. Z05 DNA polymerase ("Z05 DNA Polymerase") might tolerate alterations without compromising core enzyme function.
Comparative phylogenetic analysis does not provide specific functional information pertaining non-conserved amino acids, other than to suggest that non-conserved amino acids are not likely critical to core enzyme functions. For that reason, specific mutations were introduced into a recombinant gene encoding a variant of the Z05 DNA Polymerase ("Z05 D580G polymerase") and the resultant Z05 D580G polymerase mutants were screened for their ability to provide a reduced ability to extend an oligonucleotide primer with a 3′-mismatch to a template. Reichert et al. found that one such mutant, Z05 D580G V667E polymerase, displayed better discrimination (˜2-fold) than the parental Z05 D580G polymerase. See U.S. Pat. App. No. 2012/0015405 to Reichert et al., entitled DNA POLYMERASES WITH INCREASED 3′-MISMATCH DISCRIMINATION.
The comparative phylogenetic analysis has limitations with respect to identifying DNA polymerase activities that display improved 3′-nucleotide discrimination. This is due to the fact that all DNA polymerases of a given enzyme class are confronted with similar template substrates and nucleotide pools across the spectrum of phylogenetically diverse organisms. Given the fact that all DNA polymerases must display 3′-nucleotide mismatch discrimination to preserve high fidelity replication of daughter template strands, it is not surprising that one can apply comparative phylogenetic analysis to identify possible amino acid positions that might affect mismatch discrimination. For template substrates having different 3′-end modifications that are presented to a polymerase only in a biochemical assay, such as those used in several PCR-based assays for rare allele detection, there is a need for identifying DNA polymerases having improved 3′-nucleotide discrimination.
Taq DNA Polymerase is an enzyme that was discovered in Thermus aquaticus bacterium (Chien, A., et al., J Bacteriol. 1976, 127: 1550-1557). The enzyme is classified as deoxyribonucleic acid polymerase, class I (enzyme code, EC 126.96.36.199). Its catalytic activity is to amplify DNA sequences. The peptide and gene sequences of Taq DNA polymerase enzyme isolated from nature are well known in prior art and are listed in Table 1 (Lawyer, F. C., et al., J. Biol. Chem. 1989, 264: 6427-6437; Genbank database ID J04639.1).
TABLE 1DNA and amino acid sequence of native Taq DNA polymerase.TypeSequenceProteinMRGMLPLFEPKGRVLLVDGHHLAYRTFHALKGLTTSRGEPVQAVYGFAKSLLKALKEDsequenceGDAVIVVFDAKAPSFRHEAYGGYKAGRAPTPEDFPRQLALIKELVDLLGLARLEVPGY(N to CEADDVLASLAKKAEKEGYEVRILTADKDLYQLLSDRIHVLHPEGYLITPAWLWEKYGLterminus)RPDQWADYRALTGDESDNLPGVKGIGEKTARKLLEEWGSLEALLKNLDRLKPAIREKI[SEQ ID NO: 1]LAHMDDLKLSWDLAKVRTDLPLEVDFAKRREPDRERLRAFLERLEFGSLLHEFGLLESPKALEEAPWPPPEGAFVGFVLSRKEPMWADLLALAAARGGRVHRAPEPYKALRDLKEARGLLAKDLSVLALREGLGLPPGDDPMLLAYLLDPSNTTPEGVARRYGGEWTEEAGERAALSERLFANLWGRLEGEERLLWLYREVERPLSAVLAHMEATGVRLDVAYLRALSLEVAEEIARLEAEVFRLAGHPFNLNSRDQLERVLFDELGLPAIGKTEKTGKRSTSAAVLEALREAHPIVEKILQYRELTKLKSTYIDPLPDLIHPRTGRLHTRFNQTATATGRLSSSDPNLQNIPVRTPLGQRIRRAFIAEEGWLLVALDYSQIELRVLAHLSGDENLIRVFQEGRDIHTETASWMFGVPREAVDPLMRRAAKTINFGVLYGMSAHRLSQELAIPYEEAQAFIERYFQSFPKVRAWIEKTLEEGRRRGYVETLFGRRRYVPDLEARVKSVREAAERMAFNMPVQGTAADLMKLAMVKLFPRLEEMGARMLLQVHDELVLEAPKERAEAVARLAKEVMEGVYPLAVPLEVEVGIGEDWLSAKE DNA sequenceaagctcagat ctacctgcct gagggcgtcc ggttccagct ggcccttccc(5′ to 3′)gagggggaga gggaggcgtt tctaaaagcc cttcaggacg ctacccgggg[SEQ ID NO: 2]gcgggtggtg gaagggtaac atgaggggga tgctgcccct ctttgagcccaagggccggg tcctcctggt ggacggccac cacctggcct accgcaccttccacgccctg aagggcctca ccaccagccg gggggagccg gtgcaggcggtctacggctt cgccaagagc ctcctcaagg ccctcaagga ggacggggacgcggtgatcg tggtctttga cgccaaggcc ccctccttcc gccacgaggcctacgggggg tacaaggcgg gccgggcccc cacgccggag gactttccccggcaactcgc cctcatcaag gagctggtgg acctcctggg gctggcgcgcctcgaggtcc cgggctacga ggcggacgac gtcctggcca gcctggccaagaaggcggaa aaggagggct acgaggtccg catcctcacc gccgacaaagacctttacca gctcctttcc gaccgcatcc acgtcctcca ccccgaggggtacctcatca ccccggcctg gctttgggaa aagtacggcc tgaggcccgaccagtgggcc gactaccggg ccctgaccgg ggacgagtcc gacaaccttcccggggtcaa gggcatcggg gagaagacgg cgaggaagct tctggaggagtgggggagcc tggaagccct cctcaagaac ctggaccggc tgaagcccgccatccgggag aagatcctgg cccacatgga cgatctgaag ctctcctgggacctggccaa ggtgcgcacc gacctgcccc tggaggtgga cttcgccaaaaggcgggagc ccgaccggga gaggcttagg gcctttctgg agaggcttgagtttggcagc ctcctccacg agttcggcct tctggaaagc cccaaggccctggaggaggc cccctggccc ccgccggaag gggccttcgt gggctttgtgctttcccgca aggagcccat gtgggccgat cttctggccc tggccgccgccagggggggc cgggtccacc gggcccccga gccttataaa gccctcagggacctgaagga ggcgcggggg cttctcgcca aagacctgag cgttctggccctgagggaag gccttggcct cccgcccggc gacgacccca tgctcctcgcctacctcctg gacccttcca acaccacccc cgagggggtg gcccggcgctacggcgggga gtggacggag gaggcggggg agcgggccgc cctttccgagaggctcttcg ccaacctgtg ggggaggctt gagggggagg agaggctcctttggctttac cgggaggtgg agaggcccct ttccgctgtc ctggcccacatggaggccac gggggtgcgc ctggacgtgg cctatctcag ggccttgtccctggaggtgg ccgaggagat cgcccgcctc gaggccgagg tcttccgcctggccggccac cccttcaacc tcaactcccg ggaccagctg gaaagggtcctctttgacga gctagggctt cccgccatcg gcaagacgga gaagaccggcaagcgctcca ccagcgccgc cgtcctggag gccctccgcg aggcccaccccatcgtggag aagatcctgc agtaccggga gctcaccaag ctgaagagcacctacattga ccccttgccg gacctcatcc accccaggac gggccgcctccacacccgct tcaaccagac ggccacggcc acgggcaggc taagtagctccgatcccaac ctccagaaca tccccgtccg caccccgctt gggcagaggatccgccgggc cttcatcgcc gaggaggggt ggctattggt ggccctggactatagccaga tagagctcag ggtgctggcc cacctctccg gcgacgagaacctgatccgg gtcttccagg aggggcggga catccacacg gagaccgccagctggatgtt cggcgtcccc cgggaggccg tggaccccct gatgcgccgggcggccaaga ccatcaactt cggggtcctc tacggcatgt cggcccaccgcctctcccag gagctagcca tcccttacga ggaggcccag gccttcattgagcgctactt tcagagcttc cccaaggtgc gggcctggat tgagaagaccctggaggagg gcaggaggcg ggggtacgtg gagaccctct tcggccgccgccgctacgtg ccagacctag aggcccgggt gaagagcgtg cgggaggcggccgagcgcat ggccttcaac atgcccgtcc agggcaccgc cgccgacctcatgaagctgg ctatggtgaa gctcttcccc aggctggagg aaatgggggccaggatgctc cttcaggtcc acgacgagct ggtcctcgag gccccaaaagagagggcgga ggccgtggcc cggctggcca aggaggtcat ggagggggtgtatcccctgg ccgtgcccct ggaggtggag gtggggatag gggaggactggctctccgcc aaggagtgat accacc
Taq DNA polymerase extends primers composed from deoxyribonucleotides, however, some chemical modifications of the primer are tolerated and do not decrease much the efficiency of primer extension reactions. For example, when the nucleotide at 3′ primer terminus is ribonucleotide instead of deoxyribonucleotide, Taq DNA polymerase can extend such primer with significant efficiency and speed.

DNA polymerase mutants having enhanced template discrimination activity