A new system and method is developed for reducing the crest factor of a signal. The system includes a large signal extraction module for receiving the input signal and the magnitude of the input signal to extract a large signal greater than a predetermined threshold θ; a large signal transformation module for converting the extracted large signal to a monotonically increasing concave function; a large signal filtering module for filtering the large signal transformed by the large signal transformation module to pass a predetermined band of the large signal; a delay means for shifting the phase of the input signal; and a combiner means for combining the signal output from the large signal filtering module with the input signal whose phase has been shifted by the delay means to reduce the crest factor of the input signal.

1. Field of the Invention
The present invention relates to a system and a method for reducing the crest factor of a signal, and more particularly, to a system and a method for reducing the crest factor of a signal which can reduce the crest factor without having information about an individual channel and be applied to baseband, IF and RF signals.
2. Related Prior Art
A crest factor is an important characteristic of an RF signal in the transmission of the RF signal. The crest factor is a ratio of a peak value to an effective value of a signal waveform. That is, the crest factor is obtained by dividing a peak value of a signal by an effective rms value of the signal, and represents a degree of sharpness of a wave front. The crest factor is represented as follows.
                              CF                                (                          Crest                                                                                Factor                        )                          =                  PeakValue          RMSValue                                    [                  Equation                                                              1                ]
For example, a sine wave has a crest factor of √{square root over (2)} and a random wave can theoretically have a crest factor of an infinite value.
The crest factor is determined by a peak-to-average voltage ratio in the transmission of a multi-carrier of a digital signal. That is, the crest factor is proportional to the square root of a peak-to-average power ratio of a transmission signal.
Conventional techniques for reducing the crest factor of a signal include a clipping method and a baseband control method.
In the clipping method, a baseband signal has its amplitude clipped at a constant level. The clipping method removes the peak of a signal to reduce the crest factor of the signal. Though clipping is a simple technique because it is nonlinear processing, it should be carefully used. For example, the filter for filtering a clipped signal must be carefully selected.
The baseband control method controls the frequency band of an original signal before the modulation of the frequency band for the transmission of the signal, in order to reduce the crest factor of the signal. This method requires information about a transmission channel.
FIG. 1 is a block diagram of a conventional system and method for reducing the peak-to-average power ratio of a multi-carrier signal, which is disclosed in U.S. Pat. No. 6,314,146. In FIG. 1, reference numeral 840 denotes a receiver and a block 855 estimates a signal distorted by clipping in order to restore an original signal. This technique clips the original signal at transmission to reduce the peak-to-average power ratio of the signal. When the signal is clipped to a distorted form and transmitted to the receiver, the receiver makes an estimate of the clipped portions of the signal in order to restore the original signal. At this time, the receiver also receives information about signal clipping (clipping size, clipping peak and so on). This technique reduces the peak-to-average power ratio without having the loss of bandwidth of the original signal.

System and method for reducing crest factor of signal


First (210) and second (240) feedstocks are hydrotreated in an integrated hydrogenation plant (200) using a hot separator (230) that provides a vapor stream containing at least some of the hydrotreated first feedstock (210), wherein the second feedstock (240) is mixed with the vapor stream at a position downstream of the separator (240) and upstream of the a second hydrotreating (250) reactor to form a mixed second feedstock that is fed into the second hydrotreating reactor (250) to produce a ultra-low sulfur product.

Hydrotreating is a commonly used process in many modern refineries, in which hydrogen is contacted in the presence of a catalyst with a hydrocarbonaceous feedstock to remove impurities, including oxygen, nitrogen, sulfur, and unsaturated hydrocarbons. Consequently, hydrotreating is frequently employed to reduce the sulfur content from refined intermediates and is also commonly referred to as hydrodesulfurization. Hydrodesulfurization is typically used within a refinery in combination with processes including feed pretreatment of catalytic reformers, catalytic crackers, and hydrocrackers, product quality improvement for naphtha, diesel, jet, heating oil and residues, saturation of olefins, and polycyclic aromatics. There are numerous hydrotreating configurations and processes known in the art, and continuous efforts to reduce energy and material consumption led to integration of hydrotreating reactors in various processes.
For example, in one integration concept, a hydrotreater is combined with a hydrocracker as disclosed in U.S. Pat. No. 3,328,290 to Hengstebeck that describes a two-stage hydrocracking process wherein fresh feed is combined with effluent from the hydrocracking stage and the combined streams are then introduced into a hydrotreating stage. A higher-boiling fraction is then separated from the hydrotreater effluent and fractionated to produce a light product and a heavier-bottoms stream, which is then recycled with hydrogen-containing gas back to the hydrocracking stage.
Another example U.S. Pat. No. 6,235,190 to Bertram describes an integrated hydrotreating and hydrocracking process in which two hydrotreating catalysts of different activity are operated in series to provide improved products that are then subjected to a hydrocracking process to convert the hydrotreated effluent to lighter products with a reduced aromatic hydrocarbon content.
In a further example, as described in U.S. Pat. No. 6,261,441 to Gentry et al., a combined hydrotreating/hydrocracking process is described in which a hydrocracking stage is followed by a hydrodewaxing stage with a single feed and a bottoms fraction recycle to produce a naphtha product, a distillate boiling above the naphtha range, and a lubricant.
In yet another system, as described in U.S. Pat. No. 6,328,879 to Kalnes, two independent feedstocks are hydrocracked in a catalytic hydrocracking process that employs a hydrocracking zone, a hydrotreating zone, and a high pressure product stripper to produce various products from two feedstocks, wherein the products have a lower boiling point range than the feedatocks.
Alternatively, more than one hydrotreater reactor, and or catalyst beds may be employed for catalytic hydrogenation as described in U.S. Pat. No. 3,537,981 to Parker, or U.S. Pat. No. 6,103,105 to Cooper. While Parker's process employs a first hydrotreating reactor coupled to a separator that is in series with a second hydrotreating reactor, Cooper et al. employ two serially connected hydrotreating catalyst beds without the use of a separator. However, both Coopers and Parkers configuration are typically limited to only a single feedstock. Alternatively, as described in U.S. Pat. No. 5,958,218 to Hunter, two hydrocarbon feedstocks are hydrotreated in parallel while hydrogen flows in series between the reactors. While Hunters configuration allows hydrotreating of two at least somewhat different feedstocks, the catalytic reactors will generally operate at different hydrogen partial pressures. Consequently, the balance of the feedstocks will have to be properly balanced with the particular hydrogen partial pressures to yield the desired products.
Thus, although many integrated processes have provided at least some advantage over other known conligurations and methods, such configurations and methods are frequently limited to processes involving hydrocracking, or hydrotreating of a single boiling range (e.g., naphtha, diesel, gasoil, resid) feedetock. Consequently, all or almost all of the known hydrotreating processes require separate plants where more than one feedstock is employed. Therefore, there is still a need to provide improved configurations and methods for hydrotreating of petroleum products.


A magnetoresistive effect device includes an input port, an input-side signal line, an MR unit including a magnetoresistive effect element and a magnetic-field generating signal line, and an output unit including a magnetoresistive effect element, an output-side signal line, and an output port. The magnetoresistive effect device further includes a DC application terminal. The magnetoresistive effect element is connected to the output port via the output-side signal line in the output unit. The input-side signal line is arranged so that a high frequency magnetic field generated from the input-side signal line is applied to the magnetoresistive effect element in the MR unit. In the MR unit, the magnetoresistive effect element is connected to the magnetic-field generating signal line. The magnetic-field generating signal line is arranged so that a high-frequency magnetic field generated from magnetic-field generating signal line is applied to the magnetoresistive effect element in the output unit.

The speed of wireless communication has increased in recent years with the increasing functions of mobile communication terminals, such as mobile phones. Since the communication speed is proportional to the band width of frequencies that are used, the number of frequency bands necessary for communication is increased. Accordingly, the number of high-frequency filters that are required to be mounted in the mobile communication terminals is increased. Spintronics has been studied in recent years as a field that is probably applicable to new high-frequency components. One phenomenon that has received attention is the ferromagnetic resonance phenomenon, which is caused by a magnetoresistive effect element (refer to NPL 1). Application of an alternating magnetic field to a ferromagnetic film of the magnetoresistive effect element causes ferromagnetic resonance in magnetization of the ferromagnetic film, and the magnetization of the ferromagnetic film greatly oscillates at frequencies near a ferromagnetic resonance frequency. The ferromagnetic resonance frequency of the ferromagnetic film is generally within a high-frequency band from several gigahertz to several tens of gigahertz.


A process for applying alternating layers by chemical vapor deposition comprises the process steps of depositing an adhesion-promoter layer on a substrate and applying a barrier layer. Alternating layers comprising organic and inorganic materials are deposited alternately, and in this process the coating time for application of the adhesion-promoter layer is between 0.05 s and 4.0 s and the coating time for application of the inorganic barrier layer is between 0.1 s and 6.0 s. A composite material is also produced using the process.

The invention relates to a rapid process for the application of alternating layers by means of chemical vapor deposition processes, CVD, in particular by means of plasma enhanced chemical vapor deposition PECVD, especially plasma impulse chemical vapor deposition, PICVD, and to coatings which can be applied using this process.
To enable the permeability of packaging for gases and liquids to be reduced and to protect the packaging material from chemical attack or UV radiation, it is advantageous for this packaging to be provided with a barrier coating. In this context, by way of example, the deposition of thin SiOx coatings or coating systems on polymer substrates is of interest in particular for mass-produced plastics in order to reduce their permeability in particular to oxygen and water vapor and at the same time in particular to maintain the transparency of the material. Moreover, the contents of materials which have been coated in this way can generally be safely heated by means of microwave devices. A further advantage of these coatings lies in the large number of possible options for depositing them on polymer surfaces.
However, to improve the quality of coatings of this type, it is necessary to know the relationship between the barrier action, the morphological properties and the molecular transport through the layer, and in particular to control the influence of the deposition conditions on the coating parameters which result.

This patent is subject to a terminal disclaimer.

Rapid process for the production of multilayer barrier layers


An improved camouflage system is provided which consists of a garment worn by a person, and a structure is for producing an effective camouflage. A mechanism is for attaching the camouflage structure in slightly overlapping rows to the garment for concealment of a body part of the person wearing the garment which will create the illusion of trees, shrubs and other foliage in a three dimensional effect to blend into the terrain environment.

1. Field of the Invention
The instant invention relates generally to disruptive pattern materials and more specifically it relates to an improved camouflage system.
2. Description of the Prior Art
Numerous disruptive pattern materials have been provided in prior art that are adapted to include different colors and shades to blend into various types of terrains, so as to render the materials less easily distinguished by the eye when viewed against a background of natural foliage and the like. While these units may be suitable for the particular purpose to which they address, they would not be as suitable for the purposes of the present invention as heretofore described.


A gaming system samples information from a mobile device that is connected to a gaming network for use by at least one electronic gaming machine (EGM) included in the gaming network.

1. Field of the Invention
The present invention relates in general to gaming devices and systems, and more particularly to dynamically configuring a gaming device in a gaming system based on player mobile device (PMD) activity.
2. Description of the Related Art
Games of chance have been enjoyed by people for many years and have undergone increased and widespread popularity in recent times. As with most forms of entertainment, some players enjoy playing a single favorite game, while others prefer playing a wide variety of games. In response to the diverse range of player preferences, gaming establishments commonly offer many types of electronic games. Many electronic gaming machines (EGMs), such as slot machines and video poker machines, have been a cornerstone of the gaming industry for several years. The EGMs are computer based and contain multiple external interfaces for connecting with external devices and mobile devices.

Dynamically configuring electronic gaming machines based on player mobile device activity


A method of determining distances between communication nodes. The method includes transmitting an initial communication frame from a first node to at least one second node. The initial communication frame includes a frame preamble, first data and a first switch point. The method further includes synchronizing a local clock of the at least one second node based on the frame preamble. Switching the at least one second node to a transmitting node based on the received switch point. Transmitting a subsequent communication frame from the at least one second node to the first node. The subsequent communication frame includes a ranging preamble, second data and a second switch point. The method continues by synchronizing a local clock in the first node based on the ranging preamble from the at least one second node and determining the range between the first node and the second node based on the ranging preamble.

In many wireless networks, each node may have a need to determine the position of the node, relative to absolute and relative frames of reference. One system providing an absolute frame of reference is the Global Positioning System (GPS), using a constellation of geosynchronous satellites to provide the necessary coordinate reference points. Relative frames of reference are provided by asset tracking systems, such as those monitoring the movement of cooperating devices in a structure, for determining the building-relative position coordinates of a pre-mounted tagging device. Current systems for providing absolute and relative frames of reference typically require the cost and installation of infrastructure equipment, from satellites to fixed location beacons mounted in a structure, as used in many tracking systems.
Ad hoc wireless networks are characterized by the absence of installed infrastructure, and possibly unpredictable orientation and location of such devices. Such systems typically provide only a limited relative frame of reference, with respect to other devices in the ad hoc network. Certain ad hoc network nodes may be augmented with GPS to provide a fixed coordinate system, and hence an absolute frame of reference. However, the size and power requirements necessary for GPS are typically unavailable in energy-constrained ad hoc wireless devices. Also, the unpredictable orientation of the GPS antenna in a device, or a location without suitable GPS signals, may result in a failure to acquire sufficient GPS satellite signals for an accurate position determination by the node. Further, the ad hoc wireless network may only require a relative, not an absolute, frame of reference to perform the desired application, such as perimeter monitoring or surveillance.
In many position determination systems such as GPS, tagged asset tracking and ad hoc wireless networks, the propagation velocity of a radio frequency (RF) signal is used to measure the distance between pairs of cooperating nodes. In air, the approximate velocity of a radio wave is one foot per nanosecond (1 ft./nSec), and the distance between a transmitter and one or more cooperating receivers may be measured using methods such as Time of Flight (TOF) or Time Difference of Arrival (TDOA). The results of such ranging measurements between several pairs of nodes are required, at a minimum, for trilateration to determine the position of just a single device in systems with some fixed infrastructure. More generally, many such ranging measurements are required to perform multilateration, such as required to determine the position of each device in an ad hoc wireless network. Therefore, the accuracy of such ranging measurements is a function of the clock precision utilized for the ranging measurements.
The use of conventional, or data modulated RF bands or channels, for measuring TOF or TDOA may produce inaccurate results due to various influences and effects. Multi-path RF propagation may result in the receiver detecting a higher strength, but longer indirect path RF signal, and measuring a time interval with a larger value than for the direct RF wave path distance. Also, clock synchronization among the devices must be accomplished and maintained, typically through the use of a beacon signal transmitted by a designated node. Since such clocks are typically used for the purpose of providing a time base for demodulating the data, having a modulation frequency orders of magnitude lower than the carrier frequency, the clock is not able to resolve individual cycles of the carrier frequency. Hence, the precision of such ranging measurements is limited to an indirect detection of modulated data contained in a beacon signal, rather than arrival of an RF signal at the receiver. Further, each beacon signal will include a preamble appropriate for the data modulation technique employed
Some wireless data communications systems, such as conventional radio frequency systems, provide data communications by modulating, or coding, data signals onto a carrier frequency(s). However, other types of wireless communication systems are carrier-less and rely on time-based coding for data communications. One such communication system that relies on time-based coding to achieve reliable data communications is Ultra Wide Band ("UWB").
These UWB systems, unlike conventional radio frequency communications technology, do not use modulated carrier frequencies to transport data. Instead, UWB systems make use of a wide band energy pulse that transports data using both time-based coding and signal polarization. Time-based coding methods may include pulse-position, pulse-rate or pulse-width techniques. UWB communication systems do not provide a common clock to the transmitting and receiving nodes. Instead, a low-drift clock with a programmable offset value is implemented in each transmitter/receiver node, providing a local reference for time-based coding and decoding. Each of these multiple clock domains is subject to short-term time drift, which will exceed the necessary tolerance for accurate UWB data communication system operation after a predictable time period. As a result, precise time synchronization between the transmitting node and receiving node(s) is imperative in UWB systems to obtain accurate data communications. In order to precisely synchronize receiving node(s) with a transmitting node, UWB systems typically require preambles for each transmitted data frame. However, some applications with potential to benefit from UWB technology cannot be implemented if a preamble is required. Also, many potential applications for UWB technology are size and energy constrained, such as networks of wireless sensors and controls, which seek to minimize transmission time and to conserve energy.
In particular, wireless sensor networks might be composed of two or more communicating nodes. Further, each node may have an arbitrary location and antenna orientation, due for example, to the manner in which the nodes are emplaced. In particular, due to the ad hoc orientation of each sensor node, relative to other nodes, an omni-directional antenna is required for data communications. Further, each sensor node is typically battery-powered, and must use energy in the most efficient manner possible in order to achieve a useful lifespan for the intended application.
Ultra-WideBand (UWB) radio systems typically require a long synchronization period, or frame preamble, in order for one or more receivers (Rx) to accurately time-synchronize with the transmitter (Tx) and achieve successful digital data communications. Many applications considering UWB radios are size- and power-constrained, emphasizing the need to minimize energy consumption. Certain classes of applications require accurate determination of the distances between nodes in a wireless sensor network (i.e. multi-static radar, geo-location, etc.) For time-synchronized UWB data communications, each node's clock must have a precision in the sub-nanosecond range. This clock precision may also be used in the measurement of distance, based on TOF or TDOA measurements of UWB signals between nodes, with a sub-nanosecond clock precision providing distance measurement accuracy on the order of inches. However, each such measurement will incur the full length preamble sequence between a pair of nodes in order to perform the measurement and communicate the result. Since a wireless sensor network is typically required to exchange data as well as measuring distances, a method for achieving both, while minimizing energy consumption is essential to maximize the operational lifetime of a wireless sensor network node, many of which are battery-powered.
For the reasons stated above, and for other reasons stated below which will become apparent to those skilled in the art upon reading and understanding the present specification, there is a need in the communication industries for a method to provide collaboration among two or more transmitter/receiver nodes that utilizes the clock synchronization mechanism required for time-synchronous communication systems to provide highly accurate ranging measurements.


An apparatus for examining the electrochemical effects of (in vivo) metal implants causing allergic symptoms and/or inflamation in living organism, the apparatus containing two probes and a signal processing circuit connected thereto. One of the probes is a reference electrode (1) provided with reference electrolyte (34), which is connected to the soma tissue near the implant while the other probe is a measuring electrode (2) provided with a metal contact tip (6) to be contacted with the implant. The reference electrode (1) and the measuring electrode (2) are connected through an amplifier (20, 27) to one of the inputs of a comparing unit (22, 28). The other input of the comparing unit (22, 28) is connected the output of a memory (24, 30) containing data concerning the metal to be examined, and one of the outputs of the comparing unit (22, 28) is connected the display for the measured data (11).

1. Field of the Invention
The invention relates to an apparatus for examining the electrochemical effects of in vivo metal implants causing allergic symptoms and/or inflammation in a living organism.
2. Description of the Related Art
Recent research has demonstrated that metal implants in living organism can cause allergic symptoms as a result of ionic migration from the metal alloys implanted in vivo toward the living tissues. These migrated ions easily connect to protein bodies at the temperature of the human body. As a result, in the organism already implanted with metal, allergic symptoms can appear, i.e. local or long-distance mucosal and cutaneous reactions, eczema, dermatitis, dermatosis, etc. Under certain circumstances, this effect can cause extraordinary serious altercations, and the risk exists until the anaphylactogemic factor, i.e. the metal implants which can be dental prosthesis, filling of a tooth, screw, nail, etc., is removed from the organism.
HU-PS 192 218 describes a method and apparatus for exploring and examining such effects. By use of the mentioned apparatus and method, potential difference is measured between the implanted metal alloys and, at the same time, ion currents effected by the potential difference are measured in vivo in the neighboring soma tissues. For making the measurement, a specially formed measuring probe is applied, the current values are measured by a current/potential (I/U) converter connected to the measuring probe, and by a signal processing unit connected thereto, then displayed.
It has been found in practice that during the application of the mentioned known apparatus, the ion migration causing the harmful effects cannot be explored to the extent necessary. During the application, it has been also demonstrated that the measured values cannot be reproduced consistently, and that the measurements obtained are influenced by numerous parameters and by chance as well. Also, the measuring unit is embedded in a relatively large instrument case, so that its application becomes difficult. A further problem is that more than one person is required to perform the measurement, to accomplish suitable location of the probes and their connection, and for reading the measured values on the instrument.
WO 89100400 describes a solution where the reference electrode is integral part of the apparatus and, as such, the reference electrode cannot be releasably connected to the apparatus. Neither the reference electrode, nor the measuring electrode can be made as a disposable article.
A further serious disadvantage of both mentioned solutions is to meet the requirements of hygienic regulations, which, considering the spreading of the newer epidemic-like diseases, become increasingly strict with respect to handling of instruments and apparatuses used during medical treatment. In the case of this known solution, the sterilization of the apparatus embedded in the instrument cannot be operated in accordance with the regulations.

Apparatus for examining electrochemical effects of in vivo metal implants causing allergic symptoms and/or inflammation in a living organism


An image forming apparatus and method for controlling a charge of a toner are provided. The apparatus and method includes a toner supplying roller; a developing roller for supplying the toner to an organic photoconductor; a charge controller for adjusting the charge of the toner on the developing roller; a developing current detector for detecting a change in current for a developing voltage; and a charge controlling voltage controller for adjusting a charge controlling voltage according to the detection result. The apparatus can uniformly maintain the quality of an image by recognizing a change in the charge of toner based on a change in the environment. Also, the image forming apparatus indicates a proper toner replacement time by determining when the charge of the toner is abnormally low as a toner shortage state and displaying the toner shortage state on a panel operating unit.

1. Field of the Invention
The present invention relates to an image forming apparatus capable of controlling the charge of toner and a method thereof. More particularly, the present invention relates to an image forming apparatus and method for determining the charge of a toner from a development current with respect to a development voltage supplied to a developing roller and control the charge of the toner based thereon.
2. Description of the Related Art
Generally, an image forming apparatus adopting an electrophotographic method, such as a laser beam printer, which is also simply referred to as a laser printer, forms an image by forming an electrostatic latent image by scanning a laser beam to an organic photoconductor and attaching toner to the electrostatic latent image. Herein, the toner is typically charged with a negative (−) voltage to attach the toner to the organic photoconductor. The quality of the image formed in the image forming apparatus depends on the quantity of the toner attached to the organic photoconductor based on the charge of the toner.
FIG. 1 is a schematic diagram illustrating a conventional electrophotographic image forming apparatus.
The conventional image forming apparatus comprises a toner supplying roller 10, a developing roller 30, a blade 36, a charging roller 60, an organic photoconductor 50, and a fixing roller 70. The toner supplying roller 10 is charged with a high negative voltage, e.g., −500V, by a first power source 20 and provides toner to the developing roller 30. The developing roller 30 is charged with a high negative voltage, e.g., −300V, by a second power source 35, receives the toner attached to the toner supplying roller 10 by a potential difference from the toner supplying roller 10, and supplies the toner to an organic photoconductor (OPC) 50. The blade 36 maintains the uniform thickness of the toner attached to the developing roller 30. The charging roller 60 charges the organic photoconductor 50 with a high negative voltage, e.g., −700V, by a third power source 65. The organic photoconductor 50, which is charged by the charging roller 60, forms an electrostatic latent image in the part scanned by a laser scanning unit (LSU) 80 and attaches the toner to the electrostatic latent image. The fixing roller 70 fixes the toner attached to the electrostatic latent image which is formed in the organic photoconductor 50 on paper P. In the image forming apparatus with the above-described structure, the charge of the toner depends on a surrounding environment of the image forming apparatus, such as, a temperature and humidity. If the charge of the toner is increased, the quantity of the toner attached to the developing roller 30 is increased more than necessary and thus ghost images may be generated in the image printed on the paper P by the organic photoconductor 50 and the fixing roller 70, or the concentration of the toner on the print is increased more than necessary. If the toner has a lower charge, the toner layer attached to the developing roller 30 becomes so thin that it becomes a film on the surface of the developing film 30 or on the surface of the organic photoconductor 50. In this case, the quantity of the toner supplied to the organic photoconductor 50 is insufficient and the image printed on the paper P is indistinct or streaks are generated on the image.
FIG. 2 illustrates an example of the force acting on the toner between the organic photoconductor 50 and the developing roller 30.
The toner T is supplied to the organic photoconductor 50 by a potential difference between the developing roller 30 which is charged with about −300V and the organic photoconductor 50 which is charged with about −700V. Herein, a force Fd in which the toner T moves towards the organic photoconductor 50 is in proportion to the charge q of the toner and an electric field Ed between the developing roller 30 and the organic photoconductor 50. The force Fd can be expressed as an equation 1 below.Fd=q×Ed  Eq. 1
Meanwhile, a force Fr in which the toner T is attached to the developing roller 30 is a summation of a Van der Wall's force and an image force. The Van der Wall's force Fm is expressed as an equation 2 below.
                    Fm        =                              q            2                                4                        π            ×            ɛ                                                                                r              2                                                          Eq        .                                          2
wherein q denotes the charge of the toner, and εr denotes a dielectric rate of air.
Based on the equations 1 and 2, the charge of the toner T and the forces Fd and Fr according to the charge can be expressed as a graph shown in FIG. 3.
It can be seen from FIG. 3 that the toner T attached to the developing roller 30 is supplied to the organic photoconductor 50 in appropriate amounts at a charge q′ where the force Fd having a shape of a one-dimensional function and the force Fr having a shape of a two-dimensional function has the largest difference. The charge q′ has a different value according to electrophotographic image forming apparatuses. An electrophotographic image forming apparatuses produced by the Samsung Electronics company has a value of 12 to 20 uC/g. As shown in FIG. 3, if the toner T is charged less than 12 uC/g, or if the charge of the toner T is charged more than 20 uC/g, the quantity of the toner T attached to the developing roller 30 is decreased or increased, respectively, to an abnormal level and the toner T is not attached to the developing roller 30 uniformly and mechanical friction with the blade 36 which limits the thickness of the toner T attached to the developing roller 30 is increased. The increase in the mechanical friction between the blade 36 and the toner T can strip off the additives added to the toner T, such as SiO2 and TiO2. The additives are added to the toner T in the form of surrounding toner T particles to thereby prevent toner particles from being combined with each other and from being fused onto the organic photoconductor 50. If the additives are removed or damaged, the toner T can be fused onto the organic photoconductor 50 in the form of a film or degrades the quality of the image formed in the image forming apparatus.

Image forming apparatus controlling charge of toner and method thereof


The present disclosure relates to a method for semantic analysis performed by a computing device in the financial field. The method includes constructing a knowledge graph with a computer device. The method further includes obtaining, by the computer device, information published by the first user on a social network over the network. The method further includes the computer device generating standard information based on the information. The method further includes generating, by the computer device, a user behavior based on the standard information and the knowledge graph. The method further includes searching, by the computer device, for another user with similar user behavior based on the user behavior.

With the increasing popularity of the Internet, people are becoming more and more accustomed to publishing information on the Internet. With the increasing amount of information on the Internet, the credibility and accuracy of the information are becoming more and more uneven. More and more Internet companies want to evaluate the credibility of their users. For example, in some forums or blogs, some users' false information (including text, pictures, voice, video, etc.) often causes other users to blindly forward, which may harm the interests of others and generate a negative public opinion. The most common method of screening false information at present is to manually retrieve and filter customers' comments on the server, which is not only time-consuming and labour-intensive but also useless. Because only the contents of the false information and the single account publishing the statements are filtered, the user may continue to publish false statements by creating other accounts.
For example, in some web pages or forums that contain commodity transactions, sellers usually provide product information and their introductions. However, with the number of users increases, false product information and seller introductions make more and more customs be deceived. At present, the most common method is to manually screen and filter on the server side of the web pages or forums. However, this is not only time-consuming and labour-intensive, but also has the shortcomings such as incomplete acquisition of commodity information and large contingency of manual operation.
As another example, in some web pages or forums that contain user prediction information, such as football forums, stock forums, stock market evaluation pages, lottery forums, etc., users often have different prediction methods and prediction information. However, currently there is no system to evaluate the accuracy of the prediction based on a final real situation, and there is no system to mine user behavior hidden behind the information to associate or categorize users. That requires a system that can intelligently identify the prediction information published by users.

Systems and methods for semantic analysis based on knowledge graph


An integrated circuit includes a substrate and at least two circuits, such as a digital circuit and an analog circuit. The substrate is preferably derived from a bulk substrate wafer. The integrated circuit preferably comprises at least two islands in the substrate for noise isolation between the circuits. The two islands are buried-layers that are implanted, by preference, using conventional MeV techniques. A method of manufacturing an integrated circuit includes a substrate and at least two circuits. The method comprises the step of implanting at least two islands in the substrate for noise isolation between the circuits. The implanting is accomplished by conventional masking and high-energy implantation, such as MeV.

Noise isolation between unrelated circuits on an integrated circuit (IC) is highly desirable. This is particularly true when analog and digital circuits are on the same IC. Analog circuits are often processing minimal signals (<10 mV) and cannot tolerate interference generated by digital circuits on the same IC. FIG. 1 illustrates one of the mechanisms of coupling between separate functions on the same IC.
When digital circuitry 10 switches between high and low states, the change in the inputs and outputs of digital circuitry 10 create current impulses into the common substrate 15 via capacitor C2. Capacitor C2 represents multiple capacitors that are each associated with transistors within digital circuitry 10. The transistors are used to construct logic devices, such as inverters, NOR gates and AND gates. The current impulses flow through substrate 15 through resistors R1, R2 and R3 into ground connections 20, 25 of the IC.
The current flow through resistors R1, R2 and R3 causes voltage noise that is coupled to analog circuitry 30 through capacitor C1. This noise coupling mechanism causes noise within analog circuitry 30 which limits the minimal signal voltage that can be processed by analog circuitry 30 when provided on the same IC as digital circuitry 10.
The distribution of the current flow, and hence the amount of coupled noise, is dependent on the values of R1, R2 and R3. When RI is large and R3 is small relative to each other, the majority of current will flow through R2 and the noise is isolated from coupling to analog circuitry 30 by the effective voltage divider R3/(R1+R3). As R1 is decreased relative to R2 and R3, more noise voltage is coupled to analog circuitry 30. This typically occurs when substrate 15 is an epitaxial substrate that is used for IC processing.
Currently, two basic substrate architectures exist for standard CMOS processing: bulk substrate wafers and epitaxial layered wafers. Bulk substrate wafers consist of silicon that is uniformly doped to a constant carrier concentration. Epitaxial layered wafers have a base that is a heavily doped layer and a lightly doped epitaxial layer. The light doping of the epitaxial layer emulates the surface background carrier concentration similar to that of the bulk substrate wafers. The heavily doped layer provides a low impedance connection (R1), and thus more severe noise coupling, between analog circuitry 30 and digital circuitry 10.
Many alternatives exist to decrease the noise coupling through a common substrate. One alternative is to manufacture digital and analog circuits on separate ICs. This adds cost due to separate packaging and increased pin counts. Another alternative utilizes special processing steps, such as dielectric or junction isolation, that physically isolate unrelated circuits. However, these steps are costly compared to standard CMOS processes. Other alternatives rely on circuit design techniques, such as current-mode logic, or spacing of the unrelated circuits, but circuit noise isolation is still difficult to achieve using these alternatives.

Circuit isolation utilizing MeV implantation


The present invention permits a combination of a scalable audio coder with the TNS technique. In a method for coding time signals sampled in a first sampling rate, second time signals are first generated whose sampling rate is smaller than the first sampling rate. The second time signals are then coded according to a first coding algorithm and written into a bit stream. The coded second time signals are, however, decoded again, and, like the first time signals, transformed into the frequency domain. From a spectral representation of the first time signals, TNS prediction coefficients are calculated. The transformed output signal of the coder/decoder with the first coding algorithm, like the spectral representation of the first time signal, undergoes a prediction over the frequency to obtain residual spectral values for both signals, though only the prediction coefficients calculated on the basis of the first time signals are used. These two signals are evaluated against each other. The evaluated residual spectral values are then coded by means of a second coding algorithm to obtain coded evaluated residual spectral values, which, together with the side information containing the calculated prediction coefficients, are written into the bit stream.

The present invention relates to scalable audio coders and audio decoders and in particular to scalable coders and decoders for which at least one stage operates in the frequency domain.
Scalable audio coders are coders of modular design. An effort is therefore made to use already existing speech coders, which process signals which e.g. are sampled with 8 kHz and produce data rates of e.g. 4.8 to 8 kilobits per second. These known coders, such as e.g. the coders G.729, G.723, FS1016, CELP or parametric models for MPEG-4-Audio, which are known to persons skilled in the art, serve primarily for coding speech signals and are not generally suitable for coding higher quality music signals since they are normally designed for signals sampled with 8 kHz, so that they can only code an audio bandwidth of 4 kHz at the most. In general, however, they exhibit a low sampling rate and good quality for speech signals.
For the audio coding of music signals, e.g. to achieve HIFI quality or CD quality, with a scalable coder a speech coder is therefore combined with an audio coder, which can code signals with higher sampling rates, e.g. 48 kHz. Obviously it is also possible to replace the speech coder cited above by another coder, e.g. by a music/audio coder according to the Standards MPEG1, MPEG2 or MPEG4.
A chain circuit of this kind comprises a speech coder and a higher quality audio coder. An input signal, having a sampling rate of 48 kHz e.g., is converted by means of a downsampling filter to the appropriate sampling frequency for the speech coder. The sampling rate could, however, also be the same in both coders. The converted signal is then coded. The coded signal can be supplied directly to a bit stream formatting device for transmission. However, it only contains signals with a bandwidth of e.g. 4 kHz at the most. The coded signal is also decoded again and converted by means of an upsampling filter. Because of the downsampling filter, however, the signal now obtained only contains useful information with a bandwidth of e.g. 4 kHz. In addition it must be recorded that the spectral content of the converted coded/decoded signal in the lower band to 4 kHz does not correspond exactly to the first 4 kHz band of the input signal sampled with 48 kHz since in general coders introduce coding errors.
As has already been mentioned, a scalable coder comprises a generally known speech coder and an audio coder which can process signals with higher sampling rates. To be able to transmit signal components of the input signal which have frequencies above 4 kHz, the difference between the input signal at 8 kHz and the coded/decoded converted output signal of the speech coder is formed for each individual discrete-time sampled value. This difference can then be quantized and coded using a known audio coder, as is known to persons skilled in the art. It should be pointed out here that, apart from coding errors, the difference signal which is fed to the audio coder, which can code signals with higher sampling rates, is essentially zero in the lower frequency range. In the spectral range lying above the bandwidth of the upward converted coded/decoded output signal of the speech coder, the difference signal substantially corresponds to the true input signal at 48 kHz.
In the first stage, i.e. the speech coder stage, a coder with low sampling frequency is therefore generally used, since in general a very low bit rate of the coded signal is aimed at. At the present time a number of coders, including the cited coders, work with bit rates of a few kilobits (two to 8 kilobits or also more). Furthermore, these enable a maximum sampling frequency of 8 kHz, since more audio bandwidth is not possible anyway at this low bit rate and the coding at low sampling frequency is more advantageous as regards the computational effort. The maximum possible audio bandwidth is 4 kHz and in practice it is restricted to about 3.5 kHz. If a bandwidth improvement is to be achieved in the further stage, i.e. in the stage with the audio coder, this further stage must work with a higher sampling frequency.
The use of the so-called TNS technique in high quality audio coding to further reduce the amount of data has been known for some time (J. Herre, J. D. Johnston, xe2x80x9cEnhancing the Performance of Perceptual Audio Coders by Using Temporal Noise Shaping (TNS)xe2x80x9d, 101st AES Convention, Los Angeles 1996, Preprint 4384). The TNS technique (TNS=Temporal Noise Shaping), generally speaking, permits temporal shaping of the fine structure of the quantization noise by means of a predictive coding of the spectral values. The TNS technique is based on a consistent application of the dualism between the time domain and the frequency domain. In the technical field it is known that when the autocorrelation function of a time signal is transformed into the frequency domain it gives the spectral power density of this very time signal. The dual case hereto results when the autocorrelation function of the spectrum of a signal is formed and transformed into the time domain. The autocorrelation function transformed into or back into the time domain is also called the square of the Hilbert envelope curve of the time signal. The Hilbert envelope curve of a signal is thus connected directly with the autocorrelation function of its spectrum. The squared Hilbert envelope curve of a signal and the spectral power density of the same thus represent dual aspects in the time domain and in the frequency domain. If the Hilbert envelope curve of a signal remains constant for each partial bandpass signal over a range of frequencies, then the autocorrelation between neighbouring spectral values will also be constant. This means in fact that the series of spectral coefficients is stationary versus frequency, so that predictive coding techniques can be used efficiently to represent this signal and this, furthermore, by using a common set of prediction coefficients.
To clarify the situation, reference is made to FIG. 6A and FIG. 6B. FIG. 6A shows a short section of a temporally strongly transient xe2x80x9ccastanetxe2x80x9d signal with a duration of about 40 ms. This signal was decomposed into a multiplicity of partial bandpass signals, each partial bandpass signal having a bandwidth of 500 Hz. FIG. 6B now shows the Hilbert envelope curves for these bandpass signals with middle frequencies ranging from 1500 Hz to 4000 Hz. To make things clearer, all the envelope curves have been normalized to their maximum amplitude. Clearly the shapes of all the single envelope curves are very similar to one another, which is why a common predictor can be used within this frequency range to code the signal efficiently. Similar observations can be made for speech signals in which the effect of the glottal excitation pulses is present over the whole frequency range because of the nature of the human speech generation mechanism.
FIG. 6B thus shows that the correlation of neighbouring values e.g. at a frequency of 2000 Hz is similar to that at a frequency of e.g. 3000 Hz or 1000 Hz.
Alternatively, the property of spectral predictability of transient signals can be understood by considering the table shown in FIG. 5. At the top left of the table a continuous time signal u(t) is shown in the form of a sine wave. Next to this is the spectrum U(f) of this signal, consisting of a single Dirac pulse. The optimal coding of this signal consists in the coding of spectral data or spectral values since, for the complete time signal, only the magnitude and the phase of the Fourier coefficient have to be transmitted here in order to be able to reconstruct the time signal completely. A coding of spectral data corresponds at the same time to a prediction in the time domain. A predictive coding would thus have to take place here in the time domain. The sinusoidal time signal thus has a flat temporal envelope curve, which corresponds to a maximally non-flat envelope curve in the frequency domain.
The opposite case will now be considered in which the time signal u(t) is a maximally transient signal in the form of a Dirac pulse in the time domain. A Dirac pulse in the time domain corresponds to a xe2x80x9cflatxe2x80x9d power spectrum, while the phase spectrum rotates according to the time position of the pulse. It is clear that this signal poses a problem for the traditional methods cited above, such as e.g. the transform coding or coding of spectral data or a linear prediction coding of the time domain data. This signal can be coded best and most effectively in the time domain, since only the temporal position and the power of the Dirac pulse have to be transmitted, which, through consistent use of the dualism, means that a predictive coding in the frequency domain also constitutes a suitable method for efficient coding.
It is very important not to confuse the predictive coding of spectral coefficients over the frequency with the known dual concept of the prediction of spectral coefficients from one block to the next, which has already been implemented and is also described in the article cited above (M. Bosi, K. Brandenburg, S. Quakenbusch, L. Fiedler, K. Akagiri, H. Fuchs, M. Dietz, J. Herre, G. Davidson, Yoshiaki Oikawa: xe2x80x9cISO/IEC MPEG-2 Advanced Audio Codingxe2x80x9d, 101st AES Convention, Los Angeles 1996, Preprint 4382). In the prediction of spectral coefficients from one block to the next, which corresponds to a prediction over the time, the spectral resolution is increased, whereas a prediction of spectral coefficients over the frequency increases the temporal resolution. A spectral coefficient at e.g. 1000 Hz can therefore be determined from the spectral coefficient at e.g. 900 Hz in the same block or frame.
The above considerations thus led to the attainment of an efficient coding method for transient signals. Predictive coding techniques can, taking account of the duality between time and frequency domain, substantially be handled analogously to the already known prediction from a spectral coefficient to the spectral coefficient with the same frequency in the next block. Since the spectral power density and the squared Hilbert envelope curve of a signal are dual to each other, a reduction in a residual signal energy or a prediction gain is obtained which depends on the measure of flatness of the squared envelope curve of the signal and not on the spectral measure of flatness as in the conventional prediction method. The potential coding gain increases as the signals become more transient.
Both the prediction scheme with closed loop, also known as backward prediction, and the prediction scheme with open loop, also known as forward prediction, offer themselves as possible prediction schemes. In the case of the spectral prediction scheme with closed loop (backward prediction) the envelope curve of the error is flat. Expressed differently, the error signal energy is distributed uniformly over the time.
In the case of a forward prediction, however, as shown in FIG. 7, there is a temporal shaping of the noise introduced by quantization. A spectral coefficient to be predicted x(f) is fed to a summation point 600. The same spectral coefficient is also fed to a predictor 610, the output signal of which is also fed, with negative sign, to the summation point 600. The input signal to a quantizer 620 thus represents the difference between the spectral value x(f) and the spectral value xp(f) calculated by prediction. For forward prediction the total error energy in the decoded spectral coefficient data will remain constant. The temporal shape of the quantization error signal will, however, appear temporally shaped at the output of the decoder since the prediction was applied to the spectral coefficients, whereby the quantization noise is temporally placed under the actual signal and can thus be masked in this way problems of temporal masking e.g. for transient signals or speech signals are avoided.
This type of predictive coding of spectral values is thus called the TNS or temporal noise shaping technique. For clarification of this technique reference is made to FIG. 8A. At the top left in FIG. 8A the temporal behaviour of a strongly transient time signal is shown. Shown opposite this temporal behaviour curve at the top right in FIG. 8A is the section of a DCT spectrum. The graph at the bottom left in FIG. 8A shows the resulting frequency response of a TNS synthesis filter which was calculated by the LPC operation (LPC=Linear Prediction Coding). It should be noted that the (normalized) frequency coordinates in this diagram correspond to the time coordinates due to the time domain and frequency domain dualism. The LPC calculation obviously produces a xe2x80x9csource modelxe2x80x9d of the input signal, since the frequency response of the LPC-calculated synthesis filter resembles the envelope curve of the strongly transient time signal. A representation of the residual spectral values, i.e. of the input signal of the quantizer 620 in FIG. 7, over the frequency is shown at the bottom right in FIG. 8A. A comparison of the residual spectral values after prediction and the spectral values obtained with direct time-frequency transform shows that the residual spectral values have far less energy than the original spectral values. In the example shown the reduction in the energy of the residual spectral values corresponds to a total prediction gain of about 12 dB.
The following points should be noted in connection with the graph at the bottom left in FIG. 8A. For classical use of prediction on time domain signals, the frequency response of the synthesis filter is an approximation of the magnitude spectrum of the input signal. The synthesis filter (re)generates to some extent the spectral shape of the signal from a residual signal with an approximately xe2x80x9cwhitexe2x80x9d spectrum. When prediction is used on spectral signals as in the case of the TNS technique, the frequency response of the synthesis filter is an approximation of the envelope curve of the input filter. The frequency response of the synthesis filter is not the result of the Fourier transform of the pulse response, as in the classical case, but the result of the inverse Fourier transform. The TNS synthesis filter (re)generates so to speak the form of the envelope curve of the signal from a residual signal with an approximately xe2x80x9cwhitexe2x80x9d (i.e. flat) envelope curve. Thus the graph at the bottom left in FIG. 8A shows the input signal envelope curve as modelled by the TNS synthesis filter. This is here a logarithmic representation of the envelope curve approximation of the castanet signal shown in the figure above it.
Subsequently a coding noise was introduced into the residual spectral values such that a signal/noise ratio of about 13 dB resulted in each coding band with a width of e.g. 0.5 Bark. The error signals in the time domain resulting from the introduction of the quantization noise are shown in FIG. 8B. The left-hand diagram in FIG. 8B shows the error signal due to the quantization noise when using the TNS technique, while in the diagram on the right the TNS technique has not been used, thus providing a comparison. As expected, the error signal in the left-hand diagram is not distributed evenly over the block but is concentrated in the region in which there is a higher signal content, which will optimally mask this quantization noise in the right-hand case, on the other hand, the introduced quantization noise is distributed evenly over the block, i.e. over the time, with the result that in the region at the front where there is no signal, or scarcely so, noise is also present, which will be audible, while in the region in which there is a high signal content there is relatively little noise, through which the masking possibilities of the signal are not completely exploited.
A simple, i.e. non-scalable, audio coder with a TNS filter is described in the following.
An implementation of a TNS filter 804 in a coder is shown in FIG. 9A. This filter is arranged between an analysis filter bank 802 and a quantizer 806. The discrete-time input signal for the coder shown in FIG. 9A is fed into an audio input 800, while the quantized audio signal, i.e. the quantized spectral values, or the quantized residual spectral values are output at an output 808, which may be followed by a redundancy coder. The input signal is therefore transformed into spectral values. On the basis of the calculated spectral values a normal linear prediction computation is performed, e.g. by forming the autocorrelation matrix of the spectral values and using a Levinson-Durbin recursion. FIG. 9B shows a detailed view of the TNS filter 804. The spectral values x(1), . . . , x(i), . . . , x(n) are fed in at a filter input 810. It may happen that only a certain frequency range exhibits transient signals, whereas another frequency range is primarily of a stationary nature. This fact is allowed for in the TNS filter 804 through an input switch 812 and an output switch 814, though the primary function of these switches is to provide parallel-to-serial or serial-to-parallel conversion of the data to be processed.
If a certain frequency range is unsteady and promises a certain coding gain through the TNS technique, then only this spectral range is TNS processed, this being achieved in that the input switch 812 starts at the spectral value x(i) for example and sweeps through to the spectral value x(i+2) for instance. The inner region of the filter again comprises the forward prediction structure, i.e. the predictor 610 and the summation point 600.
The calculation for determining the filter coefficients of the TNS filter or for determining the prediction coefficients is performed as follows. The formation of the autocorrelation matrix and the application of the Levinson-Durbin recursion is performed for the highest permissible order of the noise shaping filter, e.g. 20. If the calculated prediction gain exceeds a certain threshold, TNS processing is activated.
The order of the employed noise shaping filter for the current block is then determined by subsequent removal of all coefficients with a sufficiently small absolute value from the end of the coefficient array. This results in the orders of TNS filters having values which normally lie in the range 4-12 for a speech signal.
If a sufficiently high coding gain is determined for a range of spectral values x(i) for example, this range is processed and the residual spectral value xR(i) appears instead of the spectral value x(i) at the output of the TNS filter. This residual value has a much smaller amplitude than the original spectral value x(i), as can be seen from FIG. 8A. In addition to the normal side information, the side information transmitted to the decoder thus contains a flag showing the use of TNS and, if necessary, information on the destination frequency range and on the TNS filter which was used for coding. The filter data can be represented as quantized filter coefficients.
In analogy to the coder with TNS filter, a decoder having an inverse TNS filter will now be considered.
In the decoder, which is sketched in FIG. 10A, a TNS coding is reversed for each channel. Residual spectral values xR(i) are requantized in the inverse quantizer 216 and fed into an inverse TNS filter 900 whose construction is shown in more detail in FIG. 10B. As output signal the inverse TNS filter 900 delivers spectral values again, which are transformed into the time domain in a synthesis filter bank 218. The TNS filter 900 includes an input switch 902 and an output switch 908, which again serve chiefly to provide parallel-to-serial or serial-to-parallel conversion of the data to be processed. The input switch 902 also takes account of a possible destination frequency range so as to subject only residual spectral values to inverse TNS coding whereas spectral values which are not TNS coded are allowed to pass through unchanged to an output 910. The inverse prediction filter comprises a predictor 906 and a summation point 904. In contrast to the TNS filter, however, these are connected as follows. A residual spectral value arrives via the input switch 902 at the summation point 904, where it is summed together with the output signal of the predictor 906. As output signal the predictor supplies an estimated spectral value xp(i). The spectral value x(i) is output at the output of the inverse TNS filter via the output switch. The TNS-related side information is thus decoded in the decoder, the side information including a flag indicating the use of TNS and, if necessary, information concerning the destination frequency range. In addition, the side information contains the filter coefficients of the prediction filter which was used to code a block or xe2x80x9cframexe2x80x9d.
The TNS method may thus be summarized as follows. An input signal is transformed into a spectral representation by means of a high-resolution analysis filter bank. A linear prediction is then performed in the frequency domain between spectral values which are neighbours as regards frequency. This linear prediction can be interpreted as a filter process for filtering the spectral values which is performed in the spectral domain. In this way the original spectral values are replaced by the prediction errors, i.e. by the residual spectral values. These residual spectral values, quantized and coded just like normal spectral values, are transferred to the decoder, where the values are decoded and inversely quantized. Before using the inverse filter bank (synthesis filter bank) an inverse prediction, inverse that is to the prediction carried out in the coder, is performed in that the inverse prediction filter is employed on the transmitted prediction error signal, i.e. on the requantized residual spectral values.
By employing this technique it is possible to match the temporal envelope curve of the quantization noise to that of the input signal. This permits better exploitation of the masking of the error signals for signals with a pronounced temporal fine structure or a pronounced transient behaviour. In the case of transient signals the TNS technique avoids the so-called xe2x80x9cpre-echosxe2x80x9d, for which the quantization noise already appears prior to the xe2x80x9cattackxe2x80x9d of such a signal.
As has already been mentioned, in a scalable audio coder a coder with a low sampling frequency is employed in the first stage since a very low bit rate of the coded signal is generally sought. In the second stage there is preferably an audio coder, which codes at higher bit rates but requires a much larger bandwidth and can thus code audio signals with much higher sound quality than the speech coder can. Normally an audio signal which is to be coded and which has resulted at a high sampling rate is first down-converted to a lower sampling rate, e.g. by using a downsampling filter. The reduced sampling rate signal is then fed into the coder of the first stage, the output signal of this coder being written directly into the bit stream which emerges from the scalable audio coder. This coded signal with lower bandwidth is decoded again and is then brought back to the high sampling rate, e.g. by using an upsampling filter, and is then transformed into the frequency domain. Also transformed into the frequency domain is the audio signal originally present at the input of the coder. Two audio signals are now available, the first of them suffering from the coding errors of the coder of the first stage, however. These two signals in the frequency domain can now be fed to a difference element to obtain a signal which represents only the difference between the two signals. In a switching module, which can also be implemented as a frequency selective switch, as is described later, it is possible to determine whether it is better subsequently to process the difference between the two input signals or, instead, to process directly the original audio signal transformed into the frequency domain. In any case the output signal of the switching module is fed to a known quantizer/coder, for example, which, if it functions according to an MPEG standard, performs both a quantization taking account of a psychoacoustic model and then subsequently an entropy coding, preferably using Huffman coding with the quantized spectral values. The output signal of the quantizer and coder is written into the bit stream together with the output signal of the coder of the first stage. At first sight it may seem to be a good idea to place the TNS filter described at the beginning directly behind the switching module, i.e. in front of the quantizer/coder, in order to simply imitate the structure shown in FIG. 10A. A disadvantage of this solution, however, is that the output signal of the switching module is greatly changed in relation to the original temporal audio signal at the input of the coder, with the result that a filter coefficient determination for the TNS filter is not applicable with the same quality.
It is the object of the present invention to combine the concept of scalable audio coding and the concept of temporal noise shaping so as to benefit from temporal noise shaping in the case of scalable audio coders as well.
In accordance, with a first embodiment of the present invention, this object is achieved by a method for coding discrete first time signals which have been sampled with a first sampling rate, comprising the following steps: generating second time signals, whose bandwidth corresponds to a second sampling rate, from the first time signals, the second sampling rate being equal to or less than the first sampling rate; coding the second time signals according to a first coding algorithm to obtain coded second signals; decoding the coded second signals according to the first coding algorithm to obtain coded/decoded second time signals whose bandwidth corresponds to the second sampling frequency; transforming the first time signals into the frequency domain to obtain first spectral values; calculating prediction coefficients from the first spectral values; generating second spectral values from coded/decoded second time signals, the second spectral values being a representation of the coded/decoded second time signals in the frequency domain; evaluating the first spectral values with the second spectral values to obtain evaluated spectral values whose number corresponds to the number of the first spectral values; performing a prediction of the evaluated spectral values over the frequency by means of the calculated prediction coefficients to obtain evaluated residual spectral values; and coding the evaluated residual spectral values according to a second coding algorithm to obtain coded evaluated residual spectral values.
In accordance with a second embodiment of the present invention, this object is achieved by a method for coding discrete first time signals which have been sampled with a first sampling rate, comprising the following steps: generating second time signals, whose bandwidth corresponds to a second sampling rate, from the first time signals, the second sampling rate being equal to or less than the first sampling rate; coding the second time signals according to a first coding algorithm to obtain coded second signals; decoding the coded second signals according to the first coding algorithm to obtain coded/decoded second time signals whose bandwidth corresponds to the second sampling frequency; transforming the first time signals into the frequency domain to obtain first spectral values; calculating prediction coefficients from the first spectral values; generating second spectral values from coded/decoded second time signals, the second spectral values being a representation of the coded/decoded second time signals in the frequency domain; performing a prediction of the first spectral values and the second spectral values over the frequency to obtain first residual spectral values and second residual spectral values, using the calculated prediction coefficients; evaluating the first residual spectral values with the second residual spectral values to obtain evaluated residual spectral values whose number corresponds to the number of the first spectral values; and coding the evaluated residual spectral values according to a second coding algorithm to obtain coded evaluated residual spectral values.
In accordance with a third embodiment of the present invention, this object is achieved by a method for decoding a bit stream which represents an audio signal, where the bit stream has signals coded according to a first coding algorithm, signals coded according to a second coding algorithm, and side information, where the signals coded according to the second coding algorithm have coded residual spectral values, where the residual spectral values are generated from evaluated spectral values by prediction over the frequency, where prediction coefficients of the prediction are present in the side information, comprising the following steps: decoding the coded signals which have been coded according to the first coding algorithm to obtain coded/decoded second time signals by means of the first coding algorithm; decoding the coded residual spectral values by means of the second coding algorithm to obtain the residual spectral values; transforming the coded/decoded second time signals into the frequency domain to obtain the second spectral values; performing an inverse prediction with the evaluated residual spectral values using the prediction coefficients which are present in the side information to obtain the evaluated spectral values; inversely evaluating the evaluated spectral values and the second spectral values to obtain the first spectral values; and transforming the first spectral values back into the time domain to obtain first time signals.
In accordance with a fourth embodiment of the present invention, this object is achieved by a method for decoding a bit stream which represents an audio signal, where the bit stream has signals coded according to a first coding algorithm, signals coded according to a second coding algorithm, and side information, where the signals coded according to the second coding algorithm have coded residual spectral values, where the residual spectral values are generated from evaluated spectral values by prediction over the frequency, where prediction coefficients of the prediction are present in the side information, comprising the following steps: decoding the coded signals which have been coded according to the first coding algorithm to obtain coded/decoded second time signals by means of the first coding algorithm; decoding the coded residual spectral values by means of the second coding algorithm to obtain the residual spectral values; transforming the coded/decoded second time signals into the frequency domain to obtain the second spectral values; performing a prediction with the second spectral values using the prediction coefficients which are present in the side information to obtain second residual spectral values; inversely evaluating the evaluated residual spectral values and the second residual spectral values to obtain the residual spectral values; performing an inverse prediction with the residual spectral values using the prediction coefficients which are stored in the side information to obtain first spectral values; and transforming the first spectral values back into the time domain to obtain first time signals.
In accordance with a fifth embodiment of the present invention, this object is achieved by an apparatus for coding discrete first time signals which have been sampled with a first sampling rate, comprising: a device for generating second time signals, whose bandwidth corresponds to a second sampling rate, from the first time signals, the second sampling rate being equal to or less than the first sampling rate; a first coder for coding the second time signals according to a first coding algorithm to obtain coded second signals; a decoder for decoding the coded second signals according to the first coding algorithm to obtain coded/decoded second time signals whose bandwidth corresponds to the second sampling frequency; a transformer for transforming the first time signals into the frequency domain to obtain first spectral values; a calculator for calculating prediction coefficients from the first spectral values; a device for generating second spectral values from coded/decoded second time signals, the second spectral values being a representation of the coded/decoded second time signals in the frequency domain; a device for evaluating the first spectral values with the second spectral values to obtain evaluated spectral values whose number corresponds to the number of the first spectral values; a predictor for performing a prediction of the evaluated spectral values over the frequency by means of the calculated prediction coefficients to obtain evaluated residual spectral values; and a second coder for coding the evaluated residual spectral values according to a second coding algorithm to obtain coded evaluated residual spectral values.
In accordance with a sixth embodiment of the present invention, this object is achieved by an apparatus for coding discrete first time signals which have been sampled with a first sampling rate, comprising: a device for generating second time signals, whose bandwidth corresponds to a second sampling rate, from the first time signals, the second sampling rate being equal to or less than the first sampling rate; a first coder for coding the second time signals according to a first coding algorithm to obtain coded second signals; a decoder for decoding the coded second signals according to the first coding algorithm to obtain coded/decoded second time signals whose bandwidth corresponds to the second sampling frequency; a transformer for transforming the first time signals into the frequency domain to obtain first spectral values; a calculator for calculating prediction coefficients from the first spectral values; a device for generating second spectral values from coded/decoded second time signals, the second spectral values being a representation of the coded/decoded second time signals in the frequency domain; a predictor for performing a prediction of the first spectral values and the second spectral values over the frequency to obtain first residual spectral values and second residual spectral values, using the calculated prediction coefficients; a device for evaluating the first residual spectral values with the second residual spectral values to obtain evaluated residual spectral values whose number corresponds to the number of the first spectral values; and a second coder for coding the evaluated residual spectral values according to a second coding algorithm to obtain coded evaluated residual spectral values.
In accordance with a seventh embodiment of the present invention, this object is achieved by an apparatus for decoding a bit stream which represents an audio signal, where the bit stream has signals coded according to a first coding algorithm, signals coded according to a second coding algorithm, and side information, where the signals coded according to the second coding algorithm have coded residual spectral values, where the residual spectral values are generated from evaluated spectral values by prediction over the frequency, where prediction coefficients of the prediction are present in the side information, comprising: a first decoder for decoding the coded signals which have been coded according to the first coding algorithm to obtain coded/decoded second time signals by means of the first coding algorithm; a second decoder for decoding the coded residual spectral values by means of the second coding algorithm to obtain the residual spectral values; a transformer for transforming the coded/decoded second time signals into the frequency domain to obtain the second spectral values; an inverse predictor for performing an inverse prediction with the evaluated residual spectral values using the prediction coefficients which are present in the side information to obtain the evaluated spectral values; a device for inversely evaluating the evaluated spectral values and the second spectral values to obtain the first spectral values; and an inverse transformer for transforming the first spectral values back into the time domain to obtain first time signals.
In accordance with an eight embodiment of the present invention, this object is achieved by an apparatus for decoding a bit stream which represents an audio signal, where the bit stream has signals coded according to a first coding algorithm, signals coded according to a second coding algorithm, and side information, where the signals coded according to the second coding algorithm have coded residual spectral values, where the residual spectral values are generated from evaluated spectral values by prediction over the frequency, where prediction coefficients of the prediction are present in the side information, comprising: a first decoder for decoding the coded signals which have been coded according to the first coding algorithm to obtain coded/decoded second time signals by means of the first coding algorithm; a second decoder for decoding the coded residual spectral values by means of the second coding algorithm to obtain the residual spectral values; a transformer for transforming the coded/decoded second time signals into the frequency domain to obtain the second spectral values; a predictor for performing a prediction with the second spectral values using the prediction coefficients which are present in the side information to obtain second residual spectral values; a device for inversely evaluating the evaluated residual spectral values and the second residual spectral values to obtain the residual spectral values; an inverse predictor for performing an inverse prediction with the residual spectral values using the prediction coefficients which are stored in the side information to obtain first spectral values; and an inverse transformer for transforming the firs t spectral values back into the time domain to obtain first time signals.
The present invention is based on the finding that the determination of the TNS filter coefficients or prediction coefficients must be performed on the basis of spectral values which are not affected by the coder of the first stage. A scalable audio coder should, of course, also be a flexible coder which, as coder of the first stage, can utilize one of the variants cited in the introduction to the description. According to the present invention the determination of the TNS prediction coefficients is performed on the basis of spectral values which are a direct representation of the audio signal at the input of the coder. By employing a filter bank or an MDCT a spectral representation of the audio coder input signal can be created. However, it is now no longer possible to perform the determination of the TNS filter coefficients at the same place in the coder as the actual filtering by the TNS coding filter. The determination of the TNS filter coefficients must therefore take place separately from the actual TNS coding filtering.
According to a first aspect of the present invention the determination of the TNS filter coefficients is performed directly behind the filter bank which transforms the original audio input signal into the frequency domain. Thus signals of the same type, namely signals which have not been TNS processed, are present in front of the summer or the switching module. According to the first aspect of the present invention the TNS filtering with the already determined TNS coefficients takes place behind the switching module and in front of the quantizer/coder, which might operate according to the psychoacoustic model. As will be apparent later, this implementation of the TNS technique in the scalable audio coder involves a modification of the decoder, however.
According to a second aspect of the present invention this decoding is no longer necessary, however. Here TNS prediction coefficients are again determined at the same place as for the first aspect. In contrast to the first aspect of the present invention, the two relevant spectral signals, i.e. the spectral signal with the coding error of the first stage and the spectral signal which is an essentially undistorted representation of the audio input signal, are processed by the TNS coding filter, which operates with the previously determined TNS coefficients, in front of the summing element. It is important to note that the TNS filtering of the spectral signal which bears the coding error of the coder of the first stage works without redetermination of the TNS coefficients simply using the TNS coefficients derived from the error-free audio signal. According to the second aspect of the present invention two signals of the same type, here TNS-processed signals, are again present at the input of the summer or the switching module.
Generally speaking, the first and second aspects of the present invention differ in that in one instance signals which are not TNS-processed are present in front of the summer whereas in another instance TNS-processed signals are subjected to differencing or are fed into the switching module.
The cited conditions are taken into account in the decoders according to the present invention. In the case of a decoder which decodes a signal coded according to the first aspect of the present invention, the TNS decoding, i.e. the use of the TNS decoding filter employing the TNS coefficients determined when coding, which appear again as side information in the bit stream, takes place in front of an inverse switching module which is analogous to the switching module. As for the coder, the inverse switching module is thus supplied with signals which have not been TNS processed in the case of the decoder as well.
In the case of a decoder which decodes coded signals according to the second aspect of the present invention, on the other hand, the inverse switching module is fed with TNS-processed signals. To this end the decoded signal of the coder of the first stage must be converted into the frequency domain and filtered by means of a TNS coding filter which uses the TNS filter coefficients determined in the coder. Only then are signals of the same kind, namely TNS-processed signals, compared in the inverse switching module or the adder arranged in front of it, as was the case in principle for the coder according to the second aspect of the present invention. The output signals of the inverse switching module are finally fed into a TNS decoding filter, the output signals of which are then processed by an inverse filter bank so as to reproduce the original audio signal apart from the coding errors of the whole arrangement. As has already been mentioned, the coder or decoder according to the second aspect of the present invention is preferred among the embodiments according to the present invention since no substantial modifications are necessary in the decoder as the TNS decoding filter or the inverse TNS filter is arranged in front of the inverse filter bank, which corresponds to the arrangement in FIG. 10A.

Method and a device for coding audio signals and a method and a device for decoding a bit stream

Fraunhofer-Gesellschaft zur Forderung der angewandten Forschung e.V.


A position detecting device which can increase accuracy in detecting the pole position of a motor used to perform quick acceleration and deceleration over the range from a zero speed to a high rotation speed, and a synchronous motor driving device using the position detecting device. A position detector detects basic-wave component signals in sensor signals and executes position calculation. A correcting unit calculates signal information representing at least one of a gain, an offset and a phase by a phase detector from the basic-wave component signals detected by an error calculator, and makes correction based on the calculated signal information such that a position detection error is zero.

1. Field of the Invention
The present invention relates to a position detecting device for detecting a motor pole position and a synchronous motor driving device using the position detecting device. More particularly, the present invention relates to a position detecting device and a synchronous motor driving device using the same, which are suitably used to perform acceleration and deceleration over the range from a zero speed (stopped state) to a high rotation speed.
2. Description of the Related Art
As one of devices for driving a synchronous motor (hereinafter referred to simply as a "motor"), there is known a type of driving the motor by using a position detecting device which detects a rotor pole position. Here, the term "position detecting device" means a device, such as a resolver or an absolute rotary encoder, which can not only detect a motor rotational angle during rotation, but also determine an absolute position in the stopped state. In order to achieve driving of the motor with high efficiency, however, it is essential to detect the phase of a counter electromotive voltage (i.e., the pole position of a motor rotor) by the position detecting device with high accuracy. Several techniques adapted for that purpose are also known.
For example, Patent Document 1 (Japanese Patent No. 3315847) discloses a technique of employing a detection unit for executing position calculation using a linear zone of a sensor signal, and correcting sensitivity and an offset based on maximum and minimum values of the sensor signal.
Also, Patent Document 2 (Japanese Patent No. 3397013) discloses a technique of employing a deviation detecting unit which detects a voltage and a current supplied to an armature coil to estimate a rotor pole position, and which detects a position deviation from an error between the estimated rotor pole position and a detected rotor rotational position, and correcting the detected position deviation.

Position detecting device and synchronous motor driving device using the same


A thermally conductive polymeric drinking surface for a beverage container is provided. The thermally conductive polymeric drinking surface may be an insert for a bottle or other covering configured to be formed around the mouth of a drinking container, such as a cup. The high thermal conductivity of the drinking surface contributes to the transfer of the temperature of the contents of the container to the mouth or lips of the consumer by reducing the time and energy consumption of the chilling processes being applied via the beverage or an external cooling mechanism.

Plastic bottles, glass bottles, aluminum cans and cups made from various materials ranging from paper to plastic to metal, are commonly used as beverage containers. These containers come in a variety of shapes, sizes and configurations. For cold beverages, one advantage of metal based containers, such as aluminum cans, is that the aluminum surface of the can provides the drinker with a cool drinking surface that provides the drinker's lips or mouth with the cold feeling or sensation of a cold beverage contained therein. What is therefore desired is an improved drinking surface for non-metallic containers that provides a cold drinking sensation similar to that of an aluminum can. It is also desired to provide a container having a drinking surface that has a temperature similar to that of the beverage inside the container to provide the consumer with a cool refreshing drinking sensation when the drinking surface comes into contact with the consumer's lips or mouth, thereby enhancing the overall beverage drinking experience of the consumer.

Beverage container drinking surface enhancement


Above a processing tray, a standby tray for making sheets of paper P stand by is installed. The sheets of paper P on the standby tray are dropped, fed, and moved to the processing tray, thus the conveying path from the standby tray to the processing tray is shortened. The standby tray is given a tilt angle of θ1 and the consistency is improved by the own weight of the sheets. A first and a second paper ejection tray are given a tilt angle of θ2 and the consistency is improved by the own weight of the sheets or a sheet bundle.

1. Field of the Invention
The present invention relates to a sheet post-process apparatus for post-processing sheets of paper ejected from an image forming apparatus such as a copier, a printer, or a composite device.
2. Description of the Related Art
In recent years, in an image forming apparatus, to post-process sheets of paper after image forming such as sorting or stapling sheets of paper, a sheet post-process apparatus may be installed in the neighborhood of the paper ejection unit of the image forming apparatus body. To make the succeeding sheets ejected from the image forming apparatus body stand by like this, conventionally, in Japanese Patent Publication 6-99070, an apparatus installing a shifting path halfway the path toward the stapler is disclosed. Or, in Japanese Patent Application 2004-142868, an apparatus having an installed primary loading section above a paper ejection tray to match sheets of paper at the time of paper ejection after the sheets are processed.
However, as in the conventional apparatus, when a shifting path or a buffer path is installed halfway the path toward the stapler, the length of the conveying path from the paper ejection unit of the image forming apparatus body to the stapler becomes longer, thus a problem arises that miniaturization of the apparatus is disturbed.
Therefore, a sheet post-process apparatus capable of shortening the distance from the paper ejection unit of the image forming apparatus to the processing mechanism for performing the post process and realizing miniaturization is desired.

Angled standby tray for post-process device