Embodiments disclosed herein relate to methods and compositions for species-specific detection of bacterial and fungal nucleic acids and nucleic acids of antibiotic resistance genes.

Classical Methods for the Identification and Susceptibility Testing of Bacteria
Bacteria are classically identified by their ability to utilize different substrates as a source of carbon and nitrogen through the use of biochemical tests such as the API20E™ system (bioMérieux). For susceptibility testing, clinical microbiology laboratories use methods including disk diffusion, agar dilution and broth microdilution. Although identifications based on biochemical tests and antibacterial susceptibility tests are cost-effective, at least two days are required to obtain preliminary results due to the necessity of two successive overnight incubations to identify the bacteria from clinical specimens as well as to determine their susceptibility to antimicrobial agents. There are some commercially available automated systems (i.e. the MicroScan system from Dade Diagnostics Corp. and the Vitek system from bioMérieux) which use sophisticated and expensive apparatus for faster microbial identification and susceptibility testing (Stager and Davis, 1992, Clin. Microbiol. Rev. 5:302-327). These systems require shorter incubation periods, thereby allowing most bacterial identifications and susceptibility testing to be performed in less than 6 hours. Nevertheless, these faster systems always require the primary isolation of the bacteria as a pure culture, a process which takes at least 18 hours for a pure culture or 2 days for a mixed culture. The fastest identification system, the autoSCAN-Walk-Away™ system (Dade Diagnostics Corp.) identifies both gram-negative and gram-positive bacterial species from standardized inoculum in as little as 2 hours and gives susceptibility patterns to most antibiotics in 5.5 hours. However, this system has a particularly high percentage (i.e. 3.3 to 40.5%) of non-conclusive identifications with bacterial species other than Enterobacteriaceae (Croizé J., 1995, Lett. Infectiol. 10:109-113; York et al., 1992, J. Clin. Microbiol. 30:2903-2910). For Enterobacteriaceae, the percentage of non-conclusive identifications was 2.7 to 11.4%.
A wide variety of bacteria and fungi are routinely isolated and identified from clinical specimens in microbiology laboratories. Tables 1 and 2 give the incidence for the most commonly isolated bacterial and fungal pathogens from various types of clinical specimens. These pathogens are the most frequently associated with nosocomial and community-acquired human infections and are therefore considered the most clinically important.
Clinical Specimens Tested in Clinical Microbiology Laboratories
Most clinical specimens received in clinical microbiology laboratories are urine and blood samples. At the microbiology laboratory of the Centre Hospitalier de l'Université Laval (CHUL), urine and blood account for approximately 55% and 30% of the specimens received, respectively (Table 3). The remaining 15% of clinical specimens comprise various biological fluids including sputum, pus, cerebrospinal fluid, synovial fluid, and others (Table 3). Infections of the urinary tract, the respiratory tract and the bloodstream are usually of bacterial etiology and require antimicrobial therapy. In fact, all clinical samples received in the clinical microbiology laboratory are tested routinely for the identification of bacteria and susceptibility testing.
Conventional Pathogen Identification from Clinical Specimens
Urine Specimens
The search for pathogens in urine specimens is so preponderant in the routine microbiology laboratory that a myriad of tests have been developed. However, the gold standard remains the classical semi-quantitative plate culture method in which 1 μL of urine is streaked on plates and incubated for 18-24 hours. Colonies are then counted to determine the total number of colony forming units (CFU) per liter of urine. A bacterial urinary tract infection (UTI) is normally associated with a bacterial count of 1 CFU/L or more in urine. However, infections with less than 107 CFU/L in urine are possible, particularly in patients with a high incidence of diseases or those catheterized (Stark and Maki, 1984, N. Engl. J. Med. 311:560-564). Importantly, approximately 80% of urine specimens tested in clinical microbiology laboratories are considered negative (i.e. bacterial count of less than 107 CFU/L; Table 3). Urine specimens found positive by culture are further characterized using standard biochemical tests to identify the bacterial pathogen and are also tested for susceptibility to antibiotics. The biochemical and susceptibility testing normally require 18-24 hours of incubation.
Accurate and rapid urine screening methods for bacterial pathogens would allow a faster identification of negative specimens and a more efficient treatment and care management of patients. Several rapid identification methods (Uriscreen™, UTIscreen™, Flash Track™ DNA probes and others) have been compared to slower standard biochemical methods, which are based on culture of the bacterial pathogens. Although much faster, these rapid tests showed low sensitivities and poor specificities as well as a high number of false negative and false positive results (Koening et al., 1992, J. Clin. Microbiol. 30:342-345; Pezzlo et al., 1992, J. Clin. Microbiol. 30:640-684).
Blood Specimens
The blood specimens received in the microbiology laboratory are always submitted for culture. Blood culture systems may be manual, semi-automated or completely automated. The BACTEC system (from Becton Dickinson) and the BacTAlert system (from Organon Teknika Corporation) are the two most widely used automated blood culture systems. These systems incubate blood culture bottles under optimal conditions for bacterial growth. Bacterial growth is monitored continuously to detect early positives by using highly sensitive bacterial growth detectors. Once growth is detected, a Gram stain is performed directly from the blood culture and then used to inoculate nutrient agar plates. Subsequently, bacterial identification and susceptibility testing are carried out from isolated bacterial colonies with automated systems as described previously. The bottles are normally reported as negative if no growth is detected after an incubation of 6 to 7 days. Normally, the vast majority of blood cultures are reported negative. For example, the percentage of negative blood cultures at the microbiology laboratory of the CHUL for the period February 1994-January 1995 was 93.1% (Table 3).
Other Clinical Samples
Upon receipt by the clinical microbiology laboratory, all body fluids other than blood and urine that are from normally sterile sites (i.e. cerebrospinal, synovial, pleural, pericardial and others) are processed for direct microscopic examination and subsequent culture. Again, most clinical samples are negative for culture (Table 3).
Regarding clinical specimens which are not from sterile sites such as sputum or stool specimens, the laboratory diagnosis by culture is more problematic because of the contamination by the normal flora. The bacterial pathogens potentially associated with the infection are purified from the contaminants and then identified as described previously. Of course, the universal detection of bacteria would not be useful for the diagnosis of bacterial infections at these non sterile sites. On the other hand, DNA-based assays for species or genus detection and identification as well as for the detection of antibiotic resistance genes from these specimens would be very useful and would offer several advantages over classical identification and susceptibility testing methods.
DNA-Based Assays with any Clinical Specimens
There is an obvious need for rapid and accurate diagnostic tests for bacterial detection and identification directly from clinical specimens. DNA-based technologies are rapid and accurate and offer a great potential to improve the diagnosis of infectious diseases (Persing et al., 1993, Diagnostic Molecular Microbiology: Principles and Applications, American Society for Microbiology, Washington, D.C.). The DNA probes and amplification primers which are objects of the present invention are applicable for bacterial or fungal detection and identification directly from any clinical specimens such as blood cultures, blood, urine, sputum, cerebrospinal fluid, pus and other type of specimens (Table 3). The DNA-based tests proposed in this invention are superior in terms of both rapidity and accuracy to standard biochemical methods currently used for routine diagnosis from any clinical specimens in microbiology laboratories. Since these tests are performed in around only one hour, they provide the clinicians with new diagnostic tools which should contribute to increase the efficiency of therapies with antimicrobial agents. Clinical specimens from organisms other than humans (e.g. other primates, birds, plants, mammals, farm animals, livestock and others) may also be tested with these assays.
A High Percentage of Culture Negative Specimens
Among all the clinical specimens received for routine diagnosis, approximately 80% of urine specimens and even more (around 95%) for other types of clinical specimens are negative for the presence of bacterial pathogens (Table 3). It would also be desirable, in addition to identify bacteria at the species or genus level in a given specimen, to screen out the high proportion of negative clinical specimens with a test detecting the presence of any bacterium (i.e. universal bacterial detection). Such a screening test may be based on the DNA amplification by PCR of a highly conserved genetic target found in all bacteria. Specimens negative for bacteria would not be amplified by this assay. On the other hand, those that are positive for bacteria would give a positive amplification signal with this assay.
Towards the Development of Rapid DNA-Based Diagnostic Tests
A rapid diagnostic test should have a significant impact on the management of infections. DNA probe and DNA amplification technologies offer several advantages over conventional methods for the identification of pathogens and antibiotic resistance genes from clinical samples (Persing et al., 1993, Diagnostic Molecular Microbiology: Principles and Applications, American Society for Microbiology, Washington, D.C.; Ehrlich and Greenberg, 1994, PCR-based Diagnostics in Infectious Disease, Blackwell Scientific Publications, Boston, Mass.). There is no need for culture of the bacterial pathogens, hence the organisms can be detected directly from clinical samples, thereby reducing the time associated with the isolation and identification of pathogens. Furthermore, DNA-based assays are more accurate for bacterial identification than currently used phenotypic identification systems which are based on biochemical tests. Commercially available DNA-based technologies are currently used in clinical microbiology laboratories, mainly for the detection and identification of fastidious bacterial pathogens such as Mycobacterium tuberculosis, Chlamydia trachomatis, Neisseria gonorrhoeae as well as for the detection of a variety of viruses (Podzorski and Persing, Molecular detection and identification of microorganisms, In: P. Murray et al., 1995, Manual of Clinical Microbiology, ASM press, Washington D.C.). There are also other commercially available DNA-based assays which are used for culture confirmation assays.
Others have developed DNA-based tests for the detection and identification of bacterial pathogens which are objects of the present invention: Staphylococcus spp. (U.S. Pat. No. 5,437,978), Neisseria spp. (U.S. Pat. No. 5,162,199 and European patent application serial No. EP 0 337 896 131) and Listeria monocytogenes (U.S. Pat. Nos. 5,389,513 and 5,089,386). However, the diagnostic tests described in these patents are based either on rRNA genes or on genetic targets different from those described in the present invention.
Although there are diagnostic kits or methods already used in clinical microbiology laboratories, there is still a need for an advantageous alternative to the conventional culture identification methods in order to improve the accuracy and the speed of the diagnosis of commonly encountered bacterial infections. Besides being much faster, DNA-based diagnostic tests are more accurate than standard biochemical tests presently used for diagnosis because the bacterial genotype (e.g. DNA level) is more stable than the bacterial phenotype (e.g. metabolic level).
Knowledge of the genomic sequences of bacterial and fungal species continuously increases as testified by the number of sequences available from databases. From the sequences readily available from databases, there is no indication therefrom as to their potential for diagnostic purposes. For determining good candidates for diagnostic purposes, one could select sequences for DNA-based assays for (i) the species-specific detection and identification of commonly encountered bacterial or fungal pathogens, (ii) the genus-specific detection and identification of commonly encountered bacterial or fungal pathogens, (iii) the universal detection of bacterial or fungal pathogens and/or (iv) the specific detection and identification of antibiotic resistance genes. All of the above types of DNA-based assays may be performed directly from any type of clinical specimens or from a microbial culture.
In our co-pending U.S. Ser. No. 08/526,840 and PCT (PCT/CA/95/00528) patent applications, we described DNA sequences suitable for (i) the species-specific detection and identification of 12 clinically important bacterial pathogens, (ii) the universal detection of bacteria, and (iii) the detection of 17 antibiotic resistance genes. This co-pending application described proprietary DNA sequences and DNA sequences selected from databases (in both cases, fragments of at least 100 base pairs), as well as oligonucleotide probes and amplification primers derived from these sequences. All the nucleic acid sequences described in this patent application enter the composition of diagnostic kits and methods capable of a) detecting the presence of bacteria, b) detecting specifically the presence of 12 bacterial species and 17 antibiotic resistance genes. However, these methods and kits need to be improved, since the ideal kit and method should be capable of diagnosing close to 100% of microbial pathogens and antibiotic resistance genes. For example, infections caused by Enterococcus faecium have become a clinical problem because of its resistance to many antibiotics. Both the detection of these bacteria and the evaluation of their resistance profiles are desirable. Besides that, novel DNA sequences (probes and primers) capable of recognizing the same and other microbial pathogens or the same and additional antibiotic resistance genes are also desirable to aim at detecting more target genes and complement our earlier patent application.

Species-specific, genus-specific and universal DNA probes and amplification primers to rapidly detect and identify common bacterial and fungal pathogens and associated antibiotic resistance genes from clinical specimens for diagnosis in microbiology laboratories


This arm support structure is equipped with: a bracket on which base end section of an arm of a suspension device is positioned between a pair of facing walls which face one another at a distance in vehicle chassis front-rear direction; axial support members which axially support the base end section on the bracket, by being inserted into long holes formed to pass through facing walls and through-holes formed to pass through the base end section; eccentric plate members which are formed in a circular shape and capable of integrally rotating with axial support members, and have an engaging hole capable of engaging an axial support member formed to pass therethrough at a location offset in radial direction from the circular center thereof; and a plurality of contact parts which contact the circumferential edge of the eccentric plate members, and rotatably support the eccentric plate members.

Conventionally, a double wishbone type suspension device that supports wheels by an upper arm and a lower arm has been known as an example of an independent suspension type of wheels. In general, an upper arm of this kind of suspension device is formed into a substantial V-shape or a substantial A-shape having a pair of front and rear base end portions. An upper arm is swingably attached to a side member by inserting a cam bolt into long holes of front and rear brackets fixed to the side member or the like and through-holes of base end portions, and fastening a nut to the cam bolt (see, for example, Patent Literature 1).


A multi-phase switching power conversion circuit has at least three phases and includes a plurality of switching circuits, a plurality of transformers, a plurality of output rectifier circuits, a resonant network and a control circuit. The resonant network includes a plurality of symmetrical terminals and a plurality of phase branches, which are connected in a multi-phase symmetrical relationship. Each of the symmetrical terminals is connected to the output side of respective switching circuits. The phase branches are connected to a resonant common terminal such that the phase branches are in a star connection. The resonant common terminal is different from the positive terminal and the first reference terminal of the input voltage source. The control circuit is connected to an output terminal of the multi-phase switching power conversion circuit and a plurality of the control terminals of the plurality of switching circuits. The switching circuits are conducted or shut off according to the output voltage under control of the control circuit, so that the electrical energy of the input voltage source is selectively transmitted to the resonant network.

As the power density of the power supply is increased, the demand on the operating efficiency is gradually increased. That is, the operating frequency of the power conversion circuit of the power supply needs to be enhanced. Take a power conversion circuit operated in a pulse width modulation (PWM) mode for example. Generally, as the switching frequency is increased, the size of the power supply is reduced but the switching loss is increased.
On the other hand, the resonant power conversion circuit may have higher conduction losses of switch when compared with the power conversion circuit in the PWM mode. With increasing development of the switch elements, the conduction resistances of the switch elements are decreased and thus the conduction loss is limited. Furthermore, since the power conversion circuit in the resonant mode could provide zero voltage switching or zero current switching, the switching loss is reduced and the operating efficiency at the high frequency is enhanced. As the switching frequency is increased, the size of the power conversion circuit is reduced. As a consequence, the applications of the resonant power conversion circuit are expanded. The resonant mode power conversion circuit is an important solution to achieve high frequency, high power density and high efficiency.
FIG. 1 is a schematic circuit diagram illustrating a conventional resonant DC-to-DC power conversion circuit. As shown in FIG. 1, the DC-to-DC power conversion circuit comprises a switching circuit A1, a resonant network A2, a transformer Tr and a filtering and rectifying output circuit A3. The switching circuit A1 comprises multiple switch elements for converting an input voltage source Vin into a high-frequency pulse voltage. The high-frequency pulse voltage is applied on the resonant network A2 which comprises a resonant inductor Lr and a resonant capacitor Cr. As such, the AC voltage across the primary winding of the transformer Tr transfer the electrical energy to the filtering and rectifying output circuit A3 through the secondary winding of the transformer Tr, thereby generating an output voltage Vo. The filtering and rectifying output circuit A3 comprises an output capacitor Co, a rectifying switch element Sw (e.g. a MOSFET or diode) and optionally an output filtering inductor Lo. In addition, the magnetizing inductance and the leakage inductance of transformer could be considered as portions of the resonant network A2. Take a LLC circuit for example. If the switching frequency is lower than the resonant frequency of the LLC resonant network, the magnetizing inductor of the transformer operates in resonant mode. In other words, the resonant network includes the magnetizing inductance of the transformer.
FIG. 2 is a schematic circuit diagram illustrating a conventional single-phase half-bridge LLC resonant power conversion circuit. In the power conversion circuit of FIG. 2, the switch elements at the primary side (e.g. S1 and S2) are turned on in a zero voltage switching (ZVS) manner and turned off in a resonant manner. In a case that the switching frequency is lower than the resonant frequency fr of the LLC resonant network and higher than fm, the diodes D1 and D2 at the secondary side will be turned off in a zero current switching manner, wherein
            f      r        =          1              2        ·        π        ·                                            L              r                        ·                          C              r                                            ,                      f      m        =          1              2        ·        π        ·                                            (                                                L                  r                                +                                  L                  m                                            )                        ·                          C              r                                            ,and Lm is the magnetizing inductance of the transformer Tr. The magnetizing inductance could also be obtained by connecting an external inductor with the primary winding of the transformer Tr in parallel. Since the power conversion circuit of FIG. 2 could be soft switched, the switching loss is very low. This circuit has been widely used in LCD-TV, notebook computer, telecom device or server because of its simple configuration.
Although the single-phase half-bridge LLC resonant power conversion circuit has some benefits, there are still some drawbacks. For example, according to the increasing of the power level, especially the increasing of output current, the ripple current of output filter increases a lot and thus the ripple of the output voltage Vo is increased. For reducing the output voltage ripple, the capacitance of the output capacitor Co needs to be increased. Alternatively, a complicated two-stage filter circuit at the output side is another solution for the same purpose. The means of reducing the ripple according to the prior art, however, increases the number and volume of the components and increases the overall cost of the power conversion circuit.
As the output current Io is increased, the ripple contained in the input current Iin is increased. For reducing the ripple contained in the output current and the ripple contained in the input current, a two-phase half-bridge resonant DC-to-DC power conversion circuit has been disclosed in for example EP1331723A2. In the DC-to-DC power conversion circuit of the European patent EP1331723A2, the control signals for controlling the switch elements at the primary winding side have 90-degree phase shift. In addition, the switch elements have the same switching frequency.
As the demand on the power is increased, the two-phase resonant DC-to-DC power conversion circuit is insufficient to reduce the ripple contained in the output current and the ripple contained in the output voltage. Recently, a three-phase resonant DC-to-DC power conversion circuit was disclosed for increasing the efficacy of reducing the ripples contained in the input and output currents.
FIG. 3 is a schematic circuit diagram illustrating a conventional three-phase half-bridge LLC resonant power conversion circuit. The input sides of the first phase circuit P1, the second phase circuit P2 and the third phase circuit P3 are connected in parallel. The output sides of the first phase circuit P1, the second phase circuit P2 and the third phase circuit P3 are connected in parallel. Except the connection between the input sides and the connection between the output sides, there's no additional connection between the first phase circuit P1, the second phase circuit P2 and the third phase circuit P3. In addition, the first control signals S1a, S2a, S3a and the second control signal S1b, S2b, S3b of the first phase circuit P1, the second phase circuit P2 and the third phase circuit P3 are respectively complementary to each other. The phase shift between any two adjacent first control signal S1a, S2a and S3a is 120 degrees. The phase shift between any two adjacent second control signal S1b, S2b and S3b is 120 degrees.
In a case that the power conversion circuit is applied to a high-power electronic device, the parameters of corresponding components need to be same in order to obtain the same magnitude of output currents in all of the three phase circuits. That is, the capacitance values of the resonant capacitors Cr1, Cr2 and Cr3 are same, the inductance values of the resonant inductors Lr1, Lr2 and Lr3 are same, and the magnetizing inductance values of the Lm1, Lm2 and Lm3 are same.
For mass-producing the components of the power conversion circuit, the components have respective tolerances. For example, the tolerance between the nominal inductance value and the real inductance value of an inductor is usually ranged from −15% to +15%. In addition, the tolerance between the nominal capacitance value and the real capacitance value of a capacitor is ranged from −20% to +20%. The increase of the precise will increase the fabricating cost. Due to the tolerances of the component parameters (e.g. inductance values, capacitance values or the like), the resonant frequencies of 3 phase circuits are different. If the tolerances of the component parameters are too large, the variations of the resonant properties of the phase circuits are increased.
FIG. 4 is a schematic timing waveform diagram illustrating related current signals processed in the three-phase power conversion circuit as shown in FIG. 3. Due to the tolerances of the component parameters (e.g. inductance values, capacitance values or the like), the peak values of the first phase current i1, the second phase current i2 and the third phase current i3 are distinctness. As such, the different phase current brings different current at the primary side of the transformer and different current at the secondary side of the transformer. Under this circumstance, the power loss of the power conversion circuit is increased and the operating efficiency thereof is reduced, even a component fail of the circuit happens.
For solving the above drawbacks, a power conversion circuit was disclosed is for example Japanese patent No. JP200178449, which was filed by Sanken on Mar. 23, 2001. FIG. 5 is a schematic circuit diagram illustrating a power conversion circuit disclosed in Japanese patent No. JP200178449. As shown in FIG. 5, a first coupling inductor L12, a second coupling inductor L22 and a third coupling inductor L32 are respectively connected to the resonant networks of the first phase circuit, the second phase circuit and the third phase circuit in series. The first coupling inductor L12, the second coupling inductor L22 and the third coupling inductor L32 are coupled with each other. By means of the first coupling inductor L12, the second coupling inductor L22 and the third coupling inductor L32, the current-sharing efficacy of the power conversion circuit is enhanced. Since each phase circuit needs an addition component, the operating efficiency of the power conversion circuit is reduced, and the overall volume of the electronic device having the power conversion circuit is increased.
Therefore, there is a need of providing an improved multi-phase switching power conversion circuit so as to obviate the drawbacks encountered from the prior art.

Multi-phase switching power conversion circuit


A radio communication apparatus at a transmission side that includes a plurality of transmission antennas and performs a communication using at least one carrier. The radio communication apparatus includes a channel dividing unit configured to divide a transmission signal into a plurality of channels based on channel structure information returned from another radio communication apparatus at a reception side and indicating a structure of a space-division-multiplexing-channel and a space-time-coding channel between transmission and reception antennas, and a space-time-coding unit configured to realize transmission diversity by performing a space-time-coding processing for each of the plurality of channels.

In conventional communication apparatuses, when wide band signals are transmitted and received in a mobile environment, frequency selective fading must be overcome. A multicarrier, in particular, orthogonal frequency division multiplexing (OFDM) is employed in various types of radio systems as a countermeasure technique for the frequency selective fading. On the other hand, attention is focused on multiple-input-multiple-output (MIMO) systems, which simultaneously transfer two or more signals by using multiple antennas, in order to further increase transmit capacity. MIMO systems are broadly classified into methods according to SDM and methods according to transmit diversity. A transmit diversity technique called space time coding (STC) belongs to the latter method.
An example of the SDM method is explained briefly (see "Proposal for SDM-COFDM for Wide Band Mobile Communication Realizing 100 Mbit/s by MIMO Channels". Technical Report RCS2001-135 of The Institute of Electronics, Information and Communication Engineers). In a communication apparatus at a transmitter side, error correction coding is carried out individually on data of two channels that are transmitted simultaneously, and thereafter, a predetermined modulation processing is carried out on the respective data after coding, and the results thereof are placed at corresponding subcarriers. The signals on the respective subcarriers are then individually converted into time domain (OFDM signals) by inverse Fast Fourier Transform (IFFT) processing, guard intervals are added, the signals are upconverted to a high frequency band, and thereafter, are transmitted by corresponding transmit antennas.
In the communication apparatus at the receiver side, first, high frequency signals received at different reception antennas are individually converted into baseband signals. At this time, since multiple signals (the two channels) are mixed in each baseband signal, these signals must be separated. Each baseband signal is then converted into a frequency domain signal by Fast Fourier Transform (FFT) processing. Namely, the baseband signals become signals of subcarrier units (subcarrier signals). Since signals of plural channels are multiplexed, these subcarrier signals are extracted as reception signals of respective channels by weight control. "Proposal for SDM-COFDM for Wide Band Mobile Communication Realizing 100 Mbit/s by MIMO Channels", Technical Report RCS2001-135 of The Institute of Electronics, Information and Communication Engineers, Zero-Forcing, which completely suppresses non-desired channels, is used in computing the weight. The reception signals, which are separated into channel units, are respectively subjected to metric computation by demodulation processing, and to error correction processing, and thereafter, are output as final reception signals of respective channels.
In this way, in a conventional communication apparatus that employs the SDM method, the number of transmission symbols per unit time can be increased by simultaneously transmitting different signal series by using plural channels. Namely, high-speed communication in a good channel condition can be realized.
On the other hand, in a communication apparatus that employs the STC method, generally, inverse matrix computation is not needed in channel separation at the reception side, and therefore, reception processing can be realized by a small amount of computation. Further, an apparatus configuration at the reception side can be realized by one antenna, and excellent communication quality can be ensured even in a low S/N environment. Theoretical signal processing of the STC method is disclosed in detail in S. M. Alamouti. "A Simple Transmit Diversity Technique for Wireless communications", IEEE J. Selected Areas in Communications, vol. 16, pp. 1451-1458, Oct. 1998, and V. Tarokh, H. Jafarkhani, A. R. Calderbank, "Space-time Block Coding for Wireless Communications: Performance Results", IEEE Journal On Selected Areas in Communications, Vol. 17, pp. 451-460, No. 3, March 1999.
However, in the conventional communication apparatus that employs the SDM method, since inverse matrix computation is needed for channel separation, the amount of computation increases. Further, for example, when an inverse matrix does not exist in the inverse matrix computation (or the matrix Eq. is near 0), a sudden deterioration in the signal-to-noise ratio (S/N) arises. Reception antennas in a number that is greater than or equal to the number of simultaneous transmission channels are needed.
In the conventional communication apparatus that employs the STC method, since the same signal is transmitted for plural times, it is difficult to increase the number of transmission symbols.
The communication apparatus that employs the STC method and the communication apparatus that employs the SDM method have mutually opposing features as described above. In other words, they have inherent problems. Therefore, there are more to be improved in structuring optimal MIMO channels.
The present invention has been achieved in order to solve the above problems, and it is an object of the present invention to provide a radio communication apparatus that can structure optimal MIMO channels by realizing the features of both methods and by realizing even higher speed.

Radio communication apparatus, transmitter apparatus, receiver apparatus and radio communication system


‘After Shock’ is a distinctive variety of Agonis flexuosa which is characterized by the combination of strong basal branching, short stem internodes, deep red foliage, and both shorter and narrower leaves. As a shrub form of Agonis flexuosa, ‘After Shock’ has a short, upright to slightly weeping growth habit with an overall plant height of approximately 8 ft (unpruned), a width of approximately 5 ft (unpruned).

The present invention relates to a new and distinct perennial variety of Agonis flexuosa, which has been given the variety denomination of 'After Shock'. Its market class is that of an ornamental shrub or small tree. 'After Shock' is intended for use in landscaping as a small specimen tree or hedge shrub in mild Mediterranean climates.
An application for plant breeders' rights for 'After Shock' has been lodged with the Australian Plant Breeders Rights Office and was accepted on Mar. 14, 2012 (under Application No. 2010/319).
Parentage: The Agonis flexuosa variety 'After Shock' was selected as a whole plant mutation from Agonis flexuosa 'Jervis Bay After Dark' (unpatented, Australian Plant Breeders Rights application no. 1997/225) in 2007 at a commercial plant tissue culture laboratory and propagation facility in Tumbi Umbi in the state of New South Wales, Australia, among vegetative propagules of 'Jervis Bay After Dark'. The instant plant was identified as possessing the same dark red stem and foliage color as the parent but further exhibited several improved characteristics such as a more compact and dense growth habit due to strong basal branching habit, shortened stem internode length, and smaller leaves. The instant plant was isolated and given the name 'After Shock'.
Asexual Reproduction: The new variety 'After Shock' was first asexually propagated by vegetative cuttings in 2007 and has been asexually propagated since that time by division and micropropagation in order to advance nursery and landscape trials for distinctness, uniformity and stability of the distinct characteristics. Said characteristics of the inventive 'After Shock' variety have proven to be stable through 10 successive generations produced by both cuttings as well as micropropagation.


A heat exchanger assembly that in a preferred embodiment comprises: an inlet duct lower wall interfacing with a bypass duct; an outlet duct lower wall interfacing with a bypass duct; a heat exchanger coupled between the inlet duct lower wall and the outlet duct lower wall wherein the heat exchanger is at a compound angle with respect to an inlet duct air flow direction; and a fairing coupled to the top of the heat exchanger wherein the fairing forms the inlet duct upper wall and the outlet duct upper wall.

The requirements posed by aircraft engines are changing over time, and engines have evolved dramatically in the last fifty years. As may be seen in FIG. 1, present day engines like the Rolls-Royce Trent 1000 engine 10 dwarf the size of an older fuselage like that of the Concorde 11 and its Rolls-Royce/Snecma Olympus 593 engines 12.
Traditionally, nacelles housed a multitude of components including the accessory gearbox, air-oil heat exchangers and the Full Authority Digital Engine Control (FADEC). As engine fan diameters increase, the size of the nacelle would theoretically need to increase as well. However, the drag generated by the larger nacelle eventually becomes too large. Accordingly, thinner and thinner nacelle designs have become commonplace a.k.a. slim-line nacelles. Larger engines and fans and thinner nacelles reduce the volume left to house the components traditionally housed within the nacelle. As an alternative, these components have been housed within the core zone. As the core zone already houses ducting, pipework, bleed assemblies and other components, relocated hardware previously housed within the nacelle can prove to be a challenge due to envelope constraints.
The increase in fan diameter creates changes to other assembly level requirements including a requirement to reduce the fan speed relative to the turbine speed. A reduction of the fan rotational speed with respect to the turbine rotational speed may be accomplished with an additional gearbox. Currently, heat load from the accessory gearbox, bearings and generators is typically used to pre-heat the fuel with the excess heat being fed into the bypass duct air flow, or into air flow external to the nacelle. It is estimated that the additional gearbox to reduce the fan speed will grossly increase the heat load introduced into the oil. Because the current designs already produce more heat than can be absorbed by the fuel during preheating, the additional heat load from the extra gear box must be dissipated into the bypass duct air flow.
As engine manufacturers strive towards more fuel-efficient architectures, assemblies which are usually driven by compressor discharge pressure, such as Environmental Control Systems (ECS), are being powered by electric assemblies. These assemblies put extra demand on the electrical generators; again, this additional energy results in extra heat load being dissipated into the oil.
As may be appreciated, as engine sizes increase, heat loads are increasing while the volumes available for heat exchanger assemblies for cooling are decreasing. Accordingly, the ability to better manage the heat loads within the engine becomes of paramount importance. To this end, the Applicant describes herein a completely new design for a ducted heat exchanger system. The new design is built on top of some core principals, which allow it to be more compact and more efficient than previous designs.
Applicant is no stranger to the design of heat exchangers for aerospace applications. Amongst the Applicant's portfolio of heat exchangers are numerous designs of air-oil heat exchangers which are in use on commercial and military engine platforms. In two of Applicant's prior designs, the heat exchangers reside within the secondary air flow stream (bypass duct) and are mounted on the fan case. These assemblies are described as fully protruding, that is, all the major components are mounted within the bypass duct.
FIG. 2 illustrates a three-dimensional cross-section of one of Applicant's prior heat exchanger 14 designs. The heat exchanger design depicted in FIG. 2, does not feature any fairings or aerodynamic aids. Although fulfilling the customer requirements, this design not only greatly affects the flow within the bypass duct 13 in terms of total pressure loss (ΔP/P), but also affects the thrust coefficient of the cold nozzle flow. Accordingly, the design shown in FIG. 2 negatively impacts both thrust and fuel consumption.
FIG. 3 illustrates an isometric three-dimensional view of one of Applicant's prior heat exchanger designs that includes shrouds. This design is similar to the design shown in FIG. 2 but utilizes a small shroud on the inlet 15 and outlet 15 of the heat exchanger 16. Whilst the addition of the shrouds improves the overall aerodynamic performance, the overall pressure loss is still high due to the relatively large blockage plus the high approach velocity at the heat exchange matrix.
FIG. 4 illustrates a diagram of a third previous design by Applicant, which takes a different approach to mounting the heat exchanger. As may be seen in FIG. 4, an inlet duct 24 takes secondary air into an Applicant produced heat exchanger 20 mounted on the perimeter of the engine core casing 22 and then returns the heated air back into the bypass duct via a return duct 26. The diagram of FIG. 4 highlights the relative size of the assembly to the engine core casing 22. As may be appreciated, the assembly of FIG. 4 takes up a lot of additional space outside the engine core casing 22.
In the design shown in FIG. 4, the inlet duct features a diffuser which slows the air speed before the air is ingested by the heat exchanger 20. The diffuser has a negative impact on heat exchange performance but a positive impact on pressure loss across the heat exchanger matrix. Because pressure loss is a function of velocity squared, the overall impact of the diffuser is positive. Using a diffuser in this manner is a compromise because it is long and offers many packaging constraints and the resultant diffuser design may stall under certain flow conditions effectively constricting the performance of the assembly.
As may be appreciated, it would be desirable to have a new heat exchanger design that has a reduced footprint in the overall volume of the assembly while having increased performance/efficiency to deal with the increasing heat load demands of modern engines.

High efficiency ducted heat exchanger systems


A mobile station device that receives downlink control information which is used to selectively provide downlink scheduling or a random access order, on a physical downlink control channel from a base station device. The mobile station device also transmits a random access preamble using a random access channel to the base station device based on receiving the downlink control information which provides a random access order, where the downlink control information provides a downlink resource allocation in a case that the downlink control information is used to provide the downlink scheduling and where a preset value is set for a field of the downlink resource allocation in a case that the downlink control information is used to provide the random access order.

W-CDMA has been standardized by 3GPP (3rd Generation Partnership Project), and services thereof have been sequentially provided. HSDPA (High Speed Downlink Packet Access) which is a faster communication scheme has been also standardized and services thereof are about to be provided.
Evolved Universal Terrestrial Radio Access (hereinafter, "EUTRA") has been under consideration by 3GPP. OFDM (Orthogonal Frequency Division Multiplexing) has been proposed for EUTRA downlink by 3GPP. DFT (Discrete Fourier Transform)-spread OFDM, which is a single-carrier communication scheme, has been proposed for an EUTRA uplink.
FIG. 15 illustrates an EUTRA uplink-and-downlink channel structure.
An EUTRA downlink includes a DPiCH (Downlink Pilot Channel), a DSCH (Downlink Synchronization Channel), a Downlink Common Control Channel, a PDCCH (Physical Downlink Control Channel) (L1/L2 (Layer 1/Layer 2) Control Channel), and a DL-SCH (Downlink-Shared Channel).
An EUTRA uplink includes a UPiCH (Uplink Pilot Channel), a RACH (Random Access Channel), a UL-SCH (Uplink-Shared Channel), and a PUCCH (Physical Uplink Control Channel) (see Non-Patent Document 1).
FIG. 16 illustrates an example of RACHs and UL-SCHs being allocated to radio resources. In FIG. 16, horizontal and vertical axes denote time and frequency, respectively. FIG. 16 shows a structure of one radio frame which is divided into multiple radio resources. In this case, each radio resource has a region defined by 1.25 MHz in the frequency direction and 1 ms in the time direction. RACHs and UL-SCHs explained in FIG. 15 are allocated to these regions as illustrated. Thus, the minimum unit of a RACH has 1.25 MHz. In FIG. 16, UPiCHs are shatteringly allocated in an UL-SCH region by symbol or subcarrier. Since multiple channels are prepared for RACHs in EUTRA, multiple random accesses are available at the same time. Synchronization between a mobile station device and a base station device is the primary purpose of the use of RACHs. It has also been considered that a few bits of data for requesting a scheduling of a radio resource assignment are transmitted over a RACH to reduce a connection time (see Non-Patent Document 2).
Only a preamble is transmitted over RACH for synchronization. The preamble includes a signature which is a signal pattern indicative of information. From among tens of signatures preliminarily prepared, some signatures are selected to configure a few bits of data. Currently, 6 bits of data are transmitted by signatures in EUTRA. 64 (i.e., 2 to the 6th power) signatures are prepared for 6 bits of data.
A random ID is assigned to 5 bits of 6 bits of signatures. Any of information items concerning a random access reason, a downlink path loss/CQI (Channel Quality Indicator), or the like, is assigned to the remaining 1 bit (see Non-Patent Document 3).
FIG. 17 is a sequence chart illustrating uplink synchronization using RACH. First, a mobile station device selects a signature based on a random ID, a random access reason, a downlink path loss/CQI, or the like, and transmits a preamble including the signature over the RACH (message Ma1). Upon receiving the preamble from the mobile station device, a base station device compares the preamble with a signal pattern preliminarily stored as a preamble to calculate a synchronization timing shift. Then, the base station device performs a scheduling for transmitting an L2/L3 (Layer 2/Layer 3) message, and allocates a C-RNTI (Cell Radio Network Temporary Identifier) to a mobile station device determined to require the C-RNTI based on the random access reason. Then, the base station device transmits a preamble response including synchronization timing shift information, scheduling information, the C-RNTI, and the random ID (message Ma2). The mobile station device extracts the preamble response including the random ID which is transmitted from the base station device, and transmits an L2/L3 message using the scheduled radio resources (message Ma3). Upon receiving the L2/L3 message, the base station device transmits, to the mobile station device, a contention resolution for determining whether or not a collision is occurring between mobile station devices (message Ma4) (see Non-Patent Document 3).
If multiple mobile station devices select the same signature and RACH for random accesses, the random accesses of the mobile station devices collide with one another. A sequence when a collision of random accesses occurs is explained with reference to FIG. 17. If multiple mobile station devices select the same signature and transmit preambles using the same radio resource block having the same time and frequency (i.e., the same RACH), the messages Ma1 collide. If the base station device cannot detect the message Ma1 due to the collision, the base station device cannot transmit a preamble response (message Ma2). Since the mobile station device cannot receive a preamble response (message Ma2) from the base station device, the mobile station device selects a signature and a RACH again after a given time interval, and then performs a random access. On the other hand, if the base station device can detect a preamble (Ma1) in spite of the collision, the base station device calculates a scheduling for an L2/L3 message and a synchronization timing shift, and then transmits a preamble response (message Ma2) to the mobile station devices. However, all the mobile station devices receive the preamble message, and then transmits an L2/L3 message (message Ma3) using the scheduled resource. Consequently, messages Ma3 from the mobile station devices collide. Since the base station device cannot receive the L2/L3 message due to the collision, the base station device cannot transmit a response. Since none of the mobile station devices receives a response to the L2/L3 message, each of the mobile station devices selects a signature again and then performs a random access.
When uplink synchronization between the mobile station device and the base station device is lost (for example, when data has not been received or transmitted for a long period, and the mobile station device is, for a long period, in a DRX (Discontinuous Reception) state for monitoring a downlink resource assignment signal), and when the base station device resumes a downlink data transmission, the mobile station device cannot transmit an ACK/NACK (Acknowledgement/Negative Acknowledgement) which is a reception response for an HARQ (Hybrid Automatic Repeat Request). This is because the uplink synchronization is lost, and therefore a transmission of the ACK/NACK for the HARQ causes an interference with another mobile station device. For this reason, uplink synchronization has to be established using a random access upon a downlink data transmission resuming. However, there is concern that it takes a long time for the downlink data transmission to be resumed if a collision occurs upon the random access. To prevent this, a proposition has been made in which a collision of random accesses upon a downlink data transmission resuming is prevented by, for example, using a signature dedicated to a downlink data transmission resuming.
FIG. 18 is a sequence chart illustrating a method of preventing a collision of random accesses when a downlink data transmission is resumed.
When the base station device decides to resume a downlink data transmission to a mobile station device with which uplink synchronization is lost, the base station device transmits an uplink synchronization request (message Mb1). This uplink synchronization request is transmitted using an L1/L2 (Layer 1/Layer 2) control channel. The uplink synchronization request includes the signature ID number of a random access to be performed by the mobile station device. This is called a dedicated signature. The mobile station device performs a random access (i.e., transmits a preamble) using the dedicated signature included in the received uplink synchronization request (message Mb2). Upon receiving the preamble including the dedicated signature, the base station device detects a synchronization timing shift based on the preamble. Then, the base station device transmits, as a preamble response, a TA (Timing Advance) command indicative of the synchronization timing shift (message Mb3). After the base station transmits the TA command, the base station device transmits an L1/L2 control channel including a downlink resource assignment (message Mb4), and then resumes a downlink data transmission (message Mb5) (see Non-Patent Document 4).    Non-Patent Document 1: R1-050850 "Physical Channel and Multiplexing in Evolved UTRA Uplink", 3GPP TSG RAN WG1 Meeting #42 London, UK, Aug. 29-Sep. 2, 2005    Non-Patent Document 2: 3GPP TR (Technical Report) 25.814, V7.0.0 (2006-06), Physical layer aspects for evolved Universal Terrestrial Radio Access (UTRA)    Non-Patent Document 3: 3GPP TS (Technical Specification) 36.300, V0.90 (2007-03), Evolved Universal Terrestrial Radio Access (E-UTRA) and evolved Universal Terrestrial Radio Access Network (E-UTRAN), Overall description Stage 2    Non-Patent Document 4: R2-062165 "UL Synchronization", 3GPP TSG RAN WG2 Meeting #54 Tallinn, 28 Aug. 1-Sep. 2006

Base station device, mobile station device, program, uplink synchronization requesting method, and synchronization-shift measurement signal transmitting method


Process for desulfurizing combustion gas wherein the gas is fractionated into at least two flows which pass through distinct passageways (4 and 5) and receive particles of a solid desulfurizing agent coming from distinct injectors (7 and 8) . The two particle-charged flows meet (9) , are separated again (10 and 11) and meet again thereafter (12). The process can be repeated several times. The final mixture is carried off and fractionated into desulfurized gas and recovered solid particles.

This invention relates to a process for desulfurizing combustion gas.
The strict regulations limiting in many countries and many regions the emissions of sulfur oxides through the gaseous effluents of heat generators forbid the use of fuels with a high sulfur content which, besides, may show definite economical advantages: it is the case of some coals having certain similarities to lignites, and of petroleum residues stemming from refining processes.
French patent 2,609,150 describes a process for reducing the emissions of sulfur oxides and particularly the ones of sulfur dioxide coming from a heat generator accepting fuels with a high sulfur content. According to this process, a powdered desulfurizing agent is injected into the combustion gases and the resulting mixture is circulated in a desulfurization zone, then the at least partially desulfurated gases are partly separated from the powdered desulfurizing agent which can be recycled or discharged.
A similar process is described in the French patent application EN. 88/12,363. In this patent application, the mixing of the desulfurizing agent particles with the combustion gases is carried out in a turbulence zone located upstream from the desulfurization zone.
With this type of desulfurization, the rate of use of calcic or magnesian absorbents such as calcium and magnesium oxides and carbonates remains limited. The reason for this phenomenon has not been totally cleared up, but it seems that a skin which partly prevents the reagents from getting into the grains, forms at the surface of these grains.
A process allowing to remedy the drawbacks mentioned above and particularly allowing to obtain high desulfurization rates and to increase the rate of use of the absorbent, at a lower cost, has been discovered.


A tactile sense presentation device including: a tactile-sense presentation unit having a first working electrode group of a plurality of first working electrodes arranged along a first direction; and a control unit operable to apply a first tactile sense signal having a first waveform to each of the first working electrodes. The control unit applies the first tactile sense signal, in which a phase difference corresponding to a distance between adjacent first working electrodes is applied to the first waveform, to each of the plurality of first working electrodes so as to cause the first waveform to travel along the first direction at a predetermined speed.

Recent years have seen active studies on devices for presenting a tactile sense to a user.
Patent Literature 1 discloses a device that stimulates subcutaneous Pacinian corpuscles of a user by means of capacitive coupling formed between the device and the user, thereby presenting a tactile sense to the user. The device in Patent Literature 1 includes at least one conductive electrode provided with an insulator. The conductive electrode is driven with an electrical input containing a component having a frequency in a range from 10 Hz to 1000 Hz. In accordance with an electrical input, a capacitive coupling is formed between the conductive electrode and the user's skin, which are in proximity to each other with an insulator therebetween. The user perceives the capacitive coupling formed in accordance with the electrical input as vibrations tactually.

Tactile sense presentation device and tactile sense presentation method

PANASONIC INTELLECTUAL PROPERTY MANAGEMENT CO., LTD.


Cellular communication systems supporting high utilization geographic regions having extensive cell overlap segments that collectively contain a substantial portion of the mobile units. A system and method for channel assignments incorporating selection from alternative transceivers defining overlapping cells is provided with load balancing to reduce call blocking. The system incorporates selective multiple handoffs responsive to channel assignment requests both to extend load balancing and also to substantially avoid call cutoff when active mobile units cross cell boundaries into possibly saturated cells.

1. Field of the Invention
This invention relates to cellular communication systems, such as cellular telephone or personal communication services (PCS), and more particularly relates (a) to a multilayer cellular design in which multiple cellular arrangements (each with an assigned group of frequencies) provides a substantial degree of coverage overlap, and (b) to a method for allocating and transferring calls among the cellular arrangements.
2. Discussion of the Background
Conventional cellular systems have a hierarchical system design. A mobile switching office is attached by voice and data links to a number of base stations, each of which is connected to an antenna with a set of frequencies, each of which can connect to a number of mobile units (HHTs) via a radio channel in its predetermined portion of region coverage. An HHT can be a hand-held telephone or other mobile unit communicating voice or data over an assigned frequency channel to a selected base station. Throughout this specification, when voice communication is discussed, the communication channel created and the communication links could be purely data, voice or hybrid voice and data communication.
The mobile switching office and base stations have the computing power to process communicating an HHT's requests for service and to determine which frequency channel assignment will be initially allocated for communicating with the HHT, as well as any hand-off reassignment of channel and antenna necessitated by the HHT moving beyond the cell of the currently assigned antenna.
A common approach to cellular design is illustrated in FIG. 1 and includes a hexagonal lattice of cells with a single antenna covering each cell. The actual portion of the region covered by an antenna may be slightly larger than the hexagonal cell, as shown by the circular region of radius R in FIG. 1. The overlap of cells at the cell boundaries identifies the cell segments in which conventional systems may hand-off the channel assignment and antenna for an HHT moving across a call boundary. However, this cellular overlap covers only a small portion of the geographical area of a cell.
When an HHT with an assigned channel moves to a new cell where the antenna covering the cell has an available channel, the hand-off changing the antenna and frequency for both transmit and receive is transparent to the user. If the antenna in the new cell has no available channel, the call in progress is cut off, this being an unfortunate problem with current cellular systems.
The frequencies used for channel assignments are limited. In a cellular systems the frequency set allocated to a given cell may be reused at some specified distance such as the distance D shown in FIG. 1. This distance must be large enough so as to not create co-channel interference with HHTs using the same channel in different cells. The distance D in FIG. 1 allows the bold-faced seven-cell cluster to be repeated to cover an arbitrarily large geographical region with all frequencies reused repetitively at the same distance D.
The literature teaches various systems (see, for example, FIG. 4 of "The Cellular Concept," V. M. MacDonald, Bell Systems Technical Journal, Vol. 58, No. 1, Pages 15-41, January, 1979, incorporated by reference herein) of patterns of clusters of distinct frequency sets which may be reused at a certain safe distance from each other.
In cellular systems with such frequency reuse allowing coverage of arbitrarily large regions, there is still a problem in that the number of telephone calls that may be active in a given cell at any moment are limited by the number of frequencies allocated to that cell. Some digital systems have improved the total number of calls possible in each cell by multiplexing calls and employing more complex HHTs. Still, the number of active calls in any given cell is limited. When this number is reached by active calls in a cell and not near the boundary where they could be handed off, any new HHT in that cell requesting service will be blocked. Note that blocking can occur even though neighboring cells have available channels.
One solution to the problem of excessive call blocking and call cut-offs is to reduce the cell size, providing a multitude of low power microcells which increases the total available channels over a geographical region by increasing frequency reusage. However, the power of a microcell cannot be reduced too low or the reliability of communication will suffer. Moreover, decreases in transceiver power cause the background noise to signal strength ratio to grow requiring greater HHT complexity to reject the noise levels incurred. Furthermore, as the call size decreases, moving HHTs will require more hand-offs, increasing system overhead and the chance that moving active HHTS will be cut off. This increases the risk of the entire system or portions thereof going into a "thrashing" situation. During a "thrashing" situation, cut-offs of existing calls become a real risk. The cut-off of an active telephone call is considered more disruptive than the unavailability of a channel to a new request for service. Thus, the mobility of HHTs over the region and the level of background noise serve to yield a practical limit to the minimum cell size that may be provided over a region. When the cell size is as small as practical the inability to operate using a majority of channel capacity without noticeable call blockage is a problem with current systems.
Another solution is to have different sets of frequencies occur with different reuse distances, yielding layers of various size cells, where the smaller size cells possessing increased frequency reuse may serve only a non-contiguous portion of the region supplementing the contiguous cell region of another layer. Such multiple reuse patterns add complexity to the system with the smaller cell portion still susceptible to more background noise and greater need for hand-offs. Furthermore, the number of assignable channels in multiple frequency reuse distance systems may vary so as to provide considerably less capacity in portions of each original cell causing surges in those areas to be more disruptive.
Another problem with current systems is that the boundary area between cells is a portion of the region where relatively small movement of an HHT can necessitate a hand-off, and oscillatory movement of an HHT across a boundary or circular motion around the intersection point where three adjacent hexagons meet can greatly increase the occurrence of hand-off overheads while at best preserving a low grade of signal strength to an HHT at such a local in the region. There is a need in cellular systems to avoid the disruptive behavior of service to HHTs happening at the cell boundaries.
Another problem that exists in conventional systems is that a failure, a repair, or the line of a given antenna, which takes the cell off the air, will result in a dead area of coverage in which no available service can occur in that cell for some period of time, and if an HHT moves into that cell while communication is in progress, the communication will be cut off.
The current cellular system has two service providers, and the direction of PCS service, particularly in metropolitan areas, is to have two or more providers offer competing cellular service over the same broad region. The partition of available channels to a multitude of providers, each operating independently and each subject to the degradations in service previously mentioned occurring at more exaggerated levels, compared to the channels available in each system results in poorer overall service. It is a problem to promote competition in cellular PCS service without degrading the level of service that could be provided by the total channels available.

Channel assignment selection reducing call blocking and call cutoff in a cellular communication system


A reactive energy fluid filled cushioning and stabilizing apparatus in a shoe is comprised of one or more anatomically shaped fluid filled bladders that are positioned in the forefoot, arch and/or heel areas of the shoe. The bladders assume a complementary custom fitting configuration to the contours of a foot inserted in the shoe. In variant embodiments of the apparatus, the fluid filled bladders are positioned separate from each other in the forefoot, arch and heel areas of the shoe and are interconnected in fluid communication by fluid conducting channels, or the bladders are positioned independent of each other in the forefoot, arch and heel areas of the shoe.

Field of the Invention
The present invention relates to a fluid-filled apparatus which reacts with the stimulus of an outside force, hereinafter referred to as a reactive energy apparatus, incorporated into a shoe sole and shoe upper that provides a custom fit to the shoe wearer's foot in addition to providing cushioning, support and stability to the shoe wearer's foot. In particular, the present invention pertains to an apparatus comprised of one more anatomically-shaped fluid-filled bladders that are positioned in the forefoot, arch and/or heel areas of a shoe and assume a complementary custom fitting configuration to the contours of a foot inserted in the shoe. In variant embodiments of the apparatus, the fluid-filled bladders are positioned separate from each other in the forefoot, arch and heel areas of the shoe and are interconnected in fluid communication by fluid-conducting channels, or the bladders are positioned independent of each other in the forefoot, arch and heel areas of the shoe.

Reactive energy fluid filled apparatus providing cushioning, support, stability and a custom fit in a shoe


A display device capable of switching over between vertical and horizontal display on a screen without using a frame memory is provided. A display device has a source signal line driver circuit (102), a first gate signal line driver circuit (103), and a second gate signal line driver circuit (104). The scanning direction of the first gate signal line driver circuit (103) is perpendicular to the scanning direction of the source signal line driver circuit (102), and the scanning direction of the second gate signal line driver circuit (104) is perpendicular to the scanning direction of the first gate signal line driver circuit (103). During normal display, vertical scanning of the screen is performed by the first gate signal line driver circuit (103). Images are displayed in a direction that is in accordance with the scanning direction of the first gate signal line driver circuit (103). On the other hand, vertical scanning of the screen is performed by the second gate signal line driver circuit (104) when switching over between vertical and horizontal display. Images are displayed in a direction that is in accordance with the scanning direction of the second gate signal line driver circuit (104).

1. Field of the Invention
The present invention relates to a display device structured by a pixel portion that uses light emitting elements such as liquid crystal elements or electroluminescence elements (EL elements), and to electronic equipment using the display device in a display portion. In particular, the present invention relates to a display device having a pixel portion and a driver circuit for driving the pixel portion formed on the same insulating surface, and to electronic equipment using the display device in a display portion.
2. Description of the Related Art
In recent years, display devices in which semiconductor thin films are formed on an insulating surface such as a glass substrate, in particular electronic circuits using thin film transistors (hereinafter referred to as TFTs), are in use in all fields. In particular, their use in display devices is common, and active matrix display devices such as LCDs (liquid crystal displays) are utilized in many products and widely adopted. Active matrix display devices using TFTs have several hundred thousand to several million pixels arranged in a matrix shape, and image display is performed by controlling the electric charge of each pixel by using TFTs disposed in each pixel.
Recently, techniques relating to polysilicon TFTs have progressed, which are used for simultaneously forming driver circuits on the same substrate by using TFTs in regions that are in the periphery of a pixel portion in addition to pixel TFTs that structure pixels. These techniques contribute greatly to making devices smaller and reducing their electric power consumption. Display devices have thus become indispensable devices used in display portions and the like on mobile information terminals, whose expanded fields of application have become remarkable in recent years.
An example of a general display device is shown in FIG. 2A. FIG. 2A is an example of a liquid crystal display device in which a pixel portion and a driver circuit are integrally formed on an insulating surface. A pixel portion 201 is disposed in a center portion on a substrate 200, and a source signal line driver circuit 202, gate signal line driver circuits 203, and the like are formed in the periphery of the pixel portion 201. Note that although the gate signal line driver circuits 203 are disposed symmetrically on both right and left sides of the pixel portion 201 in FIG. 2A, they may also be placed on only one side. However, it is preferable to arrange the gate signal line driver circuits symmetrically as in FIG. 2A when considering the reliability of circuit operation, efficiency, and the like.
Signals input to the source signal line driver circuit 202 and the gate signal line driver circuits 203 are supplied form the outside through a flexible printed circuit (FPC) 204.
An opposing electrode and the like are formed in an opposing substrate 210, and the opposing substrate 210 and the substrate 200 are bonded through a sealing agent 205 while maintaining a certain gap. A liquid crystal material is then injected into the gap between the substrate 200 and the opposing substrate 210 from an injection port prepared in advance. The injection port is then sealed tightly by using a sealant 206.
m source signal lines and n gate signal lines are disposed orthogonally in the pixel portion 201 as shown in FIG. 2B. There are m source signal lines and n gate signal lines in FIG. 2B. Locations 220 at which the source signal lines and the gate signal lines intersect form pixels as shown in FIG. 2C. The pixel comprises a source signal line 221, a gate signal line 222, a pixel TFT 223, a liquid crystal element 224, a storage capacitor 225, and an opposing electrode 226. The number of pixels here is m×n pixels.
Operation of the display device is explained simply with reference to FIGS. 5A to 5C. In general, screen drawing is performed on the order of 60 times per second so that pixel flicker is not recognizable by human eyes. A period denoted by reference numeral 501, that is, a period necessary to draw the screen one time, is referred to as one frame period here (see FIG. 5A).
Selection of the gate signal lines is performed in sequence from a first row in one frame period. A selection period 504 per one row is denoted as a horizontal period. A period 502 up until selection of the final row (number n row) is complete is denoted as a line scanning period. Similar operations are then performed in the next frame period, sandwiching a vertical return period 503 (see FIG. 5B).
Write-in of the image signal in sequence to the pixels of the selected row is performed in one horizontal period from the source signal lines. This period, a period 505, is denoted as a dot sampling period. A period 507 necessary for writing in an image signal to one pixel is denoted as one dot sampling period. When write-in of the image signal in one row portion of pixels is complete, similar operations are performed in the next horizontal period, sandwiching a horizontal return period 506 (see FIG. 5C).
Specific circuit operation is explained next. FIG. 6A is an example of the structure of the source signal line driver circuit of the display device, and has a shift register 602 that uses a plurality of stages of flip-flops (FFs) 601, a NAND 603, a buffer 604, and a sampling switch 605. Operation is explained with reference to FIG. 6B. The shift register 602 outputs pulses in order from the first stage in accordance with a clock signal (CK), a clock inverted signal (CKb), and a start pulse (SP).
In the case where the pulses output from the shift register 602 overlap in adjacent stages, they are input to the NAND 603 and pulses that do not overlap in adjacent stages are formed. The NAND output then passes through the buffer 604, and becomes sampling pulses.
When the sampling pulses are input to the sampling switch 605, the sampling switch 605 is turned on, and the electric potential of an image signal (Video) charges the source signal line connected to the sampling switch during that period. At the same time, the image signal is written into one pixel connected to the source signal line of the row whose gate signal line is selected. In FIG. 6B, a period denoted by reference numeral 610 is one dot sampling period.
A gate signal line driver circuit shown in FIG. 7A is explained next. The structure from a shift register to a buffer is nearly the same as that of the source signal line driver circuit, and the gate signal line driver circuit has a shift register 702 that comprises a plurality of stages of flip-flops 701, a NAND 703, and a buffer 704.
Operation is explained with reference to FIG. 7B. Similarly to the source signal line driver circuit, the shift register 702 outputs pulses in order from the first stage in accordance with the clock signal (CK), the clock inverted signal (CKb), and the start pulse (SP).
In the case where the pulses output from the shift register 702 overlap in adjacent stages, they are input to the NAND 703 and pulses that do not overlap in adjacent stages are formed. The NAND output then passes through the buffer 704, and becomes gate signal line selection pulses.
As stated above, the image signal written into the source signal line is then written into each of the pixels in the row to which a gate signal line selection pulse is input. In FIG. 7B, a period denoted by reference numeral 710 is one horizontal period, and a period denoted by reference numeral 720 is the one dot sampling period mentioned above.
In the case where the display device has many functions, such as with a personal computer, the display device may be used in a horizontal format for certain applications, and in a vertical format in other applications. For cases such as this, there is a method of displaying in a state in which a display device frame is rotated by 90°, as shown in FIG. 3A.
A pixel portion of an active matrix display device has m×n pixels arranged in a matrix shape as shown in FIG. 2B. Write-in of an image signal is performed in sequence from a pixel at coordinate (1,1), followed by (1,2), (1,3), and (1,4). One horizontal period is complete when (1,m) is reached. This is repeated n times, and write-in of one screen is complete when write-in is performed to the final pixel at coordinate (m,n).
Once again return to FIG. 3A. In the case of horizontal format display (left side) and vertical format display (right side), the pixels at coordinate (1,1) into which write-in is performed first are denoted by reference numerals 301 and 302, respectively. Considering cases of performing display of similar images in horizontal format display and vertical format display, the input order is in sequence from the upper left to the upper right, and on downward to the lower right when the image signal input corresponds to horizontal format display, as shown in FIG. 3A. In the case of performing vertical format display using this image signal, the write-in order to the display device itself does not change, and therefore the sequence for inputting the image signal must be from the upper right to the lower right, and then leftward toward the lower left.
However, it is preferable to be able to perform switch over to horizontal format display flexibly with the display device, and therefore the preparation of image signal having different formats is not efficient. Display is then performed using a frame memory, storing the image signal temporarily in the memory and then reading it out.
A frame memory is for storing the image signal of each pixel in respective memory cells, and it is therefore possible to read out from arbitrary addresses, irrespective of the write-in order. By changing the readout order of the image signal written temporarily into the frame memory, switch over between vertical and horizontal display can be performed.
The frame memory for storing one frame portion of an image signal has memory circuits managed by addresses, as shown in FIG. 3B. When the image signal is input, it is therefore written in sequence into the addresses (1,1), (2,1), . . . , (m,1), (1,2), (2,2), . . . , (m,2), (1,n), (2,n), . . . , and (m,n). For horizontal format display, readout is performed in the same sequence as the write-in.
On the other hand, for vertical format display, readout is performed in sequence from the addresses (m,1), (m,2), . . . , (m,n), (m−1,1), (m−2,2), . . . , (m−1,n), . . . , (1,1), (1,2), . . . , and (1,n), when display as in FIG. 3A is desired.
Further, the frame memory generally has at least two frame portions formed, as shown in FIG. 4A. During a period in which write-in to one of the frame memories is performed, readout from the other frame memory is performed, thus performing display.
The display device can thus perform switch over between vertical and horizontal screens by using normal drive as is. However, the number of pixels in the horizontal and vertical directions of the display device is normally different, and therefore it is necessary to change the format of the image signal at the same time as performing switch over between vertical and horizontal displays.
The image signal is structured by a total of n row portions of image signals: an image signal to be written into the pixels 1 to m of the number 1 row, an image signal to be written into the pixels 1 to m of the number 2 row, . . . , an image signal to be written into the pixels of the number n row, as shown in FIG. 4B(i). In this case, the signal corresponds to m(side)×n(length) pixels. To perform switch over between horizontal and vertical display, it is necessary to convert to a shape corresponding to −n columns of pixels horizontally by m rows of pixels vertically, as shown in FIG. 4B(ii). This operation is referred to as format conversion. Known techniques may be used for the format conversion process itself, and therefore the details are omitted here.
For instance, in case of m(side)×n(length) pixels, an image signal corresponding to one frame period is formed by gathering image signals of n lines, each image signal corresponding to one dot sampling period×m dots. When length and side of the screen change places, it becomes a display region of n(side)×m(length) pixels. The image signal for one frame period need to reconstitute so as to be an image signal corresponding to one frame period that is formed by gathering image signals of m lines, each image signal corresponding to one dot sampling period×n dots.
Here, horizontal element changes from m dots to n dots. When one dot sampling period is the same, the length of one horizontal period becomes n/m. image signal corresponding to one horizontal period is reconstituted by expanding or shortened the image signal corresponding to one horizontal period to suit the n/m length of one horizontal period and sampling of n dots.
On the other hand, it is not possible to correspond by the change in the sampling frequency because the number of lines increase and decrease to the element of the vertical direction. Therefore, it is done by increasing the number of lines a frame by thinning out a part of image signal and decreasing the number of lines a frame, and inputting a part of image signal oppositely two or more lines (For instance, two lines are continued) more than once.
Such a function is offered with IC chip installed beforehand outside, and the converted resolution has been decided to several kinds beforehand but usually;
The method of doing the format conversion high-quality and efficiently is indicated in open 2001-246939 special titles etc., and you may use such a technique.
Various types of software are recently being supplied even to small size portable terminals such as mobile phones, and there is a tendency for one unit to have more and more uses. A vertical and horizontal display switch over technique such as one described above therefore becomes very important.
However, a frame memory is prepared as externally mounted, separate from the display device. That is, the number of component parts increases. In particular, compact portable terminals have grown increasingly smaller recently, and therefore it is difficult to add an additional externally mounted frame memory to present day size terminals. Performing switch over between horizontal and vertical display by a conventional method is therefore not preferable for small size portable terminals.

This patent is subject to a terminal disclaimer.

Semiconductor Energy Laboratory Co., Ltd.


A video converter for converting a coding format of a great deal of video contents has three kinds of conversion modes including a conversion mode for performing conversion in two steps in accordance with the kinds of input pictures so that a converting operation can be completed within a limit time set in advance.

1. Field of the Invention
The present invention relates to a video converter and a method for converting video data encoded by predetermined encoding means into data encoded by another different encoding means. Particularly, it relates to a video conversion system, a conversion method and a computer program for converting MPEG-2 video data into MPEG-4 video data. Incidentally, MPEG-2 and MPEG-4 are international standards for moving picture encoding.
2. Description of the Related Art
Video contents produced and broadcast by a TV broadcast station, a CATV (cable TV) station, a Web contents provider or the like are generally encoded in MPEG (Moving Picture Experts Group) or the like and stored in digital form in various kinds of recording media such as HDD (Hard Disk Drive) and DVD (Digital Versatile Disk) so that the video contents can be used secondarily. On this occasion, particularly MPEG-2 (Moving Picture Experts Group Phase 2) is used frequently because it is known that MPEG-2 can be used in DVD and HDTV (High Definition Television) and is high in picture quality after encoding.
At present, there is a discussion about a video delivery service for delivering video contents so that the video contents stored in the aforementioned manner can be played back by a PDA (Personal Digital Assistant), a PC (Personal Computer) or the like connected to a cellular phone or the Internet.
It is however substantially difficult to deliver the video contents in the form of MPEG-2 directly because the amount of information (from the order of hundreds of kbps to twenty and several Mbps) in MPEG-2 is larger by several times than the transmission capacity (from the order of tens of kbps to ten and several Mbps) in infrastructure of these circuits. Therefore, for example, a method of once converting MPEG-2 into MPEG-4 (Moving Picture Experts Group Phase 4) higher in information compression rate than MPEG-2 and storing the MPEG-4 again (or delivering the MPEG-4 immediately) may be conceived.
Incidentally, the number of video contents handled by a TV broadcast station, a CATV (cable TV) station, a Web contents provider or the like is enormous. When these video contents are to be processed by the video delivery system, "completion of conversion of a great deal of video contents within a limit time set in advance" is requested for the sake of convenience of a broadcast schedule. Particularly it is important to achieve the system support so that a predetermined conversion process can be executed to the last without failure even in the case where a request of conversion with a load too large (to complete conversion within the limit time) is set.
In the related art, there is a system of converting MPEG-2 video contents into MPEG-4 video contents (e.g., see JP-A-2002-10267 and JP-A-2002-44669).
JP-A-2002-10267 describes a method having a procedure for limiting conversion of MPEG-2 I pictures into MPEG-4 I pictures to restrain increase in the quantity of coding generated due to direct conversion of all MPEG-2 I pictures (larger in the quantity of coding than P pictures) into MPEG-4 I pictures. Even for conversion of I pictures, the method converts the I pictures into P pictures (smaller in the quantity of coding than I pictures) on the MPEG-4 conversion side in accordance with the necessity.
JP-A-2002-44669 describes a method making use of similar correlation between motion vector information in pictures before conversion and motion vector information in pictures after conversion, that is, for performing encoding by using MPEG-2 motion vector information for decoding as MPEG-4 motion vector information. In the method, an MPEG-4 motion vector retrieving process can be omitted so that conversion time can be shortened. On the other hand, there is a video delivery system in which video data converted into MPEG-4 is not stored in a recording medium but always directly sent out to a delivery line such as the Internet (e.g., see JP-A-2002-232860). No data are held on the system side after conversion.
The method disclosed in JP-A-2002-10267 is a method in which MPEG-2 I pictures are converted into MPEG-4 P pictures in accordance with the necessity to thereby limit the quantity of coding generated in MPEG-4 as a whole. Generally, because encoding of P pictures is a process of requiring reference to other pictures, it is assumed that the time required for conversion has a tendency to increase. Hence, there is a problem that the method cannot be adapted to a system aiming at completing conversion of a great deal of video contents within a predetermined limit time.
The method disclosed in JP-A-2002-44669 is a method for performing encoding by using MPEG-2 motion vector information for decoding as MPEG-4 motion vector information to thereby shorten the conversion time. In conversion attended with conditions such as image scaling, frame skipping, etc. in the conversion process, there is however possibility that MPEG-4 may be encoded with large error because the motion vector information used is not always equal to true motion vector information. Hence, there is a problem that picture quality of video after conversion in this case deteriorates compared with that of MPEG-4 video encoded on the basis of true motion vector information. The method disclosed in JP-A-2002-232860 has a system configuration in which video contents converted in MPEG-4 are not stored. Because the method is however developed on the assumption that all MPEG-2 input frames are processed, there is no particular consideration with respect to execution of conversion of a predetermined number of video contents within a limit time. Accordingly, there is a problem that the method, like the aforementioned related-art method, cannot be adapted to a system aiming at completing conversion of a great deal of video contents within a predetermined limit time.

Video converter and computer program for executing said video converter


There is provided a stator that can suppress degradation in insulating performance and electrical strength performance, shorten a work time, and improve winding efficiency and heat radiation. The stator 1 includes a yoke part 20 formed to be annular, a plurality of tooth parts 30 protruding from the yoke part 20 in an inward direction perpendicular to an axial direction O of a rotor when the rotor is arranged in the yoke part 20, and protecting members 40 each of which is attached to both end surfaces of the tooth part 30 in the axial direction O to protect a corner of the tooth part 30. The protecting members 40 have a length in a slot 38 between the tooth parts 30 shorter than a half of an entire length of the tooth part 30 in the axial direction.

1. Technical Field
The present invention relates to a stator that is provided in an electric motor, and relates to a protecting member for a stator.
2. Description of Related Art
A stator provided in an electric motor is usually formed by stacking a plurality of steel plates, and includes teeth for winding an electric wire. Between the electric wire and the stator, insulation needs to be maintained. For this purpose, in some cases, an exposing part of the stator is coated with an insulating material, and the electric wire is coated with an insulating material to secure an insulation property. The electric wire is wound around the teeth of the stator coated with the insulating material.
When the electric wire is wound, the electric wire is pressed into edge portions of the teeth of the stator, and this results in degrading insulating performance and reducing electrical strength. In light of it, a thickness of the insulating coat formed on the stator is required to be large. As a result, a slot where the electric wire wound around the stator is accommodated is made small (wire-winding space is made small) to reduce the efficiency of the wound wire.
In view of it, a stator described in Japanese Patent Application Laid-Open Publication No. 2010-279241 is coated with a resinous insulating member at the portion contacting with an electric wire such that the electric wire does not contact directly with the stator.
However, according to Japanese Patent Application Laid-Open Publication No. 2010-279241, the insulating member is molded in conformity with the shapes of the tooth part and the slot, and the resinous member is inserted into the slot of the stator so as to cover the portion where the electric wire is wound. This technique requires a process of inserting the insulating member, and the work takes time. Furthermore, the insulating member protects the stator over the entire length of the slot in the direction of stacking steel plates, so that when the number of the stacking steel plates is changed, the shape of the insulating member also needs to be changed. This increases a cost of the insulating member. In addition, the insulating member is provided in the slot, so that the region where the electric wire is wound is limited to be narrow. As a result, a winding efficiency is reduced. Furthermore, the insulating member is provided between the stator and the electric wire, so that it becomes difficult for the heat to transmit to the stator from the electric wire that generates heat due to an electric current flowing therethrough. As a result, heat radiation performance is degraded.


A side protector for ceiling formwork. The ceiling formwork comprises at least one formwork element having a frame. The side protector has a railing. Furthermore, the side protector has a pivoting frame. The railing is connected to the pivoting frame. The pivoting frame can be connected indirectly or directly to the frame from the bottom side of the ceiling formwork. The pivoting frame can be pivoted from an installation position into an end position about an axis of rotation (A) in such a way that the ceiling formwork is laterally secured by the railing. Furthermore, a ceiling formwork system, having at least one corresponding side protector and ceiling formwork, is specified. In addition, a method for erecting such a side protector is specified.

The use of a ceiling formwork made of formwork elements to build concrete ceilings is known. In particular, frame panel formwork elements are used that at least partially form a mold to be filled with liquid concrete. To produce a concrete ceiling, a ceiling formwork supported by ceiling supports is first constructed. A desired room height can be achieved by the positioning of the ceiling formwork at a specific height. The standard room height in Germany for single-family homes, duplexes and row houses is at least 2.40 meters, wherein commercial properties can have a significantly larger room height. The liquid concrete filling operation on the ceiling formwork is carried out by workers who must issue on the ceiling formwork for this purpose. In order to protect these workers against a life-threatening fall from the ceiling formwork during the concrete filling operation, a side protector is installed on the ceiling formwork. For this purpose, side railing brackets, into which side railings are mounted, are laterally attached to the ceiling formwork. The ceiling formwork is laterally secured via the side railings and a fall of the workers from the ceiling formwork is prevented. The installation of the side protector is generally carried out by workers who must walk on the ceiling formwork to do this.
This is problematic in that there is an increased risk of the workers falling from the ceiling formwork during installation of the side protector.

Side protector, ceiling formwork system having at least one such side protector, and method for erecting such a side protector