The present invention provides a design framework that is used to develop new types of constrained turbo block convolutional (CTBC) codes that have higher performance than was previously attainable. The design framework is applied to design both random and deterministic constrained interleavers. Vectorizable deterministic constrained interleavers are developed and used to design parallel architectures for real time SISO decoding of CTBC codes. A new signal mapping technique called constrained interleaved coded modulation (CICM) is also developed. CICM is then used to develop rate matching, spatial modulation, and MIMO modulation subsystems to be used with CTBC codes and other types of codes. By way of example, embodiments are primarily provided for improved 5G LTE and optical transport network (OTN) communication systems.

Field of the Invention
The present invention relates generally to methods, apparatus and systems for communication encoders, decoders, transmitters, receivers and infrastructure and/or user devices. More particularly, aspects of the invention relate to constrained turbo block convolutional codes, constrained interleaving, and related methods, apparatus, and systems for improved constrained interleaving, encoding, decoding, signal mapping, MIMO applications, spatial modulation, and rate matching. The present invention also relates to efficient parallel ASICs and VLSI architectures and optical integrated circuit architectures to implement these methods, apparatus, and systems.
Description of the Related Art
A large body prior art includes of technical publications, patents, and standards that relate to 4G LTE (fourth generation long term evolution) wireless systems. In particular, the relevant prior art relates to encoding and decoding architectures and algorithms for use with the CTC (convolutional turbo code) specified for use with 4G LTE. Specifically, important prior art relates to algorithms and high performance ASIC architectures for CTC encoding/decoding, deterministic contention-free interleavers such as the QPP (quadratic polynomial permutation) based interleavers, and rate matching/puncturing architectures.
A parallel decoding ASIC for the CTCs used in 4G LTE can be found in C. Studer, C. Benkeser, S. Belfanti, and Q. Huang, "Design and Implementation of a Parallel Turbo-Decoder ASIC for 3GPP-LTE," IEEE J. Solid State Circuits, Vol. 46, No, 1, January 2011 (referred to as the "Studer" reference" herein). A follow on paper explains more improvements and details about efficient parallel decoding of the CTC used in 4G LTE. This second technical publication is: C Roth, S. Belfanti, C. Benkeser, and Q. Huang, "Efficient parallel turbo-decoding for high throughput wireless systems," IEEE Transactions on Circuits and Systems, 2012 (referred to as the "Roth reference" herein).
One of ordinary skill in the art would be familiar with the Studer reference which explains how to design a highly optimized parallel real time ASIC designed to implement the CTC specified for use in 4G LTE. The Roth reference provides further details and optimizations to the same architecture as described in the Studer reference. One of ordinary skill in the art would also be familiar with the following prior art reference as well: A. Nimbalker, Y. Blankenship, B. Classon, T. K Blankenship, "ARP and QPP Interleavers for LTE Turbo Coding," WCNC 2008 proceedings, (referred to as the "Nimbalker reference" herein). The architecture in the Studer reference uses the QPP interleaver as described in the Nimbalker reference. The QPP interleaver is important because it is used in the 4G LTE standard and because it can be described as a "contention free" "vectorizable" and "deterministic" interleaver.
As is well known "contention free"/"vectorizable" means that the permutation function has a particular property that aids in parallel processing implementations. Consider a case where there are N=8 parallel processors. Then, as long as N divides the frame size, K, the contention free interleaver places on a given row in memory all of the N elements to be processed by the N processors in a given clock cycle. The QPP only supports up to N=8 level vectorization.
The Studer reference also points out a very efficient way to compute the QPP address sequence. As per the Nimbalker reference, the QPP interleaved address sequence can be written asπQPP(i)=(f1i+f2i2) mod K   (1)where f1 and f2 are suitably chosen interleaver parameters that depend on the code-block size K. Note that in this notation the sequentially incremented symbol i is used to denote a coded bit position in the transmitted frame, and the permuted version of the indexing sequence, πQPP(i), is used to look up a bit position in the non-permuted sequence of input bits. The Studer reference explains a very efficient way to compute equation (1) is to use the following set of recursions which can be easily implemented in hardware. The recursions below only use additions and modulo operations which can be very efficiently implemented in hardware. Hence at runtime, in hardware, equation (1) is computed asπQPP(i+1)=(πQPP(i)+δ(i)) mod K   (2)andδ(i+1)=(δ(i)+b) mod K   (3)where πQPP(0)=0, δ(0)=f1+f2, and b=2f2.
Another prior art reference that is known to those of skill in the art and that goes into further detail about QPP recursions is: Y. Sun and J. Cavallaro, "Efficient hardware implementation of a highly-parallel 3GPP LTE/LTE-advanced turbo decoder," Integration, the VLSI Journal, No. 44, 2011, pp 305-315, (referred to as the "Sun reference" herein). This reference provides additional recursions that allow QPP addresses to be incremented by an integer, d=Δi, that can be any positive integer. This allows forward and backward sequences of QPP addresses to be generated for forward and backward recursions used in decoding. Also, this allows recursions similar to equations (2)-(3) to increment by more than one element, for example, Δi=K/M, where K is the frame size and M is the number of processors in a system. The Sun reference also explains the prior art knowledge that a set of M different QPP address generators can be run in parallel with relative offsets of one and with Δi=K/M to generate a set of M consecutive QPP addresses in parallel. The Sun reference also provides efficient hardware circuits to implement such an addressing scheme.
Another relevant field of art is called rate matching. Rate matching is also known as "puncturing." The CTC mother code defined in the LTE standards is a rate ⅓ parallel concatenated turbo code. This CTC leads to very complicated rate matching circuits at both the encoder and the decoder, thus increasing over all hardware complexity of the 4G LTE CTC encoding and decoding. A reference that discusses rate matching for LTE turbo codes is C. Ma and P. Lin, "Efficient implementation of rate matching for LTE codes," IEEE ICFCC 2010 international conference proceedings, pp. V1-704-708 (referred to as the "Ma reference" herein). FIG. 1 of the Ma reference shows the basic configuration of 4G LTE rate matching at the transmitter side. The data stream plus two streams of parity bits from the rate ⅓ parallel concatenated CTC pass through three parallel blocks labeled "sub-block interleaver." That is, three interleavers are used, one each to process the total number of bits in a non-punctured frame. Another reference that explains the rate matching used in 4G LTE is L. Yu et al., "An improved rate matching algorithm for 3GPP LTE Turbo code," Conference on Communications and Mobile Computing (CMC), pp. 345-348, April 2011. FIG. 2 of this article and the discussion thereof is very helpful in understanding the 4G LTE rate matching algorithm.
There also exists a vast body of literature related to OTN (optical transport network) applications. OTN applications are demanding because they require very high data rates and powerful codes and the frame size used in coding/decoding is long, (122,368 message bits plus coding overhead bits). OTN systems are either already available or still being researched and developed to support data rates of 100GBPS (usually referred to as 100G), 400GBPS and even up to 1000GBPS (1 Terabit per second, 1 T). These very high speed systems demand very powerful codes to achieve specified high NCGs (net coding gains) at very low BERs (bit error rates) below 10−15. High speed digital hardware that employs extensive parallel processing is needed to decode these powerful codes in real time.
It can be noted that in OTN applications, the codes being used/considered now correspond to LDPC (low density parity check) codes, concatenations of LDPC codes with one or more long block codes, or TPCs (turbo product codes). OTN applications cannot use CTCs like LTE does because the error floors required by OTN applications are far below those afforded by CTCs. Hence it would be desirable to have a much lower complexity parallel coding/decoding technique and parallel architecture than those that are currently proposed for use in or used in the OTN field. It would be desirable if this low complexity coding/decoding technique could meet the stringent NCGs requirements at BERs of 10−15 and outperform all known coding/decoding techniques that are currently proposed for use in or used in the OTN field.
The prior art also includes U.S. Pat. No. 8,537,919 "Encoding and decoding using constrained interleaving," and its continuation-in-part, U.S. Pat. No. 8,532,209, "Methods, apparatus and systems for coding with constrained interleaving, and both of these U.S. Patents are incorporated herein by reference in order to provide the reader with written description level details of known constrained interleaver design techniques, and known encoder/decoder structures that use constrained interleaving. These patents are incorporated by references, but it is to be understood that for claim construction purposes, the instant written description should be used, and not any of the written description in the incorporated-by reference patents. In this patent application, some terms are defined differently than the U.S. patents incorporated by reference herein. Therefore, it is to be understood that the interpretation of terms and phrases used in the claims herein should be taken in the context of the present application and not the references incorporated herein. The prior art also includes J. Fonseka, E. Dowling, S. I. Han and Y. Hu, "Constrained interleaving of serially concatenated codes with inner recursive codes," IEEE Communications Letters, Vol. 17, No. 7, July 2013, referred to herein as "the Fonseka [1] reference." The prior art also includes J. Fonseka, E. Dowling, T. Brown and S. I. Han, "Constrained interleaving of turbo product codes," IEEE Communications Letters, vol. 16, 2012, pp. 1365-1368, September 2012, referred to herein as "the Fonseka [2] reference." The prior art also includes S. I. Han, J. P. Fonseka and E. M. Dowling, "Constrained Turbo Block Convolutional Codes for 100G and Beyond Optical Transmissions," IEEE Photonics Technology Letters, Vol. 26, No. 10, May 2014, referred to herein as "the Fonseka [3] reference." The above-listed patents and technical publications also cite to related articles in the technical literature and to other U.S. patent references, which are also part of the prior art. It can be noted that the above referenced patents and technical papers constitute at least a portion of what would be known to one of skill in the art of CTBC (constrained turbo block convolutional) codes.
Consider FIG. 1, which corresponds to FIG. 4 in U.S. Pat. Nos. 8,537,919 and 8,532,209. FIG. 1 shows an encoder structure that can represent a method and/or an apparatus for encoding in accordance with CTBC code. The CTBC encoder embodiment of FIG. 1 makes use of an outer block code (OBC) encoder 405, that encodes in accordance with a selected OBC. For example the OBC can be a (n,k) block code, B, where n>k and n,k are positive integers. The message bit stream at the input can be considered to be a sequence of k-bit blocks consisting of message bits. Each k-bit message block is first processed by the OBC encoder 405 which, in the exemplary embodiment of FIG. 1, encodes according to an (n,k) outer code with minimum Hamming distance (MHD) given by MHD=d0. In some embodiments the outer code 405 can perform outer encoding in accordance other types of fixed-length codes, such as a finite-length convolutional code or an LDPC code, for example. A characterizing feature of the embodiment of FIG. 1 is that it also makes use of an inner recursive convolutional code (IRCC) encoder 415 that encodes its input bit stream in accordance with an inner recursive convolutional code (the selected IRCC). An appropriate IRCC is chosen to have an MHD given by MHD=di. For example, the IRCC, could be selected to be the rate-1 accumulator given by G(D)=1/(1+D). Another specific example of an IRCC is to use the rate-1 accumulator followed by a (λ,λ−1) SPC encoder (or any other block code), a finite-length (finite impulse response) convolutional code, or any other recursive convolutional code (RCC). The value of λ can be chosen to provide design flexibility to chose the IRCC to fine tune the rate and/or the di value to design a CTBC code to meet a particular set of design specifications. In some embodiments, the CTBC code is designed using the rate-1 accumulator as the IRCC, but this CTBC code is then followed by another block code like the (λ,λ−1) SPC encoder mentioned above.
Another characterizing feature of the CTBC encoder 400 is that it makes use of a constrained interleaver 410. Any specific CTBC code is defined in terms of the specifically selected outer block code B used in the OBC encoder 405, the specifically selected recursive convolutional code (RCC) used in the IRCC encoder 415, and a specifically selected constrained interleaver having a specified size and permutation function used in block 410. The constrained interleaver 410, and various forms of its interleaver constraints are described in the above-cited prior art references. The constrained interleaver 410 can be designed to provide an interleaver gain, Gl, similar to uniform interleaving, but also can be designed to ensure that the net MHD of the entire CTBC code satisfies some target MHD, dt≧d0di. It can be noted that if the constrained interleaver used in the CTBC were to be replaced by a uniform interleaver of the same length, a "Uniform-interleaved Turbo Block Convolutional" (UTBC) code would result, and the MHD of this corresponding UTBC code would typically be close to MHD, dt=di.
Various forms of constrained interleavers are defined in the above-referenced U.S. patents and the three above-cited references related to constrained interleaving. A constrained interleaver type 2, i.e., the "CI-2" is introduced and used in the block 401 of FIG. 1. The above-referenced U.S. patents teach how CI-2 interleaver constraints can be defined to design the constrained interleaver 410 to enforce the property MHD, dt≧d0di. In U.S. Pat. No. 8,532,209, the term and notation "Constrained interleaver type 2" and its abbreviation "CI-2" are introduced. In the Fonseka [2] reference, it is shown that CI-2s can be designed to achieve a specified target MHD that satisfies d0di≦MHD≦d02di. CI-2s use inter-row constraints in order to achieve this. Note that the constrained interleaver block 410 in FIG. 1 is labeled "r×ρn constrained interleaver." This is because, as discussed in the above-referenced U.S. patents, the constrained interleaver's permutation function is designed using a r×ρn row-column matrix structure. That is, the prior art relies upon the CI-2 design matrix, [A]r×ρn and requires certain relations to hold for coded bit positions from different codewords of the OBC that are loaded into [A]r×ρn. In the Fonseka [1] reference, the symbol for the number of rows of the CI-2 design matrix was changed to the symbol, "L," and the CI-2 design matrix is thus written as of [A]L×ρn. In the rest of this patent application, from here forward, the symbol L will be used to refer to the number of rows in the CI-2 design matrix.
An objective of the CI-2 interleaver is to create CTBC codes that simultaneously provide a specified high MHD while achieving as high of an interleaver gain as possible. The high MHD provides a lower error floor and has other desirable effects in various types of channels, and the high interleaver gain ensures a high coding gain for the CTBC code. However, the interleaver gain attainable by the CI-2 is limited to a large extent by the number of rows, L in the CI-2 design matrix. The lower the number L, for a fixed frame size K, the higher the CI-2 interleaver gain. However, when CI-2 interleavers are used, lowering L will eventually limit the achievable MHD.
It would be desirable to have improved constrained interleavers that do not require a CI-2 design matrix, but instead use L=1, and can thus lead to improved CTBC codes that have higher interleaver gains as compared to a CI-2 interleaver of the same length. It would be desirable to further include improved signal mapping methods, apparatus and systems to map a CTBC code onto a target signal constellation in such a way as to provide a constellation mapping gain, similar to the kinds of gains provided by trellis coded modulation (TCM) and bit interleaved coded modulation (BICM). It would also be desirable to have new rate matching algorithms that could efficiently interoperate with these new and improved CTBC codes and signal mapping subsystems. It would also be desirable to have algorithms developed for applications in multiple input multiple output (MIMO) systems and spatial modulation and subsystem for use communications devices that include in multi-antenna subsystem.
Next consider FIG. 2, which corresponds to FIG. 5 in U.S. Pat. Nos. 8,537,919 and 8,532,209. FIG. 2 shows a prior art receiver method and apparatus for a receiver 500 used to receive and decode a signal r(t) which was generated in accordance with FIG. 1 or a version of a serial concatenated code whose inner coded is a block code that is also discussed in U.S. Pat. Nos. 8,537,919 and 8,532,209. It is important to note that when CTBC codes as generated using an IRCC as shown in FIG. 1 herein, block 510 and the connection between block 510 and 525 will be missing. The block 510 is only used to decode coded signals generated by an alternative embodiment shown in FIG. 2 of the above two referenced patents. So herein, block 510 and should be ignored.
Block 1105 processes or otherwise demodulates a received signal r(t) to generate an initial vector rs, which preferably corresponds to a vector of bit metrics. The bit metrics are preferably used in decoding of the component codes using an α-posteriori probability (APP) decoding technique.
The IRCC soft in soft out (SISO) decoder 515 can implement a well known soft decoding algorithm such as the BCJR algorithm, or a soft output Viterbi algorithm (SOVA), the min sum algorithm. Such algorithms are known to generate extrinsic information indicative of the reliability of the soft decoded results. The BCJR algorithm can be embodied using any of the MAP, Log-MAP, or the Max-Log-Map algorithms. For example, if the IRCC SISO decoder 515 involves the BCJR algorithm, then the IRCC SISO decoder 515 will need to compute a sequence of branch transition probabilities, γ's, that each are a function of a respective element of the received signal metrics, rs, and a corresponding respective element of updated or initial extrinsic information, the Le's. The IRCC SISO decoder 515 will use this sequence of branch transition probabilities, γ's, while making one forward recursion pass to update a set of state metrics, α's, and one backward recursion pass algorithm to update a set of state metrics, β's . Such concepts are well known in the art in the context of decoding convolutional turbo codes (CTCs). Using the calculated α's, β's and γ's values, the BCJR decoding of the IRCC decoder calculates the extrinsic information of all its input bits. For example, see P. Robertson, et al., "A comparison of optimal and sub-optimal MAP decoding algorithms operating in the log domain, " IEEE ICC 1995, pp. 1009-1013.
The IRCC SISO decoder 515 couples its extrinsic information output to a constrained deinterleaver 520 which deinterleaves the extrinsic information received from the IRCC SISO decoder 515, for example, in accordance with the inverse CI-2 permutation function. The OBC SISO decoder 525 is coupled to receive the deinterleaved extrinsic information from the constrained deinterleaver 520. The OBC SISO decoder 525 also preferably implements a known soft decoding algorithm such as the well known Chase-Pyndiah algorithm (also referred to as the Pyndiah algorithm), low complexity Chase-Pyndiah algorithm, the OSD algorithm and its low complexity variations, or any similar soft decoding algorithm for decoding of block codes, for example. In general, different well known (or proprietary) soft decoding algorithms can be used in the blocks 515 and 525. All such algorithms are well known to those of skill in the art, for example, see J. Cho and W. Sung, "Reduced complexity Chase-Pyndiah decoding for turbo product codes," pp. 210-215, IEEE workshop on signal processing systems, October, 2011.
It would be desirable to have a decoding architectures that could be used to efficiently decode the new improved CTBC codes. It would be desirable to have additional efficient algorithms and parallel architectures to decode the improved CTBC codes that have undergone additional constrained interleaving based signal mapping and/or rate matching and/or constrained interleaving based spatial modulation.
While the above mentioned prior art relating to constrained interleaving for use with an OBC and an IRCC provide very powerful CTBC codes, the CI-2 is based on the CI-2 design matrix, [A]L×ρn, and the concept of a random interleaver. The construction of the CI-2 requires many randomization operations performed in the CI-2 design matrix and a complicated process of ensuring that randomizations do to not violate any constraints in the CI-2 design matrix. As discussed below, this CI-2 design matrix and design process actually limits BER performance. Also, the CI-2 is not a vectorizable/contention free interleaver. Herein a "random interleaver" is also defined in opposition to a "deterministic interleaver" that uses a mathematical formula to generate the deterministic interleaver permutation. A random interleaver is thus often implemented as a table look up or with a state-machine logic circuit whose sequencing logic does not use a fixed mathematical equation but whose state transition logic needs to be specifically designed for each is frame size.
It would be desirable to have a family of a contention free, vectorizable constrained interleavers, both deterministic and semi-random. It would be desirable to have an SCC that is constructed by coupling the output of the OBC to the IRCC via a contention free, vectorizable and deterministic version of a constrained interleaver. It would further be desirable to be able to design a system that could achieve the memory efficient benefits of the Studer reference, and to also greatly simplify the rate matching requirements of the system. It would be desirable to have a parallel architecture that could meet the encoding and decoding performance requirements of the 4G LTE CTC encoders and decoders, but with simpler computational functional units, less overall computational complexity, and thus lower power consumption. It would be desirable to have a CTBC encoder/decoder architecture that could eliminate the complicated and hardware intensive rate matching and inverse rate matching subsystems required by 4G LTE encoders and decoders. It would also be desirable if the parameters of this same CTBC encoder/decoder architecture could be scaled to higher values of N levels of parallelism and designed to provide the NCGs need at BERs of 10−15 for 400 GHz and beyond OTN applications. It would be desirable to also have a new coded modulation techniques that could be used to map codes onto higher order constellations and to implement advanced functions such as rate matching, spatial modulation, and MIMO systems. It would be desirable if the advanced modulation technique could be used along with optical integrated circuits and similar technology to implement higher capacity optical communication channels, for example 400 GHz and beyond, and 1 Tera Hz and beyond. It would be desirable to have a constrained interleaver design process that did not rely on the CI-2 design matrix and was able to provide higher BER performance for random and deterministic constrained interleavers.

Constrained interleaving for 5G wireless and optical transport networks


A work device, including: a reciprocable movable member; a lengthy member connected to the movable member at its first-end-side portion, so as to extend in a first direction, and is elastically curved toward a second direction different from the first direction so that its second-end-side portion is fixed at a prescribed position, the lengthy member being operable to follow a reciprocating movement of the movable member; and a guide member having a first guide surface extending along the second direction, a second guide surface extending in a third direction intersecting the second direction, and a third guide surface contiguous to the first and second guide surfaces, wherein the third guide surface is configured such that a boundary point in the lengthy member transits on the third guide surface from the first guide surface toward the second guide surface when the lengthy member contacting the first guide surface moves so as to come into contact with the second guide surface following the movable member, the boundary point defining a boundary between a region of the lengthy member that is in contact with the third guide surface and a region of the lengthy member that is not in contact with the third guide surface.

1. Field of the Invention
The present invention relates to a work device including a lengthy member which follows a movable member configured to reciprocate in a certain direction and which is guided by a guide member and also relates to an image recording apparatus equipped with the work device.
2. Discussion of Related Art
There has been conventionally known an image recording apparatus of an ink-jet system configured to record an image on a recording medium by ejecting ink based on input signals. As the ink-jet system, there has been known a system in which the ink is introduced to an actuator such as a piezoelectric element or an electrostrictive element disposed in a recording head and droplets of the ink are ejected from nozzles by deformation or deflection of the actuator in accordance with the input signals. There has been also known, as the ink-jet system, a system in which the ink pressurized utilizing local boiling thereof by a heat-generating element is ejected as the ink droplets.
The ink-jet recording head is mounted on a carriage so as to be moved relative to the recording medium such as a recording sheet, for instance. The carriage is configured to be driven by a drive force transmitted from a drive source such as a motor for thereby reciprocating in a certain direction while being guided by guide shafts, guide rails, or the like. During the reciprocating movement of the carriage, the ink droplets are selectively ejected from the recording head toward the recording medium, so that an image is recorded by the ink droplets attached to the recording medium.
For supplying the ink from an ink cartridge to the recording head, there is employed a method using ink tubes. Each of the ink tubes defines a flow path through which the ink flows to the recording head from the ink cartridge disposed separately from the recording head. The ink tube has flexibility that allows the ink tube to follow the reciprocating movement of the carriage. The ink tube has a length that is determined considering a distance between the carriage and the ink cartridge when the carriage is located at a position that is the most distant from the ink cartridge. When the carriage is located at a position that is the nearest to the ink cartridge, the ink tube is curved into a substantially U-shape, for instance.
Patent Document 1 discloses an arrangement in which inks of four colors accommodated in respective ink cartridges are supplied to a carriage (1) via respective four ink supply tubes (87). Each of the four ink supply tubes is curved into a generally U-shape that has a bent portion (W). In the disclosed arrangement, owing to a guide member (89), a direction of a restoring force of the bent portion (W) of the ink supply tube that acts on the carriage is made equal to a direction substantially perpendicular to a direction of the reciprocating movement of the carriage, namely, made equal to a direction substantially perpendicular to a main scanning direction (X). Accordingly, the restoring force of the bent portion (W) does not act obliquely on the carriage, so that the carriage is prevented from being inclined, avoiding a deterioration in the accuracy with which the liquid or ink is ejected.
Patent Document 1 JP-A-2004-268346

Work device and image recording apparatus equipped with the work device


A Low Dropout Regulator (LDO) with Less Quiescent Current in the Dropout Region is described, including an error amplifier configured to compare a reference voltage to an LDO output voltage across a resistive divider, a current mirror configured to mirror a first output of the error amplifier to a first and second output of the current mirror, and a comparator configured to compare the LDO output voltage to a second output of the error amplifier, which has been compared to the second output of the current mirror, and configured to output a control voltage to the error amplifier, where a low quiescent current is maintained when an LDO input voltage is near or less than the LDO output voltage.

A low dropout regulator (LDO) regulates the output voltage when the supply voltage is very close to the output voltage. The dropout region of the LDO is the operation region in which the input voltage is near or less than the target output voltage. The quiescent current of the LDO is the operation current when absent a load current. In the dropout region, no matter how efficient the LDO is, the LDO cannot output the exact target voltage.
FIG. 1 shows a conventional low dropout regulator. Vref is the reference voltage used to define the output voltage Vout, and 'A' is an error amplifier. This is a kind of OTA (Operational Transconductance Amplifier) but its output current depends negatively on its input voltage. R1 and R2 are resistors to generate a divided voltage from the output voltage, to make Vout equal to the desired target voltage Vref×[(R1+R2)/R1]. Imirror1 is a current mirror circuit designed to supply current to the load, and to keep the output voltage equal to the target voltage.
FIG. 2 shows an example of the current mirror circuit Imirror1, of FIG. 1. Devices P1 and P2 are PMOSFETs, and Vin is the input voltage of the LDO. Note that 'in' is the current input terminal, and 'out' is the current output to supply current to the load. Typically, the output current, source-to-drain current of P2 is more than the input current, source-to-drain current of P1. If the input current is 'm' amperes and the output current is 'n' amperes, the current mirror ratio is n/m.
The conventional LDO of FIG. 1, including FIG. 2, increases the source-to-drain current of P1, and as a result, its operation current increases, if Vin becomes near or less than the target output voltage. This means that the LDO operates in its dropout region. For a conventional current mirror circuit, the input voltage, Vin, must be sufficiently high in order to keep the desired current mirror ratio n/m. For example, in the circuit shown in FIGS. 2, P1 and P2 need to be saturated to get the desired current mirror ratio. Once Vin becomes less than a certain voltage, P2 cannot be saturated, and its source-to-drain current decreases according to the decrease of Vin. The source-to-drain current of P1 does not decrease, and P1 is always saturated, if P1 is a usual enhancement type MOSFET.
If Vin of the conventional LDO decreases, and becomes near or less than Vref×[(R1+R2)/R1], which is the target output voltage, the input current of Imirror1, namely the output current of error amplifier A, increases up to a maximum, because the feedback divided voltage of the output is kept less than Vref. In this case, if the load current is low, the operation current of the LDO can be significantly higher than the load current, which is undesirable.

Low dropout regulator with less quiescent current in dropout region


An apparatus includes a magnetic adjustment lens positioned at the electron beam path between the electron source and sample, the magnetic adjustment lens excited by an electric coil, and a permanent magnet lens positioned below the magnetic adjustment lens to focus the electron beam onto the sample surface, the permanent magnet lens excited by one or more permanent ring magnets enclosed except on a bottom surface by a magnetic field conductor. The magnetic adjustment lens may be excited to eliminate magnetic field leakage of the permanent magnet lens.

Defects in integrated circuit chips may arise during a patterned wafer fabrication process. Defect information may be obtained from wafer inspection tools to improve production yield. Two types of defect detection tools are commonly used: (1) optical inspection tools, which illuminate a wafer using optical light, and (2) scanning electron beam inspection tools, which illuminate a wafer using electrons.
Optical inspection systems provide throughput on the order of a few hours for a 300 mm wafer inspection task, utilizing parallel data acquisition. However, the resolution of optical inspection tools is restricted by the diffraction limit imposed by the (relative to electrons) large wavelength of photons. Currently, optical inspection systems are not able to identify all of the defects at fourteen nanometer and smaller design scales.
The diffraction limit imposed by scanning electron beam inspection tools is significantly lower than that of optical tools because the wavelength of electrons is on the order of angstroms. Scanning electron beam inspection tools provide resolution on the order of a few nanometers, which is enough resolution to detect defects at fourteen nanometer and lower patterned wafer design scales. Unfortunately, the throughput of scanning electron beam inspection tools is low because imaging data is obtained in a serial manner during the scanning process. Depending on the signal-to-noise ratio required for the inspection process, the data acquisition rate of scanning electron beam inspection tools is commonly from 100 mega pixels to about one giga pixel per second. Currently, it takes more than a week to inspect a 300 mm wafer with a ten nanometer pixel size using a conventional scanning electron beam inspection system.
A multiple electron beam inspection system utilizing parallel data acquisition may increase the throughput and shorten the time to inspect a wafer. One parallel approach utilizes a common electric current coil to generate magnetic excitations at an array of holes on a magnetic pole piece, creating an array of magnetic electron focusing lenses. The current coil has only one axis of symmetry, which means that only one lens in the array can be excited symmetrically. The other lenses are excited asymmetrically because the lens optical axis does not overlap with the geometric axis of the excitation coil. This symmetry mismatch causes magnetic field distortion along the optical axis, degrades resolution, and introduces non-uniform performance among different lenses. Another approach creates a small current-driven electron magnetic focusing lens unit, and assembles an array of such lens units together. The lens excitation is limited by the current density in a limited space; resolution of the lens is limited as a result. Yet another approach utilizes non-shielded permanent magnets and shared magnetic conductors to create a focusing lens array. All of these prior approaches either have two focusing lens fields, complicating deflection and detection system design, or they exhibit a strong leakage field along the optical axis, which degrades resolution and field of view.

Permanent magnet based high performance multi-axis immersion electron lens array with low axial leakage field


A biological substance quantification method of quantifying an expression amount of a specific biological substance in a target sample stained by using a fluorescent dye accumulated particle bondable to the specific biological substance. The method includes: inputting a first fluorescent image obtained by image capturing of the target sample; extracting a bright spot portion from the first fluorescent image and calculating a first luminance value; and calculating a number of the fluorescent dye accumulated particle included in the bright spot portion by using the first luminance value, a second luminance value of a bright spot portion extracted from a second fluorescent image obtained by image capturing of a standard sample, and a distribution of a third luminance value of each bright spot portion in a third fluorescent image obtained by image capturing of a preparation on which the fluorescent dye accumulated particle is dispersed without aggregating.

In pathological diagnosis, quantifying the expression amount of biological substance overexpressing in a tissue slice is very important information for prognostic expectation and for determination of future treatment plan. In such quantification of biological substance, development of the method that enables accurate quantification of biological substance and extraction of region of interest has been desired for analysis of the expression amount of a specific biological substance in the region of interest that is an analysis target region set in the tissue slice.
Conventionally, there has been used a method of extracting fluorescent bright spots from a fluorescent image obtained by image capturing of a tissue sample stained with a specific protein using fluorescent dye accumulated particles, and counting the number of bright spots to perform the quantitative evaluation of biological substance. However, there may be cases where a single bright spot is seen on the fluorescent image but actually a plurality of fluorescent dye accumulated particles are accumulated. Thus, accurate quantification by merely simply counting the number of bright spots is difficult.
Thus, for example, Patent Document 1 describes the method of extracting the fluorescent bright spots from the fluorescent image similarly to the above method and calculating the number of particles included in each bright spot on the basis of the average luminance value per fluorescent dye accumulated particle. The method described in Patent Document 1 calculates the luminance value in each bright spot and uses the luminance value to be a mode as the average luminance value per fluorescent dye accumulated particle.
However, since the luminance value varies according to the difference of luminance per fluorescent dye accumulated particle; difference of light source intensity, microscope individual difference, difference of creation condition of fluorescent dye accumulated particles, difference of staining condition of the tissue slice to be a target, and the like, the luminance variations due to these factors need to be corrected in order to improve the accuracy of quantification. With respect to such a problem, the correction of luminance variation is performed by a method of performing sensitivity calibration of the fluorescent detector (for example, refer to Patent Document 2) and a method of performing standardization regarding the intensity of excitation light source and microscope specific factors (for example, refer to Patent Document 3).

Biological substance quantification method, image processing device, pathological diagnosis support system, and recording medium storing computer readable program


The present invention sets forth a cover for orthopedic splinting rods which in its broadest sense consists of a single sheet of material which is formed into a cylinder having a longitudinal slit up its side and a curved top of the cap to allow for natural resilience of the cap to hold the walls of the cap against the edges of the ends of the splinting rods to insure retention of the cap on the ends of the splinting rods. An embodiment for external use encases the metal cap in a plastic housing having a gripping means to assist in the installation of the cap on the ends of the splinting rod. The external housing has smooth surfaces to avoid and minimize any trauma to surrounding body tissue. The present invention also sets forth a method of installing a cover for orthopedic splinting rods on the clipped end of the rod, comprising the steps of rotating of the metal cap down over the burred end of the clipped splinting rods and rotating the cap to cause the burred end of the cap to form a helical groove or type of screw thread in the wall of the cap to secure the cap. The cap tapers at the end near the top so as to cause ever increasing pressure on the burred end of the splinting rod as it is advanced into the cap.

1. Field of the Invention
This invention relates to appliances for orthopedic surgery and more specifically to devices to cover the ends of splinting rods used to assist in the healing of bone fractures and the like.
2. Description of the Prior Art
Orthopedic splinting rods have gained wide application in orthopedic surgical procedures. Initially used for large fractures such as hip pinnings, these rods have become ever more frequently used and are now commonplace for many types of procedures seeking to properly set bone fragments o segments and fix the position of these segments during the knitting process. As these splinting rods have become ever more widely used, problems have been noticed with respect to the presence of the rod in the patient or extending out of the patient during the prolonged period of healing. The end of the rod is cut, often by clipping, to shape so as not to protrude for an extended length. But the end is often sharp and, if inside the body, can produce extensive trauma to surrounding tissues by causing lacerations and/or also acting as a site for infection. If the end of the rod is positioned outside the body as, for example, in the pinning of finger bones or toe bones, the ends of the rods can lacerate the surrounding tissue mass, such as adjoining fingers or toes, etc., and, again, are a hazard.
Many attempts have been made to provide appropriate covers for these types of splints, but none of them have been totally successful. An example of such a device is shown in U.S. Pat. No. 4,688,560 to Schultz.
The ideal device should be small, light, securely held to the end of the orthopedic splint and be easy to apply and difficult to remove. Many of the prior devices achieve some or many of these objectives. However, the cost and ease of manufacturing is another factor which has to be considered and simplicity of the product is also of paramount importance both from the standpoint of reliability, ease of application, and also from the aspect of per unit cost.

Cover for orthopedic splinting rods and method of installation


A method for operating a satellite (10, 34) so that a predetermined amount of propellant is left to perform end-of-life maneuvers is provided. A first pressure differential is established between a first propellant tank (74) and a second propellant tank (76) by pressurizing the first tank (76) with a pressurant. Propellant is transferred from the first tank (74) to the second tank (76). Using precise control during transfer the amount of propellant in the tanks may be determined. The second tank is used for orbit maintenance. The first tank is used for end-of-life maneuvers. Propellant may be transferred between the tanks using a latching process so that a predetermined amount of propellant may be transferred from a non-empty tank to a full tank.

The present invention relates to a method for controlling the operation of a satellite and, more particularly, to a method for controlling the operation and consumption of propellant of a satellite near the end of its useful life.
Spacecraft propulsion systems typically use at least two tanks that store propellant therein. The propellant is used to maintain the satellite orbit due to various perturbations in space. A small amount of propellant is required to maintain an orbit. When a satellite nears the end of its propellant, the satellite may be placed in a graveyard orbit.
A standard method for determining the amount of remaining propellants on a spacecraft in orbit is to keep track of the propellant by bookkeeping. That is, keeping track of the burn times and thus deducing the amount of propellant used which can be subtracted from the propellant loaded prior to launch. This method is typically no more accurate than one percent of the loaded propellant. This amount may be equivalent to several months of station keeping. Because the spacecraft is deorbited several months before the end of its useful life, potentially millions of dollars in revenue are lost.
Another method for performing determining the amount of propellant is described in the parent application. The method described therein set forth a method that empties a tank. In some situations, it may not be desirable to empty a tank.
The present invention proposes a method that can be used on existing and future spacecraft to predict the amount of propellant therein without incorporating expensive fuel gauge type devices and without emptying a fuel tank.

This patent is subject to a terminal disclaimer.

Method of operating a satellite for end-of-life maneuvers


Embodiments of the present invention disclose a system and method for multi-viewpoint video capture. According to one embodiment, the system includes a camera housing for accommodating both a first multi-imager set and a second multi-imager set, with each multi-imager set including a plurality of optical cameras having different viewpoint directions and configured to produce a source image. Furthermore, each camera in the first multi-imager set and the second multi-imager set include corresponding cameras facing in approximately the same viewpoint direction. The first multi-imager set is positioned laterally adjacent to the second multi-imager set such that lines joining a center of projection of corresponding cameras in the first multi-imager set and second multi-imager set are approximately parallel.

The functionality and performance of video acquisition systems has changed drastically within the last decade. For example, such video systems have advanced from standard definition video capture to high definition video capture and from two-dimensional video acquisition to three-dimensional video capture and output. In particular, sports broadcast and video capture methods require large numbers of operations personnel, detailed and taxing mechanical procedures, and complex equipment. Furthermore, as the demand for three-dimensional video increases, this task becomes even more challenging and expensive.

System and method for multi-viewpoint video capture

Hewlett-Packard Development Company, L.P.


A plasma ignition system includes an ignition plug attached to an engine and having a center electrode, a ground electrode, and a discharge space, a discharge power source circuit, a plasma generation power source circuit, a resistance element between the discharge power source circuit and the center electrode, a rectifying device between the plasma generation power source circuit and the center electrode, and an element receiving portion in a periphery of the center electrode. The plug puts gas in the discharge space into a plasma state to ignite a fuel/air mixture in the engine, as a result of application of high voltage to the plug by the discharge power source circuit and supply of high current to the plug by the plasma generation power source circuit. The resistance element and the rectifying device are placed in the receiving portion.

1. Field of the Invention
The present invention relates to measures to prevent leakage of electromagnetic wave noise in a plasma ignition system, which is used for ignition in an internal combustion engine.
2. Description of Related Art
Recently, from a standpoint of environmental protection, lean mixture combustion or supercharged mixture combustion, for example, is required in an internal combustion engine to reduce emissions in combustion exhaust gas or to improve fuel mileage, so that an ignition condition is becoming severe. Accordingly, an ignition system, in which stable ignitionability is achieved, is required in an engine of poor ignitionability.
In the case of ignition of the engine, an ignition system using an ordinary spark plug 10z shown in FIG. 10A includes a battery 31z, an ignition switch 32z, an ignition coil 33z, an electronic control unit (ECU) 35z, an ignition coil drive circuit (transistor) 34z, a rectifying device 21z, and the spark plug 10z. As shown in FIG. 10B, when the ignition switch 32z is thrown, a primary voltage having a low voltage is applied to a primary coil 331z of an ignition coil 33z from the battery 31z in response to an ignition signal from the ECU 35z. Subsequently, when the primary voltage is cut off through the switching of the ignition coil drive circuit 34z, a magnetic field in the ignition coil 33z changes, and thereby a secondary voltage in a range of −10 to −30 kV is generated in a secondary coil 332z of the ignition coil 33z. As a result, electric discharge takes place in a center electrode 110z and a ground electrode 131z, and accordingly a high-temperature region is generated in a small area. In the case of the ignition by the ordinary spark plug 10z, the above high-temperature region serves as a source of ignition to excite ignition and explosion of a compressed air-fuel mixture. Meanwhile, a current of about 35 mA rectified through a diode 21z passes through the secondary coil 332z during a conducting period of about 2 ms, which is a relatively long duration, and energy of about 35 mJ is released to the spark plug 10z.
In the case of ignition by a plasma ignition system 1x shown in FIG. 12A, when an ignition switch 31x is thrown (see FIG. 12B), a primary voltage having a low voltage is applied to a primary coil 321 of an ignition coil 32x from a discharge battery 30x. By switching of an ignition coil drive circuit (transistor) 33x controlled by an electronic control unit (ECU) 34x, the primary voltage is cut off and thereby a magnetic field in the ignition coil 32x changes. Consequently, a secondary voltage in a range of −10 to −30 xV is generated in a secondary coil 322x of the ignition coil 32x. The insulation in a discharge space 140x breaks down and electric discharge is started when the secondary voltage reaches a discharge voltage proportional to a discharging gap in the discharge space 140x formed between a center electrode 110x and a ground electrode 130x. Meanwhile, energy (e.g., −450V, 120 A) stored in a capacitor 42x from a plasma energy supply battery 40x, which is provided separately from the discharge battery 30x, is released to the discharge space 140x at once. Accordingly, gas in the discharge space 140x enters into a high-temperature and pressure plasma state, and is injected through an opening 132x formed at a leading end of the discharge space 140x. As a result, a very high temperature range in a range of thousands to tens of thousands of degrees Celsius and having great directivity is generated in a wide range of volume. Thus, such a plasma ignition system is expected to be applied to an ignition system in an internal combustion engine of difficult ignitionability in which lean mixture combustion or supercharged mixture combustion, for example, is performed. In addition, when the plasma ignition system is applied to the ordinary spark plug, plasma having high energy is generated between electrodes of the plug. Therefore, improvement in ignitionability is expected.
However, in the conventional plasma ignition system Ix, the energy stored in the capacitor 42x for plasma generation is instantaneously supplied to a plasma ignition plug 10x. Consequently, as shown in FIG. 12B, a high current of about 120 A is passed for a conducting period of about 8 μsec, which is an extremely short duration. Since the above passing of high current is periodically repeated according to rotation of the engine, an electromagnetic wave noise of high frequency is generated. Malfunction of the electronic control unit installed in a vehicle or the like is caused by such an electromagnetic wave noise, and as a result, an accidental fire of the engine may be caused. As a method for preventing the above electromagnetic wave noise, a method for blocking the electromagnetic wave noise is disclosed in JP55-172659U corresponding to U.S. Pat. No. 4,327,702. The electromagnetic wave noise is blocked, by using a shielding wire for a wiring for plasma generation connecting a plasma generation power source and a plug, giving an electromagnetic wave shield to cover the whole plug, and using a resistance wire for a wiring for electric discharge connecting an electric discharge power source and the plug.
Nevertheless, the internal combustion engine such as a car motor usually includes a plurality of cylinders, and accordingly, the electromagnetic wave shield needs to be given over a very wide range when the conventional method illustrated in JP55-172659U is employed. In a plasma ignition system, in which a plurality of plasma ignition plugs 10x (1), 10x (2), 10x (3), 10x (4) is connected to an ignition coil 32x via a distributor 60x, as shown in FIG. 11, when a shielding wire is used for a plasma generation wiring 400x connected to each plug, the whole plug is covered with an electromagnetic wave shield, and a resistance wire 36x is used for a high voltage supply wiring, in order to restrict the generation of the electromagnetic wave noise, stray capacitances Cs (1 to 6) in electromagnetic wave shield parts Sd (1 to 6) are not constant since the length of each shielding wire differs. Accordingly, it is difficult to maintain an earth potential of each electromagnetic wave shield part at the same electric potential, and thereby an electric potential difference is generated between the electromagnetic wave shields. Such an electric potential difference serves as a generation source of a new electromagnetic wave noise. Also, electric field concentration is generated in a connection part of each electromagnetic wave shield part, and it is difficult to block the electromagnetic wave noise completely.
In addition, a transmit circuit is formed from the ignition coil 32x and the plasma ignition plug 10x as a discharging space. When high voltage is applied from the ignition coil 32x and electric discharge is started, the electromagnetic wave noise is generated and may leak to the outside because a plasma generation wiring connecting a center-electrode terminal area 112x and the capacitor 42x for plasma generation serves as an antenna. In the ordinary spark plug, such transmission of the electromagnetic wave noise is prevented by interposing a resistance element between the ignition coil and the plug. However, as mentioned above, the high current must be passed through the plasma generation wiring. Thus, the electromagnetic wave noise at the time of starting of the electric discharge cannot be absorbed by interposing the resistance element on the plasma generation wiring.


According to one aspect of an example, there is provided a transaction gateway in a first network for receiving a transaction from the first network and for sending the transaction to a transaction processor in a second network. The transaction gateway is arranged to identify restricted data in the transaction, to modify the received transaction by replacing identified restricted data with replacement data different to the identified restricted data, and to send the modified transaction to the transaction processor in the second network.

In order to conduct business or to operate, businesses and organizations often collect data about individuals or objects. The collected data enables individuals or objects to be identified, and may also be used to enable the individuals or objects to be communicated with. Types of data collected in this context may include, for example, personally identifiable information (PII) that may uniquely identify an individual, or that may be used with other data to identify an individual. In the context of an object, such as a computing device, identifying data may include an Internet protocol (IP) address, a host name, and location information.
In many countries the use of personally identifiable information may be governed by legislation, such as data privacy or data protection legislation. One example of such legislation is the United Kingdom's Data Protection Act. In many countries, legislation may prohibit personally identifiable information from being sent outside of a country or region (such as the European Economic Area) borders.
Such legislation, however, may be inconvenient for businesses and organizations that operate in and obtain personally identifiable information from multiple countries or regions.


The subject disclosure provides for a method of eccentricity correction of a borehole shape computation. The method includes deploying a caliper tool into a borehole penetrating a subterranean formation and acquiring field measurements with the deployed caliper tool. The method includes applying, in a processor circuit, an eccentricity correction algorithm to one or more standoff samples from the obtained field measurements, wherein the eccentricity correction algorithm produces a shape fitted curve that represents a measured borehole with a least number of points outside of the shape fitted curve and a least amount of error. The method includes determining eccentricity-corrected borehole coordinates with the applied eccentricity correction algorithm and determining a borehole shape from the eccentricity-corrected borehole coordinates. The method includes determining tool location coordinates relative to the borehole with the determined borehole shape.

Modern oil field operations demand a great quantity of information relating to the parameters and conditions encountered downhole. Such information typically includes characteristics of the earth formations traversed by the borehole, and data relating to the size and configuration of the borehole itself. The collection of information relating to conditions downhole, which commonly is referred to as "logging," can be performed by several methods including wireline logging and "logging while drilling" (LWD).
During exploration and recovery operations, the standoff data of the borehole may be used as an indication of formation stress, compaction, and other mechanisms that operate to deform the borehole. In these and other logging environments, an image of the borehole wall can be constructed with the standoff data. Among other things, such images reveal the fine-scale structure of the penetrated formations. However, assessing the standoff data of the borehole rapidly and accurately, especially when the logging tool acquiring the associated data moves off-center, can be difficult.
In one or more implementations, not all of the depicted components in each figure may be required, and one or more implementations may include additional components not shown in a figure. Variations in the arrangement and type of the components may be made without departing from the scope of the subject disclosure. Additional components, different components, or fewer components may be utilized within the scope of the subject disclosure.

Eccentricity correction algorithm for borehole shape and tool location computations from caliper data


A handle returning apparatus includes an input member rotating in conjunction with an operation handle operated to rotate in first and second directions including a neutral position, and a spring member including a plurality of spring portions integrally formed and including different spring forces, each of the spring portions including first and second spring end portions separating from each other in a circumferential direction and being configured to generate a returning force for returning the operation handle to the neutral position by being resiliently deformed on a basis of the rotation operation of the operation handle to bias the input member in a state where one of the first and second spring end portions engages with the input member and the other engages with a fixation member. The plurality of spring portions is selectively switchable therebetween to generate the returning force depending on a condition of the rotation operation.

A handle returning apparatus for returning an operation handle that is rotationally operated to a neutral position by an elastic force of a spring member is known. For example, JP2006-83983A which will be hereinafter referred to as Reference 1 discloses a seat lifting apparatus including a lever-type operation handle that is rotationally operated in a first direction and a second direction (i.e., an up-down direction) including a neutral position. The seat lifting apparatus disclosed in Reference 1 also includes an input member that rotates in conjunction with the operation handle and a spring member that includes a spring portion in an arc form (substantially in a C-shape) extending around a rotation axis of the input member. The spring member includes opposed end portions which are away from each other in a circumferential direction, one of the end portions engaging with the input member and the other of the end portions engaging with a fixation member. Accordingly, the input member is biased so that a returning force for returning the operation handle to the neutral position is generated in the handle returning apparatus.
Specifically, the input member rotates relative to the fixation member based on the rotation operation of the operation handle so that the spring member is resiliently or elastically deformed in a manner that the opposed end portions of the spring members are resiliently forced apart from each other. Based on a returning force, i.e., an elastic force, generated by the spring member that is resiliently deformed, the operation handle which is pivotally operated may return to the neutral position.
Nevertheless, in the seat lifting apparatus disclosed in Reference 1, in a case where a difference in operation amount (rotation operation amount) of the operation handle occurs depending on a direction of rotation operation of the operation handle, a difference in possible returning force, specifically, in maximum value of possible returning force, may be generated. For example, in a configuration where the operation handle is provided at a side of a vehicle seat in a width direction thereof as in the seat lifting apparatus in Reference 1, it may be difficult to increase the rotation operation amount of the operation handle in a case where the operation handle is pulled downward as compared to a case where the operation handle is lifted upward. Thus, the resulting difference in returning speed and operation reaction force of the operation handle between the cases where the operation handle is pulled downward and where the operation handle is lifted upward affects returning performance of the operation handle, which may be a cause of decrease of operation feeling of the operation handle.
A need thus exists for a handle returning apparatus and a seat lifting apparatus for a vehicle which are not susceptible to the drawback mentioned above.

Handle returning apparatus and seat lifting apparatus for vehicle


A MCM (multi-carrier modulation) receiver that utilizes a plurality sub-carriers (e.g., tones) to transmit information in a frame-by-frame manner. Identify a first subset of sub-carriers that have negligible ISI (inter-symbol interference) and ICI (inter-carrier interference), and a second subset of sub-carriers that ISI/ICI cancellation is needed to improve the performance. For sub-carriers in the first subset, conventional equalization is performed to obtained soft decisions. For those sub-carriers in the second subset, perform ISI cancellation then ICI cancellation along with equalization. For sub-carriers in the second subset, identify a series of third subsets (one for each of the sub-carriers in the second subset) that cause interference to the sub-carriers in the second set. For sub-carriers in the third subset, identify a series of fourth subsets from a previous frame that cause interference to the sub-carriers in the third set. For each element in the third subset, perform ISI cancellation to eliminate the interference from the elements in the fourth subset from the previous frame to obtain a plurality of intermediate decisions. For each element in the second subset, perform a combined equalization and ICI cancellation using said intermediate decisions. The selection of the first subset, the second subset, the third subset, and the fourth subset are based on examining the frequency response of the communication channel.

1. Field of the Invention
The present invention relates to multi-carrier modulation receivers, in particular to apparatus and methods of canceling the inter-symbol and inter-carrier interference in a multi-carrier modulation receiver.
2. Description of Related Art
MCM (multi-carrier modulation) is a modulation scheme that is widely used in high-speed data communications. It has two flavors: OFDM (orthogonal frequency division multiplexing), which is currently used in wireless local area network standards IEEE 802.11a and 802.11g, and DMT (discrete multi-tone), which is currently used in ADSL (asymmetrical digital subscriber line) standards and VDSL (very-high-speed digital subscriber line) standards. While the DMT system is discussed throughout the specification and is used as an example, one of ordinary skill in the art will realize that the techniques disclosed by the present invention are applicable to a general MCM system.
A DMT transmitter utilizes a plurality of tones, (or so-called sub-carriers, which are sinusoidal waves), which are orthogonal to one another. Each tone may carry a certain bit-load of information using a certain modulation scheme, such as 4-QAM (4-point quadrature amplitude modulation, which carries 2-bit information), 16-QAM (16-point quadrature amplitude modulation, which carries 4-bit information), 64-QAM (64-point quadrature amplitude modulation, which carries 6-bit information), and so on. The total bit loads provided by all the tones determines the total number of data bits that a DMT symbol carries. For example, if there are 256 tones and each of them uses 16-QAM, then the total amount of data a DMT symbol carries is 256×4=1024 bits. A DMT transmission system thus operates on a per-frame basis. Each frame consists of a block of data bit stream whose length is equal to the total number of bits that a DMT symbol carries. For example, if each DMT symbol carries 1024 bits, then the data to be transmitted from the transmitter are divided into many blocks; each block has 1024 bits and is carried by a DMT symbol.
FIG. 1 depicts a typical block diagram of the physical layer implementation of a DMT transmitter 1000. A SIPO (serial-in parallel-out) buffer 1100 converts a block of data bit stream (that a DMT symbol carries) into N parallel data banks, labeled 0, 1, 2, to N−1. Each entry of the N banks is mapped into a respective QAM (quadrature amplitude modulation) constellation point by the subsequent QAM mapper 1200, resulting in N complex (frequency domain) data, labeled 0, 1, 2, to N−1. A constellation diagram is a representation of a digital modulation scheme in the complex plane. The points on a constellation diagram are called constellation points. Constellation points are a set of modulation symbols that comprise a modulation alphabet.
For example, if tone number 5 uses 4-QAM to carry two bits of information, the QAM mapper 1200 will map the two-bit input corresponding to tone number 5 into one of the four constellation points: 1+j, 1−j, −1+j, −1−j. The N complex data from QAM mapper 1200 are converted into N time-domain samples by N-point IFFT (inverse Fast Fourier Transform) 1300. The last N_CP time-domain samples of the IFFT output are pre-pended to the beginning of the N samples, resulting in N+N_CP time-domain samples, labeled 0, 1, 2, to N+N_CP−1. These N_CP pre-pended samples are referred to as "cyclic prefix" (CP) of the corresponding IFFT frame. A subsequent PISO (parallel-in serial-out) buffer 1500 converts the N+N_CP time domain samples into N+N_CP serial samples, which is converted into analog voltage by a DAC (digital-analog converter) 1600. An amplifier 1700 amplifies the output from DAC 1600 to make it suitable for transmission over the communication medium 2020 (for example, communication lines such as telephone lines). The N+N_CP serial samples constitute a DMT symbol that carries a certain block of information.
FIG. 2 illustrates the construction of two successive DMT symbols. The last N_CP samples of the IFFT output corresponding to the first block of data bit stream are pre-pended to the beginning of the first DMT symbol. Similarly, the last N_CP samples of the IFFT output corresponding to the second block of data bit stream are pre-pended to the beginning of the second DMT symbol. "Cyclic prefix" carries redundant information that is readily available. However, it provides a "guard interval" that serves as a buffer between two successive DMT symbols. Without the guard interval, the information carried by the 1st DMT symbol will leak to the 2nd DMT symbol and causes interference when the two successive DMT symbols are transmitted over the communication channel 2020, since every practical communication channel has a non-zero length in its impulse response. When a CP is employed with a length is longer than the impulse response of the communication channel, the leakage of the information carried by the first DMT symbol will be contained within the guard interval between the two DMT symbols. The CP, however, is an overhead to the system. A longer CP allows the system to tolerate more dispersion from the communication channel at the cost of a lower data throughput.
FIG. 3 depicts a typical block diagram of a DMT receiver. The received signal from the communication medium 2020 is amplified by an amplifier 2050, then filtered by a filter 2100, and then converted into digital samples by a ADC (analog-digital converter) 2200. As mentioned above, the cyclic prefix in the transmitter needs to be longer than the length of the impulse response of the communication channel 2020 for the leakage of information from one DMT symbol to be contained within the guard interval. In practice, however, the impulse response of the communication channel may be longer than the CP because the CP is usually not long enough (otherwise the sacrifice in data throughput will be prohibitively high). Therefore, a TEQ 2300 (time-domain equalizer) is often used. TEQ 2300 is essentially a FIR (finite impulse response) filter whose purpose is to effectively shorten the impulse response of the communication channel, so that the leakage of the information from one DMT symbol can be contained within the guard interval.
From the output of the TEQ 2300, the frame boundary between two successive DMT symbols is detected. The output of TEQ 2300 is then converted by a SIPO (serial-in parallel-out) 2400 into successive blocks of time domain samples; each block consists of N+N_CP time-domain samples, labeled 0, 1, 2, . . . , N+N_CP−1. The 1st N_CP samples of each frame, which corresponds to the samples within the guard interval, are discarded in the subsequent CP removal 2500, resulting in N samples, labeled 0, 1, 2, to N−1. The N time-domain samples are transformed into N frequency-domain samples by the N-point FFT (fast Fourier transform) 2600.
Ideally, we would like the N frequency-domain samples at the output of FFT 2600 to exactly match the N frequency-domain data at the input of IFFT 1300 in the transmitter depicted by FIG. 1. Unfortunately, due to the communication channel, the amplitude and phase of each sub-carrier is altered and therefore the frequency domain sample at the receiver will not exactly match that at the transmitter. A FEQ (frequency domain equalizer) 2700 is used to equalize the frequency-domain samples. The amplitude and phase change experienced by each tone is thus corrected independently on a per-tone basis. After frequency domain equalization, a slicer 2750 is used to decide the most likely constellation point that the transmitter originally uses for each tone. For example, if the output of FEQ 2700 is 0.9+1.1j for tone number 5 which uses 4-QAM to carry two-bit information, then slicer 2750 would decide that the most likely constellation point that the transmitter originally uses for tone number 5 is 1+1j. The slicer is an example embodiment of a "decision device," as it makes a most likely decision for each tone. The output from slicer 2750 is mapped to N data banks by the subsequent "QAM demapper" 2800. The N data banks from "QAM demapper" 2800 are converted back into a block of data bit stream by PISO 2900, which ideally will match that of the input of SIPO 1100 in the transmitter.
To effectively shorten the length of the effective impulse response of the communication channel, many algorithms for calculating the coefficients for TEQ are proposed, including minimum mean square error (MMSE), maximum shortening SNR (MSSNR), minimum ISI (mini-ISI), and maximum bit rate (MBR). Among these, MBR offers the best performance, but the computation complexity is too high to be implemented in a commercial MCM receiver. In many cases, unfortunately, none of the algorithms are able to lead to a practical solution that completely contains the leakage of the information from a DMT symbol within the guard interval (i.e. CP of the next DMT symbol). Under these circumstances, a DMT symbol causes interference to the next DMT symbol. This phenomenon is known as "inter-symbol interference" (ISI).
When the length of impulse response of the communication channel exceeds the CP length, a DMT symbol will fail to settle into steady state within its guard interval (i.e. the CP portion of this DMT symbol). In other words, there is still some transient behavior within the "useful" part, i.e. the last N samples, of the DMT symbol. DMT modulation, as a special of MCM, relies on the orthogonality of carriers to faithfully deliver the information. The orthogonality between two tones holds only when they are both in steady state, where both become purely sinusoidal. Whenever the DMT symbol fails to settle into steady state within its guard interval, the orthogonality among tones used by this DMT symbol fails. There is then coupling between the information carried by any two tones that it uses. This phenomenon is known as "inter-carrier interference" (ICI).
One way to alleviate the ISI/ICI problem is to employ multiple TEQ's (time-domain equalizers). For example, a dual-TEQ architecture is shown in FIG. 4. This receiver divides the tones into two groups and employs TEQ1 and TEQ2. Each TEQ is optimized to minimize the ISI/ICI for one group of tones. Each TEQ output is converted into frequency domain samples in a respective FFT. On a per-tone basis, this receiver determines which output of the two paths yields the best SNR (signal-noise ratio). Once the better path for each tone is determined, the output from the path is equalized by the subsequent FEQ. A drawback for this type of receiver is that the hardware cost of this architecture is rather high.
Another way to alleviate the ISI/ICI problem is to employ a so-called "per-tone frequency domain equalizer" (PTFEQ), as shown in FIG. 5. The principle in this design is to eliminate TEQ by replacing it with a plurality of tapped delay lines 5010 in the frequency domain at the FFT output. The hardware cost, however, is prohibitively high for a commercial MCM receiver.
U.S. patent application Ser. No. 11/256,707 filed Oct. 24, 2005, titled "Inter-Symbol and Inter-Carrier Interference Canceller for Multi-Carrier Modulation Receiver," by inventors Chia-Liang Lin, Heng-Cheng Yeh and Cheng-Hsian Li, taught a cost effective method for performing ISI/ICI cancellation. However, in that scheme, the inputs to the ICI canceller contain ISI, therefore, there are certain interplays between ICI and ISI. Also, the method of finding the ISI/ICI coefficients taught in that application is based on LMS (least mean square) adaptation, which can be prone to noise and may take a long time to converge, given that the coefficients can only be adapted once per DMT symbol and each DMT period is usually relatively long.
There remains a need in the art for a low-cost, robust, and effective scheme for performing ISI/ICI cancellation.

Low noise inter-symbol and inter-carrier interference cancellation for multi-carrier modulation receivers


An oscillation circuit for producing an output whose frequency accords with an input voltage value, comprises a first constant current source whose current has a value according to the input voltage value; a charge capacitor to be charged by the first constant current source; a comparator having one input terminal connected to a charge terminal of the charge capacitor to be charged and an other input terminal supplied with a first reference voltage, for comparing inputs to both input terminals with each other and outputting an output signal having a high level or a low level; and first switch means for pulling down a potential of the charge terminal of the charge capacitor to a second reference voltage lower than the first reference voltage under control by an output of the comparator.

1. Field of the Invention
The present invention relates to a voltage-controlled oscillation circuit for producing an output having a frequency proportional to the value of the input voltage and a PLL (Phase Locked Loop) circuit using this oscillation circuit, and, more particularly, an improvement of an oscillation circuit capable of producing an output in a sufficient frequency range even on a low supply voltage.
2. Description of the Related Art
PLL circuits are widely used in, for example, digital display devices, radio devices and the like, as circuits to produce an output synchronous with a received signal. This PLL circuit generally has a voltage-controlled oscillation circuit, which compares the phase of an input signal with the phase of a comparison signal that is acquired by frequency-dividing an output signal by N, and produces a voltage proportional to a voltage value corresponding to the difference between those phases.
FIG. 11 is a circuit diagram exemplifying a conventional voltage-controlled oscillation circuit. This oscillation circuit comprises a voltage-current converter 1, a charge/discharge capacitor C1, comparators comp1 and comp2 and a latch circuit 16. The voltage-current converter 1 produces currents I1 and I2 proportional to the voltage value of an input voltage Vin and alters the direction of the current output according to the output of the latch circuit 16. The comparators comp1 and comp2 respectively use a lower reference voltage VRL and a higher reference voltage VRH for comparison. The voltage-current converter 1 includes a differential amplifier 15, current sources P1, P2 and P3, each comprised of a P channel transistor, which respectively produce currents i10, i11 and Il proportional to the voltage value of the input Vin, N channel transistors Q1 and Q2, which constitute a current mirror circuit, and switches SW1 and SW2.
FIG. 12 is a waveform diagram for explaining the operation of the voltage-controlled oscillation circuit. When an output Vout is at an H level, for example, the switch SW1 is closed and the switch SW2 is opened, causing the current I1 to flow toward the capacitor C1 from the current source P3. Consequently, the potential at a node A rises toward the reference voltage VRH. When the potential at the node A reaches the reference voltage VRH, the output of the comparator comp2 changes its level to an L level from an H level, causing the level of the output Vout to be changed to an L level from the H level via the latch circuit 16. As a result, the switch SW1 is open and the switch SW2 is closed, causing the current I2 to flow from the capacitor C1 to the current source I2. When the potential at node A reaches to the reference voltage VRL, the output of the comparator comp1 changes to an L level from an H level, causing the level of the output Vout to be changed to an L level from the H level via the latch circuit 16. As a result, the switch SW2 is opened and the switch SW1 is closed. The operation then returns to the initial operation.
As apparent from the above, the potential at the node A oscillates between the reference voltages VRH and VRL in accordance with the charging and discharging of the capacitor C1. This provides a clock pulse having H and L levels as the output Vout. The frequency of the output Vout then is determined by the charging and discharging speeds of the capacitor C1, which are proportional to the current values I1 and I2 proportional to the input voltage Vin. Thus, the output Vout has a frequency according to the value of the input voltage Vin.
Because of the recent tendency of the supply voltage becoming lower and lower, however, the voltage-controlled oscillation circuit shown in FIG. 11 suffers a narrower output frequency range. As the supply voltage becomes lower, particularly, the low frequency range tends to be lost.
The two comparators comp1 and comp2 receive the high reference voltage VRH and low reference voltage VRL respectively as their one input signals and receive the voltage at the node A as the other input signals. The comparator comp1 performs comparison on a voltage near the low reference voltage VRL, and the comparator comp2 performs comparison on a voltage near the high reference voltage VRH. From the viewpoint of the circuit characteristics of comparators, therefore, it is necessary to set the reference voltages VRH and VRL within the input range that guarantees the linear operation of comparators.
As the supply voltage decreases, the input range of comparators is inclined to become narrower. If the reference voltages VRH and VRL are designed to lie within that narrow input range, therefore, their voltage difference .DELTA.V (=VRH-VRL) becomes smaller. This means that the pulse width of the output Vout shown in FIG. 12 is reduced, and the output frequency range shifts toward the higher level, thus losing the low frequency band.
If the reference voltages VRH and VRL are set to the upper and lower limits of the input range that can ensure the linear operation of comparators, the operation of comparators becomes slower. This slows the response speed of the voltage-controlled oscillation circuit, shifting the oscillation frequency.
This problem may be overcome by increasing the capacitance of the capacitor C1. This solution however increases the area of the capacitor formed in the integrated circuit, and thus stands in the way of improving the integration. Alternatively, the currents I1 and I2 may be designed to be smaller, which however is not so preferable because the current values that are proportional to the input voltage Vin should be set as large as possible in order to increase the operational range of the voltage-controlled oscillation circuit.

Oscillation circuit and PLL circuit using same


Compositions and methods are provided for the introduction and the regulated expression of genes in plants. Compositions include promoter constructs that provide a level of activity useful for the regulated expression of site-specific recombinases, while avoiding premature excision. Further provided are isolated polynucleotides encoding novel babyboom polypeptides, expression cassettes, and plants comprising the same. Methods for the introduction of genes into plants are provided, including methods for plastid transformation and methods for the transformation of tissues from mature seeds and leaves.

Current transformation technology provides an opportunity to engineer plants with desired traits. Major advances in plant transformation have occurred over the last few years. However, most transformation methods rely on the introduction of polynucleotides into embryonic tissues that are rapidly proliferating. Methods that allow for the transformation of more mature tissues would save considerable time and money. Accordingly, methods are needed in the art to increase transformation efficiencies of plants and allow for the transformation of more mature tissues.
Further, it is often necessary to reduce the activity of a transgene because the transgene may negatively affect the growth or fertility of the plant. Recombination systems can be used to excise the transgene, wherein the expression of a site-specific recombinase is regulated by an inducible promoter. Often, these systems are associated with premature excision. Accordingly, methods are needed in the art to efficiently excise transgenes with limited premature excision.

Methods and compositions for the introduction and regulated expression of genes in plants


A conformable pod and related method of production are provided, wherein the pod defines a soft structure adapted for deformation in response to minimal pressure applied manually, i.e., by contact with some portion of the body, and for retaining a deformed configuration substantially in the absence of continued application of manual pressure. The pod includes a viscous and tacky gel material encased within a closed chamber defined at least in part by a relatively thin and easily deformed outer skin providing minimal deformation resistance. In a preferred form, the encased gel material is a coagulated or partially coagulated gel such as a silicone-based gel material adapted for deformation in response to minimal applied manual pressure, and for substantially pressureless retention of the as-deformed shape, followed by relatively slow return (at least two seconds or more) to an initial nondeformed shape upon manual release of the pod.

This application claims the benefit of U.S. Provisional Application No. 60/582,263, filed Jun. 22, 2004.
This invention relates generally to improvements in user contact surfaces and related shape-conforming or biomorphic pods and grips for use with manually grasped implements and the like, such as a cooking vessel, hair brush, paint brush, sports racquet or bat, scissors or other hand tool, musical instrument, or virtually any other type of manually grasped device having a handle or related structure for manual grasping or engagement as by means of the hand or fingertips, or alternatively for contact and/or manipulation by a portion of the human body.
More specifically, this invention relates to an improved bio-conforming or biomorphic pod for mounting onto an appropriate implement surface, and a related method of pod production, wherein the conformable pod incorporates a viscous and tacky gel material adapted for custom-fit shape deformation in response to a minimal applied inertia or pressure, and for substantially pressureless retention of the deformed shape until the pod is released by the user. When released, the pod exhibits a memory characteristic for subsequent relatively slow return to an initial nondeformed shape.
Manual implements generally incorporate a handle or related structure having a size and shape for appropriate manual grasping and manipulation of the implement during normal intended use. In one common form, the handle structure comprises an elongated rigid element having one end securely connected to the implement, such as a cooking pot or pan, a hair brush, a tool head, etc. In another common form, the manual implement is normally grasped directly about the body thereof, such as in the case of writing and other marking instruments, baseball and softball bats, etc. Other manual implements are adapted for fingertip manipulation and thereby include one or more keypads for fingertip depression, e.g., keyboards, musical instruments such as a clarinet, saxophone, trumpet and the like. Still other manual implements incorporate handle-like structures of alternative or complex shapes, such as circular structures adapted for finger or fingertip insertion to manipulation the implement, e.g., scissors and shears, and the like. In this regard, a variety of improved grips and the like have been developed in recent years for use with such manual implements, primarily to provide a relatively soft and resilient grip structure for enhanced user comfort and/or improved control of the manual implement during normal use thereof.
In the past, grip structures for relatively large manual implements such as sports equipment, hand tools, cookware, and hair brushes and the like have generally been limited to relatively soft and resilient rubber or rubber-like elastomer sleeves or sleeve-like wraps mounted onto the implement handle. While such grip structures beneficially provide a degree of cushioning for improved user comfort, the relative deformation of such grip structures during normal use is inherently limited. In particular, such cushioned grip structures are generally incapable of significant shape deformation, and thus have generally not provided a high degree of shape deformation needed to assume a customized ergonomical shape conforming closely to the anatomical contours of an individual user's hand and/or fingers. Such customized ergonomic shape is beneficially conducive to substantially optimized user comfort with substantially minimized user fatigue, by providing anatomical pressure relief while enhancing manual dexterity and improving manual control of the related device or implement.
In relatively small manual implements such as writing instruments and the like, cushioned grip structures have been similarly provided in the form of resilient elastomer sleeves and the like. In some designs, an outer resilient sleeve encloses an internal cavity which is filled with a flowable substance such as a deformable putty or relatively viscous liquid adapted to displace and thereby achieve substantial shape deformation when the grip structure is manually grasped. With such deformable grip structures, the outer resilient sleeve can exhibit a relatively high stiffness in comparison with the deformable putty or flowable substance contained within the internal cavity, whereby a significant minimum applied pressure for grip deformation is defined by the structure of the outer sleeve and not by the flowable material contained within said outer sleeve. In this regard, the structural thickness and thus the stiffness of the outer sleeve is often increased with a view toward preventing or minimizing inadvertent sleeve puncture and resultant undesirable escape of the flowable material from the internal cavity. Increased sleeve stiffness is, of course, contrary to a desire to provide a significantly reduced and subtantially minimized pressure threshold for initiating deformation of the deformable grip. Moreover, such grip structures having a flowable material encased within a resilient outer sleeve are difficult to manufacture and assemble.
Exemplary grip structures having a flowable material encased within an outer resilient sleeve or the like are shown and described in U.S. Pat. Nos. 5,000,599 and 6,647,582 and 6,725,505; U.S. Publication Nos. US2003/0123917; and US2003/0051316; and in copending U.S. Ser. No. 09/484,127, filed Jan. 15, 2000 and Ser. No. 10/678,148, filed Oct. 2, 2003; and U.S. Prov. Applns. Nos. 60/473,250, filed May 20, 2003 and 60/477,233, filed Jun. 9, 2003.
There exists, therefore, a need for further improvements in and to deformable pods and grips of the type used with manual implements and the like, particularly wherein a pod or grip structure can be initially deformed to a customized ergonomic shape upon application of an extremely light and substantially minimal pressure, and thereafter retain the deformed shape in substantial conformance with the unique anatomical contours of a specific user, substantially without requiring further or continued application of pressure, followed by relatively slow return upon release of the pod or grip structure substantially to its initial non-deformed shape. The present invention fulfills these needs and provides further related advantages.


One illustrative embodiment of an end effector generally comprises a tip, a housing, and a main insert. The housing may include a housing neck, one or more housing sections, and an end section. A main insert may be positioned within the housing. One or more intermediate inserts may also be positioned in the housing, as may an end insert. During use, it is contemplated that the end effector will provide a user a more ergonomic and comfortable experience, requiring less effort from the user resulting in a less fatigue during use. Additionally, the illustrative embodiments of the end effector may be adjusted for optimal use in an infinite number of applications. Additionally, the locator ensures that the user properly locates the distal end of a rivet to create a uniform nugget as the distal end of the rivet is spread.

By way of background and without limitation, the End Effector 10 disclosed may be used for the installation of rivets which are a permanent mechanical fastener. Before being installed, a rivet consists of a smooth cylindrical shaft with a head on one end. The end opposite the head is called the buck-tail. On installation the rivet is placed in a punched or drilled hole, and the tail is upset, or bucked (i.e., deformed), so that it expands to about 1.5 times the original shaft diameter, holding the rivet in place. To distinguish between the two ends of the rivet, the original head is called the factory head and the deformed end is called the shop head or buck-tail. Because there is effectively a head on each end of an installed rivet, it can support tension loads (loads parallel to the axis of the shaft); however, it is much more capable of supporting shear loads (loads perpendicular to the axis of the shaft). A flush rivet is used primarily on external metal surfaces (aka "work piece") where good appearance and the elimination of unnecessary aerodynamic drag are important. A flush rivet takes advantage of a countersink hole; they are also commonly referred to as countersunk rivets. Countersunk or flush rivets are used extensively on the exterior of aircraft for aerodynamic reasons. Additional post-installation machining may be performed to perfect the airflow. (As discussed in further detail at http://en.wikipedia.org/wiki/Rivet)