A switching power supply circuit in which, in order to both provide high power conversion efficiency of a complex resonant converter having a synchronous rectifier circuit and reduce a circuit scale and cost by simplifying the circuit, a synchronous rectifier circuit of a winding voltage detection system is provided on a secondary side of the complex resonant converter, a coupling coefficient is decreased to about 0.8 by setting a gap length in an isolated converter transformer to about 1.5 mm, and numbers of turns of a primary winding and secondary windings are set such that a level of a voltage induced per turn (T) of the secondary windings is 2 V/T. Thus, since magnetic flux density at a core of the isolated converter transformer is decreased to a certain value or lower, a secondary side rectified current can be in a continuous mode even under a condition of heavy load.

Switching power supply circuits employing switching converters such for example as flyback converters and forward converters are widely known. These switching converters form a rectangular waveform in switching operation, and therefore there is a limit to suppression of switching noise. It is also known that because of their operating characteristics, there is a limit to improvement of power conversion efficiency.
Accordingly, various switching power supply circuits using a resonant converter have been proposed (see for example Japanese Patent Laid-Open No. Hei 11-332233), and put to practical use. A resonant converter can readily provide high power conversion efficiency, and achieve low noise because the resonant converter forms a sinusoidal waveform in switching operation. The resonant converter has another advantage of being able to be formed by a relatively small number of parts.
FIG. 24 is a circuit diagram showing an example of a conventional switching power supply circuit having a resonant converter. In the power supply circuit shown in the figure, a partial voltage resonant circuit is combined with a current resonant converter of an externally excited type.
In the power supply circuit shown in this figure, a full-wave rectifying and smoothing circuit comprising a bridge rectifier circuit Di and one smoothing capacitor Ci is provided to a commercial alternating-current power supply AC. As a result of a full-wave rectifying operation by the bridge rectifier circuit Di and the smoothing capacitor Ci, a rectified and smoothed voltage Ei (direct-current input voltage) is obtained across the smoothing capacitor Ci. This rectified and smoothed voltage Ei has a level equal to that of an alternating input voltage VAC.
The current resonant converter supplied with the direct-current input voltage and switching the direct-current input voltage has two MOS-FET switching devices Q1 and Q2 connected to each other by half-bridge coupling as shown in the figure. Damper diodes DD1 and DD2 formed by body diodes are connected in a direction shown in the figure in parallel with the switching devices Q1 and Q2 between a drain and a source of the switching devices Q1 and Q2, respectively.
A partial resonant capacitor Cp is connected in parallel with the switching device Q2 between the drain and the source of the switching device Q2. A capacitance of the partial resonant capacitor Cp and a leakage inductance L1 of a primary winding N1 form a parallel resonant circuit (a partial voltage resonant circuit). Then, a partial voltage resonant operation, in which voltage resonance occurs only when the switching devices Q1 and Q2 are turned off, is obtained.
The power supply circuit is provided with an oscillation and drive circuit 2 formed by a general-purpose IC, for example, for switching-driving the switching devices Q1 and Q2. The oscillation and drive circuit 2 has an oscillating circuit and a driving circuit. The oscillating circuit and the driving circuit apply a drive signal (a gate voltage) of a required frequency to gates of the switching devices Q1 and Q2. Thereby the switching devices Q1 and Q2 perform switching operation so as to be turned on/off alternately at the required switching frequency.
An isolated converter transformer PIT is provided to transmit a switching output of the switching devices Q1  and Q2 to a secondary side. One end of the primary winding N1 of the isolated converter transformer PIT is connected via a series connection of a primary side parallel resonant capacitor C1 to a junction (a switching output point) between the source of the switching device Q1 and the drain of the switching device Q2, whereby the switching output is transmitted.
Another end of the primary winding N1 is connected to a primary side ground.
A capacitance of the series resonant capacitor C1 and the leakage inductance L1 of the isolated converter transformer PIT including the primary winding N1 form a primary side series resonant circuit for converting an operation of a primary side switching converter into a current resonance type operation.
According to the above description, the primary side switching converter shown in this figure obtains the current resonance type operation by the primary side series resonant circuit (L1-C1) and the partial voltage resonant operation by the partial voltage resonant circuit (Cp//L1) described above.
That is, the power supply circuit shown in this figure employs a form in which a resonant circuit for making the primary side switching converter a resonant converter is combined with another resonant circuit. In the present specification, such a switching converter will be referred to as a complex resonant converter.
Though not described with reference to a figure, the isolated converter transformer PIT has an E-E-shaped core formed by combining E-shaped cores made of a ferrite material with each other. The isolated converter transformer PIT has a primary side winding part and a secondary side winding part divided from each other. The primary winding N1 and a secondary winding (N2A and N2B) to be described next are wound around the central magnetic leg of the E-E-shaped core.
Two secondary windings N2A and N2B divided by being provided with a center tap are wound as the secondary winding of the isolated converter transformer PIT. An alternating voltage corresponding to the switching output transmitted to the primary winding N1 is induced in the secondary windings N2A and N2B.
In this case, the center tap of the secondary windings N2A and N2B is connected to a secondary side ground. A full-wave rectifier circuit comprising rectifier diodes D01 and D02 and a smoothing capacitor C0 is connected to the secondary windings N2A and N2B as shown in the figure. Thereby a secondary side direct-current output voltage EO is obtained as a voltage across the smoothing capacitor C0. The secondary side direct-current output voltage EO is supplied to a load side not shown in the figure, and also inputted from a branch point as a detection voltage for a control circuit 1 to be described next.
The control circuit 1 supplies a detection output corresponding to change in level of the secondary side direct-current output voltage EO to the oscillation and drive circuit 2. The oscillation and drive circuit 2 drives the switching devices Q1 and Q2 so as to vary switching frequency according to the supplied detection output of the control circuit 1. Thus varying the switching frequency of the switching devices Q1 and Q2 stabilizes the level of the secondary side direct-current output voltage.
Operating waveforms when the power supply circuit having the circuit configuration shown in this figure meets a load condition of low voltage and high current are shown in FIG. 25. The operating waveforms shown in FIG. 25 were obtained when a measurement was performed under a condition of the alternating input voltage VAC=100 V and load power Po=125 W. The condition of the low voltage and the high current in this case is a condition of the secondary side direct-current output voltage Eo=5 V and a primary side series resonance current Io as a switching current of the primary side switching converter=25 A.
In obtaining the experimental result of the operating waveforms shown in FIG. 25, conditions and parts and elements and the like of the power supply circuit are selected as follows.
First, numbers of turns of the secondary windings N2A and N2B and the primary winding N1 are set such that a voltage level induced per T (turn) of the secondary side winding is 5 V/T. Specifically, the secondary winding N2A=N2B=1 T and the primary winding N1=30 T.
Then, a gap about 1.0 mm is formed in the central magnetic leg of the E-E-shaped core of the isolated converter transformer PIT. Thereby a coupling coefficient of about 0.85 is obtained between the primary winding N1 and the secondary windings N2A and N2B.
The primary side series resonant capacitor C1=0.068 g F and the partial voltage resonant capacitor Cp=330 pF are selected. As the rectifier diodes Do1 and Do2, 50A/40V Schottky diodes are selected.
A voltage V1 across the switching device Q2 in the waveform chart of FIG. 25 corresponds to an on/off state of the switching device Q2. Specifically, the voltage V1 is a rectangular wave, having a zero level during a period T2 when the switching device Q2 is turned on and being clamped at a predetermined level during a period T1 when the switching device Q2 is turned off. A switching current IDS2 flowing through the switching device Q2//the damper diode DD2 has a waveform such that during the period T2, the switching current IDS2 flows through the damper diode DD2 and is thus of a negative polarity at a time of turn-on, and is then inverted to flow through the drain and the source of the switching device Q2 with a positive polarity, and such that the switching current IDS2 is at a zero level during the off period T1.
The switching device Q1 performs switching so as to be turned on/off alternately with the switching device Q2. Therefore a switching current IDS1 flowing through the switching device Q1//the damper diode DD1 has a waveform shifted 180° in phase with respect to the switching current IDS2.
A primary side series resonance current Io flowing through the primary side series resonant circuit (C1-L1) connected between the switching output point of the switching devices Q1 and Q2 and the primary side ground is a waveform resulting from a synthesis of a sinusoidal wave component as the resonance current of the primary side series resonant circuit (C1-L1) and a sawtooth wave component produced by an exciting inductance of the primary winding N1, the waveform corresponding to a synthetic waveform of the switching current IDS1 and the switching current IDS2.
A measurement condition of the load power Po=125 W in this case is a condition of a heavy load close to a maximum as a load condition met by the power supply circuit shown in FIG. 24. In the condition of the heavy load in a range of load power handled by the power supply circuit, a rectified current on the secondary side is in a discontinuous mode.
Specifically, as shown in FIG. 25, a secondary winding voltage V2 occurring at the secondary winding N2A has a waveform clamped at a predetermined absolute value level only during a period when the primary side series resonance current Io flows in the form of a sinusoidal wave, and has a zero level during an interval when the sawtooth wave component produced by the exciting inductance flows as the primary side series resonance current Io between the periods when the primary side series resonance current Io flows in the form of a sinusoidal wave. A waveform obtained by inverting the secondary winding voltage V2 occurs at the secondary winding N2B.
Thus, a rectified current 11 flowing through the rectifier diode Do1 and a rectified current 12 flowing through the rectifier diode Do2 flow only in periods DON1 and DON2, respectively, during which the primary side series resonance current Io flows in the form of a sinusoidal wave. Neither of the rectified currents I1 and I2 flows in other periods. That is, the secondary side rectified current flows into the smoothing capacitor discontinuously.
A forward-direction voltage drop of the rectifier diodes Do1 and Do2 formed by Schottky diodes is 0.6 V. In the above-described operation on the secondary side, since the rectified currents I1 and I2 have a fairly high level of 35 Ap as shown in the figure, a conduction loss by these rectifier diode devices is noticeable, and thus a power loss is increased. As a result of an actual measurement, DC-to-DC power conversion efficiency when the direct-current input voltage (rectified and smoothed voltage Ei)=130 V is only about 86%.
As techniques for reducing the conduction loss of the rectified current on the secondary side, a synchronous rectifier circuit is known which performs rectification by MOS-FETs having a low on resistance. FIG. 26 shows an example of configuration of such a synchronous rectifier circuit using a winding voltage detection system.
Incidentally, FIG. 26 shows only a configuration on a secondary side of an isolated converter transformer PIT. A configuration on a primary side is the same as in FIG. 24. As a constant-voltage control system, a switching frequency control system is employed which variably controls switching frequency of a primary side switching converter according to level of a secondary side direct-current output voltage Eo.
A power supply circuit employing the secondary side configuration shown in FIG. 26 also meets the same condition of low voltage and high current (VAC=100 V, load power Po=125 W, Eo=5 V, and Io=25 A) as in the case of FIG. 24.
Also in this case, one end of each of secondary windings N2A and N2B having the same number of turns as a secondary winding is connected to a center tap. However, an output of the center tap is connected to a positive electrode terminal of a smoothing capacitor C0. Another end of the secondary winding N2A is connected to a secondary side ground (a negative electrode terminal side of the smoothing capacitor C0) via a drain and a source of an N-channel MOS-FET Q3. Similarly, another end of the secondary winding N2B is connected to the secondary side ground (the negative electrode terminal side of the smoothing capacitor C0) via a drain and a source of an N-channel MOS-FET Q4. That is, in this case, the MOS-FETs Q3 and Q4 are inserted in series in respective rectified current paths of the secondary windings N2A and N2B on the negative electrode side. Body diodes DD3 and DD4 are connected to the drain and the source of the MOS-FETs Q3 and Q4, respectively.
A driving circuit for driving the MOS-FET Q3 is formed by connecting a gate resistance Rg1 between a junction between the secondary winding N2B and the drain of the MOS-FET Q4 and a gate of the MOS-FET Q3, and connecting a resistance R11 between the gate of the MOS-FET Q3 and the secondary side ground.
Similarly, a driving circuit for driving the MOS-FET Q4 is formed by connecting a gate resistance Rg2 between a junction between the secondary winding N2A and the drain of the MOS-FET Q3 and a gate of the MOS-FET Q4, and connecting a resistance R12 between the gate of the MOS-FET Q4 and the secondary side ground.
When an on voltage is applied to a gate of a MOS-FET, a region between a drain and a source of the MOS-FET becomes equivalent to a mere resistor, so that a current flows bidirectionally. When the MOS-FET is to be made to function as a secondary side rectifying device, the current needs to flow in only a direction to charge the positive electrode terminal of the smoothing capacitor C0. If a current flows in an opposite direction, a discharge current flows from the smoothing capacitor C0 to the isolated converter transformer PIT side, so that power cannot be efficiently transmitted to a load side. Also, the opposite current causes heat generation of the MOS-FET, noise and the like, and invites a switching loss on the primary side.
The driving circuits described above are circuits for switching-driving the MOS-FETs Q3 and Q4 so that current flows in only the direction to charge the positive electrode terminal of the smoothing capacitor C0 (that is, from the drain to the source) on the basis of detection of a voltage of the secondary winding.
A waveform chart of FIG. 27 shows operation of the power supply circuit employing the secondary side configuration shown in FIG. 26 (the primary side configuration is the same as in FIG. 24) when the load power Po=125 W. As described above, the load power Po=125 W in this case is a condition of substantially a maximum load.
In this figure, a voltage V1 across a switching device Q2 and a corresponding secondary winding voltage V2 obtained across the secondary windings N2A and N2B are in timing similar to that in FIG. 24. Incidentally, the secondary winding voltage V2 shown in FIG. 27 has a polarity as viewed from a junction between the secondary winding N2A and the gate resistance Rg2. The secondary winding voltage V2 is of an opposite polarity as viewed from a junction between the secondary winding N2B and the gate resistance Rg1.
At the time of arrival of a period during which the secondary winding voltage V2 of the polarity shown in this figure is clamped at a predetermined level of negative polarity, the driving circuit for driving the MOS-FET Q4 operates to apply an on voltage at a level set by the gate resistance Rg2 and the resistance R12 to the gate of the MOS-FET Q4.
Similarly, at the time of arrival of a period during which the secondary winding voltage (V2) of the polarity opposite to that shown in this figure is clamped at a predetermined level of negative polarity, the driving circuit (the gate resistance Rg1 and the resistance R11) for driving the MOS-FET Q3 operates to apply an on voltage to the gate of the MOS-FET Q3.
Thus, rectified currents I1 and I2 of positive polarity flow through the MOS-FETs Q3 and Q4 in periods DON1 and DON2, respectively, as shown in the figure. The rectified currents I1 and I2 are 35 Ap as in the case of the circuit of FIG. 24 (the rectified currents I1 and I2 in the waveform chart of FIG. 25). However, the MOS-FETs Q3 and Q4 have a low on resistance, and thus a conduction loss of the rectified currents can be greatly reduced as compared with the rectifier diodes Do1 and Do2 formed by Schottky diodes. In addition, as is understood from the fact that the driving circuits are formed by only resistive elements, the winding voltage detection system has an advantage in that the driving circuit system is of a simple configuration.
However, in the condition of a heavy load corresponding to FIG. 27 (the load power Po=125 W), the secondary side rectified current in this power supply circuit is also in a discontinuous mode. This is indicated by discontinuity between the periods DON1 and DON2 in FIG. 27.
In the discontinuous mode, even when a current for charging the smoothing capacitor Co as the rectified currents I1 and I2 becomes a zero level, a current flows through the primary winding N1 of the isolated converter transformer PIT in the same direction. This indicates that in the foregoing waveform chart of FIG. 25, the sawtooth wave current component produced by the exciting inductance of the primary winding N1 flows as the primary side series resonance current Io in periods other than the periods DON1 and DON2 with the same polarity as in immediately preceding timing. Therefore, in actuality, the polarity of a voltage induced in the secondary windings N2A and N2B is not inverted, and in the meanwhile the MOS-FETs Q3 and Q4 maintain an on state without being completely turned off. Thus, a current in an opposite direction flows as the rectified currents I1 and I2 in the periods other than the periods DON1 and DON2 as shown in the figure. The rectified currents I1 and I2 in the opposite direction in the periods other than the periods DON1 and DON2 cause ineffective power. Since the rectified currents I1 and I2 in these periods have a relatively high level of 8 Ap, an amount of ineffective power is correspondingly large.
Thus, when the synchronous rectifier circuit employs the winding voltage detection system, while the conduction loss of the rectified currents is reduced, it is difficult to effectively improve total power conversion efficiency because of occurrence of ineffective power as described above.
A waveform chart of FIG. 28 shows operation of the power supply circuit employing the secondary side configuration shown in FIG. 26 under a condition of a light load.
As with the configuration of the power supply circuit shown in FIG. 24 as described above, the power supply circuit shown in FIG. 26 in practice performs constant-voltage control by controlling switching frequency. When the secondary side direct-current output voltage is increased under a condition of a light load, the power supply circuit operates to increase the switching frequency to decrease the secondary side direct-current output voltage, whereby the secondary side direct-current output voltage is stabilized.
In such a light-load condition, the secondary side winding voltage V2 is inverted in substantially the same timing as the voltage V1 across the switching device Q2 as shown in FIG. 28. Accordingly, the secondary side rectified currents I1 and I2 flow so as to continuously charge the smoothing capacitor Co without a period of discontinuity between periods DON1 and DON2. That is, a continuous mode is obtained. In this case, there are no periods when the rectified currents I1 and I2 in the opposite direction flow as in the operation at the heavy load as shown in FIG. 27, and thus correspondingly no ineffective power is produced.
Thus, the power supply circuit having the configuration obtained by replacing the secondary side rectifier circuit system with the synchronous rectifier circuit using the winding voltage detection system still has the problem of decrease in power conversion efficiency at the time of heavy load.
As techniques for solving the problem of occurrence of ineffective power caused by the rectified currents in the opposite direction as shown in FIG. 27, a synchronous rectifier circuit using a rectified current detection system is known. The rectified current detection system is a technique that turns off a MOS-FET before a rectified current for charging a smoothing capacitor Co becomes a zero level.
FIG. 29 shows an example of configuration of a synchronous rectifier circuit using this rectified current detection system. Incidentally, for simplicity of description, this figure shows a configuration for half-wave rectification.
In the rectified current detection system, a current transformer TR is provided to detect a current flowing through a secondary winding N2. A primary winding Na of the current transformer is connected to an end part of the secondary winding N2 and a drain of a MOS-FET Q4. A source of the MOS-FET Q4 is connected to a negative electrode terminal of a smoothing capacitor Co.
A secondary winding Nb of the current transformer is connected in parallel with a resistance Ra, and also connected in parallel with diodes Da and Db such that directions of forward voltages of the diodes Da and Db are opposite to each other, whereby a parallel connection circuit is formed. Further, the parallel connection circuit is connected with a comparator 20. A reference voltage Vref is inputted to an inverting input of the comparator 20. A junction between the reference voltage Vref and the inverting input of the comparator 20 is connected to an end part of a side where an anode of the diode Da and a cathode of the diode Db are connected in the parallel connection circuit. A non-inverting input of the comparator 20 is connected with an end part of a side where a cathode of the diode Da and an anode of the diode Db are connected in the parallel connection circuit.
In this case, an output of the comparator 20 is amplified by a buffer 21 and then applied to a gate of the MOS-FET Q4.
FIG. 30 shows operation of the circuit having the configuration shown in FIG. 29.
When a voltage induced in the secondary winding N2 becomes higher than a voltage (Eo) across the smoothing capacitor Co, a rectified current Id first starts to flow in a direction from an anode to a cathode of a body diode of the MOS-FET Q4 so as to charge the smoothing capacitor Co. Since the rectified current Id flows through the primary winding Na of the current transformer, a voltage Vnb corresponding to the rectified current Id flowing through the primary winding Na is induced in the secondary winding Nb of the current transformer. The comparator 20 compares the voltage Vnb with the reference voltage Vref. When the voltage Vnb exceeds the reference voltage Vref, the comparator 20 outputs an H level. This H level output is applied as an on voltage from the buffer 21 to the gate of the MOS-FET Q4 to turn on the MOS-FET Q4. Thus the rectified current Id flows from the drain to the source of the MOS-FET Q4. FIG. 30 shows the rectified current Id flowing with positive polarity.
Then, as the level of the rectified current Id is lowered with passage of time and correspondingly the voltage Vnb becomes lower than the reference voltage Vref, the comparator 20 inverts the output. The inverted output is output via the buffer 21 to discharge a gate capacitance of the MOS-FET Q4 and thereby turn off the MOS-FET Q4. Incidentally, the remaining rectified current Id at this point in time flows via a body diode DD4 within a short time.
With such an operation, the MOS-FET Q4 is turned off before the rectified current Id becomes a zero level. Thus, the flowing of the currents in the opposite direction through the MOS-FETs during the periods of discontinuity of the rectified currents as shown in FIG. 27 does not occur, so that ineffective power is not produced and power conversion efficiency is correspondingly increased.
For example, a measurement result obtained shows that DC-to-DC power conversion efficiency when the configuration on the secondary side of the power supply circuit shown in FIG. 24 is a synchronous rectifier circuit using the rectified current detection system for full-wave rectification which circuit is based on the configuration shown in FIG. 29 is improved to about 90% under the same condition as that of FIG. 25, FIG. 27 and the like.
As is understood from FIG. 29, however, the synchronous rectifier circuit of the above-described rectified current detection system requires, for one MOS-FET, at least one current transformer set and a relatively complex driving circuit system for driving the MOS-FET by an output of the current transformer. This complicates the circuit configuration, leading to disadvantages of decrease in manufacturing efficiency, increase in cost, increase in size of a circuit board, and the like.
In particular, when the synchronous rectifier circuit of the rectified current detection system is provided on the secondary side with the configuration of the primary side switching converter shown in FIG. 24 as a basis, a double-wave rectifier circuit needs to be formed on the secondary side. Therefore two current transformers and two driving circuit systems as described above are required for the MOS-FETs Q3 and Q4, thus aggravating the above problem.
Thus, the winding voltage detection system and the rectified current detection system are in a tradeoff relation in that the winding voltage detection system is disadvantageous in terms of power conversion efficiency because of ineffective power but enables a simple circuit configuration, whereas the rectified current detection system is advantageous in terms of power conversion efficiency because ineffective power is not produced but makes circuit configuration complex.


A method for the preparation of silicon dioxide dispersions is described, wherein the surface of the silicon dioxide is modified by a treatment with the reaction products of a compound of trivalent aluminum with least one aminoorganosilane. These reaction products are formed in a separate reaction step. The present invention also relates to recording sheets for ink jet printing having such a dispersion incorporated in an ink-receiving layer.

Ink jet printing processes are mainly of two types: continuous stream and drop-on-demand.
In continuous stream ink jet printing, a continuous ink stream is emitted under pressure through a nozzle. The stream breaks up into droplets at a certain distance from the nozzle. If a specific location on the recording sheet has to be printed the individual droplets are directed to the recording sheet, otherwise they are directed to a collecting vessel. This is done for example by charging unnecessary droplets in accordance with digital data signals and passing them through an static electric field which adjusts the trajectory of these droplets in order to direct them to the collecting vessel. The inverse procedure may also be used wherein uncharged droplets are collected in the vessel.
In the non-continuous process, or the so-called "drop-on-demand" process, a droplet is generated and expelled from the nozzle in accordance with digital data signals only if a specific location on the recording sheet has to be printed.
The printing speed of modern ink jet printers is ever increasing for economical reasons. Recording sheets suitable for these printers therefore need to absorb the inks very quickly. Especially suitable are recording sheets containing nanocrystalline, nanoporous inorganic compounds, preferably oxides such as aluminum oxides or silicon dioxide, or oxide/hydroxides such as aluminum oxide/hydroxides. Such recording sheets are known as "nanoporous" recording sheets.
Such recording sheets available today do not meet all of the required demands. In particular, in the case where dye-based inks are used for recording, the water fastness and the diffusion fastness of images printed on these recording sheets have to be improved. In the case where pigment-based inks are used for recording, the surface gloss of images printed on these recording sheets has to be improved, because very often the required photo quality is not obtained due to a bad compatibility between the ink-receiving layer and the ink. Particularly disturbing are gloss differences between different parts of the image. Furthermore, the manufacturing process of the known nanoporous recording sheets is not well mastered on an industrial scale.
Patent application DE 10,020,346 describes a recording sheet, which contains silicon dioxide prepared in the gas phase with a size of the primary particles of at most 20 nm, wherein the surface of the silicon dioxide has been modified by a treatment with aluminum chlorohydrate.
Patent application WO 00/20,221 describes the reaction of silicon dioxide prepared in the gas phase with aluminum chlorohydrate. The modified silicon dioxide is incorporated afterwards into an ink-receiving layer of nanoporous recording sheets for ink jet printing.
The described modification procedures show, however, the disadvantage that high quantities of aluminum chlorohydrate are needed in the modification step. Reaction speed is low and the resulting dispersions show bad storage stability due to their high salt content. The dispersions may be used only at a low value of pH, because they gel at higher values of pH. High quantities of the hardener boric acid are needed for hardening these recording sheets.
Patent application WO 02/094,573 describes the use of silicon dioxide prepared in the gas phase in recording sheets for ink jet printing, wherein the surface of the silicon dioxide has been modified by a treatment with aminoorganosilanes.
Patent application WO 01/05,599 describes the use of silicon dioxide pigments in recording sheets for ink jet printing, wherein the surface of the silicon dioxide has been modified by a treatment with cationic aminoorganosiloxanes.
Patent application EP 0,983,867 describes the use of colloidal silicon dioxide in recording sheets for ink jet printing, wherein the surface of the silicon dioxide has been modified by a treatment with silanes of general formula (R1)nSi(OR2)4-n, wherein at least one of the substituents R1 contains an amino group.
The described modification procedures show the advantage that only relatively small quantities of aminoorganosilanes are needed in the modification step. The modification reaction occurs in a relatively broad range of values of pH. Only small quantities of the hardener boric acid are needed for hardening these recording sheets due to their higher value of pH.
The described modification procedures show, however, the disadvantage that the aminoorganosilanes are expensive and that the modified dispersions fix negatively charged species only in a relatively narrow range of values of pH.


Briefly, in accordance with one embodiment of the invention, a circuit includes: at least one differential amplifier. The differential amplifier is coupled in a circuit configuration so that the differential output voltage signal of the differential amplifier circuit includes a scalable second-order harmonic component of the differential input voltage signal applied to the differential amplifier circuit. Briefly, in accordance with another embodiment of the invention, a method of applying a differential input voltage signal to a differential amplifier circuit to produce a differential output voltage signal includes the step of: driving the differential amplifier circuit so that the differential output voltage signal of the differential amplifier circuit includes a second-order harmonic component of the differential input voltage signal.

Amplifiers, and, particularly, operational amplifiers and differential amplifiers, are well-known in the art. Such amplifiers have a variety of applications, such as described, for example, in Design of MOS VLSI Circuits for Telecommunications, edited by Y. Tsividis and P. Antognetti, published by Prentice-Hall, Inc. (1985), and herein incorporated by reference. Typically, it is desirable to ensure that such differential amplifiers and operational amplifiers operate linearly so that the output voltage signal is approximately an amplified version of the applied input voltage signal. However, it is sometimes also desirable to produce harmonic components of an applied voltage signal.

Differential amplifier circuit for use in a read channel for a magnetic recording system


A method includes obtaining, by a management device, sub-interface information that includes a correspondence between an identifier of a first sub-interface of a first device and a first network slice identifier, a correspondence between an identifier of a second sub-interface of a second device and a second network slice identifier, and information indicating that the first sub-interface is directly connected to the second sub-interface, and determining, by the management device, a network slice topology based on the obtained sub-interface information.

As service requirements and application scenarios are increasingly diversified, to meet differentiated requirements of different services in a same network, a concept of network slicing is proposed in the industry. A network slicing technology provides a capability of implementing differentiated service-level agreement (SLA) assurance for a plurality of services when network resources are limited. The network slicing technology is applied to fields such as a 5th generation (5G) technology and an enterprise private line, to provide sliced pipe assurance based on SLA requirements of services. However, there is no technical solution for collecting and presenting a sub-interface-based network slice topology.

Method and system for determining network slice topology, and device


Systems, a ballistic device, and methods of use are provided herein. The system includes at least one first ballistic device having a first payload configured to detonate, and a second ballistic device configured to track at least one of a position and a movement of an object, and enable the at least one first ballistic device to identify at least one of the position and the movement of the object. The second ballistic device includes a second payload configured to detonate.

The present disclosure relates to swarm technology and, more particularly, to a projectile system and methods of use.
At least some known projectile systems include a plurality of projectiles (e.g., ballistic devices) that each includes at least one sensor, a boost package, and a payload. That is, in known ballistic defense projectile systems, each projectile has the same configuration such that one projectile includes every feature included in the other projectiles. Incorporating every feature into each projectile, however, is expensive and may be cost-prohibitive, particularly when the projectile includes a payload that is configured to detonate.


The invention concerns microcavity plasma devices and arrays with thin foil metal electrodes protected by metal oxide dielectric. Devices of the invention are amenable to mass production techniques, and may, for example, be fabricated by roll to roll processing. Exemplary devices of the invention are flexible. Embodiments of the invention provide for large arrays of microcavity plasma devices that can be made inexpensively. The structure of preferred embodiment microcavity plasma devices of the invention is based upon thin foils of metal that are available or can be produced in arbitrary lengths, such as on rolls. In a device of the invention, a pattern of microcavities is produced in a metal foil. Oxide is subsequently grown on the foil and within the microcavities (where plasma is to be produced) to protect the microcavity and electrically isolate the foil. A second metal foil is also encapsulated with oxide and is bonded to the first encapsulated foil. For preferred embodiment microcavity plasma device arrays of the invention, no particular alignment is necessary during bonding of the two encapsulated foils. A thin glass layer or vacuum packaging, for example, is able to seal the discharge medium into the array.

Microcavity plasma devices have several distinct advantages over conventional discharges. The small physical dimensions of microcavity plasma devices allows them to operate at pressures much higher than those accessible to conventional, macroscopic discharges. When the diameter of a cylindrical microcavity plasma device is, for example, on the order of 200-300 μm or less, the device will operate at pressures as high as atmospheric pressure and beyond. In contrast, standard fluorescent lamps, for example, operate at pressures typically less than 1% of atmospheric pressure. Microcavity plasma devices may be operated with different discharge media (gases or vapors or a mixture thereof) to offer output in the visible and nonvisible (ultraviolet, vacuum ultraviolet, and infrared, for example) wavelength ranges. Microcavity plasma devices are able to produce light more efficiently than other conventional discharge systems and do so on the microscopic scale.
Microcavity plasma devices developed over the past decade have been demonstrated to be well-suited for a wide variety of applications. An exemplary application for a microcavity plasma device array is to a display. Since the diameter of single cylindrical microcavity plasma devices, for example, is typically less than 400-500 μm, devices or groups of devices offer a spatial resolution that is desirable for a pixel in a display. In addition, the efficiency of a microcavity plasma device exceeds that of a conventional plasma display panel, such as those used in high definition televisions.
Early microcavity plasma devices exhibited short lifetimes because of sputtering that damaged metal electrodes used in the early, DC-driven devices. Polycrystalline silicon and tungsten electrodes extend lifetime but are higher cost materials and difficult to fabricate with present techniques.
Research by the present inventors and colleagues at the University of Illinois has pioneered and advanced the state of microdischarge devices. This development has led to practical devices including one or more important features and structures. For example, microcavity plasma devices can be operated continuously at gas pressures beyond one atmosphere at power loadings exceeding 100 kW/cm3. The ability to interface plasma in the gas or vapor phase with an e-h+ plasma in semiconductor devices has been demonstrated. MEMs and semiconductor processes have been applied to the fabrication of devices and arrays.
This research by present inventors and colleagues at the University of Illinois has resulted in exemplary practical devices. For example, semiconductor fabrication processes have produced exemplary densely-packed arrays of uniform microcavity plasma devices. An example array fabricated in silicon has demonstrated 250,000 discharge devices in a 25 cm2 active area. It has been demonstrated that the arrays can be used to excite phosphors in a manner analogous to plasma display panels, but at luminous efficacy levels that are not achievable with conventional plasma display panels. Another important device is a microcavity plasma photodetector that exhibits high sensitivity. Phase locking of microcavity plasma devices has also been demonstrated. Devices have been fabricated in ceramic material systems.
The following U.S. patents and patent applications describe microcavity plasma devices resulting from these research efforts. Published Applications: 20050148270—Microdischarge devices and arrays; 20040160162—Microdischarge devices and arrays; 20040100194—Microdischarge photodetectors; 20030132693—Microdischarge devices and arrays having tapered microcavities; U.S. Pat. No. 6,867,548—Microdischarge devices and arrays; U.S. Pat. No. 6,828,730—Microdischarge photodetectors; U.S. Pat. No. 6,815,891—Method and apparatus for exciting a microdischarge; U.S. Pat. No. 6,695,664—Microdischarge devices and arrays; U.S. Pat. No. 6,563,257—Multilayer ceramic microdischarge device; U.S. Pat. No. 6,541,915—High pressure arc lamp assisted start up device and method; U.S. Pat. No. 6,194,833—Microdischarge lamp and array; U.S. Pat. No. 6,139,384—Microdischarge lamp formation process; and U.S. Pat. No. 6,016,027—Microdischarge lamp.
U.S. Pat. No. 6,541,915 discloses arrays of microcavity plasma devices in which the individual devices are fabricated in an assembly that is machined from materials including ceramics. Metallic electrodes are exposed to the plasma medium which is generated within a microcavity and between the electrodes. U.S. Pat. No. 6,194,833 also discloses arrays of microcavity plasma devices, including arrays for which the substrate is ceramic and a silicon or metal film is formed on it. Electrodes formed at the tops and bottoms of cavities, as well as the silicon, ceramic or glass microcavities themselves, contact the plasma medium. U.S. Published Patent Application 2003/0230983 discloses microcavity plasma devices produced in low temperature ceramic structures. The stacked ceramic layers are arranged and micromachined so as to form cavities and intervening conductor layers excite the plasma medium. U.S. Published Patent Application 2002/0036461 discloses hollow cathode plasma devices in which electrodes contact the plasma/discharge medium.
Additional exemplary microcavity plasma devices are disclosed in U.S. Published Patent Application 2005/0269953, entitled "Phase Locked Microdischarge Array and AC, RF, or Pulse Excited Microdischarge"; U.S. Published Patent Application No. 2006/0038490, entitled "Microplasma Devices Excited by Interdigitated Electrodes;" U.S. patent application Ser. No. 10/958,174, filed on Oct. 4, 2004, entitled "Microdischarge Devices with Encapsulated Electrodes,"; U.S. patent application Ser. No. 10/958,175, filed on Oct. 4, 2004, entitled "Metal/Dielectric Multilayer Microdischarge Devices and Arrays"; and U.S. patent application Ser. No. 11/042,228, entitled "AC-Excited Microcavity Discharge Device and Method."

Arrays of microcavity plasma devices with dielectric encapsulated electrodes

The Broad of Trusstees of the University of Illinois


A method for avoiding singularities in the movement of a set of collinear CMGs in an array of CMGs in a spacecraft is provided. First, a command to adjust an orientation of the spacecraft is received. Then, the momentum needed from the set of collinear CMGs is determined. A dynamic control that determines a momentum path and avoids singularities by rotating the singularities out of the momentum path is used and a required gimbal movement for each of the CMGs in the set of collinear CMGs from the momentum path is calculated.

In order to control the attitude of a spacecraft, various rotating inertia memberscan be used. One such inertia member is a control moment gyroscope (CMG). A CMG typically comprises a flywheel with a fixed or variable spin rate mounted to a gimbal assembly. The spin axis of the CMG can be tilted by moving the CMG using the gimbal assembly. This motion produces a gyroscopic torque orthogonal to the spin axis and gimbal axis.
To achieve full attitude control of a spacecraft, a CMG array, including a minimum of three CMGs may be arranged such that each CMG in the CMG array imparts torque about a linearly independent axis. Typically, additional CMGs are provided for redundancy purposes and to assist with singularity avoidance. A singularity occurs when the momentum vectors of the CMGs line up such that one or more components of the requested torque can not be provided.
Several different techniques have been developed to avoid singularities. In one method it is first noted that a Jacobian A maps the CMG gimbal rates into a three dimensional array torque:Aω=τ  (1)where A is a 3×n Jacobian matrix, ω is a n×1 array of gimbal rates for the n gimbals, and τ is a 3×1 array of torque components to be imparted to the spacecraft. From the above equation and with a known torque command, τ, the individual gimbal rates for each CMG can be calculated. Using the known Moore-Penrose pseudoinverse to invert the Jacobian matrix, a set of possible gimbal rates is:ω=AT(AAT)−1τ.  (2)
As discussed previously, inherent in the use of CMGs is the possibility that the CMGs' momentum vectors may line up in such a way that a singularity condition is reached. Mathematically, singularities can occur when the eigenvalues of AAT approach zero, causing (AAT)−1 to approach infinity. Equivalently, singularities occur when the determinant of the matrix AAT is equal to zero (expressed algebraically as det (AAT)=0). In the case of a 3×n matrix A, this is equivalent to the rank of the matrix AAT being two or less.
Different approaches have been devised to avoid singularities in the movement of CMGs. In one approach, to ensure that (AAT)−1 never is zero, (AAT)−1 is replaced by (AAT+εI)−1 where I is the identity matrix and ε is a small number. The use of a positive ε ensures that det (AAT+εI)−1 never becomes 0.
While useful in some instances, a drawback to this approach is that this approach changes the gimbal rate calculation. In the case of the Jacobian A, the use of the pseudoinverse means that gimbal rates are no longer exactly mapped into the commanded torques because of the error ε introduces. This resulting error steers the spacecraft in the wrong direction and can introduce significant, undesired torque, especially near the singularity.
A second approach is to limit the CMG array's momentum output to a smaller area within a momentum envelope. The momentum envelope is the momentum provided in all possible combinations of the CMGs in the CMG array. In one exemplary embodiment, depending on the CMG arrangement, by operating within one-third or less of the total momentum envelopes, singularities can be avoided. However, this approach wastes potential torque and results in systems that are much larger and heavier than needed.
In view of the foregoing, it is desirable to provide a method for determining a singularity free momentum path that addresses one or more of the foregoing deficiencies or other deficiencies not implicitly or expressly described. It is also desirable to provide a system for determining a singularity free momentum path that addresses one or more of the foregoing deficiencies or other deficiencies not implicitly or expressly described. Furthermore, other desirable factors and characteristics of the present invention will become apparent from the subsequent detailed description and the appended claims, taken in conjunction with the accompanying drawings and the foregoing technical field and background.

Method and system for determining a singularity free momentum path


An application running on a UE receives a command through a user interface or from a server to initiate the locating of a tracking device. The tracking device determines one or more hot spots it is close to. A server determines whether to instruct the tracking device to use its GNSS circuitry to determine its location based on a comparison of location information corresponding to the hot spot(s) and user selectable accuracy criteria. The UE may detect a beacon signal and determine an initial value corresponding to an initial signal strength of the beacon. The beacon may only broadcast for a predetermined period based on location scenario. The application produces a default initial indication, based on the initial signal strength and uses updated beacon signal strength values to adjust the indication relative to the default indication based on the updated signal strength value relative to the initial signal strength value.

Wireless communications devices serve many purposes in today's connected world. In addition to providing communication between people using such devices, wireless communications devices may use sensors to provide information regarding the environment surrounding the device to a user. Such information may be provided wirelessly to another wireless communication devices, such as a user equipment device ("UE"). Examples of user equipment devices include smart phones, tablets, smart watches, and the like. Information that may be provided to a UE includes the location of the device that is providing the information. A device may use myriad means and methods for determining its location, such as GNSS and triangulation between cellular towers or triangulation between Wi-Fi hot spots.
However, although the resolution of location information determined using GNSS or triangulation techniques may be acceptable for use in locating a large object such as a vehicle or an item that is being shipped in a vehicle, such resolution may not be good enough to locate a small item being tracked inside of a large vehicle like a truck, a ship, or a plane. GNSS resolution may also not be adequate for locating a small item being tracked, such as a key ring or other small item of personality that often gets lost (or hides in plain sight) in someone's home, office or yard among clutter, furniture, or vegetation.
GNSS and Cellular Radios require relatively a relatively large amount of battery energy to operate and after most battery power is exhausted, the battery may not have enough energy to power a cellular radio or GNSS for locating a small item and subsequently the item being tracked may be lost forever.

Method and system for determining the location of a tracking device


An infrared-reflective film includes a substrate film composed of a polyolefin film or a polycycloolefin film. The substrate film has two main surfaces and an infrared-reflective layer is formed on one main surface and the other main surface faces air, nitrogen gas, inert gas or a vacuum. A surface of the infrared-reflective layer faces either of air, nitrogen gas, inert gas or a vacuum.

Field of the Invention
The present invention relates to an infrared-reflective film (heat reflective film).
Description of the Related Art
An infrared-reflective film is adhered to a building window, a vehicle window so as to be used to improve cooling or heating effect. Further, an infrared-reflective film is adhered to a window of a refrigerated (freezing) counter display to be also used to improve cold reserving effect.
FIG. 5 is a cross-sectional view of a conventional infrared-reflective film 70. In the conventional infrared-reflective film 70, an infrared-reflective layer 72 is formed on one surface of a substrate film 71 (transparent polymer film). The substrate film 71 is to be used as a base of lamination and a polyethylene terephthalate film is preferably used as the substrate film 71.
The infrared-reflective layer 72 is a laminated layer in which a metal thin layer is interposed between transparent dielectric layers respectively having a high reflective index. The infrared-reflective layer 72 transmits visible light, however, reflects infrared-reflective beams. The infrared-reflective layer 72 is formed on the substrate film 71 by a sputtering method or the like.
Far-infrared rays included in irradiation light 73 from above the infrared-reflective layer 72 are reflected by the infrared-reflective layer 72. Far-infrared rays included in irradiation light 74 from below the substrate film 71 are, however, mostly absorbed in the substrate film 71 as mentioned below.
Since polyethylene terephthalate contains plenty of C═O groups, C—O groups, and aromatic groups, the polyethylene terephthalate exhibits vibration absorption of a far-infrared region of 5 μm to 25 μm. Accordingly, polyethylene terephthalate has a property of absorbing far-infrared rays.
In the infrared-reflective film 70 shown in FIG. 5, a polyethylene terephthalate film is used as the substrate film 71. Accordingly, the substrate film 71 absorbs a part of far-infrared rays included in the irradiation light 74 from below the substrate film 71 to increase the temperature.
The temperature further increases because the substrate film 71 absorbs a part of far-infrared rays included in reflective light emitted to the lower side of the infrared-reflective layer 72. As a result, the irradiation light 74 from below the substrate film 71 is mostly absorbed in the substrate film 71. The substrate film 71 itself thereby re-emits infrared rays.
While the infrared-reflective film 70 shown in FIG. 5 reflects far-infrared rays when the irradiation light 73 comes from an infrared-reflective layer 72 side (upper side), the infrared-reflective film 70 does not reflect far-infrared rays when the irradiation light 74 comes from a substrate film 71 side (lower side). Accordingly, the infrared-reflective film 70 shown in FIG. 5 does not have enough infrared-reflective properties.
FIG. 6 is a cross-sectional view of another conventional infrared-reflective film 80 (Japanese Unexamined Patent Application Publication No. 2011-104887 A). In the infrared-reflective film 80 shown in FIG. 6, an infrared-reflective layer 82 and a protective layer 83 are formed on one surface of a substrate film 81. The substrate film 81 is a film to be a base of lamination and a polyethylene terephthalate film is preferably used as the substrate film 81.
The infrared-reflective layer 82 is a laminated layer in which a metal thin layer is interposed between transparent dielectric layers with a high refractive index. The infrared-reflective layer 82 transmits visible light. However, the infrared-reflective layer 82 reflects infrared-reflective rays. The infrared-reflective layer 82 is formed on the substrate film 81 by the sputtering method or the like.
In the infrared-reflective film 80 shown in FIG. 6, a polycycloolefin layer is used as a protective layer 83. Since the basic chemical constitution of polycycloolefin consists of carbon atom and hydrogen atom, polycycloolefin exhibits a little absorption of a far-infrared region. Accordingly, far-infrared rays included in irradiation light 84 from above the protection layer 83 reaches the infrared-reflective layer 82 without mostly being absorbed in the protective layer 83 to be reflected at the infrared-reflective layer 82. Far-infrared rays included in reflective light 85 reflected by the infrared-reflective layer 82 are also scarcely absorbed in the protective film 83 and are emitted outward. Accordingly, there is almost no increase in the temperature of the protective layer 83.
However, as mentioned below, far-infrared rays included in irradiation light 86 from below the substrate film 81 are mostly absorbed in the substrate film 81.
In the infrared-reflective film 80 shown in FIG. 6, a polyethylene terephthalate film is used as the substrate film 81. Accordingly, the temperature of the substrate film 81 rises by absorbing a portion of far-infrared rays included in irradiation light from below the substrate film 81. The temperature of the substrate film 81 further rises by absorbing a portion of far-infrared rays included in reflective light emitted to the lower side of the infrared-reflective layer 82. As a result, irradiation light 86 from below the substrate film 81 is mostly absorbed in the substrate film 81. This substrate film 81 itself thereby re-emits infrared rays.
While the infrared-reflective film 80 shown in FIG. 6 reflects far-infrared rays when the irradiation light 84 comes from a protective film layer 83 side (upper side), the infrared reflective film 80 does not reflect far-infrared rays when the irradiation light 86 comes from a substrate film 81 side (lower side). Consequently, the infrared-reflective film 80 shown in FIG. 6 does not have sufficient infrared-reflective properties.
FIG. 7 is a cross-sectional view showing still another conventional infrared-reflective film 90 (an infrared rays cut filter) (Japanese Unexamined Patent Application Publication No. 2006-30944 A). The infrared-reflective film 90 shown in FIG. 7 is a film in which infrared-reflective layers 92, 93 are formed on both surfaces of a substrate film 91 (a transparent resin film). The substrate film 91 is a film to be a base of lamination and a norbornene resin film or a polyether sulfonic resin film is used as the substrate film 91.
The infrared-reflective layers 92, 93 are both dielectric multi-layers, in which a dielectric layer A and a dielectric layer B with a refractive index higher than the dielectric layer A are alternately formed. Although the infrared-reflective layers 92, 93 allow visible light to transmit, the infrared-reflective layers 92, 93 reflect far-infrared rays. The infrared-reflective layers 92, 93 are formed on the substrate film 91 by the deposition evaporation method.
The infrared-reflective film 90 shown in FIG. 7 has the infrared-reflective layers 92, 93 on both surfaces thereof. Accordingly, regardless of a material of the substrate film 91, the infrared-reflective film 90 similarly reflects both far-infrared rays included in the irradiation light 94 emitted from the upper side and far-infrared rays included in the irradiation light 95 emitted from the lower side. Consequently, the infrared-reflective film 90 shown in FIG. 7 has superior infrared rays reflective properties.
However, it costs expensive to produce an infrared-reflective film 90 shown in FIG. 7 because the infrared-reflective layers 92, 93 have to be formed on both surfaces thereof.
Since the infrared-reflective film 90 shown in FIG. 7 has the infrared-reflective layers 92, 93 on both surfaces thereof, the infrared-reflective film 90 has a low transmittance of visible light. Accordingly, the room gets dark when the infrared-reflective film 90 is used for a window of a building. Further, when the infrared-reflective film 90 is adhered to a refrigerated (freezing) counter display, it becomes hard to see inside the refrigerated (freezing) counter display. Accordingly, the infrared-reflective film 90 shown in FIG. 7 has a drawback in practicability.


A golf training system includes at least one target/connecting member having a top wall defining a central opening and a perimeter wall defining a plurality of openings spaced apart from each other about the circumference of the perimeter wall. The target/connecting member is configured to be used as a target by fastening the target/connecting member to the ground using a golf tee or to other surfaces using an integrated anchoring system. Alignment rods are configured to be inserted through the openings in the perimeter wall. Target/connecting members can attach multiple alignment rods together in predetermined configurations for use as golf training aids.

Golf requires the ability to swing a variety of different types of golf clubs in a variety of different ways in an effort to produce a desired and consistent result when hitting a golf ball. Each facet of the game of golf, e.g., driving, pitching, chipping and putting, has different aspects, such as stance, grip, ball position, swing tempo, head position, etc., and may require the use of a different type of club. As a result, each facet of the game of golf utilizes vastly different techniques and swing mechanics.
Golf training aids have been developed to help with virtually every aspect of each facet of a golfer's game. However, most training aids are directed to a very specific aspect of one facet of a golfer's game. In addition, different training aids may be designed to help with a particular aspect in different ways. Therefore, a golfer looking to improve in all or a few aspects of a particular facet of golf, such as driving, typically needs to purchase multiple different training aids. A golfer looking to improve in more than one facet of golf would be faced with the prospect of purchasing even more training aids.
A golfer therefore can end up with a large supply of different types and kinds of training aids. A large amount of space may therefore be required just for storing a golfer's training aids. In addition, multiple training aids would typically have to be transported at the same time by a golfer or golf instructor to and from a practice area which can be cumbersome, especially when considering the size and/or complexity of some training aids that are available on the market.


Two bushings are installed concentrically in the two ends of a bore with an interference fit using an elongated mandrel which is maintained coaxially on the bore centerline by a guide inserted in one end of the bore while the bushing is pressed into the other end of the bore. The force to press the bushings into the bore is exerted by a pulling tool to which the mandrel is coupled, and that force is transmitted through the mandrel to the bushing by way of a collar attached to the mandrel. The force of the pulling tool exerted on the mandrel is reacted through an anvil attached to the barrel of the pulling tool and effectively in contact with the side of the part opposite to which the bushing is being pressed into the bore. The faces of the collar and the anvil are machined flat and perpendicular to the axis of the mandrel so that the squeezing force exerted between them on the bushing and the part is parallel to the axis of the bore. This ensures that the bushing is pressed straight into the bore and is held in a concentric orientation during the installation so there is no gouging of the walls of the bore and the bushing is installed exactly concentric with the bore and coaxial to the other bushing.

This invention relates to installation of bushings in a bore, and more particularly to a method and apparatus for installing two spaced bushings concentrically in the two ends of a bore, coaxially with each other.
In the aerospace and other industries, there has long been a requirement for the mounting of bushings in bores in parts, usually to support a shaft which is mounted in the part for supporting or connection to some other part. An example is a flap assembly which requires the installation of pintel pins. The pintel pins must be installed in the bushings in a clevis flap assembly with a very close tolerance sliding fit, and if the bushings are not concentric in the bore and coaxial with each other, the installation of the pins will be difficult or impossible and require extensive and expensive rework on the part in order to be able to install the pintel pins.
The conventional method of installing bushings in such parts is with the use of a press. The part is positioned on the press platen and spacers are positioned to support the part against deflection when the ram of the press is actuated to press the bushing into the bore. A bushing is positioned at the end of the bore and the part is aligned under the ram of the press, then the press is operated to drive the ram down and press the bushing into the bore.
This system works best when jigs and fixtures are available for positioning the part in the press and aligning the bore and bushing with the press ram. However, in many manufacturing plants where bushings are used, the cost of manufacturing jigs and fixtures for each of the numerous parts that have bushings installed becomes very expensive, especially since each bushing in the part must have a different installation fixture and every revision of the part would usually require a redesign of the fixture. More importantly however, even with the use of jigs and fixtures, the conventional process is imperfect because of the impossibility of maintaining alignment of the bushings with the bore and each other while exerting a pressing force on the bushing during installation which is exactly parallel to the axis of the bore.
When the axis of the bushing is misaligned with the axis of the bore or the direction of force exerted on the bushing during installation, or if the pressing surfaces are nonperpendicular to the axis of the bore, the bushing will not be pressed straight into the bore, but will be cocked slightly and gouge the walls of the bore as it is pressed into the bore, and will be pressed into a slightly skewed position in the bore so that it is nonconcentric with the bore and not coaxial with the other bushing inserted at the other end of the bore. This nonconcentricity and/or noncoaxial orientation prevents or makes difficult the proper installation of the pin, shaft, or bolt through the two bushings, requiring removal of the bushings, reboring of the bore in the part, and reinstallation of oversized bushings in the new bore. This rework adds considerably to the cost of the part and slows the production speed of the manufacturing operation.
One of the principle reasons for using bushings in a bore is that they can be replaced when the bushing becomes worn. This makes it possible to replace only the portion of the part which is worn instead of the entire part, which could be very expensive. However, the replacement of bushings is considerably more difficult in the field than it is in the factory. In the field, the bushings are forced into the bore with whatever tools are at hand, including hammers making correct alignment of the bushings in the bore and with each other unusual and primarily a matter of good luck. Another technique in wide use is to insert a bolt through the bushings and the bore, and tighten a nut on the bolt to force the bushings into the bore. This is preferable to pounding the bushings in with a hammer, but is slow and does not ensure either concentricity or coaxial alignment because there is no guidance for the bushings and nothing to keep the bolt coaxially aligned in the bore.
Various other schemes have been devised to solve this problem, but none of these schemes provides the desirable attributes of low cost, simplicity, speed of operation and flexibility of use with numerous different parts. Thus, there has been a long unfulfilled need in the industry for a simple, reliable, inexpensive and flexible process and apparatus for installing bushings concentrically in a bore in which the bushings are held in exact coaxially alignment with each another during installation so that the installed bushings are properly oriented in the bore and to each other.

Apparatus for installing bushings concentrically in a bore


A polymer dispersed liquid crystal (PDLC) display element for use in an electronic apparatus wherein the display element operates in a PDLC mode or reverse PDLC mode providing for reduction of visual display haze, improved display contrast. The particular medium includes a liquid crystal portion and a polymer portion and may include a chiral component. In cases where visual display haze and contract is a particular problem in the utility of the reverse PDLC display element, such as in the case where the display element is combined with a solar battery, a sufficient amount of dichroic dye is added to the liquid crystal portion of the medium, and, further, a compound showing no fluorescence emission with respect to a liquid crystal portion, or a polymer or a chiral polymer precursor portion of the medium, is employed. The resulting electronic apparatus can, therefore, incorporate a reverse PDLC type display element in combination with a solar battery or other component that affects display quality in such elements to improve the overall visual quality and contrast in the display rendering the display more useful in more miniaturized applications.

1. Related Application
This application is a continuation of International Application No. PCT/JP94/00505, filed Mar. 29, 1994.
2. Technical Field
This invention relates to reflective type, bright display elements useful as a display in wrist watches, meter display panels of vehicles, information display terminals for electronic memorandum books or computer notebooks, computers and televisions, or information bulletin boards, and to the construction and configuration of electronic apparatus employing such display elements as provided in a visual display, such as a LCD panel.

Polymer dispersed liquid crystal (PDLC) display element for use in an electronic apparatus


A ceramic fiber band that can fulfill a plurality of functions in connection with the injection and/or extrusion process of an injection molding or extrusion apparatus (i.e. heat, cool, and insulate), thereby reducing overall energy costs and creating a safer working environment. The ceramic band, being an insulator and having heating and/or cooling elements with a faster response contained therein, exert the desired heating/cooling effect upon the barrel portion of the injection/extrusion machine thereby raising, lowering, or maintaining the temperature of the fluid plastic compound passing through the barrel.

This application is not referenced in any microfiche appendix.
1. Field of the Invention
The present invention relates generally to vacuum cast ceramic fiber insulators, specifically those having the ability to raise, lower or maintain temperature by utilizing heating and/or cooling controls in a low temperature apparatus.
2. Prior Art
The use of injection/extrusion machines to create plastic parts is well known. Plastic compounds are subjected to heat in order to render it into a fluid state. After it reaches a fluid state, the compound is then forced under pressure into mold cavities or through dies. It is well known in the art that the fluid resin compounds are directed into the mold cavities or dies by way of injection or extrusion barrels. In order to insure the optimum flow characteristics of the compound, the barrels are equipped with heating mechanisms in order to control the temperature of the compound. Multiple types of heating bands have been disclosed in the prior art. The most common is the heating coil that surrounds the barrel and conducts heat to the compound as it flows through the barrel.
The optimization of the flow characteristics of the fluid compound is dependent upon the ability to control the amount of heat applied to the barrel as the fluid compound passes through it. Therefore, an improved heating and cooling mechanism is needed to improve the efficiency in the attempt to achieve the optimal flow characteristics for the fluid compound.
The present invention is directed to a ceramic insulating and heating/cooling band. The prior art does not demonstrate the use of such bands with low temperature apparatus, in the range of less than 1000xc2x0 C./1600xc2x0 F. Thus, there is a need for a temperature control mechanism for use in low temperature apparatus, such as injection/extrusion machines.
The present invention is generally directed toward a device for effectively insulating while heating and/or cooling a fluid plastic compound more efficiently. Specifically, the present invention is directed toward a device having the ability to maximize the controllability of the temperature of a fluid plastic compound in connection with the injection/extrusion process of low temperature apparatus, such injection molding and extrusion machines
A discussion concerning specific terms used herein is in order. The term xe2x80x9cceramic bandxe2x80x9d is synonymously referred to as a vacuum cast ceramic fiber heater band, machinable ceramic fiber board or other materials of the ceramic genre. While extrusion machines can be different from injection molding machines, the present invention can be utilized in connection with both types of machines. Therefore, reference to one type of machine should be viewed as reference to both types of machines. The use of the terms xe2x80x9clow temperaturexe2x80x9d and xe2x80x9clow temperature apparatusxe2x80x9d herein refers to those apparatus which cannot obtain temperatures of 1000xc2x0 C./1600xc2x0 F. and higher.
The present invention utilizes a ceramic band which surrounds the barrel of the injection/extrusion machine. The ceramic band acts as an insulator for the heat contained on and within the barrel. Further, the ceramic band incorporates heating elements and cooling mechanisms to provide the ability to accurately and quickly change the temperature of the barrel.
The heating/cooling mechanisms can be, but not necessarily, cast within the ceramic band providing an efficient temperature control unit.
The ceramic band can take a plurality of different forms, including but not limited to a half-section configuration or individual block configuration. In any configuration, the band fits around the exterior portion of the barrel and provides a heating and/or cooling effect to such barrel, thereby heating or cooling the fluid plastic compound flowing through such barrel.
With respect to the heating mechanisms, standard heating elements are cast within the ceramic band. With respect to the cooling mechanisms, air or fluid can be utilized to decrease the temperature of the barrel. Suitable apparatus for providing air or fluid cooling ability is provided through a manifold apparatus either in the interior or on the exterior of the ceramic band.
In addition, monitoring devices such as LED lights, can be utilized to monitor the present invention and advise the operator its current operational condition.
The primary objective of the present invention is to provide a self-insulated heating and cooling unit used in connection with low temperature plastics injection/extrusion machines.
Another object of the present invention is to provide a device which insulates and provides improved temperature controllability for the barrel portion of injection and extrusion machines.
Another object of the present invention is to provide a faster responding device utilizing insulation, heating elements and air and/or water cooling devices to alter the controlled temperature of the barrel portion of injection and extrusion machines.
Other objects and further scope of the applicability of the present invention will become apparent from the detailed description to follow, taken in conjunction with the accompanying drawings wherein like parts are designated by like reference numerals.

Vacuum cast ceramic fiber insulated band having heating and cooling elements


One gate driver includes an output node configured to be coupled to a gate line and to provide power to the gate line for driving thin-film transistor (TFT) gates of a display. An input node of the gate driver is configured to receive an input signal. The gate driver includes a first field-effect transistor (FET) having a gate, a drain, and a source. The drain may be coupled to the input node and the source may be coupled to the output node. The gate driver also includes a second FET having a gate, a drain, and a source. The drain may be coupled to the input node. The gate driver includes a capacitor having a first end coupled to the gates of the FETs and a second end coupled to the source of the second FET. Using the gate driver power consumption of the display may be reduced.

The present disclosure relates generally to electronic displays and, more particularly, to reducing power consumption and size of gate drivers of a display.
This section is intended to introduce the reader to various aspects of art that may be related to various aspects of the present disclosure, which are described and/or claimed below. This discussion is believed to be helpful in providing the reader with background information to facilitate a better understanding of the various aspects of the present disclosure. Accordingly, it should be understood that these statements are to be read in this light, and not as admissions of prior art.
Electronic displays, such as liquid crystal displays (LCDs) or organic light emitting diode (OLED) displays, are commonly used in electronic devices such as televisions, computers, and handheld devices (e.g., cellular telephones, audio and video players, gaming systems, and so forth). Such display devices typically provide a flat display in a relatively thin package that is suitable for use in a variety of electronic goods. In addition, such display devices typically use less power than comparable display technologies, making them suitable for use in battery-powered devices or in other contexts where it is desirable to minimize power usage.
LCDs typically include an LCD panel having, among other things, a liquid crystal layer and various circuitry for controlling orientation of liquid crystals within the layer to modulate an amount of light passing through the LCD panel and thereby render images on the panel. The LCD panel may include gate driver circuitry for driving gates of thin-film transistors (TFTs) of the LCD panel. Specifically, the gate driver circuitry may select among rows of TFTs to activate to enable data to be provided to a selected row of TFTs. OLED displays may also include gate driver circuitry for selecting among rows of TFTs to activate to enable data to be provided to a selected row of TFTs. Unfortunately, the gate driver circuitry may use a high power supply voltage thereby consuming a substantial amount power. Furthermore, the gate driver circuitry may occupy a large amount of space. Accordingly, there is a need for low power techniques to improve reliability and to decrease the amount of power consumed and space used by the gate driver circuitry, and thereby decreasing the amount of power consumed and space used by an electronic display.

Devices and methods for reducing power consumption and size of gate drivers


A mechanical power breaker for a vehicle has a casing, a power input, a master power output, and a breaker positioned in the casing and electrically connected between the power input and master power output. The breaker has a seat, a conductive connector, a moving element, and a first and second resilient supports. The conductive connector is held in seat by the first resilient support and formed by multiple conductive blocks stacked together. The conductive connector and first resilient support are electrically connected between the power input and the master power output. The hollow moving element is held in the seat by the second resilient supports to surround the conductive connector. If the casing is shaken, the moving element is moved to hit the conductive connector. The conductive connector is separated into the multiple conductive blocks to disconnect the electronic connection between the power input and master power output.

1. Field of the Invention
The present invention relates to a power breaker, and more particularly to a mechanical type power breaker for use in a vehicle.
2. Description of Related Art
How to ensure the vehicle safety in driving situation is an important issue for most of the vehicle manufacturing factories. The vehicle manufacturing factories are dedicated to improve two important aspects. One aspect is mechanical safety structure and the other is a vehicle circuit protection system.
The mechanical safety structure is assembled to the inside of the vehicle to prevent the driver from being hurt when a collision suddenly occurs. The vehicle circuit protection system prevents the vehicle circuit from being short that may cause a fire when the vehicle is crashed and the gasoline leaks from a fuel tank of the vehicle. However the vehicle circuit protection system is not common.
Therefore, the present invention provides a mechanical power breaker for a vehicle to prevent the vehicle circuit from a short circuit when the vehicle is crashed.


A local network, particularly an Ethernet network having redundancy properties, in which coupling devices (K1, K2, K3) and a redundancy manager (RM) are interconnected in a ring-shaped topology. The redundancy manager opens the ring to create a linear topology if test messages (T11, T12) emitted by the redundancy manager (RM) are received at the other port (P42, P41) within a given period of time, otherwise the redundancy manager (RM) closes the connection. As a so-called redundancy manager observer, at least one of the coupling devices (K3) is configured such that it evaluates messages received at the port (P32) thereof, the port (P32) being connected to the redundancy manager, opens the ring to create a linear topology, and signals an error if more than initial test messages (T11) have been received at this port (P32) and if the status “ring opened” has been indicated by the redundancy manager (RM) in the last-received initial test message (T11), whereby messages are prevented from circulating on the ring.

The present invention is related to a local network, particularly an Ethernet network, having redundancy properties and a coupling device for such a network.
A local Ethernet network with redundancy properties is known from WO 99/46908. In the network described therein, coupling devices and a redundancy manager, each having at least two ports and being configured as layer 2 components, are interconnected in a ring-shaped topology by connecting two ports of adjacent devices. The term "layer 2 components" means that at least the layers 1 and 2 of the ISO OSI 7-layer model are implemented in the devices. The devices thus perform an address evaluation and message routing. In normal operation, i.e., if no errors have occurred in the network, the redundancy manager opens the ring. This means that messages received by the redundancy manager at its one port located in the ring are not forwarded via its other port located in the ring, but are blocked. This status is symbolized by an open switch. Thus, logically, the network has a linear topology in which the two line ends of the network are connected to a redundancy manager, which in the error-free case separates the two line ends from one another. In the event of an error, i.e., if an interruption occurs at some point in the network outside the redundancy manager, the redundancy manager connects the two line ends to again form a linear topology.
Physically, however, this is a ring-shaped network topology in which the devices form a ring by connecting two ports of adjacent devices. The redundancy manager ensures that at any given time the ring is interrupted only at a single location. To check whether there is already an interruption of the ring outside the redundancy manager, i.e., to check whether the network is operating error free, the redundancy manager sends test messages into the ring at defined time intervals via the two ports with which it is connected to the ring, and it opens the ring to form a linear topology with respect to the transmission logic if the test messages are received at its other port within an additional predefined time interval. Otherwise, i.e., if at least one test message has been received at the other port within the additional time interval, the redundancy manager closes the connection, such that the network has again a logically linear topology consisting of a continuous line. With respect to further embodiments and advantages of such a local network and a redundancy manager, reference is made to the above-cited publication WO 99/46908.
Networks in which messages are distributed from node to node and in which the response to the failure of a communication link between two nodes is to add a redundant communication path, have the risk that messages are copied or circulate within the network. Such errors can be triggered when a redundant communication path is mistakenly added. Thus, as a result of a malfunction of the conventional redundancy manager, the two line ends could inadvertently be connected and a logically ring-shaped network topology could be created. As a result, messages can circulate within the ring. In the worst case, the maximum bandwidth of the network available for the data traffic is occupied, such that further payload traffic among the users connected to the network can no longer be transmitted. When such an error occurs, the network is not available, or at least not fully available, for the connected users.

Local network, particularly Ethernet network having redundancy properties, and coupling device for such a network


A medical balloon for deployment of stents is disclosed. The medical balloon includes a central section; a proximal and a distal cone section; a proximal and a distal shaft section; a proximal sleeve which is arranged coaxial to and radially inside the proximal shaft section. The proximal end of the proximal shaft section is permanently connected and/or unitarily formed with the proximal end of the proximal sleeve, and distal thereto, at least part of the proximal cone section is connected with the proximal sleeve by an adhesive bond, and a distal sleeve which is arranged coaxial to and radially inside the distal shaft section. Methods of manufacturing a medical balloon are disclosed.

In medicine, a natural conduit in a body may be locally flow constricted or damaged. A stent or stent graft may be inserted into this natural conduit, in order to counteract such a flow constriction or to repair damage. Stents may be deployed by means of a balloon catheter. The stent is crimped around the balloon before deployment, the catheter is introduced into a body lumen and once the balloon is positioned appropriately, the balloon is inflated and expands the stent.
FIGS. 1a to 1d illustrate a few exemplary chronologic steps of inflating a balloon from the state of the art. The balloon shown in these Figures is used for deploying a stent which is crimped around the balloon. FIG. 1a shows the unexpanded state of the balloon and stent. As shown in FIG. 1b, when starting to introduce pressure into the balloon, its expansion begins at the outer ends. This phenomena is called dog boning and is caused by the resistance for expanding this part being smaller than in the center of the balloon/stent assembly (the struts in the middle are held in position from both sides, whereas at the end struts are only held from one side). From there, the expansion of the balloon/stent proceeds towards a center of the balloon/stent until the entire balloon is expanded, as shown in FIG. 1d. The dog-boning of the balloon cones has a so-called foreshortening effect for the stent, which means that the stent is shortened longitudinally to a small extend. This foreshortening effect is undesirable, because it makes the expanded length deviate from the unexpanded length.
FIG. 2 schematically shows a typical diagram of the relationship between the pressure inside a balloon and its diameter. As illustrated, the pressure inside the balloon increases up to a nominal pressure 1, at which the balloon is considered to be fully expanded. If the pressure is increased further, the balloon diameter does not increase substantially any more. When exceeding a burst pressure 2, the balloon may burst.

ABBOTT LABORATORIES VASCULAR ENTERPRISES LIMITED