A monolithic electronic device includes a substrate, a semi-insulating, piezoelectric Group III-nitride epitaxial layer formed on the substrate, a pair of input and output interdigital transducers forming a surface acoustic wave device on the epitaxial layer and at least one electronic device (such as a HEMT, MESFET, JFET, MOSFET, photodiode, LED or the like) formed on the substrate. Isolation means are disclosed to electrically and acoustically isolate the electronic device from the SAW device and vice versa. In some embodiments, a trench is formed between the SAW device and the electronic device. Ion implantation is also disclosed to form a semi-insulating Group III-nitride epitaxial layer on which the SAW device may be fabricated. Absorbing and/or reflecting elements adjacent the interdigital transducers reduce unwanted reflections that may interfere with the operation of the SAW device.

The present invention relates to nitride-based acoustic wave devices. Acoustic wave devices form a class of electronic devices that process signals that exist as acoustic (i.e. sound or compression) waves traveling in piezoelectric crystals. Piezoelectric crystals are characterized by the fact that when the material is mechanically stressed (i.e. compressed or placed under tension), an associated electric field is induced. Likewise, when an electric field is applied to a piezoelectric crystal, the material becomes mechanically stressed in a predetermined manner. It is possible to exploit these characteristics to perform many different functions with a piezoelectric crystal.
For example, piezoelectric microphones convert acoustic waves traveling though air into electronic signals. Piezoelectric speakers and buzzers perform the opposite function. Piezoelectric sensors detect changes in pressure, temperature, torque, humidity and/or a wide range of other phenomena.
Common piezoelectric materials include quartz (SiO2), zinc oxide (ZnO), barium titanate (BaTiO3), lithium tantalate (LiTaO3) and lithium niobate (LiNbO3). However, other materials, most notably silicon carbide (SiC) and the Group III-nitride materials such as aluminum nitride (AlN) and Gallium Nitride (GaN) are piezoelectric and may be used to form acoustic wave devices.
When a time-varying electric field is applied to a portion of a piezoelectric crystal, the applied electric field induces an acoustic wave that propagates through the crystal. Acoustic waves may travel through a piezoelectric material in a number of modes. For example, acoustic waves may travel through the body of the material—so-called "bulk" waves—or on the surface of the material. Waves that travel along the surface of the piezoelectric material are generally referred to as surface acoustic waves (or SAWs), and devices that process surface acoustic waves are referred to as surface acoustic wave devices, or SAW devices.
A simple surface acoustic wave device comprises a piezoelectric crystal or a thin film of piezoelectric material on a substrate. Interdigitated metal stripes on the surface of the crystal form transmitting and receiving electrodes. The metal electrodes convert electrical energy into mechanical stress in the crystal and vice versa. Hence, the interdigital electrodes formed on a piezoelectric material are referred to as interdigital transducers, or IDTs.
A simple surface acoustic wave device is illustrated in perspective in FIG. 10. The SAW device comprises a piezoelectric film 2 formed on a substrate 1. A metal (usually aluminum) is deposited on the film and patterned using standard photolithographic or liftoff techniques to form the input IDT 3 and the output IDT 4. The thickness of the piezoelectric film is generally on the order of one SAW wavelength.
In operation, an electric signal may be applied to the input IDT 3. The input signal causes a surface acoustic wave to be induced in the piezoelectric film 2 and propagate along the surface of the film 2 towards the output IDT 4. The shape of the generated wave depends on the electric signal applied to the input IDT, the design and orientation of the IDT fingers, and the piezoelectric material used. When the wave reaches the output IDT 4, a voltage is induced across the fingers of the IDT 4 which is then output from the device. The shape of the output wave is affected by the design of the output IDT 4.
FIG. 11 illustrates some design parameters for IDTs. The finger period D determines the wavelength λ of the SAW generated by the IDT. The linewidth L and space S of the fingers are generally equal to λ/4. The number of fingers determines the coupling efficiency of the IDT, and the width W of the overlap of fingers affects the frequency response of the finger pair. By changing the overlap of finger pairs within an IDT, various filter functions can be realized.
Surface acoustic wave devices have many different applications in digital and analog electronics. For example, surface acoustic wave devices may be used as bandpass or bandstop filters, duplexers, delay lines, resonators and/or impedance elements among other things. They may also be used to perform digital functions such as convolution, correlation, pulse compression and/or digital filtering (for example in spread-spectrum communication systems) depending on the design of the device, and in particular depending on the layout of the interdigital transducers. The design and fabrication of surface acoustic wave devices are described in Chapter 66 of K. Ng, Complete Guide to Semiconductor Devices, McGraw Hill (1995).
The velocity of surface acoustic waves in a device depends on the material from which the device is constructed and the mode of propagation of the SAW. For example, the propagation velocity (also called the SAW velocity) of first order Rayleigh-mode acoustic waves in GaN is around 3600 m/s, while the corresponding SAW velocity in AlN is about 5800 m/s, and over 6800 m/s in SiC. For RF devices, the SAW velocity determines the bandwidth of signals that can be processed by the device. The fundamental operating frequency (f0) of a SAW device is given by the formula:
      f    0    =      v    λ  where ν is the SAW velocity and λ is the wavelength. As discussed above, the wavelength of the device is determined by the finger period of the IDT. The width and spacing of IDT fingers (and thus the finger period) is limited by the resolution of photolithographic techniques. Thus, for a given finger period, increasing the SAW velocity increases the fundamental operating frequency of the device. Stated differently, having a higher SAW velocity permits a device to process higher-frequency signals for a given device geometry. Accordingly, the Group III-nitrides and SiC may be desirable piezoelectric materials for the fabrication of SAW devices.
Group III-nitrides and silicon carbide may also be desirable materials for the fabrication of high power, high temperature and/or high frequency devices. These wide bandgap materials have high electric field breakdown strengths and high electron saturation velocities as compared to other semiconductor materials such as gallium arsenide and silicon.
Electrical circuits requiring high power handling capability (>20 watts) while operating at high frequencies such as radio frequencies, including for example, S-band (2–4 GHz) and X-band (8–12 GHz), have in recent years become more prevalent. Because of the increase in high power, high frequency circuits there has been a corresponding increase in demand for transistors that are capable of reliably operating at radio frequencies and above while still being capable of handling higher power loads. Previously, bipolar transistors and power metal-oxide semiconductor field effect transistors (MOSFETs) have been used for high power applications but the power handling capability of such devices may be limited at higher operating frequencies. Junction field-effect transistors (JFETs) were commonly used for high frequency applications but the power handling capability of previously known JFETs may also be limited.
Recently, metal-semiconductor field effect transistors (MESFETs) have been developed for high frequency applications. The MESFET construction may be preferable for high frequency applications because only majority carriers carry current. The MESFET design may be preferred over current MOSFET designs because the reduced gate capacitance permits faster switching times of the gate input. Therefore, although all field-effect transistors utilize only majority carriers to carry current, the Schottky gate structure of the MESFET may make the MESFET more desirable for high frequency applications.
In addition to the type of structure, and perhaps more fundamentally, the characteristics of the semiconductor material from which a transistor is formed also affects the operating parameters. Of the characteristics that affect a transistor's operating parameters, the electron mobility, saturated electron drift velocity, electric breakdown field and thermal conductivity may have the greatest effect on a transistor's high frequency and high power characteristics.
Electron mobility is the measurement of how rapidly an electron is accelerated to its saturated velocity in the presence of an electric field. In the past, semiconductor materials which have a high electron mobility were preferred because more current could be developed with a lesser field, resulting in faster response times when a field is applied. Saturated electron drift velocity is the maximum velocity that an electron can obtain in the semiconductor material. Materials with higher saturated electron drift velocities are preferred for high frequency applications because the higher velocity may translate to shorter times from source to drain.
Electric breakdown field is the field strength at which breakdown of the Schottky junction and the current through the gate of the device suddenly increases. A high electric breakdown field material may be preferred for high power, high frequency transistors because larger electric fields generally can be supported by a given dimension of material. Larger electric fields may allow for faster transients as the electrons can be accelerated more quickly by larger electric fields than by smaller.
Thermal conductivity is the ability of the semiconductor material to dissipate heat. In typical operations, all transistors generate heat. In turn, high power and high frequency transistors usually generate larger amounts of heat than small signal transistors. As the temperature of the semiconductor material increases, the junction leakage currents generally increase and the current through the field effect transistor generally decreases due to a decrease in carrier mobility with an increase in temperature. Therefore, if the heat is dissipated from the semiconductor, the material may remain at a lower temperature and may be capable of carrying larger currents with lower leakage currents.
In the past, high frequency MESFETs have been manufactured of n-type III–V compounds, such as gallium arsenide (GaAs) because of their high electron mobilities. Although these devices provided increased operating frequencies and moderately increased power handling capability, the relatively low breakdown voltage and the lower thermal conductivity of these materials have limited their usefulness in high power applications.
Silicon carbide (SiC) has been known for many years to have excellent physical and electronic properties which should theoretically allow production of electronic devices that can operate at higher temperatures, higher power and higher frequency than devices produced from silicon (Si) or GaAs. The high electric breakdown field of about 4×106 V/cm, high saturated electron drift velocity of about 2.0×107 cm/sec and high thermal conductivity of about 4.9 W/cm-K indicate that SiC would be suitable for high frequency, high power applications.
SiC-based MESFET structures and their fabrication are described in U.S. Pat. No. 5,270,554 to Palmour et al. and U.S. Pat. No. 5,925,895 to Sriram et al., both of which are incorporated herein by reference as if fully set forth herein. SiC MESFET structures and fabrication are also described in U.S. application Ser. No. 09/567,717 filed May 10, 2000 by Allen, et al., the disclosure of which is incorporated herein by reference as if fully set forth herein.
In the nitride arena, a device of particular interest for high power and/or high frequency applications is the high electron mobility transistor (HEMT), which is also known as a heterostructure field effect transistor (HFET). These devices may offer operational advantages under a number of circumstances because a two-dimensional electron gas (2DEG) is formed at the heterojunction of two semiconductor materials with different bandgap energies, and where the smaller bandgap material has a higher electron affinity. The 2DEG is an accumulation layer in the undoped, smaller bandgap material and can contain a very high sheet electron concentration in excess of, for example, 1013 carriers/cm2. Additionally, electrons that originate in the wider-bandgap semiconductor transfer to the 2DEG, allowing a high electron mobility due to reduced ionized impurity scattering.
This combination of high carrier concentration and high carrier mobility can give the HEMT a very large transconductance and may provide a strong performance advantage over metal-semiconductor field effect transistors (MESFETs) for high-frequency applications.
High electron mobility transistors fabricated in the gallium nitride/aluminum gallium nitride (GaN/AlGaN) material system have the potential to generate large amounts of RF power because of the combination of material characteristics that includes the aforementioned high breakdown fields, their wide bandgaps, large conduction band offset, and/or high saturated electron drift velocity.
In electronic communication systems, it is usually desirable to amplify signals before transmission or after reception. It is also often desirable to filter such signals immediately before or after amplification. In high frequency communication systems, such amplification may be performed efficiently using an amplifier circuit incorporating a SiC MESFET or a Group III-nitride based transistor. Filtering may be efficiently performed using a SAW filter.
In order to minimize the number of circuit elements required to implement a communication system and simplify its design, it is desirable to integrate as many components as possible on a single chip. Attempts to integrate SAW devices with other devices have been made. However, such devices have typically required that the piezoelectric crystal be bonded onto a semiconductor substrate (such as silicon) on which active electronic components are formed.

Integrated nitride-based acoustic wave devices and methods of fabricating integrated nitride-based acoustic wave devices


A valve device, particularly for a breathing regulator for divers, including an inlet chamber which can be connected to a source of gas under super-atmospheric pressure, and an outlet chamber which is under a regulated pressure. A seating, which is movable under the influence of the super-atmospheric pressure and the regulated pressure, seals between the inlet chamber and the outlet chamber and has a through-passing passageway that interconnects the chambers. Also included is a valve body which is movably arranged in the seating such as to open and close the through-passing passageway in the seating. A servo element exerts on the valve body a force that depends on the position of the movable seating, so that varying force-influence on the valve body from the super-atmospheric pressure will be compensated for by a corresponding varying force from the servo element. This results in an essentially constant valve characteristic, while providing a mechanically simple solution at the same time.

Diving equipment includes a so-called breathing regulator which is connected to one or more diving tanks or their technical equivalence and which is intended to adjust the tank pressure to a predetermined regulated pressure. The breathing regulator is provided with a valve device to this end. FIG. 1 illustrates diagrammatically one such valve device constructed in accordance with known technology. As will be seen, the valve includes an inlet side 11, which is under tank pressure Pinlet, and an outlet side 12, which is under the regulated pressure Poutlet. A conical valve body 14 is movable in a seating 15. The outlet side 12 is delimited by a flexible diaphragm or membrane 16, which is connected to a spring 17 that exerts a spring force Fspring on the diaphragm 16. Thus, one side of the diaphragm 16 is subjected to forces from the regulated pressure Poutlet, while the other side of the diaphragm is subjected to forces from the ambient pressure Pambient. The spring force, together, e.g., with forces exerted by the diaphragm 16, acts/act on a rod 18 connected to the valve body and thus exerting a force Frod onto said body.
This known valve has the following modus operandi. Movement of the valve body 14 is determined by the forces to which it is subjected. These forces include (as shown in the figure) an upwardly directed force Finlet determined by the tank pressure. This force is counteracted essentially by downwardly acting forces Foutlet, which are comprised generally of the effect of the regulated pressure Poutlet on the valve body and of the rod force Frod. When a diver who has a nozzle connected to the outlet side inhales, therewith causing a decrease in the regulated pressure, the pressure on the diaphragm 16 decreases and the diaphragm then exerts an increasing force on the rod 18. The force Frod then increases and, in the case of a functioning valve, the valve body will be moved downwards, thereby allowing tank gas to flow in through the seating 15 until the regulated pressure has increased to an extent at which the valve body returns to the position shown in FIG. 1.
One problem with such known valve devices is that movement of the valve body is dependent on the tank pressure, which in the case of a full tank can correspond to a super atmospheric pressure of about 300 bar and may fall to close on 0 bar during use. This means, in turn, that the valve characteristic will vary, together with the regulated pressure.
Several solutions to this problem have been proposed. One example of these proposed solutions is illustrated in FIG. 2. The valve shown in FIG. 2 corresponds to the valve shown in FIG. 1 in many aspects, although in this latter case the homogenous valve body 14 has been replaced with a valve body 14′ that includes a through-passing passageway 14a′. This passageway connects the outlet 12 with a space 19, which is sealed against the inlet 11 by means of an O-ring 20. Consequently, those forces Finlet acting on the valve body from the inlet side have essentially no axial component, thereby providing a balanced valve that gives a regulated pressure generally independent of Pinlet.
The use of a valve body of this design, however, has the drawback that the sealing surface between the valve body and the seating ultimately tends to leak. There are several reasons for this. The fit between the hole and the cone of the valve body must be perfect—no irregulatories can be permitted. After having been in use for a longer period of time, the cone becomes damaged, in the form of scratches and ruts in the hole-defining edge and the cone, this damage contributing to seal leakage. Furthermore, the valve body must be straight, meaning that the body must be guided with utmost precision. The O-ring may begin to leak as a result of abrasion and other type of wear, thereby preventing achievement of the desired balancing effect and sealing effect.
These drawbacks associated with the use of a conical valve body are avoided when using a spherical sealing body. A spherical body is self-guiding, thereby obviating the need of the accurate guide required by a conical valve body.
It will be realised, however, that the balancing solution illustrated in FIG. 2 cannot be applied when a spherical body is used as a valve-closing means.

Balanced valve device and breathing apparatus comprising such valve device


A method and apparatus for implanting a testicular prosthetic device includes creating a passageway into the interior of an animal's scrotum which contains a tunica vaginalis; incising the tunica vaginalis containing a testis attached to a spermatic cord; severing the spermatic cord; removing the testis through the passageway and inserting a prosthetic device through the passageway and into the lumen of the tunica and closing the tunica. The second testis is similarly removed from its tunica vaginalis through the same passageway and a similar prosthetic device is placed in the cavity created. The tunica may be drawn outwardly, thought the passageway, and removal of the testis by incising the tunica and severing the spermatic cord may be performed outside the animal. The tunica may also be severed and the prosthetic device placed directly within the scrotal sac. The testicular implant is generally kidney-shaped and is sized to fit the tunica.

In order to control population growth, it is commonly recommended that animals not kept for breeding purposes be neutered. Unrestricted breeding among animals leads to overpopulation, particularly of "stray" or abandoned animals. In metropolitan areas, such overbreeding necessitates costly animal control programs to maintain public health and control nuisances. To avoid such problems, neutering is particularly encouraged for small animals kept as household pets, such as dogs and cats. In male animals, this is accomplished by orchidectomy, or surgical removal of the testes from the scrotal sac.
Such surgery is generally safe for the animal and relatively inexpensive. However, the surgery does alter the appearance of the animal. In large dogs, such as bloodhounds, mastiffs and shepherds, the empty scrotal sac is particularly noticeable. While recognizing that responsible pet ownership requires neutering, nonetheless, many pet owners find themselves uncomfortable with such a disfiguring procedure. The permanently altered appearance of their pet serves as a constant reminder of their surgical choice. Comments by strangers who misidentify the gender of the pet may serve to exacerbate this discomfort.

Surgical method and apparatus for implantation of a testicular prosthetic device


A thermostat failure detection device sequentially calculates a normal-time minimum water temperature on an assumption that a thermostat is normal and an internal combustion engine is operated in a state where the engine water temperature is less likely to rise, sequentially calculates a failure-time maximum water temperature on an assumption that the thermostat is in a stuck-open failure state and the internal combustion engine is operated in a state where the engine water temperature is likely to rise, determines the failure of the thermostat if the engine water temperature is lower than the normal-time minimum water temperature, determines the normality of the thermostat if the engine water temperature is higher than the failure-time maximum water temperature, and determines neither the normality nor the failure if the engine water temperature is between the normal-time minimum water temperature and the failure-time maximum water temperature.

A thermostat is provided in a cooling water flow passage of an internal combustion engine system. The thermostat closes a water passage to a radiator when an engine is cold. By doing so, cooling water is not circulated to the radiator. As a result, the warm-up of the engine is promoted. When the temperature of the cooling water reaches a preset valve opening temperature of the thermostat, the thermostat adjusts a flow rate of the cooling water to the radiator by opening the water passage to the radiator. As a result, the cooling water is maintained at a suitable temperature.
If the thermostat breaks down, a control as described above cannot be executed. Accordingly, JP2004-316638A proposes a technique for determining a failure of a thermostat. In this JP2004-316638A, the failure of the thermostat is determined if a detected engine water temperature is lower than a reference determination temperature calculated in consideration of the influence of traveling wind.

Thermostat failure detection device and thermostat failure detection method


In accordance with one aspect of the present disclosure, an apparatus and a method of testing one of aerodynamic and acoustic characteristics of a vehicle is described. The method includes measuring a vehicle center using a gauge. The gauge includes a carrier section from which a pair of arms extend in opposed directions. Each of the arms has an axle center locator. The carrier section has a vehicle center marker. Upon determination of the vehicle center, the vehicle can be located in a wind tunnel at a predetermined location and exposed to an enhanced air flow.

The present exemplary embodiment relates to an apparatus associated with aerodynamic and acoustic testing. It finds particular application in conjunction with aerodynamic and acoustic testing of automotive vehicles, and will be described with particular reference thereto. However, it is to be appreciated that the present exemplary embodiment is also amenable to other like applications.
A wind tunnel is often used to test the aerodynamics and the acoustics of an automotive vehicle. A wind tunnel can include a test section housing within which a vehicle to be tested is positioned. An air flow generator such as a fan or blower is used to create an air stream that is directed through a contraction cone and the test section within which the vehicle is positioned. Wind tunnels may be of an open circuit design in which atmospheric air is drawn into the contraction cone and discharged from the diffuser back to the atmosphere. Alternatively, wind tunnels may be designed to recirculate air in a loop or return flow configuration. A wind tunnel also includes instrumentation for measuring the desired information, such as the wind drag created by a vehicle or sounds created as the wind flows over the vehicle.
Wind tunnels are used extensively in the automotive industry for the purpose of determining the effect of air flow on the major system components of a particular vehicle design. For example, testing may involve evaluating the effect of air flow on the vehicles ability to cool the vehicle engine. In this instance, a "climatic" wind tunnel will be used to verify system performance through a range of simulated environmental conditions.
A second category of vehicle evaluations in a wind tunnel involves evaluating or testing the aerodynamic and acoustic effect of the air flow as it flows around the vehicle. For example, the design of windshield wipers, the design of radio antennas, the overall exterior configuration of the vehicle, the effect of open vs. closed windows, etc. are important and regularly evaluated features of the vehicle's design.
Wind tunnels for use in the aerodynamic and acoustic testing of vehicles require careful design to ensure accurate and reproducible simulation of road conditions at the applicable speeds and thus provide reliable design data. To achieve accurate and reliable results and to avoid resonance problems for aerodynamic and acoustic testing, the ability to precisely position a vehicle at a specific location within the test section of the wind tunnel is important. Moreover, by way of example, if a feature on the surface of a vehicle is being modified to evaluate a change in aerodynamic performance, it is important that the vehicle be positioned at the same location within the testing section for each comparative evaluation.
Presently, a vehicle center is positioned at a designated location within the testing section. A vehicle center is identified on the vehicle manually by using measurement squares and a tape measure. The axle centers are projected down to the floor and the distance between the marks is measured. The measurement is divided in half and the calculated sum is indexed from the front axle center and manually marked onto the vehicle to be tested. This methodology has proven to be time consuming and a source of human error.

Apparatus for location of vehicle center for aerodynamic testing


A projector includes a measurement unit having a function of measuring at least two of a period during which the projector remains powered on, a period of an in-use state in which a user is using the projector, and a period of a non-use state in which the user is not using the projector; and a control unit having a function, of determining an in-use state ratio or a non-use state ratio associated with the projector based on the periods measured by the measurement unit and a function of executing an energy saving setting based on the in-use state ratio or the non-use state ratio.

1. Technical Field
The present invention relates to a projector and a projector controlling method.
2. Related Art
In a projector that projects an image to be projected, a state in which the projector remains powered on is classified into tire following two states: an in-use state in which a user is using the projector; and a non-use state in which the user is not using the projector.
Conceivable examples of the case where the projector is in the in-use state may include a case where the projector is projecting images to be projected (motion images and still images) based, for example, on an external signal and a case where the user is operating the projector.
Further, conceivable examples of the case where the projector is in the non-use state may include a case where a signal or any other form of information is not externally inputted and the user is not operating the projector and a case where the user temporarily stops projecting motion images to be projected and leaves the projector powered on.
The projector in the non-use state is, so to speak, an apparatus that simply projects meaningless light. When the proportion of the non-use state is large, unnecessary electric power is undesirably consumed and the life of each component that forms the projector (life of light source section that outputs light, in particular) is undesirably wasted.
In view of the circumstances described above, as a projector having a function of reducing unnecessary power consumption and preventing the life of each component that forms the projector from being wasted, a projector that measures an accumulated period of the non-use state and acquires other types of information and displays them has been developed (see JP-A-2G12-132977, tor example). The projector of the related art is intended to solve the problems of unnecessary power consumption and waste of the life of each component that forms the projector (life of light source section that outputs light, in particular).
To solve the problems described above, the projector of the related art essentially requires user's active actions operation performed by user themselves), such, as reviewing a method for using the projector based on the displayed information and changing the setting of the projector. Therefore, the problems described above cannot be solved unless the user takes active actions.

Projector and projector controlling method


To surely detect a frequency shift and a timing difference of a received signal making use of a chirp signal even when the C/N ratio of the received signal is low by resolving the problem of resolution of FFT processing, a frequency and timing synchronization circuit comprises: a chirp signal generator (6) for generating a reference chirp signal and a conjugate complex signal of the reference chirp signal; a first demodulator (7) for obtaining a first in-phase complex signal and a first quadrature-phase complex signal by demodulating the received signal with the reference chirp signal and the conjugate complex signal a frequency offset circuit (5) for obtaining an offset signal by shifting a frequency of the received signal; a second demodulator (8) for obtaining a second in-phase complex signal and a second quadrature-phase complex signal by demodulating the offset signal with the reference chirp signal and the conjugate complex signal; and a detector unit (9) for detecting the frequency shift and a difference of the synchronization timing according to a first peak frequency giving a maximum power spectrum among frequency components of the first and the second in-phase complex signal, and a second peak frequency giving a maximum power spectrum among frequency components of the first and the second quadrature-phase complex signal.

The present invention relates to a frequency and timing synchronization circuit to be applied to a TDMA (Time Division Multiple Access) receiver making use of a chirp signal.
In a radio communication system such as a mobile communication system, where high-speed digital signals are transmitted through multi-paths changing their status every moment, it is very important for a receiver how to quickly and exactly establish initial synchronization of detection frequency and symbol timings with a receiving radio signal even in a low C/N (Carrier to Noise) ratio.
In a Japanese patent application entitled xe2x80x9cFrequency Correction Apparatusxe2x80x9d and laid open as a Provisional Publication No. H09-008765 (hereinafter called the first prior art), there is disclosed a method of correcting a reference frequency for detecting an OFDM (Orthogonal Frequency Division Multiplex) signal.
The frequency correction apparatus of the first prior art comprises a frequency converter having a controllable local oscillator for converting the OFDM radio-frequency signal into an intermediate-frequency signal, a quadrature demodulator for splitting the intermediate-frequency signal into an in-phase and a quadrature-phase signal, a complex FFT (Fast Fourier Transform) circuit for performing complex Fourier transformation of the in-phase and the quadrature-phase signal, and a power measurement circuit for obtaining a frequency distribution of power spectrums of the OFDM signal in a frequency domain from the output of the complex FFT circuit.
By detecting a center position of the frequency distribution when multi-carriers are arranged symmetrically to a center frequency of the OFDM signal in the frequency domain, or by detecting a position of a reference carrier when it is inserted in the OFDM signal with an intensity different to other carriers, the actual center frequency of the OFDM signal is determined and the local oscillator is feedback-controlled for acquiring frequency synchronization with the receiving signal, in the first prior art.
In another Japanese patent application entitled xe2x80x9cA Frequency Error Detection Circuit by Detecting a Correlation Peakxe2x80x9d and laid open as a Provisional Publication No. H09-200081 (hereinafter called the second prior art), a method of detecting a frequency error for controlling an AFC (Automatic Frequency Control) circuit is disclosed to be applied to a DSSS (Direct Sequence Spread Spectrum) receiver.
The frequency error detection circuit according to the second prior art comprises;
a first complex code generator for generating a first complex code which is obtained by giving a certain frequency offset to a pseudo random scramble code toward a positive side,
a second complex code generator for generating a second complex code obtained by giving the same frequency offset to the pseudo random scramble code toward a negative side,
a first and a second complex matched filter each calculating complex correlation of a base-band complex signal obtained through a quadrature demodulator with the first complex code and. the second complex code;
a peak timing detector for outputting a peak timing for every symbol where either or both of outputs of the first and the second complex matched filter show a maximum power value,
a first and a second peak average detector for outputting a first and a second peak average by calculating averages, for several symbols, of peak values which are extracted from outputs of the first and the second complex matched filter at each peak timing detected by the peak timing detector,
a first and a second power calculator for calculating power values of the first and the second peak average, respectively,
a power difference calculator for obtaining a normalized power difference by dividing a difference between outputs of the first and the second power calculator by a sum thereof, and
a frequency error converter for outputting a frequency error corresponding to the normalized power difference referring to a table memory.
The complex matched filter functions as a band-pass filter. Therefore, the power value of the first peak average becomes larger than that of the second peak average when a positive frequency error remains in the base-band complex signal, and vice versa, which is detected by the frequency error converter and converted into the frequency error to be fed back to the quadrature detector, in the frequency error detection circuit of the second prior art.
There is also a method of establishing frequency and timing synchronization making use of a chirp signal, which is to be applied to a mobile terminal, for example, receiving a SCH (Synchronization Channel) signal according to the TDMA communication system.
FIG. 3 is a block diagram illustrating a conventional frequency and timing synchronization circuit 104 making use of the chirp signal.
The frequency and timing synchronization circuit 104 of FIG. 3 comprises a chirp signal generator 6, a demodulator 7 and a detector unit 9.
From a transmitter 1, a radio signal, wherein certain symbols are scrambled with a chirp signal s(t) represented by following equation (1), is transmitted.                                                                         s                                                  (                  t                  )                                            =                                                2                                                                  cos                                                        (                                                                  π                                                                          xe2x80x83                                                                                t                          2                                                                                μ                                                                          xe2x80x83                                                                                T                          2                                                                                )                                                                                =                                                1                                      2                                                                                {                                                            exp                                                                    (                                                                              j                                                                                xe2x80x83                                                                                π                                                                                xe2x80x83                                                                                t                              2                                                                                μ                                                                                xe2x80x83                                                                                T                              2                                                                                )                                                              +                                          exp                                                                    (                                                                                -                              j                                                                                xe2x80x83                                                                                π                                                                                xe2x80x83                                                                                t                              2                                                                                μ                                                                                xe2x80x83                                                                                T                              2                                                                                )                                                                              }                                                                                (        1        )
Here, t, xcexc and T is a time counted from beginning of each symbol, a constant called a chirp rate, and a symbol cycle, respectively.
The radio signal arrives through transmission paths to a receiver added with a frequency shift xcex94f due to fading, for example, and therefore, the chirp signal r(t) in the received signal is expressed as follows.                                                                         r                                                  (                  t                  )                                            =                                                s                                                        (                    t                    )                                                  ·                                  exp                                                        (                                          j2πΔ                                                                    xe2x80x83                                                                  f                                                                    xe2x80x83                                                                  t                                        )                                                                                =                                                1                                      2                                                                                {                                                            exp                                                                    (                                                                              j                                                                                xe2x80x83                                                                                2                                                        πΔ                                                                                xe2x80x83                                                                                f                                                                                xe2x80x83                                                                                t                                                    +                                                                                j                                                                                xe2x80x83                                                                                π                                                                                xe2x80x83                                                                                t                                2                                                                                μ                                                                                xe2x80x83                                                                                T                                2                                                                                )                                                              +                                          exp                                                                    (                                                                              j                                                                                xe2x80x83                                                                                2                                                                                xe2x80x83                                                                                πΔ                                                                                xe2x80x83                                                                                f                                                                                xe2x80x83                                                                                t                                                    -                                                                                j                                                                                xe2x80x83                                                                                π                                                                                xe2x80x83                                                                                t                                2                                                                                μ                                                                                xe2x80x83                                                                                T                                2                                                                                )                                                                              }                                                                                (        2        )
The chirp signal generator 6 generates a reference chirp signal having the same chirp rate xcexc with the transmission signal and expressed as       exp              (                        j                                xe2x80x83                              π                                xe2x80x83                                                        (                              t                +                                  Δ                                                        xe2x80x83                                                      t                                            )                        2                                    μ                                xe2x80x83                                          T            2                              )        ,
and its conjugate complex signal expressed as       exp              (                                    -            j                                          xe2x80x83                              π                                xe2x80x83                                                        (                              t                +                                  Δ                                                        xe2x80x83                                                      t                                            )                        2                                    μ                                xe2x80x83                                          T            2                              )        ,
xcex94t being a difference of symbol timings. The reference chirp signal and the conjugate complex signal are multiplied onto the received signal by the demodulator 7 for obtaining following two complex signals P(t) and Q(t) (hereinafter called the in-phase complex signal and the quadrature-phase complex signal, respectively) to analyzed by the detector unit 9;                                           P                                      (              t              )                                =                                    r                                            (                t                )                                      ·                          exp                                            (                                                      j                                                              xe2x80x83                                                            π                                                              xe2x80x83                                                                                (                                                  t                          +                                                      Δ                                                                                xe2x80x83                                                                                t                                                                          )                                            2                                                                            μ                                                              xe2x80x83                                                                                T                      2                                                                      )                                                    ,                            (        3        )                                          Q                                (            t            )                          =                              r                                      (              t              )                                ·                                    exp                                            (                                                                            -                      j                                                                                xe2x80x83                                                            π                                                              xe2x80x83                                                                                (                                                  t                          +                                                      Δ                                                                                xe2x80x83                                                                                t                                                                          )                                            2                                                                            μ                                                              xe2x80x83                                                                                T                      2                                                                      )                                      .                                              (        4        )
FIG. 4 is a block diagram illustrating a configuration of the detector unit 9 of FIG. 3, comprising a complex FFT circuit 10, a power detector 11 and a comparator 12.
Each of the in-phase and the quadrature-phase complex signal P(t) and Q(t) is transformed into a frequency domain signal through the complex FFT circuit 10, whereof a power value of each frequency component is calculated by the power detector 11 and a frequency component giving a maximum power spectrum is detected by the comparator 12.
As to in-phase complex signal P(t) of equation (3), it is expressed as P(t)=A+B, where;
                              A          =                                    {                              first                                                  xe2x80x83                                                term                                                  xe2x80x83                                                of                                                  xe2x80x83                                                equation                                                  xe2x80x83                                                                  (                  2                  )                                            }                        ·                          exp                                            (                                                      j                                                              xe2x80x83                                                            π                                                              xe2x80x83                                                                                (                                                  t                          +                                                      Δ                                                                                xe2x80x83                                                                                t                                                                          )                                            2                                                                            μ                                                              xe2x80x83                                                                                T                      2                                                                      )                                                    ,                            (        5        )
and
                              B          =                                    {                              second                                                  xe2x80x83                                                term                                                  xe2x80x83                                                of                                                  xe2x80x83                                                equation                                                  xe2x80x83                                                                  (                  2                  )                                            }                        ·                          exp                                            (                                                      j                                                              xe2x80x83                                                            π                                                              xe2x80x83                                                                                (                                                  t                          +                                                      Δ                                                                                xe2x80x83                                                                                t                                                                          )                                            2                                                                            μ                                                              xe2x80x83                                                                                T                      2                                                                      )                                                    ,                            (        6        )
The term A of equation (5) is calculated as;                                                         A              =                                                1                                      2                                                                                xe2x80x83                                                                                exp                                                              (                                                                        j                                                                                xe2x80x83                                                                              2                                                    πΔ                                                                                xe2x80x83                                                                              f                                                                                xe2x80x83                                                                              t                                                +                                                                              j                                                                                xe2x80x83                                                                                π                                                                                xe2x80x83                                                                                t                              2                                                                                μ                                                                                xe2x80x83                                                                                T                              2                                                                                )                                                        ·                  exp                                                                  (                                                            j                                                                    xe2x80x83                                                                  π                                                                    xe2x80x83                                                                                (                                                      t                            +                                                          Δ                                                                                xe2x80x83                                                                                t                                                                                )                                                2                                                                                μ                                                                    xe2x80x83                                                                                T                        2                                                                              )                                                                                =                                                1                                      2                                                                                xe2x80x83                                                                  exp                                                        (                                                                                j                                                                                xe2x80x83                                                                                2                                                        π                                                                                xe2x80x83                                                                                t                              2                                                                                +                                                      j                                                                                xe2x80x83                                                                                2                                                        π                                                                                xe2x80x83                                                                                t                                                                                xe2x80x83                                                                                Δ                                                                                xe2x80x83                                                                                t                                                    +                                                      j                                                                                xe2x80x83                                                                                π                                                                                xe2x80x83                                                                                Δ                                                                                xe2x80x83                                                                                t                              2                                                                                μ                                                                                xe2x80x83                                                                                T                            2                                                                                +                                              j                                                                          xe2x80x83                                                                        2                                                π                                                                          xe2x80x83                                                                        Δ                                                                          xe2x80x83                                                                        f                                                                          xe2x80x83                                                                        t                                                              )                                                                                (        7        )
which is a product of exp(2jxcfx80t2) whereof frequency varies according to time passage, and therefore, the term A makes no contribution to the maximum power spectrum after FFT.
The term B of equation (5) is calculated as;                                                         B              =                                                1                                      2                                                                                xe2x80x83                                                                                exp                                                              (                                                                        j                                                                                xe2x80x83                                                                              2                                                    πΔ                                                                                xe2x80x83                                                                              f                                                                                xe2x80x83                                                                              t                                                -                                                                              j                                                                                xe2x80x83                                                                                π                                                                                xe2x80x83                                                                                t                              2                                                                                μ                                                                                xe2x80x83                                                                                T                              2                                                                                )                                                        ·                                      exp                                                              (                                                                        j                                                                                xe2x80x83                                                                              π                                                                                xe2x80x83                                                                                (                                                              t                                +                                                                  Δ                                                                                xe2x80x83                                                                                t                                                                                )                                                        2                                                                                μ                                                                                xe2x80x83                                                                                T                            2                                                                                )                                                                                =                                                1                                      2                                                                                xe2x80x83                                                                  exp                                                        (                                                                                j                                                                                xe2x80x83                                                                                2                                                        π                                                                                xe2x80x83                                                                                t                                                                                xe2x80x83                                                                                Δ                                                                                xe2x80x83                                                                                t                                                    +                                                      j                                                                                xe2x80x83                                                                                π                                                                                xe2x80x83                                                                                Δ                                                                                xe2x80x83                                                                                t                              2                                                                                μ                                                                                xe2x80x83                                                                                T                            2                                                                                +                                              j                                                                          xe2x80x83                                                                        2                                                π                                                                          xe2x80x83                                                                        Δ                                                                          xe2x80x83                                                                        f                                                                          xe2x80x83                                                                        t                                                              )                                                                                (        8        )
whereof   exp        (                  j                          xe2x80x83                        π                          xe2x80x83                        Δ                          xe2x80x83                                  t          2                            μ                          xe2x80x83                                  T          2                      )
is a constant term to be cancelled in the power spectrum after FFT.
Therefore, a frequency dependent component P, which defines the maximum power spectrum after FFT, of the in-phase complex signal P(t) can be expressed as follows;                     P        =                              1                          2                                                      exp                                      (                                                j2π                                                        (                                                                  Δ                                                                          xe2x80x83                                                                        f                                            +                                                                        Δ                                                                                xe2x80x83                                                                              t                                                                          μ                                                                                xe2x80x83                                                                                T                            2                                                                                )                                                                  t                            )                                                          (        9        )
In the same way, a frequency dependent component Q, which defines the maximum power spectrum after FFT, of the quadrature-phase complex signal Q(t) can be expressed as follows;                     Q        =                              1                          2                                                      exp                                      (                              j                                                  xe2x80x83                                                2                                                  xe2x80x83                                                                  π                                                        (                                                                  Δ                                                                          xe2x80x83                                                                        f                                            -                                                                        Δ                                                                                xe2x80x83                                                                              t                                                                          μ                                                                                xe2x80x83                                                                                T                            2                                                                                )                                                                  t                            )                                                          (        10        )
Therefore, the frequency domain signal of the in-phase complex signal P(t) after processed by the complex FFT circuit 10 has a power spectrum peak at a frequency Fp=xcex94f   Fp  =            Δ                    xe2x80x83                  f        +                  Δ                          xe2x80x83                        t                    μ                          xe2x80x83                                  T          2
as shown in FIG. 5A, while that of the complex signal Q(t) has a power spectrum peak at a frequency Fq=xcex94f   Fp  =            Δ                    xe2x80x83                  f        -                  Δ                          xe2x80x83                        t                    μ                          xe2x80x83                                  T          2
as shown in FIG. 5B, which are detected by the comparator 12.
From these frequencies Fp and Fq, the frequency shift xcex94f and the difference of symbol timing At are calculated as follows;
xcex94f=(Fp+Fq)/2,xe2x80x83xe2x80x83(11)
xcex94t=(Fpxe2x88x92Fq)xc2x7xcexcT2/2,xe2x80x83xe2x80x83(12)
According to the frequency shift xcex94f and the difference xcex94t of the symbol timings thus obtained, the frequency and timing synchronization is performed in the conventional frequency and timing synchronization circuit 104 of FIG. 3 making use of the chirp signal.
However, there is a problem of frequency resolution of the complex FFT circuit 10 in the conventional frequency and timing synchronization circuit 104 of FIG. 3, when the C/N ratio of the received signal is low.
FIG. 6 is a spectrum chart for illustrating the above problem, whereof Y-axis represents frequency offset to a reference frequency and X-axis represents a power value of a signal according to the frequency offset. The resolution of the FFT processing is determined by the sampling frequency fs and the number m of samples, and a vector value is calculated for each frequency component ranged with a frequency difference fs/m, as represented by black dots in FIG. 6.
Therefore, when a signal has a peak value p at the same frequency with the reference frequency or at a frequency just shifted by n (n being an integer) times of fs/m from the reference frequency as illustrated in FIG. 6 with a dashed line denoted as Best Case, the peak value p is output as it is, at the frequency component. However, when the signal has the peak value p at a frequency in a middle of two consecutive resolution steps of the FFT as illustrated with a solid line denoted as Worst Case, the peak value p is to be represented by two off-peak values a of two frequency components nearest to the frequency offset fp giving the peak value p, that is, frequency components having frequency offsets 0 and +fs/m, in the example of FIG. 6.
Therefore, when the signal has a sufficient C/N ratio, the actual offset fp of the peak frequency can be estimated to be fp=+fs/2m, in the example, from frequency offsets 0 and +fs/m of the two off-peak values xcex1. However, when the C/N ratio is low, the off-peak values xcex1 may become equivalent to the noise level and become impossible to be detected by the comparator 12.
This is the problem.
Therefore, a primary object of the present invention is to deal with the above problem and provide a frequency and timing synchronization circuit which can surely detect the frequency shift and the timing difference of a received signal making use of the chirp signal, even when the C/N ratio of the received signal is low.
In order to achieve the object, a frequency and timing synchronization circuit making use of a chirp signal according to the invention, for detecting a frequency shift and a synchronization timing of a received signal modulated with a transmission chirp signal, comprises:
a chirp signal generator for generating a reference chirp signal having the same chirp rate with the transmission chirp signal and a conjugate complex signal of the reference chirp signal;
a first demodulator for obtaining a first in-phase complex signal and a first quadrature-phase complex signal by demodulating the received signal with the reference chirp signal and the conjugate complex signal, respectively;
a frequency offset circuit for obtaining an offset signal by shifting a frequency of the received signal by an offset frequency;
a second demodulator for obtaining a second in-phase complex signal and a second quadrature-phase complex signal by demodulating the offset signal with the reference chirp signal and the conjugate complex signal, respectively; and
a detector unit for detecting the frequency shift and the synchronization timing according to a first and a second peak frequency, the first peak frequency giving a maximum power spectrum among frequency components of the first and the second in-phase complex signal, and the second peak frequency giving a maximum power spectrum among frequency components of the first and the second quadrature-phase complex signal.
a detector unit for detecting the frequency shift and a difference of the synchronization timing according to a first peak frequency giving a maximum power spectrum among frequency components of the first and the second in-phase complex signal after processed through FFT, and a second peak frequency giving a maximum power spectrum among frequency components of the first and the second quadrature-phase complex signal after processed through FFT.
The offset frequency is determined in consideration of resolution of the complex FFT circuit defined by a sampling frequency and a number of samples applied in the complex FFT circuit.
Therefore, the effective resolution step of the FFT can be made fine, and the frequency shift and the timing difference of a received signal can be detected surely and exactly making use of the chirp signal, even when the C/N ratio of the received signal is low.
In place of the frequency offset circuit and the second demodulator, there may be comprised:
a plurality of frequency offset circuits each obtaining respective one of offset signals by shifting a frequency of the received signal by a distinct offset frequency; and
second demodulators each obtaining respective one of offset in-phase complex signals and respective one of offset quadrature-phase complex signals by demodulating respective one of the offset signals with the reference chirp signal and the conjugate complex signal, respectively.
In this case, the first peak frequency is detected from a frequency component giving a maximum power spectrum among frequency components of the first in-phase complex signal and the offset in-phase complex signals after processed through FFT, and a second peak frequency is detected from a frequency component giving a maximum power spectrum among frequency components of the first quadrature-phase complex signal and the offset quadrature-phase complex signals after processed through FFT.

Frequency and timing synchronization circuit making use of a chirp signal


A circuit configuration for measuring air humidity includes a measuring system which includes an oscillator unit and a signal-processing unit, as well as a frequency-determining impedance, which is determined largely by a parallel connection of a capacitive humidity sensor and a parasitic resistance. The frequency-determining impedance is connected to the oscillator unit, which includes an oscillator having an oscillation frequency that is a function of the frequency-determining impedance. The oscillator unit includes a switchable impedance, which can be connected in addition to the frequency-determining impedance, the oscillator thereby being switchable from a first frequency range to a second frequency range. An oscillator signal having the oscillation frequency of the oscillator is supplied to the signal-processing unit in which the frequency of the oscillator signal is able to be measured, and a correction value to compensate for the influence of the parasitic parallel resistance on the oscillation frequency of the oscillator is ascertainable as a function of the difference between a first measurement of the frequency of the oscillator signal in the first frequency range and a second measurement of the frequency of the oscillator signal in the second frequency range.

Capacitive humidity sensors are mainly used for measuring the air humidity, such as, for example, in air-conditioning technology for controlling an interior environment. Generally, they are thin-film sensors whose measured quantity is the moisture-dependent capacitance of the polymer layer between the electrodes. Sensors of this type are described, for example, in H.-R. Traenkler, E. Obermeier, "Sensortechnik, Handbuch fuer Praxis und Wissenschaft" ("Sensor Technology, Handbook for Practice and Science"), Springer Verlag, 1998, pages 1247 to 1250.
In order to be able to measure the capacitance of the sensor, oscillator circuits are often used in which the capacitive sensor is the frequency-determining member. In this case, a value for the air humidity may be ascertained simply by measuring the oscillator frequency and assigning a humidity value to the result of the measurement. Examples for such circuits in the field of moisture measurement can be found, for example, in H. Lemme, "Sensoren in der Praxis" ("Sensors in Practice"), Franzis-Verlag, 1993, pages 327 to 336.
U.S. Pat. No. 5,844,138 describes a device for measuring humidity, in which a capacitive humidity sensor is the frequency-determining element of an oscillator. The instantaneous air humidity is ascertained by measuring the oscillator frequency and converting it into a humidity value with the aid of calibration data stored in a memory unit.
Particularly in an industrial environment, parasitic parallel resistances, caused by contamination of the sensor connections or by material properties of the sensor itself, appear at the capacitive humidity sensor and also influence the oscillator frequency of the measuring circuit, and thereby falsify the measurement result. Such changes of the impedance connected to the oscillator are not described as being taken into account in U.S. Pat. No. 5,844,138.
The measuring error which comes about due to the parallel connection of a parasitic resistance to the capacitive humidity sensor can be corrected if not only the capacitance, but also the impedance of the measuring sensor is measured, since phase information is also acquired at the same time, and the error produced by the parasitic parallel resistance can be eliminated. However, circuit configurations for measuring impedance are quite complicated and cost-intensive.
European Published Patent Application No. 1 411 349 describes a method for recording the air humidity, in which the charging duration and/or discharging duration of a capacitive humidity sensor is recorded via two different measuring resistors, and the capacitance of the sensor is calculated from the results. In this case, the capacitive sensor is modeled by a parallel connection of an ideal capacitor with an ohmic resistance, both the resistance value and the capacitor value being a function of the moisture and the temperature.

Circuit configuration and method for measuring humidity


The invention provides a recordable display apparatus, which uses a converting unit to convert the digital image signal into a specific image signal, in which the specific image signal has a signal format acceptable by the multimedia processor. The multimedia processor is used for integrating the specific image signal and the sound signal produced by the sound-receiving unit during receiving and recording the ambient sound into an audio/video (AV) recording data.

1. Field of the Invention
The invention generally relates to a recordable display apparatus, and more particularly, to a display apparatus with audio/video (AV) recording function.
2. Description of Related Art
The projector is used as a display apparatus for producing a large-size image. The imaging principle of a projector is based on converting an illumination beam generated by a light source module into an image beam through a light valve, and then, projecting the image beam onto a screen or wall through a projection lens so as to form images. With the advances in projection technology and lower manufacturing costs, the application of the projector has been gradually expanded to domestic purposes from commercial purpose.
Although today it is very convenient to play image information such as the briefing or movie in a meeting or class by using the projector, however, during the process of the meeting or class, participants are often unable to catch recording the speech content of the meeting host or teaching presentation. At the time, the above-mentioned problem can be solved by additionally erecting a video camera, but it is troublesome to erect the video camera on site and the effect of the recorded projection image is limited (for example, the clarity is not good and it is affected by the brightness of the ambient light in site).
U.S. Pat. No. 7,303,282 discloses a multimedia display apparatus, used to produce image signal and audio signal when a projection apparatus is electrically connected to a docking station. U.S. Pat. No. 6,827,447 discloses a recordable projector. U.S. Pat. Nos. 6,966,651 and 7,052,136 disclose a portable multimedia projection system, which includes a portable housing, a media player, a sound system, a projection apparatus, a controller and a damping apparatus, wherein the sound system has at least one internally-mounted speaker.


An automotive starting system includes an accumulator, a hydraulic motor, and an engine controller. The accumulator is adapted for holding a charge of pressurized fluid. The hydraulic motor has a fluid input operatively connected to the accumulator and has an output shaft operatively connected to an internal combustion engine of an automotive vehicle. The engine controller shuts off the engine if an idle engine condition exists and restarts the engine using the hydraulic motor following an idle engine shut-off if a predetermined engine-start condition exists. A method for starting an internal combustion engine of an automotive vehicle includes having the engine controller shut off the engine if an idle engine condition exists and thereafter having the engine controller restart the engine using the hydraulic motor if a predetermined engine-start condition exists.

The automotive industry has utilized an electric starter engine powered by an automotive vehicle's 12-volt battery for an operator to start the internal combustion engine of the vehicle. However, known automotive designs include using an engine controller to temporarily shut off the engine to reduce fuel consumption and emissions if the engine is in an idle-engine state. An example of an idle-engine state includes the vehicle being stopped for a predetermined time threshold such as can occur when the vehicle encounters a traffic light. These designs usually employ an integrated starter/generator either on the front end accessory drive or engaged with the flywheel in a suitable arrangement. This typically requires using a nominal 42-volt electrical system instead of a nominal 12-volt electrical system for all but smaller displacement engines. Such designs add considerable complexity to the vehicle and add corresponding cost and weight. After an idle engine shut-off has occurred, the engine controller uses the electric starter motor to restart the engine when a predetermined engine-start condition exists. An example of an engine-start condition includes the presence of the key in the ignition ("Key-On") since the idle engine shut-off and the absence of brake pressure.
Known designs of certain diesel trucks (such as that disclosed in U.S. Pat. No. 5,528,901) include using an electric starter motor to start an auxiliary diesel engine to power an air compressor to power a separate pneumatic starter motor of a main diesel engine to start the main diesel engine. The operator uses only the auxiliary diesel engine when the truck is at a truck stop to power auxiliary equipment, and the operator starts and uses the main diesel engine when the truck is traveling on the road.
Still, scientists and engineers continue to seek improved automotive starting systems.


There is provided mechanisms for remote provision of a subscriber entity to a PLMN in a communications network. A method is performed by the subscriber entity. The method comprises obtaining a first list of PLMN IDs in the communications network for which a signal quality criterion is fulfilled. The method comprises providing the first list of PLMN IDs and eUICC ID of the subscriber entity to a network server. The method comprises receiving a third list of PLMN IDs from the network server, wherein the third list of PLMN IDs is based on the first list of PLMN IDs. The method comprises selecting, when the third list of PLMN IDs is a non-empty list, one PLMN from the third list of PLMN IDs to remotely provision the subscriber entity.

Mobile networks are being used to connect all sorts of devices; automated reading of utility meters, intelligent connectivity of cars and commercial vehicles to enable drivers to access navigation, infotainment or breakdown services, traffic lights, home security and assisted living.
A subscriber identity module or subscriber identification module (SIM) is an integrated circuit chip that is intended to securely store the international mobile subscriber identity (IMSI) number and its related key, which are used to identify and authenticate subscribers on the devices. The SIM circuit is part of the function of a Universal Integrated Circuit Card (UICC) physical smart card. Traditional SIM cards are predicated on only associating with one network operator.
The GSMA specifies an embedded SIM service. Further, OEMs are providing managed connectivity by embedding the connectivity into the device, using a "blank" eSIM, and then depending on where (in which country) the device is sold, the device/eSIM can be finally localized to a local operator in that country. Here, localization involves the device entering a new country from its home/manufacture country to localize its eSIM profile to be the local operator profile, so that the device is attached by the localized eSIM profile to the local operator network, in order to meet local regulatory/commercial requirements.
Hence, there is still a need for improved localization of a subscriber entity.


A hand held implement for raking is provided with an apparatus for cleaning of debris which becomes lodged on and among the tines by means of an actuator, requiring a minimum of effort and disruption of the raking rhythm on the part of the operator in order to effect cleaning. The tines are transversely flexible, allowing the movement of a cleaning bar, which also serves as a stress distributor and tine separator, along and about and to the distal ends of the tines for the purpose of removing lodged debris. The cleaning bar is attached to and operated by a hollow actuator, which moves along and about the rake handle, by an intervening spring, which applies downward tension to the tines. The tines, in turn diverging and downward arcing, provide tension to help keep the cleaning apparatus as positioned by the operator anywhere along its travel upon the tines. The cleaning apparatus is further kept at the proximal end of its travel by the raking motion of the operator and is activated by rotation of the lower arm about the elbow without requiring a change of grip by either hand of the operator.

A. Field of the Invention
This invention relates to a lawn or leaf rake, more particularly to a rake which can be cleaned of leaves which become lodged on the tines during the raking process, by operation of an integral actuating handle without the need for shifting of the operator's hands from the position customarily used in raking. The invention embodies significant improvements in simplicity of design and economy of construction over the existing state of the art of self cleaning rakes.
B. Description of the Prior Art
The art of self-cleaning rakes is documented from the early years of the U.S. Patent Office. However, prior to the most recent patents, "Hand rakes with cleaner", that is prior to U.S. Pat. No. 2,705,394, practically all of the prior patents pertain to garden rakes, those with rigid tines set at right angles to the handle, as differentiated from lawn or broom rakes with flexible tines, which are generally longer, more slender, and in the same general plane as that of the rake handle.
This flexible aspect of the lawn rake tines presents additional and unique circumstances which must be addressed in the design of a self cleaning apparatus for such rakes. These circumstances are further compounded by the predominant radial tine placement or fan shape of most broom rakes and the convention of including a downward bent section at the end of each tine to enhance leaf gathering efficiency.
U.S. Pat. Nos. 3,884,023 and 4,850,185, in addition to one other recent patent for self-cleaning leaf rake, U.S. Pat. No. 4,776,158, address these problems in different ways. All three accept the existence of the downward angled tines, and their arts embody means of dealing with the problem of moving a cleaning device past the angled crooks to reach the very end of the tines in order to effect thorough removal of lodged debris. The instant invention substitutes gradually downward arcing tines of flexible material to enhance the leaf gathering capacity of the rake normally accomplished by the angled ends of most tines. The existence of a gradual arc as opposed to an abrupt crook in the tines allows a simpler cleaning device to be moved along the tines in conjunction with a more direct actuating mechanism.
U.S. Pat. No. 3,884,023 utilizes a cleaning apparatus with actuating linkage (including handle) involving 7 sections or parts comprising at least 10 separate pieces plus fasteners. The apparatus as taught involves an estimated 11 steps in the assembly of the separate pieces, (not including the tine/head subassembly), in addition to special preparation of the handle with a groove to accept the actuating linkage. The construction of the tines and tine head assembly is not taught in full detail.
U.S. Pat. No. 4,776,158 is not of fan or radial shape tine placement, as is the instant invention, having instead a rectangular shape with parallel tines. Such shape tends to be less manageable in some raking circumstances than the more customary fan shape. This patent and U.S. Pat. No. 4,850,185 is otherwise distinguished from the instant invention by the existence of a separate handle for the actuating mechanism requiring release of the down side hand of the operator from its customary position on the rake handle during raking in order to grasp the actuator handle to effect cleaning. With the instant invention, it is not necessary to release or reposition the hand in order to effect cleaning.
U.S. Pat. No. 4,850,185 maintains a stationery cleaning apparatus attached to the handle and effects removal of debris by withdrawing the tine assembly into the cleaner by an actuator linkage attached to a lever handle. Other than the tine/head subassembly, which is not taught in detail, there are 8 pieces plus fasteners in the handle/cleaner and actuating linkage, including the tension spring and tine spreader, which require an estimated 9 assembly steps.
In addition to rakes classified as self cleaning, there are a number of patents that relate to adjustable, collapsible, or retractable rakes. These are of a broom or lawn rake variety and have similarities of mechanism with this and some of the prior self cleaning rake inventions. U.S. Pat. No. 2,149,429 includes a guide bar that moves to the distal end of the rake tines. It does not, however, move to the very end of the crooked tines, as does the mechanism in U.S. Pat. No. 3,884,023 or in the instant rake, making it ineffective in cleaning. The same lack of movement to the very ends of the tines makes the adjustable or collapsible rakes taught in U.S. Pat. Nos. 3,727,389, 3,701,243, and 4,901,515 ineffective in cleaning. In addition, all four of the above incorporate a catch or locking mechanism which must be operated in order to move the spreader bar along the tines. This intervening mechanism is not required in U.S. Pat. No. 3,884,203 or in the instant invention.
U.S. Pat. No. 4,236,742 relates to a retractable rake for use as a portable golf rake or for light raking chores. This rake has a hollow shaft, flared into a rake head at one end to enable a plurality of tines to be retracted into the head. The tines are locked into position in both their open and retracted position, by means of a leaf spring and button arrangement, thus requiring operation of the button in order to retract the tines. This invention, as taught, is not designed for general lawn usage. In addition, the rake does not incorporate as economical a use of materials or ease of manufacture or use as does the instant invention.
The lack of wide spread commercialization of existing designs for a function whose utility is readily recognized by anyone familiar with leaf raking suggests the restrictive manufacturing costs and/or lack of operational ease of use and effectiveness of the existing designs. The instant invention utilizes a cleaning apparatus with actuating linkage (including handle) involving 4 sections or parts comprising 4 separate pieces plus fasteners, in its spring steel version and 2 parts comprising 2 separate pieces, in its molded plastic version. Manufacturing steps are necessarily reduced in both cases, with an estimated 5 steps for the spring steel version and 3 steps for the molded plastic version. Operation of the cleaning mechanism requires only one small movement of the operator's lower arm, pivoting at the elbow, without appreciable disruption of the raking rhythm.

Raking implement with integral tine cleaner