A method for generating a radio frequency signal, wherein a signal to be transmitted is decomposed into a weighted sum of periodic basic signals of different frequencies.

1. Technical Field
The present disclosure relates to the field of wireless communications, and more specifically aims at methods and devices for transmitting radio frequency signals.
2. Discussion of the Related Art
FIG. 1 is a simplified block diagram of a radio frequency signal transceiver device 100 where the processing of the radio frequency signals is of essentially digital nature.
Device 100 comprises an antenna 102 and a digital signal processor 104 (DSP) for example comprising a microprocessor. In the receive direction, the analog signal received by antenna 102 crosses a low-noise amplifier 106 (LNA), and is then directly converted into a digital signal by an analog-to-digital converter 108 (ADC) having its output connected to an input of digital processor 104. The basic signal processing operations, and especially carrier demodulation operations, are digitally carried out by device 104. In the transmit direction, device 104 directly generates a digital signal having the shape of a carrier wave modulated by the data to be transmitted, ready to be transmitted over the network. This signal is simply converted into an analog signal by a digital-to-analog converter 110 (DAC) placed at the output of device 104, and then amplified by a power amplifier 112 (PA), before being transmitted by antenna 102.
This type of device is sometimes called "radio software" since the processing implemented by the receiver and by the transmitter are essentially software in nature.
An advantage of such a device is that it is sufficient to reprogram the software part to make the device compatible with new communication standards (new carrier frequencies, new modulations, etc.).
However, in practice, the use of transceiver devices of purely software nature often may not be considered since this may require extremely fast converters and a digital processor capable of providing considerable computing power. Indeed, present communication standards use carrier frequencies on the order of a few GHz. To be able to process such signals in real time, the bandwidth of the converters and of the calculation device should be at least equal to 10 GHz. Further, to have satisfactory signal quality, a sampling over at least 16 bits should generally be provided. Converters and calculation devices capable of fulfilling such constraints have a considerable power consumption, conventionally ranging from 500 to 1,000 watts. Such a power consumption is incompatible with most network devices, and in particular with portable terminals.
FIG. 2 is a simplified block diagram of a radio frequency signal transceiver device 200, illustrating a solution which has been provided to decrease the constraints on converters and on the signal digital processor.
On the receive chain side, device 200 comprises the same elements as device 100 of FIG. 1, and further comprises a device 202 (SASP—Sampled Analog Signal Processor) for pre-processing the analog signal, arranged between the output of low-noise amplifier 106 and the input of analog-to-digital converter 108. Device 202 is configured to perform an analog pre-processing of the signal, enabling to lower the operating frequency to be able to return to conditions compatible with low power consumption conversion and digital processing devices. Functionally, device 202 selects a frequency envelope (or several envelopes in the case of a multistandard terminal) of the signal received by antenna 102, and lowers the frequency of the signal contained in this envelope. To achieve this, device 202 comprises a sampling circuit capable of delivering analog samples of the input signal, and a processing circuit capable of performing a discrete Fourier transform processing on the signal samples and of delivering first intermediate analog samples. Device 202 further comprises a processing circuit capable of modifying the spectral distribution of the first intermediate samples and of delivering second intermediate analog samples, and a processing circuit capable of performing an inverse discrete Fourier transform on the second intermediate samples and of delivering analog samples of an output signal having a lower frequency than the input signal. Detailed examples of embodiment of device 202 are described in patent application WO 2008/152322 and in article "65 nm CMOS Circuit Design of a Sampled Analog Signal Processor dedicated to RF Applications" by François Rivet et al.
The receive chain of device 200 has the advantage of providing a particularly advantageous rapidity and consumed power saving, especially in mobile telephony applications, while allowing a multistandard use and being easily reconfigurable in case of a modification of a communication standard or in case of the occurrence of a new standard.
On the transmit chain side, device 200 comprises conventional means for modulating a carrier signal with digital data. In the shown example, device 200 can alternately or simultaneously transmit data on two carrier waves P1 and P2 having different frequencies. Carrier signals P1 and P2 are respectively generated by a wave generator 204 and by a wave generator 206. Each wave generator for example comprises a voltage-controlled oscillator controlled by a quartz. A first modulator 205, for example comprising a multiplier, receives on the one hand signal P1 provided by generator 204, and on the other hand a bit train D1 of data to be transmitted provided by digital processor 104. Modulator 205 generates a signal P1′ corresponding to carrier P1 modulated by data D1 to be transmitted. A second modulator 207, for example comprising a multiplier, receives on the one hand signal P2 provided by generator 206, and on the other hand a bit train D2 of data to be transmitted provided by digital processor 104. Modulator 207 generates a signal P2′ corresponding to carrier P2 modulated by data D2 to be transmitted. Signals P1′ and P2′ are added by an adder 208, and the resulting signal is amplified by power amplifier 112, and then emitted by antenna 102.
The transmit chain of device 200 is fast and saves consumed power but has the disadvantage of not being easily reconfigurable in case of a modification of communication standards or in the case where new standards appear.
In the example of FIG. 2, the transmit chain of device 200 further comprises a counter-feedback loop enabling to verify that the signal transmitted by antenna 102 comprises no error. The counter-feedback loop comprises a coupler 210 which samples part of the output signal of power amplifier 112 (signal transmitted by antenna 102). The signal sampled by coupler 210 crosses a low-noise amplifier 212 (LNA) and a demodulation and digitization circuit 214. The digitized signal provided by circuit 214 is sent to digital processor 104, which verifies whether the signal actually coincides with that which was desired to be transmitted.
The provision of the counter-feedback loop, which actually corresponds to a simplified receive chain arranged in parallel with the main receive chain, has the disadvantage of increasing the bulk, the cost, and the power consumption of the device.
Another disadvantage is that circuit 214 generally comprises, for each communication standard capable of being used in transmit mode, a specific analog hardware demodulator. Circuit 214 is thus not easily reconfigurable in the case of a modification of communication standards.

Radio frequency signal transmission method and device

CENTRE NATIONAL DE LA RECHERCHE SCIENTIFIQUE


An antenna mounting system including at least one level indicating device mounted on a mast of the antenna for providing a constant indication of the alignment of the mast relative to a selected orientation of the mast with respect to the horizon.

When installing a terrestrial satellite antenna (TSA), the base and/or mast of the antenna must be accurately aligned relative to the horizon in order to facilitate alignment of the antenna dish mounted on the mast with an orbiting satellite. For example, when the base is mounted on the ground, it is generally preferable to align the mast perpendicular to the horizon. Similarly, if the base is mounted on a vertical structure such as a wall, it is generally preferable for the mast to be oriented parallel to the horizon. In the past, proper installation of a TSA required the services of a skilled installer utilizing a variety of leveling tools to ensure optimal mast orientation. As a result, customers and other novices were often discouraged from installing and aligning such antennas.
Moreover, after the initial installation of a TSA, its orientation may shift due to a variety of causes, such as impact with external objects, strong winds, ground settling, and the like. Therefore, realignment of the mast of a TSA is required from time to time. The novice is then faced with the choice of attempting a complicated alignment procedure, or hiring a costly installer to realign the TSA. More likely, the novice simply lets the orientation of the antenna continue to stray with ever degenerating equipment performance.
In response to the problems faced by the novice installer, several prior devices have been proposed for aiding the alignment of terrestrial satellite antennas (TSA's). For example, U.S. Pat. No. 4,495,706 to Kaminski discloses an alignment gauge which uses a pendulum and compass to indicate the azimuth and elevation angles for orienting an antenna receiving dish. U.S. Pat. No. 4,237,465 to Shibano et al. discloses a map plate having polar coordinates for use with a pendulum styled pointer to adjust the angle of elevation of an antenna dish.
U.S. Pat. No. 5,274,926 to Dillon discloses an instrument for aiming an antenna on the earth's surface toward an antenna on a satellite in geostationary orbit about the Earth. The instrument is said to include a bubble level (not shown) to align a sphere showing a map such that a line through the center of the bubble and the center of the sphere will be aimed at the center of the earth.

Antenna mast with level indicating means


The method and system provide a perpendicular magnetic recording (PMR) head having an air bearing surface (ABS). The PMR head includes first and second poles each having front and back gap regions, a first nonmagnetic insertion layer between the back gap regions, a magnetic pole layer with a front terminating at the ABS, a write gap, shield(s), a second nonmagnetic insertion layer between the second pole back gap region and the shield(s), and coil(s) between the shield(s) and the first pole. The magnetic pole layer terminates between the ABS and the second pole back gap region and has pole angle(s) of at least thirty and not more than fifty degrees. At least part of the magnetic pole layer resides on the second pole. Part of the shield(s) are adjacent to the write gap. Another part of the shield(s) is coupled with the second pole back gap region.

FIG. 1 depicts a conventional perpendicular magnetic recording (PMR) head 10 used in recording a PMR media (not shown). The conventional PMR head 10 is typically used as a write head in a merged head including the PMR head and a read head. The conventional PMR head 10 includes a first pole (P1) 12, P1 pad 14, a first coil 16, insulator 18, a second pole (P2) 20, a magnetic pole layer (main pole) 22, write gap 24, a shield pad 26, a second coil 28, and shield 30. Although depicted as a single shield 30, it is typically composed of two portions 30A and 30B that are formed separately. The PMR head 10 is also depicted with two coils 16 and 28. However, PMR heads having a single coil may also be used.
In order to write data to a PMR media, the coils 16 and 28 are energized. Consequently, the main pole 22 is magnetized and the media written by flux from the pole tip 22A. Based on the direction of current through the coils 16 and 28, the direction of magnetic flux through the main pole 22 changes. Thus, bits having opposing magnetization can be written and the desired data stored on the PMR media. When the conventional PMR head 10 is not writing, no current is driven through the coils 16 and 28. When in this quiescent state, the remanence (zero current) magnetization of the poles 12, 20, and 22 is desired to be approximately zero.
The conventional PMR head 10 is desired to be used at higher recording densities.
In such applications, domain lockup, also termed remanent erasure, is an issue. Domain lockup occurs when the conventional PMR head 10 inadvertently erases data in the PMR media when no current energizes the PMR head 10. This occurs due to a remanent field remaining the main pole 22. Domain lockup is sensitive to the shape anisotropy of the pole tip 22A. A long nose length, NL, or the length of the pole tip 22A from the air-bearing surface (ABS) to the flaring point, is more likely to cause domain lockup. Without lamination of the main pole 22, the nose length typically must be no greater than about twice of the physical track width (perpendicular to the page in FIG. 1). Consequently, most PMR heads 10 have a short nose length. For such PMR heads 10, the primary cause of pole erasure is the magnetic domains in the yoke that may not fully relax after writing. Stated differently, the main pole 22 may not completely demagnetize after writing. Further, the pole tip 22A is sufficiently small that such deviations of the magnetization domains in the pole 22 from a perfectly demagnetized state may produce significant magnetization in the pole tip 22A. As a result, a high remanent field may be present in the PMR media even when no current is driven through the coils 16 and 28. This remanent field may erase data recorded on the PMR media after the head 10 passes over the media for many revolutions. Because it involves this inadvertent erasure, domain lockup is undesirable.
Domain lockup may result not only in inadvertent erasure of data, but also failure of the PMR media. The servo areas (not shown) of the PMR media are usually written at much lower linear density than the areas that store user data. Consequently, the servo areas are more subject to being erased by the remanent field of the PMR head 10. Erasure of servo areas may cause complete drive failure. Therefore, it would be highly desirable for domain lockup to be eliminated.
Various methods have been used to reduce domain lockup. These methods may have significant drawbacks. For example, FIG. 2 depicts a conventional PMR head 10′ in which domain lockup is attempted to be addressed. The PMR head 10′ contains components that are analogous to the conventional PMR head 10. Consequently, the PMR head 10′ is labeled similarly. The PMR head 10′ includes P1 12′, P1 pad 14′, first coil 16′, insulator 18′, a magnetic pole layer (main pole) 22′, write gap 24′, shield pad 26′, second coil 28′, and shield 30′. Note that the PMR head 10′ does not include a second pole analogous to P2 20. In addition, the conventional PMR head 10′ includes a small nonmagnetic layer 34 directly behind the main pole 22′. Although the PMR head 10′ has somewhat reduced domain lockup, it suffers from other drawbacks. In addition, the amount by which domain lockup is reduced may be relatively small.
Accordingly, what is needed is a system and method for reducing domain lockup in a PMR head.

Perpendicular magnetic recording head having nonmagnetic insertion layers


Aspects of configurable logic for hardware bug workaround in integrated circuits may comprise detecting within a chip at least one condition that would likely result in an occurrence of a hardware bug prior to the hardware bug occurring. Upon the detection of the condition, at least one trigger event may be generated within the chip via at least one debug signal, and the trigger event may be utilized to execute workaround code that may prevent the occurrence of the hardware bug. The debug signal may be generated inside the chip and/or outside the chip. The trigger event may be generated by combining a plurality of debug signals within the chip with at least one input or output signal of the chip.

Chips today are very complex devices, often having tens of thousands of transistors, if not hundreds of thousands or millions of transistors, that comprise complicated logic circuits and functionality. Although any bug in a chip or functionality not specified for the original chip can theoretically be worked around by external circuitry, practically speaking, only the simplest of these can be fixed externally because of the large amount of external circuitry that may be required. Therefore, when a hardware bug is found in a chip, great cost may be incurred in redesigning, laying out and manufacturing a new chip. Production of a system, of which the chip may be an integral part, may potentially be delayed by several months while waiting for the new chip, with even more cost incurred resulting from lost sales.
Conventional chips may provide the capability to observe critical internal logic signals on external pins for debugging purposes. These signals may be referred to as debug signals and may be accessible via a debug bus or debug port. A chip designer may also provide external access to at least the most important signals associated with the different hardware functions or blocks in the chip, since access to these signals may also be helpful during debugging.
FIG. 1 is a block diagram of a typical architecture of an exemplary chip with an embedded processor. Referring to FIG. 1, there is shown a processor 102, an interrupt controller block 104, a memory block 106, a control register block 108, a plurality of functions 110, 112, 114, a debug port multiplexer 116, a plurality of debug pins 120 and a plurality of general purpose I/O (GPIO) pins 122. The interrupt controller block 104 may comprise an interrupt controller register (ICR) 124. The functions may comprise Function_1 110, Function_2 112 and Function_n 114.
The processor 102 may be an embedded processor such as an ARM processor. The interrupt controller 104 may comprise suitable logic, circuitry and/or code that may be adapted to provide an interrupt signal INT to the processor 102, which indicates that certain functionality may have been completed or that there may be a problem that may require the attention of the processor 102. The memory 106 may be dynamic random access memory (DRAM), read-only memory (ROM), a non-volatile writeable random access memory (FLASH), and/or other types of memory. The functions 110, 112, 114 may comprise suitable logic, circuitry and/or code that may be adapted to perform specific functionalities needed for this system such as, for example, video processing, audio processing, or data encryption/decryption, as well as general functionalities needed for the chip to operate. The control register 108 may comprise suitable logic, circuitry and/or code that may be adapted to control the output of the debug signals from the debug ports multiplexer 116 to the debug pins 120. The multiplexer 116 may comprise suitable logic, circuitry and/or code that may be adapted to select which of the inputs from the functions 110, 112, 114 may be output to the debug pins 120.
The GPIO pins 122 may be coupled to functionality external to the chip, which may be adapted to provide input signals to the chip and/or receive output signals from the chip. The functionality of the GPIO pins 122 may be controlled by the control register block 108. The ICR 124 may comprise suitable logic, circuitry, and/or code that may be adapted to control occurrence and/or handling of interrupts by the interrupt controller 104.
In the exemplary system of FIG. 1, the processor 102 may execute instructions and/or use data in memory block 106, and may respond to at least one interrupt signal referred to as INT from the interrupt controller 104. The processor 102 may additionally control the functionality of the interrupt controller 104, may read and/or write to the control register block 108 in order to output the correct debug signals from the debug port multiplexer 116, and may control the functionality of the functions 110-114. The interrupt controller 104 may receive interrupt signals from the functions 110-114, and may generate the interrupt signal INT to the processor 102.
If a hardware bug exists in the exemplary system illustrated in FIG. 1, and the bug must be fixed, the only option may be to re-design the chip to remove the bug, and re-spin the chip.
Further limitations and disadvantages of conventional and traditional approaches will become apparent to one of skill in the art, through comparison of such systems with some aspects of the present invention as set forth in the remainder of the present application with reference to the drawings.

Method and system for configurable trigger logic for hardware bug workaround in integrated circuits


In a switching power supply circuit including a primary side rectifying and smoothing circuit, a switching element, a converter transformer, a secondary side rectifying and smoothing circuit, and a switching element controlling section, a first primary side series resonant circuit is formed by a choke coil and a primary side series resonant capacitor, a second primary side series resonant circuit is formed by a leakage inductance occurring in a primary winding and the primary side series resonant capacitor, and a secondary side series resonant circuit is formed by a leakage inductance occurring in a secondary winding and a secondary side series resonant capacitor.

1. Field of the Invention
The present invention relates to a switching power supply circuit.
2. Description of the Related Art
As so-called soft-switching power supply of a resonant type, a current resonant type and a voltage resonant type are widely known. In a present situation, a current resonant converter of a half-bridge coupling system using two switching devices is in wide use because such a current resonant converter is easily put to practical use.
However, the characteristics of a high withstand voltage switching element, for example, are now being improved, and therefore the problem of withstand voltage in putting a voltage resonant converter to practical use is being cleared up. In addition, a voltage resonant converter formed as a single-ended type with one switching element is known to be advantageous as compared with a current resonant forward converter having one switching element in terms of input feedback noise, the noise component of a direct-current output voltage line, and the like.
FIG. 17 shows an example of configuration of a switching power supply circuit having a voltage resonant converter of the single-ended type. Such a voltage resonant converter is referred to as a multiple resonant converter with a series resonant circuit formed by a leakage inductor L2 on a secondary winding side and a secondary side series resonant capacitor C2 on a secondary side.
In the switching power supply circuit shown in FIG. 17, a rectifying and smoothing circuit formed by a bridge rectifier circuit Di and a smoothing capacitor Ci rectifies and smoothes a commercial alternating-current power AC, and thereby generates a direct-current input voltage Ei as a voltage across the smoothing capacitor Ci. Incidentally, a noise filter that is formed by a set of common mode choke coils CMC and two across capacitors CL and removes common mode noise is provided in the line of the commercial alternating-current power AC.
The direct-current input voltage Ei is input as a direct-current input voltage to the voltage resonant converter. As described above, the voltage resonant converter employs a configuration of the single-ended type with one switching device Q1. The voltage resonant converter in this case is an externally excited converter. The switching device Q1 of a MOSFET is switching-driven by an oscillating and driving circuit 2.
A body diode DD1 of the MOSFET is connected in parallel with the switching device Q1. A primary-side parallel resonant capacitor Cr is connected in parallel with the source and drain of the switching device Q1. The primary-side parallel resonant capacitor Cr and a leakage inductor L1 of a primary winding N1 of an isolated converter transformer PIT form a primary side parallel resonant circuit (voltage resonant circuit). This primary side parallel resonant circuit provides a voltage resonant operation as the switching operation of the switching device Q1.
The oscillating and driving circuit 2 applies a gate voltage as a driving signal to the gate of the switching device Q1 to switching-drive the switching device Q1. Thus the switching device Q1 performs switching operation at a switching frequency corresponding to the cycle of the driving signal.
The isolated converter transformer PIT transmits the switching output of the switching device Q1 to the secondary side. As for the structure of the isolated converter transformer PIT, as shown in FIG. 18, for example, the isolated converter transformer PIT has an EE-shaped core formed by combining an E-type core CR1 and an E-type core CR2 of ferrite material with each other. A winding part is divided into a primary side winding part and a secondary side winding part. The primary winding N1 and a secondary winding N2 are wound on a bobbin B covering the central magnetic leg of the EE-shaped core. In addition, a gap G of about 0.8 mm to 1.0 mm is formed in the central magnetic leg of the EE-shaped core of the isolated converter transformer PIT. Thereby a coupling coefficient k=about 0.80 to 0.85 is obtained between the primary side and the secondary side. The coupling coefficient k at this level may be considered to represent loose coupling, and correspondingly a state of saturation is not easily obtained. The value of the coupling coefficient k is a factor in setting leakage inductance (the value of inductance of the leakage inductor L1).
One end of the primary winding N1 of the isolated converter transformer PIT is inserted between the switching element Q1 and the positive electrode terminal of the smoothing capacitor Ci. Thereby, the switching output of the switching element Q1 is transmitted to the primary winding N1. An alternating voltage induced by the primary winding N1 occurs in the secondary winding N2 of the isolated converter transformer PIT.
In this case, a secondary side series resonant capacitor C2 is connected in series with one end of the secondary winding N2. Thus, the leakage inductor L2 of the secondary winding N2 and the capacitance of the secondary side series resonant capacitor C2 form a secondary side series resonant circuit (current resonant circuit).
In addition, a voltage doubler half-wave rectifier circuit is formed by connecting rectifier diodes Do1 and Do2 and a smoothing capacitor Co to the secondary side series resonant circuit as shown in the figure. This voltage doubler half-wave rectifier circuit generates a direct-current output voltage Eo having a level corresponding to twice a secondary winding voltage V3 induced in the secondary winding N2 as a voltage across the smoothing capacitor Co. The direct-current output voltage Eo is supplied to a load, and is also input to a control circuit 1 as a detection voltage for constant-voltage control.
The control circuit 1 inputs a detection output obtained by detecting the level of the direct-current output voltage Eo input as the detection voltage to the oscillating and driving circuit 2. The oscillating and driving circuit 2 outputs a driving signal varied in frequency or the like according to the level of the direct-current output voltage Eo indicated by the detection output input to the oscillating and driving circuit 2. The oscillating and driving circuit 2 thereby controls the switching operation of the switching element Q1 so as to make the direct-current output voltage Eo constant at a predetermined level. Thereby control is performed to stabilize the direct-current output voltage Eo.
FIGS. 19A, 19B, and 19C and FIG. 20 show results of experiments on the power supply circuit having the configuration shown in FIG. 17. In conducting the experiments, principal parts of the power supply circuit of FIG. 17 are set as follows.
The isolated converter transformer PIT has an EER-35 as the core, and has a gap length of 1 mm set for the gap of the central magnetic leg. As for the respective numbers T of turns of the primary winding N1 and the secondary winding N2, N1=39 T and N2=23 T. The level of an induced voltage per turn (T) of the secondary winding N2 is set to 3 V/T. The coupling coefficient k of the isolated converter transformer PIT is set to k=0.81.
The capacitance of the primary side parallel resonant capacitor Cr is selected to be Cr=3900 pF (picofarads), and the capacitance of the secondary side series resonant capacitor C2 is selected to be C2=0.1 pF (microfarads). Accordingly, the primary side parallel resonance frequency fo1=230 kHz (kilohertz) of the primary side parallel resonant circuit and the secondary side series resonance frequency fo2=82 kHz of the secondary side series resonant circuit are set. In this case, relative relation between the primary side parallel resonance frequency fo1 and the secondary side series resonance frequency fo2 can be expressed as fo1≈2.8×fo2.
The rated level of the direct-current output voltage Eo is 135 V. Load power handled by the power supply circuit is in a range of maximum load power Pomax=200 W to minimum load power Pomin=0 W.
FIGS. 19A, 19B, and 19C are waveform charts showing the operations of principal parts in the power supply circuit shown in FIG. 17 on the basis of the switching period of the switching element Q1. FIG. 19A shows a switching voltage V1 applied to the switching element Q1, a switching current IQ1, a primary winding current I2, a secondary winding current I3, a rectified current ID1, and a rectified current ID2 at the maximum load power Pomax=200 W. FIG. 19B shows the switching voltage V1, the switching current IQ1, the primary winding current I2, and the secondary winding current I3 at an intermediate load power Po=120 W. FIG. 19C shows the switching voltage V1 and the switching current IQ1 at the minimum load power Pomin=0 W.
The switching voltage V1 is a voltage obtained across the switching element Q1. The switching voltage V1 is at a zero level during a period TON during which the switching element Q1 is on, and forms a sinusoidal resonant pulse waveform during a period TOFF during which the switching element Q1 is off. The resonant pulse waveform of the switching voltage V1 indicates that the operation of the primary side switching converter is of a voltage resonant type.
The switching current IQ1 flows through the switching element Q1 (and the body diode DD1). The switching current IQ1 flows with waveforms shown in the figures during the period TON, and is at a zero level during the period TOFF.
The primary winding current I2 flowing through the primary winding N1 is obtained by combining a current component flowing as the switching current IQ1 during the period TON with a current flowing through the primary side parallel resonant capacitor Cr during the period TOFF. Though shown in only FIG. 19A, in the operation of the secondary side rectifier circuit, the rectified current ID1 and the rectified current ID2 flowing through the rectifier diodes Do1 and Do2 each flow sinusoidally, as shown in the figure. In this case, the resonant operation of the secondary side series resonant circuit appears in the waveform of the rectified current ID1 more dominantly than in the rectified current ID2.
The secondary winding current I3 flowing through the secondary winding N2 has a waveform obtained by combining the rectified current ID1 and the rectified current ID2 with each other. FIG. 20 shows switching frequency fs, the period TON and the period TOFF of the switching element Q1, and AC-to-DC power conversion efficiency (ηAC→DC) of the power supply circuit shown in FIG. 17 with respect to load variation.
First, looking at AC-to-DC power conversion efficiency (ηAC→DC), high efficiencies of 90% or more are obtained over a wide range of load power Po=50 W to 200 W. The present inventor has previously confirmed by experiment that such a characteristic is obtained when a secondary side series resonant circuit is combined with a voltage resonant converter of a single-ended type.
The switching frequency fs, the period TON, and the period TOFF in FIG. 20 indicate the switching operation of the power supply circuit shown in FIG. 17 as a characteristic of constant-voltage control dealing with load variation. In this case, the switching frequency fs is substantially constant with respect to load variation. On the other hand, the period TON and the period TOFF linearly change in manners opposite to each other as shown in FIG. 20. This indicates that the switching operation is controlled by holding the switching frequency (switching period) substantially constant while the direct-current output voltage Eo is varied, and changing a time ratio between the on period and the off period. Such control can be regarded as PWM (Pulse Width Modulation) control that changes the on period and the off period within one cycle. The power supply circuit shown in FIG. 17 stabilizes the direct-current output voltage Eo by this PWM control.
FIG. 21 schematically shows the constant-voltage control characteristic of the power supply circuit shown in FIG. 17 by relation between the switching frequency fs (kHz) and the direct-current output voltage Eo.
The power supply circuit shown in FIG. 17 has the primary side parallel resonant circuit and the secondary side series resonant circuit. Therefore the power supply circuit shown in FIG. 17 has, in a composite manner, two resonant impedance characteristics, that is, a resonant impedance characteristic corresponding to the primary side parallel resonance frequency fo1 of the primary side parallel resonant circuit and the secondary side series resonance frequency fo2 of the secondary side series resonant circuit. Since the power supply circuit shown in FIG. 17 has the relation fo1≈2.8×fo2, the secondary side series resonance frequency fo2 is lower than the primary side parallel resonance frequency fo1, as shown in FIG. 21.
As for constant-voltage control characteristics with respect to the switching frequency fs under a condition of a constant alternating input voltage VAC, as shown in FIG. 21, constant-voltage control characteristics at the maximum load power Pomax and the minimum load power Pomin under the resonant impedance corresponding to the primary side parallel resonance frequency fo1 of the primary side parallel resonant circuit are represented by a characteristic curve A and a characteristic curve B, respectively, and constant-voltage control characteristics at the maximum load power Pomax and the minimum load power Pomin under the resonant impedance corresponding to the secondary side series resonance frequency fo2 of the secondary side parallel resonant circuit are represented by a characteristic curve C and a characteristic curve D, respectively. When constant-voltage control is to be performed at tg, which is a rated level of the direct-current output voltage Eo, under the characteristics shown in FIG. 21, the variable range (necessary control range) of the switching frequency necessary for the constant-voltage control can be represented as a section denoted by Δfs.
The variable range Δfs shown in FIG. 21 extends from the characteristic curve C at the maximum load power Pomax which characteristic curve corresponds to the secondary side series resonance frequency fo2 of the secondary side series resonant circuit to the characteristic curve B at the minimum load power Pomin which characteristic curve corresponds to the primary side parallel resonance frequency fo1 of the primary side parallel resonant circuit. Crossed between the characteristic curve C at the maximum load power Pomax and the characteristic curve B at the minimum load power Pomin are the characteristic curve D at the minimum load power Pomin which characteristic curve corresponds to the secondary side series resonance frequency fo2 of the secondary side series resonant circuit and the characteristic curve A at the maximum load power Pomax which characteristic curve corresponds to the primary side parallel resonance frequency fo1 of the primary side parallel resonant circuit.
Thus, as the constant-voltage control operation of the power supply circuit shown in FIG. 17, switching-driving control is performed by PWM control that holds the switching frequency fs substantially fixed, and changes a time ratio (period TON/period TOFF) in one switching period. Incidentally, this is indicated by the fact that the period length of one switching period (TOFF+TON) shown at the times of the maximum load power Pomax=200 W, load power=100 W, and the minimum load power Pomin=0 W in FIGS. 19A, 19B, and 19C is substantially constant, while the widths of the period TOFF and the period TON are changed.
Such an operation is obtained by making a transition between a state in which the resonant impedance (capacitive impedance) at the primary side parallel resonance frequency fo1 of the primary side parallel resonant circuit is dominant and a state in which the resonant impedance (inductive impedance) at the secondary side series resonance frequency fo2 of the secondary side parallel resonant circuit is dominant, as resonant impedance characteristics according to load variation in the power supply circuit, in the narrow variable range Δfs of the switching frequency. (See Japanese Patent Laid-Open No. 2000-134925 as Patent Document 1)
The power supply circuit shown in FIG. 17 has the following problems.
The switching current IQ1 at the maximum load power Pomax shown in FIG. 19A is at a zero level until an end of the period TOFF as turn-on timing. When the period TON is reached, the switching current IQ1 operates such that a current of negative polarity first flows through the body diode DD1, and is then inverted to flow from the drain to the source of the switching element Q1. Such an operation indicates that ZVS (Zero Voltage Switching) is performed properly.
On the other hand, the switching current IQ1 at Po=120 W corresponding to an intermediate load which current is shown in FIG. 19B operates so as to flow as noise before the end of the period TOFF as turn-on timing is reached. This operation is an abnormal operation in which ZVS is not properly performed.
That is, it is known that a voltage resonant converter provided with a secondary side series resonant circuit as shown in FIG. 17 causes the abnormal operation in which ZVS is not properly performed at the time of the intermediate load. It is confirmed that the actual power supply circuit shown in FIG. 17 causes such an abnormal operation in a load variation range as a section A shown in FIG. 20, for example.
As described earlier, a voltage resonant converter provided with a secondary side series resonant circuit inherently has, as a tendency thereof, a characteristic of being able to maintain high efficiency in an excellent manner as the load is varied. However, as shown by the switching current IQ1 in FIG. 19B, a considerable peak current flows at the time of turning on the switching element Q1, thereby inviting an increase in switching loss and constituting a factor in lowering power conversion efficiency.
At any rate, an abnormal operation as described above causes a shift in phase-gain characteristics of a constant-voltage control circuit system, for example, thus resulting in a switching operation in a state of abnormal oscillation. Thus, in the present situation, it is generally recognized to be difficult to put the power supply circuit of FIG. 17 to practical use in actuality.


A control device for an automatic transmission capable of operating in an automatic shift mode in which a gear ratio is automatically selected based on driving conditions of a vehicle, and also capable of operating in a manual shift mode in which the gear ratio is changed based on an upshift command or a downshift command by manual operation of a manual operation device, the control device includes a manual shift control unit that changes and sets the gear ratio after downshift depending on road conditions when the downshift command has been operated by the manual operation device in the manual shift mode, and performs the downshift to the gear ratio that has been changed and set.

The present invention relates to a control device and method for an automatic transmission.
An automatic transmission mounted on a vehicle or the like is structured so as to be capable of driving the vehicle at an appropriate gear ratio without a shift operation by a driver, by automatically judging a gear ratio (shift speed) based on an accelerator opening and a vehicle speed particularly during forward driving. However, in recent years, in order to meet such a demand for sports driving and a delicate selection of engine braking for example, automatic transmissions have been proposed that also allow a so-called manual shift operation in which the driver can select a gear ratio (shift speed) (see, for example, Japanese Patent Application Publication No. JP-A-10-324169).
In recent years, since improvements in vehicle fuel consumption and so on are required, multi-speed automatic transmissions, such as those achieving six or more forward speeds, are becoming prevailing in the category of stepped automatic transmissions. Moreover, automatic transmissions such as belt type CVTs, that steplessly change speed, are also structured so as to be capable of selecting simulated shift speeds by providing finely spaced gear ratios (for example, 6 stages or more).

Control device for automatic transmission


A tilting converter comprising a container (2), defining a first longitudinal axis X, having a bottom (2′); a support ring (3), coaxial to the container (2) and distanced from said container, provided with two diametrically opposite supporting pins (6), defining a second axis Y orthogonal to the first axis X, adapted to allow a rotation of the converter about said second axis Y; a suspension system, connecting said container (2) to said support ring (3), comprising groups (12) of first suspension devices (7), said groups (12) being arranged substantially equidistant to each other along a cylindrical side surface coaxial to the first axis X, in a position between the support ring (3) and the bottom (2′); each of said first suspension devices (7) being provided with a plurality of longitudinal elastic elements, each longitudinal elastic element being arranged alongside the next so as to define a laying plane, and a gap (15, 15′) is provided between one longitudinal elastic element and the next.

Field of the Invention
The present invention relates to a tilting oxygen converter provided with a suspension system of the container of the converter connecting said container to a support ring.
State of the Art
The main object of an oxygen converter is to convert the cast iron produced in the furnace into raw liquid steel, which may be subsequently refined in the secondary steel production department.
The main functions of the oxygen converter, also known as B.O.F. (Basic Oxygen Furnace), are to decarburize and remove phosphorous from the cast iron and to optimize the temperature of the steel so that further treatments may be carried out before casting with minimum heating and cooling of the steel.
The exothermal oxidation reactions which are generated in the converter produce a great deal of thermal energy, more than that needed to reach the established temperature of the steel. This extra heat is used to melt scrap and/or additions of ferrous material. The B.O.F. container, being substantially a furnace, is also subject to high thermal expansions.
The converter consists of a container, defining the reactor and having a substantially cylindrical shape, supported by a support ring, surrounding the container and appropriately distanced therefrom, provided with two diametrically opposite supporting pins, all supported by two supports anchored to the ground. The rotating control of the container is fitted onto one of the two supporting pins.
An example of oxygen converter of the prior art is described in WO2008/092488A1. The container of such a converter is supported by means of an outer support ring and a plurality of suspension elements connecting said container to said support ring, the suspension elements being restrained at a first end to the container and at a second end to the support ring.
First suspension elements are formed by packs of lamellae which are arranged in a distributed manner under the support ring, considering the converter in its upright position with the loading mouth facing upwards. One end of the packs of lamellae is fixed onto a lower surface of the support ring; the other end of the packs of lamellae is fixed to a lower frustoconical portion of the container. In such a configuration, the packs of lamellae are arranged inclined by an angle of approximately 20-25° with respect to a respective vertical plane containing the longitudinal axis defined by the converter.
Second suspension elements are formed instead by pendulum rods, which are hinged at one end to the support ring and at the other end to the container. Considering the conveyor in its upright position with the loading mouth facing upwards, such pendulum rods are arranged horizontally at the two supporting pins above and under the support ring.
Disadvantageously, in the packs of lamellae made and arranged in the manner described above, excessively concentrated loads are created at the fastening points of the lamellae to the support ring and to the structure of the container of the converter. In particular, such packs of lamellae so configured are transversally rigid and do not allow the flexion and torsion of the lamellae themselves, and thus the redistribution of the stresses when the converter is turned by 90° from its vertical position to carry out the step of tapping.
Furthermore, the different thermal expansion of the container and of the support ring determines high concentrated loads in the fastening points of the lamellae due to the fact that the attachment of the lamellae themselves on the container tends to shift because it expands more, while the attachment of the lamellae on the support ring tends to remain in position because it expands less.
Another example of oxygen converter of the prior art, similar to that disclosed in WO2008/092488A1 (FIG. 1a), is described in US2012/0223107A1 (FIG. 3).
A further example of oxygen converter of the prior art is disclosed in GB2492735. In this converter the converter vessel or container is secured to the trunnion ring by means of a plurality of connecting elements, and each connecting element is rigidly secured by means of its opposing ends to the underside of the trunnion ring at one end and to the converter vessel at the other end. Each connecting element comprises precisely one lamella which is shielded from the converter vessel by means of a protective element that is connected by means of its opposing ends both to the converter vessel and to the trunnion ring, but only one of these ends is rigidly connected either to the trunnion ring or to the converter vessel. Therefore said protective element does not itself have any bearing function. Each lamella defines an own laying plane that is clearly different from the laying plane of the adjacent lamellae, being each lamella distanced from the adjacent ones and fixed in an area of the converter angularly distanced from the fixing areas of the adjacent lamellae. Moreover each lamella has respective distinct fastening supports. Even if this solution allows to slightly reduce the concentrated loads at the fastening points of the lamellae, it is still felt the need to make a tilting converter which allows to better overcome the aforesaid drawbacks.
The centering between container and support ring is also important to suitably allow deformations or thermal expansions of the container caused by the high temperatures reached during the refining process.


A drive circuit (18) is provided that utilizes a power supply (30) to generate an output voltage that effects a precise displacement in a physical device. The drive circuit (18) includes storage circuitry (26) having a capacitive load (54) for storing the output voltage, a supply switch (34) connected the to the storage circuitry (26), the power supply (30), and control circuitry (22) that controls the supply switch (34) in response to an input signal. The control circuitry (22) controls the supply switch (34) in response to the input signal such that a supply voltage (31) from the power supply (30) is intermittently applied to the capacitive load (54) of the storage circuit (26) for generation of the output voltage that is used to effect a precise displacement in a physical device.

1. Field of the Invention
The present invention relates generally to devices which control the charge stored at a capacitive load by controlling the application of a supply voltage to the load and, more particularly, to a minimum power piezoelectric actuator circuit that quickly and efficiently charges an actuator output voltage in response to an input signal.
2. Discussion of Related Art
Piezoelectric actuators are often used to effect small, but well-controlled displacements of physical objects, such as optical components and machining tools. These mechanical displacements must be very precise and controlled, and are typically on the order of 1-100 microns. In operation, the actuator functions based on an applied voltage. The actuator itself is a capacitor used to store a potential which drives the mechanical displacement. Since capacitors draw no DC current, the theoretical power necessary for actuator operation is very low.
Despite the low power requirements of the actuator, the conventional circuits used to drive piezoelectric actuators unnecessarily dissipate significant power in a number of ways. First, the conventional systems often use power supplies having a voltage higher than is necessary to charge the actuator. These high-voltage power supplies are often separate from other subsystems and dedicated specifically to driving the piezoelectric actuator, resulting in waste and increased manufacturing costs. The conventional piezoelectric actuator systems also typically use resistive elements in portions of the system with significant current flows, thereby causing unnecessary dissipation of power. Furthermore, the conventional circuits used to drive piezoelectric actuators often draw current even when it is not needed to charge the actuator.
In addition to being very precise and accurate, the mechanical displacements effected by a piezoelectric actuator must be well-controlled in the sense that they are properly timed. A piezoelectric actuator must therefore be able to respond quickly to an input drive signal used to control the charging of the actuator. However, many conventional load charging circuits are unable to provide the fast response times required for piezoelectric actuation.
In view of the foregoing, it is an object of the present invention to provide a piezoelectric drive circuit that quickly responds to an input signal and that employs a low voltage power supply and low impedance and reactive components to avoid unnecessary power dissipation. Power is also conserved by drawing current only when necessary to charge the piezoelectric actuator. It is also an object of the present invention that these power saving advantages allow a piezoelectric drive circuit as taught herein to be incorporated into small, portable devices which run on battery power.


A process for producing fluoride crystals, in particular calcium fluoride crystals, having high radiation resistance to ultraviolet radiation, which includes: provisioning of a crystal powder (6) containing alkali metal fluoride or alkaline earth metal fluoride to form a raw crystal mass, melting of the raw crystal mass in a crystal growing unit (11) and solidifiying of the molten raw crystal mass by cooling. In the process, an ammonium salt (7) of a complex fluoro acid and an aliphatic alcohol (8) are added to the crystal powder (6) or to the raw crystal mass, to decrease oxidic impurities. A fluoride crystal produced by the process and also an optical component formed from such a fluoride crystal are also disclosed.

The invention relates to a process for producing fluoride crystals, in particular calcium fluoride crystals, having high radiation resistance to ultraviolet radiation, which comprises: provisioning of a crystal powder containing alkali metal fluoride or alkaline earth metal fluoride to form a raw crystal mass, melting of the raw crystal mass in a crystal growing unit and solidifying of the molten raw crystal mass by cooling. The invention also relates to a fluoride crystal produced by this process and also an optical component produced from such a fluoride crystal.
Large format, highly homogeneous and ideally scattering-free single crystals (or ingots) of alkali metal fluorides or alkaline earth metal fluorides, in particular CaF2, are typically produced in a one-stage or multistage process from a raw crystal mass in a growing furnace. Such crystals, in particular calcium fluoride crystals, have a high transmission at wavelengths in the (near) UV range, e.g. at 193 nm. These can therefore be used as transmissive optical elements, e.g. as lenses, prisms or the like in projection or illumination units or as beamforming elements in laser systems for microlithography. Apart from single crystals, it may also be possible to use crystalline fluoride materials which consist of a plurality of phases for producing optical components, as is described, for example, in GB 1 104 182.
Particularly when the crystal material is used in lens, illumination and laser system positions having a high radiation exposure, e.g. at energy densities greater than 50 mJ/cm2, there is the problem that the crystal material is damaged by the hard UV radiation used, which is, for example, produced by an excimer laser, if the crystal material has structural crystal defects (flaws). Such structural crystal defects are defects in the crystal structure which can be caused, inter alia, by incorporation of foreign atoms or foreign particles during crystallization. Under intensive UV irradiation, color centers which irreversibly reduce the transmission of the crystal with increasing radiation dose are formed at the flaws. In the case of high-purity crystals, the absorption coefficient tends to a saturation value with increasing radiation dose; this saturation value can serve as a characteristic measure of the laser resistance of the fluorite material, as is described in DE 10 2009 030 203 A1 of the applicant.
It is basically known that structural crystal defects can be influenced by an appropriately controlled growth and heat treatment process. Thus, for example, EP 0 939 147 A2 discloses a process in which a single crystal of calcium fluoride having improved optical properties is produced by carrying out a heat treatment process having a specific temperature profile on this.
In particular, it is also known that even tiny amounts of oxygen (in the ppm range), which are introduced as a result of the technology into the raw crystal mass in a growing apparatus, cause undesirable chemical secondary reactions in the melt of the crystal starting material or the raw crystal mass. Such secondary reactions lead to formation of crystal defects by incorporation of foreign atoms or ions into the crystal lattice and also to precipitation of microscopically small particles of metal oxides, e.g. CaO, PbO, or MgO, and thus lead to impairment of the optical properties of the crystal due to absorption or scattering.
The formation of such oxygen defect sites can be influenced by use of an appropriate protective gas atmosphere and use of scavengers. A scavenger is a purifying agent, i.e. a substance which serves to remove impurities from the raw crystal material. In the case of fluoride-containing crystals, an oxygen scavenger such as PbF2 is usually used for this purpose. It has been found that the addition of PbF2 brings about chemical reactions which lead to substantial removal of oxides, hydroxides or water. Here, for example, substances which are volatile even several hundred degrees Celsius below the melting point of calcium fluoride, e.g. PbO etc., are formed in a temperature range from 600° C. to 900° C.
Here, the impurities vaporize or sublime into the atmosphere surrounding the crystal and can there be removed from the plant (the growing furnace). Such a procedure is described, for example, in K. Th. Wilke, Kristallzüchtung, Deutscher Verlag der Wissenschaft, Berlin 1988, page 630 ff. Impurities in the crystal which, for example, deposit along small-angle grain boundaries can also be minimized by such purification processes.
However, it is known from, for example, DE 101 42 651 A1 that the use of a crystalline scavenger in the form of PbF2 can lead, from a laser resistance point of view, to not insignificant residues of foreign metal oxide (PbO) or fluoride (PbF2) in the crystal and these cannot be completely removed by distillation/sublimation.
To solve this problem, it has been proposed in, for example, the article "Czochralski growth of VV-grade CaF2 single crystals using ZnF2 additive as a scavenger", J. Chryst. Growth 222, (2001), pages 243-248, that another metal fluoride, namely ZnF2, be used instead of PbF2 as scavenger. EP 0 869 203 A2, too, describes the production of a fluoride crystal, in which a scavenger in the form of ZnF2 is used in a plurality of process steps. However, ZnF2 and also the ZnO formed therefrom vaporize at higher temperatures than PbF2 and PbO. In addition, Zn residues also cause absorptions in the UV wavelength range from 150 nm to about 170 nm and influence the laser resistance at 193 nm.
EP 0 919 646 A1 describes the production of calcium fluoride crystals using scavengers, e.g. Teflon, and/or metal fluorides, e.g. PbF2, CoF2 and MnF2, but the above-described problems can also occur in the case of these scavenger materials. WO 01/025001 A1 also describes a process for producing fluoride crystals, in which use is made of a fluorinating material which is provided in the form of a solid, e.g. PbF2, NH4F, NH4F. HF or polytetrafluoroethylene (Teflon), or in the form of a gas, e.g. in the form of HF, F2 or NF3.
DE 101 42 651 A1 proposes using a reactive gas which comprises at least one substance containing reactive fluorine in addition to the crystalline scavenger in order to avoid problems with scavenger residues. This is said to be able to minimize the proportion of residues of the scavenger, e.g. PbO, in the crystal. Thus, for example, it is proposed that CF4, which reacts according to the equation 2PbO+CF4→2PbF2+CO2, be used as reactive gas. PbF2 reformed in this reaction can react again with oxygen-containing compounds or, should a reaction partner no longer be present, the fact that it has a somewhat lower boiling point than PbO and is therefore easier to remove can be exploited. The second compound formed, viz. CO2, is gaseous and can easily be removed under reduced pressure.
However, the process described in DE 101 42 651 A1 also has a number of disadvantages: since PbF2 has, at atmospheric pressure, a boiling point which is only about 200° C. below that of PbO, the PbF2 can remain and thus reduce the laser resistance. Although the vapor pressure of PbO (vapor pressure at 1500° C.-53.3 mbar (40 torr)) is a factor of about 10 lower than that of PbF2 (vapor pressure at 1500° C.-600 mbar (450 torr)), it cannot be assumed that all of the lead (Pb) can be removed from the plant, since lead compounds can deposit, for example, in graphite linings or on difficult-to-access places in the growing furnace. Furthermore, a secondary reaction in the form of PbO+CF4→PbF4+CO (redox reaction) can occur and carbon is incorporated in the form of graphite in the crystal (as black inclusions) as a result of the Boudouard equilibrium (CO2+C2CO), which likewise reduces the transmission and impairs other optical properties of the crystal.

Production of radiation-resistant fluoride crystals, in particular calcium fluoride crystals


A transimpedance amplifier includes a current regulator having a first current mirror of a first conduction type and a first current mirror of a second conduction type. The first current mirror stage of a first conduction type receives a reference current. The first current mirror stage of a second conduction type is connected to the first current mirror stage of the first conduction type, and receives an output current from the first current mirror stage of the first conduction type, and to generate a current to be used as a current source by a transimpedance amplifier. Each of the current mirror stages includes a first transistor and a second transistor of the same conduction type having their gate terminals connected wherein the first transistor receives an input current and the second transistor provides an output current that is a factor of the received input current.

Transimpedance amplifiers (TIA) are commonly used for providing a voltage output signal proportional to a current signal and are normally implemented by providing a feedback resistor across the input and output nodes of an operational amplifier. TIAs are often known as current to voltage converters. One use of a TIA is to convert an input current signal into an output voltage signal.
FIG. 1 shows a transimpedance amplifier 100 implemented with MOSFETs. The TIA comprises an n-channel MOSFET (NMOS) 110 and a p-channel MOSFET (PMOS) 120 arranged such that the gate terminals 112, 122 of the MOSFETs are tied to an input vin, and the drain terminals 114, 124 of the MOSFETs are tied to an output vout. The source terminal 126 of the PMOS is connected to a current source 130 and a discharge capacitor 140. A feedback resistor 150 is connected across the TIA 100. With this arrangement, the gain Gm of the amplifier is controlled by the current IS provided to the TIA by the current source 130 where Gm∝√{square root over (IS)}.
In RF receiver modules, TIAs generally function as low-noise pre-amplifiers, which largely determine the overall performance of the module. In the past, because of the wide bandwidth and the high gain necessary for sensitive data links, TIAs have been implemented with bipolar and gallium arsenide (GaAs) metal-semiconductor field effect transistors (MESFET). These implementations result in high speed devices, but they can be costly and the production processes lack high-yield manufacturability. Recently, metal-oxide-semiconductor (MOS) technology has become popular for the design of TIAs because of its low cost and high-yield manufacturability. However, using a single MOS gain stage fails to provide enough gain for multi-gigabit operation, because MOS transistors have a lower transconductance than bipolar transistors. Hence, successful high-speed MOS implementations have relied on multiple gain stages.
At present, the current supplied to a TIA is fixed at a maximum value that ensures the highest gain Gm(max) from the TIA. However, this gain is not always required and so during normal operation of the TIA, this high current is mostly taken up by the load. This unnecessarily increases the power consumption of the TIA when Gm(max) is not required.
In view of the above, there remains a need to provide a TIA which is driven by a current source that is adaptive to the requirements of the TIA circuit such that a high current source is only provided to a component of the circuit (when Gm(max) is required) while supplying the remaining components of the circuit with lower current sources so as to minimise the overall power consumption.

Current regulated transimpedance amplifiers


A package for a medical device having an elongated shaft and a proximal luer fitting, the package includes a one-piece body formed as a single structure. The one-piece body includes at least a spiral casing formed therein, the spiral casing defining a spiral lumen configured to receive the elongated shaft of the medical device. The package includes a luer retaining portion at one end of the spiral casing with barrier film sealed to the luer retaining portion to maintain the sterility of the medical device. The package further including a cap placed over the luer retaining portion and barrier film to protect the integrity of the barrier film and prevent perforation of the barrier film which would compromise the sterility of the medical device.

Catheters may be inserted into a patient's vasculature and deployed at various locations within the patient for a wide variety of purposes and medical procedures. For example, one type of catheter is used in percutaneous catheter intervention (PCI) for the treatment of a vascular constriction generally known as a stenosis. In this instance, the catheter has a distally mounted balloon that can be placed, in a deflated or unexpanded condition, within the stenosis, and then inflated or expanded to dilate the narrowed lumen of a blood vessel. This type of balloon dilation therapy is generally referred to as percutaneous transluminal angioplasty (PTA). When the treatment is more specifically intended for vessels of the heart, the process is known as percutaneous transluminal coronary angioplasty (PTCA). In other PCI procedures, a stent is expanded into contact with the vessel wall to prevent narrowing or restenosis of the artery.
For example, FIGS. 1 and 2 illustrate the deployment of a PTCA treatment catheter 16 within a patient's vasculature. To treat small diameter vessels remote from an entry point into a patient, a guiding catheter 10 may be used to span the distance. Guiding catheter 10 is typically inserted into a large artery 12 near the patient's groin and is then advanced towards the heart H to the entry opening or ostium of a diseased coronary artery. Guiding catheter 10 provides a conduit through which catheters and guidewires, such as treatment catheter 16 and a guidewire 18, can be passed from outside the patient to the vessel being treated. Treatment catheter 16 generally includes a flexible elongated tubular shaft and a luer fitting 14. For certain interventional procedures, treatment catheter 16 may include a dilatation balloon and/or a stent disposed along a distal portion thereof.
Referring now to FIGS. 3-4, catheters are commonly packaged and stored in a packaging hoop 11 as shown in FIG. 3 in accordance with the teachings of the prior art. Packaging hoop 11 consists of a coiled tube 13 having a proximal opening 17 through which treatment catheter 16 is inserted. Several clips 15 are coupled to tube 13 to maintain the tube in the coiled configuration. The overall diameter of packaging hoop 11 is selected to be as small as possible without imparting a lasting curve shape to the generally straight medical device during the shelf life thereof. An overly small diameter hoop 11 may also create excessive friction forces and thereby possibly cause product damage during loading or unloading of the medical device. Luer fitting 14, located at the proximal end of treatment catheter 16, has an area 19 that fits into proximal opening 17 of tube 13, as shown in FIG. 4, in order to secure treatment catheter 16 in packaging hoop 11 until it is removed for clinical use. One potential issue with utilizing clips 15 for holding tube 13 in the coiled configuration is that the packaging may not stay flat because clips 15 cannot grip tube 13 with much force without undesirably pinching or kinking the tubing and thereby restricting the loading and unloading of treatment catheter 16. Furthermore, tube 13 may separate from clips 15 during distribution.
In addition to coiled tube 13 and clips 15, catheter packaging typically requires additional parts or components such as a sealed pouch that completely envelopes the entire hoop 11. In addition, a sealed pouch further requires that the pouch be fully contained within a box or carton in order to protect the pouch. One potential issue with this packaging configuration is that the pouch can move inside the box during transport. This creates the possibility for the tube 13 or clips 15 to pierce the pouch and compromise the sterility of the catheter. If this were to occur, a breach of the pouch may be undetectable by the operator.
Accordingly, there remains a need in the art for improved packaging that overcomes these disadvantages of currently available catheter packaging, as well as a need to reduce the number of packaging components and the cost of packaging.


Measurements of pressure relative to atmospheric reference pressure are enabled by a sealed enclosure, which in one aspect is a variable volume enclosure including a movable element responsive to atmospheric pressure, to at least partially equalize pressure internal to the variable volume enclosure to the pressure external to the variable volume enclosure, such that a pressure sensor pneumatically connected to the internal volume can use its pressure a reference in measuring atmospheric pressure. The variable volume enclosure may house electronics and/or electrical terminations for pressure sensing, or the variable volume enclosure may be a sub-enclosure connected pneumatically to the pressure sensor, with another enclosure housing electronics or electrical terminations. In another embodiment, the sealed enclosure includes an absolute pressure sensor for sensing atmospheric pressure for use as a reference, and need not have variable volume.

Many applications require the measurement of pressure at a remote location, but some of these applications are more difficult to implement due to atmospheric conditions where the measurement is made. For example, pressure measurements are needed in sewers to control and manage water and sewage flow, but the environment in the sewer is moist and damaging to electronics, as well as potentially explosive. Although the pressure measuring electronics can be placed inside a housing, in order to measure pressure relative to atmospheric pressure, it has been necessary to include a vent to atmosphere to permit detection of atmospheric pressure in order to make the measurement, which means that moisture from the surrounding atmosphere can enter the housing and damage the electronics, as well as posing a threat of ignition of explosive gasses in the environment.
In the prior art, electronics for a pressure sensor have been positioned within a housing, and a vent tube has been pneumatically connected from the pressure sensor to the housing, to carry atmospheric pressure to the pressure transducer so that measured pressures may be referenced to atmospheric pressure. In this approach, to control moisture, a desiccant has typically been installed at the housing end of the pressure transducer's vent tube, to absorb moisture and prevent moisture from entering the vent tube. While this solution addresses the infiltration of moisture, it requires regular replacement of the desiccant when its moisture absorbing capacity is diminished. Furthermore, because the vent tube carries pressure from inside the housing to the sensor, if the housing is sealed, the housing pressure, not atmospheric pressure, is delivered to the sensor, causing inaccuracy. If, however, the housing is unsealed to allow its internal pressure to equalize with atmosphere, the venting exposes electronics and electrical connections in the housing to outside atmosphere, which can present problems to the electronics in an environment containing moisture, and more severe problems when the environment contains other more corrosive gasses. Furthermore, if flammable gasses can infiltrate the housing the electronics can pose a risk of igniting those gasses.
In other prior art, a pressure transducers for sensing external air pressure have been mounted in a bellows or bladder inside of a generally unsealed electrical control housing. While this solution isolates the pressure transducer from moisture, it does not prevent the infiltration of moisture to other electronic components, nor does it address the problems inherent with atmospheric exposure of electronics and electrical connections in potentially explosive or corrosive environments.

Variable volume enclosure to terminate vented pressure transducers


The present invention relates to an electric motor and an electric power steering apparatus including the same. According to an embodiment of the present invention, problems such as a malfunction of an electric motor, damage to a component of the electric motor, and lowering of an output of the electric motor can be solved by preventing penetration of foreign substances into the electric motor, so that a steering stability of a vehicle can be improved.

1. Field of the Invention
The present invention relates to an electric motor and an electric power steering apparatus including the same, and more particularly to an electric motor and an electric power steering apparatus including the same, by which problems such as a malfunction of an electric motor, damage to a component of the electric motor, and lowering of an output of the electric motor can be solved by preventing penetration of foreign substances into the electric motor, so that a steering stability of a vehicle can be improved.
2. Description of the Prior Art
FIG. 1 is a schematic diagram of a rack-driven power steering apparatus according to the related art.
As illustrated in FIG. 1, the rack-driven power steering apparatus, according to the related art, includes a driving unit 120 controlled by an electronic control unit (ECU) configured to determine a steering degree of a steering wheel 113 through a torque sensor 111 mounted to an upper end of a steering column 110, for generating power, a rack bar 130 for moving tie rods 133 connected to wheels 131 according to the steering degree of the steering wheel 113, and a driven unit 140 for receiving a rotational force of the driving unit 120, and converting the received rotational force into an axial movement force to transmit the axial movement force to the rack bar 130.
Here, the driving unit 120 includes an electric motor 121 controlled by the electronic control unit (ECU), a driving pulley 125 coupled to a shaft 123 of the electric motor 121, and a driving belt 127 wound on the driving pulley 125, and the driven unit 140 include a ball nut 141 for supporting the rack bar 130 inside a rack housing surrounding the rack bar 130, and a driven pulley 143 assembled on an outer peripheral surface of the ball nut 141.
The driven pulley 143 coupled to the driving pulley 125 connected to the shaft 123 of the electric motor 121 and the driven pulley 143 coupled to the ball nut 141 are disposed in parallel, and the driving belt 127 is mounted on the driving pulley 125 and the driven pulley 143 to transmit the rotational force of the electric motor 121 to the rack bar 130. The rack bar 130 is moved leftwards and rightwards by an operation of the ball nut 141 to generate an assistant force.
However, in the rack-driven power steering apparatus according to the related art, foreign substances are introduced between the housing and the shaft of the electric motor which causes electric motor malfunction, damage to a component of the electric motor, and lowers the output of the electric motor, and accordingly, the steering stability of a vehicle cannot be secured.

Motor and electric power steering apparatus including the same


The present invention teaches a method and system of encrypting a set of data having a test bit. The method comprises the step of generating a first and a second data string from the set of data. Once generated, the test bit is tested, and should it comprises a first value, the first data string is shifted in a first direction to transfer a carry over bit from a carry over register into the first data string while transferring a first bit of the first data string to said carry over register. This shifting step is followed by the step of shifting the second data string in a second direction to transfer a first bit of the second data string to the carry over register while transferring the carry over bit from the carry over register into the second data string. However, if the test bit comprises a second value, a first logic operation is executed on the first data string with a first number, and a second logic operation is executed on the second data string with a second number. Thereafter, the first data string is shifted in the second direction such that said carry over bit from said carry over register is transferred onto the first data string while the first bit of the first data string is transferred to the carry over register. Subsequently, the second data string is shifted in the first direction such that the carry over bit from the carry over register is transferred onto the second data string while the first bit of the second data string is transferred to the carry over register.

In the automotive industry, remote keyless entry ("RKE") systems have become standard equipment on many new vehicles. Comprising a receiver within the car and a number of fob transmitters for transmitting a message to the receiver, RKE systems enable users to control several vehicle functions remotely, such as the door locks and trunk, for example.
In providing remote control to vehicle functions, a problem arises as to restricting remote access to the automobile's owners and other authorized users. To prevent unauthorized access, an identification system is incorporated with a security code or codes within both the fob transmitter and receiver. The receiver receives a transmitted signal having a command and and identification or security code and compares the received code with the security code stored in its memory. If the receiver determines the received security code to match the stored code, the command is initiated for. execution.
As the demand for RKE systems has evolved in the marketplace, greater emphasis has been placed on increased security, reliability and flexibility. With the development of sophisticated electronics, presently, a transmitted message may be decoded and retransmitted at a later time. This is in part because in these known systems the transmitted message, does not change between transmissions.
As such, one area of focus has been the incorporation of encryption techniques to decrease the likelihood of unauthorized reception and retransmission of the originally transmitted signal comprising both the command and security code. Therefore, there is a demand for an encryption method designed specifically for RKE systems. Moreover, a need further exists for a RKE system which encrypts the transmitted message.

Remote keyless entry encryption algorithm


A touch panel includes a plurality of first touch electrode patterns including a plurality of first touch units and a connection unit connecting two adjacent first touch units. The touch panel further includes a plurality of second touch electrode patterns insulated from and intersected with the first touch electrode patterns, and the second touch electrode patterns includes a plurality of second touch units and a bridge unit connecting two adjacent second touch units, and a plurality of bridge units are arranged along a preset direction and an intersection angle between the preset direction and the second direction is greater than 0° and less than 90°.

In the prior art, a flexible screen is often used in wearable devices due to low power consumption and bendability. In order to satisfy an interaction between a user and the flexible screen, a touch panel is usually arranged by overlapping on the flexible display screen in order to receive touch operation instructions such as clicking and sliding on the touch screen.
The existing touch panel often has a first direction electrode 11 and a second direction electrode 12. As shown in FIG. 1a, wherein the enlarged view of the specific structure is shown in FIG. 1b, the first direction and the second direction are perpendicular to each other. The first direction electrode 11 is continuously arranged, and the second direction electrode 12 is arranged at intervals, and a bridge unit 13 is arranged between the adjacent electrodes 121 of the second direction electrode 12 to make the second direction electrode 12 be connected to each other and be insulated from the first direction electrode 11. In the actual application process, an electronic device often has a habitual bending direction, such as vertical bending or horizontal bending. Under the action of external forces, the touch panel tends to bend accordingly. When bending, the bridge unit 13 is subjected to bending stress.