A single port multi-resonator acoustic resonator device (200, 300, 400, 490) possesses an input impedance that exhibits precisely designed electrical resonances. The device contains at least three parts: a transducer/resonator (201, 301, 401. 491) used both to interface to an external electrical circuit and to transform electrical energy into mechanical (i.e. acoustic) vibrations (and vice versa), and also function as a resonator; a mechanical (i.e. acoustic) resonator (203, 303, 460, 480) and an acoustic coupler (202, 302, 404, 494) that controls the acoustic interaction between the transducer/resonator and the mechanical resonator.

Communications systems have need for components, such as resonators and filters, that control frequency generation or that limit the range of frequencies used in the systems. Resonators are not only formed of electrical inductor and capacitor circuits, but can also be electromechanical in nature, such as quartz crystals, surface acoustic wave devices, or thin film bulk acoustic resonators. Frequency filters also may be based upon that core resonator technology. Resonators exhibiting multiple resonances at nearby frequencies within a single structure would be particularly useful in frequency filters to produce multiple pass-bands and/or stop-bands of frequency. Both the resonators and the filters formed with those resonators may take advantage of the piezoelectric type of resonator, a form of electromechanical resonator.
In the simplest form, a piezoelectric electromechanical resonator is composed of a piezoelectric plate sandwiched between a pair of electrodes formed in a unitary assembly. Resonator 1, illustrated in FIG. 1a, representative of prior art to which reference is made, is composed of a piezoelectric plate 2 that possesses sufficiently aligned and smooth surfaces 3 and 4 to which metal plates or electrodes 5 and 6 are respectively attached. An electromechanical thickness-mode resonance is established by the piezoelectric transduction process, wherein electrodes 5 and 6 are electrically driven by a signal of such frequency that an acoustical standing wave is established across the thickness dimension of the piezoelectric plate, perpendicular to the plane of the electrodes. The acoustic deformation produced in the foregoing transduction propagates in a direction that is substantially normal to the piezoelectric plate and the electrodes, and is referred to herein as a bulk acoustic wave or simply a bulk wave. The bulk wave may be an acoustic shear wave which exhibits mechanical deformation in directions that are substantially transverse to the direction of propagation or an acoustic longitudinal wave that exhibits mechanical deformations in a direction that is substantially parallel to the direction of propagation.
The fundamental (e.g. the lowest frequency) resonance of the plate occurs when a half wavelength of standing acoustic wave is produced between the outer surfaces 7 and 8 of respective electrodes 5 and 6. Other higher-mode resonances occur at those higher frequencies at which an integral number of half-wavelengths of acoustic vibration exists between surfaces 7 and 8. Resonators having perfect physical symmetry, as suggested in FIG. 1a, exhibit only resonances at odd multiples of the fundamental frequency because of the coincidence of the odd symmetry of the applied voltage ("+" on one electrode, "−" on the other electrode) and the structural symmetry of the device. If the resonator is not structurally symmetric, then resonances may also occur at even order multiples of the fundamental frequency. In either case, the resonances are separated in frequency by a relatively wide margin, a margin that is approximately equal to the fundamental resonant frequency of the device.
In literature describing the prior art, the word resonator is sometimes used in two different senses creating a possible ambiguity. The first is to refer to that portion of the device that exhibits substantial mechanical vibratory deformations (i.e. acoustic vibrations) when periodically perturbed with a mechanical (acoustical) force at rates or frequencies near the mechanical resonant frequency of that portion of the device. Second, that literature also sometimes uses the word resonator to refer to the entire device. To avoid such an ambiguity, in this specification, unless the context indicates otherwise, the term "mechanical resonator" is intended to refer to a portion of the entire device that exhibits substantial acoustic vibrations. The term "resonator device" will refer to the entire device that contains that mechanical resonator together with a transducer, and that transducer may also function in the device as a second resonator. Further, with regard to the present invention, the term "resonator device or the term "dual-frequency resonator device" means the entire device, unless the context indicates otherwise.
Some additional definitions should be helpful to more quickly achieve an understanding of the invention. The term "transducer/resonator" usually means that portion of the acoustic resonator device that exhibits substantial acoustic vibrations near resonance and in which the transducer physically constitutes all or a substantial part of the transducer/resonator. The transducer is piezoelectric in character and includes electrically conductive electrodes or terminals for applying electrical voltages to (or extracting electrical voltages from) the piezoelectric material. The term "mechanical resonator", as earlier described, usually refers to a portion of a resonator device that exhibits substantial acoustic vibrations when perturbed at a frequency near the frequency of resonance of that portion of the device. The mechanical resonator does not include a transducer and is physically separate from the transducer/resonator, even though the mechanical resonator may be acoustically coupled to that transducer/resonator.
For clarity of illustration, the figures greatly exaggerate the thickness of the layers relative to the lateral extent of the layers. For example, in the bulk-wave piezoelectric resonator device of the type depicted in FIG. 1a, the lateral dimension of the electrodes, typically, will be of the order of fifty or so times greater than the thickness of the layer of piezoelectric material. Although not depicted in FIG. 1a, the piezoelectric layer also may extend laterally substantially beyond the right and left-hand boundaries of the electrodes. These extensions of the piezoelectric layer may be used to mechanically support the resonator device.
An acoustic device can include two or more portions that form resonators. Each of these portions exhibits an acoustic resonance. If one resonator portion is acoustically coupled to a second resonator portion, the two resonator portions interact. The interaction is similar to the effect that occurs with electrical coupling between two resonant electrical circuits (e.g. two circuits each circuit consisting of a series connection of an inductance and a capacitance) that are, individually, resonant at the same frequency. In such coupled electrical circuits, when one compares the magnitude of an electrical signal input to one resonant circuit to the electrical signal output from the second resonant circuit, as a function of frequency, one observes a narrow peak in the output signal near the resonant frequency of the two circuits when the electrical coupling between these two circuits is small, which level of coupling is commonly referred to as "under-coupled." As the electrical coupling is increased, the peak in the output signal increases in level and broadens out as a function of frequency. As the coupling is further increased the peak in the output signal versus frequency eventually broadens and splits into two separate peaks, peaked at different frequencies, with a dip in signal level between the two peaks. This circumstance is commonly referred to as "over-coupled." At the level of coupling where the peak is the broadest but does not yet exhibit a central dip, the circuits are said to be "critically-coupled." Acoustically coupled resonators exhibit similar effects.
Resonators of the conventional form (of FIG. 1a) are also known to exhibit additional resonances due to standing waves distributed parallel to the principal plate (e.g. electrode) surfaces 7 and 8. These resonances are formed by plate wave reflections at the edge of the electrodes or mounting structures. Such resonances are generally unwanted or spurious and are not used, except in some specialized applications. These resonances are normally avoided by using electrodes whose lateral dimensions are many times greater than the thickness of the piezoelectric layer. In a structure similar to that depicted in FIG. 1a, but in which the lateral extent of the electrodes is of the order of the plate thickness, the acoustic resonances are two dimensional in nature. Due to the construction of such a plate wave resonator, the lateral extent of the acoustic vibrations are substantially confined, or "trapped", by the width of the electrodes or some other physical feature. However, if two such resonators are placed in close lateral proximity, a small portion of the acoustic vibrations in the first resonator are acoustically coupled to the second resonator and induce some acoustical vibrations in the second resonator. In that configuration the lateral spacing of the two resonators controls the amount of acoustic coupling.
So-called monolithic, two-port, crystal filters have been fabricated from two or more "trapped energy" resonators placed in close lateral proximity. Wave coupling lateral to the plate thickness direction produces a split in the resonance and thus a multi-pole filter response when one resonator is driven and the other resonator is loaded into a circuit. The resonators are placed side-by-side and the coupling is perpendicular to the primary thickness resonance direction. The degree of resonator coupling and number of resonators affects the overall response.
In contrast to the side-by-side configuration of resonators in the filter described in the preceding paragraph, the present invention utilizes a transducer/resonator and a mechanical resonator that are vertically stacked, instead of being placed side by side, and, in the present invention, the transducer/resonator is separated from the mechanical resonator by layers of material that function as an acoustic coupler. The layers of the acoustic coupler are selected so as to obtain a desired degree of acoustic coupling between the transducer/resonator and the mechanical resonator. Additionally, the present invention is a single-port device.
In this specification the term "transducer" is intended to refer to a device (or that portion of a device) that converts electrical signals into mechanical vibrations (and vice-versa). The term "transduction" refers to the physical process of converting or transforming electrical signals into mechanical vibrations (and vice-versa). Thus, in the prior art transducer of FIG. 1a, the transducer is composed of electrodes 5 and 6 and piezoelectric material 2, and the resonator is bounded by surfaces 7 and 8 that reflect acoustic waves generated by the transduction process. That confines most of the acoustic vibrations to the region bounded by surfaces 7 and 8. In this simple case, the transducer constitutes the entire resonator portion of the device and is referred to herein as the transducer/resonator.
Reference is made to FIG. 1b, labeled "prior art". FIG. 1b illustrates an acoustic resonator device that constitutes a variation of the simple, thickness-mode plate resonator of FIG. 1a. Here, transducer 51 is attached to layer 59 in a vertically stacked unitary assembly so that the combination forms a resonator device 50. At resonance, a half wavelength (or appropriate multiple of a half wavelength) acoustic wave is produced between surfaces 57 and 60. Although transducer 51 may, alone, be substantially less than a one-half (acoustic) wavelength in thickness, the distance between surface 57, the upper surface of the transducer, and surface 60, the bottom surface of layer 59 comprises one-half acoustic wave-length or a multiple thereof. The principal acoustic vibrations occur between surfaces 57 and 60. The combination of transducer and layer 59, which together have an acoustic thickness of one-half acoustic wavelength, is also referred to herein as a "transducer/resonator." It should be remembered, however, that in many instances the transducer may constitute all of the resonator.
As a further variation of the foregoing configuration, transducer 51, alone, may be a half wavelength thick, and layer 59 is then a half wavelength thick or an integral multiple thereof. In this latter case these resonator devices are conventionally referred to as "overmoded" resonators, see K. M. Lakin, G. R. Kline and K. T. McCarron, "High Q Microwave Acoustic Resonators and Filters", IEEE Trans. Microwave Theory Tech. Vol. 41 no. 12, December 1993, pp. 2139–2146.
A further variation on the thickness mode plate resonator device is shown in FIG. 2. Here a single port, resonator device 90 comprises substrate 80, acoustic isolator 89, and transducer/resonator 79 in a vertically stacked integral (e.g. unitary) assembly. Substrate 80 is the general support structure and may consist of any of a wide variety of materials, such as silicon, alumina, sapphire, or other material, generally in a wafer form that is compatible with integrated circuit processing techniques. Acoustic isolator 89 may consist of a single layer of material or of a sequence of quarter wavelength thick layers of material, illustrated in this example as layers 81 through 85, in sufficient numbers and having sufficient relative impedance discontinuities between adjacent layers, to thereby acoustically isolate transducer 79 from substrate 80. Such layers are often designed to have alternating extremes of acoustic impedance such that significant wave reflections occur at surfaces 91 through 96 such that little or no vibrations reach substrate 80. Typically layers of high impedance materials such as aluminum nitride ("AlN") or tungsten ("W") alternate with layers of low impedance materials such as silicon dioxide ("SiO2"). An acoustic isolator is sometimes referred to herein as a reflector.
Transducer 79 is composed of piezoelectric region 87 and interfaces 97 and 98, to which are attached associated electrodes 86 and 88. Electrodes 86 and 88 include external points of electrical connection to other circuitry, not illustrated. Transducer 79 also serves as a resonator. The acoustic vibrations at resonance are substantially confined between surface 96, i.e. the inner or lower boundary of the transducer, and the air or vacuum interface at surface 99, i.e. the outer or upper surface of the transducer, in a manner similar to that of the conventional resonator illustrated in FIG. 1a.
The acoustic vibrations in device 80 rapidly diminish with distance from resonator 86 within the isolator reflector array 81 through 85, as illustrated in FIG. 3, to which reference is made. In FIG. 3 transducer region 79, which is about 4.00 μm thick at 1600 mhz, shows a half wavelength of acoustic vibration. The standing wave rapidly diminishes in amplitude throughout acoustic isolator 89 with little wave amplitude reaching substrate 80. For this example, electrodes 86 and 88 are each 0.3 μmeters of aluminum, piezoelectric material 87 is 3.0 μmeters of AlN, with the reflector sequence starting with 0.7 μm of SiO2 and then 1.76 μmeters of AlN alternating until the final layer of SiO2 is reached. The wave distribution was calculated for 1600 MHz with +1.0 Volt on electrode 88 and −1.0 Volt on electrode 86.
In contrast to the resonator devices of FIGS. 1a and 1b, the graph of the amplitude of acoustic vibration vs distance presented in FIG. 3 illustrates that the acoustic vibrations in resonator device 90 of FIG. 2 extend beyond the boundaries of transducer/resonator 79 and, in fact, penetrate into isolator 89. Because the acoustic vibrations are not entirely confined to transducer/resonator 79, and because a small portion of these vibrations extend into the isolator, the electrical and acoustic properties of the device are, to some extent, affected by the presence of and the acoustical properties of the layers of material that constitute the isolator. Resonators of the type described in FIG. 2 have been disclosed in U.S. Pat. Nos. 3,414,832 and 5,373,268 and 5,821,833.
Because the present invention possesses more complex properties, to aid in understanding the present disclosure, electrical resonances are defined in terms of the electrical impedance of the structure rather than by the more conventional half-wavelength or frequency-thickness descriptions. The electrical characteristics of a piezoelectric resonator can be described by the electrical impedance of the input port, i.e. at the input terminals to the electrodes, and in a manner analogous to well known inductor-capacitor resonant circuits. Most useful in identifying and defining the meaning of electrical resonance in such a one-port device is the nature of the phase of the electrical input impedance, Zin. as a function of frequency, f. As an example, reference is made to FIG. 4 showing the computed phase and magnitude of the electrical input impedance of a resonator near the fundamental resonant frequency of approximately 1600 MHz. The magnitude of the input impedance is shown as a dashed line and the phase of the input impedance is shown as a solid line. At frequencies 120 below electrical resonance, the phase is near −90 degrees (lagging), which is analogous to a capacitive reactance. At frequency 121, the phase slope is positive, the phase is zero degrees, and the impedance is purely resistive and at a minimum value. The conditions of positive phase slope, zero phase, and minimum impedance characterize series resonance.
Between frequencies 121 and 123, the phase 124 is positive, analogous to an inductance. At frequency 123 the phase slope is negative, the phase is zero, and the impedance is resistive and of high value relative to the resistive impedance at 121. The impedance conditions at 123 characterize parallel resonance. At higher frequencies 125, the phase is again negative, analogous to a capacitive reactance. For the purpose of this specification, series resonance means the frequency at which the input impedance exhibits zero phase and a positive phase slope with a relatively low value of resistive impedance. In contrast, parallel resonance means a frequency at which the input impedance exhibits zero phase, negative phase slope and a relatively high resistive impedance.
A simple resonator of the prior art whose input impedance exhibits an input phase that undergoes a phase excursion from approximately −90 degrees to approximately +90 degrees and then back to approximately −90 degrees as the frequency increases, such as that depicted in FIG. 4, is referred to in this specification as a single-frequency electrical resonator. The single frequency resonator exhibits an electrical resonance that includes both a series resonance and a parallel resonance at a slightly higher frequency. For the purpose of this specification, the frequency increment between series resonance, 121, and parallel resonance 123 is defined as the resonator bandwidth. The mean of the series resonance frequency and the parallel resonance frequency is referred to as the nominal resonant frequency. Although the prior art device depicted in FIG. 1a is referred to as a single-frequency electrical resonator, the device exhibits similar phase excursions at each a series of the harmonic electrical resonance's. Those harmonic electrical resonances occur at approximately odd integer multiples of the frequency of the fundamental electrical resonance for the simple resonator.
A purely mechanical resonator possesses resonances that are defined by peaks in acoustic standing wave amplitude. The concept of electrical series or parallel resonance is inapplicable to mechanical resonators. The frequency at which a mechanical resonator, in the absence of acoustic coupling to other portions of the device, would exhibit a peak in the amplitude of the acoustic standing wave is referred to herein as the mechanical resonant frequency of the isolated mechanical resonator.
Methods for fabrication of piezoelectric resonators for use at microwave frequencies are well known in the prior art. See, e.g., the descriptions of such devices in the specification of U.S. Pat. No. 5,894,647 for a "Method for Fabricating Piezoelectric Resonators and Product", Lakin, and see the references to prior art cited therein. See also "Microwave Acoustic Resonators and Filters," by Lakin, Kline and McCarron, IEEE Trans. on Microwave Theory and Techniques, Vol. 41, No. 12, December 1993, p. 2139; Guttwein, Ballato and Lubaszek, U.S. Pat. No. 3,694,677; and "Acoustic Bulk Wave Composite Resonators", Applied Physics Letters 38(3) by Lakin and Wang, Feb. 1, 1981. Such resonators also may be fabricated on, and supported by, a substrate by including a set of intervening layers of material having alternating high and low acoustic impedances, which layers have thickness' of a quarter wavelength. The intervening layers act as an acoustic mirror or reflector that acoustically isolates the resonator from the underlying substrate. See, e.g., U.S. Pat. Nos. 3,414,832 and 5,373,268 and 5,821,833 and 6,291,931. For methods of analysis and further descriptions of reflectors and resonators see Lakin, "Solidly Mounted Resonators and Filters, 1995 IEEE Proc. Ultrasonics Symposium, pp. 905–908 and Lakin et al. "Development of Miniature Filters for Wireless Applications", IEEE Trans. on Microwave Theory and Techniques, Vol. 43, No. 12, December 1996, pp. 2933–2939. It is expressly understood that the content of the foregoing prior art publications, as well as the content of any other prior art cited herein by reference to a publication or a patent is incorporated herein by reference in the entirety.
The present invention relates to single-port microwave acoustic resonator devices and, more particularly, to obtaining electrical resonances at the input impedance of the single-port electro-acoustic resonator device that have a more diverse range of properties or that are more complex than the properties exhibited by a simple, single resonator device. That result is obtained using techniques of controlling the resonant frequencies of the transducer/resonator and of the mechanical resonator and by controlling the amount of acoustic coupling between the transducer/resonator and the mechanical resonator.
Accordingly, a principal object of the invention is to provide a single port, piezoelectric resonator device, useful at microwave frequencies, whose input impedance exhibits a more complex and diverse character than can be obtained from a prior art one-port device comprising a simple transducer/resonator.
A further object of the invention is to design a single port, piezoelectric resonator device, useful at microwave frequencies, whose input impedance exhibits two or more electrical resonances that are not harmonically related and are located relatively close together in frequency.
A further object of the invention to provide in a one-port microwave acoustic resonator device whose input impedance exhibits at least two frequencies of electrical resonance that are spaced apart in the frequency spectrum by a fractional increment of the first resonance.

Single-port multi-resonator acoustic resonator device


A full bridge circuit includes a first switch, a second switch, a third switch, a fourth switch, a first controller, a second controller, a first operational amplifier, a first multiplexer, a second operational amplifier, and a second multiplexer. The first controller includes a first output coupled to the first switch. The second controller includes a first output coupled to the third switch. The first operational amplifier includes a first input coupled to the first switch and the second switch, and a second input for receiving a first reference voltage. The second operational amplifier includes a first input coupled to the third switch and the fourth switch, and a second input for receiving a second reference voltage. The first multiplexer is coupled to the first operational amplifier, the first controller, and the second switch. The second multiplexer is coupled to the second operational amplifier, the second controller, and the fourth switch.

1. Field of the Invention
The present invention relates to a DC motor and a full bridge circuit, and more particularly, to a DC motor and a full bridge circuit utilizing operational amplifiers and multiplexers to fix output voltages and avoid reverse current.
2. Description of the Prior Art
A DC motor driver is a necessary power transformation device in modern industries and the information society. The DC motor is capable of transforming electricity into kinetic energy required for driving devices. Conventional motors include DC motors, AC motors, and stepping motors. DC motors and AC motors are often applied in products not requiring delicate manipulations. For example, blades of an electric fan are rotated with a DC motor or an AC motor. As the technology of digital products grows, a rotation rate of a DC motor or an AC motor is required to be faster and faster. However, with a high rotation rate of a motor, the current of the motor cannot be consumed completely. The unconsumed and therefore remaining currents reversely flow to a corresponding power supply. This scenario leads to damages of controllers and drivers of the motor. Therefore, the damages caused by reverse current under a high rotation rate of the motor have to be avoided.
Please refer to FIG. 1, which is a diagram of a DC motor 10 in the prior art. The DC motor 10 comprises a power supply 12, an input capacitor C1, a hall sensor 16, a first controller 20, a second controller 21, and a full bridge circuit 14. The power supply 12 is utilized for generating an input voltage Vin. The input capacitor C1 is coupled to the power supply 12. A voltage difference between both terminals of the input capacitor C1 is a supply voltage VDD. The Hall sensor 16 has a first output end 162 for generating a first timing control signal H+, and a second output end 164 for generating a second timing control signal H−.
The first controller 20 has a first input end 102 coupled to the first output end 162 of the Hall sensor 16, a second input end 104 coupled to the second output end 164 of the Hall sensor 16, a first output end 106 for generating a first switch control signal, and a second output end 108 for generating a second switch control signal. The second controller 21 has a first input end 112 coupled to the second output end 164 of the Hall sensor 16, a second input end 114 coupled to the first output end 162 of the Hall sensor 16, a first output end 116 for generating a third switch control signal, and a second output end 118 for generating a fourth switch control signal.
The full bridge circuit 14 has an input end 142 coupled to the power supply 12 and the input capacitor C1, and the voltage at the input end 142 is the supply voltage VDD. The full bridge circuit 14 comprises a first switch SW1, a second switch SW2, a third switch SW3, a fourth switch SW4, and a motor loading Le. The first switch SW1 has a control terminal 132 coupled to the first output end 106 of the first controller 20, an input end 134 coupled to the power supply 12 and the input capacitor C1, and an output end 136 for generating a first output voltage Vout1. The second switch SW2 has a control terminal 152 coupled to the second output end 108 of the first controller 20, an input end 154 coupled to ground, and an output end 156 coupled to the output end 136 of the first switch SW1. The third switch SW3 has a control terminal 172 coupled to the first output end 116 of the second controller 21, an output end 174 coupled to the power supply 12 and the input capacitor C1, and an output end 176 for generating a second output voltage Vout2. The fourth switch SW4 has a control terminal 192 coupled to the second output end 118 of the second controller 21, an input end 194 coupled to ground, and an output end 196 coupled to the output end 176 of the third switch SW3. The motor loading Le has a first terminal 182 coupled to the first switch SW1 and the second switch SW2, and a second terminal 184 coupled to the third switch SW3 and the fourth switch SW4.
The first switch SW1, the second switch SW2, the third switch SW3, and the fourth switch SW4 may be metal-oxide semiconductor transistors, for example, the first switch SW1 and the third switch SW3 are P-type metal-oxide semiconductor transistors, and the second switch SW2 and the fourth switch SW4 are N-type metal-oxide semiconductor transistors. The first switch SW1, the second switch SW2, the third switch SW3, and the fourth switch SW4 may also be bipolar-junction transistors, for example, the first switch SW1 and the third switch SW3 are NPN bipolar-junction transistors, and the second switch SW2 and the fourth switch SW4 are PNP bipolar-junction transistors. In the same way, the first switch SW1 and the third switch SW3 are PNP bipolar-junction transistors, and the second switch SW2 and the fourth switch SW4 are NPN bipolar-junction transistors. Even the four switches SW1-SW4 are all PNP bipolar-junction transistors or NPN bipolar-junction transistors.
Please refer to FIG. 1 and FIG. 2. FIG. 2 is a timing diagram of the signals shown in FIG. 1. The DC motor 10 controls the switches of the full bridge circuit 14 and the waveforms of the first output voltage Vout1 and the second output voltage Vout2 are square waves. The first timing control signal H+ and the second timing control signal H− outputted from the Hall sensor 16 control turning on and turning off the first switch SW1, the second switch SW2, the third switch SW3, and the fourth switch SW4. The first switch SW1 and the fourth switch SW4 are turned on and the third switch SW3 and the second switch SW2 are turned off, when the first timing control signal H+ is low and the second timing control signal H− is high. The direction of a motor current ILe flows from the first output voltage Vout1 to the second output voltage Vout2. At this time, the first output voltage Vout1 is high and the second output voltage Vout2 is low. During the transition of the first timing control signal H+ and the second timing control signal H−, the second switch SW2 and the fourth switch SW4 are turned on, and the first switch SW1 and the third switch SW3 are turned off. The motor current ILe weakens gradually through the second switch SW2 and the fourth switch SW4. At this time, both the first output voltage Vout1 and the second output voltage Vout2 are low. When the first timing control signal H+ is high and the second timing control signal H− is low, the second switch SW2 and the third switch SW3 are turned on, and the first switch SW1 and the fourth switch SW4 are turned off. The motor current ILe flows from the second output voltage Vout2 to the first output voltage Vout1, at this time, the first output voltage Vout1 is low, and the second output voltage Vout2 is high.
Please refer to FIG. 3, which is a waveform diagram of the signals of FIG. 1 when a high rotation rate of the DC motor results in a voltage spike. During the first stage, the first timing control signal H+ is low, and the second timing control signal H− is high. The motor current ILe flows from the first output voltage Vout1 to the second output voltage Vout2, at this time, the first output voltage Vout1 is high, and the second output voltage Vout2 is low. During the second stage and the transition of the first timing control signal H+ and the second timing control signal H−, the motor current ILe weakens gradually through the second switch SW2 and the fourth switch SW4. However, since the rotation rate of the DC motor is high, the motor current ILe cannot be weakened to be zero after the transition of the switches. Therefore, during the third stage, the motor current ILe flows reversely to the supply voltage VDD through the second switch SW2 and the fourth switch SW4, and charges the input capacitor to result in a voltage spike. As shown in FIG. 3, the magnitude of the voltage spike depends on the magnitude of the reverse current flowing into the input capacitor C1 and the capacitance of the input capacitor C1. During the fourth stage when the first timing control signal H+ turns to high and the second timing control signal H− turns to low, the motor current ILe flows from the second output voltage Vout2 to the first output voltage Vout1, at this time, the first output voltage Vout1 is low, and the second output voltage Vout2 is high.
Please refer to FIG. 4, which is a diagram illustrating the flow of the motor current ILe during the first stage shown in FIG. 3. During the first stage, the first switch SW1 and the fourth switch SW4 are fully on. The motor current ILe flows from the first output voltage Vout1 to the second output voltage Vout2.
Please refer to FIG. 5, which is a diagram illustrating the flow of the motor current ILe during the second stage shown in FIG. 3. During the second stage, the second switch SW2 and the fourth switch SW4 are turned on. At this time, the motor current ILe weakens gradually. The motor current ILe continues to flow from the first output voltage Vout1 to the second output voltage Vout2.
Please refer to FIG. 6, which is a diagram illustrating the flow of the motor current ILe during the third stage shown in FIG. 3. During the third stage, motor current ILe continues to flow from the first output voltage Vout1 to the second output voltage Vout2 due to the motor current ILe not decaying to zero yet. The second switch SW2 and the third switch SW3 are turned on, and the motor current ILe flows from the second switch SW2 to the third switch SW3. At this time, the motor current ILe charges the input capacitor C1 to increase the supply voltage VDD and to result in a voltage spike.
Please refer to FIG. 7, which is a diagram illustrating the flow of the motor current ILe during the fourth stage shown in FIG. 3. During the fourth stage, since the motor current ILe has weakened to be zero, the second switch SW2 and the third switch SW3 are turned on whereas the first switch SW1 and the fourth switch SW4 are turned off. The motor current ILe flows from the second output voltage Vout2 to the first output voltage Vout1.
For a DC motor having a low rotation rate, the motor current ILe can decay to zero during the transition of the first timing control signal H+ and the second timing control signal H−. However, in modern applications, the rotation rate of a modern motor is ever increasing. When the rotation rate of the motor exceeds a limit, the motor current ILe has not weakened to be zero after the transition of the switches. At this time, the motor current ILe flows reversely to the supply voltage VDD through the second switch SW2 and the third switch SW3, and charges the input capacitor C1 to result in the voltage spike. Therefore, the controller and the driver of the DC motor 10 would be damaged or burned down, the power supply 12 may be burned down also, and the reliability and the effective operational range of the system of the DC motor 10 would be degraded.

Full bridge circuit and DC motor capable of fixing output voltage and avoiding reverse current


An artificial lift assembly and method relating thereto is designed to prevent inadvertent electrical discharge caused by fluids flowing through an electrical submersible pumping system, which is part of the assembly. The assembly uses a rupture disc to prevent fluid flow during introduction of the assembly into a wellbore and a dart and sleeve to prevent fluid flow during removal of the assembly from the wellbore.

In subsurface wells, such as oil wells, an electrical submersible pump with a motor (ESP) is often used to provide an efficient form of artificial lift to assist with lifting the production fluid to the surface. ESPs decrease the pressure at the bottom of the well allowing for more production fluid to be produced to the surface than would otherwise be produced if only the natural pressures within the well were utilized.
The typical electrical submersible pump installation consists of a downhole gauge (sensor) to monitor pressure and temperature, connected to a motor that drives a single or double seal, also known as a protector. The protector inhibits oil ingress into the motor while permitting pressure equalization between the well annulus and motor connected to the downhole pump, typically a centrifugal pump but sometimes a progressing cavity pump, or other centrifugal or positive displacement pumps. Historically, the motor has been a 2-Pole Induction motor that has existed in the marketplace for over fifty years.
Recently, the use of permanent magnet motors has come to the forefront for use in electrical submersible pumping (ESP) in oil and gas wells. Replacing the induction motor with a permanent magnet motor is new to the oil and gas industry and offers several benefits including a higher efficiency, power factor, and increased reliability. The foundation of a permanent magnet motor is that it utilizes rare earth magnets in the rotor to enable better synchronization with the electrical current flowing through the stator thereby increasing the efficiency and power factor.
One of the pitfalls with permanent magnet motors is that during installation or pump removal, the wellbore equalizes pressure through the pump which causes rotation of the pump and subsequently the motor. When the motor spins, the magnets within the rotor spin thereby generating power which is transmitted up the cable to the surface. This can present safety issues caused by technicians being unaware that the pumping system is spinning downhole and transmitting electrical power to the surface.

This patent is subject to a terminal disclaimer.


A photocatalyst comprising silver orthophosphate (Ag3PO4) is used for selective and exhaustive conversion of alcohols into their corresponding carbonyl compounds; namely, benzyl alcohol, 4-methoxy benzyl alcohol and/or cinnamyl alcohol with >99% yield. The reactions were carried out in double distilled water at room temperature and pressure under sunlight-type excitation. Oxidation of alcohols is selective and Ag3PO4 may be active for a variety of other essential oxidation reactions.

1. Technical Field
The present disclosure relates to a method for the selective oxidation of alcohols into their corresponding carbonyl compounds using Ag3PO4 as a photocatalyst to oxidize at least 90% of the alcohols into their corresponding carbonyl compounds.
2. Description of the Related Art
The "background" description provided herein is for the purpose of generally presenting the context of the disclosure. Work of the presently named inventors, to the extent it is described in this background section, as well as aspects of the description which may not otherwise qualify as prior art at the time of filing, are neither expressly or impliedly admitted as prior art against the present invention.
Selective, efficient and complete oxidation of alcohols into corresponding carbonyl compounds, such as aldehydes, ketones etc., is of paramount significance for chemical industries because carbonyl compounds are used in food, beverage, and pharmaceutical industries as well as a raw material in chemical industries (Miyamura, H.; Matsubara, R.; Miyazaki, Y.; Kobayashi, S. Angew. Chem. 2007, 119, 4229; Hundlucky, M. Oxidations in Organic Chemistry, American Chemical Society, Washington, D.C., 1990—each incorporated herein by reference in its entirety). To achieve the afore-mentioned conversions, the use of stoichiometric inorganic reagents, such as KMnO4, K2CrO4 etc., is predominantly used in these industries. Although such reagents offer high activity and selectivity, accumulation of waste products arising from the use of these inorganic reagents poses a threat to the environment. Intensive effort has been carried out in past years to develop 'green oxidation processes'. Although there are active heterogeneous metal catalysts being developed for the aerobic oxidation of various alcohols, the reactions are carried out in harmful organic solvents and/or under vigorous conditions (Kwon, M. S.; Kim, N.; Park, C. M.; Lee, J. S.; Kang, K. Y.; Park, J. Org. Lett. 2005, 7, 1077; Enache, D. I.; Edwards, J. K.; Landon, P.; Solsona-Espriu, B.; Carley, A. F.; Herzing, A. A.; Watanabe, M.; Kiely, C. J.; Knight, D. W.; Hutchings, G. J. Science 2006, 311, 362; Yamada, Y. M. A.; Arakawa, T.; Hocke, H.; Uozumi, Y. Angew. Chem. 2007, 119, 718—each incorporated herein by reference in its entirety).
Since the discovery of water splitting on TiO2 electrode in 1970 under light (Fujishima, A.; Honda, K. Nature 1972, 238, 37—incorporated herein by reference in its entirety), the photocatalytic process has been widely investigated owing to its renewable attributes. Most of the earlier studies involving photocatalysis focused on environmental cleanup, H2 production, CO2 reduction etc. (Hoffmann, M. R.; Martin, S. T.; Choi, W.; Bahnemann, D. W. Chem. Rev. 1995, 95, 69; Maeda, K.; Teramura, K.; Lu, D.; Takata, T.; Saito, N.; Inoue, Y.; Domen, K. Nature 2006, 440, 295; Varghese, O. K.; Paulose, M.; LaTempa, T. J.; Grimes, C. A. Nano Lett. 2009, 9, 731—each incorporated herein by reference in its entirety). Recently, utilization of the photocatalytic process for the synthesis of fine chemicals in an environmentally friendly fashion is desirable (Yoon, T. P.; Ischay, M. A.; Du, J. Nature Chemistry 2010, 2, 527; Palmisano, G.; Augugliaro, V.; Pagliaro, M.; Palmisano, L. Chem. Commun. 2007, 3425—each incorporated herein by reference in its entirety). Attempts have been made to achieve various functional group transformations such as amine to imine (Lang, X. J.; Ji, H. W.; Chen, C. C.; Ma W. H.; Zhao, J. C. Angew. Chem. Int. Ed. 2011, 50, 3934; Su, F. Z.; Mathew, S. C.; Mohlmann, L.; Antonietti, M.; Wang, X. C.; Blechert, S. Angew. Chem. Int. Ed. 2011, 50, 657; Wang, C.; Xie, Z. G.; deKrafft, K. E.; Lin, W. B. J. Am. Chem. Soc. 2011, 133, 13445—each incorporated herein by reference in its entirety), nitro to azo (Zhu, H.; Ke, X.; Yang, X.; Sarina, S.; Liu, H. Angew. Chem. Int. Ed. 2010, 49, 9657—incorporated herein by reference in its entirety), aniline to azobenzene conversion (Li, S.; Diebold, U. J. Am. Chem. Soc. 2010, 132, 64—incorporated herein by reference in its entirety), hexane to hexanone and hexanol (Ide, Y.; Kawamoto, N.; Bando, Y.; Hattori, H.; Sadakane, M.; Sano, T. Chem. Commun. 2013, 49, 3652—incorporated herein by reference in its entirety), alcohols to their corresponding aldehydes (Yurdakal, S.; Palmisano, G.; Loddo, V.; Augugliaro, V.; Palmisano, L. J. Am. Chem. Soc. 2008, 130, 1568; Maldotti, A.; Molinari, A.; Amadelli, R. Chem. Rev. 2002, 102, 3811; Palmisano, G.; Garcia-Lopez, E.; Marci, G.; Loddo, V.; Yurdakal, S.; Augugliaro, V.; Palmisano, L. Chem. Commun. 2010, 46, 7074; Augugliaro, V.; Caronna, T.; Loddo, V.; Marc, G.; Palmisano, G.; Palmisano, L.; Yurdakal, S. Chem Eur. J. 2008, 14, 4640; Wang, Q.; Zhang, M.; Chen, C.; Ma, W.; Zhao, J. Angew. Chem. Int. Ed. 2010, 49, 7976; Palmisano, G.; Yurdakal, S.; Augugliaro, V.; Loddo, V.; Palmisano, L. Adv. Synth. Catal. 2007, 349, 964—each incorporated herein by reference in its entirety), and so on. Selective oxidation of alcohols into corresponding aldehydes is one of the indispensable transformations in organic synthesis because aldehydes are being used in food, beverage, and drug industries as well as raw materials in chemical industries. Examples of the photocatalysts that are studied for alcohol oxidation include CdS/graphene (Zhang, N.; Zhang, Y.; Pan, X.; Fu, X.; Liu, S.; Xu, Y-J. J. Phys. Chem. C 2011, 115, 23501—incorporated herein by reference in its entirety), CdS/graphene (Zhang, N.; Zhang, Y.; Pan, X.; Yang, M.-Q.; Xu, Y.-J. J. Phys. Chem. C 2012, 116, 18023—incorporated herein by reference in its entirety), and Au/CeO2 (Tanaka, A.; Hashimoto, K.; Kominami, H. Chem. Commun. 2011, 47, 10446; Tanaka, A.; Hashimoto, K.; Kominami, H. J. Am. Chem. Soc. 2012, 134, 14526—each incorporated herein by reference in its entirety).
Xu et. al., studied the conversion of benzyl alcohol into benzaldehyde in the presence of CdS assembled on two-dimensional graphene scaffold under visible light in organic solvent (namely benzotrifluoride). Although the selectivity was ˜80%, the conversion and the yield were <50%. Furthermore, Xu et. al., synthesized a ternary photocatalyst involving CdS/graphene/TiO2 for the oxidation of benzyl alcohol to benzaldehyde and demonstrated that the ternary had better photocatalytic activity as compared to binary CdS/graphene. The conversion of benzyl alcohol was ˜80% while selectivity towards benzaldehyde and yield of benzaldehyde were ˜80% and <80% respectively, but the reaction was carried out in the organic solvent benzotrifluoride.
Kominami et. al. deposited Au nanoparticles on a CeO2 photocatalyst surface and studied photocatalytic chemoselective oxidation of alcohols to their corresponding aldehydes caused by plasmonic effect of Au under irradiation by green light. Although high selectivity (>99%) and a complete (>99%) oxidation of benzyl alcohol to benzaldehyde was obtained, the rate of reaction or benzaldehyde formation was extremely low (3.0 μmolh−1). In addition, disparate forms of TiO2, namely rutile,1 anatase, and brookite (Addamo, M.; Augugliaro, V.; Bellardita, M.; Paola, A. D.; Loddo, V.; Palmisano, G.; Palmisano, L.; Yurdakal, S. Catal. Lett. 2008, 126, 58—incorporated herein by reference in its entirety), and surface modified TiO2, such as Nb2O5/TiO2 (Furukawa, S.; Shishido, T.; Teramura, K.; Tanaka, T. ACS Catal., 2012, 2, 175—incorporated herein by reference in its entirety), and Pt/TiO2 (Zhai, W.; Xue, S.; Zhu, A.; Luo, Y.; Tian, Y. ChemCatChem. 2011, 3, 127—incorporated herein by reference in its entirety), have been investigated and it has been found that the rutile form is the most selective photocatalyst for the oxidation of the alcohol to its corresponding aldehyde.
In most of the studies, organic solvents were used to achieve high selectivity. The highest conversion and selectivity in water, which was 50% and 65% respectively, for the oxidation of alcohol to its corresponding aldehyde has been reported with rutile TiO2 under UV light irradiation at room temperature.
The development of active catalysts for selective and efficient conversion of alcohols to aldehydes in energy efficient and environmentally benign ways is highly desirable by chemical industries. The current disclosure describes the selective (>99%) and the exhaustive conversion (>90%) of benzyl alcohol and 4-methoxy benzyl alcohol to their corresponding aldehydes, namely benzaldehyde and 4-methoxybenzaldehyde respectively, with >99% yield, photocatalyzed by Ag3PO4 in water at room temperature and pressure under sunlight-type excitation. Moreover, >90% conversion of cinnamyl alcohol to cinnamaldehyde with >90% selectivity and >90% yield is also achieved in aqueous suspension of Ag3PO4 at room temperature and pressure under sunlight-type excitation.

Photocatalyst for selective oxidation of alcohols: a green synthesis approach

King Fahd University of Petroleum and Minerals


A sheet feeding device includes a stacking plate, a sheet stacker, and a damping member. The stacking plate is liftably supported in the sheet stacker and is adapted for sheets to be fed into a sheet processing apparatus to be stacked thereon. The sheet stacker is adapted to be movable between a housed position where the stacker is housed in a housing and an exposed position where the stacking plate is exposed outside of the housing. The damping member is adapted to exert on the sheet stacker a damping force according to moving speed of the sheet stacker, thereby limiting the moving speed.

The present invention relates to sheet feeding devices, such as large capacity cassettes (hereinafter merely referred to as LCCs), adapted for use in sheet processing apparatus, such as image forming apparatus, to store therein a large number of sheets to be fed into the apparatus. The present invention further relates to image forming apparatus provided with such sheet feeding devices.
Conventional sheet feeding devices are positioned beside sheet processing apparatus for storing sheets of a size that is most frequently used therein. For example, LCCs adapted for use in copying machines as image forming apparatus generally have a capacity of approximately 2,000 sheets of A4-size plain paper placed in landscape orientation.
Designed for multipurpose use and to perform various functions such as of printing or facsimile communication, recent image forming apparatus tend to handle an increasing number of sheets of various sizes and types.
In light of the foregoing, there have been developed LCCs with a capacity of 4,000 or more sheets of various sizes. With 4,000 to 5,000 sheets of A3-size plain paper stored therein, such LCCs have a total weight of approximately 100 kg.
On the other hand, there has been strong demand for smaller and lighter image forming apparatus. Suppose an image forming apparatus is provided with an LCC. The LCC includes a sheet stacker for stacking sheets, and the sheet stacker is removable from a housing of the LCC. When the sheet stacker with a large number of sheets stacked therein is pulled out of, or pushed into, the housing, the impact of collision between the sheet stacker and the housing causes the image forming apparatus to vibrate or move. Such vibration or movement causes components inside the image forming apparatus to become loosely mounted or prevents the image forming apparatus from being maintained in a horizontal position.
To solve the foregoing problems, JP H11-208902A discloses an LCC that has an elastic member arranged in a housing so as to face a rear side surface of a sheet stacker. The elastic member is intended to cushion an impact of collision caused between the housing and the sheet stacker when the stacker is moved. JP 2003-267565A discloses an LCC that has a housing with an openable upper surface. The openable upper surface allows access to a sheet stacker from above, thereby eliminating the need to remove the sheet stacker from the housing for sheet replenishment or any other operation.
However, it is hard to determine an optimum shape, material, size, etc., for the elastic member in order to ensure that the elastic member cushions the impact of collision between the housing and the sheet stacker. Also, the construction as disclosed in JP 2003-267565A involves complicated arrangement of sheet feeding members and also makes it difficult to stack sheets in the sheet stacker without causing damage, such as bent corners, to the sheets.
A feature of the invention is to provide an LCC that ensures that a collision impact on a sheet processing apparatus is cushioned with a damping member provided in a sheet stacker. The damping member is adapted to act on the sheet stacker a damping force according to moving speed of the sheet stacker as being moved in and out of a housing of the LCC. Another feature of the invention is to provide an image forming apparatus that prevents components therein from becoming loosely mounted and is allowed to be maintained in a horizontal position.

Sheet feeding device and image forming apparatus


A method for 3DTV multiplexing according to the present invention comprises, deriving a delay value in units of frames for the left and right image based on a left image PES (Packetized Elementary Stream) corresponding to a left image and a right image PES corresponding to a right image, carrying out synchronization of the left image PES and the right image PES based on the delay value in units of frames, and generating 3DTV transport streams (TSs) by carrying out multiplexing of the synchronized left image PES and the synchronized right image PES. According to the present invention, efficiency of video services can be improved.

1. Field of the Invention
The present invention relates to image processing. More specifically, the present invention relates to a method for 3DTV multiplexing and an apparatus thereof.
2. Discussion of the Related Art
Three dimensional (3D) digital broadcasting services are getting attention as one of the next generation broadcasting services together with UDTV services subsequent to HDTV. It is expected that owing to the advancement of related technologies such as release of high definition, commercial stereoscopic display, 3DTV service which enables people to enjoy 3D images will be provided for every home in the coming years. In particular, those 3D broadcasting services currently provided in the form of a commercial service or a trial service usually make use of stereoscopic videos consisting of left and right images.
A plurality of images correlated with each other can be stored or processed at the same time during 3D image processing. In addition to 3D images, free-viewpoint images, panoramic images, multi-view images, and multi-segmented images correlated with each other may also be stored or processed at the same time. Here, a multi-segmented image refers to an ultra-high resolution image whose resolution is four to sixteen times the resolution of an HD image divided into a plurality of HD images. As described above, in case of storing or processing a plurality of images correlated with each other at the same time, the plurality of images should be synchronized with each other in units of frames.

Method for 3DTV multiplexing and apparatus thereof

Electronics and Telecommunications Research Institute


A lyre mounting bracket (14) is provided for acoustically coupling a musical instrument tuner (12) to a lyre (16) for a musical instrument (10). The tuner (12) includes an electronics section (26) having a vibratory sensor. A hinged elbow member (44) secures the mounting bracket (14) to the tuner (12). The mounting bracket (14) has a main body (42) which includes two oppositely facing, planar surfaces (50, 52) for securing directly between a clamp (20) of the lyre (16). Support braces (58, 60) extend outward from opposite sides of the main body (42), each having arcuately shaped sides (66-76) which extend orthogonal to and adjacent to the peripheral edges of respective ones of the planer mounting surfaces (50, 52). The sides (66-76) have profiles which fit substantially flush with portions of a profile (94) defined by the peripheral edges of a first half (90) of the lyre clamp (20). The first half (90) of the lyre clamp (20) includes two lobes (96, 98) which fit substantially flush against mounting ears (54, 56) defined by the main body (42) of the mounting bracket (14). The support braces (58, 60) further include protuberances (78, 80) which extend into crooks (97, 99) of the profile (94) of the first half (90) of the lyre clamp (20). The arcuately shaped sides (66-76) and the protuberances (78, 80) of the support braces (58, 60), and the planar mounting surfaces (50, 52) together define two oppositely facing sockets (62, 64) for receiving the two parts (90, 92) of the lyre clamp (20) in a closely fitting relation, such that the lyre mounting bracket (14) is acoustically coupled to the lyre (16) for transmitting mechanical vibration therebetween.

Electro-mechanical musical instrument tuners have been provided for acoustically coupling to musical instruments for sensing mechanical vibrations of the musical instruments to determine the pitch of tones being emitted by the musical instruments. The detected pitch is then utilized to provide an output indicating the pitch of the tone being played. The musical instrument tuners can be utilized both for tuning the musical instruments and for developing a person's ear for detecting the pitch being played, such as for teaching a person to recognize various intonations. Musical instrument tuners have been clamped directly to the instruments using C-type clamps to acoustically couple the tuners to the instruments, such that mechanical vibrations will pass to a vibratory motion detector mounted within the housing of the tuner. The C-type clamps typically have a threaded clamping member which is subject to over tightening of the threaded coupling, which may cause damage to the musical instruments from excessive forces being applied to the instrument. Alligator clips have also been used to clip musical instrument tuners to instruments, which may result in teeth of the alligator clips placing scratch marks on the exterior of the instruments.
Musical instruments such as brass and woodwind instruments often have an attached clamp for holding sheet music, which is called a lyre. Such clamps have a lower clamping member which is typically mounted to the instrument and clamps an elongated member to the musical instrument. The opposite end of the elongated member has a second clamp which is spring biased and which is utilized to clamp sheet music to the instrument, such that a person playing the instrument may view the sheet music. Such lyres have often been made such that the upper clamps have an arcuately shaped profile defined by peripheral edges of one of the clamp members which is of a particular shape of an ancient Greek lyre, or harp. The arcuately shaped profile includes two lobes on an upper portion thereof, which are pressed against the back of the sheet music to hold the sheet music in position. The upper clamps also typically include a clamp member having two arcuately shaped prongs which extend in an upwardly direction for clamping against the front side of the sheet music, while causing minimal interference with viewing of the outer page of the sheet music.

Lyre mounting bracket for a musical instrument tuner


A pneumatic tire has a conductive portion formed by a conductive rubber is provided in the tread rubber formed by a nonconductive rubber. The conductive portion has an outer skin portion and a projection portion. The outer skin portion is formed on a surface of the tread rubber, and extends to an inner side in a tire width direction from a side surface of the tread rubber so as to terminate at a position which goes beyond a contact end. The projection portion spirally extends from a position in an outer side than contact end so as to reach a position in an inner side in the tire width direction than the contact end, and protrudes to an inner side in a tire diametrical direction from the outer skin portion so as to terminate without reaching a bottom surface of the tread rubber.

Field of the Invention
The present invention relates to a pneumatic tire which can discharge a static electricity generated in a vehicle body and a tire, to a road surface, and a manufacturing method of the pneumatic tire.
Description of the Related Art
Recently, there have been proposed a pneumatic tire including a tread rubber blended with silica at a high ratio in order to reduce the rolling resistance that largely affects a fuel consumption performance. However, compared to a tread rubber blended with carbon black at a high ratio, the electric resistance of such tread rubber is high, and accordingly, static charge generated on a vehicle body or the tire is prevented from being released to the road surface. As a result, problems like radio noises tend to occur.
Accordingly, there has been developed a pneumatic tire structured such that a tread rubber constructed by a nonconductive rubber blended with a silica or the like is provided with a conductive portion constructed by a conductive rubber blended with a carbon black or the like, whereby an electrical conduction performance can be achieved. For example, in a pneumatic tire described in the Japanese Unexamined Patent Publication No. 2000-190709, a surface of a nonconductive tread rubber is provided with a thin conductive portion in which one end is connected to a wing rubber made of a conductive rubber, and another end is terminated at an inner side in a tire width direction than a contact end, thereby forming a conduction route for discharging a static electricity.
However, in the pneumatic tire as mentioned above, when a wear makes progress to such a degree that a surface of the tread rubber disappears by a thickness of the conductive portion, the conductive portion becomes short, a position of a terminal end moves close to a contact end, and a frequency at which the conductive portion comes into contact with a road surface is widely reduced. Therefore, there has been a tendency that an electrical conduction performance after the wear is lowered. On the contrary, there can be considered to employ a strategy of forming the thick conductive portion, however, in the light of enhancing an improving effect by forming the tread rubber by the nonconductive rubber, it is desirable to employ a means which can make a volume of the conductive portion as small as possible.
Further, in a pneumatic tire described in the following Japanese Unexamined Patent Publication No. 2008-13000, a conductive route for discharging a static electricity is formed by embedding a conductive portion which is spirally continuous along a tire circumferential direction in a nonconductive tread rubber, however, in the structure mentioned above, since the conductive route becomes long in distance, there has been a tendency that an electric resistance is increased. In order to well secure the electrical conduction performance, it is desirable to employ a means which can suppress the long distance of the conductive route at least in an early stage of the wear.

Pneumatic tire and manufacturing method of the same


An apparatus for and method of arbitrating bus conflicts is disclosed. The system can use information about the status of a data stream to switch between more or less restrictive prioritization schemes. In one particular embodiment, the arbitration system can also be programmed by causing it to store information on overall system requirements for data which the arbitration system can then use to set a current prioritization scheme.

In many computer systems, components are connected by a shared bus. In such an architecture, only one unit at a time can exert control over the shared bus. Contention among the units which request use of the bus at the same time must be resolved in some manner, i.e., arbitrated.
The most basic arbitration scheme is the so-called "first in first out" arbitration scheme. In this scheme, requests to use the bus are granted strictly in the order in which they arise. This scheme has the advantage of simplicity, but the disadvantage that the relative importance of a device is not taken into account in determining when to grant its request for access to the bus. Thus, devices urgently requiring access to the bus may have to wait while other lower priority devices complete their use of the bus.
A commonly used prior art method for bus arbitration overcomes the above disadvantage by using a central arbiter communicating with all of the devices requesting use of the bus. The central arbiter determines which device has the highest bus use priority taking into account relative importance of the device and how long the device has been waiting to use the bus.
Another known arbitration system can be referred to as "daisy chaining". In this method, a signal indicating that the bus is available for use is transmitted from device to device in priority order. Any device requiring access to the bus captures the signal during its turn and uses the bus. Another prior art scheme offers a rotating priority method wherein bus access is offered to each device in a strict rotation.
The demands for efficient arbitration of bus contentions are especially pronounced in a computer system which processes real time data streams, such as video data streams, and audio data streams, animation data streams. It is crucial for real time data streams to be transferred without interruption since an interruption may be visually or audibly noticeable to an observer. Furthermore, if an arbiter is unable to provide a relatively rapid grant to a device transferring real time data streams, large buffers within the device may be required to store the real time data before it can be transferred. For example, if the arbiter were not assuredly able to provide a relatively swift grant to a video input DMA controller, the video input DMA controller would need large data FIFO buffers to detain high bandwidth video input until it can be transferred via the bus. Large FIFO buffers add significant size and thus cost to the design of a video input controller.
As a result, processing real time data streams requires special control of the system bus beyond what is necessary for non-real time data streams. In some prior art systems, this need is addressed by permitting a real time data stream controller such as a video input DMA controller to retain exclusive use of the bus once granted control until the entire video data stream has been transferred.

Apparatus for and method of arbitrating bus conflicts


A power-split transmission having several ranges of transmission ratios with continuously variable speed reductions. A drive shaft is connected to two shafts of a variable speed drive unit via a distributor gear whose shafts are in turn connected with the input shafts of a parallel gearbox via respective clutches. Two ranges of ratios only differ from each other through the reduction of the input shaft of the parallel gearbox and their output shafts. An additional embodiment includes a variable speed drive unit, a first planetary drive that is connected with a drive shaft, an input shaft of the variable speed drive unit and a connecting shaft, as well as a second planetary drive that is connected with the connecting shaft of the output shaft of the variable speed drive unit, and an output shaft and an electric motor that is nonrotatably connected with the connecting shaft.

1. Field of the Invention
The invention concerns a branched power transmission with several transmission ratio ranges and with a steplessly adjustable transmission ratio, as well as a parallel gearbox, especially for use in such a transmission. The invention further concerns a method for operating a branched power transmission. Additionally, the invention concerns a method for controlling the contact pressure or the contact force between the conical disks and the endless torque-transmitting means of a variable speed unit in such a branched power transmission. The invention further concerns such a transmission that additionally contains an electrical machine. Furthermore, the invention also concerns a method for the low-loss production of an output torque of the transmission when a motor vehicle is stationary as well as a method for operating a transmission with a steplessly variable transmission ratio and an electrical machine.
2. Description of the Related Art
Automatic transmissions with continuously adjustable transmission ratios are gaining increasing interest in motor vehicles not only because of the high driving comfort, but also because of the lower fuel consumption in comparison to conventional automatic transmissions that operate with planetary gear sets. Such transmissions contain, for example, a variable speed unit that is formed by two conical disk pairs enwrapped by an endless torque-transmitting means, wherein the spacing between the conical disks of the conical disk pairs is adjustable in opposite directions for adjusting the transmission ratio.
One problem with variable speed units lies in their limited transmission ratio adjustment range as well as in their limited torque transmitting capability. In order to enlarge the gear spread, meaning the transmission ratio adjustment range, and the torque transmitting capability, branched power automatic transmissions were produced in which the variable speed unit can be connected in various ways with a gear transmission through at least one clutch, wherein the adjustment range of the variable speed unit traverses is carried out during the change of the transmission ratio of the overall transmission in one direction or the other in accordance with the clutch position, so that an enlarged gear spread results in that case by a reduced variable speed unit spread. Further, at least in a branched power transmission in which a part of the drive torque is transmitted parallel to the variable speed unit through a clutch directly to the gear transmission or the output, the variable speed unit does not have to transmit the entire drive torque, whereby the torque transmitting capability of the transmission is enlarged.
FIG. 1 shows the basic structure of a first embodiment of a known branched power transmission.
The drive torque originating from an engine that is not shown and that is transmitted by a input shaft 2 is transmitted on the one hand directly to a variable speed unit 4 and on the other hand to a gear transmission 8 constructed as a summing transmission or a coupling transmission through a clutch 6, of which one input is connected with the output of the variable speed unit 4 and whose output is constituted by an output shaft 10, by which the motor vehicle is driven. One characteristic of the branched power transmission with the basic structure in accordance with FIG. 1 is that the variable speed unit 4 must transmit the full engine torque in the non-split region (clutch 6 disengaged) and must, therefore, be correspondingly designed for high torques and high rotational speeds, which means expense. Known branched power transmissions of that structure have a maximum spread of 7 and a torque transmitting capability of about 500 Nm.
A modified basic structure of a branched power transmission is shown in FIG. 2. In that structure, an input shaft 2 directly drives an input-side distributor transmission 12, for example a planetary transmission that has two outputs, one of which is connected with a variable speed unit 4 and the other is connected through a clutch 6 with the output shaft 10 which is non-rotatably engaged with the output of variable speed unit 4. Also in this embodiment, variable speed unit 4 must transmit the entire engine power when clutch 6 is disengaged.

Transmission with continuously variable transmission ratio

LuK Lamellen und Kupplungsbau Betilligungs KG


Automatic testing a group of line circuits on a smart line card served by a central switch over a PCM bus is localized by using the microprocessor/DSP that provides codec functions for the group of line circuits. Tests such as DC overhead that are not performable by the central switch over the PCM bus are performed by the microprocessor/DSP when informed by the central switch that at least one of the channels is not processing voice signals. The microprocessor/DSP controls test switches on the line card to set-up various test terminations for the idle circuit as well as to selectively control the gain of the idle channel's amplifiers. The microprocessor/DSP synthesizes test tones and executes test tone detection processes within the time slot intervals accorded by the central switch to the idle channel. Among the tests performed are Forward Loop Gain with different resistance terminations, cable and fuse tests, DC overhead, and idle channel noise utilizing a limited number of test routines such as tone generation, DC generation, discrete Fourier transform, peak detection and variance from acceptable limits. The pass/fail condition of the idle channel under test is reported to the central switch.

The testing of customer loops by the central switch in modern digital telephone switching systems, such as the 5ESS Switch manufactured by Lucent Technologies Inc., is performed on both an analog and digital basis. Digital testing alone does not provide sufficient fault coverage of line card circuitry, and therefore, the performance of additional tests requires the direct connection of a test unit to check the electrical parameters of the subscriber loop. Digital testing is disclosed in Marsh et al, U.S. Pat. No. 5,596,322 issued Jan. 21, 1997 which describes a 16-channel programmable codec known as the T7531/35 manufactured by Lucent Technologies Inc. The Marsh patent employs the codec's DSP to calibrate or trim the amplifier gains of 14 of the channels based on the factory calibration of only two of the channels. In the Marsh method, the DSP applies analog calibration signals to each of the A/D paths and compares the output signals received from the paths with the output signal received from the pre-trimmed path to determine the A/D gain correction coefficients for the paths. The DSP also applies a digital calibration signal to each D/A path and sets up a loop-back condition for routing the output signal from the D/A paths to one of the A/D paths and then compares the results to set the D/A coefficients.
Another digital testing technique is disclosed in Lopresti et al U.S. Pat. No. 5,559,440 issued Sep. 24, 1996 and assigned to Lucent Technologies Inc. The Lopresti patent describes tests of the transmit and receive paths that traverse the line circuits by temporarily plugging in a personal computer to the line circuit's PCM bus to ascertain whether the A/ID and D/A channel gains, return loss, terminal balance return loss, noise and distortion parameters are within acceptable limits.
While the Marsh and Lopresti arrangement operate satisfactorily, the need exists for an automated, self-testing capability on the line card that does not require the connection of a personal computer to perform floating point calculations or to generate and detect the test tones. In addition, it would be desirable to perform certain tests on the line circuits, such as DC overhead measurement and peak amplitude measurement that cannot be performed by a central switching facility linked to the line circuits by a PCM bus and which introduces its attendant complications of companding, filtering, limited data sampling rate and the inability to convey DC signals.
While it might appear to be desirable to perform such tests locally at the line card and thereby eliminate the effects of the PCM bus, the implementation of such tests at the line card presents several problems. Among the problems presented are how to implement with limited processing capability the necessary digital tone generation and detection, how to compute the variance of signals having different DC offsets, how to accomplish peak detection all without substantially interfering with voice processing.


A slope rate compensation circuit includes a source follower level-shift amplifier, a capacitor, a first resistor and a second resistor. The source follower level-shift amplifier includes a first transistor and a second transistor. The first transistor allows a first current to flow therein, the second transistor allows a second current to flow therein, and the first current increases with the second current. The capacitor is connected to the source terminal of the first transistor. The first resistor is connected to the source terminal of the second transistor. The second resistor allows a third current to flow therein, and the third current increases with the second current. The second resistor is related to the output voltage of the slope rate compensation circuit.

1. Field of the Invention
The present invention relates to a slope rate compensation circuit, method thereof and pulse width modulation (PWM) boost converter circuit, and more particularly, to a slope rate compensation circuit, method thereof and pulse width modulation boost converter circuit using a constant increasing slope rate.
2. Description of the Related Art
FIG. 1 shows a known pulse width modulation boost converter circuit 1, which comprises a boost converter circuit 10, a pulse width modulator 11, an oscillator 12, an error amplifier 13, a voltage divider 14 and a compensation circuit 15. When the pulse width modulation boost converter circuit 1 is activated, a reference voltage Vref is applied to the non-inverting input of the error amplifier 13. Meanwhile, the inverting input of the error amplifier 13 is connected to a feedback voltage VFB of the voltage divider 14 to define the magnitude of the DC output voltage Vout. The oscillator 12 outputs a pre-oscillating signal SOSC to the pulse width modulator 11 so as to generate a PWM signal SPWM. The PWM signal SPWM enables the transistor switch SW1, increasing the inductor current and storing energy in the inductor magnetic field. The output Eo of the error amplifier is compared with a constant-frequency sawtooth wave to determine the disabling duration of the switch SW1, so that the inductor current IL, generated by the input voltage Vin, flowing through the boost inductor L, is capable of intermittently charging the capacitor C11 and thus increases the DC output voltage Vout. The diode D11 limits charge current direction of the capacitor C11. When SW1 is disabled, D11 is forward biased.
A PWM boost converter circuit is applied to either the circuit mode PWM control or to voltage mode PWM control. The current mode PWM control uses inductor current as a feedback signal, and thus exhibits a superior loop performance over the voltage mode PWM control. In addition, without a leading compensation, the current mode PWM control has a phase margin 90 degrees more than does the voltage mode PWM control, and thus is more stable.
Generally, if the duty ratio of a PWM switching regulator is greater than 50% in the current mode PWM control, it is necessary to conduct a slope rate compensation. The compensated slope rate ma must satisfy the criteria
            m                                      2        2    <  maso as to avoid unstable and inefficient situations. When ma is equal to m2, it represents the optimal compensation slope rate, where −m2 is the slope rate of the inductor current when the transistor switch SW1 is disabled.
FIG. 2 shows a waveform of the inductor current IL of the PWM boost converter circuit 10. As shown in FIG. 2, the slope rate m1 of the inductor current is
      V          i                                      n        Lwhen the transistor switch SW1 is enabled, the slope rate −m2 of the inductor current is
            V      out        -          V      IN        Lwhen the transistor switch SW1 is disabled, and
      V    out    =            V              i                                                  n              ×                  1                  (                      1            -                          duty                                                                                ratio                                )                    .      Therefore, if ma is equal to m2,
  ma  =                    V                  i                                                              n                    L        ×                            duty                                                              ratio                          (                      1            -                          duty                                                                                ratio                                )                    .      The primary difficulty in determining proper slope rate compensation is that the input and output voltages of the PWM boost converter circuit 10 are variable, as is the duty ratio; therefore it is not easy to locate a corresponding variable slope rate compensation.
U.S. Pat. No. 6,611,131 discloses a piecewise-linear slope rate compensation, which adopts resistors to detect inductor current and uses a BJT transistor to conduct the slope rate compensation. First, it results in a greater power consumption. Second, the level of voltage clamping shifts with the voltage source, so the design is more complicated. Third, the current limit is achieved by clamping the feedback signal of the inductor current, but the current limit level is distorted after a summation is performed with the slope rate compensation signal.
FIG. 3A shows output voltage waveforms of high duty ratio and low duty ratio without slope rate compensation, and FIG. 3B shows output voltage waveforms of high duty ratio and low duty ratio with slope rate compensation. As can be seen in FIG. 3B, the output voltage of high duty ratio with slope rate compensation reaches the circuit limit level more easily than the output voltage of low duty ratio with a slope rate compensation voltage since a larger slope rate is added.
U.S. Pat. No. 6,522,116 discloses another slope rate compensation technique, which must detect the variation of the output voltage before a slope rate compensation is conducted. However, detecting the variation of the output voltage itself is a difficult task. Since generally, the monolithic IC has no access to the output voltage, if the detection circuit exhibits an inaccurate performance due to oversimplification, the overall performance is decreased. In addition, this prior art adopts a large resistor to perform a linear effect, thus resulting in excessive power consumption.
U.S. Pat. No. 5,717,322 discloses another slope rate compensation technique. Since its compensated slope rate is constant, its performance varies with the duty ratio.
Because the known prior arts have these drawbacks listed above, it is necessary to design an easily-manufactured, power-saving slope rate compensation technique whose slope rate is variable with the working duty ratio.

Slope rate compensation circuit, method thereof and pulse width modulation boost converter circuit


A process for generating a visible image that involves forming an electrostatic latent image upon the surface of a charge receptor medium consisting of a thin dielectric layer adjacent the metallized surface of a non-conducting substrate. The latent image is then developed to form a visible image. The thin dielectric layer may be in the form of a thin plastic film which may be delaminated from the metallized layer after image development and then laminated to the face of graphics display media such as a pressure sensitive paper. Alternately, the dielectric layer may be delaminated from the metallized layer, inverted, and laminated back to the metallized layer so that the developed visible image is sandwiched between the metallized layer and the thin plastic film. The process produces a latent image receptor comprising a support base having a metallized layer and a dielectric layer for use in producing electrographic images.

Electrography may be defined as an imaging process in which a latent electrostatic charge image is formed on the surface of dielectric medium and made visible by applying oppositely charged toner particles.
The inventor of the general technology, P. Selenyi, introduced the term "electrography". His classic studies are reported in P. Selenyi "On the Electrographic Recording of Fast Electrical Phenomena", Journal of Applied Physics, Vol. 9, p. 637-641, October, 1938. He describes forming an electrostatic image upon insulating plastic films supported on the surface of a rapidly moving metal surface. Charge was supplied by an image generating charge source mounted above the dielectric film. After charging, the latent charge image was developed and fixed. Selenyi formed images at the incredible recording speed of ten meters per second.
In U.S. Pat. No. 3,714,665, Mutschler et al disclose a recording system using charging styli to deposit a charge on ordinary paper moved in contact over the surface of a metal ground plane.
The use of a charge deposition source to form an electrostatic latent image on the surface of a dielectric film or paper which is moved over the surface of a conducting ground plane is described in U.S. Pat. Nos. 4,463,363 (Gundlach et al), 3,714,665 (Mutschler et al), and 4,521,791 (Day). Moving a relatively thick film or a paper web over a conducting ground plane is easily accomplished because of the high tensile strength of such materials. If, however, a relatively thin film is to be printed in registration using two or more print stations, film stretching and tensile strength considerations limit the effectiveness of this approach. In printing color images, which have large charged areas, the electrostatic tacking forces can be considerable.
Lewicki et al U.S. Pat. Nos. 5,124,730 and 5,187,501 describe a charge deposition printing system employing a plastic film temporarily affixed to the surface of an endless metal belt. In wide format printing, use of an endless metal belt becomes difficult because of problems associated with controlling belt tracking when belt width exceeds belt circumference.
The first commercial applications of electrographic printing employed direct charging of dielectric coated conducting base paper by an array of conducting styli by means of small air gap breakdown. In such processes, the rough paper surface provides an air effective gap spacing and this minimizes air breakdown voltage. Multiplexed driving of such an array of styli is possible since there is a well-defined minimum voltage for initiating electrical breakdown across small air gaps. Multiplexed addressing of writing styli is accomplished by connecting the styli in equal and electrical parallel groups, which are addressed simultaneously, thus significantly reducing the number of high voltage stylus drivers. The stylus groups are positioned adjacent a series of counter-electrodes which either contact a semi-conducting paper base or are capacitively coupled through the image receiving layer to the paper or film base. The counter-electrodes are electrically addressed in such a way as to select the stylus groups in sequence for writing. Each time a stylus group is selected for writing, the digital data for that group is applied to the stylus drivers groups in parallel but only the selected group actually puts latent image charge onto the dielectric receptor. All of the other, non-writing groups have their ability to write suppressed by applying write-suppressing voltage to the counter-electrodes. The net effect of multiplexed stylus driving is to reduce the cost of the drive electronics at the expense of writing speed since many groups must be written in sequence in order to write a single raster line having the length of the entire stylus array. In addition to speed reduction, multiplexed writing introduces a number of deleterious imaging artifacts, discussed below, which degrade the final visible image.
The paper or film base dielectric substrate which supports the dielectric image receiving layer must possess a certain minimum level of resistivity in order to prevent crosstalk between counter-electrodes yet a certain level of conductivity is necessary to enable writing at all. A narrow range of base paper resistivity between about 3 and 10 megohm per square is required of multiplexed, direct charging printers. This limited range of substrate resistivity is difficult to control resulting in imaging defects, described in more detail below. Typical base papers are described in Barr et al U.S. Pat. No. 4,868,048 and Reiche et al U.S. Pat. No. 3,995,083.
Recently introduced direct charging printers do not employ multiplexing since low cost highly integrated high voltage driver switches are now available. These printers may employ base paper having a much higher conductivity. Nevertheless, image defects remain which are associated with use of a resistive paper base. Among these are; image flares due to excess field emission, dropouts due to insufficient field emission, poor and variable dot formation due to charge spreading (blooming) caused by excess field emission, high background from toner stain caused by conductive salts offsetting from back side of the media, field-emission streaks in background areas from "off" styli, a limited variety of directly printable materials due to exacting electrical and mechanical characteristics needed for "contact" latent image creation, limited humidity range, lot-to-lot media variations, media which are difficult to manufacture consistently, printer contamination due to conductive coating which rubs off the back side of the media, printing speed which is limited by base sheet resistivity, and poor sheet aesthetics due to the "chemical" coating on the back side. When multiplexed writing is employed, additional image defects occur which are commonly referred to as "multiplexing striations", speeds are further reduced, and the other imaging defects inherent in all contact writing are considerably exacerbated
Early applications of small-gap-breakdown charging of dielectric paper were in the plotter field. Developments over the last several years have seen resolution increased to 16 points per millimeter, printing widths growing to 1.8 meters, and the availability of color.
Electrostatic printing technology was originally developed for text printing and CAD applications. Its advantages of speed, image durability, and good color gamut at moderate cost led to its early dominance in wide-format graphics' applications. The problems listed above, however, are currently causing its market share, in terms of both printer equipment and square footage of output, to erode in favor of simpler, but slower, methods such as ink jet. The total amount of printed output produced using electrostatic printers will continue to increase for the next several years in spite of these problems but, clearly, what is needed is a technology with the advantages of electrostatics but without its drawbacks.
Another method of forming electrostatic latent images employs a gated charge source spaced a tenth of a millimeter or more from the dielectric receptor surface. Here, a charge is formed using a low energy spark, corona wire, or silent electric discharge and the flow of these charges to the recording surface is controlled by fields other than those directly responsible for charge generation.
The silent electric discharge method of latent image charge generation has been commercially successful. This technology has been referred to as; ion printing, charge deposition printing, ion projection printing, and electron beam imaging. Fotland et al U.S. Pat. Nos. 4,155,093 and Carrish 4,160,257 disclose this charge image generation method.
Charge deposition printing provides for very high current density imaging that, in turn, allows one to print electrostatic images at extremely high speeds. Commercial printers presently operate as fast as 2.3 meters per second speed and even higher speeds have been attained in the laboratory. An image element, or dot, may be written in periods as short as 100 nanoseconds. In this case, the time constant of the dielectric receptor media must be less than about 10 nanoseconds. Since the relative dielectric constant of paper is approximately 2, the paper resistivity must be less than 5 megohm-cm. in order to prevent transient voltage drops across the paper base. Such voltage drops have the effect of reducing the extraction electric field of the charge deposition printhead
Presently employed dielectric media, and particularly dielectric coated paper, exhibit the many problems listed above when employed with either high speed charge deposition imaging systems or non-multiplexed direct charging apparatus.


Disclosed is a high-strength conductive film having excellent conductivity which can be applied in electronic devices, and which further reduces unevenness in conduction. The conductive film comprises modified fine cellulose having at least a carboxyl group and one type or two or more types of conductive substance. Further, the conductive film is formed by a process involving a step for oxidizing cellulose to prepare modified cellulose, a step for making finer the modified cellulose by dispersing the same in a dispersion medium to prepare modified fine cellulose, a step for mixing the modified fine cellulose and the conductive substance to prepare a dispersion liquid, and a step for drying the dispersion liquid to form a conductive film.

Polyethylene terephthalate (PET) is most commonly used as a base material of conductive films, and this is followed by such materials as polyethylene sulfonate (PES), polycarbonate (PC) or polyarylate (PAR), which are used less frequently than PET. In addition, although examples of conductive substances include inorganic materials such as indium tin oxide (ITO), tin oxide (SnO), zinc oxide (ZnO) or cadmium tin oxide (CdSnO4), ITO is used most commonly from the viewpoints of high conductivity and visible light transmission. Under the present circumstances, ITO-PET conductive films are applied in the form of flexible devices to various electronic devices. However, since PET uses fossil fuels in the manufacturing process thereof, it has a considerable effect on the environment with respect to reducing levels of fossil fuels, global warming and the like. In addition, since there are limitations on indium reserves, there is the risk of these reserves being depleted in the near future if indium continues to be used in this manner, and due to its price becoming extremely high, there is a pressing need to find an alternative among other materials.
On the other hand, growing attention is being focused on polysaccharides derived from natural materials for use as environmentally-friendly materials. Cellulose is contained in such sources as plant cell walls and external secretions of microorganisms or the mantle of sea squirts, is the most commonly found type of polysaccharide in the world, and is expected to be applied and deployed in carious fields due to its biodegradability, high crystallinity and superior stability and safety.
Since cellulose contains strong hydrogen bonds within its molecules and has high crystallinity, making it practically insoluble in water and ordinary solvents, considerable research has been conducted to improve its solubility. In particular, a technique that introduces a carboxyl group via an aldehyde by oxidizing the primary hydroxyl group at position C6 using a TEMPO (2,2,6,6-tetramethylpiperidinooxy radical) catalyst system makes possible to selectively oxidize only the primary hydroxyl group, and this technique has attracted attention in recent years since the reaction can also be carried out under mild conditions (aqueous system, room temperature). When TEMPO oxidation is carried out using natural cellulose, only the surface can be oxidized on the nanometer level while maintaining the crystallinity of cellulose. After washing, fine, modified cellulose can be uniformity dispersed in water by simply adding slight mechanical treatment.
An aqueous cellulose dispersion prepared using this method has a particle width on the nanometer order and is homogeneous, and as a result of drying, demonstrates high transparency in the visible light range and allows the obtaining of a film having high strength, thereby resulting in expectations for its application and deployment in various fields.
An example of a conductive base material using cellulose is described in Patent Document 1, which describes a conductive polymer composite paper produced by incorporating a conductive polymer in paper pulp. In addition, Patent Document 2 describes conductive paper obtained by mixing cellulose with a conductive substance.

Conductive film and manufacturing method thereof


The present invention relates to new crystalline molecular sieve SSZ-75 having STI topology prepared using a tetramethylene-1,4-bis-(N-methylpyrrolidinium) dication as a structure-directing agent, and methods for synthesizing SSZ-75.

1. Field of the Invention
The present invention relates to new crystalline molecular sieve SSZ-75, a method for preparing SSZ-75 using a tetramethylene-1,4-bis-(N-methylpyrrolidium) dication as a structure directing agent ("SDA") and uses for SSZ-75.
2. State of the Art
Because of their unique sieving characteristics, as well as their catalytic properties, crystalline molecular sieves and zeolites are especially useful in applications such as hydrocarbon conversion, gas drying and separation. Although many different crystalline molecular sieves have been disclosed, there is a confirming need for new molecular sieves with desirable properties for gas separation and drying, hydrocarbon and chemical conversions, and other applications. New molecular sieves may contain novel internal pore architectures, providing enhanced selectivities in these processes.

Molecular sieve SSZ-75 composition of matter and synthesis thereof