The invention relates to a method for coupling a signal to an antenna structure, as well as to an antenna structure, which comprises at least two antenna elements (101, 102), a ground plane (105) for grounding the antenna structure, a coupling line (106) for coupling a first antenna element and a second antenna element to each other, and a feeding line (107) for feeding the antenna structure through one feeding point. The first antenna element (101) is next to the ground plane and perpendicular to the ground plane (105). The second antenna element (102) is above the ground plane and parallel to the ground plane. The first antenna element is arranged to receive information on a reception band of a broadband radio system and the second antenna element is arranged to transmit information on a transmission band of said broadband radio system. By arranging the second antenna element to be adjustable and by adding antenna element to the antenna structure, the antenna structure according to the invention can be used, for example, in mobile stations of 2nd and 3rd generation mobile communication systems.

The present invention relates to small-sized microstrip antennas that operate on many different frequency bands. In particular, the invention relates to internal antennas used in mobile phones, which are fed from one feeding point.
In the present patent application, a frequency range comprises one or more frequency bands, i.e. a frequency band is part of the frequency range. Furthermore, by the reception band is meant a frequency band reserved for downlink data transmission and by the transmission band is meant a frequency band reserved for uplink data transmission.
In mobile stations, there is going on a changeover to terminals that operate in several frequency ranges. Solutions of several frequency ranges like this include so-called dual band terminals currently in use, which operate in two frequency ranges.
Dual band terminals have been implemented by both an external and internal antenna. The external antenna, which can be, for example, monopole, helix or their combination, is demanding as for its manufacturing technique, and it breaks easily. Therefore, in mobile stations, there is going on an increasing changeover to internal antenna structures implemented by microstrip antennas. The advantage of internal antennas compared to external antennas is the ease of the manufacturing technique and the speeding up of the serial production as the degree of integration increases, as well as the more durable structure than that of the external antennas.
A conventional microstrip antenna comprises a ground plane and a radiating antenna element that is insulated from the ground plane by an insulating layer. The resonance frequency of the microstrip antenna is determined on the basis of the physical dimensions of the antenna element and the distance between the antenna element and the ground plane. The operating principle and dimensioning of microstrip antennas are well known and they are described in the literature relating to the field.
FIGS. 1a and 1b show a microstrip antenna and an L-plane antenna according to prior art, which hereinafter in the present patent application will be called an L-antenna.
The microstrip antenna consists of a ground plane, a radiating antenna element, as well as a feeding line. In between and above the ground plane and the antenna element, there is either air or some other dielectric agent as an insulating material.
Traditionally, the L-antenna is a whip antenna that is bent near the ground plane parallel to the ground plane, whereupon the antenna has a low feed impedance. It is also possible to build of the L-antenna a microstrip antenna that consists of a ground plane, a radiating antenna element as well as a feeding line.
Normally, the length of the resonant proportion of the antenna in wavelengths is defined as the difference between the microstrip antenna and the L-antenna. The electric length of the microstrip antenna is half a wavelength whereas, traditionally, the electric length of the L-antenna is a quarter of a wavelength. From the electric length of the L-antenna it follows that the maximum current of the L-antenna is at the input.
Normally, the microstrip antenna is made on a double-sided substrate, one metallisation of which acts as the ground plane and on the other, the pattern of the antenna element is made by etching. The antenna element is fed by the feeding line, which is coupled to the antenna element either from one side (FIG. 1a) or by taking the feeding line through the ground plane and the insulating material (FIG. 1b). The resonance frequency of the microstrip and L-antennas is affected by the physical dimensions of the antenna element, the place of the feeding point, as well as, to some extent, the location of the antenna element with respect to the ground plane.
The size of the microstrip antenna has been reduced by developing a so-called PIFA antenna (PIFA, Planar Inverted F-Antenna), shown in FIG. 2b. In the PIFA antenna, the antenna element is coupled to the ground plane by a grounding line. This being the case, the actual antenna element can be dimensioned so that it is considerably smaller than in the case of the microstrip antenna. Furthermore, by optimising the place of the feeding point, the feed impedance of the antenna can be changed to the desired impedance level, which is not possible in the L-antenna. The resonance frequency of the PIFA antenna is affected by the physical dimensions of the antenna element and the ground plane, as well as by the distance of the antenna element from the ground plane. The antenna element is fed either from one side (FIG. 2a) or by taking the feeding line through the ground plane and the insulating material (FIG. 2b). When narrowing the width of the grounding line, the resonance frequency of the antenna decreases. The grounding line can be as wide as the whole antenna element or, at its narrowest, merely a conductor.
Furthermore, it is well known to feed a microstrip antenna capacitively. In a capacitively fed microstrip antenna, there is a feeding element in between the antenna element and the ground plane, whereupon a capacitive coupling is formed between the antenna element and the feeding element. The feeding line is coupled to the feeding element, which radiates power further to the antenna element. The capacitive coupling can be implemented both in the microstrip antenna (FIG. 3) and the PIFA antenna (FIG. 4).
The problem of microstrip antennas is the narrow bandwidth. The frequency ranges of 2nd generation mobile communication systems are reasonably narrow and, therefore, they can be implemented by microstrip antennas. For example, the frequency range of the GSM system is 890-960 MHz, wherein a transmission band is 890-915 MHz and a reception band is 935-960 MHz. Thus, the bandwidth required of one antenna element is no less than 70 MHz. Due to the production tolerances and the objects in the vicinity of the antenna, for example, the hand of a user, the bandwidth of the antenna element must be even wider. The frequency ranges required by 3rd generation mobile communication systems, for example, broadband CDMA systems are still considerably wider than, for example, the GSM system""s and, therefore, their implementation with microstrip antennas is difficult. For example, a transmission band of the WCDMA system is 1920-1980 MHz and a reception band is 2110-2170 MHz. This being the case, the whole width of the frequency range is 250 MHz. This is why the bandwidth of microstrip antennas according to prior art described above has been increased as far as possible with solutions, where several resonance frequencies close to each other are implemented in one antenna element.
Solutions are known from prior art, where several resonance frequencies close to each other are implemented in one antenna element. In one solution, the number of resonance frequencies is increased by adding slots to the antenna element. However, the slots easily act in the case of small antennas as slot radiators, whereupon antenna elements that are resonating close to each other are strongly coupled to each other and form a resonator around the slot. This further follows that at the frequency in question the radiation resistance is low and the current densities in the vicinity of the slot are high, whereupon the loss of the antenna increases. Consequently, the adding of the bandwidth of a microstrip antenna in the manner in question only succeeds at the cost of gain and radiation efficiency. Hence, with the solution in question, for example, the gain values required by 3rd generation broadband CDMA systems cannot be achieved.
Of the microstrip antennas described above, an attempt has also been made to develop antenna structures that operate in several frequency ranges. For example, an antenna structure of two frequency ranges can be implemented by one common feeding point and an antenna element the resonance frequency of which can be adjusted by a switch and an electric load to the frequency range of another mobile communication system. A second alternative is to use one antenna element and two separate feeding points, whereupon two different resonance frequencies are generated in the antenna element. A third alternative is to use two antenna elements, which are coupled to a common feeding point. In this case, both antenna elements have one resonance frequency.
FIG. 5 shows a PIFA antenna of two frequency ranges according to prior art, which is fed from one feeding point. The resonance frequency of the antenna element is adjusted either by coupling in between the antenna element and the ground plane an electric load. Alternatively, the load can also be coupled as part of the feeding line. The load can be some reactive component, for example, a capacitance or inductance. The size of the change in the resonance frequency is determined on the basis of the electric load.
A solution according to FIG. 5 is described, for example, in the publication xe2x80x9cElectrical Tuning of Integrated Mobile Phone Antennas,xe2x80x9d Louhos, J-P, Pankinaho, I, Proceedings of The 1999 Antenna Applications Symposium, Allerton Park, Monticello, Ill., Sep. 15-17, 1999. In the solution in question, it is possible to operate with one PIFA antenna element both on a transmission and reception band of the GSM900 system. The antenna element is dimensioned so that the first resonance frequency is selected from the reception band of the GSM900 system. The resonance frequency is adjusted to a lower resonance frequency by coupling the capacitive load C with a switch S between the antenna element and the ground plane, whereupon the resonance frequency of the antenna element changes to the transmission band of the GSM900 system.
FIGS. 6 and 7 describe the antenna structures described in the publication xe2x80x9cDual Frequency Planar Inverted F-Antennaxe2x80x9d (Liu Z., et al., IEEE Transactions on Antennas and Propagation, No. 10, October 1997, pages 1451-1458), wherein two resonance frequencies are implemented in one PIFA antenna.
In the solution according to FIG. 6, from a PIFA antenna E1, a part E2 is separated, which is dimensioned for a higher frequency range. The first antenna element E1 is fed from a feeding point F1 and the second antenna element E2 is fed from a second feeding point F2. Both antenna elements are grounded and dimensioned so that they have different resonance frequencies. For grounding, a plurality of ground pins G1, G2 are used. The antenna elements"" polarisations are the same.
In the solution according to FIG. 7, the antenna elements are coupled to each other, whereupon one antenna element E3 is formed, which is fed from one feeding point F3. For grounding, a plurality of ground pins G3, G4, G5 are used. In this case, in one slotted PIFA antenna, two resonance frequencies can be implemented. However, the dimensioning of the antenna elements becomes considerably more difficult, because the antenna elements are coupled to the same feeding point and the antenna elements"" gain, impedance and bandwidths depend on each other. Also in this solution, the antenna elements"" polarisations are the same.
The advantage of one feeding point compared to solutions of a plurality of feeding points is that the manufacturing of the antenna elements becomes easier and the need for contact surfaces decreases. The required area also becomes smaller. In addition, production, operators and the authorities want to measure the operation of an antenna, as well as the strength and quality of the signal transmitted and received by a mobile phone from one feeding point.
In the case of one feeding point and several antenna elements, the biggest problem is the inter-coupling of the antenna elements, which impairs the radiation efficiency of the antenna structure. Due to the inter-coupling of the antenna elements, from the antenna element that operates at a first frequency range, power is coupled to the antenna element of a second frequency range and vice versa. Therefore, in the solutions of several antenna elements in question, the harmful inter-coupling of antenna elements must be reduced in order to achieve good radiation efficiency.
In the solutions according to prior art described above, the antenna elements are parallel to the ground plane, whereupon the coupling between the antenna elements and the ground plane is highly capacitive. The capacitive coupling in turn follows that the antenna elements are unilateral. The transmitting antennas used in mobile stations should be unilateral, whereas their receiving antennas should be as isotropic, i.e. omnidirectional as possible. For example, the antenna structure according to FIG. 5 operates well when information is transmitted from a mobile station to a base transceiver station, but information transmitted by the base transceiver station should be received in all the different operating positions of the phone.
Although, in the solutions mentioned above, it is possible to change from one frequency range into another, the solutions are implemented in the GSM system, i.e. with reasonably narrow bandwidths. In addition, the antenna elements are unilateral, whereupon they do not necessarily operate sufficiently well when receiving a broadband signal. On the other hand, the problem with the antenna structure of two antenna elements fed from one feeding point is, in addition to those mentioned above, also the inter-coupling of the antenna elements. Hence, it has not been possible to implement antenna solutions required by 3rd generation mobile stations that meet the gain, radiation efficiency and bandwidth values, by microstrip antennas according to prior art.
Due to the factors mentioned above, by microstrip antennas according to prior art, it has neither been possible to implement an antenna structure comprising one feeding point that would operate optimally enough in both 2nd and 3rd generation mobile stations.
In the present invention, an antenna structure fed from one feeding point that operates on several different frequency bands with which in addition to a good bandwidth also unilaterality in transmitting and isotropy in receiving is achieved, is implemented in a new way. The antenna structure""s gain and radiation efficiency are made good by reducing the interfering inter-coupling of the antenna elements. In addition, due to the positioning of the antenna elements, the space required by the whole antenna structure is smaller compared to the antennas of a corresponding frequency range. Consequently, it is easy to position an antenna structure according to the invention, for example, inside a mobile phone or an antenna unit to be coupled to a mobile phone.
The objectives of the invention are achieved by both a new frequency band solution and a new positioning of antenna elements, which enables the implementation of an antenna structure that operates on a broad band. In the frequency band solution, the antenna""s transmitting antenna element of a lower frequency range is more unilateral than the receiving antenna element of a higher frequency range. In addition, the positioning of antenna elements according to the invention reduces the inter-coupling between at least two antenna elements, whereupon the antenna structure""s gain and radiation efficiency become good.
The basic idea of the invention is to use, instead on one transmitting and receiving antenna element, two antenna elements coupled to each other with a coupling line so that a first antenna element is used to receive information from a reception band of a first radio system and a second antenna element is used to transmit information on a transmission band of the first radio system. In a preferred embodiment of the invention, the first reception band is a reception band of some broadband CDMA system of a 3rd mobile station generation and the first transmission band is a transmission band of the same broadband CDMA system. In this way, the antenna structure is made to operate on a broad band and it is possible to operate in a broad frequency range.
According to the invention, the antenna elements are positioned so that the first antenna element, which preferably is a receiving antenna element, is on the side of the ground plane and perpendicular to the ground plane and the second antenna element, which preferably is a transmitting antenna element, is in turn above the ground plane and parallel to the ground plane. This being the case, the first antenna element can be made omnidirectional and the second antenna element unilateral. There is also little harmful inter-coupling between the antenna elements, whereupon a good gain and radiation efficiency are achieved by the antenna structure.
Harmful inter-coupling can be further reduced by designing the polarisations of the first and second antenna elements to differ from each other, whereupon a good polarisation attenuation is produced between the antenna elements.
By improving the coupling between the resonances of the first antenna and the ground plane, the efficiency and omnidirectionality of the antenna can be improved on the reception band. This can be best implemented so that the open end of the first antenna element is located in the vicinity of the upper edge of the printed board, whereupon the electric fields of the antenna and the ground plane are strongly coupled to each other at the xe2x80x9copenxe2x80x9d end of both radiators. This being the case, the antenna element acts as a feeding element for the ground plane, which acts as a main radiator.
The coupling between the second antenna element and the ground plane can again be reduced by placing the second antenna element on the ground plane so that the open end, feeding point and ground point of the second antenna element are located more in the centre of the ground plane. In this case, according to a preferred embodiment, the antenna structure can be placed in a mobile station that has, for example, a camera and a GPS antenna.
In the preferred solution, the adaptation of the first antenna element can be improved further by designing a coupling line connecting the antenna elements from the input to the second antenna element and a grounding line reaching from the second antenna element to the ground so that their common electric length is a quarter of a wavelength at the resonance frequency of the first antenna. This being the case, the first antenna element sees the grounding in question as open and the antenna operates more efficiently as a monopole-type (e.g. folded monopole) antenna. This also follows that although the grounding line of the first antenna element is slightly shorter than a quarter of a wavelength, its effect is smaller on the adaptation of the first antenna element than on the adaptation of the second antenna element and, thus, the capacitance of the first antenna element with respect to the ground plane is lower in the optimum location of the first antenna element so that radiation resistance and feed impedance of the first antenna element are sufficiently high.
The suitability of the antenna solution according to the invention for end products can be further improved with a preferred embodiment according to the invention, wherein the second antenna element is arranged to also operate in the frequency range or part of the frequency range of a second mobile communication system. In this case, for example, an antenna structure can be implemented, wherein by the first antenna element a reception band of a broadband radio system is implemented. By the second antenna element, both a transmission band of a broadband radio system and at least one transmission band of a second radio system, which is e.g. a transmission band, a reception band or both of the GSM1800 or GSMA1900 system, are implemented.
There always remains a little harmful, lossy inter-coupling between the antenna elements, which makes it more difficult to implement the second antenna element as adjustable. In the case in question, however, the implementation of the second antenna element becomes easier due to the first antenna element, because the first antenna element improves slightly the adaptation of the second antenna element at a lower resonance frequency on said frequency band of the GSM1800 or GSMA1900 systems and, thus, simultaneously adds to said bandwidth. Consequently, by the antenna structure according the invention, it is possible to implement an antenna structure that operates both in 2nd and 3rd generation mobile communication systems.
In the antenna structure according to the invention, the antenna elements do not significantly impair each other""s properties, whereupon it is easy to add to the same feeding point antenna elements that operate below and above the first transmission band. Thus, the operation of the antenna structure according to the invention can be extended, for example, into the frequency ranges of the GSM900 or PDC800 systems by using antenna elements dimensioned for the frequency ranges in question. The adding of antenna elements that operate above the first frequency range is even easier, because as the frequencies increase, the size of the antenna elements becomes smaller. It is easy to implement in the antenna structure, for example, at least one of the antenna elements of the following systems: Bluetooth, WLAN (Wireless Local Area Network) or GPS (Global Positioning System).
According to a first aspect of the invention, there is implemented an antenna structure, which comprises a first antenna element, a second antenna element, a ground plane for grounding the antenna structure, a coupling line for coupling the first antenna element and the second antenna element to each other, and a feeding line for feeding the antenna structure through one feeding point, in which antenna element (=antenna structure!), the first antenna element is next to the ground plane and perpendicular to the ground plane and the second antenna element is above the ground plane and parallel to the ground plane.
According to a second aspect of the invention, there is implemented a method for coupling a signal to an antenna structure, which comprises a first antenna element, a second antenna element, a ground plane for grounding the antenna structure, a coupling line for coupling the first antenna element and the second antenna element to each other, a feeding line for feeding the antenna structure, and which method comprises coupling signals to be transmitted and received to the antenna structure through one feeding point, the method comprising positioning the first antenna element next to the ground plane and perpendicular to the ground plane and positioning the second antenna element above the ground plane parallel to the ground plane.
According to a third aspect of the invention, there is implemented an antenna unit, which comprises an antenna structure, which antenna structure comprises a first antenna element, a second antenna element, a ground plane for grounding the antenna structure, a coupling line for coupling the first antenna element and the second antenna element to each other, and a feeding line for feeding the antenna structure through one feeding point, and which antenna structure is manufactured on an insulating material which has a base and at least one wall region, which wall region reaches in a direction deviating from the base, and the shape of which antenna structure follows the shapes of the base and the wall region, and in which antenna structure the first antenna element is next to the ground plane and perpendicular to the ground plane and the second antenna element is above the ground plane and parallel to the ground plane.
According to fourth aspect of the invention, there is implemented a mobile station, which comprises an antenna structure, which antenna structure comprises a first antenna element, a second antenna element, a ground plane for grounding the antenna structure, a coupling line for coupling the first antenna element and the second antenna element to each other, and a feeding line for feeding the antenna structure through one feeding point, and in which antenna structure the first antenna element is next to the ground plane and perpendicular to the ground plane and the second antenna element is above the ground plane and parallel to the ground plane.

Method for coupling a signal and an antenna structure


CMUT devices are used in many applications e.g. for ultrasound imaging and pressure measurement. These devices operate by sensing a change in capacitance caused by deflection of a membrane (32) comprising one of a pair of electrodes in the device by ultrasound exposure of or pressure applied on, the membrane. The CMUT device may be susceptible to the effects of changing temperature.

A CMUT (capacitive micromachined ultrasound transducer) is a type of MEMS (Micro-Electro-Mechanical Systems) device. The CMUT utilizes electrical forces for the transducer effect. Construction is usually done by manufacturing the CMUT on a silicon substrate. A cavity is formed in the region of the substrate and a thin layer suspended across the cavity to form a membrane. Electrodes are placed at the membrane and at the substrate to form an electrode pair.
Vibration of the membrane can be triggered by applying pressure (for example using ultrasound) or can be induced electrically. Electrical connection to the CMUT, often by means of an ASIC (application specific integrated circuit) facilitates both transmission and reception modes of the device. In reception mode, changes in the membrane position cause changes in electrical capacitance, which can be registered electronically. In transmission mode, applying an electrical signal casses vibration of the membrane.
CMUT devices generally operate with a biasing voltage applied. The CMUT can be operated in so called collapsed mode where the biasing voltage applied is increased above the collapse voltage to restrict the membrane and confine it against the substrate. The frequency of operation of the CMUT device is characterised by the material and physical properties of the membrane, such as the stiffness, and the size of the cavity. The bias voltage and application of the CMUT device also influence the operation mode.
A CMUT is often used in devices for ultrasound imaging applications and in other applications where the CMUT is used to detect fluid or air pressures. A pressure causes a deflection of the membrane that is electronically sensed as a change of capacitance. A pressure reading can then be derived.
A problem with CMUT devices is that a temperature change may also be the root cause of membrane deflection. In reception mode, this gives an inaccurate reading of information derived from the CMUT device. In particular, such a problem can be very limiting for CMUT based pressure sensors used in medical devices. In transmission mode, this causes a reduced efficiency of the electrical to presure power (such as ultrasound waves) conversion

Temperature compensation in a CMUT device


A developer apparatus for a printing device includes a rotatable developer carrying member which carries developer thereon. A magnet assembly is disposed in the developer carrying member. The magnet assembly includes a plurality of magnets for conveying the developer. The magnets generate a magnetic field over a larger proportion of a circumference of the developer carrying member in an edge region adjacent an end of the developer carrying member than in a mid-region longitudinally spaced from the end of the member by the edge region. The apparatus enables unwanted toner emissions to be reduced by the development of a stable annular magnetic brush in the edge regions.

The exemplary embodiment relates to printing systems. It relates particularly to a magnetic brush roll and method for suppressing unwanted toner emissions from a developer unit and will be described with reference thereto.
In typical electrophotographic image forming devices, such as copy machines and laser beam printers, a photoconductive insulating member is charged to a uniform potential and thereafter exposed to a light image of an original document to be reproduced. The exposure discharges the photoconductive insulating surface in exposed or background areas and creates an electrostatic latent image on the member, which corresponds to the image areas contained within the document. Subsequently, the electrostatic latent image on the photoconductive insulating surface is made visible by developing the image with a developer material. Generally, the developer material comprises toner particles adhering triboelectrically to carrier granules. The developed image is subsequently transferred to the print medium, such as a sheet of paper. The fusing of the toner image onto paper is generally accomplished by applying heat to the toner with a heated roller and application of pressure. In multi-color printing, successive latent images corresponding to different colors are recorded on the photoconductive surface and developed with toner of a complementary color. The single color toner images are successively transferred to the copy paper to create a multi-layered toner image on the paper. The multi-layered toner image is permanently affixed to the copy paper in the fusing process.
The toner is stored in a developer unit and delivered from the unit onto the photoreceptor by a magnetic brush system. The brush rotates and is replenished with toner and carrier granules as it reenters the developer unit. There is a tendency for toner particles to be emitted from the magnetic brush system or from the developer housing into the surrounding atmosphere. Toner emissions from a xerographic magnetic brush system can contaminate the developer and reduce the reliability of other components of the image forming device.
The toner emissions may coat surfaces which are sensitive to such contamination, causing them to malfunction. One such surface is the scorotron grid, where, for example, if too much toner contamination alights, an overvoltage condition on the charged photoreceptor occurs. This, in turn, can give rise to image quality problems or induce a sudden and uncontrolled loss of developer material into the marking engine cavity, which may necessitate a service call. Even if the level of contamination does not cause a reliability issue, for example if it alights on a handle or door visible to the customer, it can present an impression of a poorly operating machine.
Unwanted or uncontrolled toner emissions from within the developer housing can often be inhibited by implementation of various sealing solutions. Emissions from the brush itself have proved more difficult to control. In particular, toner may escape at the edges of the magnetic brush into the atmosphere.

Local suppression of unwanted toner emissions


A ring optical transmission network access node (ANi) comprises:                two identical band splitters (BSa, BSb), each provided with homologous first, second and third ports (Ba, BWa, BPa; Bb, BWb, BPb),        a first coupler (CW) coupling the second port (BWb) of the second splitter (BSb), on the one hand, to the second port (BWa) of the first splitter (BSa) via a first channel and, on the other hand, to an access point (A2) via a second channel,        a second coupler (CP) coupling the third port (BPa) of the first splitter (BSa), on the one hand, to the third port (BPb) of the second band splitter (BSb) via a first channel and, on the other hand, to another access point (A1) via a second channel.

1. Field of the Invention
The field of the invention is optical connection transmission networks. The invention relates more particularly to optical networks adapted to relatively limited geographical dimensions, such as metropolitan access networks.
2. Description of the Prior Art
As a general rule, an optical network consists of a plurality of stations able to send and receive optical signals to and from other stations of the network. These exchanges of information are effected by means of optical connections to which are connected access nodes that serve the respective stations.
To exploit the bandwidth capacity of the optical connections, wavelength division multiplexing (WDM) is advantageously used. Accordingly, the optical connections carry multiplex signals formed of a combination of optical signals each consisting of an optical carrier wave modulated as a function of the information to be sent. Each carrier wave has a specific wavelength that defines a corresponding spectral channel.
Moreover, if the network is of sufficiently limited size, providing devices for individual regeneration of the channels may be avoided. Such a network, which is then referred to as "transparent", may nevertheless include optical amplifiers disposed to amplify simultaneously the channels of the WDM multiplexes transmitted. In the absence of such amplifiers, the network is referred to as "passive".
Accordingly, the field of the invention is that of transparent networks, passive or otherwise.
A first type of prior art transparent WDM network can use a ring configuration. The network then includes, for example, an optical connection one end whereof is coupled to a send interface of a hub and the other end whereof is coupled to a receive interface of the same hub. The hub is also normally adapted to communicate with an external interconnection network.
FIG. 1 represents one example of such a network diagrammatically in the most simple situation where the looped connection consists of a single fiber F to which are coupled access nodes AN1-AN3 for receiver terminals RX and sender terminals TX of associated stations ST1-ST3.
The optical connection therefore consists of a plurality of fiber sections FS1-FS4 separated by the nodes (and where applicable by optical amplifiers, not shown). The connection has a first end E1, called the upstream end, connected to the send interface HT of the hub HUB and a second end E2, called the downstream end, connected to the receive interface HR of the hub.
The send interface HT is provided with a plurality of senders using carrier waves with different send wavelengths. Also, each station includes at least one receive-wavelength-selective receiver. Accordingly, each sender of the hub can inject into the connection a signal of given wavelength and when that signal reaches an access node via the fiber, that signal may be processed by the associated station if it includes a receiver tuned to that wavelength.
Conversely, the receive interface HR of the hub is provided with a plurality of respective receive-wavelength-selective receivers, and each station includes at least one sender TX of given send wavelength. Accordingly, a sender of a station can inject into the connection, toward the second end E2, a signal of given wavelength and when that signal reaches the receive interface, it can be processed by the hub thanks to one of its receivers sensitive to that wavelength. Given that all the signals propagate in the connection in the same direction, measures must be taken to avoid interference and conflicts at the level of the receivers. For example, a rule may be imposed whereby all the send wavelengths of the hub and of the stations are all different from each other. Time-division multiplexing may also be used for signals sent by a plurality of senders that would be carried by the same wavelength. Each station could include a receiver terminal and a sender terminal respectively consisting of a plurality of receivers and a plurality of senders tuned to a plurality of wavelengths.
Exchanges of information between stations are effected via the hub in the following manner. Each sender station sends the hub a signal carried by one of the receive wavelengths of the hub. That signal contains an address indicative of the destination station. After reception of the signal and its conversion to electrical form by the hub, the destination address is processed by the management means of the network to determine a receive wavelength of the destination station. The signal is then reconverted to optical form by means of a carrier wave having that wavelength and sent as a downlink signal.
All signals on the connection coming from the hub constitute information traffic known as "downlink" traffic and all signals on the connection coming from the stations constitute "uplink" information traffic.
In the embodiment shown diagrammatically in FIG. 1, the nodes ANi (where i is equal to 1, 2 or 3 in the example represented) each consist of a simple 2-to-2 type coupler. Each coupler has a first input connected to the upstream section FSi, a first output connected to the downstream section FSi+1, a second input connected to the senders TX of the associated station and a second output connected to the receivers RX of that station. Hereinafter, the first inputs and outputs are called the first and second "connection points" of the node to the connection and the second inputs and outputs are called first and second "access points" of the station to the connection.
Thus two adjacent sections FSi, FSi+1 are coupled to each other via a first channel of the 2-to-2 coupler, the upstream section FSi is coupled to the receiver terminals RX via a second channel, and the sender terminals TX are coupled to the downstream section FSi+1 via a third channel, these three channels enabling couplings that are not wavelength-selective.
Consequently, each station may insert into the downstream section signals produced by a variable number of senders TX, tuned to any wavelength. Similarly, for reception, each station may process a variable number of signals by providing the same number of photodetectors coupled to the outputs of means for filtering (or demultiplexing) signals received from the upstream section. It is therefore a simple matter to change the capacity of the network by adding senders and photodetectors, without disturbing traffic in transit through the node. Note that a plurality of stations can process the same channel, to enable the broadcasting of the same signal to a plurality of receivers of different stations.
This embodiment provides great flexibility in the choice of send and receive wavelengths.
In a second type of network (not shown) similar to the previous one, the uplink and downlink traffic are contra-directional, i.e. propagate in opposite directions. In this case, the senders and the receivers of the hub are coupled to the same end of the fiber, for example the end E1, and for the downlink traffic each node ANi performs the function of a "1-to-2" coupler from the hub, via the end E1 and the upstream sections, to the downstream section, on the one hand, and to the receiver terminal of the associated station, on the other hand. Conversely, for the upstream traffic of signals going from the sender terminal to the hub, each node ANi performs the function of a "2-to-1" coupler in the opposite direction, i.e. from the senders of the stations to the receivers of the hub, via the same end E1. It may be noted that the "1-to-2" and "2-to-1" coupling functions referred to above may be effected as in the preceding embodiment by means of a 2-to-2 type directional coupler of which only three ports are used, the senders TX and receivers RX of the associated station being connected to the same second access point. It may also be noted that this second type of network no longer has a ring physical topology but instead a tree topology. In this second type of network, the end E2 is not connected to the hub. In fact, all the senders and receivers of the hub are connected to the same end E1.
As in the first embodiment, the nodes provide couplings that are not wavelength-selective. This latter embodiment therefore has the same advantages in terms of the flexibility in the choice of send and receive wavelengths. It further enables the use of common wavelengths for carrying uplink and downlink traffic.
Other considerations must be taken into account in network design, however, in particular the design of transparent and passive networks.
An important aspect in the dimensioning of a network is the "power budget", i.e. the processing of the minimum permissible optical powers for the photodetectors of the receivers, and consequently the send powers to be provided and/or the maximum possible lengths of the connections between senders and receivers.
In this regard, the last embodiment mentioned has the advantage that it makes it possible to optimize the optical power budget by appropriate dimensioning of the coupling ratios of the various 1-to-2 couplers constituting the nodes of the network. For example, in the case of a network with no in-line amplification, the optimized coupling ratios may be calculated in the following manner.
There is taken as the minimum receive power a common value for all of the receivers of the network, that value being sufficiently high to take into account the spreads of the characteristics of the photodetectors and aging phenomena. Similarly, the senders in the same network are normally chosen to deliver substantially the same power, which also makes it possible to define a nominal send power value common to all the senders of the network.
Given these conditions, for a network having N nodes, for example, there will be chosen N couplers the N coupling ratios whereof have respective values such that the various optical paths via these couplers between the senders and receivers of the hub and the senders and receivers of the terminals have transmission coefficients having substantially the same common value. This is reflected in N equations in which the unknowns are the N coupling ratios and the parameters are the transmission coefficients of the elements constituting the network.
As the uplink and downlink traffic take the same paths, the coupling ratios are optimized for both types of traffic.
On the other hand, this kind of optimization is generally not possible in the case of the first network. In fact, except in special circumstances, if couplers are chosen the coupling ratios whereof are optimized for the downlink traffic, for example, the coupling ratios for the uplink traffic are then fixed, generally at non-optimum values, and vice-versa.
Another important aspect of network design relates to protection against faults or interruption of the connection.
In the case of ring topologies such as that of FIG. 1, prior art solutions are based on the use of a connection consisting of a plurality of optical fibers, which are normally part of the same cable.
One possibility is to provide two fibers respectively dedicated to uplink and downlink traffic, for example. The senders of the hub are then each coupled to the two ends of the fiber dedicated to downlink traffic so that the downlink signals propagate in the fiber in opposite directions. Similarly, the senders of each station are coupled to the fiber dedicated to uplink traffic at two connection points such that the uplink signals propagate in the fiber in opposite directions also.
Means are then provided in each station for routing received couples enabling each receiver to be selectively coupled to one or the other of the coupling points of the station to the fiber dedicated to downlink traffic. Similarly, the hub is provided with means for routing received signals for selectively coupling each of its receivers to one or the other of the two ends of the fiber dedicated to uplink traffic.
Accordingly, if a section of the ring is broken, appropriate switching of the routing means enables the exchanges of signals between the hub and each station to be maintained.
An alternative solution consists in duplicating the network by dedicating one fiber, called the "working" fiber, to the normal mode of operation and the other fiber, called the "protection" fiber, to the protection mode of operation.
There are then provided in the hub and in each station means for routing the signals sent that selectively couple according to the mode of operation each sender to one or the other of the fibers. There are similarly provided means for routing received signals that selectively couple each receiver to one or the other of the fibers.
The two networks advantageously each conform to the second type described hereinabove i.e. use as access nodes optimized 2-to-1 (or 1-to-2) couplers adapted to route bidirectional uplink and downlink traffic.
Given that the working and protection fibers normally belong to the same cable, a break occurring in one section generally leads to unavailability of both fibers in the same section. It will therefore be necessary in each station for the access couplers to the respective fibers to be disposed symmetrically. Thus, in protection mode the uplink and downlink traffic can always propagate between the hub and each station by using fiber sections that are not faulty (provided of course that only one section is faulty).
Although these solutions involving a plurality of fibers are effective in terms of protection, they lead to an overcost that it is desirable to be able to avoid.
To this end, there may be envisaged a protection system applied in the case of a single-fiber connection. The principle is analogous to the preceding situation, except for switching the coupling of the signals sent from one fiber to the other. There has to be used for each station an access node to the same fiber that consists either of a single 2-to-2 coupler or of two 2-to-1 couplers disposed symmetrically.
In the former situation, the coupling structure is that of the network from FIG. 1 with the drawback already indicated: it is generally not possible to dimension the couplers so that they have coupling ratios optimized both for traffic exchanged between each station and the first end E1 of the fiber, on the one hand, and between each station and the second end E2, on the other hand. The same drawback applies in the latter situation.
An object of the invention is to propose a solution enabling the implementation of a network in which are reconciled properties that are incompatible with the prior art networks described hereinabove, i.e. a network provided with a single-fiber connection, in a ring configuration enabling protection in the event of interruption at one point of the connection, and in which to access the fiber the stations use couplers having coupling ratios optimized both in the normal mode of operation and in the protection mode, i.e. both for traffic exchanged between each station and the first end E1 of the fiber, on the one hand, and between each station and the second end E2, on the other hand.

Ring optical transmission network access node

Abstract: Such nodes enable the implementation of a passive ring network comprising only one fiber and in which the coupling ratios of the couplers (CW, CP) are optimized for two opposite traffic propagation directions.

Abstract: Application in particular to optical fiber metropolitan access networks, with broken fiber protection device.


A template for radiography and methods of using and making a template are described. The template can include a contoured sheet having first portions and second portions. The first portions are radiodense and the second portions are radiolucent to provide markings on a radiograph. At least one surface of the contoured sheet can correspond to at least one surface of a target part to provide alignment of the contoured sheet to the target part.

The statements in this section merely provide background information related to the present disclosure and may not constitute prior art.
Defects in castings can form as a result of the casting process. Internal defects may not be found from merely inspecting the surface of the casting. Therefore, radiography is often used to inspect castings in order to identify and locate internal defects that a casting may have.
Radiography includes exposing the casting to x-rays in order to produce a radiograph. The radiograph can be reviewed to see if the casting has any defects. However, accurately locating the defects in the casting that are found on the radiograph can be difficult. For example, the radiograph may be a different size than the casting. Furthermore, the orientation of the casting when the radiograph was taken may be difficult to determine.
In order to more accurately locate internal defects of a casting, lead indicators can be taped to the casting. However, numerous radiographs may need to be taken when using lead indicators. A first radiograph is typically taken without any lead indicators taped to the casting. Lead indicators may then be taped to the casting around where the defects are believed to be located, and a second radiograph can be taken. Therefore, more than one radiograph is typically needed to accurately identify locations of defects.
An additional radiograph can also be taken with lead indicators in the area of the defect to perform a parallax shot with the intent of determining the defects relative depth. Parallax principles and methods are further described in Barry, R. and Ruescher, E., "Stereo Radiography," Nondestructive Testing Handbook, third edition, Vol. 4, Radiographic Testing, Columbus, Ohio, American Society for Nondestructive Testing (2002), p 419-423.
After defect locations are identified, the defect locations of the casting can be reworked or remedied. Although the methods described above can identify locations of defects, at least two radiographs typically need to be taken of the casting in order to obtain an accurate location of the defects which can add cost and complexity to the inspection process of the casting.


A non-invasive gas analyte sensing and monitoring system and method is provided that is particularly applicable to transcutaneous monitoring of arterial blood gases in a mammal. The system and method relies on diffusion of the analyte to be measured into a diffusion chamber and remote sensing of the analyte in the diffusion chamber using optical chemical sensors and associated optoelectronics.

1. Field of the Invention
The present invention relates to non-invasive sensing of substances such as, for example, non-invasive sensing of transcutaneous blood gases for respiratory management in patients.
2. Background of the Related Art
The Background of the Related Art and the Detailed Description of Preferred Embodiments below cite numerous technical references, which are listed in the Appendix below. The numbers shown in parenthesis at the end of some of the sentences refer to specific references listed in the Appendix. For example, a "(1)" shown at the end of a sentence refers to reference "1" in the Appendix below. All of the references listed in the Appendix below are incorporated by reference herein in their entirety.
Challenges of Preterm Birth
Optimal cellular functioning depends on an adequate supply of oxygen and acid-base balance. Highly complex biologic systems have evolved to maintain cellular functions during aberrant conditions of low and high oxygen exposures and to correct acid-base imbalance. When these systems are overwhelmed, cellular damage occurs. The fetus, which is dependent on the placenta for gas exchange, is exposed to relatively low oxygen concentrations and anti-oxidant defense mechanisms are under-developed (1).
Preterm birth requires technologies for supporting immature organ systems on the one hand, while minimizing technology-induced injury on the other that may contribute to long-term adverse outcomes. Specific problems of this kind are extremes in arterial partial pressure of oxygen (PaO2) and carbon dioxide (PaCO2). A variety of pathological conditions are known to result from both hypoxemia and hyperoxia. However, current understanding is that the range for acceptable PaO2 is quite narrow. Similarly, hypocarbia or hypercarbia can result in brain or ocular pathology. The optimal range of PaCO2 has not been established, but avoiding extremes is critical for survival.
Consequences of Hypoxia and Hyperoxia
Severe hypoxemia that can occur in utero as a result of acute or chronic placental dysfunction, or postnatally in infants with cyanotic congenital heart disease or neonatal lung disease, may cause brain injury, multi-organ failure, pulmonary hypertension, and death. Although oxygen therapy for newborns has been used in the United States since the 1930s, its relationship to retinopathy of prematurity (ROP) (previously retrolental fibroplasis), a potentially blinding disease, was not recognized until the 1950s (2). Fluctuations in arterial partial pressure of oxygen (PaO2)(i.e. swings from hypoxia to hyperoxia) also increase the risk for ROP, underscoring the need for monitoring devices that are responsive to rapid changes in PaO2 (3). Efforts to define an optimal oxygen target range to minimize the incidence of ROP continue to the present (4, 5, 6).
Pulmonary oxygen toxicity contributes to bronchopulmonary dysplasia (BPD), the major respiratory morbidity of prematurity characterized by an arrest of alveolar development (7, 8). As summarized by Finer and Leone, "the therapeutic index for oxygen use seems to be much narrower than previously realized, and we have yet to define the optimal oxygen exposure for the most premature infants" (1).
Consequences of Hypocarbia and Hypercarbia
Extremes in arterial partial pressure of carbon dioxide (PaCO2) also contribute to neonatal morbidities. Hypocarbia in the first few days of life may contribute to altered cerebral blood flow in the preterm infant. Potential consequences include intraventricular hemorrhage (IVH), periventricular leukomalacia (white matter injury), and cerebral palsy (9, 10, 11, 12). Hypocarbia resulting from overventilation contributes to volutrauma and the development of BPD (13). Permissive hypercapnea has been proposed as a strategy to limit lung injury, but the safety of this practice has not been established (14, 15, 16). Indeed, early exposure to hypercarbia and associated acidosis increases the risk for severe ROP and IVH (17, 15).
The optimal range of PaCO2 has not been established in the neonatal population, but clearly, rapid recognition of and response to extremes in PaCO2 is critical for survival and minimizing morbidities.
Status of Neonatal Blood Gas Monitoring
For the reasons discussed above, arterial blood gas (ABG) measurements are indispensable for respiratory management in the neonatal intensive care unit (NICU). Although arterial blood gas (ABG) measurements remain the gold standard for guiding respiratory management in the NICU, the necessity for placement of indwelling arterial lines or intermittent arterial or heel blood sampling are associated with potential complications such as infection, thrombus formation, bleeding, and procedure-associated pain. In addition, the frequency of blood sampling often necessitates blood transfusions to replace blood volume. Further, ABGs, at best provide only intermittent information concerning dynamic changes in blood gases.
To address these concerns, non-invasive devices have been developed to continuously monitor blood gases. In the 1970s, transcutaneous monitors were developed to measure the amount of O2 (tcPO2) and CO2 (tcCO2) dissolved in tissue that approximated arterial values when the skin underneath the sensor was heated to 43-44° C. (18). Disadvantages included the potential for skin burns, necessitating frequent sensor site changes and re-calibration. Accuracy was affected by skin thickness, peripheral perfusion, and use of vasopressors (18).
Since its introduction in the 1980s, pulse oximetry, which measures the proportion of hemoglobin in arterial blood that is bound to oxygen, has largely replaced tcPO2 monitors in most NICUs. Pulse oximeters are easy to use, and do not require heating the skin or calibration. However, false alarms may occur due to motion artifacts. In addition, due to the shape of the hemoglobin-oxygen dissociation curve, small changes in oxygen saturation (SpO2)>95% can fail to detect large increases in PaO2, thus limiting the usefulness of pulse oximetry for detection of hyperoxia (19). In addition, compared to SpO2 monitoring, tcPO2 monitoring was associated with less variability in oxygen tension and greater proportion of time spent within the oxygen target range (19).
TcPCO2 monitoring is more accurate than end-tidal CO2 monitors in patients with shunt or ventilation/perfusion inequalities, and can be used in situations such as high frequency or non-invasive ventilation where ET-CO2 cannot (20). Recently, a SpO2/tcPCO2 combination sensor and a neonatal sized tcO2/tcPCO2 combination sensor based on current technology have been introduced (21, 22).
To address some of the above concerns, transcutaneous monitors for O2 (tcpO2) and CO2 (tcCO2) using electrochemical sensors have been used. However, these electrodes have at least the following disadvantages: (1) they suffer from calibration drift due to depletion of the electrolyte; (2) they lack sensitivity at low O2 due to consumption of O2 by the Clark electrode; (3) the potential for membrane failure; (4) the use of adhesives to maintain direct skin contact; and (5) the possibility of burns from raising skin temperature to 43° C. for tcpO2 monitoring. The last two issues, in particular, can lead to serious skin injury. Given the drawbacks of current technologies, it is clear that a new generation of neonatal monitoring devices is needed that is: (1) non-invasive/painless; (2) more accurate and sensitive than current sensors; (3) easy to use by staff; and (4) and can follow dynamic changes in blood gases for longer periods of time.

Non-invasive analyte sensing system and method


An efficient method and apparatus that operates directly upon compressed blocks of image data to transform them into compressed blocks of scaled image data. Decompression of the compressed blocks of scaled image data yields blocks of scaled image data, which are then used to display a scaled down version of an original image. The present invention is more computationally efficient than a previously known scaling method, which required that compressed blocks of image data be uncompressed prior to any scaling operations.

Digital images and video are often transmitted and stored in a compressed data format, conforming to compression standards such as the MPEG and JPEG standards for motion and still images. Both of these standards utilize the discrete cosine transform (DCT) to compress eight by eight (8.times.8) blocks of image data. In this context, there often arises the need to scale down an image that is provided in a compressed data format in order to achieve a suitable image.
For example, where an image is to be sent in compressed data format to receivers of different computational and display capabilities, it is necessary to scale down the image to match the capabilities of each receiver. As another example, in multi-party video conferencing, an image of each conference participant is transmitted in compressed data format. Down-scaling is needed because the images are displayed together in a mosaic comprising a large image of the current speaker and smaller, scaled down, images of the other conference participants.
A previously known method for scaling down an image provided in a compressed data format includes: un-compressing the compressed blocks of image data, scaling down the blocks of image data to produce one or more blocks of scaled image data, and then compressing the blocks of scaled image data.
For example, given an image in a DCT compressed data format of four compressed 8.times.8 blocks of image data, scaling down the image by a factor of two using the previously known method requires: four 8.times.8 inverse discrete cosine transforms to un-compress the compressed blocks of image data, 192 add operations and 16 shift operations for bilinear interpolation to scale down the blocks of image data into one 8.times.8 block of scaled image data, and an 8.times.8 discrete cosine transform to compress the block of scaled image data. Since a full 8.times.8 discrete cosine transform (or its inverse) requires 80 multiply operations and 464 add operations, and since a multiply operation can be represented as 4 shift operations and 3 add operations, in this example the previously known scaling method requires a total of 5,276 operations, comprising 1,704 shifts and 3,572 adds. Such a large number of operations is computationally time consuming and is therefore expensive.
Because the previously known method must first decompress the compressed blocks of image data before any scaling operations, the previously known method is computationally inefficient. What is needed is an efficient method and apparatus that operates directly upon compressed blocks of image data to transform them into compressed blocks of scaled image data.

Method and apparatus for scaling down images that are provided in a compressed data format


A light-activated illuminating member having a support member and a phosphorescent material supported thereby. The phosphorescent material can form a luminescent image having a varying luminescent effect after the phosphorescent material is exposed to a light of constant intensity. One or more such light-activated illuminating members can be assembled with a light source, such as a UV-LED pen, and be formed into a book-type writing/drawing toy which is capable of providing a varying luminescent effect. When a plurality of such illuminating members are used in the book-type toy, such a toy is capable of providing a different, combined luminescent effect.

Conventional drawing boards, such as Etch-A-Sketch®, MagnaDoodle®, or LiteBright®, provide a writing/drawing surface allowing users to thereby serve amusement and education function to users, especially to children. However, such drawing boards can not be used in a dark environment because the images formed on the drawing boards cannot be seen without additional lighting aid.
Luminescent materials have been used to aid in the writing and reading in the dark. For example, U.S. Pat. No. 4,590,381 issued to Mendelson provides a phosphorescent marking system on which luminescent images are formed and retained for a period of time.
The present invention provides a luminescent device capable of forming a luminescent image having a varying luminescent effect. Additionally or alternatively, the present invention provides a writing/drawing toy kit capable of forming a luminescent image that has a varying luminescent effect so as to provide a comprehensive entertainment and education purpose.


An apparatus, system, and method are disclosed for managing configuration parameters of a non-volatile storage device. The method includes storing a first set of configuration parameters for a non-volatile recording device. The first set of configuration parameters are configured for a storage operation on a storage element of the non-volatile recording device. The method also includes storing a second set of configuration parameters for the non-volatile recording device during execution of the storage operation on the storage element. The second set of configuration parameters are configured for a second storage operation on the storage element of the non-volatile recording device.

Configuration parameters may be used to manage various properties of a storage device. Configuration parameters often may be set during operation of the storage device. However, a storage device may take time to set individual parameters. When many parameters are being set, the aggregated time to set each individual parameter may form a significant delay.
There are several scenarios where multiple configuration parameters may be set for a storage device, even for a single storage operation. For example, properties of a storage device, such as stored values, read voltage levels, resistivity, or the like, can shift over time due to storage cell damage, storage cell leakage, temperature, or other disturbances that alter the state of the storage cells. Therefore, to counter or rectify such disturbances, customized configuration parameters may be used for a storage operation, such as customized read voltage thresholds or the like. However, setting different configuration parameters per storage operation may have a negative impact on performance, due to the delay described above.

Managing configuration parameters for a non-volatile medium


A filtration system is provided with filter media operable to remove solids, particulate and colloidal matter from a process fluid. The filtration system may include one or more porous membranes which are anchored or secured on their periphery with a large open work area operable to remove selected components from the process fluid. The filter media may include an envelope formed from two membrane sheets with a spacer element disposed there between. A chamber or collection zone may be formed within the envelope to receive clarified fluid separated from the process fluid. Alternatively, the filter media may be multiple layers of spiral wound membranes or a roll of membranes.

An ongoing goal in the filtration industry is to perform filtration continuously for sustained periods, even when processing fluids with high amounts of solids or colloidal materials. A wide variety of filter media designs and configurations have been used in attempts to provide a continuous filtration process. This goal has led to several known techniques for continuously inhibiting the buildup of scale, cake or films which tend to deposit on and block passage of desired fluid flow through associated filter media. In some cases, these techniques are used intermittently, to perform what is called cyclic cleaning of filter media surfaces, usually when an associated filtration process has been suspended for such cleaning.
In order to achieve sufficient surface area to serve in commercial scale filtration applications, multiple layers of filter media such as flat sheet membranes are often collected together within a single filtration device. Combining a plurality of sheet membranes may include arranging them in parallel stacks, mounted either horizontally or vertically, or, winding a relatively long single sheet into a spiral configuration.
Various techniques for continuous and cyclic cleaning of associated filter media have been used. U.S. Pat. Nos. 4,872,988; 4,952,317; 5,014,564 and 5,725,767 teach using reciprocating torsional vibration applied parallel to planes of stacked filter elements. Flat filter leaf elements may be stacked within and rigidly attached to a pressure vessel, whereby the entire filtration device is torsionally shaken in a back and forth or reciprocating manner generally parallel to the face of the leaf elements and orbitally around a vertical axis through the center of the stack. These reciprocating orbital shaking motions may produce boundary layer turbulence or interference on upstream surfaces of each membrane or leaf element to continuously scour associated surfaces of any scaling, cake or film. In order to transmit and to withstand the relatively violent motions being applied by external vibration devices, the leaf elements may include a support mounted at an interior surface of the associated membrane. The membranes may be sealed or laminated to each other over a support element or plate. The support element may be constructed of a relatively rigid plastic material, metal or other suitable materials.
U.S. Pat. No. 6,322,698 teach vibratory cleaning by applying reciprocating torsional vibration parallel to the planes of stacked flat filter elements. The flat filter elements includes frame elements, multiply-stacked, with outer edges sealingly combine to form an integral pressure vessel. The entire stacked frame device may be torsionally shaken parallel to the planes of the filter elements. This shaking motion may produce boundary layer scouring for the purpose of preventing accumulation of surface scaling, caking or filming on upstream surfaces of associated filter media.
U.S. Pat. No. 6,287,467 teaches cleaning parallel mounted, flat leaf filter elements using air bubbles forcefully directed between and sweeping across the upstream surfaces of their filter leaf elements. A solid plate-like stiffening element may be disposed parallel to the filter media of each element to provide overall stiffness with desired spacing between upstream surfaces through which cleaning air bubbles are directed. Conventional filter element design and construction often includes rigid support structures with internal reinforcing plates and/or rigid mechanical elements disposed within the central work area of such filter elements

Filtration system and dynamic fluid separation method


A crease ironing apparatus is provided that is adaptable to be either attached to an existing ironing board or as a stand alone apparatus, wherein the apparatus includes two elongated members, while one member's longitudinal transverse cross section resembles a U forming a longitudinal trough and the other member's longitudinal transverse cross section is dimensioned to fit within the trough of the other member along with the fabric to be clamped while the two members are hingedly secured at one end allowing the two members to scissor together or apart to receive fabric to be clamped while further comprising a means for securing the two members in the clamping position and if desired a means for clamping the device to an existing ironing board.

Hand ironing creases that are both straight and aligned with seams in trouser legs, shirt sleeves and skirts is extremely difficult. The problem encountered by those attempting to iron straight creases arises from the difficulty of securing the garment in position with one hand while ironing with the other. Securing the garment usually takes more than one hand especially when long creases such as trouser legs are attempted to be ironed. The present invention solves this problem by securing the garment in position by clamping both sections of fabric forming the crease in alignment while the crease is ironed. The crease ironing apparatus is provided as an ironing board attachment which is either conveniently clamped to an ironing board or not clamped as an independent device which allows the entire garment to be moved around while the device clamps fabric sections in alignment for creases to be ironed.
Numerous prior art devices have been patented for attaching to an ironing board to improve ironing for certain garments or conditions. Breen et al, U.S. Pat. No. 5,016,367 discloses an ironing board with two swingable board extensions for assisting in ironing all portions of trouser legs. Theeten, U.S. Pat. No. 4,779,365 discloses an ironing board with two parallel ironing surfaces useful for ironing clothes of varying dimensions. Vennen, U.S. Pat. No. 2,888,759 discloses an ironing board with a built-in midget board permitting the ironing of various items which are ordinarily too difficult to iron on a large ironing surface. Hipp, U.S. Pat. No. 2,869,259 discloses an ironing board attachment comprising an apron sling adapted transversely under the ironing board to receive clothes to be iron keeping the clothes off the floor. Kleinsorge, U.S. Pat. No. 2,729,906 discloses an ironing device attachable to an ironing board which is adapted for ironing garments having curved surfaces. And finally, Hedges, U.S. Pat. No. 2,554,983 discloses an ironing board attachment which includes an adjustable support means adjacent to the side of the ironing board which can be extended in an outward and upward direction to support large articles when ironing such as sheets, and curtains. These prior devices in the field of art concerned herein are extremely useful, however these prior devices do not provide an ironing board attachment means for ironing straight and aligned creases in garments.


Substituted hydrofluoroalkylphenol compounds and methods of using the compounds, e.g., for anesthetizing a subject, are disclosed.

Alkyl phenols have a broad range of medicinal properties ranging from central nervous system (CNS) effects to antioxidant activities. The effects of alkyl phenols on the CNS are generally sedative in nature. For example, the dialkylphenol, propofol (2,6-diisopropylphenol) is used as an anesthetic agent in both humans and animals. This compound also serves as a muscle relaxant, anti-epileptic, anti-emetic, anti-spasmotic and as a bronchodilator.
Diisopropylphenol also has uses in the treatment of pathologies relating to the presence of free oxygen radicals (see, see, e.g., U.S. Pat. No. 5,308,874; U.S. Pat. No. 5,461,080; and Aarts et al., 1995, FEBS Let. 357(1): 83-5). For example, propofol has been used to inhibit inflammatory responses of the upper respiratory tract due to oxidative stress (see, e.g., Zaloga et al., The Internet Journal of Emergency and Intensive Care Medicine ISSN 1092-4051; Borgeat et al., 1992, American J. of Gastroent. 87(5): 672-674). Propofol also has been shown to repair neural damage caused by free oxygen radicals in vitro (Sagara et al., 1999, J Neurochem. 73(6): 2524-30; Jevtovic-Todorovic et al., 2001, Brain Res. 913(2): 185-9) and has been used in vivo to treat head injury (see, e.g., Kelly et al., Journal of Neurosurgery 90: 1042-1052, 1999
Although propofol is considered to be relatively safe, side effects have been reported, including dose-dependent hypotension, hypertriglyceridemia and pancreatitis. In addition, high doses of propofol can result in "propofol syndrome", a serious condition characterized by severe metabolic acidosis and circulatory collapse.

Hydrofluoroalkyl phenols having anesthetic properties


The output of a differential amplifier circuit group (PA) that amplifies signals inputted to input PINs is connected to a final-stage differential amplifier circuit (PAn). The output of the differential amplifier circuit (PAn) is connected to a detection (DET) circuit. A detect signal sent from the DET circuit is outputted to the (−) side input of a comparator. A bias signal (BP) outputted from a bias circuit is inputted to the base of a PMOS transistor of a source follower circuit. An output signal (SFOUT) outputted from a source terminal thereof is inputted to the (+) side input of the comparator. A result of comparison between the bias signal (BP) and the output signal (SFOUT) is outputted from the comparator as an output signal (COMPOUT).

1. Field of the Invention
The present invention relates to a semiconductor integrated circuit including differential amplifying means for amplifying input signals.
This application is counterpart of Japanese patent application, Serial Number 36659/2004, filed Feb. 13, 2004, the subject matter of which is incorporated herein by reference.
2. Description of the Related Art
A configuration of a differential amplifier circuit having a conventional constitution will be explained with reference to FIG. 13. The circuit 1300 shown in the same figure includes a differential amplifier circuit group 1310 and a final-stage differential amplifier circuit 1320, a detection (DET) circuit 1330 that detects an output 1320 of the differential amplifier circuit, and a bias circuit 1340. The output of the bias circuit 1340 is connected to a comparator 1350.
The differential amplifier circuit group 1310 is a circuit group in which a circuit similar to the differential amplifier circuit 1320 is included thereinside and connected in plural form. Therefore, an internal configuration of the differential amplifier circuit 1320 will be described. In the differential amplifier circuit 1320, a constant current circuit (PTn) connected to a power supply VDD is connected to respective source terminals of a PMOS transistor P(n1) and a PMOS transistor P(n2). Drain terminals of these transistors are connected to a ground GND via a resistor PR(n1) and a resistor PR(n2) respectively. Incidentally, "n" indicates the whole number indicative of the number of connecting stages of the differential amplifier circuits 1320.
An input PIN(1) or an output P0(n1) from a pre-stage circuit is connected to the PMOS transistor P(n1) and an input PIN(2) or an output P0(n2) from a pre-stage circuit is connected to the PMOS transistor P(n2).
Connecting points of the drain terminals of the PMOS transistors P(n1) and P(n2) and the resistors PR(n1) and PR(n2) respectively form the outputs P0(n1) and P0(n2) of each differential amplifier circuit and are connected to the input of the nest-stage differential amplifier circuit. An output P0(n2) of the differential amplifier circuit placed in the final stage is connected to the detection circuit 1330.
In the detection (DET) circuit 1330, a source follower circuit is formed by both a constant current circuit (PT) 1332 connected to the power supply VDD, and a PMOS transistor 1334 having a source terminal to which the constant current circuit (PT) 1332 is connected, a gate terminal to which the output signal P0(n2) of the differential amplifier circuit PAn is connected and a drain terminal connected to the ground GND. An output 1336 for outputting a detect signal DET is outputted from the source terminal of the PMOS transistor 1334. Also the output 1336 is connected to the ground GND via a condenser 1338. The output (DET) 1336 of the detection circuit 1330 is connected to the comparator 1350.
The bias circuit 1340 includes a constant current circuit 1342 connected to the power supply VDD and a resistor 1344 connected between the output of the constant current circuit 1342 and the ground GND. The bias circuit 1340 outputs a bias signal BP from a connecting point 1346 at which the potential is determined by a current value of the constant current circuit 1342 and the resistance value of the resistor 1344. The bias signal BP is sent to the comparator 1350 where it serves as a reference potential of the comparator 1350. In the comparator 1350, the bias signal BP of the bias circuit 1340 is inputted to the input thereof on the plus (+) side, and the output signal DET of the DET circuit 1330 is inputted to the input thereof on the minus (−) side. The comparator 1350 compares these inputs and outputs an output signal (COMPOUT) to an output 1360.
A circuit, which comprises these differential amplifier circuit group (PA) 1310, differential amplifier circuit (PAn) 1320, detection (DET) circuit 1330, bias circuit 1340 and comparator 1350, has the relationship of a positive phase and a negative phase with respect to the input PIN(1) and the input PIN(2). When an amplitude-modulated signal is inputted thereto, output signals (P0n1) and (P0n2) amplified by the differential amplifier circuit group (PA) 1310 and the differential amplifier circuit (PAn) 1320 are outputted. The output signal (P0n2) delivered from the detection (DET) circuit 1330 is peak-detected on the amplitude Low side by the source follower circuit of the detection (DET) circuit 1330 and the condenser 1338 thereof, followed by demodulation. The demodulated signal is sent to the comparator 1350 as a detect signal (DET). The bias circuit 1340 generates a bias-signal (BP) corresponding to an intermediate potential between an "H" signal of the detect signal (DET) and an "L" signal thereof and supplies it to the comparator 1350. In the comparator 1350, a signal antiphase to the detect signal (DET) is brought to a signal between the power supply VDD and the ground GND.
A conventional configuration of a differential amplifier circuit shown in FIGS. 14 and 15 made use of a configuration wherein a plurality of differential amplifier circuits PA1 through PAn (where n: the number of connecting stages of differential amplifier circuits) were connected and a final-stage differential amplifier circuit was connected to an automatic gain control (AGC) circuit. Incidentally, both figures are combined together as shown in FIG. 16.
Described specifically, the differential amplifier circuits PA1, PA2, . . . , PAn respectively include PMOS transistors (Pn) each having a source terminal to which a power supply VDD is connected and a gate terminal to which a bias signal input (PB) 1410 is connected, and PMOS transistors (Pn1), (Pn2), (Pn3) and (Pn4) each having a source terminal to which a drain terminal of the PMOS transistor (Pn) is connected.
The drain terminals of the PMOS transistors (Pn1) and (Pn3) are interconnected with each other and connected to the ground GND via a resistor (PRn1). Also the drain terminals of the PMOS transistors (Pn1) and (Pn4) are interconnected with each other and connected to the ground GND via a resistor (PRn2). The gate terminals of the PMOS transistors (Pn3) and (Pn4) are interconnected with each other and connected with a connecting line 1420 for inputting an AGC signal.
The differential amplifier circuits PA1 through PAn configured of these PMOS transistors (P1(P2 through Pn)), (P11(P21 through Pn1)), (P12(P22 through Pn2)), (P13(P23 through Pn3)), (P14(P24 through Pn4)) and resistors (PR11(PR21 through PRn1)) and (PR12(PR22 through PRn2)) output outputs P011(P021 through P0n1) from nodes to which the drain terminals of the PMOS transistors (P11(P21 through Pn1)) and (P13(P23 through Pn3)) are connected, and output outputs P012(P022 through P0n2) from nodes to which the drain terminals of the PMOS transistors (P12(P22 through Pn2)) and (P14(P24 through Pn4)) are connected, respectively.
An input PIN(1) for inputting an input signal PIN(1) is connected to the gate of the PMOS transistor (P11) of the first-stage differential amplifier circuit PA1. An input PIN(2) for inputting an input signal PIN(2) is connected to the gate terminal of the PMOS transistor (P12).
Outputs P011 through P0(n-1)1 of the pre-stage differential amplifier circuits are connected to their corresponding gate terminals of the PMOS transistors (P21) through (Pn1) of the second-stage and later differential amplifier circuits PA2 through PAn. Outputs P012 through P0(n-1)2 of the pre-stage differential amplifier circuits are connected to their corresponding gate terminals of the PMOS transistors (P22) through (Pn2). An output signal P0n2 outputted from the output of the final-stage differential amplifier circuit PAn is inputted to the AGC circuit 1430.
The automatic gain control (AGC) circuit 1430 includes a PMOS transistor 1432 having a source terminal to which the power supply VDD is connected and a gate terminal to which a bias signal input (PB) is connected, and a PMOS transistor 1434 connected to a drain terminal of the PMOS transistor 1432. The output P0n2 of the differential amplifier circuit PAn is connected to its corresponding gate terminal of the PMS transistor 1434, and its drain terminal is connected to the ground.
A node at which the drain terminal of the PMOS transistor 1432 and the source terminal of the PMOS transistor 1434 are connected, constitutes an output 1440 that outputs a gain control signal (AGC). The output 1440 is also connected to the ground GND via a condenser 1450. The gain control signal (AGC) outputted to the output 1440 is supplied to the respective differential amplifier circuits PA1 through PAn.
In the circuit comprising these differential amplifier circuits PA1 through PAn and the AGC circuit 1430, a sine wave is inputted to the input PIN(1) and a sine wave antiphase to that at the input PIN(1) is inputted to the input PIN(2). A signal amplified by each of the differential amplifier circuits PA1 through PAn is transmitted to the AGC circuit 1430 as an output signal P0n2 of the final-stage differential amplifier circuit PAn. The output signal P0n2 is inputted to the gate terminal of the PMOS transistor 1434 of the AGC circuit 1430 and raised by a VT potential of the PMOS transistor. The Low-side peak of the amplified sine wave is detected by the condenser 1450 so that a gain control signal (AGC) is outputted. The gain control signal (ABC) is inputted to the gate terminals of the PMOS transistors (P13(P23 through Pn3)) and (P14(P24 through Pn4)) of the differential amplifier circuits PA1(PA2 through PAn), respectively.
When the potential of the gain control signal (AGC) is sufficiently higher than the potentials of the signals inputted to the gate terminals of the PMOS transistors (P11(P21 through Pn1)) and (P12(P22 through Pn2)) in the differential amplifier circuits PA1(PA2 through PAn), the currents equivalent to ½ of current values defined by the PMOS transistors (P1(P2 through Pn)) respectively flow into the PMOS transistors (P11(P21 through Pn1)) and (P12(P22 through Pn2)). The gains and potentials of the output signals P011(P021 through P0n1) and P012(P022 through P0n2) are respectively determined based on the current values of the PMOS transistors (P11(P21 through Pn1)) and (P12(P22 through Pn2)) and the resistance values of the resistors (PR11(PR21 through PRn1)) and (PR12(PR22 through PRn2).
When the gain control signal (AGC) becomes low, the current flows into each of the PMOS transistors (P13(P23 through Pn3)) and (P14(P24 through Pn4)). Therefore, the currents that flow through the PMOS transistors (P11(P21 through Pn1)) and (P12(P22 through Pn2)) become small and hence the gains of the output signals P011(P021 through P0n1) and P012(P022 through P0n2) become low. At this time, their output potentials remain unchanged.
Thus, when the amplitudes of the input signal PIN(1) and the input signal PIN(2) has changed, the potential of the gain control signal (AGC) varied correspondingly, thereby changing the gains of the differential amplifier circuits PA1 through PAn, whereby a stable output signal could be obtained.
Patent Document 1    Japanese Unexamined Patent Publication No. Hei 3(1991)-278110
In such a conventional circuit configuration shown in FIG. 13, however, when, for example, the total gain of the differential amplifier circuits of n stages is 80 [dB] and a very small ASK-modulated wave which is 30 [μV] at the maximum amplitude and 3 [μV] at the minimum amplitude with respect to each of the input PIN(1) and the input PIN(2), is inputted, a signal amplified to a maximum amplitude of 300 [mV] and a minimum amplitude of 30 [mV] is outputted to the output P0n2 of the differential amplifier circuit (PAn). In this case, the output signal P0n2 sent to the detection (DET) circuit 1330 is peak-detected by the source follower circuit and the condenser 1338 and raised by the VT potential of the PMOS transistor 1334, after which it is demodulated into a signal having an amplitude of 135 [mV], which in turn is outputted as a detect signal (DET).
In the bias circuit 1340, the current value of the constant current circuit 1342 and the resistor 1344 are set in such a manner that an intermediate potential of the amplitude 135 [mV] of the detect signal DET is reached, whereby a bias (BP) signal is outputted. When, at this time, a VT variation range of the PMOS transistor is ±100 [mV], for example, the VT potential of the PMOS transistor 1334 varies. Therefore, the DET signal of the detection (DET) circuit 1330 is affected by the VT variations and hence the output potential of the DET signal varies within the range of ±100 [mV].
On the other hand, since the output signal of the bias circuit 1340 is not affected by the VT variations of the PMOS transistor, the output signal results in a constant value even where the VT variations of the PMOS transistor take place.
A problem arises in that the detect signal (DET) varies within the range of ±100 [mV] when the VT variations take place in these ways, whereas since the amplitude of the detect signal (DET) is 135 [mV], the bias (BP) signal does not fall into the middle of the amplitude of the detect signal (DET) so that the output signal (CMPOUT) of the comparator 1350 can not provide or reach an expected value.
In the circuit configuration of the prior art shown in FIGS. 14 and 15, the output voltage of each of the differential amplifier circuits PA1 (PA2 through PAn) remains unchanged where the VT value of the PMOS transistor varies, whereas the potential of the gain control signal (AGC) changes by the VT variations as a result of the PMOS transistor 1434 provided within the AGC circuit 1430 being affected by the VT variations. Therefore, a problem arises in that when the VT variations of the PMOS transistor take place, the gain of each of the differential amplifier circuits PA1 (PA2 through PAn) inputted with the gain control signal AGC changes, so that a stable output signal is not obtained.


An integrated circuit (IC) package having a spark gap feature for electro static discharge protection is disclosed. The IC package includes an integrated circuit package substrate. An integrated circuit die is placed on the package substrate. Also on the package substrate are a first lead trace connected to the integrated circuit die and a second lead trace adjacent the first lead trace. The second lead trace is also connected to the integrated circuit die. The second lead trace has a spark gap feature allowing at least a portion of electro static spike on the second lead trace to jump to the first lead trace. The spark gap feature can be implemented in various ways including a protrusion from the second lead trace to the first lead trace, the protrusion having a sharp vertex proximal to the first lead trace.

The present invention relates to integrated circuit (IC) packaging technology, and more particularly, to an IC package having improved protection from electro-static discharges.
Integrated circuits (IC) having thousands or even millions of electronic components and circuit elements are fabricated on an IC die over a semiconductor substrate such as silicon. In this document, the phrase "IC die" includes the electronic components and circuit elements within the IC die. To electrically connect the IC die to other electronic components and to protect the IC die, the IC die is typically packaged within an IC package. A sample IC package 10 is illustrated in FIG. 1A in a perspective view. The IC package 10 often has plastic or ceramic body 14 and encases the IC die within the IC package 10. Further, the IC package 10 includes connection means 11 and 12 to connect, electrically and mechanically, the IC package 10 to other electronic components. For convenience, the connection means 11 and 12 are illustrated as pins 11 and 12; however, it is understood in the art that the connection means 11 and 12 can include other implementations such as ball grid array, connection pad, or others depending in the desired application. To avoid clutter, only two of the twelve illustrated pins are designated with the reference numerals 11 and 12; however, it is known in the industry that the IC package 10 can include any number of pins or connections means including, for example, hundreds of pins.
FIG. 1B illustrates a cut-away top view of the IC package 10 showing the IC die, designated with reference numeral 20, encased within the IC package 10. The IC die 20 is connected to the pins 11 and 12 via lead traces 31 and 32 fabricated over a bottom portion 16 of the IC package 10. Again, to avoid clutter, only two of the twelve illustrated lead traces are designated with the reference numerals 31 and 32; however, it is known in the industry that the IC package 10 can include any number of lead traces. In general, reference number 30 is used to refer to all the lead traces of the IC package 10.
The bottom portion 16 is often called a package substrate 16. Each of the lead traces 31 and 32 are electrically and mechanically connected to an input-output (IO) port (via connection pads on the IC die 20) of the IC die 20 via wire bonds 22 or other suitable methods. To avoid clutter, the electronic and circuit elements within the IC die 20 are not illustrated in the Figures. Also to avoid clutter, only two of the multiple wire bonds are desiganted with reference number 22; however, it is know in the industry that any or all traces can be connected to the IC die 20 via wire bonds 22.
The IC die 20 typically operates at 1.5 to 5 volts (v) depending on implementation. That is, each of the lead traces 30 of the IC package 10 typically carries electrical signals at 1.5 to 5 volts. However, the IC die 20 is often designed and manufactured to withstand temporary voltage spikes of up to two (2) kilovolts (kV) introduced to each of its IO ports. This is because surrounding components or environmental factors may introduce electrostatic discharges (ESD) to pins 11 and 12 (of the IC package 10) which are connected to the IO ports of the IC die 20 via the lead traces 30. Electrostatic discharge (ESD) is the release of static electricity when two objects come into contact. In fact, the IC package 10 may be exposed to even higher level of ESD voltage spikes from which the IC die 20 should protected. To protect the IC die 20 from failure, it would be desirable to reduce or minimize the severity of the ESD spike reaching the IO ports of the IC die 20.

Integrated circuit package with improved electro-static discharge protection

Avago Technologies General IP (Singapore) Pte. Ltd.


A method applying a line lock feature, i.e. a wheel hold mode, with a brake system for a vehicle comprises initiating a wheel hold mode for the brake system and applying pressure to the wheels of a non-driven axle. The wheel brake pressure is maintained at the wheels of the non-driven axle independent of application of the brake pedal. The wheel brake pressure at the wheels of the non-driven axle is released when a release trigger event occurs. The release trigger event may be an automatic trigger event or a driver actuated trigger event.

The present invention relates to an electronically controlled hydraulic brake system. Maintaining optimal performance for vehicle brake systems is important. For example, in some situations it is desirable for a driver of the vehicle to provide conditioning of the tire, such as increasing a surface temperature of the tire prior to driving. Also, in a competition drag racing environment, a driver is required to position the car at the starting line and maintain that position until the signal to accelerate is given. Failure to do so results in disqualification. In such situations of tire conditioning, it is desirable to hold some tires of the vehicle from moving to maintain the vehicle position, while allowing other tires to freely rotate and, thus, increase the surface temperature of the tire.
Vehicles equipped with manual transmissions require the driver to operate the throttle, brake, and clutch pedal simultaneously in order to achieve optimum tire-conditioning. The driver's left foot must manage clutch engagement while their right foot is forced to engage both the throttle to manage engine speed and the brake pedal to prevent the vehicle from moving. To assist a driver in such conditions mechanical and electro-mechanical valves and solenoids are installed on the vehicles to override the braking system and to assist the driver to maintain brake pressure at the appropriate wheels.
The background description provided herein is for the purpose of generally presenting the context of the disclosure. Work of the presently named inventors, to the extent it is described in this background section, as well as aspects of the description that may not otherwise qualify as prior art at the time of filing, are neither expressly nor impliedly admitted as prior art against the present disclosure.

Electronic brake system line lock feature


Embodiments relate to a method and computer program product for prefetching data on a chip having at least one scout core and a parent core. The method includes saving a prefetch code start address by the parent core. The prefetch code start address indicates where a prefetch code is stored. The prefetch code is specifically configured for monitoring the parent core based on a specific application being executed by the parent core. The method includes sending a broadcast interrupt signal by the parent core to the at least one scout core. The broadcast interrupt signal being sent based on the prefetch code start address being saved. The method includes monitoring the parent core by the prefetch code executed by at least one scout core. The scout core executes the prefetch code based on receiving the broadcast interrupt signal.

The present invention relates generally to multi-core chips having a parent core and a scout core, and more specifically, to specific prefetch algorithms for a parent core in a multi-core chip.
Multiple cores may be located on a single chip. In one approach, a second core on the same chip as the parent core may be provided as a scout core. In one approach to leverage or utilize the existing scout core, the scout core is used to prefetch data from a shared cache into the parent core's private cache. This approach may be especially useful in the event the parent core encounters a cache miss. A cache miss occurs when a particular line of data causes a search of a directory of the parent core, and the requested line of cache is not present. One typical approach to obtain the missing cache line is to initiate a fetch operation to a higher level of cache. The scout core provides a mechanism that is used to prefetch data needed by the parent core.
It should be noted that various application behave differently, and as a result one prefetching algorithm or approach may not always improve latency of accessing cache content. Specifically, for example, if the parent core is executing several different applications, then the prefetching algorithm used to monitor the different applications may provide various latencies for accessing cache content depending on the specific application that is being executed. For example, an application designed to search through a sparsely populated database may behave differently (e.g., the prefetching algorithm may provide a higher or lower latency of accessing cache content) when compared to an application designed to perform image color correction.

This patent is subject to a terminal disclaimer.

Specific prefetch algorithm for a chip having a parent core and a scout core

International Business Machines Corporation