A process of providing a disposable core for use in die casting processes. A salt material is molten and cast into a core of a desired configuration under exacting conditions. The fluidity of the molten salt is controlled enabling casting the salt material into a core by die casting methods. The die casting method provides a core with a high surface finish and strength. The core is evenly cooled subsequent to it being cast and is maintained at an elevated temperature to maintain its surface finish and structural integrity. The cast core is inserted into the dies of a metal die casting machine to facilitate casting a metal product having internal forms not otherwise attainable. The core is removed from the metal product by simply dissolving and flushing the core out of the casting. The salt material may be reclaimed by a de-salination process for further use.

Internal passages are not commonly provided in products produced by a die casting process. Casting an internal passage requires the use of a core produced from a material that can be removed from the cast product without damaging the product. Sand and foam are common materials used for making cores. Sand is removed by demolishing it and foam is removed by burning. However, neither is considered suitable for die casting. The die casting process subjects the cores to high pressures during the fill process that are typically beyond that which can be withstood by sand and foam cores. Sand cores are also difficult to demolish and the destructions of foam produces toxic contaminants that are environmentally undesirable.
Accordingly, die casting is most commonly achieved without the production of internal passages or cavities. Products produced by die casting that have internal passages require follow-up processes. More often such products are produced by methods other than die casting.
An alternative material was developed some years ago as having properties desirable for producing cores for casting. This material has a salt base (soda ash or perhaps more accurately an ionic compound) that is dissolvable in water but in a dry state can be provided in powder-like form. The dry powder form can be melted into a liquid or molten state, poured into a mold and cooled to a solid state. That is, the core is produced by "casting" the desired core shape and the core produced thereby is in turn used for the casting of a metal product requiring internal passages. Such a core can be inserted into a casting mold, and following the casting process, it can be simply washed out of the product to leave the internal passages or cavities. Such a material is available from Park Chemical Company of Detroit, Mich. and the material is known as Aluminum Casting Salt. Hereafter this material is often referred to simply as casting salt but includes all materials having similar properties, e.g., an ionic compound that can be molded into a solid state and washed away with water.
The problem with this material is that the casting salt cores previously produced with any degree of success have been produced by the known method of gravity flow casting. These have been found to be too fragile for use in die casting. Whereas the die casting process as compared to gravity flow casting is known to produce a stronger casting, that has not been the case for producing cores of casting salt. Prior attempts to produce cores from casting salt have resulted in disaster. The liquid molten casting salt has such a low viscosity that the material cannot be contained in the die casting molds. When the material is converted to its molten state and injected into a casting mold under the required high pressures, the material simply blows or spits out the vent holes (or out the seams of the mold in a natural venting process). Thus, the use of casting salt for making cores for die casting has been extremely limited. It is believed that there is a single application where such cores are die cast but under extremely unsatisfactory conditions. It is understood that in this application the molten casting salt spits or blows out of the seams and the process is both wasteful and messy and requires special preparation and cautious handling.


A vacuum tube for amplifying an r.f. signal includes an assembly containing a cathode and grid for current modulating an electron beam derived from the cathode. One of the electrodes of the assembly includes a slow wave structure approximately resonant to the frequency of the signal. A cavity resonant to the frequency of the signal, positioned between the grid and a collector for the beam, is coupled to the beam. In one embodiment, the slow-wave structure is mounted in a support for the grid, while in a second embodiment, the grid is configured as plural, parallel meander lines forming the slow-wave structure. In the latter embodiment, the beam is preferably annular and the meander line geometry, in certain modifications, is adjusted so that there is a relatively small electric-field variation with radius over the portion of the grid through which the annular beam passes. In a further embodiment, the grid is configured as two interlaced spirals, driven by complementary replicas of the r.f. signal so the beam is formed at twice the frequency of the r.f. signal. Focusing electrodes configured as a perforated sheet, contacting the cathode, or as electrodes just downstream of the control grid, or both, collimate ,the beam, whether hollow or not.

A recently developed vacuum tube for handling r.f. signals includes a cathode for emitting a linear electron beam, a grid positioned at right angles to the direction of flow of the beam in close proximity to the cathode (no farther than the distance an emitted electron can travel in a quarter of an r.f. cycle at the highest frequency being handled by the tube) for current modulating the beam, and a cavity resonant to the frequency of the signal positioned between the grid and a collector electrode for the beam. The grid is coupled by a structure resonant to the frequency being handled by the tube to an r.f. input signal to be amplified by the tube. To prevent electron emission from the grid, it is formed of a non-emissive material, such as pyrolytic graphite or molybdenum coated with zirconium.
As applied to the electron beam flowing beyond the grid, the terms "current-modulated," "space-charge-modulated," "density-modulated" and "intensity-modulated" are synonymous, and refer to concentrations (or "bunches") alternating with depletions of particle density (or space-charge density) along the beam. Speeding and slowing of particle velocity is indicated by the term "velocity modulation."
Very high efficiency is achieved with such a tube by biasing the grid so that current flowing from the cathode toward the grid occurs for no more than one half cycle of the r.f. signal handled by the tube. Typically, the bias voltage between the grid and cathode is very small or zero.
In one prior art configuration, the resonant input circuit supplies electric fields having opposing phases between the cathode and grid and between the grid and an accelerating anode positioned between the grid and the output cavity. In another prior art modification, a second resonant cavity positioned between the output cavity and the accelerating anode is adjusted so the resonance frequency thereof is above the frequency being handled by the tube, to increase the average efficiency of the tube. These prior art structures are disclosed in the commonly assigned U.S. Pat. Nos. 4,480,210, 4,527,091 and 4,611,149. Devices incorporating the teachings of at least some of these patents are commercially available from the assignee of the present invention under the registered trademark KLYSTRODE.
While the prior art tubes have performed admirably, they are rather large. One of the factors contributing to the size of the prior art tubes of the general type disclosed in said patents is an input resonant cavity coaxial with the cathode and the electron beam emitted from it. The resonant coaxial cavity couples an input signal to an assembly including the cathode and grid. This resonant cavity has a length in the direction of the beam axis that is nominally either a half-wavelength at the frequency handled by the tube or a full wavelength at this frequency. In practice, it is most usually the latter.
The r.f. input signal to be amplified is transformer-coupled to the input resonant cavity which couples the field established in the cavity to the grid-cathode and grid-anode regions, in response to the input signal. In this document, the phrase "transformer coupled to the cavity" signifies that the r.f. power coming into or going out of a coaxial cable is coupled by r.f. magnetic fields to the cavity via loop coupling or by r.f. electric fields via probe coupling. While the size constraint associated with the input resonant cavity is not an impediment to many commercial uses of the KLYSTRODE brand tube, it is a substantial detracting factor for many military and space applications.
It is, therefore, an object of the present invention to provide a new and improved vacuum tube for handling r.f. signals wherein the vacuum tube includes a grid for current-modulating an electron beam, in combination with a resonant input structure and a resonant output cavity, wherein the tube has a smaller volume and length than prior art tubes of this type.
It is important for the reduced-size tube of the aforementioned type to have a relatively high input impedance across a signal source, i.e. for the grid-cathode impedance to be relatively high, as in the prior art.
It is, therefore, a further object of the invention to provide a new and improved r.f. amplifying tube including a relatively small resonant structure for coupling an input signal to a grid for current-modulating a beam without excessively loading the input signal source.

Vacuum tube including grid-cathode assembly with resonant slow-wave structure


A DTD-compliant XML document editor is disclosed. The XML document editor generates hints for required elements and required element slots automatically in the process of document creation and edition so as to guide the user to produce a valid document, while syntactic violations are avoided at the same time. The editor also suggests optional elements that may be added into the document to be edited. The user requires no knowledge about XML and DTD to edit DTD-compliant XML documents. The editing process is user-interface-neutral, being able to support user interfaces based on tree views, presentation views and forms. By combining the DTD-compliant editing process with the simplicity of forms, a simple XML document editor with forms as its user interface is developed.

Extensible markup language (XML) is a new generic representation for data and documents. It has emerged as a standard for information exchange over the network. Moreover, many application domains have developed DTD (Document Type Definition) or schema-based XML vocabularies for their data and documents to be coded in XML and processed by standard software packages. XML thus has great impact on all these application domains.
Fundamental to XML applications is the creation and editing of XML documents. This is evidenced by the large number of commercial and public-domain XML editors released so far. An XML document editor (hereinafter referred to as "XML editor") allows its user to produce either well-formed and/or syntax valid XML documents. Such an XML editor typically maintains a well-formed working document and provides validity checking upon the user's request. The user has all the freedom to edit the document. To ensure the document to be valid, the user must check for validity and correct the syntactic violations in the document reported by the validity checker. This requires the user's knowledge about the DTD or schema of the document. Another approach to relieve the user from syntactic concerns is to provide extensive customization specific to XML vocabularies. However, this requires the intervention of a technical expert.
XML editors in use today typically support the so-called context-sensitive editing. While a user is editing a document, the system provides guidance or hints regarding what the user can do next so as to produce a valid document eventually. This policy is helpful. However, the guidance or hints these XML editors provide are typically too loose to guarantee the validity of the resulting document. That becomes a major reason why these XML editors need a validity checker.
Current XML editors appear to differ more in their user interfaces than their underlying editing methodologies and, thus, may be classified into 4 types according to their user interface. They are: user interfaces based on text views, tree views, presentation views and forms. Some systems support multiple types of user interfaces. A text view XML editor allows the user to edit tags as in a text editor. A tree view editor displays the hierarchical structure of an XML document as a tree and provides operations for inserting and deleting tree nodes. A presentation view applies some kinds of transformations, including XSLT, CSS and proprietary ones, to an XML document in order to present a "WYSIWYG" user interface to its user. A presentation view specific to a vocabulary typically requires customization and support by technical experts.
A form-based XML editor is essentially a form generator based on a data model specified by an XML schema (or DTD). Current form-based XML editors are not robust enough to handle general XML data, not to mention their ability to avoid syntactic violations upon data entry.
These XML editors do not enforce syntactic constraints strictly during document construction and provide validity checking as warning/hints for correcting syntactic violations among other typically inaccurate context-sensitive guidance and hints. These XML editors allow users to create and edit both well-formed and valid XML documents in a single editing mode.


A wheel adapted for use on a roller skate or the like includes a wheel member made of elastomer having shock-absorbing property, and a pair of disk members embedded into a central hole of the wheel member, and combinable with each other for firmly clamping two central rim portions on two opposite sides of the wheel member, so that the wheel member secured with the two disk members will be stably rotatably mounted on an axle provided on a mounting frame of a roller skate, a skate board or any other wheeled sporting equipments for enhancing a better stability, heat resistance, shock-absorbing property and ground-treading property of the wheel.

A conventional roller or wheel W secured with a hub H is rotatably mounted on an axle A of a roller skate as shown in FIGS. 1 and 2, in which the roller or wheel W may be made of polyurethane (PU) or polyvinyl chloride (PVC) material. However, such a conventional roller has the following drawbacks:
1. Even polyurethane, for instance, may have good elasticity and resistance to abrasion, but is generally considered to be less resistant to high or low temperature. During a high-speed running of a skate roller, a frictional heat may be produced to deteriorate or damage such PU rollers.
2. In order to reinforce the strength and stability of the PU roller, a rigid central hub H with extension E is integrally secured with the ring-shaped wheel W. Since the wheel W may be made of elastomer foam material, a poor foam product such as unhomogeneous porosity or shrunk molding product will cause instability of the wheel when running the roller skate or even cause sport accident or injury to a player.
3. The wheel W is integrated with hub H so that once the wheel W is damaged the whole wheel unit should be deleted to waste resource.
4. The hub H is merely a circular-shaped disk, having no clamping portions for efficiently clamping the wheel W, so that the wheel unit will be vibrated when gliding a roller skate to thereby influence a stable playing of the skate since the hub H is formed as a "flat" disk without having any clamping flanges provided thereon for clamping the wheel W.
5. No tread grooves exist in the wheel surface to easily cause slipping on a ground surface, also lacking of waterrepelling capability.
The present inventor has found the drawbacks of the rollers or wheels of a conventional roller skate and invented the present wheel having better properties.

c/o Hung Hsing Patent Service Center, P.O. Box 55-1670


Apparatus for forming plastic preforms into plastic containers, with at least one blowing station arranged on a moving carrier, wherein the blowing station has a blow mould carrier for holding a blow mould and the blow mould carrier has a first blow mould carrier part and a second blow mould carrier part, wherein the second blow mould carrier part can be moved away from the first blow mould carrier part in order to open the blow mould for removal of a plastic container and/or for insertion of a preform, and wherein the second blow mould carrier part can be moved towards the first blow mould carrier part in order to transfer the blow mould into a closed state for forming the preform, wherein that the second blow mould carrier part has a lower density and/or strength than the first blow mould carrier part.

The present invention relates to an apparatus for forming plastic preforms into plastic containers and a blow mould carrier. Such devices has been known for some time in the prior art. Normally, plastic preforms are formed into plastic containers by pressurization with compressed air in a plurality of blowing stations. As part of the expansion process, a number of successive movement processes take place. Normally the blow mould carrier halves are pivoted apart for insertion of a plastic preform and then pivoted together in order to perform the expansion process. After the end of the expansion process, the blow mould carrier parts are pivoted apart again to be able to remove the finished blown plastic container. When the blow mould carrier parts are in the closed state, in particular during the expansion process, in addition the blow mould carrier parts are locked together in order to be able to withstand the not inconsiderable pressures. Normally a pressure pad mechanism is arranged on one of the two blow mould carrier parts in order, for the duration of the prevailing interior pressure in the blowing process, to press the two blow moulds arranged on the blow mould carrier parts together with greater force than that provoked by the blowing pressure. Thus visible joint seams on the containers produced are kept as small as possible.
Furthermore sterile applications are known from the prior art in which the individual blowing stations are conducted inside a clean room. For example WO 2010/020529 A2 describes an apparatus for forming plastic preforms into plastic containers, which has a transport device comprising several blowing stations and a clean room.
By means of the clean room, aseptic conditions can be guaranteed in the production of the plastic containers and also during their filling with drinks.
The subject of WO 2010/020529-A2 is hereby made fully the subject of the present disclosure by reference.
As stated above, an important component of a blowing station is the locking mechanism which connects or locks the two mould carrier halves together during the blowing process. After the end of the blowing process, the lock opens. After opening the mould, the bottle can be removed and a new preform inserted. In the prior art, these locking mechanisms normally have a locking shaft mounted in one of the two mould carriers. Several bearing points distributed over the height of the mould carrier guarantee an even force distribution in the components. This is important since the lock must withstand a holding force of several 10,000 N. The locking process consists for example of a rotary or linear movement of the lock. This movement is normally generated by a stationary curve attached to the base structure of the machine, and transferred to the locking shaft via various deflections.
The opening and closing movement of the mould carrier is independent of this and controlled separately. Control takes place via a stationary control curve. For complete control therefore two curves are required, one to control the two mould carrier halves and one to control the lock.
It may also be necessary to provide a tempering of the blow moulds and/or the blow mould carrier shells in order to cool or heat the plastic surface quickly on expansion.
It has been found that the two blow mould carrier parts are subject to different requirements.
For example it is possible that asymmetric opening of the blow mould carrier parts offers kinematic advantages, in order to be able to insert the preforms in the blow mould gently (for the preform and control curves of the blow mould and the transfer stars for insertion and removal). If for this, one blow mould carrier part must be pivoted further than the other, the forces on the one are higher.
Furthermore a blow mould carrier part must resist the above-mentioned pressure pad mechanism, which again makes this heavier.
Furthermore it is possible to pivot the above-mentioned locking mechanism with the blow mould carrier parts. This locking mechanism normally has moving parts (e.g. a rotatable hook) which must be held by at least one blow mould carrier part, which again makes this heavier, while the other blow mould carrier part need merely provide a surface to create a form fit for the moving parts of the other (e.g. a groove to receive the hook).
It is therefore desirable to adapt the blow mould carrier parts of a blowing station to the different requirements, in particular with regard to the kinematics and weight.

Device for forming plastic preforms into plastic containers and blow mould carrier


A rule-based search engine is used in conjunction with an automated network-based workflow system (which in turn is interfaced with an organizational database) to efficiently determine service routing requests from users/clients. The search engine employs search techniques adapted for use with multi-dimensional tree structures that define the matrix organizational model. Workflow services are preferably represented by roles that can be used to represent workflow actors in the workflow routing rules. These roles are preferably evaluated at run-time to best match recipients depending on the organization context from which the routing request is made.

Workflow is generally defined as a series of tasks within an organization to produce a final outcome. In particular, workflow, as defined by the Workflow Management Coalition®, is the automation of a business process, in whole or in part, during which documents, information, or tasks are passed from one participant to another for action, according to a set of procedural rules.
A workflow management system, as defined by the Workflow Management Coalition® is a system that defines, creates, and manages the execution of workflows through the use of software, running on one or more workflow engines, which is able to interpret the process definition, interact with workflow participants and, where required invoke the use of information technology (IT) tools and applications.
Therefore, a workflow model is a description of how different tasks are sequenced in a business process and who performs these tasks. Since tasks involve information that is exchanged between people or systems, it is necessary to extend this definition to include the data that is being exchanged. From this explanation we can define three separate components of most workflow models:                Organization (e.g., who does what and who knows what in the organization?)        Process (e.g., what are the tasks and how are they sequenced?)        Application (e.g., what data is to be exchanged?)Any number of organizational data sets, business processes or applications can be used in a single workflow process, and further, these components may be re-used in any number of workflow processes.
Sophisticated workgroup computing applications facilitate defining different workflows for different types of jobs. For example, in a publishing setting a document may be automatically routed from writer to editor to proofreader to production. At each stage in the workflow, one individual or group is responsible for a specific task. Once the task is complete, the workflow system ensures that the individuals responsible for the next task are notified and receive the data they need to execute their stage of the process. Some of the common workflow systems include FLOWMARK® and POWERFLOW®, available from International Business Machines and Percussion Software, respectively.
IBM FLOWMARK® is a workflow management system that helps organizations to define, document, test, control, execute, and improve their business processes. Furthermore, it helps define control flow and data flow via a graphical interface, thereby avoiding writing of code. Flowmark is usually used in conjunction with Lotus NOTES®.
Percussion POWERFLOW® is a tool for adding workflow logic to any Lotus Domino® application via a graphical interface. POWERFLOW also performs statistical analysis on business processes. POWERFLOW requires a Domino server (4.5 or above).
The business world has long recognized the benefits of employing workflow systems to automate internal business processes. By defining an internal process and automating the flow of information, these tools have improved the way business is done. Paper-intensive, manual tasks, such as expense reporting or claims processing, have been replaced with a more efficient workflow model. In addition, with the growing emphasis on electronic business enterprise strategies, workflow technology has also become the integration platform to support business-to-business (i.e., external) relationships, as well as internal processes.
As companies have recognized the benefits of workflow enabled electronic document exchange, several workflow automation tools have emerged in the market. These tools are widely employed, despite their inherit weaknesses. One limitation of most workflow tools is that even though they allow companies to implement workflow processes, they fail to provide a good platform-independent solution. Furthermore, corporate re-organizations and employee turnover are frequent events in today's business world. In order to compete in the expanding global market, companies must be able to adapt to changing organization structures. Another limitation of traditional workflow tools is that they suffer from lengthy implementations—both a direct cost and an opportunity cost in today's fast-paced electronic business economy. Secondly, and perhaps most importantly, most tools are often too complex for the typical business user to administer. That is, specialized IT personnel are often required to implement, maintain and troubleshoot workflow enabled technology—either because relationships are hardcoded in the workflow tool, or because the technology is embedded in a more complex application. When IT personnel administer the applications, the business-users' input in the business event becomes limited. It is easy to see how the benefit of business-user's input into improving the process and responding more quickly to changes in the organization is lost when the business-user has no direct control or oversight of the business event.
Conventional automated workflow tools generally provide descriptions of an organizational structure that are cumbersome to modify to account for organizational changes and have limited organizational modeling ability. For example, prior art systems allow fewer dimensions of organizational definition, and significantly, they fail to provide the ability to define easily referenced functional roles with the organization. Another shortcoming is seen in cases where the added dimensions can be defined, but cannot be identified "in context". As an example, a functional role of "Direct Supervisor" should not yield the same result for all personnel within an organization. The correct, contextual answer varies depending upon the person for whom that relative role is being resolved. The prior art also lacks the ability to provide "functional links" between organization units. These links allow a portion of an organization tree/sub-tree to be made available as a provider of some service or function to another sub-tree that is part of a completely different organization (e.g., a different company engaged to provide out-sourced services).
Therefore, there is a need for a system that provides the ability to provide functional links between organizational units, and allows organizational changes to be made easily and to be correctly modeled by the automated workflow system.
Furthermore, there is a need for a tool that allows a business(es) to build and maintain web enabled workflow solutions to support electronic business applications, as well as business-to-business relationships. Furthermore, it is beneficial to have an enterprise workflow tool that provides a clear separation between the application, the processes and the organization. The present invention provides for an easily implemented, maintained and administered workflow model that maximizes user group control, and can dynamically adapt to changes in the enterprise.

Workflow system matrix organization search engine

International Business Machines Corporation


A unit for recovering thermal energy from exhaust gas of an internal combustion engine including an inlet, an outlet, a heat exchanger arranged downstream of the inlet and upstream of the outlet, and a valve capable of being switched between a recovery position. The exhaust gas flows from the inlet through the heat exchanger to the outlet, and a bypass position, in which the exhaust gas flows from the inlet through a bypass to the outlet. The valve includes separate first and second valve bodies and associated separate first and second valve seats arranged upstream and downstream of the heat exchanger, respectively. The first and second valve bodies are adapted to be switched only simultaneously from the recovery position to the bypass position and vice versa.

The automotive industry is more and more oriented in the design of engines, on-board systems and the vehicle itself, towards the reduction of fuel consumption, emission of pollutants, recyclability of materials and the recovery and conversion of thermal energy that would otherwise be dissipated.
As regards the recovery and conversion of thermal energy, the exhaust gas exiting from the internal combustion engine carries, after being treated by a catalyst or by an even more complex treatment unit, an amount of thermal energy that can be recovered for various useful purposes. For example, it has been suggested to recover the thermal energy from the exhaust gas for the purpose of heating the cabin (by more rapidly heating up the cooling water of the engine) of the vehicle, for the purpose more rapidly heating up the lubricating oil to the optimum temperature (lower viscosity) after a cold start of the engine, or for converting the thermal energy into electrical energy (with the aid of a suitable converter).
Such unit for recovering thermal energy is known, for example, from EP-A-2 381 083. The unit disclosed in this reference comprises an inlet for the exhaust gas, an outlet for the exhaust gas, and a heat exchanger arranged between the inlet and the outlet for the exhaust gas. In addition, the unit comprises a single rotatable valve flap that can be rotated to a recovery position in which the exhaust gas is directed along a U-shaped path through the heat exchanger while exhaust gas is prevented from flowing through the bypass. Alternatively, the valve flap can be rotated to a bypass position in which the exhaust gas is directed along a straight bypass of the heat exchanger, although the U-shaped flow path through the heat exchanger is not physically closed. As a further alternative, the valve flap can be rotated to any angular position between the recovery position and the bypass position for partial heat recovery.
However, the unit disclosed in EP-A-2 381 083 suffers from a number of disadvantages. First of all, the single valve flap may only substantially completely close the bypass even when the valve flap is in the recovery position, since the valve flap must be capable of being rotated to a desired position at any operating condition (e.g. after a cold start of the engine as well as when the engine is at the desired operating temperature, at which the inner diameter of the bypass may be larger than after a cold start due to the change in temperature). Accordingly, at least some clearance must be provided between the valve flap and the inner wall of the bypass so that the valve flap cannot completely close the bypass even when the valve flap is in the recovery position (bypass closed). As a consequence of this clearance, there is some backflow of exhaust gas that—after having flown through the heat exchanger—has been cooled to a temperature which is considerably lower than that of the exhaust gas coming from the engine or the catalyst and entering the unit. This leads to a loss of heat of the exhaust gas flowing through the heat exchanger which is unwanted and reduces the efficiency of the heat recovery process. Also, as the valve is in the recovery position the valve flap is rotated to a position in which it is arranged perpendicular to the direction flow of exhaust gas through the bypass. This may generate an unwanted backpressure which must be overcome thus reducing the efficiency of the engine.
As the valve is in the bypass position the U-shaped flow path through the heat exchanger is not completely closed, although any further heating up of the cooling water of the engine (which is the working fluid flowing through the heat exchanger) may be unwanted since the cooling water already has the desired temperature. Needless to say that in any position between the recovery position and the bypass position there may be backflow of exhaust gas which has been cooled—after having flown through the heat exchanger—to a temperature which is considerably lower than the temperature of the exhaust gas entering the system, thus substantially reducing the efficiency of the heat recovery process.

Unit for recovering thermal energy from exhaust gas of an internal combustion engine


A process for the continuous production of a thermoplastic molding composition containing a mixture of a TPU and a thermoplastic (co)polymer is disclosed. The process comprises. (i) introducing into an extruder via a first feed point component I) with at least 3% by weight, with respect to component I), of at least one member selected from the group consisting of components A), B) and C), and PA1 (ii) metering in the balance of said A), B) and C) into said extruder via one or more subsequent feed points, PA1 (iii) extruding components I) and II) under conditions sufficient to cause A), B) and C) reaction to form TPU, and PA1 (iv) upon the completion of said reaction, removing the resulting mixture of TPU and (co)polymer from the extruder,. wherein component I) comprises 1 to 60% by weight, with respect to the total mixture, of a thermoplastic (co)polymer, and wherein component II) comprises 99 to 40% by weight, with respect to the total mixture, of the polyurethane-forming components including A) an organic diisocyanate, B) a linear polyol having terminal hydroxyl groups and a number average molecular weight of 500 to 5000, and C) a chain extender having a molecular weight of 60 to 500, and wherein the molar ratio of NCO groups in A) to the groups which react with NCO in B) and C) is in the range of 0.9 to 1.2.

Thermoplastic polyurethane elastomers (TPUs) have long been known. They are of commercial importance due to their combination of high-grade mechanical properties with the known advantages of cost-effective thermoplastic processability. A wide range of variation in their mechanical properties can be achieved by the use of different chemical synthesis components. A review of TPUs, their properties and applications is given in Kunststoffe [Plastics] 68 (1978), pages 819 to 825, and in Kautschuk, Gummi, Kunststoffe [Natural and Vulcanized Rubber and Plastics] 35 (1982), pages 568 to 584.
TPUs are synthesized from linear polyols, mainly polyester diols or polyether diols, organic diisocyanates and short chain diols (chain extenders). Catalysts may be added to the reaction to speed up the reaction of the components.
The relative amounts of the components may be varied over a wide range of molar ratios in order to adjust the properties. Molar ratios of polyols to chain extenders from 1:1 to 1:12 have been reported. These result in products with hardness values ranging from 80 Shore A to 75 Shore D.
TPUs can either be produced in stages (prepolymer method) or by the simultaneous reaction of all the components in one step (one shot). In the former, a prepolymer formed from the polyol and diisocyanate is first formed and then reacted with the chain extender.
TPUs may be produced continuously or batch-wise. The best-known industrial production processes are the so-called belt process and the extruder process. By mixing them with other polymers, the favorable properties of TPUs can be modified. In particular, the admixture of graft rubbers improves the low temperature properties of TPUS. A TPU/ABS mixture is described in U.S. Pat. No. 4,317,890. The molding compositions obtained, however, exhibit unsatisfactory impact strength at low temperatures and are not free from segregation phenomena, the so-called "mother of pearl" effect. The use of TPUs of low density (EP-A 0,152,049) or the addition of auxiliary processing agents (polyacrylates in U.S. Pat. No. 4,179,479, styrene-MSA copolymers in EP-A 0,443,432) does not impart a significant improvement to the compositions, which are characterized, in particular, by their modest low-temperature impact-strength.
Another route is described in DE-OS 2,854,409 (U.S. Pat. No. 4,342,847). The TPU-polymer mixtures are produced in an extruder reaction process by adding all the TPU raw materials to the pre-produced polymer, which has previously been melted in the extruder. The mixtures produced in this manner exhibit improved behavior at low temperatures. Their disadvantages, however, are the high melting temperatures required for the copolymer, which result in thermal degradation of the product, or the friction effects which occur at lower melting temperatures, which likewise result in degradation of the product. Moreover, during a continuous process at the low temperatures which are customary for a mixed phase of a TPU reaction (100.degree. to 200.degree. C.), extreme variations occur in the viscosity of the copolymer melt, which cause pressure fluctuations when metering in the TPU monomer. In a continuous process, this does not permit the TPU components to be metered in uniformly over an extended period, which can lead to inhomogeneities in the product or even to a stoppage of the continuous process.
Therefore, no success has hitherto been achieved in producing TPU-copolymer mixtures having good mechanical and elastic properties, particularly at low temperatures, in a continuous process over an extended period without interruptions.

Process for preparing TPU molding compositions


A deposition module within a web press for inhibiting media deformation may include a fluid deposition device located on a side of a media, and a processing device to control the fluid deposition device to selectively apply a wetting agent to the media. The deposition of the wetting agent may be based on the location of content printed on the media as defined by print data.

An inkjet web press is a high-speed, digital, industrial printing device that prints on a continuous media web at speeds of hundreds of feet per minute. A roll of media such as a paper on an unwinding device supplies the press with a web which is conveyed through the press along a media path. Stationary printheads along the media path may eject droplets of printing fluid onto the web to form images. The web may be conveyed through a drying area and out of the press through rollers to be rewound on a rewinding device.
Throughout the drawings, identical reference numbers designate similar, but not necessarily identical, elements. The figures are not necessarily to scale, and the size of some parts may be exaggerated to more clearly illustrate the example shown. Moreover, the drawings provide examples and/or implementations consistent with the description; however, the description is not limited to the examples and/or implementations provided in the drawings.

HEWLETT-PACKARD DEVELOPMENT COMPANY, L.P.


An apparatus and method for estimating a decision boundary in symbol units in a mobile communication system. The method for estimating a decision boundary of reception symbols in a receiver of a mobile communication system includes the step of calculating a preliminary decision boundary for every symbol entered a decision boundary estimator; and reflecting the preliminary decision boundary calculated for every symbol in a previous symbol's decision boundary having been calculated in consideration of a channel environment until reaching the previous symbol, and calculating a new decision boundary for the entered symbol.

1. Field of the Invention
The present invention relates to a mobile communication system, and more particularly to an apparatus and method for estimating a decision boundary of a specific signal to which a higher order modulation scheme is applied, in a receiver for use in the mobile communication system.
2. Description of the Related Art
Mobile communication systems have evolved from typical voice-centered mobile services to high-speed packet transmission services to make it possible to transmit not only voice data but also high-speed packet data. The 3rd Generation Partnership Project (3GPP2) and the 3GPP2 for use in the third-generation mobile communication standard have recently proposed a variety of high-speed packet communication systems. For example, a High Speed Downlink Packet Access (HSDPA) standardization is in progress in the 3GPP, and a Evolution Data and Voice (1xEV DV) standardization is in progress in the 3GPP2.
Typically, a digital signal transmission system is superior to an analog signal transmission system because it is less sensitive to noise, has lower distortion, and has higher transmission efficiency as compared to the analog signal transmission system. However, the digital signal transmission system has a disadvantage in that it requires a wider bandwidth and higher system complexity as compared to the analog signal transmission system. As a variety of circuit technologies have been developed, the digital signal transmission system has been widely used in a variety of communication schemes because it generates less errors and has higher reliability compared to the analog signal transmission system.
A representative example of such a digital modulation/demodulation technique capable of transmitting two-bit input signals as a single signal is a Quadrature Phase Shift Keying (QPSK) scheme. The QPSK scheme transfers an odd bit of input bits to an In-phase (I)—channel, and transfers an even bit of the input bit to a Quadrature (Q)—channel, such that a specific value '0' is changed to '−1', and other value '1' remains. In this case, the input bits each are matched with a predetermined phase and amplitude to create modulated signals. These modulated signals each are typically called a symbol.
However, in order to implement high-speed data transmission for the aforementioned wireless communication system, there has recently been proposed a higher-order modulation scheme such as an M-Quadrature Amplitude Modulation (M-QAM) scheme capable of improving a data transfer rate.
The M-QAM scheme matches input bits with a plurality of phases and amplitudes to create modulation signals, and is classified into a variety of modulation schemes (for example, 4 Quadrature Amplitude Modulation (QAM), 16QAM, 32QAM, 64QAM, 128QAM, and 256QAM, etc.) according to a modulation rate.
The 16QAM scheme included in the M-QAM scheme will hereinafter be described with reference to FIG. 1. It should be noted that the 16QAM scheme has been disclosed for illustrative purposes and therefore the present invention can also be applicable to the remaining M-QAM schemes other than the 16QAM scheme.
FIG. 1 is a diagram illustrating a constellation of a conventional 16QAM scheme.
Referring to FIG. 1, an input signal of 4 bits includes 16 values (i.e., 24=16) ranging from '0000' to '1111', such that the 16 values can be mapping-processed on 16 positions contained in a complex plain. Two preceding bits from among the four bits are separated from each other according to individual quadrants contained in a complex plain, and two following bits from among the four bits are separated from each other according to four fields partitioned in a prescribed quadrant. For example, two preceding bits of all symbols contained in a first quadrant of the complex plain are assigned '00', the first quadrant is divided into four fields, and these four fields are respectively assigned '00', '01', '10', and '11' each serving as two following bits.
In this manner, a signal modulated into 16 symbols is transmitted to a receiver. The receiver demodulates the signal composed of 16 symbols into corresponding bits according to the amplitude and phase of individual symbols. Individual symbol values are determined on the basis of a specific position having an amplitude of 2A. For example, if a Q-channel amplitude is higher than the amplitude of 2A in the first quadrant of the complex plain, symbol values '0001' and '0011' are separated from each other on the basis of a specific position having an I-channel amplitude of 2A. In this case, such a boundary value (e.g., 2A) capable of distinguishing a plurality of symbols from each other is called a decision boundary.
In case of transferring the aforementioned symbols over a wireless channel environment, the symbol signals incur considerable distortion due to signal fading or noise.
Therefore, the signal mapping-processed with the aforementioned symbols may be unexpectedly higher than the decision boundary such that it could be mistaken for wrong symbols. A reception error rate is determined according to the number of wrong symbols.
Therefore, the higher-order modulation scheme such as the 16QAM scheme can enhance a data transfer rate, but it has a disadvantage in that it must estimate individual phases and amplitudes of plural symbols in order to allow the receiver to perform symbol decision whereas the QPSK scheme can discriminate between such symbols on the basis of only the phase. Therefore, there must be newly developed a technique capable of estimating more reliable and reasonable decision boundaries in consideration of signal distortion caused by channel environments.
As described above, a demodulator of the receiver adapts the decision boundary to perform soft or hard decision of symbols, and it is preferable for information related to the demodulator to be estimated from a reception signal.
Provided that there are numerous transmission symbols, they are stochastically and equally distributed on the constellation shown in FIG. 1. There has recently been proposed a variety of schemes for estimating such a decision boundary using the aforementioned characteristic, for example, a first scheme for calculating a mean absolute value by accumulating absolute values of reception symbols during a predetermined period of time, and an unbiased estimation scheme considering noise dispersion to remove a bias component caused by noise, etc.
Typically, the first scheme for calculating a decision boundary of reception symbols by acquiring a mean value of absolute values of reception signals is effectively used in the case of a high Signal to Noise Ratio (SNR), and the unbiased estimation scheme is effectively used in the case of a low SNR.
The aforementioned estimation schemes are different in terms of performance according to the SNR. Provided that a decision boundary estimation process is performed on the condition that a sufficient accumulation period has been provided, the first scheme and the unbiased estimation scheme converge on an ideal decision boundary contained in the symbol constellation of symbols.
However, the aforementioned schemes must accumulate symbols for a long period of time to estimate the decision boundary, such that they require a long processing-time to accumulate the reception symbols. In addition, if the reception symbols are unevenly created on the constellation, the aforementioned schemes may cause undesired performance deterioration.
For example, the above scheme for acquiring an amplitude level of a reception symbol by averaging absolute values of reception signals is applied to the High Speed Downlink Packet Access (HSDPA), one of the high-speed data transmission schemes, and problems associated with this exemplary scheme will hereinafter be described in detail.
First of all, the HSDPA, one of the high-speed data transmission schemes, will hereinafter be described.
The HSDPA scheme is a general term of a prescribed data transmission scheme including an High Speed-Downlink Shared Channel (HS-DSCH) serving as a downlink data channel for supporting high-speed downlink packet data transmission in a Universal Mobile Terrestrial System (UMTS) and control channels associated with the HS-DSCH. There have recently been proposed an Adaptive Modulation and Coding (AMC) scheme, a Hybrid Automatic Retransmission Request (HARQ) scheme, and an Fast Cell Select (FCS) scheme to support the HSDPA scheme, and their detailed description will herein be omitted because they have no connection with the present invention.
The HSDPA scheme transmits data over a High-Speed Physical Downlink Shared CHannel (HS-PDSCH) functioning as a traffic channel, and at the same time transmits data over a (Physical Common PIlot CHannel) (PDPICH) functioning as a pilot channel. The HS-PDSCH can transmit data, and can also transmit such data according to 16QAM and QPSK schemes. For the purpose of simplicity, it is assumed that data transmission is performed using only the 16QAM scheme.
The PCPICH is a pilot channel, and is adapted to estimate a phase of a reception signal by transmitting prescribed symbols (e.g., an operation for continuously transmitting signals of '1') between a transmitter to a receiver.
In the meantime, a narrow-band modulation signal dk of a k-th HS-PDSCH from among a plurality of HS-PDSCHs can be represented by the following Equation 1:
                                          d            k                                          (            t            )                          =                              ∑                          -              ∞                        ∞                                                                                A              d                        ·                                          g                k                                                          (                i                )                                      ·                          exp                                            [                                  j                                                                                Φ                      k                                                                                (                      ⅈ                      )                                                                      ]                                      ·                          u                                            (                                                      t                    /                                          T                      s                                                        -                  ⅈ                                )                                                                        [                  Equation                                                              1                ]
where, gk(i)=√{square root over (Ik2+Qk2)}, |I|,|Q|ε{A,3A},
                    Φ        k                          (        i        )              =                  tan                  -          1                                            Q          k                          I          k                      ,and u(t) is a unit step function. As is well known in the art, the unit step function u(t) can be represented by the following Equation 2:
                              u                                (            t            )                          =                  (                                                                      1                  ,                                                                                for                                                                                0                                    ≤                  t                  <                  1                                                                                0                  ,                                                                              otherwise                                                                                )                                                                [                  Equation                                                              2                ]
Referring to Equations 1 and 2, Ad is a constant where an amplitude of a transmission signal is reflected, gk is an amplitude of a corresponding symbol, and Φk(i) is a phase of the corresponding symbol.
Spread code waveform associated with the k-th HS-PDSCH from among a plurality of HS-PDSCHs is represented by the following Equation 3:
                                          c            k                                          (            t            )                          =                              ∑                          -              ∞                        ∞                                                        exp                                            [                                  j                                                                                Φ                      k                                                                                (                      ⅈ                      )                                                                      ]                                      ·                          u                                            (                                                      t                    /                                          T                      c                                                        -                  ⅈ                                )                                                                        [                  Equation                                                              3                ]
where, Φk is a complex channelization code, and is denoted by Φk(i)ε{νπ/2+π/4; ν=0,1,2,3}.
The PCPICH serving as a pilot channel for channel estimation is represented by the following Equation 4:
                                          d            cpich                                          (            t            )                          =                              ∑                          -              ∞                        ∞                                                        A              p                        ·                          g              cpich                        ·                          exp                                            (                                  j                                                                                π                    /                    4                                                  )                                      ·                          u                                            (                                                      t                    /                                          T                      cpich                                                        -                  ⅈ                                )                                                                        [                  Equation                                                              4                ]
Equation 4 is denoted in a manner similar to the traffic modulation signal of the HS-PDSCH shown in Equation 1. As can be seen from Equation 4, the PCPICH continuously transmits specific symbol signals, such that the reference character 'g' indicative of an amplitude and the reference character 'Φk' indicative of a phase are denoted by constants, respectively. For example, 'g' may be set to 'gcpich' and 'Φk' may be set to π/4. The other reference character 'Ap' is a constant for reflecting an amplitude of a transmission signal associated with a pilot signal in the same way as in 'Ad'.
Similarly to the spread code waveform associated with the HS-PDSCH, spread code waveform of the PCPICH is denoted in a manner similar to the traffic modulation signal, and different code values are assigned to individual channels. In this case, codes assigned to individual channels are orthogonal codes as well known in the art. In conclusion, the spread code waveform of the PCPICH can be represented by the following Equation 5:
                                          c            cpich                                          (            t            )                          =                              ∑                          -              ∞                        ∞                                                        exp                                            [                                  j                                                                                Φ                      cpich                                                                                (                      ⅈ                      )                                                                      ]                                      ·                          u                                            (                                                      t                    /                                          T                      cpich                                                        -                  ⅈ                                )                                                                        [                  Equation                                                              5                ]
where, gcpich is an amplitude of the PCPICH, and Tcpich is a symbol period.
Provided that one HS-PDSCH and one PCPICH are transmitted over a transmitter, a transmission signal r(t) can be represented by the following Equation 6:r(t)=d1(t)·c1(t)+dcpich(t)·ccpich(t)  [Equation 6]
With reference to Equation 6, the modulation signal di(t) of the HS-PDSCH is multiplied by the channelization code waveform c1(t), thereby creating a signal denoted by d1(t)·c1(t). The modulation signal dcpich(t) of the PCPICH is multiplied by the channelization code waveform ccpich(t), thereby creating the other signal denoted by dcpich(t)·ccpich (t). If the signal denoted by d1(t)·c1(t) is added to the other signal denoted by dcpich(t)·ccpich(t), there is provided the resultant signal r(t) transferred over the transmitter.
In this case, if the signal r(t) transferred over the transmitter is received in the receiver, it is affected by multi-path fading and noise, such that it can be represented by the following Equation 7:r(t)=h(t)·[d1(t−τ1)·c(t−τ1)+dcpich(t−τ1)·c(t−τ1)]n(t)  [Equation 7]
where, h(t) is a complex channel gain, τ1 is a predetermined time delay, and n(t) is a noise component added to the reception signal r(t). Typically, n(t) indicates a noise component having spectrum density of No/2.
The reception signal r(t) shown in Equation 7, that has been affected by the fading and noise and then transmitted to the receiver, is adapted to estimate a channel environment in the receiver, such that an original transmission signal is effectively demodulated.
The receiver will hereinafter be described with reference to FIG. 2.
Referring to FIG. 2, the despreader 100 despreads the reception signal r(t). In this case, the output signal z(n) of an n-th symbol that has been despread by the despreader 100 and then transmitted to the channel compensator 120 is represented by the following Equation 8:
                              z                                (            n            )                          =                              1                          T              s                                                                    ∫                                                T                                                                                s                                +                                  τ                  ^                                                                                (                                          n                      +                      1                                        )                                                      T                                                                                s                                +                                  τ                  ^                                                                                r                                                        (                    t                    )                                                  ·                                                      C                    *                                                                          (                                          t                      -                                              τ                        ^                                                              )                                                                                ⅆ                t                                                                        [                  Equation                                                              8                ]
With reference to Equation 8, the despread output signal z(n) can be calculated by performing a convolution operation between the reception signal r(t) and the channelization code during a predetermined symbol period Ts. In this case, '*' is a complex conjugate, and {circumflex over (τ)} is an estimated time delay.
The PCPICH signal transferred to perform channel estimation is despread by the despreader 100, and is then transmitted to the channel estimator 110. The PCPICH signal Zcpich(n) having been despread and transmitted to the channel estimator 110 can be represented by the following Equation 9:
                                          z            cpich                                          (            n            )                          =                              1                          T              cpich                                                      ∫                                                            r                                                        (                    t                    )                                                  ·                                                      C                    cpich                    *                                                                          (                                          t                      -                                              τ                        ^                                                              )                                                                                ⅆ                t                                                                        [                  Equation                                                              9                ]
where, zcpich(n) is an output value of the n-th symbol of the PCPICH serving as a pilot channel.
In this case, provided that channel estimation is ideally performed and no noise occurs, ĥ(n) created by transmitting the n-th symbol output signal zcpich(n) of the PCPICH over the channel estimator 110 can be represented by the following Equation 10:ĥ(n)=Ap·h(n)  [Equation 10]
where, Ap is a constant indicative of an amplitude of a transmission signal associated with a pilot signal, and h(n) is a complex channel gain.
ĥ(n) generated from the channel estimator 110 performs channel estimation on a reception signal generated from the despreader 100. Specifically, the reception signal is affected by the channel environment according to the channel environment condition having been estimated by the channel estimator 110, such that a distortion component can be compensated. For example, provided that the reception signal causes an undesired phase variation of θ over a wireless environment, it can be restored to the original transmission signal by compensating with a phase value of −θ. The above signal ĥ(n) is applied to the channel compensator 120 and the decision boundary estimator 130. The reception signal z(n) having been despread by the despreader 110 as shown in Equation 8 performs a predetermined operation on the channel estimation value ĥ(n), thereby creating a channel-compensated output signal.
The output value d(n) having been channel-compensated by the channel compensator 120 can be represented by the following Equation 11:d(n)=ĥ*(n)·z(n)=∥h(n)∥2ApAd·g(n)[Equation 11]
A predetermined scalar product operation between a complex conjugate of the signal ĥ(n) generated from the channel estimator 110 and the output value z(n) of the despreader 100 is performed, resulting in a channel-compensation signal. With reference to Equation 11, channel distortion contained in the channel-estimation signal is normally compensated, such that the channel-estimation signal is composed of only four signals, i.e., h(n), Ap, Ad and g(n).
The demodulator 140 determines a mapping position contained in the constellation in association with the n-th symbol signal d(n) generated from the channel compensator 120. In the meantime, provided that the reception signal is an ideal signal having no distortion over a wireless channel, it can be correctly determined by a specific decision boundary (e.g., 2A).
However, since the reception signal is affected by real-time distortion created over a wireless channel, it is preferable for a substantial decision boundary to be adaptively determined according to the channel environment.
Therefore, upon receiving the channel-compensation reception signal d(n) from the channel compensator 120, the decision boundary estimator 130 preferably estimates a decision boundary in consideration of channel environment. The demodulator 140 modulates reception signal d(n) into bit values according to the decision boundary estimated by the decision boundary estimator 130. For example, in case of the 16QAM scheme, the reception signal d(n) is modulated into four bits for every symbol.
Bit data demodulated by the demodulator 140 is decoded by the decoder 150. The UMTS system typically adapts a turbo decoder as a decoder.
The decision boundary estimator will hereinafter be described with reference to FIG. 3.
FIG. 3 is a detailed block diagram of the decision boundary estimator 130 shown in FIG. 2.
Referring to FIG. 3, a complex signal shown in Equation 11 has been generated from the channel estimator 120 shown in FIG. 2, and is classified into an In-phase component and a Quadrant-phase component to perform a real-number operation. In more detail, the In-phase component contained in the output signal of the channel compensator 120 is transmitted to a first absolute value calculator 200, and the Quadrant-phase component is transmitted to a second absolute value calculator 220. In the meantime, the output signal of the channel estimator 110 is a channel gain denoted by Equation 10, and is transmitted to a third absolute value calculator 240.
Upon receiving the channel-compensated output signals (i.e., In-phase and Quadrature-phase components) from the channel compensator 120, the first and second absolute value calculators 200 and 220 calculate individual absolute values of the received signals, and the first accumulator 210 and the second accumulator 230 perform an accumulation and dump operation on the calculated absolute values.
The signals having been applied to the first and second absolute value calculators 200 and 220 each include a real part and an imaginary part, and are processed while being classified according to the real part and the imaginary part. However, the signals are equivalent to the complex symbol expression, such that the same signal processing step is applied to them. Therefore, the first accumulator 210 is adapted to accumulate the absolute value of N signals having the In-phase component, and the second accumulator 230 is adapted to accumulate the absolute value of N signals having the Quadrature-phase component. The absolute-value accumulated signals created from the first and second accumulators 210 and 230 are added to each other in the adder 260, this value is divided by 4 using the ¼ divider 270, and a resultant signal is transmitted to another divider 280.
In the meantime, if the output signal of the channel estimator 110 is transmitted to the third absolute value calculator 240, the third absolute value calculator 240 numerically squares an absolute value of the received signal, and the squared signal is transmitted from the third absolute value calculator 240 to the third accumulator 250. The third accumulator 250 accumulates the received signal by N symbols, and transmits the accumulated signal Y to the divider 280.
A division operation between the two signals X and Y applied to the divider 280 is performed. In more detail, the divider 280 divides the sum value X created by accumulating the absolute value of the complex signal by the output value Y of the channel estimator 110.
In this case, the signal is generated from the channel compensator 120, and is then transmitted to the divider 280 after accumulating its own absolute value. This signal can be represented by the following Equation 12:
                                                                                ∑                                          n                      =                      1                                        N                                                                                                                                                                                                                      h                                                                                (                              n                              )                                                                                                                                2                                            ·                                              A                        d                                            ·                                              A                        p                                            ·                                                                                                Re                                                                                {                                                          g                                                                                (                                n                                )                                                                                }                                                                                                                                                                                          +                                                                                ∑                                      n                    =                    1                                    N                                                                                                                                                                                                                  h                                                                                (                            n                            )                                                                                                                      2                                        ·                                          A                      d                                        ·                                          A                      p                                        ·                                                                                        Im                                                                          {                                                      g                                                                                (                              n                              )                                                                                }                                                                                                                                                                                                4                            [                  Equation                                                              12                ]
With reference to Equation 12, the output signal of the channel compensator 120 is divided into a real part and an imaginary part, and its absolute value is calculated while being classified according to the real part and the imaginary part. The calculated absolute value is accumulated during a predetermined time corresponding to N symbols, and the accumulated result is divided by 4.
In the meantime, an output signal of the channel estimator 110 is applied to the decision boundary estimator 130, an absolute value of the output signal is accumulated, and the accumulated result is applied to the divider 280. In this case, the signal applied to the divider 280 is represented by the following Equation 13:
                              ∑                      n            =            1                    N                                                                                        h                                                  (                  n                  )                                            ·                              A                p                                                          2                                    [                  Equation                                                              13                ]
With reference to Equation 13, an absolute value of ĥ(n) denoted by Equation 10 is squared, the squared result is accumulated during a predetermined time corresponding to N symbols, and the accumulated result is transferred to the divider 280 in such a way that the signal denoted by Equation 13 is created.
Finally, the divider 280 divides the signal denoted by Equation 12 by the other signal denoted by Equation 13. That is, the divider 280 performs division operation on the output signal of the decision boundary estimator 130, such that the last output signal of the decision boundary estimator 130 can be represented by the following Equation 14:
                                          θ            ^                    1                =                                            A              d                                      A              p                                          A                                    [                  Equation                                                              14                ]
As described above, considering the fact that the absolute value of mean amplitudes of the real and imaginary parts converge on a specific value of 2A after the lapse of a long period of time, the decision boundary estimator 130 normalizes the sum of individual absolute values with a channel gain, and divides the normalized value by 4. The normalized value is divided by 4 because the sum of a mean value (A+3A)/2 of the real part and a mean value (A+3A)/2 of the imaginary part converges on a specific value of 4A on the condition that the value of N is sufficiently high and creation frequencies of all symbols are equal to each other.
As previously shown in FIG. 2, the output signal of the channel compensator 120 and the output signal of the decision boundary estimator 130 are transmitted to the demodulator 140, and are adapted to set up a reference amplitude for determining a symbol. Therefore, a soft-decision process of the output value of the channel compensator 120 is performed by the output value of the decision boundary estimator 130.
For reference, there are two quantization methods for symbols received over the channel, i.e., a hard decision method and a soft decision method. In more detail, data received over the channel is slightly different from a predetermined data value modulated in a transmission step. The hard decision method determines the reception data using only two levels '1' and '0' according to a predetermined reference. The soft decision method converts a baseband signal into digital data having a specific level of 2n using a soft-decision threshold value predetermined by an N-bit Analog-to-Digital Converter (ADC). There is a little difference in a coding gain according to a quantization method of the demodulated data. For example, throughput performance of a 16-level soft decision method is superior to that of the hard decision method by about 2 dB, such that most digital communication systems mainly prefer the soft decision method to the hard decision method.
Provided that the aforementioned conventional decision boundary {circumflex over (θ)}1, and the output signal z(n) are adapted as input signals and a simple metric method is adapted to calculate a soft Long Likelihood Ratio (LLR), a signal composed of 4 bits is created by demodulating a predetermined symbol. This 4-bit signal can be represented by the following Equations 15˜18:Λ0(z(n))=In  [Equation 15]Λ1(z(n))=Qn  [Equation 16]Λ2(z(n))=2·A−|In|  [Equation 17]Λ3(z(n))=2·A−|Qn|  [Equation 18]
Referring to the above Equations 15˜18, Λi(z(n)) indicates an LLR of an i-th bit of the n-th symbol. In more detail, Equation 15 indicates a first bit from among the output bits (i.e., 4 bits) associated with a single symbol, and Equation 18 indicates the last bit from among the four bits.
In this case, Equation 15 and Equation 16 indicate two preceding bits from among the above four bits, respectively, such that they are adapted to determine a quadrant of a complex plain. Equation 17 and Equation 18 indicate two following bits from among the four bits, such that they are adapted to discriminate among four fields of a predetermined quadrant. In this case, 'In' indicates an In-phase component of the n-th symbol, and is represented by the following Equation 19. 'Qn' indicates a Quadrature-phase component of the n-th symbol, and is represented by the following Equation 20.In=Re{z(n)}  [Equation 19]Qn=Im{z(n)}  [Equation 20]
Since 'In' and 'Qn' are channel-compensated signals, Λ2 can be represented by the following Equation 21 using its corresponding gain and an estimation value {circumflex over (θ)}1, of a reference level, and Λ3 can be represented by the following Equation 22 using its corresponding gain and the estimation value {circumflex over (θ)}1 of the reference level.Λ2(z(n))=2·{circumflex over (θ)}1·Ap2·∥h(n)∥2−|In|  [Equation 21]Λ3(z(n))=2·{circumflex over (θ)}1·Ap2·∥h(n)∥2−|Qn|  [Equation 22]
Finally, the aforementioned four-bit output signals associated with 16QAMs of the Equations 15, 16, 21, and 22 are applied to the decoder 150, and are then decoded by the decoder 150.
As described above, the scheme for averaging the accumulated absolute values of reception symbols and the other scheme for estimating the decision boundary using a mean accumulation value of the squared absolute value are different in performance according to a reception SNR, however, they are all designed on the assumption that transmission symbols are evenly distributed on the constellation and are then transmitted to a target object. Therefore, the conventional decision boundary estimator necessitates numerous reception symbol samples (e.g., symbols corresponding to about one packet), such that it unavoidably create an unnecessary time delay as long as a prescribed accumulation time needed for the decision boundary estimation process. Furthermore, the conventional decision boundary estimator unexpectedly incurs performance deterioration depending on the SNR.
In conclusion, the scheme for averaging the accumulated absolute value of reception symbols and the other scheme for adapting the square of the mean accumulation value each incur an undesired time delay corresponding to the accumulated samples, such that it is difficult for either one of them to be applied to hardware (e.g., an interference canceller for requesting a real-time processing) to be processed at a high speed, for example, an interference canceller for requesting a real-time processing.

Apparatus and method for estimating a decision boundary in symbol units in a mobile communication system


A method of determining exposure of an occupant (8) to equivalent melanopic lux, EML, in an environment (2) comprising one or more indoor spaces and zero or more outdoor spaces, wherein the environment comprises one or more illumination devices (4, 5) wherein said one or more illumination devices (4, 5) comprise one or more luminaires (4) illuminating the environment with artificial illumination. The method comprises: —accessing light level information from one or more information sources (6, 12) wherein the light level information comprises information at least on the artificial illumination, and based at least in part on said information on the artificial illumination determining where in the environment at least a predetermined EML level is present; —using a tracking system (16) to track the occupant within the environment; and —based on this accessing and tracking, monitoring whether the occupant has been exposed to at least a minimum EML quota, the minimum EML quota comprising the occupant being exposed to at least the predetermined EML level for at least a predetermined time period.

WO 2017/042682 A1 discloses a system for computing solar radiation seen by an individual, the system comprising: portable location means for locating and tracking the position of an individual in the course of a day of direct or indirect exposure to the sun; data-collection means containing satellite data d2 regarding solar radiation in the traced position of the individual; computing means operatively connected to said location means and to said data-collection means for receiving data d1 regarding the position of the individual and data d2 regarding solar radiation on the ground in the same position where the individual is in order to compute the dose R1 of solar radiation seen by the individual and to compute a dose R3 of solar radiation that is missing or is in excess with respect to an expected dose R2 corresponding to a desired personal circadian rhythm of the individual.
New lighting recommendations are published in the WELL Building Standard (http://standard.wellcertifled.com/light/circadian-lighting-design). This defines a recommended melanopic daylight (D65) equivalent, referred to as EML (equivalent melanopic lux), for a variety of scenarios such as work places (e.g. work stations in offices). EML is measured in the vertical plane at a predetermined height above the finished floor corresponding to a typical eye level of the occupant. EML also takes into account luminous intensity and colour temperature. The WELL building standard specifies that EML levels are to be measured during performance verification (the commissioning stage) in order to ensure compliance.
Luminous flux is the measure of the perceived power of light. It is equal to the electromagnetic radiant flux in the visible spectrum weighted per unit frequency according to the luminosity function, which represents the human eye's response to different wavelengths of light. The SI unit is Lumen (lm). Lux (Lx) is the SI unit of luminous flux per unit area (illuminance), a quantity which may be referred to by the symbol E. Sometimes as a shorthand the quantity E itself (as opposed to the SI unit) is referred to as the "lux" (strictly the illuminance). EML is defined as the lux Ev in a vertical plane or vertical cylindrical surface multiplied by a dimensionless ratio MEF (melanopic equivalence factor), referred to in the WELL Building Standard as the melanopic ratio R. The MEF is a function of correlated colour temperature (CCT) of the lux component. It is a metric used to measure the effect of light on a human's circadian rhythm.
The WELL Building Standard®, Version 1.0 of Oct. 20, 2014, downloadable at https://greenplantsforgreenbuildings.org/wp-content/uploads/2014/09/THE-WELL-BLDG-STD-102414-FINAL_sml.pdf discloses on page 189-191 the following:
TABLE L1: MELANOPIC RATIO
This unit Equivalent Melanopic Lux (EML) was proposed by Lucas and others (Lucas et al., "Measuring and using light in the melanopsin age." Trends in Neuroscience, Jan 2014). The authors provided a toolbox which for a desired spectrum derives equivalent "α-opic" lux for each of the five photoreceptors in the eye (three cones, rods, and the ipRGCs). The authors selected scaling constants such that each of the values would be identical to each other and the standard definition of lux for a light spectrum of perfectly uniform energy (CIE Standard Illuminant E).
Given a spectrum of light, each equivalent α-opic lux is related to each other by a constant. The table below shows the example ratios between the equivalent melanopic lux and the standard visual lux for several sources.
To calculate the equivalent melanopic lux (EML), multiply the visual lux (L) designed for or measured in a building by this ratio (R): EML=L×R. For example, if incandescent lights provide 200 lux in a space, they will also produce 108 equivalent melanopic lux. If daylight is modeled to provide the same visual brightness (200 lux), it will also provide 220 equivalent melanopic lux.
Similar melanopic ratios can be determined by incorporating the spectrum of the desired source into the calculations in Table L2. Projects are encouraged to use this approach to obtain more accurate results. Both the authors of the journal article and the IWBI have spreadsheets to aid in this calculation.
TABLE L2MELANOPIC AND VISUAL RESPONSECCT (K)LIGHT SOURCERATIO2950Fluorescent0.432700LED0.452800Incandescent0.544000Fluorescent0.584000LED0.765450CIE E (Equal Energy)1.006500Fluorescent1.026500Daylight1.107500Fluorescent1.11
To calculate the melanopic ratio of light, start by obtaining the light output of the lamp at each 5 nm increment, either from manufacturer or by using a spectrometer. Then, multiply the output by the melanopic and visual curves given below to get the melanopic and visual responses. Finally, divide the total melanopic response by the total visual response.
LightMelanopicMelanopicVisualVisualWavelengthOutputCurveResponseCurveResponse380. . .0.00015. . .0.00004. . .385. . .0.00031. . .0.00006. . .390. . .0.00063. . .0.00012. . .395. . .0.00133. . .0.00022. . .400. . .0.00285. . .0.0004. . .405. . .0.00625. . .0.00064. . .410. . .0.01384. . .0.00121. . .415. . .0.02601. . .0.00218. . .420. . .0.04887. . .0.004. . .425. . .0.07232. . .0.0073. . .430. . .0.10624. . .0.0116. . .435. . .0.14492. . .0.01684. . .440. . .0.19552. . .0.023. . .445. . .0.24805. . .0.0298. . .450. . .0.31075. . .0.038. . .455. . .0.37816. . .0.048. . .460. . .0.45413. . .0.06. . .465. . .0.53676. . .0.0739. . .470. . .0.62557. . .0.09098. . .475. . .0.70857. . .0.1126. . .480. . .0.79016. . .0.13902. . .485. . .0.85767. . .0.1693. . .490. . .0.91449. . .0.20802. . .495. . .0.95668. . .0.2586. . .500. . .0.98081. . .0.323. . .505. . .0.98453. . .0.4073. . .510. . .0.96695. . .0.503. . .515. . .0.9224. . .0.6082. . .520. . .0.86033. . .0.71. . .525. . .0.78346. . .0.7932. . .530. . .0.69706. . .0.862. . .535. . .0.60543. . .0.91485. . .540. . .0.51274. . .0.954. . .545. . .0.42335. . .0.9803. . .550. . .0.33998. . .0.99495. . .555. . .0.26511. . .1. . .560. . .0.20105. . .0.995. . .565. . .0.14819. . .0.9786. . .570. . .0.10672. . .0.952. . .575. . .0.0752. . .0.9154. . .580. . .0.05217. . .0.87. . .585. . .0.03578. . .0.8163. . .590. . .0.02434. . .0.757. . .595. . .0.01645. . .0.6949. . .600. . .0.01109. . .0.631. . .605. . .0.00747. . .0.5668. . .610. . .0.00503. . .0.503. . .615. . .0.0034. . .0.4412. . .620. . .0.00231. . .0.381. . .625. . .0.00157. . .0.321. . .630. . .0.00107. . .0.265. . .635. . .0.00073. . .0.217. . .640. . .0.0005. . .0.175. . .645. . .0.00035. . .0.1382. . .650. . .0.00024. . .0.107. . .655. . .0.00017. . .0.0816. . .660. . .0.00012. . .0.061. . .665. . .0.00008. . .0.04458. . .670. . .0.00006. . .0.032. . .675. . .0.00004. . .0.0232. . .680. . .0.00003. . .0.017. . .685. . .0.00002. . .0.01192. . .690. . .0.00002. . .0.00821. . .695. . .0.00001. . .0.00572. . .700. . .0.00001. . .0.0041. . .705. . .0.00001. . .0.00293. . .710. . .0. . .0.00209. . .715. . .0. . .0.00148. . .720. . .0. . .0.00105. . .725. . .0. . .0.00074. . .730. . .0. . .0.00052. . .735. . .0. . .0.00036. . .740. . .0. . .0.00025. . .745. . .0. . .0.00017. . .750. . .0. . .0.00012. . .755. . .0. . .0.00008. . .760. . .0. . .0.00006. . .765. . .0. . .0.00004. . .770. . .0. . .0.00003. . .775. . .0. . .0.00002. . .780. . .0. . .0.00001. . .Totals:. . .. . .
Note that EML may equivalently also be defined based on other intensity and spectral information indicative of the lux and colour temperature. E.g. using foot-candles as the measure of intensity in the visible spectrum (i.e. illuminance), a factor equivalent to the MEF can be defined which gives the same EML value for illuminance measured in foot-candles as the above equation would for the same illuminance measured in lux.
EML is thus a measure of adjusted lux per unit area in a defined vertical plane or surface. In the norm today it is measured in the vertical plane but it is also possible this will be changed or added to it that it can can also be measured in the cylindrical way. The WELL standard also specifies that such EML levels should be met at specific heights above the finished floor. This is illustrated schematically in Figure la and lb. Figure la shows a vertical plane VP through a region of a space such as a room, and a horizontal component Ev of the lux in that space as projected onto an elementary area dA in the plane VP at a height h above the floor. The EML is defined as the lux E falling on dA (the luminous flux per unit area) multiplied by MEF(CCT) where CCT is the colour temperature of the lux component E. FIG. 1b shows the alternative measure of EMF, based on a horizontal component of the lux Ev as projected onto an elementary area dA at a height h on the surface of a vertical cylindrical surface VC. Note the cylindrical surface over which the component is measured doesn't have to be a full cylinder. E.g. the EMF may be averaged over a half cylindrical surface, or measured at an elementary area dA on a cylindrical surface at a particular point.
For work areas, the WELL standard specifies that at least one of the following requirements is met. (a) At 75% or more of workstations, at least 200 EML is present, measured on the vertical plane facing forward, 1.2 m above finished floor. This light level may incorporate daylight, and is present for at least the hours between 9:00 am and 1:00 pm for every day of the year. (b) For all workstations, electric lights (which may include task lighting) provide maintained illuminance on the vertical plane facing forward of 150 EML or greater.
Breakrooms: for workplaces where employees spend most of their time in spaces with light levels limited by work type (such as restaurant servers or hospital ward workers), the WELL standard specifies that such workplaces have break rooms which meet the following requirement. Lights provide a maintained average of at least 250 EML as measured on the vertical plane facing forward at surfaces 1.2 m above finished floor. The lights may be dimmed in the presence of daylight, but are able to independently achieve these levels. In living environments (e.g. bedrooms, bathrooms, and rooms with windows), the WELL standard specifies the following. (a) 200 or more EML as measured facing the wall in the centre of the room 1.2 m above the finished floor. The lights may be dimmed in the presence of daylight, but are able to independently achieve these levels. (b) Lights provide not more than 50 EML (to the extent allowable by code) as measured 0.76 m above the finished floor.
In learning areas, the WELL standard specifies that at least one of the following requirements is met. (a) Early education, elementary, middle and high schools, and adult education for students primarily under 25 years of age: the light (which may incorporate daylight) must provide at least 125 EML at 75% or more of desks, on the vertical plane facing forward 1.2 m above finished floor. This light level is present for at least 4 hours per day for every day of the year. (b) Ambient lights provide maintained illuminance on the vertical plane of EML greater than or equal to the lux recommendations in the Vertical (Ev) Targets in Table 3 of IES-ANSI RP-3-13, following the age group category most appropriate for the population serviced by the school. For example, art studios in elementary school, middle school, or high school are provided with 150 EML from the electric lights.
Separately from the WELL standard, one known technique for monitoring a person's exposure to light is to wear a wearable light sensor device on a pendant around the neck.


The present invention relates to biomarkers associated with CTCL, including TOX, PLS3, KIR3DL2, GATA3 and RUNX3, where increased expression, relative to normal control, of one or more of TOX, PLS3, KIR3DL2, and/or GATA3 is associated with CTCL and decreased expression of RUNX3, relative to normal control, is associated with CTCL. One or more of these biomarkers may be used to diagnose CTCL and/or design and monitor treatment.

T-cell lymphoma that involves the skin is generally known as cutaneous t-cell lymphoma (CTCL). The term CTCL encompasses a number of disorders, including mycosis fungoides (MF), which is the most common form of CTCL. Sézary syndrome (SS) is an advanced, variant form of mycosis fungoides, characterized by the presence of malignant lymphocytes in the blood. See "Getting the Facts" monograph for "Cutaneous T Cell Lymphoma" published by the Lymphoma Research Foundation, 115 Broadway Suite 1301, New York N.Y. 10006 (last update January 2013). No specific diagnostic or prognostic markers exist to enable early diagnosis of MF, SS and CTCL. The pathogenesis of CTCL is unknown and there are no therapeutic targets available for disease-specific therapies. Because no specific diagnosis is possible, CTCL is often confused with other inflammatory dermatoses, such as psoriasis or eczema, and patients often live with the disease for years before a correct diagnosis is established. If diagnosed early, CTCL patients have a longer lifespan, but the survival rate goes down as CTCL reaches advanced stages. Thus, a diagnostic test that could easily confirm the presence of CTCL early in disease course is needed.

This patent is subject to a terminal disclaimer.

Method for diagnosis, prognosis and determination of treatment for cutaneous t-cell lymphoma

UNIVERSITY OF PITTSBURGH—OF THE COMMONWEALTH SYSTEM OF HIGHER EDUCATION


A multi-radio device includes two or more radio subsystems coupled to a packet traffic arbitration (PTA) module. The PTA module receives a first request for radio communication from a first radio subsystem, where the first request indicates a first usage duration. After granting the first request, the PTA module receives a second request for radio communication from a second radio subsystem. The second request indicates a second usage duration. The PTA module sends a rejection of the second request to the second radio subsystem, where the rejection indicates a remaining time of the first usage duration.

Wireless local area networks (WLAN) connect multiple devices via wireless communication within a limited area (e.g., ˜100 meters) such as a home, school, laboratory, campus, or office building. A WLAN enables its users to move around within the WLAN area and remain connected to the network. A wireless device in a WLAN can connect to the Internet via a router or an access point. Many modern WLANs are based on the IEEE 802.11 standards, commonly known as Wi-Fi.
Wireless personal area networks (WPANs) provide wireless communication over short distances (e.g., from a few centimeters to a few meters) among a group of devices, such as computers, smartphones, tablets, laptops, headsets, and personal digital assistants (PDAs). A WPAN provides a power-efficient and inexpensive communication mechanism for a wide range of devices. Examples of WPAN protocols include Bluetooth, ZigBee, wireless USB, ultra-wideband (UWB), etc.
Modern communication and computing devices are typically equipped with multiple radios to enable wireless connections according to a variety of radio protocols. For example, a multi-radio device such as a smartphone typically includes radio circuitry for Wi-Fi and Bluetooth connections. In addition, a multi-radio device also typically includes radio circuitry for connecting to broadband wireless networks such as a 4G LTE network and/or a 5G NR network. The radios for different wireless networks co-exist in close physical proximity. The radios in some scenarios may also share the same frequency band. If two or more radios operate simultaneously, they can potentially interfere with each other. For example, when one radio transmits a signal as another radio receives a signal, the received signal may be corrupted by the transmitted signal. There is a need for coexistence of radios to operate in a shared environment of a multi-radio device. Furthermore, there is a need for providing a coexistence mechanism to improve or reduce interference among the radios in a multi-radio device.

Low-power coexistence mechanism based on packet traffic arbitration