A method and apparatus for a Gigabit Ethernet transceiver that has a Class-B amplifier in a main transmitter for higher efficiency and power handling capabilities. The output current of the main transmitter is produced by a reference voltage applied across a resistor, where the reference voltage generator, resistor and amplifier are fabricated on the same substrate, such that the output current is constant across process voltage and temperature. The transceiver also has a replica transmitter whose signal is used to cancel the main transmitter signal at the input of the receiver section of the transceiver. The replica transmitter is fabricated on the same substrate as the main transmitter, such that its output signal reflects non-linearities in the main transmitter across process voltage and temperature.

A local area network operating under the Gigabit Ethernet standard includes transceiver nodes communicating with each other in a full-duplex mode; that is, transmitting in both directions simultaneously at full bandwidth within a particular frequency band. The transceivers typically have a transformer at their output, with a communication cable electrically connecting the transformers of the two transceivers.
Within the transceiver, both a transmitter section and a receiver section are coupled to the transformer and, necessarily, to each other. Without additional circuitry, the signal at the input to the receiver will include both a received signal from the communication cable and a signal from the transmitter section. The received signal, having been attenuated in the cable, will typically be at a lower signal strength than the signal from the transmitter section within the transceiver.
Signal isolation between the transmitter and receiver sections of a transceiver may be obtained with a costly, transformer-based, hybrid electrical bridge circuit. However, the degree of isolation provided by such bridge circuits is dependent upon how closely the terminating impedance of the bridge matches the impedance of the communication cable. Because the impedance of the communication cable is dynamic, over time more or less of the transmitter signal is present at the input of the receiver to interfere with detection of the received signal.
An alternative technique for reducing the signal strength of the signal from the transmitter section involves generating within the transceiver a signal to cancel the transmitter signal. A signal that is identical to the transmitter signal, but having inverse polarity, is mixed with the combination of transmitter and received signal in order to cancel some portion of the transmitter signal. As a result, the signal-to-noise ratio of the received signal is improved.
However, the efficacy of this technique depends upon the degree to which the cancellation signal matches the transmitter signal. Differences between the two signals will result not only in some portion of the transmitter signal not being cancelled out, but may also inject additional noise into the signal at the input of the receiver. Merely matching the transmitter input signal is not enough, the non-linearities of the transmitter circuitry must also be matched in order for the technique to be fully effective.
Other challenges in implementing a Gigabit Ethernet network arise from the transformer coupling of the transceiver nodes. A transformer presents the transmitter circuitry with a characteristic impedance load. Because transmitters are typically implemented as current source/sink circuits, and the communication standards require that signal levels of a certain voltage be produced in the network, the power-handling requirement of transmitters is directly related to the impedance load presented by the transformers. As the impedance load is reduced, the transmitter must source/sink proportionately more current to produce the required signal voltages.
Since the communication cable is double terminated (i.e., is transformer coupled at both ends of the cable), the two transformers are in parallel and together present to the transmitter half the impedance of a single transformer. This requires a transmitter design that provides twice the power. Furthermore, where a transmitter is implemented with circuitry that only sinks current, the transformer is typically provided with a center tap connected to a voltage source. Because the transmitter is only driving half the transformer, the load impedance is halved again, requiring another doubling of the power handled by the transmitter circuitry.
Circuit designs typically used in Ethernet transmitters may not successfully scale up to the power levels required in such networks. The amplifiers and circuit sources used may become non-linear at these increased power levels, resulting in degraded functionality or failure to meet communications standards.
Before undertaking the DETAILED DESCRIPTION OF THE INVENTION below, it may be advantageous to set forth definitions of certain words and phrases used throughout this patent document: the terms "include" and "comprise," as well as derivatives thereof, mean inclusion without limitation; the term "or," is inclusive, meaning and/or; "each" means every one of at least a subset of the identified items; the phrases "associated with" and "associated therewith," as well as derivatives thereof, may mean to include, be included within, interconnect with, contain, be contained within, connect to or with, couple to or with, be communicable with, cooperate with, interleave, juxtapose, be proximate to, be bound to or with, have, have a property of, or the like; and the term "controller" means any device, system or part thereof that controls at least one operation, such a device may be implemented in hardware, firmware or software, or some combination of at least two of the same. It should be noted that the functionality associated with any particular controller may be centralized or distributed, whether locally or remotely. Definitions for certain words and phrases are provided throughout this patent document, those of ordinary skill in the art should understand that in many, if not most instances, such definitions apply to prior, as well as future, uses of such defined words and phrases.

Class-B transmitter and replica transmitter for gigabit ethernet applications


A hanger for supporting a plurality of pipettes adjacent a horizontal support surface. The hanger comprising contiguous mounting and pipette supporting portions. Each supporting portion comprises a pair of substantially parallel, laterally separated support arms extending forward from a back plate to vertically support a pipette body there between. Each mounting portion comprises a spacer extending rearward from the back plate spacing the back plate from a vertical side of a channel frame to which the hangers are mounted. The hangers are connected to the vertical side of the channel frame by screws extending through openings in the back plates of the hangers and into the vertical side of the channel frame. The channel frame, in turn, is secured to a horizontally extending support member by laterally spaced screw members extending vertically through a bottom side of the channel frame to engage the support member.

In order to repeatedly dispense measured quantities of liquid in clinical and diagnostic laboratories, it is common today to utilize manual and electronic pipettes. In use, a technician grasps the body of a pipette and places a removable tip end thereof in the fluid to be aspirated and dispensed by the pipette. The technician either allows manual or motor actuated movement of a piston within the body of the pipette to draw a measured quantity of liquid into the pipette tip. The technician then reverses the movement of the piston to dispense a precise quantity of liquid into a receptacle. When the dispensing operation is complete, the removable tip is ejected from the pipette and a new tip inserted onto the tip end of the pipette.
When the pipette is not in use, it is usually placed on a counter or bench top or in a drawer of a table or on a shelf in the laboratory where the pipette is normally used. Because it is important to avoid damage to the delicate mechanisms comprising the pipette and to maintain the pipettes in close proximity to the technician for ready access for operation, various racks, stands, and carrousels have been developed for supporting pipettes on table and counter tops and even by hanging on walls.
Typical of the pipette stands are the "Pipettor Stand/Work Station" model 290-002 and "Pipettor Stand" model 290-001 of PCG Scientifics, Gathersburg, Md., and the "Scienceware Micropipettor Racks" catalog numbers 14-791-8A and 14-791-8D from Fischer Scientific of Pittsburgh, Pa. The Pipettor Stand-Work Station comprises a small cabinet for resting on a table or counter top. A top of the cabinet includes a plurality of openings for each vertically receiving a lower body portion of a pipette to support a series of pipettes when not in use. The Pipettor Stand includes a flat base for resting on a table or counter top and a sloping back carrying upper and lower support arms with vertically aligned front facing notches for receiving and supporting the main bodies of a plurality of pipettes. The Scienceware Micro Pipettor Racks include an L-shaped base for resting on a table or counter top with an upper lateral flange containing a series of top openings for receiving the lower body portion of a plurality of separate pipettes.
Typical of the wall mounted racks are the "Magnetic Pipet Holder" model 290-009 and the "Wall Mounted Pipettor Rack" models 01-024 and 01-026 of PCG Scientifics. The Magnetic Pipet Holder includes a flat support plate carrying a magnetic strip for attaching the support plate to a metal surface. Upper and lower arms extend forward from the support plate and include vertically aligned notches for receiving and supporting a plurality of pipettes. The Wall Mounted Pipettor Racks comprise metal racks with round or key-hole shaped top opening holes for vertically receiving and supporting a number of pipettes. The racks are attachable to a wall by screws or by double-sided tape.
Typical of a pipette carousel stand are the "Carousel Pipettor Stand" model 02-254 of PCG Scientifics and the "Carousel Pack GR-2" of the Rainin Instrument Co., Inc., the assignee of the present invention. Such pipette carousels include a base and a vertically extending pole with outwardly extending arms with notches for receiving and supporting a plurality of pipettes.
Each of the foregoing stands, racks and carousel pipette holders serve a useful purpose. However, they are each specially designed for a particular support setting. The stands, racks and carousel structures are designed for supporting a plurality of pipettes on a counter or a tabletop. The Magnetic Pipette Holder is specifically designed for attachment to a metal surface. The wall mounted racks are designed for attachment to a flat wall surface by screws or tape.
None of the foregoing pipette holders is designed to provide multiple modes of attachment to the various support surfaces commonly found in clinical and diagnostic laboratories such as the vertical front surfaces of refrigerators, cabinet doors, or fume hoods (all of which are usually formed of a ferrous material), or hollow walls or wooden surfaces or the edges of shelves or table tops. Accordingly, there is a need for a simple and inexpensive support apparatus for securely and safely supporting pipettes when not in use and which possesses the ability of attachment in a number of different manners to a number of different surfaces typically found in laboratory settings. The present invention satisfies such needs.

Hanger for supporting a plurality of pipettes adjacent a horizontal support surface


A CCD shift register which is improved in the transfer efficiency with a minimal decrease in the amount of electric charge that can be handled. The CCD shift register has an array of transfer electrodes, each comprising a pair of storage and transfer gate electrodes, which are formed on a semiconductor substrate through a gate insulator. A semiconductor region under each storage gate electrode is divided into a plurality of subregions by using impurities.

1. Field of the Invention
The present invention relates to a CCD shift register which may be applied to CCD solid-state imaging devices or other similar devices.
2. Description of the Prior Art
FIG. 1 shows one example of frame interline (FIT) type CCD solid-state imaging devices. In the figure, reference numeral 1 denotes an imaging section, 2 denotes a storage section, and 3 denotes an output section, that is, a horizontal shift register section having a CCD structure. The imaging section 1 has a multiplicity of light-receiving elements 4 arranged in a matrix to form pixels, and vertical shift registers 5 having a CCD structure, each disposed at one side of a vertical line of the light-receiving elements 4 to transfer signal charges from the light-receiving elements 4 to the storage section 2 in the vertical direction. The storage section 2 is disposed at the lower side of the imaging section 1, as viewed in the vertical direction, for temporary storage of signal charges generated in the imaging section 1. The storage section 2 has a plurality of vertical shift registers 6 similarly having a CCD structure which are disposed in one-to-one correspondence to the vertical shift registers 5 in the imaging section 1. Both the vertical shift registers 5 and 6 in the imaging and storage sections 1 and 2 employ a four-phase driving system, for example, in which the shift registers 5 and 6 are controlled with four-phase driving pulses .phi.IM.sub.1, .phi.IM.sub.2, .phi.IM.sub.3 and .phi.IM.sub.4 and .phi.ST.sub.1, .phi.ST.sub.2, .phi.ST.sub.3 and .phi.ST.sub.4, respectively. The horizontal shift register section 3, serving as an output section, employs a two-phase driving system, for example, in which it is controlled with two-phase driving pulses .phi.H.sub.1 and .phi.H.sub.2.
In solid-state imaging devices for high-definition (high-resolution) imaging applications, the horizontal shift register section 3 has a multichannel horizontal shift register structure in which two horizontal shift registers 3A and 3B, for example, are provided in parallel, as shown in FIGS. 1 and 2, in order to lower the horizontal transfer frequency. In this horizontal shift register section 3, a pair of first and second horizontal shift registers 3A and 3B are disposed with a transfer gate section 7 interposed therebetween, as shown in FIG. 2. A plurality of transfer electrodes 9, each comprising a storage gate electrode 9S and a transfer gate electrode 9T, are formed in common to the horizontal shift registers 3A and 3B. The driving pulse .phi.H.sub.1 is applied to a first group of alternate transfer electrodes 9, and the driving pulse .phi.H.sub.2 is applied to a second group of remaining alternate transfer electrodes 9. Hatched regions 10 are channel stop regions.
In this FIT type solid-state imaging device, fast sweep-away transfer is carried out during the vertical blanking period to sweep away the smear component from the vertical shift registers 5 and 6 in the imaging and storage sections 1 and 2 to a smear drain region 12 through a smear gate section 11. Thereafter, the signal charges in the light-receiving elements 4 are transferred from the imaging section 1 to the storage section 2 where the charges are stored temporarily. Then, for each horizontal blanking period, the storage section 2 distributes signal charges for one horizontal line to the first and second horizontal shift registers 3A and 3B in the horizontal shift register section 3. For example, signal charges e.sub.1 in alternate vertical shift registers 6 are transferred to the first horizontal shift register 3A, while signal charges e.sub.2 in the remaining alternate vertical shift registers 6 are transferred to the second horizontal shift register 3B. The signal charges e.sub.1 and e.sub.2 for one horizontal line transferred to the first and second horizontal shift registers 3A and 3B are transferred therethrough horizontally and then sequentially output alternately through switching means (not shown).
Conventional two-phase CCD shift registers which are applicable to the above-described horizontal shift registers 3A and 3B are shown in FIGS. 3 and 4.
A two-phase CCD shift register 14 shown at A in FIG. 3 has a semiconductor substrate 15 of a first conductivity type, e.g., p-type, a gate insulator 16 formed on one surface of the semiconductor substrate 15, a plurality of transfer electrodes 17 arranged in one direction, each transfer electrode 17 comprising a storage gate electrode 17S formed from a first-level polycrystalline silicon layer and a transfer gate electrode 17T formed from a second-level polycrystalline silicon layer, and p-type impurity regions 18 formed only under the transfer gate electrodes 17T with a higher impurity concentration than that of the substrate 15, thereby forming transfer sections, i.e., a first storage section st.sub.1, a second transfer section tr.sub.1, a second storage section st.sub.2 and a second transfer section tr.sub.2. The first storage and transfer gate electrodes 17S and 17T are connected to a bus line to which a driving pulse .phi..sub.1 is applied, and the second storage and transfer gate electrodes 17S and 17T are connected to a bus line to which a driving pulse .phi..sub.2 is applied.
In the two-phase CCD shift register 14, when charge transfer is to be effected, e.g., when the pulse .phi..sub.1 is at the high level, while the pulse .phi..sub.2 is at the low level, a staircase surface potential profile 20 such as that shown at B in FIG. 3 is formed owing to a potential difference based on the impurity concentration difference between the storage sections st.sub.1 and st.sub.2 on the one hand and the transfer sections tr.sub.1 and tr.sub.2 on the other, so that electric charge e is transferred in the direction of the arrow a. In this arrangement, a region A corresponding to the storage section st.sub.1 is a region which is capable of storing the signal charge e, as shown at B in FIG. 3.
A two-phase CCD shift register 21 shown at A in FIG. 4 is arranged so as to increase the transfer efficiency and the driving frequency. In this prior art, the p-type impurity regions corresponding to the transfer sections tr.sub.1 and tr.sub.2 are each divided into two subregions by varying the impurity concentration. More specifically, a first p-type impurity region 22 of relatively low impurity concentration and a second p-type impurity region 23 of relatively high impurity concentration are formed under each transfer gate electrode 17T, and the gate length l.sub.2 of the storage sections st.sub.1 and st.sub.2 is made shorter than the gate length l.sub.1 of those shown at A in FIG. 3, thereby allowing the transfer electric field to be readily applied. The other arrangements are the same as those shown in FIG. 3. Accordingly, a three-step potential profile is formed under each transfer section, as shown by a staircase surface potential profile 24 at B in FIG. 4, thus increasing the transfer efficiency and the driving frequency.
With the arrangement shown in FIG. 4, since the charge storing capacity of the region B decreases owing to the impurities, the region A mainly serves as a region capable of storing the electric charge e. More specifically, gate electrodes 26 and 27 are usually isolated from each other by an insulator 28, as shown at A in FIG. 5, and hence the insulator 28 is locally thick. When the driving pulses .phi..sub.1 =.phi..sub.2, the potential is modulated at the region between the gate electrodes 26 and 27, resulting in a potential barrier (or a potential dip) 29, which interferes with the charge transfer, as shown by a potential diagram at B in FIG. 5. Accordingly, it is necessary to establish either an electric potential difference between the gate electrodes 26 and 27 or an impurity concentration difference between the respective regions under the gate electrodes 26 and 27 and hence a surface potential difference so that the potential barrier (or the potential dip) 29 is canceled (see the potential diagram shown at C in FIG. 5). It should be noted that the chain line 30 represents a state where the barrier 29 is not canceled because the potential difference is not adequately large, whereas the solid line 31 represents a state where the barrier 29 is satisfactorily canceled with an adequate potential difference.
For this reason, the two-phase CCD shift register 21 shown in FIG. 4 necessitates raising the impurity concentration in each p-type impurity region 22 to thereby increase the potential difference between the regions A and B in order to cancel the potential barrier (or the potential dip) in addition to the purpose of obtaining a necessary transfer electride field (in this case the potential difference between the regions B and C decreases conversely). Accordingly, the region A mainly serves as a region capable of storing electric charge, as has been described above, and the amount of electric charge that can be handled decreases by an amount corresponding to the reduction in the gate length l.sub.2 in comparison to the arrangement shown in FIG. 3. In general, however, the decrease in the amount of electric charge can be compensated for by increasing the gate width of the CCD shift register.
In a case where the two-phase CCD shift register 21 shown in FIG. 4 is employed for the multichannel horizontal shift register section 3 of the solid-state imaging device shown in FIG. 1, if the gate width is increased in order to ensure the amount of electric charge that can be handled, the efficiency of charge transfer in the vertical direction (i.e., the direction V) from the first horizontal shift register 3A to the second horizontal shift register 3B lowers. An effective way of improving the vertical transfer efficiency is to shorten the vertical gate length (and hence the gate width of the horizontal shift register). As a consequence, the prior art suffers from the problems that if the two-phase CCD shift register 14 shown in FIG. 3 is employed for the multichannel horizontal shift register section 3, the horizontal transfer efficiency lowers, whereas, if the two-phase CCD shift register 21 shown in FIG. 4 is employed therefor, the amount of electric charge that can be handled decreases.

CCD shift register having a plurality of storage regions and transfer regions therein


An aspect includes a hybrid gas turbine engine system of a hybrid electric aircraft. The hybrid gas turbine engine system includes a gas turbine engine having a low speed spool and a high speed spool, a generator operably coupled to the low speed spool, a high spool electric motor operably coupled to the high speed spool, and a controller. The controller is configured to control the hybrid gas turbine engine system in a powered warm-up state to add heat to one or more components of the gas turbine engine by operating the gas turbine engine with a higher engine power setting above idle to drive rotation of the generator, transfer power from the generator to the high spool electric motor, and produce thrust. The gas turbine engine transitions from the powered warm-up state after reaching a target temperature of the one or more components in the powered warm-up state.

The subject matter disclosed herein generally relates to engine systems and, more particularly, to a method and apparatus for hybrid gas turbine engine system powered warm-up.
Conventional gas turbine engines are typically operated at an idle power level during taxiing from a gate to a runway and can remain at idle power for a substantial period of time before takeoff, which consumes a quantity of fuel prior to flight. Hybrid electric aircraft use electricity to provide a portion of the power needed for aircraft propulsion by converting electricity into a propulsive force. A hybrid electric aircraft can use an electric drive to support taxiing operations on a runway and prevent fuel burn typical of conventional gas turbine engine idling.

Hybrid gas turbine engine system powered warm-up


A Chinese to English Translation Tool (CETT) locates pre-existing translations of phrases and increases probabilities for a match by searching independently of proper nouns and Western characters. The CETT comprises an enhancement program, a search program and a display program. The CETT is adapted to access a translation database, a proper noun table, and an unmatched translation table. The translation database is a central repository of an accumulated body of translated literature. The proper noun database is a table showing the associations between English and Chinese for proper nouns.

Companies offering web services and service oriented architecture for international businesses include services that translate foreign languages into English. Automated translation tools can provide real time translation capabilities. Asian languages such as Chinese pose special problems for automated translation into English. For example, current automated techniques for translation from Chinese to English do not take into account the idiomatic differences between the languages. Consequently, automated literal translation of Chinese to English bears little resemblance to normal, everyday English, and proper translations can only be achieved by manual translation by Chinese language experts.
However, there exists a large body of Chinese literature translated into English. In the translation of the literature, idiom has been taken into account. There is a considerable wealth of such translated literature today. These translated literary works have already been expertly translated from Chinese to English and vice versa. Moreover, the technology exists to capture and store these translations digitally so that the data can be searched at high speed. For example, the availability of 64 bit computing and very large memories allow for efficient high speed searching of such captured translations. A variety of methods for converting Chinese literature to digital format are known to persons skilled in the art.
Chinese is a well structured language with specific character orders. Chinese has no spaces between words, but it does use commas to separate clauses and periods at the end of sentences. There are no spaces between words and each character carries a specific meaning. These characteristics of the Chinese language raise the possibility of using electronic files of translated literature for automated Chinese to English translation because the chance of finding a match in the wealth of literature is very high. In addition, the fact that Chinese has no real tenses, gender, cases or plurals, reduces the variety of different sentence structures considerably.
If such a search could be made, several problems exist. First, proper nouns may be translated in several ways. For example, in English our reference to the main Chinese river is "Yangtze". However, in Chinese this refers to a fish that lives at the mouth of the river. The actual river itself is called the "Chang Jiang," which literally means "long river." Another example is Peking versus Bei Jing, or the "Imperial Palace." Although Chinese has no capital letters, proper nouns can easily be identified from the structure of the sentence.
Second, modern Chinese in use today has many embedded western characters, such as numbers, names and web addresses. Embedded Western characters do not require translation, yet their positioning within the Chinese text to be translated can lessen the likelihood of a match. The accuracy of this translating can be improved if it is made independent of embedded Western characters. Modern Chinese uses many of these, particularly numbers.
What is needed is a tool or method to increase the accuracy of automated translation of Chinese into English by taking advantage of existing expert translations. In addition, need exists for further improvements in accuracy by addressing the impact of proper nouns and embedded Western characters to such a search.

International Business Machines Corporation


A thermally stable polyimide which is blocked at the polymer terminal with a dicarboxylic acid anhydride represented by the formula (III): ##STR1## wherein Z is a divalent radical selected from the group consisting of a monoaromatic radical which is substantially unsubstituted or substituted with a radical having no reactivity with amine or dicarboxylic acid anhydride and has from 6 to 15 carbon atoms, a condensed polyaromatic radical or a noncondensed aromatic radical connected each other with a direct bond or a bridge member, and has a fundamental skeleton represented by recurring structural units of the formula (IV): ##STR2## and a process of preparing the polyimide.

1. Field of the Invention
The present invention relates to polyimide resin and particularly relates to a thermally stable polyimide and a process of preparing the polyimide.
2. Description of the Related Art
Conventionally, polyimide obtained by reaction of tetracarboxylic acid dianhydride with diamine is excellent in mechanical strength and dimensional stability in addition to its high heat resistance and flame retardance and electrical insulation property.
Because of these preferred properties, polyimide has been used in various fields such as electric and electronic appliance, space and aeronautic equipment, and transfer machinery. Polyimide is hence expected to be widely used in fields where heat resistance is required.
For example, polyimide having a fundamental skeleton of the following formula (V): ##STR3## has been widely used in the form of films and molded materials due to its excellent heat resistance, and mechanical and electrical properties. However, polyimide has a disadvantage that continuous use of the polyimide for a long time at high temperature in air leads to gradual reduction of these excellent properties and impairs essential properties of the polyimide.
Attempts have also been carried out to improve properties by blocking the polymer terminal of the polyimide having the fundamental skeleton of the formula (V). For example, Japanese Laid-Open Patent HEI 2-60931 and 2-60932 (1990) have disclosed polyimide and its precursor polyamic acid which have been blocked with a monoamine such as p-fluoroaniline in order to improve alkali resistance. U.S. Pat. No. 4,742,153 has also disclosed blocking the polyamic acid precursor of the formula (V) with pyromellitic anhydride. The invention, however, relates to a process for carrying out, in a drying step after blocking, solvent removal, imidization and simultaneous high polymer formulation of polyimide by intermolecular reaction of terminal carboxylic acid and terminal amine. That is, the invention relates to thermosetting polyimide.
Consequently, it has not been investigated to block the polymer terminal of polyimide having the formula (V) with dicarboxylic acid anhydride in order to improve thermal stability at high temperature.

Thermally stable polyimide and preparation process of polyimide


A method and system are provided for the reception of a biorthogonally-modulated RF signal in a simplex free-space system. The method is found to substantially diminish the coupling between the error probabilities of those bits associated with signal selection and those bits associated with signal polarity in prior art biorthogonal receivers. By decoupling the error probabilities, the polarity information error rate is reduced and consequently the overall error rate of the receiver system is improved. The receiver system preserves soft-decision reliability information for the signal selection bits in a manner so that the information is insensitive to uncompensated amplitude fluctuations in the received signal. A symbol deinterleaver in the receiver is located between the correlators and the bit estimators. The method and system are applicable to all known biorthogonal signal sets but requires a specific interleaving structure in the corresponding transmitter system.

Robust mobile digital communication is difficult to accomplish because of the deleterious effects of the environment on the propagation characteristics of the radio-frequency (RF) channel (from about 0 to 100 Gigahertz). Atmospheric conditions and variations in terrain, foliage, and the distribution and density of man-made structures and operating machinery strongly affect the characteristics of the medium which RF signals traverse. Satellite communication systems are typically limited by insufficient electromagnetic radio-frequency (RF) field strength due to the presence of background cosmic, atmospheric, and man-made noise which together establish an irreducible noise floor. Satellite systems may also be limited by the thermal noise of the active electronic components used in the implementation of the receiver system. Terrestrial mobile communications systems are also subject to performance degradation due to the effects of background and thermal noise. However, terrestrial mobile receiver systems are further degraded by multipath propagation due to the reception of signals corresponding to paths other than the line-of-sight (LOS) propagation path.
A method which is used to combat the deleterious effects of multipath propagation, as well as interference from other sources, is "spread spectrum" modulation [reference: R. L. Pickholtz, D. L. Schilling, and L. B. Milstein, "Theory of spread-spectrum communications--a tutorial, IEEE Transactions on Communications, Vol. 30, No. 5, pp. 855-884, May 1982]. See also U.S. Pat. Nos. 5,063,560; 5,081,543; 5,235,614; and 5,081,645. In a spread spectrum communication system, the bandwidth occupied by the digital data message is expanded (spread) in the transmitter by multiplying the data message by a spreading signal or sequence which is unrelated to the data message. The spreading effect is collapsed in the receiver by the process of correlation. Multipath and other forms of interference which are frequency-selective then only perturb part of the spread signal. However, spread spectrum methods are no more effective than narrowband modulation methods, for example, the method of Orthogonal Frequency Division Multiplexing (OFDM) [reference: W. Y. Zou and Y. Wu, "COFM: an overview," IEEE Transactions on Broadcasting, Vol. 41, No. 1, pp. 1-8, March 1995], in combating the effects of wideband noise.
A disadvantage of spread spectrum modulation is that it typically has relatively poor spectrum efficiency (i.e. less than one bit per Hertz per information symbol). The ability of the spread signal to combat narrowband forms of interference is determined by the "processing gain" of the spread spectrum system, which is essentially the ratio of the spread signal bandwidth to that of the original data message. Therefore, in order to ensure sufficient processing gain, the data message bandwidth is a small fraction of the spread signal bandwidth (e.g. less than about one-tenth). As a result, it may be difficult to use single-carrier spread spectrum modulation in circumstances where the available bandwidth is strictly limited and the necessary data throughput requires high spectrum efficiency.
A spread spectrum communication system requires the use of at least one transmitted spreading signal. It is possible to increase the throughput of a spread spectrum communication system through the consideration of additional spreading signals in the determination of the transmitted signal. Two prior art methods for increasing the information efficiency of a spread spectrum communication system are "m-ary orthogonal" modulation and "m-ary biorthogonal" modulation [reference: W. C. Lindsey and M. K. Simon, Telecommunication Systems Engineering. Englewood Cliffs, N.J.: Prentice-Hall, Inc., 1973, p. 198, pp. 210-225]. Both of these methods differ from methods of simultaneously multiplexing spreading signals, in which a plurality of spread signals are simultaneously transmitted in a single information symbol or baud. In m-ary orthogonal modulation, only one signal selected from a plurality of spreading signals is transmitted for the duration of the baud. If there are a number, denoted M, of signals in the signal set, then with m-ary orthogonal modulation, each information baud conveys log.sub.2 M information bits because the determination as to which one of the M signals (i.e. a 1-of-M choice) can be represented with log.sub.2 M decision bits, hereafter known as the "selection" bits.
Prior art m-ary biorthogonal modulation is similar to m-ary orthogonal modulation except that the polarity of the transmitted signal is also modulated by some bits in the data message. While the m-ary orthogonal system conveys log.sub.2 M information bits in a baud through the selection of the transmitted signal, the m-ary biorthogonal system conveys (log.sub.2 M)+1 information bits through the selection of the transmitted signal (log.sub.2 M bits as before) and determination of the transmitted signal polarity by one bit per information symbol, hereafter known as the "polarity bit". The specific prior art embodiment of biorthogonal modulation where M is one (1) corresponds to zero (0) selection bits and only one polarity bit. This is equivalent to prior art "antipodal" (also known as bipolar, biphase, BPSK) modulation where the signal is always transmitted and only the polarity of said signal is modulated. The embodiment with M equal to one (1) of biorthogonal modulation is a degenerate example and is not considered further. In the instant application, the minimum valid number M of signals for m-ary orthogonal or m-ary biorthogonal modulation is two (2). Furthermore, only binary powers of two are considered for values of M (i.e. two, four, eight, sixteen, and so forth). The methods of m-ary orthogonal and m-ary biorthogonal modulation are unrelated to the modulation method of "m-ary amplitude" modulation, where one of a plurality of amplitude scale factors are applied to a spread signal in order to increase the information density.
In order to implement a receiver for transmitted m-ary orthogonal or biorthogonal signals, the signals in the signal set are identically embodied in both the transmitter and the receiver system and are pairwise distinct for all M signals. Furthermore, the signals must all be pairwise orthogonal or approximately orthogonal (AO). Mutually orthogonal or orthogonal signals are defined as having a crosscorrelation value (sum) of about zero, when properly synchronized. The crosscorrelation value of two signals is proportional to the sum or equivalently the integration of the pairwise product of the two signals (i.e. the inner product). AO signals are distinguished from orthogonal signals by having the characteristic of a small-magnitude crosscorrelation value, which is non-zero. Either orthogonal or AO signals may be used with either m-ary orthogonal or m-ary biorthogonal modulation. As the value of M is increased, the maximum permitted crosscorrelation value among signal pairs for proper operation of the receiver system is decreased. Whereas perfectly orthogonal signals do not interfere with one another in a correlation receiver, when AO signals are implemented, error correcting codes are typically required in order to reduce the irreducible error level (known as the code noise or self-noise) due to mutual crosscorrelation interference between the non-orthogonal AO signals.
Spreading signals can be determined by various methods. Some spreading signals or sequences which have been implemented in prior art spread spectrum systems include, but are not limited to, distinct m-sequences, which are also known as pseudonoise (PN) sequences or maximal-length sequences [reference: D. V. Sarwate and M. B. Pursley, "Crosscorrelation properties of pseudorandom sequences," Proceedings of the IEEE, Vol. 68, No. 5, pp. 593-619, May 1980], distinct phases of a single m-sequence [reference: S. L. Miller, "An efficient channel coding scheme for direct sequence CDMA systems," Proceedings of MILCOM '91, pp. 1249-1253, 1991], orthogonal m-sequences with bit stuffing as disclosed by Gutleber in U.S. Pat. No. 4,460,992, issued Jul. 17, 1984, Gold codes, Kasami codes, Hadamard codes, and Bent codes [reference: S. Tachikawa, "Recent spreading codes for spread spectrum communications systems," (translated) Electronics and Communications in Japan, Part 1, Vol. 75, No. 6, pp. 41-49, June 1992]. It is also possible to determine spreading sequences which occupy the available bandwidth and which are orthogonal using matrix eigenvector methods, such as disclosed in U.S. Pat. No. 4,403,331 by P. H. Halpern and P. E. Mallory, or by using orthonormal wavelets as disclosed by Resnikoff, et al. in U.S. Pat. No. 5,081,645. Some of the previously disclosed methods use orthogonal or approximately orthogonal spread spectrum signals together with antipodal data modulation in a simultaneously multiplexed system. The orthogonal spreading sequences determined for such methods (sometimes referred to as "multiple access" methods) may also be suitable for m-ary orthogonal or m-ary biorthogonal modulation, where only one of a plurality of signals is transmitted (i.e. non-multiplexed). The above-listed spreading sequences or signals may be used in this invention, as discussed hereinafter.
Error correction digital codes (ECC) are an important method of combating the effects of noise, outside interference, and self-noise in spread spectrum communication systems, regardless of the modulation method. An ECC method consists of a system (i.e. an algorithm and its implementation) of encoding an original data message through the incorporation of redundancy information together with a system for decoding the encoded message so as to recover the original message. The bit rate throughput required to represent the encoded message is larger than that of the message itself. The average ratio of the original message length to the encoded message length is known as the "code rate". As the amount of redundancy is increased, the code rate decreases, which diminishes the decoded message throughput of the system. However, the probability of error in determining the message at the receiver, after ECC decoding, is typically diminished by a much greater amount, so that the loss of some throughput in order to dramatically improve the reliability of the communication system is acceptable. A measure of the improvement is the "coding gain" of the ECC. The coding gain may be interpreted as the effective increase in the signal-to-noise ratio (SNR) brought about by the use of the ECC in a equivalent system without ECC or with reference to some other ECC method. ECC methods in which the amount of ECC applied to message bits is approximately equal for all bits are known as equal error protection (EEP) ECC methods. ECC methods in which the amount of ECC redundancy varies according to the function or purpose of the bit are known as unequal error protection (UEP) ECC methods. UEP ECC methods are typically used when the relative importance of the transmitted bits varies considerably. The conditions for continuous mobile reception may be so difficult that practical code rates are usually about one-half or less, which is equivalent to one hundred (100) percent redundancy.
A block diagram of a prior art receiver system for a biorthogonally modulated signal is shown in FIG. 1. The received RF signal is first amplified and filtered with a bandpass filter in tuner 1 to remove interference and noise which is outside of the bandwidth of the spread spectrum signal. The RF signal is typically frequency-translated to a lower frequency, known as the intermediate frequency (IF), for further processing, which simplifies the implementation. In some digital systems, the IF frequency is zero, which requires that the remaining processes be implemented with complex digital arithmetic (i.e. real and imaginary components). The tuner also accomplishes the function of automatic gain control (AGC) so that the received signal energy is made to be approximately constant for subsequent processing. The combined RF functions are abbreviated as RF tuner 1.
In many conventional FIG. 1 systems, the signal is converted (quantized) from an analog representation (i.e. voltage or current) to a digital representation by analog-to-digital converter (ADC) 3. The number of bits in the implementation of the ADC is chosen to preserve sufficient dynamic range in the digitized signal so that the irreducible error level caused by ADC quantization noise does not significantly degrade the receiver system performance. Typically, the number of bits is between six (6) and twelve (12), inclusive. The received and quantized signal is made synchronous with the transmitter in baud frequency and carrier frequency by baud clock recovery 5 and carrier frequency recovery 7 subsystems. Typically, these functional blocks are implemented with early/late or pulse-swallowing algorithms, phase-lock loops (PLLs) and voltage-controlled oscillators (VCOs), and/or frequency-lock loops (FLLs). The overall function of recoveries 5 and 7 is to eliminate frequency offsets caused by variation in components and the effect of Doppler frequency shift. The synchronization also establishes the proper timing (i.e. phase relationship) so that the magnitude of the crosscorrelation values between the possible signals in the biorthogonal signal set are at a minimum at an instance in the baud interval known as the "sampling point".
The digitized and synchronized signal is optionally equalized by adaptive equalizer 9 in order to partially correct for the effects of RF signal dispersion caused by the frequency-selective characteristics of the RF propagation channel. The equalizer also mitigates phase and amplitude errors caused by implementation loss due to analog components in the transmitter and receiver systems. Equalizer 9 may be implemented with a finite-impulse response (FIR) transversal filter or an infinite-impulse response (IIR) recursive filter or a combination thereof. The coefficients of the equalization filter are determined by a tap-weight update algorithm and are updated at a rate sufficient to reasonably track changes in the RF propagation characteristics. Some prior art methods for equalization include, but are not limited to, minimum mean square estimation (MMSE), least mean square (LMS), and recursive least square (RLS) algorithms, all of which are known.
The equalized signal is propagated to biorthogonal demodulator 11. The function of biorthogonal demodulator 11 is to determine estimates of the log.sub.2 M encoded selection bits and one encoded polarity bit in each information baud. The biorthogonal transmitter system (not shown) transmits one of the plurality of M signals from the biorthogonal signal set in each baud. The bit representation of the index of the biorthogonal signal (i.e. 1-of-M choice) is represented by the encoded selection bits. The polarity of the resulting signal is modulated in the transmitter system by multiplying the determined signal by a factor of positive unity (one) or negative unity, arbitrarily corresponding to an encoded polarity bit value of zero or one, respectively. After demodulation, the estimated section and polarity bits are organized as a serial bit sequence and processed by deinterleaver 13. The function of deinterleaver 13 is to reverse the effect of the shuffling process of the corresponding interleaver, if present, in the transmitter system (not shown). Interleaving is a method of time-diversity in which groups consisting of one or a plurality of bits are re-arranged by way of a shuffling algorithm which is a bijection. The incorporation of an interleaver in the transmitter system and consequently a deinterleaver in the receiver system is optional, but they are frequently used in mobile communication systems to combat burst errors. The deinterleaved sequence of bits is substantially similar to that after ECC encoding in the transmitter system, except for the occurrence of errors.
After deinterleaving 13, the selection and polarity bit estimates are decoded by error correction code decoder 15 in order to recover the original data message. The ECC decoder reverses the effects of the ECC encoder (not shown) in the transmitter system according to the error correction code, substantially reducing the number of bit errors after decoding. The size of the decoded message is made smaller than the message prior to decoding by an amount corresponding to the code rate factor.
The system of ECC decoder 15 implements an algorithm which is essentially the reverse of the ECC encoder in the transmitter (not shown). In the transmitter system, ECC encoding is performed after bit scrambling and prior to interleaving and biorthogonal modulation. The optimum ECC decoder is related to the optimum ECC encoder. The optimum encoder for a specific communication system depends on the expected characteristics of the error distribution at the receiver, which requires knowledge of the specific modulation method and the expected RF channel impairment. Typically, convolutional codes or block codes, or a combination thereof, are implemented in the encoder in the transmitter and corresponding ECC decoder in the receiver. Convolutional codes with soft-decision Viterbi decoding in the receiver have been found to be optimum in circumstances where the error distribution is substantially random, uncorrelated, and resembles a Gaussian function. However, in bursty environments, block codes may be preferable (e.g. binary BCH codes, Reed-Solomom codes, concatenated parity-Reed-Solomon codes, and/or quadratic residue codes). In the ECC encoder in the transmitter (not shown), the length of the scrambled source message is increased by a factor which is the reciprocal of the code rate.
After ECC decoding 15, the decoded message is propagated to descrambler 17. Scrambling in the transmitter and consequently descrambling 17 in the receiver is optional. Scrambling is typically used to eliminate long runs of consecutive binary ones and zeroes and to cause approximately equal probabilities for the occurrence of the binary digits zero and one in the transmitted source message. Scrambling may be accomplished in the transmitter by multiplying the source message bit sequence by a binary polynomial, for example a PN-sequence, using binary arithmetic. Similarly, in order to descramble the decoded bit sequence message, the decoded and synchronized bit sequence is multiplied by the same binary polynomial. The distribution of the sequence generated by the binary scrambling polynomial is approximately uniform and has a very long period. After descrambling 17, the estimated message 19 substantially resembles the source data message that was transmitted, except for the occurrence of errors.
FIG. 2 is a block diagram of the prior art biorthogonal demodulator 11 in FIG. 1 where M is two (2). The equalized signal 21 is applied to a plurality of correlators 23 and 25. Each correlator determines the inner product between the input signal 21 and a spreading signal, when properly synchronized about the sampling point. Correlators are typically implemented with integrators and sample-hold circuits in analog embodiments and multiply-accumulate (MAC) circuits in digital embodiments. The dynamic range (analog embodiment) or bit width (digital embodiment) is determined so that the irreducible error level due to the implementation does not significantly degrade the receiver performance. The spreading signals which are input to the correlators 23 and 25 are generated by signal generators 27 and 29, respectively, resulting in correlation sums 31 and 33. In general, for a biorthogonal signal set with M orthogonal or AO signals, M correlators and M signal generators are required in the receiver in order to generate M correlation sums. The signal generators are identical to those in the corresponding transmitter system where the result of only one of the signal generators is transmitted, together with polarity modulation, in each baud. For the specific embodiment where M=2, the received signal corresponds to one of two possible transmitted orthogonal or AO signals in each baud, together with polarity modulation. The correlation sums 31 and 33 are propagated to the absolute-value functions 35 and 37, respectively, which remove the polarity information from the correlation sums by discarding sign-bit information. The correlation sums are also propagated to limiters 39 and 41. The limiters (also known as 1-bit quantizer or hard-limiters) operate in a manner opposite to that of the absolute-value functions. While the absolute-value functions remove polarity information and preserve magnitude information in order to determine the selection bits, the limiters remove magnitude information and preserve only polarity information in order to determine the polarity bits.
The correlation sum magnitudes are compared by the comparison operator 43 in order to determine which of the two correlation sum magnitudes is the largest. The signal in the biorthogonal signal set corresponding to the largest correlation sum magnitude is determined to be the signal which is most likely to have been transmitted in the baud interval. The comparison operation is typically implemented with a comparator circuit. The comparator results in a binary digit which corresponds to the index of the signal which was estimated to have been transmitted. For example, in the biorthogonal transmitter system, if an encoded selection bit value of one results in the transmission of the signal corresponding to signal generator 27 in the receiver, then the comparator 43 is implemented so that it results in the binary digit one when the magnitude of the correlation sum 31 is greater than the magnitude of the correlation sum 33. Correspondingly, if an encoded selection bit value of zero in the transmitter results in the transmission of the signal corresponding to signal generator 29 in the receiver, then the comparator 43 is implemented so that it results in the binary digit zero when the magnitude of the correlation sum 33 is greater than the magnitude of the correlation sum 31. If the result of the comparison operator is ambiguous (i.e. equal correlation sum magnitudes), then the value of the comparison operator is determined randomly. The result 45 of the comparison operator 43 is the selection bit estimate.
The selection bit estimate is the control input for polarity bit multiplexor 47. Multiplexor 47 propagates only one of the plurality of limiter values, which are the multiplexor data inputs, according to the selection bit. For all representations of a multiplexor in the block diagrams in this disclosure, the control input is distinguished from the data inputs by the label "S". For polarity bit multiplexor 47, the control bits are the selection bits. Multiplexor 47 is implemented so that the limiter value corresponding to the correlation sum magnitude with the largest value is propagated. In the previous example, multiplexor 47 propagates the result of limiter 39 if the result of comparison operator 43 is binary value one. Multiplexor 47 propagates the result of limiter 41 if the result of comparison operator 43 is binary value zero. The result 49 of the multiplexor is the polarity bit estimate. The selection bit 45 and polarity bit estimates 49 are together propagated beyond biorthogonal demodulator 11 to deinterleaver 13 shown in FIG. 1.
In the general case where M is greater than two, comparison operator 43 is replaced with a "maximum-value" function. The maximum-value function results in the index value of the signal whose correlation sum magnitude is the largest. The index value is represented by log.sub.2 M selection bits (i.e. a 1-of-M choice). Index values are organized so that the bit sequences in the transmitter and those recovered by the receiver system are identical in the absence of demodulation errors (i.e. for a particular index, the same signal is selected in the biorthogonal modulator as is determined in the biorthogonal demodulator). In general, there are a plurality of M signal generators, M correlators, and M absolute-value functions in the biorthogonal demodulator for a biorthogonal signal set with M possible orthogonal or AO signals. In general, polarity multiplexor 47 selects from one of M inputs according to log.sub.2 M estimated selection bits. The polarity of the correlation sum with the largest magnitude is propagated as the demodulated polarity bit. The result of the demodulation of a single information baud is a sequence of (log.sub.2 M)+1 bits.
A disadvantage of the prior art biorthogonal demodulator in FIG. 2 is that soft-decision reliability information for the selection bit estimate, beyond the minimum log.sub.2 M selection bits which represent the signal choice, is not useful in the determination of which polarity bit estimate is selected by the polarity bit multiplexor 47. The selection and polarity bit estimates are both determined in the prior art biorthogonal demodulator 11, while improvements due to soft-decision reliability information are typically not produced until after ECC decoding 15. Thus, even if the selection bit error rate is eventually reduced by ECC decoding, the selection bit error rate prior to ECC decoding, in the biorthogonal demodulator 11, may be high (e.g. 1.times.10.sup.-3 to 1.times.10.sup.-2), which results in a degradation in the polarity bit error rate performance.
The probability of error in determining the selection bit in prior art FIGS. 1-2, P.sup.S.sub.b, prior to error decoding, for the specific embodiment where M=2 and the RF channel impairment is additive white Gaussian noise is known to be [reference: G. R. Cooper and C. D. McGillem, Modern Communications and Spread Spectrum. New York: McGraw-Hill, Inc., 1986, pp. 233-234]: ##EQU1## where E.sub.S is the energy in any one of the signals in the biorthogonal signal set, which are assumed to be equiprobable and equal in energy, N.sub.b0 is the noise spectral density, and Q is the complementary cumulative distribution function for the Gaussian probability density function (a.k.a. the Marcum Q function) [reference: G. R. Cooper, et al., ibid., pp. 423-425]. The noise spectral density is determined by measuring only the noise power through the bandwidth of the received signal. In the general case where M is greater than two, the error event probability for the selection bits considered together as the selection word, P.sup.S, is given only approximately by the union bound: ##EQU2## For the specific embodiment where M=2, the selection word error rate and the selection bit error rate are equal. However, in general, for M greater than two, the selection word error event probability, P.sup.S, and the selection bit error probability, P.sup.S.sub.b, are related but unequal. In some literature references to biorthogonal modulation (e.g. G. R. Cooper, et al., ibid. pp. 233-234), the term "N" or "M" refers to the total size of the signal set, including the orthogonal or AO signals with both positive and negative polarities. In this disclosure, the term "M" refers only to the number of distinct orthogonal signals (i.e. the dimension of signal set and N=2M) since the polarity of the signal is determined by the modulation and not the signal design.
An exemplary distribution of the correlation sum values determined in the prior art biorthogonal demodulator 11 is shown in FIG. 3 with one of two possible orthogonal signals transmitted in the presence of additive white Gaussian noise. Since the signals are assumed to be equiprobable, it is sufficient to consider the possible correlation sums in the event of the transmission of only one of the signals for all baud intervals and with a fixed arbitrarily positive polarity. The abscissa values are normalized by a factor equal to the signal energy, E.sub.S in Equation (1). In the absence of noise, there would be two point masses, centered at zero (0) and positive one (+1), respectively. The presence of noise smears the point masses out into continuous distributions. Distribution region 51 illustrates the correlation between the received signal and the orthogonal signal which corresponds to the correct selection bit. Distribution region 53 represents the correlation between the received signal and the other orthogonal signal (i.e. corresponding to the incorrect determination of the selection bit). The mean abscissa value of region 53 is about zero because the two signals in the biorthogonal signal set are presumed to be orthogonal. Since the detection method for biorthogonal modulation is a comparison of correlation sum magnitudes, errors occur in overlapping region 55.
However, biorthogonal system errors also occur in regions 57 and 59 which correspond to the images of the overlapping region 55 for the opposite polarity. These additional error events are caused because the correlation sum polarity information cannot be used to determine which of the two signals was most likely to have been sent. Instead, the polarity of the correlation sum is used to determine the remaining information bit, the polarity bit. The mean abscissa values of regions 51 and 53 are separated by Euclidean distance 61 which is a distance equivalent to the signal energy when the normalization is taken into consideration.
The probability of error in determining the polarity bits in the biorthogonal receiver system of prior art FIGS. 1-2 is more complicated and depends upon the probability of error for the selection bits. FIG. 5 is a graph which illustrates an exemplary distribution of the correlation sums in the determination of the polarity bits for the biorthogonal receiver when the impairment is AWGN and the selection bit is correctly demodulated. The abscissa values are normalized as in prior art FIG. 3. Region 63 corresponds to the distribution for a positive polarity modulation (arbitrarily, binary value zero) of the transmitted signal and region 65 corresponds to the distribution for a negative polarity modulation (arbitrarily, binary value one) of the transmitted signal. If the selection bit is correctly identified, which occurs with probability (1-P.sup.S.sub.b), the conditional polarity bit error probability, P.sup.P.vertline.S.sub.b, is: ##EQU3## The Marcum Q function decreases monotonically as its argument is increased. Since the numerator of the radical in Equation (3) is twice that in Equation (1), the conditional error probability in determining the polarity bits is significantly smaller than that of the selection bits in Equation (1), if the selection bit is correctly identified. The argument of the Q function is typically considered as a measure of the receiver signal-to-noise ratio (SNR). The conditional polarity bit error probability given by Equation (3) has more than a three decibel advantage (i.e. factor of two in SNR), when compared to the selection bit error probability given by Equation (1). A precise determination of the SNR advantage requires inversion of Equation (1), which has no closed-form solution and can only be determined numerically. The improvement in SNR is evident by the larger Euclidean distance 67 between the possible distribution regions 63 and 65 shown in FIG. 4.
However, when the selection bits are incorrectly identified in the biorthogonal demodulator, which happens with probability P.sup.s.sub.b the conditional probability of error for the polarity bits is one-half (1/2) because the polarity of the correlation sum corresponding to the incorrect biorthogonal signal is unrelated to the correlation sum of the correct signal. The polarity determination is then essentially random; in other words, a coin toss. The overall error probability for the polarity bits, P.sup.P.sub.b, is the weighted sum of the two polarity bit conditional error probabilities: ##EQU4##
By inspection of Equation (4), even when the conditional probability of error in the determination of the polarity bits is zero, which would require an infinite SNR, the polarity bit error is still non-zero and is substantially determined by the probability of error of the selection bits, P.sup.S.sub.b. Thus, the benefit of the SNR advantage in the conditional polarity bit error rate determination shown in Equation (3) is diminished by the selection bit error rate of Equation (1), which is typically substantially worse. This is a disadvantage to the prior art method of m-ary biorthogonal demodulation when compared to simultaneously multiplexed modulation and demodulation methods such as OFDM.
The prior art biorthogonal receiver system has been previously described and shown in FIGS. 1-2. U.S. Pat. No. 4,247,943 to Malm discloses the use of orthogonal or biorthogonal codewords in a receiver system for a signal that is generated by frequency-shift keying together with local phase-shift-keying. The receiver makes use of crosscorrelation sums determined between the received signal and four orthogonal signals, corresponding to the in-phase and quadrature component at two frequencies, to demodulate the signal.
Biorthogonal codewords [reference: W. C. Lindsey and M. K. Simon, Telecommunications Systems Engineering, ibid., pp. 188-194] are sets of binary sequences (i.e. ones and zeroes) which are constructed so that the permissible codewords are either pairwise orthogonal or complementary to each other. Biorthogonal codes are typically used to phase-modulate a narrowband sinusoidal signal. Methods for demodulating a biorthogonal signal set and decoding a biorthogonal codeword are unrelated except superficially in that both methods require magnitude comparisons among a plurality of possible signals or codewords.
U.S. Pat. No. 4,730,344 to Saha discloses a four dimensional modulation method known as Quadrature--Quadrature Phase-Shift Keying (QQPSK). The system of the '344 patent makes use of the in-phase and quadrature components of a signal with further data shaping of each component according to two orthogonal pulse shapes. This is equivalent to simultaneous multiplexing of antipodal-modulated orthogonal signals and not biorthogonally modulated signals. The reception of antipodal signals is unrelated to biorthogonal demodulation and the receiver of the instant invention.
U.S. Pat. No. 4,700,363 to Tomlinson et al discloses a method of m-ary phase modulation and unequal error protection. The system of '363 uses m-ary phase-shift keying together with the possible use of biorthogonal binary phase codewords. In general, m-ary phase modulation is different from m-ary biorthogonal modulation because the possible phase states in m-ary phase modulation are not pairwise orthogonal or approximately orthogonal. Hence, m-ary phase modulation is unrelated to the instant invention.
Accordingly, it is apparent from the above that there exists a need in the receiving art for: (i) limiting or diminishing the coupling between the selection and polarity bit error probabilities; (ii) preserving selection bit soft-decision reliability information for use in the determination of the polarity bits; (iii) reducing the polarity bit error rate; and (iv) reducing the overall bit error rate in biorthogonal demodulation.

RF simplex spread spectrum receiver and method with symbol deinterleaving prior to bit estimating


A method is provided for fabricating a battery using an anode preloaded with consumable metals. The method forms an ion-permeable membrane immersed in an electrolyte. A preloaded anode is immersed in the electrolyte, comprising MeaX, where X is a material such as carbon, metal capable of being alloyed with Me, intercalation oxides, electrochemically active organic compounds, and combinations of the above-listed materials. Me is a metal such as alkali metals, alkaline earth metals, and combinations of the above-listed metals. A cathode is also immersed in the electrolyte and separated from the preloaded anode by the ion-permeable membrane. The cathode comprises M1YM2Z(CN)N.MH2O. After a plurality of initial charge and discharge operations are preformed, an anode is formed comprising MebX overlying the current collector in a battery discharge state, where 0≦b<a.

1. Field of the Invention
This invention generally relates to electrochemical cells and, more particularly, to a method of fabrication and associated battery with an anode that is preloaded with consumable metals.
2. Description of the Related Art
The rechargeable lithium ion battery (LIB) has triggered the portable electronic devices revolution due to its high power density, long cycling life, and environmental compatibility. The rechargeable LIB consists of a cathode (positive electrode) and an anode (negative electrode), separated by a Li+-ion permeable membrane. A solution or polymer containing lithium-ions is also used in the battery so that Li+-ions can "rock" back and forth between the positive and negative electrode freely. The positive materials are typically transition-metal oxides such as lithium cobalt oxide (LiCoO2), lithium manganese oxide (LiMn2O4), lithium iron phosphate (LiFePO4), and their derivatives. Lithium-ions can move in their interstitial space freely and reversibly. The negative electrode materials can use lithium-metal, alloys, and carbonaceous materials. During discharge, Li+-ions are extracted from the negative electrode and inserted into the positive electrode. In the meantime, electrons pass through an external circuit from the negative electrode to the positive electrode and generate electric power. During a charge, ions and electrons move along the reverse direction and go back to their original places.
Although LIBs have been successfully used, the conflict between lithium demand and its scarcity surges its cost, which hinders the further application of lithium-ion batteries on a large scale. Therefore, a low-cost rechargeable battery is urgently needed as an alternative to expensive LIBs. Under the circumstance, sodium-ion batteries are attracting more attention because sodium has very similar properties to lithium, but a cheaper cost. Like lithium-ion batteries, sodium-ion batteries need Na+-host materials as their electrode. Much effort has been expended to directly duplicate the Li+-host structures, using Na+-host electrode materials for the sodium-ion batteries. For example, NaCoO2, NaMnO2, NaCrO2 and Na0.85Li0.17Ni0.21Mn0.64O2, all having a layered-structure similar to LiCoO2, have been developed for sodium-ion batteries. Similarly, Co3O4 with a Spinel structure, Na3V2(PO4)3 with a NASICON structure, and NaFePO4 with an Olivine structure have been employed in sodium batteries. In addition, sodium fluorophosphates, such as Na2PO4F, NaVPO4F and Na1.5VOPO4F0.5, have also used as the positive electrode in sodium batteries.
However, it is impractical to copy the structures of Li+-host compounds for Na+ or K+-host compounds. Sodium and potassium ions are much larger than lithium ions, and severely distort the structure of the Li*-host compounds. Thus, it is very important for the advancement of sodium/potassium-ion batteries to develop new Na+/K+-host materials with large interstitial spaces in which sodium/potassium-ions can easily and reversibly move. Na+/K+— ions have been observed to intercalate into metal cyanide compounds. Transition metal hexacyanoferrates (TMHCFs) with large interstitial spaces have been investigated as cathode materials for rechargeable lithium-ion batteries [1,2], sodium-ion batteries [3,4], and potassium-ion batteries [5]. With an aqueous electrolyte containing proper alkali-ions or ammonium-ions, copper and nickel hexacyanoferrates [(Cu,Ni)-HCFs]demonstrated robust cycling life with 83% capacity retention after 40,000 cycles at a charge/discharge current of 17C [6-8]. In spite of this, the materials demonstrated low capacities and energy densities because (1) only one sodium-ion can be inserted/extracted into/from per Cu-HCF or Ni-HCF formula, and (2) these TM-HCFs electrodes must be operated below 1.23 V due to the water electrochemical window. The electrochemical window of a substance is the voltage range between which the substance is neither oxidized nor reduced. This range is important for the efficiency of an electrode, and once out of this range, water becomes electrolyzed, spoiling the electrical energy intended for another electrochemical reaction.
In order to compensate for such shortcomings, manganese hexacyanoferrate (Mn-HCF) and iron hexacyanoferrate (Fe-HCF) have been used as cathode materials in a non-aqueous electrolyte [9, 10]. When assembled with a sodium-metal anode, Mn-HCF and Fe-HCF electrodes cycled between 2.0V and 4.2 V delivered capacities of ˜110 mAh/g.
FIG. 1 is a diagram depicting the crystal structure of a metal hexacyanometallate (MHCM) (prior art). The aforementioned TMHCFs can be categorized into a more general group of MHCMs that have the open framework as shown. In general, Berlin Green and its analogues can be expressed as M1YM2Z(CN)N, where M1m+ and M2n+ are transition metals with different or same formal oxidation numbers (m and n). M1 and M2 may be (Ti), vanadium (V), chromium (Cr), manganese (Mn), iron (Fe), cobalt (Co), nickel (Ni), copper (Cu), zinc (Zn), Ca, Mg, strontium (Sr), or barium (Ba). The ratio of M1 and M2 can vary. In the case of Berlin Green, M1 and M2 are both Fe. In addition, various amounts of water (H2O) can be stayed in their interstitial or lattice positions.
Without the interstitial ions, Berlin Green and its analogues can be used as cathode in any metal-ion batteries. The electrochemical reactions during discharge can be expressed as follows,
Cathode:M1x[M2(CN)z]y.dH2O+bMea++bae−=MebM1x[M2(CN)z]y.dH2O
Anode:Me−Ø=Mea++ae−+Ø.
Where Me is the alkali, alkaline earth, or aluminum, and Me−Ø is the metal loaded anode that has been described in parent application entitled, BATTERY ANODE WITH PRELOADED METALS, Ser. No. 14/198,663, which is incorporated herein by reference.
It would be advantageous if a metal-ion battery with a cathode of Berlin Green or its analogues could be made with a metal preloaded in the anode, for higher energy, longer cycling life, and lower costs.    [1] V. D. Neff, "Some Performance Characteristics of a Prussian Blue Battery", Journal of Electrochemical Society 1985, 132, 1382-1384.    [2] N. Imanishi, T. Morikawa, J. Kondo, Y. Takeda, O. Yamamoto, N. Kinugasa, and T. Yamagishi, "Lithium Intercalation Behavior into Iron Cyanide Complex as Positive Electrode of Lithium Secondary Battery", Journal of Power Sources 1999, 79, 215-219.    [3] Y. Lu, L. Wang, J. Cheng, and J. B. Goodenough, "Prussian Blue: a New Framework for Sodium Batteries", Chemistry Communications 2012, 48, 6544-6546.    [4] L. Wang, Y. Lu, J. Liu, M. Xu, J. Cheng, D. Zhang, and J. B. Goodenough, "A Superior Low-Cost Cathode for a Na-ion Battery", Angewandte Chemie International Edition 2013, 52, 1964-1967.    [5] A. Eftekhari, "Potassium Secondary Cell Based on Prussian Blue Cathode", Journal of Power Sources 2004, 126, 221-228.    [6] C. D. Wessells, R. A. Huggins, and Y. Cui, "Copper Hexacyanoferrate Battery Electrodes with Long Cycle Life and High Power", Nature Communications 2011, 2, Article number: 550.    [7] C. D. Wessells, S. V. Peddada, R. A. Huggins, and Y. Cui, "Nickel Hexacyanoferrate Nanoparticle Electrodes for Aqueous Sodium and Potassium Ion Batteries", Nano Letters 2011, 11, 5421-5425.    [8] C. D. Wessells, S. V. Peddada, M. T. McDowell, R. A. Huggins, and Y. Cui, "The Effect of Insertion Species on Nanostructured Open Framework Hexacyanoferrate Battery Electrodes", Journal of the Electrochemical Society 2012, 159, A98-A103.    [9] T. Matsuda, M. Takachi, and Y. Moritomo, "A Sodium Manganese Ferrocyanide Thin Film for Na-ion Batteries", Chemical Communications 2013, 49, 2750-2752.    [10] S—H. Yu, M. Shokouhimehr, T. Hyeon, and Y-E. Sung, "Iron Hexacyanoferrate Nanoparticles as Cathode Materials for Lithium and Sodium Rechargeable Batteries", ECS Electrochemistry Letters 2013, 2, A39-A41.

This patent is subject to a terminal disclaimer.

Battery with an anode preload with consumable metals


An method for evaluating sensitivity of a photoresist includes transferring an exposure dose monitor mark onto an inspection resist film with an inspection setting exposure dose using an exposure tool. Inspection sensitivity index varying according to the inspection setting exposure dose is measured, using an inspection transferred image of the exposure dose monitor mark delineated on the inspection resist film. An inspection photoresist sensitivity of the inspection resist film is calculated using sensitivity calibration data, based on the inspection sensitivity index.

1. Field of the Invention
The present invention relates to a method for evaluating sensitivity of a photoresist used in a photolithography process for manufacturing a semiconductor device, a method for preparation of a photoresist and a manufacturing method of a semiconductor device.
2. Description of the Related Art
In a photolithography process for manufacturing semiconductor device such as a semiconductor integrated circuit, a circuit pattern delineated on a photomask is transferred onto a resist film coated on a semiconductor substrate by an exposure tool. In accordance with recent higher integration and higher performance of a semiconductor device, it is required to transfer a finer circuit pattern delineated on a photomask to a semiconductor substrate. The imaging performance of a reduction projection exposure tool can be evaluated using an optical imaging theory derived from the Rayleigh equation. A resolution R of an exposure tool is proportional to a wavelength λ of the exposure light and inversely proportional to a numerical aperture NA. Furthermore, a depth of focus DOF is proportional to the exposure wavelength λ and inversely proportional to the square of the numerical aperture NA. Accordingly, in response to requirements for miniaturization of a semiconductor device, improvements in the process have been achieved, including shortening of the exposure wavelength λ and achieving a higher NA of a projection lens. However, along with the recent demands for further miniaturization of the semiconductor device, it has become extremely difficult to ensure an exposure latitude and a depth of focus DOF. Thus, the miniaturization of a semiconductor device significantly reduces the manufacturing yield.
To implement a photolithography process with a small process margin, great importance has been attached to a precise analysis of errors that decrease the process margin and error allocation (error budget). For example, even when many chips on a semiconductor substrate have been exposed with the same prescribed exposure dose, the effective appropriate exposure dose varies due to avariation in photoresist sensitivity, post exposure bake (PEB), nonuniformity of development in a substrate, a variation in thickness of a resist film on a substrate, and the like. Consequently, a decrease in manufacturing yield occurs. To efficiently apply the small process margin and to prevent a decrease in the manufacturing yield, a method for implementing feedback or feed forward control by highly accurate monitoring an exposure dose and focusing is needed. At the same time, it is also necessary to perform a precise analysis on error factors which decrease the process margin for each process unit and to correct the major error factors based on results of the analysis.
For controlling an exposure dose and focusing in a reduction projection exposure tool, an exposure dose monitoring method and a focus monitoring method have been proposed (refer to Japanese Patent Laid-Open No. 2001-102282). In the proposed exposure dose monitoring method, an image is transferred with a stepped exposure distribution by using a photomask having a pattern therein with a dimensional ratio (a duty ratio) of a transparent portion and an opaque portion that is continuously changed in one direction by a pitch that can not resolve images on the semiconductor substrate. In the proposed focus monitoring method, focus is monitored using a focus monitor mark having diamond-shaped patterns that shift each phase of the exposure light so as to show different characteristics of pattern dimensions for defocus.
In addition, as a sensitivity control method for a photoresist, the sensitivity is evaluated by measuring a dimension of a delineated pattern on a resist film, which is transferred by a prescribed mask pattern using an exposure tool. Another sensitivity control method for a photoresist has been provided, where the sensitivity is evaluated by determining a clearing dose by a pattern sufficiently larger than a minimum feature size corresponding to a resolution of an exposure tool. Here, the "clearing dose" refers to the lowest exposure dose for which all of the resist film is removed.
As described above, in microfabrication it is important to control exposure conditions for photolithography highly accurately in order to achieve processing accuracy and uniformity in pattern dimensions of semiconductor devices. Particularly, an evaluation method for photoresist sensitivity becomes more important for a fine pattern with smaller exposure latitude and smaller focus margin, or for an isolated fine pattern. However, in the evaluation method for photoresist sensitivity by measuring a dimension of a delineated pattern, the dimension may vary depending on not only the exposure dose but also focusing of an exposure tool. Therefore, using the method for measuring the dimension of the delineated pattern, it is difficult to determine variations in the photoresist sensitivity, separating from the focus variation. Furthermore, with regard to the evaluation method by determining the clearing dose, it is difficult to determine an accurate exposure dose corresponding to the remaining thickness of the resist film, since the remaining thickness changes abruptly in the vicinity of the clearing dose.

Method for evaluating sensitivity of photoresist, method for preparation of photoresist and manufacturing method of semiconductor device


A distinct cultivar of Osteospermum plant named xe2x80x98Seidacrexe2x80x99, characterized by its uniformly mounded and outwardly spreading plant habit; freely branching growth habit; freely flowering habit; and pale yellow-colored ray florets and dark purple-tipped disc florets.

Osteospermum ecklonis cultivar Seidacre.
The present Invention relates to a new and distinct cultivar of Osteospermum plant, botanically known as Osteospermum ecklonis, and hereinafter referred to by the name xe2x80x98Seidacrexe2x80x99.
The new Osteospermum is a product of a planned breeding program conducted by the Inventor in Gunma-den, Japan. The objective of the breeding program is to create new Osteospermum cultivars with interesting floret colors.
The new Osteospermum originated from a cross made by the Inventor in April, 1986 of two unidentified selections of Osteospermum ecklonis, not patented. The new Osteospermum was discovered and selected by the Inventor as a single flowering plant within the progeny of the stated cross grown in a controlled environment in Gunma-den, Japan.
Asexual reproduction of the new Osteospermum by terminal vegetative cuttings was first conducted in Gunma-den, Japan on Jun. 1, 1991. Asexual reproduction by cuttings has shown that the unique features of this new Osteospermum are stable and reproduced true to type in successive generations.
The cultivar Seidacre has not been observed under all possible environmental conditions. The phenotype may vary somewhat with variations in environment such as temperature, daylength and light intensity without, however, any variance in genotype.
The following traits have been repeatedly observed and are determined to be the unique characteristics of xe2x80x98Seidacrexe2x80x99. These characteristics in combination distinguish xe2x80x98Seidacrexe2x80x99 as a new and distinct Osteospermum:
1. Compact, mounded and outwardly spreading plant habit.
2. Freely branching growth habit.
3. Freely flowering habit.
4. Pale yellow-colored ray florets and dark purple-tipped disc florets.
Plants of the new Osteospermum differ from plants of the female parent, an unidentified selection, in the following characteristics:
1. Plants of the new Osteospermum are shorter than plants of the female parent.
2. Plants of the new Osteospermum flower earlier than plants of the female parent.
3. Ray florets of plants of the new Osteospermum are pale yellow in color whereas ray florets of plants of the female parent are white in color.
Plants of the new Osteospermum differ from plants of the male parent, the unidentified Osteospermum selection, primarily in ray floret coloration.
Plants of the new Osteospermum are similar in ray floret coloration to plants of the cultivar Cape Daisy Maputo, not patented. However, plants of the new Osteospermum are more outwardly spreading and more freely flowering than plants of the cultivar Cape Daisy Maputo.


The peripheral pump includes a wall bounding a pump chamber and provided with a substantially annular delivery port with a base having a circular segment with a center; a circular cylindrical impeller revolving in the pump chamber, having two end faces and a rim of blades disposed at intervals along a circumference of the impeller so that a groove-like gap is formed between two adjacent blades, the groove-like gap having a base, as seen in a longitudinal section through the pump, having a circular segment with a center substantially coincident with the center of the circular segment of the delivery port, the center of the circular segments being located within an outer contour of the impeller.

The present invention relates to a peripheral pump. More particularly it relates to a pump for delivering fuel from a storage tank to the internal combustion engine of a motor vehicle.
In particular it relates to a peripheral pump of the above mentioned general type which has a circular-cylindrical impeller revolving in a pump chamber and provided at least on one of its two end faces with a rim of blades disposed at intervals along the circumference of the impeller. Such a pump in which the common center of the two partial circles lies on the axial gap between the impeller and the chamber wall is already known (U.S. Pat. No. 3,315,607). But the efficiency of such peripheral pumps may not be satisfactory.

Peripheral pump, particularly for delivering fuel from a storage tank to the internal combustion engine of a motor vehicle


A power conversion device includes a wire connected to an electrical component, a casing to accommodate therein the electrical component, the casing being provided with an opening through which the wire is led out, a cover component removably attached to an exterior of the casing to cover the opening, and provided with a wiring hole through which the wire passes, and a pressing plate that is a fixing component to fix the wire inside the cover component. The cover component is attachable in two or more different orientations.

There is a commonly-known power conversion device that is connected to an external device such as an electric vehicle having a storage battery installed therein, and that performs power conversion for charging the storage battery from a commercial power supply and power conversion for power feed from the storage battery to a house. A connector is provided at the end of a wire led out of, or withdrawn from, a casing of the power conversion device. The connector is used to connect the wire to the external device. The connector provided at the end of the wire is fitted into a connector of the external device, so that the power conversion device can transmit and receive power to and from the external device.
Patent Literature 1 discloses a charger including a charging cable led out, or withdrawn, from the right side panel of the casing. The casing is constituted by the front panel, the back panel, the right side panel, the left side panel, and the top panel. In the charger disclosed in Patent Literature 1, the charging cable that is a wire used to transmit and receive power is led out from a portion of the right side panel closer to the top panel in a direction perpendicular to the right side panel.