An optimized harmonic drive (“OHD”) includes a wave generator, a flex spline, and a circular rigid spline. The wave generator includes a wave generator contour that minimizes a velocity profile of the wave generator during a high load condition. The flex spline is attached to the wave generator and the circular rigid spline is mechanically engaged to the flex spline. The flex spline includes a plurality of flex spline teeth and the circular rigid spline includes a plurality of rigid spline teeth. The plurality of rigid spline teeth is greater than the plurality of flex spline teeth and the high load condition exists when the rigid spline teeth of the plurality of rigid spline teeth are fully engaged with flex spline teeth of the plurality of flex spline teeth.

1. Field
The present disclosure relates to harmonic drives, and in particularly relates to designing tooth profiles for harmonic drives.
2. Related Art
Harmonic drives are utilized throughout different industries and offer good performance in many categories. In general, harmonic drives are commonly known as devices that are utilized for gear reduction, increasing rotational speed, and differential gearing. Harmonic drives are also known as harmonic or strain wave gears systems and include three basic components that includes a wave generator, a flex spline, and a circular rigid spline. Generally, the wave generator consists of two separate parts that include an elliptical disk and an outer bearing where the elliptical disk is inserted into the outer bearing to produce a bearing with an elliptical shape.
The flex spine is a ring or cup that generally has sides that are thin with teeth positioned radially around the outside of the flex spline. The flex spline fits tightly over the wave generator, so that when the wave generator plug is rotated, the flex spline deforms to the shape of a rotating ellipse and does not slip over the outer elliptical ring of the bearing. The bearing lets the flex spine rotate independently to the wave generator's shaft. The circular spline is a rigid circular ring with teeth on the inside. The flex spline and wave generator are placed inside the circular spline, meshing the teeth of the flex spline and the circular spline. Because the flex spline is deformed into an elliptical shape, the teeth of the flex spline only actually mesh with the teeth of the circular spline in two regions on opposite sides of the flex spline located on the major axis of the ellipse. If the input rotation is produced by the wave generator, as the wave generator plug rotates, the flex spline teeth, which are meshed with the teeth of the circular spline, slowly change position. The major axis of the ellipse of the flex spline rotates with the wave generator, so the points where the teeth mesh revolve around the center point at the same rate as the shaft of the wave generator.
In the design of harmonic drives, there are fewer teeth (e.g., two fewer) on the flex spline than there are on the circular spline. As a result, for every full rotation of the wave generator, the flex spline will rotate a slight amount backward relative to the circular spline (e.g., two teeth in this example). Thus the rotation action of the wave generator results in a much slower rotation of the flex spline in the opposite direction.
As a result, harmonic drives have: no backlash; high compactness and light weight; high gear ratios; reconfigurable ratios within a standard housing; good resolution and repeatability when repositioning inertial loads; high torque capability; possible high gear reduction ratios; and coaxial input and output shafts. However, harmonic drives often suffer from reduced efficiency that may be low as 80%, which is below the more typical 93% efficiency of standard non-harmonic drive gearing.
In FIG. 1, a system diagram of an example of an implementation of a known harmonic drive 100 is shown. The harmonic drive 100 includes a wave generator 102, flex spline 104, and circular rigid spline 106. In this example, the wave generator 102 includes a wave generator plug 108 to couple the wave generator 102 to a drive shaft (not shown) and an outer bearing 110 that is an elliptical disk. The flex spline 104 includes a plurality of teeth 112 and the circular rigid spline 106 also includes a plurality of teeth 114, where the plurality of teeth 112 of the flex spline and plurality of teeth 114 of the circular rigid spline 106 are configured to mechanically interface with each other. Moreover, in this example, a zoomed in view 116 is also shown of an example profile of a tooth 118 of the teeth 112 of the flex spline 104 interfacing with a profile of a tooth 120 of the teeth 114 of the circular rigid spine 106.
In general, much of the loss in efficiency is attributed to friction between the teeth during the meshing process. Known approaches to improve this efficiency by optimizing the tooth shape have been extensively researched with mixed results. Moreover, known harmonic drives (such as harmonic drive 100) have a movement locus of the teeth of the harmonic drive 100 that corresponds to a prolate trochoid motion that has points within the movement locus that change in direction.
FIG. 2 shows a system diagram of a zoomed in view 200, which is similar to zoomed in view 116 shown in FIG. 1. The view 200 includes a plot of the movement locus 202 of a tooth 204 of teeth 112 of the flex spline 104 in relation to a first tooth 206 and a second tooth 208 of teeth 114 of the circular rigid spline 106. In between the first tooth 206 and the second tooth 208 of teeth 114 of the circular rigid spline 106 is a tooth cavity 210 that is configured to receive the tooth 204. In this example, the tooth 204 has a first center position 212 and the tooth cavity 210 has a second center position 214. In this example, the movement locus 202 is obtained when the relative movement 216 of the two gears (i.e., the flex spline 104 and circular rigid spline 106) of the harmonic drive 100 is approximated by a rack. The movement locus 202 is obtained in a principal cross section (i.e., the cross section perpendicular to an axis 218 and selected at a given position of the tooth 204 in the tooth trace direction shown by relative movement 216), which generally constitutes a basis for forming the tooth profiles of the two gears—the flex spline 104 and circular rigid spline 106. In general, the resulting plot of the movement locus 202 is a prolate trochoid motion curve that corresponds to a curve traced by a point extending from the rim of a circular wheel as the wheel rolls along a straight line without slipping where the curve includes loops 220, 222, and 224.
FIG. 3A shows a graph 300 of a plot of a velocity profile 302 of the harmonic drive 100 versus the position of the tooth 204 along a horizontal axis 304 as a result of the prolate trochoid motion along the movement locus of FIG. 2. In this example, the velocity of tooth 204 starts at zero (or approximately zero) at point 306 and increases rapidly along the velocity profile 302 plot until a maximum velocity is reached at a point 308 (along the velocity profile 302) that corresponds to the first center position 212 of tooth 204 reaching the second center position 214 within the tooth cavity 210 while traveling along the plot of the movement locus 202. As such, at point 308, the tooth 204 reaches full engagement with the first tooth 206 and second tooth 208 within the tooth cavity 210. Since the tooth 204 is fully engaged with the first tooth 206 and second tooth 208, this results in a high load condition between the flex spline 104 and circular rigid spline 106. However, since this motion is a prolate trochoid motion that includes at least loop 222, the movement of the tooth 204 relative to the first tooth 206 and second tooth 208 causes an immediate change in direction at point 310 and causes immediate travel at high velocity in the opposite direction while rapidly reducing velocity until reaching a point 312 where the velocity of the velocity profile 302 is about zero. A resulting efficiency from this movement locus 202 is shown in FIG. 3B.
FIG. 3B shows a graph 314 of a plot of a load multiplied by a velocity 316. The area 318 under the plot of the load multiplied by velocity 316 represents the frictional loss that is proportional to the efficiency of the harmonic drive 100 based on the movement locus 202.
Unfortunately, as a result of these issues, at present the best efficiencies for harmonic drives are approximately 90% and these high efficiencies are difficult to achieve consistently because of typical manufacturing variances. As such, there is a need for a method of producing harmonic drives that are more efficient than harmonic drives made using existing approaches.


A lock assembly includes a base member having a bottom with a peripheral wall extending from a periphery defining the bottom, two rods extending from the bottom and two openings defined through the peripheral wall. Each of the rods has a board and a disk respectively and rotatably mounted thereto which is accessed via the openings. Each of the boards has a protrusion extending therefrom which has two sides. A slide has a slot defined therethrough so as to slidably mount to the rods. A periphery defining the slot has two pairs of opposite sides, a width between each of the two opposite sides being smaller than a length of the protrusion and larger than a width between the two sides of the protrusion. The movement of the slide actuates a limit device disposed to the lock assembly.

1. Field of the Invention
The present invention relates to a lock assembly and, more particularly, to an improved lock assembly for a suitcase and/or letter box and including a slide movably disposed relative to at least one set of number setting means which is controllably adjusted to control the slide.
2. Brief Description of the Prior Art
A lock assembly is commonly used to limit two parts, such as a base and a cover of a letter box or a suitcase, being separated to each other so as to keep from access in the letter box. Some of the lock assemblies are equipped with a key which is used to open the lock assembly. However, most of the keys of the lock assemblies are very small and are easily to be lost. Furthermore, if the lock assembly is disposed to a letter box which generally stands outside, the lock assembly tends to get rust. Some lock assemblies need no key at all, but they involve complicated structure and high manufacturing cost.
The present invention intends to provide an improved lock assembly to mitigate and/or obviate the above-mentioned problems.


A method for maintaining the rotational speed of a crankshaft of an internal combustion engine having a plurality of cylinders each having a spark plug wherein a predetermined amount of delivered fuel is to be combusted at a firing time within each of the plurality of cylinders with each rotation of the camshaft or crankshaft includes the step of operating the internal combustion engine, measuring the rotational speed of the crankshaft, defining an expected engine speed, calculating a speed error as the rotational speed of the crankshaft less the expected engine speed, and changing the predetermined amount of delivered fuel to be combusted in each of the plurality of cylinders to reduce the speed error. The preferred embodiment is implemented in fuzzy logic.

1. Field of the Invention
The present invention relates generally to methods of starting and idling an internal combustion engine for an automotive vehicle. More particularly, the present invention relates to a method for starting and idling an internal combustion engine utilizing a dynamic fuel source.
2. Description of the Related Art
Conventional fuel delivery systems for internal combustion engines adjust fuel delivered by a feedback signal created by an exhaust gas oxygen (EGO) sensor to maintain desired stoichiometric combinations. During starting and cold idling, such feedback from the EGO sensor is not available. Therefore, only open loop calculations of fuel requirements are available. A problem with open loop calculation is that such an open loop calculation will not compensate or vary as a function of the fuel blend currently being consumed. This insensitivity to fuel blend varies the operation of the internal combustion engine.
An example of an open loop system is U.S. Pat. No. 5,229,946 which discloses a method for optimizing engine performance for internal combustion engines. This method accounts for different blends of fuel; namely, pure fuels and different blends of fuel and alcohol. This method utilizes specific engine parameters to determine what type of fuel is being combusted. This method utilizes a different engine map for each blend of fuel. This approach is not flexible in that it requires a specific blend of fuel before it can look up a value in a specific map. This method also relies on sensing the amount of fuel in a fuel tank to determine whether a sensing event should even occur.
The method disclosed in U.S. Pat. No. 5,229,946 which fails to immediately determine the composition of the fuel to better enable the internal combustion engine to operate during start-up and idling situations. In fact, this disclosed method does not identify the fuel composition until the fuel tank is refilled. Further, there is no provision to measure the performance of the internal combustion engine. The method merely estimates the performance based on the last identification of fuel composition.

Method for improving spark ignited internal combustion engine starting and idling using poor driveability fuels


An electrical connector electrically connecting an IC package to a printed circuited board comprises a base member (1), a cover member (2) moveably assembled over the base member, an actuator (3) driving the cover member to move over the base member and a plurality of terminals. The base member defines a plurality of receiving cavities (100) therethrough receiving terminals therein. The cover member defines pin mounting holes (200) corresponding to the receiving cavities of the base member. The base member comprises a head portion (13) and a main portion (14) and the cover member correspondingly comprises a side portion (23) and a leading portion (24) respectively. Latching devices, which are formed on the main portion and the leading portion of the base member and the cover member, each comprises a latching hole (5) and a hook (6) mating with the latching hole. Latching holes and hooks are formed on different ones of the base and the cover members. Each latching hole comprises an assembling portion (51) and a retaining portion (52) along a movement direction of the cover member, and a width of the assembling portion is larger than a width of the retaining portion.

1. Field of the Invention
The present invention relates to an electrical connector, and particularly to an electrical connector having latching devices.
2. Description of Related Arts
A typical electrical connector connecting an IC package to a printed circuit board includes a plate-like base, a cover member and a plurality of terminals received in the base. The electrical connector is mounted on the printed circuit board, which has a plurality of conductive pads electrically connecting the terminals of the electrical connector, and the terminals also conductively contact the IC package mounted on the electrical connector, thus permit the interconnecting between the IC package and the printed circuit board.
To adapt the miniaturization of the computer's size, electrical connectors are designed to be smaller day after day, meanwhile the densities of the terminals are becoming much higher. It is becoming more and more important to accurately position the IC package to the electrical connector to ensure the reliability of transmission of electrical signals. Commonly, the cover member of electrical connector is driven by an actuator in a plane parallel to the underlying base, which makes the IC package, whose pins are received in the cover, interconnect with the terminals mounted on the base of the electrical connector without applying any insertion force to the pins of the IC package. It is noted that close coupling of the base and the cover is quite significant to make the better interconnection. However, in some cases, there are only some simple latching members formed on the head or the edges of the connectors. Because of the differences of the structure strength between the components coupling together, when distortion takes place in the base or cover of the electrical connector, the coupling of the base and cover is not quite steady, which may result in unreliable engagement between the terminals and the pins of the IC package. In addition, the transmission of electrical signals would be infected and it may lead to shut off electricity instantly.
Chinese Patent Application No. 01255457.X discloses an electrical connector having improved latching members. Hooks and corresponding latching holes formed on the head of an electrical connector latch with each other to dismiss the distortion which may take place when a cover of the electrical connector is driven by an actuator lever to move over the underlying base. However, the latching members formed only on the head of the electrical connector are not able to effectively eliminate the distortion after the cover and the base are securely coupled together. Furthermore, the same width of the latching holes may lead the hooks to get out from the latching holes while the cover is moving over the underlying base.
Hence, an electrical connector having improved lathing devices is desired.

Electrical connector having latching devices


An acoustic wave identification transponder device, having a substrate, an electroacoustic transfer generating an acoustic wave in said substrate and a set of encoding elements disposed in a path of the acoustic wave for modifying the acoustic wave, having elements which reflect portions of the acoustic wave having a component orthogonal to the incident wave. The reflectors may be, for example, trackchangers or reflective arrays compressors (RAC). The wave may be phase-amplitude modulated for increased efficiency expressed in bits per tap. Such phase amplitude modulation is preferably imposed by partial beam width phase delay elements disposed within an acoustic beam path.

A known radio frequency passive acoustic transponder system produces individualized responses to an interrogation signal. The code space for these devices may be, for example 216 codes, or more, allowing a large number of transponders to be produced without code reuse. These devices provide a piezoelectric substrate on which an aluminum pattern is formed, for example by a typical microphotolithography process, with a minimum feature size of, for example, one micron.
These codes are imposed upon a received signal by signal transforming elements formed on the substrate. Typically, for each encoded symbol, a separate transforming element is formed. In order to facilitate decoding, the transforming elements interact with the interrogation signal at different respective time delays, which are ensured by the acoustic propagation delay on the substrate. These time delays therefore dictate the minimum path length, and therefore size of the substrate. As in many microphotolithographic systems, the substrate size is related to system cost, and smaller substrates imply lower costs. Therefore, a tradeoff exists between substrate size and available encoding complexity.
The aforementioned transponder devices including a surface acoustic wave device, in which an identification code is provided as a characteristic time-domain reflection, attenuation, phase delay, and/or transducer interaction pattern in a retransmitted signal, in a system which generally requires that the signal emitted from an exciting antenna be non-stationary with respect to a signal received from the tag. This ensures that the reflected signal pattern is distinguished from the emitted signal, and can be analyzed in a plurality of states. This analysis reveals the various delay components within the device. In such a device, received RF energy, possibly with harmonic conversion, is emitted onto a piezoelectric substrate as an acoustic wave by means of an interdigital electrode system, from whence it travels through the substrate, interacting with reflecting, delay or resonant/frequency selective elements in the path of the wave. A portion of the acoustic wave is ultimately received by an interdigital electrode system, which may be the same or different than the launch transducer, and retransmitted. These devices do not require a semiconductor memory nor external electrical energy storage system, e.g., battery or capacitor, to operate. The propagation velocity of an acoustic wave in a surface acoustic wave device is slow as compared to the free space propagation velocity of a radio wave. Thus, the time for transmission between the radio frequency interrogation system and the transponder is typically short as compared to the acoustic delay of the substrate. This allows the rate of the interrogation frequency change to be based primarily on the delay characteristics within the transponder, without requiring measurements of the distance between the transponder and the interrogation system antenna.
The interrogation frequency is controlled to change sufficiently from the return or "backscatter" signal from the transponder, so that a return signal having a minimum delay may be distinguished from the interrogation frequency, and so that all of the relevant delays are unambiguously received for analysis. The interrogation frequency thus should not return to the same frequency within a minimum time-period. Generally, such systems are interrogated with a pulse transmitter or chirp frequency system.
Systems for interrogating a passive transponder employing acoustic wave devices, carrying amplitude and/or phase-encoded information are disclosed in, for example, U.S. Pat. Nos. 4,059,831; 4,484,160; 4,604,623; 4,605,929; 4,620,191; 4,623,890; 4,625,207; 4,625,208; 4,703,327; 4,724,443; 4,725,841; 4,734,698; 4,737,789; 4,737,790; 4,951,057; 5,095,240; and 5,182,570, expressly incorporated herein by reference. Other passive interrogator label systems are disclosed in the U.S. Pat. Nos. 3,273,146; 3,706,094; 3,755803; and 4,058,217, expressly incorporated herein by reference.
Passive transponder tag interrogation systems are also known with separate receiving and transmitting antennas, which may be at the same frequency or harmonically related, and having the same or different polarization. Thus, in these systems, the transmitted and received signals may be distinguished other than by frequency. The acoustic wave is often a surface acoustic wave, although acoustic wave devices operating with various other wave types, such as bulk waves, are known.
The information code associated with and which identifies the passive transponder is built into the transponder at the time that a layer of metallization is finally defined on the substrate of piezoelectric material. This metallization also defines the antenna coupling, launch transducers, acoustic pathways and information code elements, e.g., reflectors and delay elements. Thus, the information code in this case is non-volatile and permanent. The information representing these elements is present in the return signal as a set of characteristic perturbations of the interrogation signal, such as a specific complex delay pattern and attenuation characteristics. In the case of a transponder tag having launch transducers and a variable pattern of reflective elements, the number of possible codes is N>2M were N is the number of acoustic waves launched by the transducers and M is the number of reflective element positions for each transducer. Thus, with four launch transducers each emitting two acoustic waves, and a potential set of eight variable reflective elements in each acoustic path, the number of differently coded transducers is 2048. Therefore, for a large number of potential codes, it is necessary to provide a large number of launch transducers and/or a large number of reflective elements. However, efficiency is lost with increasing acoustic path complexity (i.e., power splitting), and a large number of distinct acoustic waves reduces the signal strength of the signal encoding the information in each. Therefore, the transponder design is a tradeoff between device codespace complexity and efficiency.
the known passive acoustic transponder tag thus typically includes a multiplicity of "signal conditioning elements", i.e., delay elements, reflectors, and/or amplitude modulators, which are coupled to receive the first signal from a transponder antenna. Each signal conditioning element provides an intermediate signal having a known delay and a known amplitude modification to the acoustic wave interacting with it. Even where the signal is split into multiple portions, it is advantageous to reradiate the signal through a single antenna. Therefore, a "signal combining element" coupled to the all of the acoustic waves, which have interacted with the signal conditioning elements, is provided for combining the intermediate signals to produce the radiated transponder signal. The radiated signal is thus a complex composite of all of the signal modifications, which may occur within the transponder, modulated on the interrogation wave.
In known passive acoustic transponder systems, the transponder remains static over time, so that the encoded information is retrieved by a single interrogation cycle, representing the state of the tag, or more typically, obtained as an inherent signature of an emitted signal due to internal time delays. In order to determine a transfer function of a passive transponder device, the interrogation cycle may include measurements of excitation of the transponder at a number of different frequencies. This technique allows a frequency domain analysis, rather than a time domain analysis of an impulse response of the transponder. This is particularly important since time domain analysis requires very high time domain resolution, e.g., <100 nS, to accurately capture the characteristics of the encoding, while frequency domain analysis does not impose such stringent requirements on the analysis system.
Passive transponder encoding schemes include selective modification of interrogation signal transfer function H(s) and delay functions f(z). These functions therefore typically generate a return signal in the same band as the interrogation signal. Since the return signal is mixed with the interrogation signal, the difference between the two will generally define the information signal, along with possible interference and noise. By controlling the rate of change of the interrogation signal frequency with respect to a maximum round trip propagation delay, including internal delay, as well as possible Doppler shift, the maximum bandwidth of the demodulated signal may be controlled. Thus, the known systems seek to employ a chirp interrogation waveform which allows a relatively simple processing of limited bandwidth signals.
Typically, the interrogator transmits a first signal having a first frequency that successively assumes a plurality of incremental frequency values within a prescribed frequency range. This first frequency may, for example, be in the range of 905–925 MHz, referred to herein as the nominal 915 MHz band, a frequency band that is commonly available for unlicensed use. Of course, other bands may be used, and preferably these are bands which do not require a license, and are available worldwide for use. These bands extend, for example, from 100 MHz to 25 GHz. Of course, licensed bands and locally available bands may be used. The response of the tag to excitation at any given frequency is distinguishable from the response at other frequencies, due to the relationship of the particular frequency and fixed time delays.
Preferably, the passive acoustic wave transponder tag includes at least one element having predetermined characteristics, which assists in synchronizing the receive and allowing for temperature compensation of the system. As the temperature changes, the piezoelectric substrate may expand and contract, altering the characteristic delays and other parameters of the tag. Variations in the transponder response due to changes in temperature thus result, in part, from the thermal expansion of the substrate material. Although propagation distances are small, an increase in temperature of only 20° C. can produce an increase in propagation time by the period of one entire cycle at a transponder frequency of about 915 MHz; correspondingly, a change of about 1° C. results in a relative phase change of about 18°.
A known transponder is constructed such that ith delay time ti=T0+KΔT+ΔVi, where K is a proportionality constant, ΔT is the nominal, known difference in delay time between the intermediate signals of two particular successive ones of the signal delay elements in the group, and □Vi is a modification factor due to inter-transponder variations, such as manufacturing variations. By measuring the quantities ΔT and ΔVi, it is possible, according to known techniques, to determine the expected delay time ti-T0 for each and every signal delay element from the known quantities K, ΔT and ΔVi. The manufacturing variations ΔVi comprise a "mask" variation ΔMi due to imperfections in the photolithographic mask; an "offset" variation ΔOi which arises from the manufacturing process used to deposit the metal layer on the piezoelectric substrate; and a random variation ΔRi which is completely unpredictable but usually neglectably small. Specific techniques are available for determining and compensating both the mask variations ΔMi and the offset variations ΔOi.
The known chirp interrogation system for interrogating surface acoustic wave transponder system provides a number of advantages, including high signal-to-noise performance. Further, the output of the signal mixer—namely, the signal which contains the instantaneous difference frequencies of the interrogating chirp signal and the transponder reply signal, typically fall in (or may be made to fall in) the range below 3000 Hz, and thus may be transmitted over inexpensive, shielded, audio-grade twisted-pair wires, and indeed may possibly be transmitted over the telephony infrastructure. Furthermore, since signals of this type are not greatly attenuated or dispersed when transmitted over long distances, the signal processor may be located at a position quite remote from the signal mixer, or provided as a central processing site for multiple interrogator antennae.
Another known type of interrogation system employs impulse excitation. These systems require broadband transponder signal analysis, and thus cannot typically employ audio frequency (low frequency) analysis systems. This impulse excitation interrogation system does not seek to analyze the response of fixed elements within the passive transponder to a plurality of different excitation challenges.
A known surface acoustic wave passive interrogator label system, as described, for example, in U.S. Pat. Nos. 4,734,698; 4,737,790; 4,703,327; and 4,951,057, expressly incorporated herein by reference, includes an interrogator having a voltage controlled oscillator which produces a first radio frequency signal determined by a control voltage. This first signal is amplified by a power amplifier and applied to an antenna for transmission to a remote transponder as an interrogation signal. As is known, the voltage controlled oscillator may be replaced with other oscillator (or signal generator) types.
The first signal is received by an antenna of the remote transponder and passed to a signal transforming element, which converts the first (interrogation) signal into a second (reply) signal, encoded with a characterizing information pattern. The information pattern is encoded by a series of elements having characteristic delay periods T0 and ΔT1, ΔT2, . . . ΔTN.
Two common types of systems exist. In a first, the delay periods correspond to physical delays in the propagation of the acoustic signal. After passing each successive delay, a portion of the signal I0, I1, I2 . . . IN is tapped off and supplied to a summing element. The resulting signal, which is the sum of the intermediate signals I0 . . . IN, is fed back to a transponder tag antenna, which may be the same or different than the antenna which received the interrogation signal, for transmission to the interrogator/receive antenna. In a second system, the return signal is composed of sets of reflected signals, resulting from reflectors in the path of the signal which reflect portions of the acoustic wave back to the launch transducer, where they are converted back to an electrical signal and emitted by the transponder tag antenna. The second signal is passed either to the same or different antenna of the remote transponder for transmission back to the interrogator/receiver apparatus. In both cases, between the taps or reflectors, signal modification elements, such as delay pads, selectively modify the signal. This second signal carries encoded information which, at a minimum, identifies the particular transponder.
The transponder serves as a signal transforming element, which comprises N+1 signal conditioning elements and a signal combining element, where necessary. The signal conditioning elements are selectively provided to impart a different response code for different transponders, and which may involve separate intermediate signals I0, I1 . . . IN within the transponder. Each signal conditioning element comprises a known delay Ti and a known amplitude modification Ai (either attenuation or amplification). The respective delay Ti and amplitude modification Ai may be functions of the frequency of the received first signal, may provide a constant delay and constant amplitude modification, respectively, independent of frequency or may have differing dependency on frequency. The order of the delay and amplitude modification elements may be reversed; that is, the amplitude modification elements Ai may precede and the delay elements Ti. Amplitude modification Ai can also occur within the path Ti. The signals are combined, where necessary, in a combining element which combines these intermediate signals (e.g., by addition, multiplication or the like) to form the second (reply) signal S2 and the combined signal is emitted by the transponder antenna.
The second signal is picked up by a receiving antenna of the interrogation apparatus. Both this second signal and the first signal (or respective signals derived from these two signals) are applied to a mixer (four quadrant multiplier) to produce a third signal containing frequencies which include both the sums and the differences of the frequencies contained in the first and second signals. The third signal is then low-pass filtered (to pass the difference frequency and block the input signals and sum frequency), digitized and passed to a digital signal processor which determines the amplitude ai and the respective phase φi of each frequency component fi among a set of frequency components (f0, f1, f2 . . . ) in the filtered third signal. The filter thus distinguishes the sum and difference components, and prevents aliasing in the analog-to-digital converter. Typically, the low pass filter is set to have a narrow passband, to filter transients and reduce Gaussian noise. For example, in a known system with a frequency hopping rate of 8,000 per second, the filter has a cutoff of about 3,000 Hz. This narrow bandwidth allows a relatively slow analog to digital converter, e.g., about 10 ksps, to be employed to digitize the signal.
Each phase φi is determined with respect to the phase φ0=0 of the lowest frequency component f0. The third signal may be intermittently supplied to the mixer by means of a switch, and indeed the signal processor may be time-division multiplexed to handle a plurality of mixed (demodulated) signals from different antennas.
The information determined by the digital signal processor is passed to a microprocessor computer system. This computer system analyzes the frequency, amplitude and phase information and makes decisions based upon this information. For example, the computer system may determine the identification number of the interrogated transponder. This identification number and/or other decoded information is made available at an output.
In one known interrogation system embodiment, the voltage controlled oscillator is controlled to produce a sinusoidal RF signal with a frequency that is incrementally swept in 128 equal discrete steps from 905 MHz to 925 MHz. Each frequency step is maintained for a period of 125 microseconds so that the entire frequency sweep is carried out in 16 milliseconds, with a step rate of 8 kHz. Thereafter, the frequency is dropped back to 905 MHz in a relaxation period of 0.67 milliseconds. This stepwise frequency sweep approximates a linear frequency sweep. In this embodiment, each delayed component within the replay (second) signal has a different frequency with respect to the instantaneous interrogation (first) signal.
Assuming a round-trip, radiation transmission time of t0, the total round-trip times between the moment of transmission of the first signal and the moments of reply of the second signal will be t0+T0, t0+T1, . . . t0+TN, for the delays T0N, T . . . , T1 respectively. Considering only the transponder delay TN, at the time tR when the second (reply) signal is received at the antenna, the frequency of this second signal will be ΔfN less than the instantaneous frequency of the first signal transmitted by the interrogator system antenna. Thus, if the first and second signals are mixed or "homodyned", this frequency difference ΔfN will appear in the third signal as a beat frequency. Understandably, other beat frequencies will also result from the other delayed frequency spectra resulting from the time delays T0, T1 . . . TN-1. In the case of a "chirp" waveform, the difference between the emitted and received waveform will generally be constant, and therefore the relationship of each delayed component can be determined.
As can be seen, in this embodiment, all significant components of the third (mixed) signal will be within a limited range defined by the maximum delay within the transponder signal transformer and the chirp rate. Thus, this signal may be band limited within this range without loss of significant information. In a known system with a chirp range of 20 MHz, over a cycle period of 16 mS, with 128 transitions, the frequency difference per transition is 156,250 Hz.
In one embodiment of a passive transponder, the internal circuit is a surface acoustic wave device which operates to convert the received first signal into an acoustic wave, and then to reconvert the acoustic energy back into the second signal for transmission via a dipole antenna. The signal transforming element of the transponder includes a piezoelectric substrate material such as a lithium niobate (LiNbO3) crystal, which has a free surface acoustic wave propagation velocity of about 3488 meters/second. The substrate is, for example, a 3 mm by 5 mm rectangular slab having a thickness of 0.5 mm. On the surface of this substrate is deposited a layer of metal, such as aluminum, forming a pattern which includes transducers and delay/reflective elements. Each delay element has a width sufficient to delay the propagation of the surface acoustic wave from one tap transducer to the next by one quarter cycle or 90° with respect to an underlayed wave at the frequency of operation (in the 915 MHz band). By providing locations for three delay pads between successive tap transducers, the phase φ of the surface acoustic wave received by a tap transducer may be controlled to provide four phase possibilities, zero pads=0°; one pad=90°; two pads=180°; and three pads=270°. These pads may be selectively deposited as a metallization layer during manufacture, or formed in a complete complement and selectively removed during a secondary process to encode the transponder. Where a reflective element returns the signal to the launch transducer, the delays are calculated based on two passes over the pad. Typically, a reflective or semireflective element is provided between each set of delay pads to allow them to be distinguished, and allowing, in the case of semireflective elements, for a series of sets of delay pads to be disposed along the path of an acoustic wave. As the number of sets of delay pads increases, the signal to noise ratio in the transponder relay signal is severely degraded. This limitation on the number of tap transducers places a limitation on the length of the informational code imparted in the transponder replies.
A plurality of launch transducers may be connected to common bus bars which, in turn, are connected to the dipole antenna of the transponder. Each launch transducer may have a forward and backward wave, and, indeed, care must be taken to damp a reverse wave where this emission is undesired in order to reduce interference. Thus, the codespace of the transponder may include a plurality of sets of encoding elements, each associated with a particular wavepath. Opposite each launch transducer is one or more reflectors, which reflect surface acoustic waves back toward the transducers which launched them. Since the transducers are connected in parallel, a radio frequency interrogation pulse is received by all the transducers essentially simultaneously. Consequently, these transducers simultaneously generate surface acoustic waves which are transmitted outward in both directions. The system is configured so that the reflected surface acoustic waves are received by their respective transducers at staggered intervals, so that a single interrogation pulse produces a series of reply pulses after respective periods of delay.
FIG. 1 shows a known an interrogator system comprising a voltage controlled oscillator 10 which produces a first signal S1 at a radio frequency determined by a control voltage V supplied by a control unit 12. This signal S1 is amplified by a power amplifier 14 and applied to an antenna 16 for transmission to a transponder 20.
The signal S1 is received at the antenna 18 of the transponder 20 and passed to a signal transforming element 22. This signal transformer converts the first (interrogation) signal S1 into a second (reply) signal S2, encoded with an information pattern. The information pattern is encoded as a series of elements having characteristic delay periods T0 and ΔT1, ΔT2, . . . ΔTN. Two common types of transducer devices exist. In a first, shown schematically in FIG. 5, the delay periods correspond to physical delays in the propagation of the acoustic signal. After passing each successive delay, a portion of the signal I0, I1, I2 . . . IN is tapped off and supplied to a summing element. The resulting signal S2, which is the sum of the intermediate signals I0 . . . IN, is fed back to a transponder tag antenna, which may be the same or different than the antenna which received the interrogation signal, for transmission to the interrogator/receive antenna. In a second system, shown schematically in FIG. 4, the delay periods correspond to the positions of reflective elements, which reflect portions of the acoustic wave back to the launch transducer, where they are converted back to an electrical signal and emitted by the transponder tag antenna.
The signal S2 is passed either to the same antenna 18 or to a different antenna 24 for transmission back to the interrogator/receiver apparatus. This second signal S2 carries encoded information which, at a minimum, identifies the particular transponder 20.
The signal S2 is picked up by a receiving antenna 26. Both this second signal S2 and the first signal S1 (or respective signals derived from these two signals) are applied to a mixer (four quadrant multiplier) 30 to produce a third signal S2 containing frequencies which include both the sums and the differences of the frequencies contained in the signals S1 and S2. The signal S3 is passed to a signal processor 32 which determines the amplitude ai and the respective phase φ, of each frequency component fi among a set of frequency components (f0, f1, f2 . . . ) in the signal S3. Each phase φi is determined with respect to the phase φ0=0 of the lowest frequency component f0. The signal S3 may be intermittently supplied to the mixer by means of a switch.
The information determined by the signal processor 32 is passed to a computer system comprising, among other elements, a random access memory (RAM) 34 and a microprocessor 36. This computer system analyzes the frequency, amplitude and phase information and makes decisions based upon this information. For example, the computer system may determine the identification number of the interrogated transponder 20. This I.D. number and/or other decoded information is made available at an output 38.
The transponder, as shown in FIG. 2, serves as a signal transforming element 22, which comprises N+1 signal conditioning elements 40 and a signal combining element 42. The signal conditioning elements 40 are selectively provided to impart a different response code for different transponders, and which may involve separate intermediate signals I0, I1 . . . IN within the transponder. Each signal conditioning element 40 comprises a known delay Ti and a known amplitude modification Ai (either attenuation or amplification). The respective delay Ti and amplitude modification Ai may be functions of the frequency of the received signal S1, or they may provide a constant delay and constant amplitude modification, respectively, independent of frequency. The time delay and amplitude modification may also have differing dependency on frequency. The order of the delay and amplitude modification elements may be reversed; that is, the amplitude modification elements Ai may precede the delay elements Ti. Amplitude modification Ai can also occur within the path Ti.
The signals are combined in combining element 42 which combines these intermediate signals (e.g., by addition, multiplication or the like) to form the reply signal S2 and the combined signal emitted by the antenna 18.
In one embodiment, as shown in FIGS. 3A and 3B, the voltage controlled oscillator 10 is controlled to produce a sinusoidal RF signal with a frequency that is swept in 128 equal discrete steps from 905 MHz to 925 MHz. Each frequency step is maintained for a period of 125 microseconds so that the entire frequency sweep is carried out in 16 milliseconds. Thereafter, the frequency is dropped back to 905 MHz in a relaxation period of 0.67 milliseconds. The stepwise frequency sweep 46 shown in FIG. 3B thus approximates the linear sweep 44 shown in FIG. 3A.
Assuming that the stepwise frequency sweep 44 approximates an average, linear frequency sweep or "chirp" 47, FIG. 3B illustrates how the transponder 20, with its known, discrete time delays T0, T1 . . . TN produces the second (reply) signal 52 with distinct differences in frequency from the first (interrogation) signal 51. Assuming a round-trip, radiation transmission time of ta, the total round-trip times between the moment of transmission of the first signal and the moments of replay of the second signal will be t0+T0, t0+T1, . . . t0+TN, for the delays TON, T . . . T1 respectively. Considering only the transponder delay TN, at the time tg when the second (reply) signal is received at the antenna 26, the frequency 48 of this second signal will be ΔfN less than the instantaneous frequency 47 of the first signal S1 transmitted by the antenna 16. Thus, if the first and second signals are mixed or "homodyned", this frequency difference ΔfN will appear in the third signal as a best frequency. Understandably, other beat frequencies will also result from the other delayed frequency spectra 49 resulting from the time delays T0, T1 . . . TN-1. thus, in the case of a "chirp" waveform, the difference between the emitted and received waveform will generally be constant.
In mathematical terms, we assume that the phase of a transmitted interrogation signal is φ=2πfτ, where τ is the round-trip transmission time delay. For a ramped frequency df/dt or f, we have: 2πfτ=dφ/dt=ω. ω, the beat frequency, is thus determined by τ for a given ramped frequency or chirp f. In this case, the (mixed) signal S3 may be analyzed by determining a frequency content of the S3 signal, for example by applying it to sixteen bandpass filters, each tuned to a different frequency, f0, f1 . . . fE, fF. the signal processor determines the amplitude and phase of the signals that pass through these respective filters. These amplitudes and phases contain the code or "signature" of the particular signal transformer 22 of the interrogated transponder 20. This signature may be analyzed and decoded in known manner.
In one embodiment of a passive transponder, shown in FIGS. 6 and 7, the internal circuit operates to convert the received signal S1 into an acoustic wave and then to reconvert the acoustic energy back into an electrical signal S2 for transmission via a dipole antenna 70, connected to, and arranged adjacent a SAW device made of a substrate 72. More particularly, the signal transforming element of the transponder includes a substrate 72 of piezoelectric material such as a lithium niobase (LiNbO3) crystal, which has a free surface acoustic wave propagation velocity of about 3488 meters/second. On the surface of this substrate is deposited a layer of metal, such as aluminum, forming a pattern which includes transducers and delay/reflective elements.
One transducer embodiment includes a pattern consisting of two bus bars 74 and 76 connected to the dipole antenna 70, a "launch" transducer 78 and a plurality of "tap" transducers 80. The bars 74 and 76 thus define a path of travel S2 for a surface acoustic wave which is generated by the launch transducer and propagates substantially linearly, reaching the tap transducers each in turn. The tap transducers convert the surface acoustic wave back into electric energy which is collected and therefore summed by the bus bars 74 and 76. This electrical energy then activates the dipole antenna 70 and is converted into electromagnetic radiation for transmission as the signal S2.
The tap transducers 80 are provided at equally spaced intervals along the surface acoustic wave path 82, as shown in FIG. 6, and an informational code associated with the transponder is imparted by providing a selected number of "delay pads" 84 between the tap transducers. These delay pads, which are shown in detail in FIG. 7, are preferably made of the same material as, and deposited with, the bus bars 74, 76 and the transducers 78, 80. Each delay pad has a width sufficient to delay the propagation of the surface acoustic wave from one tap transducer 80 to the next by one quarter cycle or 90° with respect to an undelayed wave at the frequency of operation (in the 915 MHz band). By providing locations for three delays pads between successive tap transducers, the phase f of the surface acoustic wave received by a tap transducer may be controlled to provide four phase possibilities, zero pads=0°; one pad=90°; two pads=180°; and three pads=270°.
The phase information φ0 (the phase of the signal picked up by the first tap transducer in line), and φ1, φ2 . . . φN (the phases of the signals picked up by the successive tap transducers) is supplied to the combiner (summer) which, for example, comprises the bus bars 74 and 76. This phase information, which is transmitted as the signal S2 by the antenna 70, contains the informational code of the transponder. The Signal S2 also includes interrogation frequency, dependent amplitude variations as a result of the encoding elements, which typically also provide information for decoding the transponder identity.
As shown in FIG. 7, the three delay pads 84 between two tap transducers 80 are each of such a width L as to each provide a phase delay of 90° in the propagation of an acoustic wave from one tap transducer to the next as compared to the phase in the absence of such a delay pad. This width L is dependent upon the material of both the substrate and the delay pad itself as well as upon the thickness of the delay pad and the wavelength of the surface acoustic wave.
The transducers are typically fabricated by an initial metallization of the substrate with a generic encoding, i.e., a set of reflectors or delay elements which may be further modified by removal of metal to yield the customized transponders. Thus, in the case of delay pads, three pads are provided between each set of transducers or taps, some of which may be later removed. Where the code space is large, the substrates may be partially encoded, for example with higher order code elements, so that only the lower order code elements need by modified in a second operation.
While a system of the type described above operates satisfactorily when the number of tap transducers does not exceed eight, the signal to noise ratio in the transponder reply signal is reduced as the number of tap transducers increases. This is because the tap transducers additionally act as launch transducers as well as partial reflectors of the surface acoustic wave so that an increase in the number of tap transducers results in a corresponding increase in spurious signals in the transponder replies. This limitation on the number of tap transducers places a limitation on the length of the informational code imparted in the transponder replies.
Spurious signals as well as insertion losses may be reduced in a passive transponder so that the informational code may be increased in size to any desired length, by providing one or more surface acoustic wave reflectors on the piezoelectric substrate in the path of travel of the surface acoustic wave, to reflect the acoustic waves back toward a transducer for reconversion into an electric signal.
Surface acoustic waves may encounter frequency selective filtering structures, partial reflectors, full reflectors, phase delay pads or electroacoustic transducing elements as they travel across the substrate, which is typically lithium niobate (LiNbO3) which has a surface acoustic wave propagation velocity of 3488 m/sec and is piezoelectric. The system may have a single acoustic path or sets of acoustic paths which are, for example, parallel, as shown in FIG. 8A.
A wavefront produced by reflections from the leading and trailing edges of transducer fingers will be formed by the superposition of a first wave reflected from the first leading edge and successive waves reflected from successive edges and having differences in phase, with respect to the first wave, of −λ/4, λ/2, −3 λ/4, λ, etc. As may be seen, the wave components having a phase −λ4, λ/2 and −λ/4 effect a cancellation, or at least an attenuation of the wave component reflected from the leading edge.
The interdigital fingers of the transducers may therefore be advantageously split to reduce reflections. Conventional interdigital finger transducers which are constructed to operate at a fundamental, resonant frequency of 915 MHz, have a finger width (λ/4 ) of approximately 1 micron; a size which approaches the resolution limit of certain photolithographic fabrication techniques (the selective removal of metallization by (1) exposure of photoresist through a mask and (2) subsequent etching of the metallized surface to selectively remove the metal between and outside the transducer finger). If the fingers are split, the width of each finger (λ/8) for a fundamental frequency of 915 MHz would be approximately 0.5 micron. The size would require sophisticate photolithographic fabrication techniques. In order to increase the features sizes, the transducers in the transponder are constructed with a resonant frequency f0 of 305 MHz. In this case, the width of each finger is three times larger than transducer fingers designed to operate at 915 MHz, so that the width (λ/8) of the split fingers is approximately 1.5 microns. This is well within the capability of typical photolithographic fabrication techniques. Although such transducers are constructed with a resonant frequency of 305 MHz, they are nevertheless driven at the interrogation frequency of approximately 915 MHz; i.e., a frequency 3f0 which is the third harmonic of 305 MHz. The energy converted by a transducer, when driven in its third harmonic 3f0 (915 MHz), is about ⅓ of the energy that would be converted if the transducer were driven at its fundamental frequency f0 (305 MHz). Accordingly, it is necessary to construct the transducers to be as efficient as possible within the constraints imposed by the system. As is well known, it is possible to increase the percentage of energy converted, from electrical energy to SAW energy and vice versa, by increasing the number of fingers in a transducer. In particular, the converted signal amplitude is increased by about 2% for each pair of transducer fingers (either conventional fingers or split fingers) so that, for 20 finger pairs for example, the amplitude of the converted signal will be about 40% of the original signal amplitude. Such an amplitude percentage would be equivalent to an energy conversion of about 16%. In other words, the energy converted will be about 8 dB down from the supplied energy.
The edge portions of the delay pads, as well as the lateral edges of the bus bars and (i.e., the edges transverse to the SAW paths of travel) are advantageously provided with two levels of serrations, which substantially reduce SAW reflections from these edges. The serrations include for example, two superimposed "square waves" having the same pulse height but different pulse periods. For example, the pulse height for both square waves is λ/4, and the pulse period is λ/3 for one square wave and 6 λ for the other, where λ is the SAW wavelength at 915 MHz. The first level of serrations serves to reduce reflections, while the second level serves to break up the average reflection plane.
The addition of finger pairs to the transducers therefore advantageously increases the energy coupling between electrical energy and SAW energy. However, as explained above in connection with FIGS. 1, 3A and 3B, the system according to the invention operates to excite the transducers over a range of frequencies between 905 MHz and 925 MHz. This requires the transducer to operate over a 20–25 MHz bandwidth: a requirement which imposes a constraint upon the number of transducer finger pairs because the bandwidth of a transducer is inversely proportional to its physical width. This relationship arises from the fact that the bandwidth is proportional to 1/τ, where τ is the SAW propagation time from one side of the transducer to the other (the delay time across the transducer).
The transducer may be divided into several separate sections: a central section, two flanking sections and two outer sections. The central section includes interdigital transducer fingers which are alternately connected to two outer bus bars and to a central electrical conductor. This central section comprises a sufficient number of finger pairs to convert a substantial percentages of electrical energy into SAW energy and vice versa. By way of example and not limitation, there may be 12 finger pairs so that the converted amplitude is approximately 24% of the incoming signal amplitude. Flanking the central section, on both sides, are sections containing "dummy" fingers; that is, fingers which are connected to one electrode only and therefore serve neither as transducers nor reflectors. The purpose of these fingers is to increase the width of the transducer so that the outer sections will be spaced a prescribed distance, or SAW delay time, from the central section. For example, there may be 7 dummy fingers (or, more particularly, split fingers) in each of the sections. Finally, each of the outer sections of the transducer contains a single transducer finger pair which is used to shape the bandwidth of the transducer, e.g., maintain an effective bandwidth of about 25 MHz.
The transducer system preferably has an electrical impedance at the design frequency which matches the impedance of the antenna coupling, to maximize the power transfer between the antenna system and transducer. This matching is accomplished by forming series connections of transducer structures, which present as capacitive loads, to reduce impedance, as necessary, and providing heavy metal traces for the bus bars to reduce Ohmic losses. The bus bars are, for example, made approximately twice as thick as the other metallized elements on the substrate.
In practice, the metallization is deposited on the substrate surface using a two-layer photolithographic process. Two separate reticles are used in forming the photolithographic image: one reticle for the transducers, reflectors and phase pads as well as the alignment marks on the substrate, and a separate reticle for the bus bar. The process thus comprises the steps of depositing a 300 Angstrom layer of chromium and then 1000 Angstrom layer of aluminum on the substrate, followed by UV-exposure solubilizing resin spin coating, masking and etching of the bus bars, followed by deposition of 1000 Angstroms of aluminum and another US activated resin spin coating, masking and etching to form the transducers, reflectors and phase pads as well as the alignment marks on the substrate, doubling the thickness of the bus bar structures.
Each two successive fingers of a transducer may be shorted at one or more locations between the bus bars. The shorts between successive fingers reduce energy loss due to Ohmic resistance of the fingers and render the reflector less susceptible to fabrication errors.
These various techniques and systems, described above, may advantageously be employed or combined with aspects of the present invention in known manner to achieve desired results.
The embodiment of FIG. 8A comprises a substrate 120 of piezoelectric material, such as lithium niobate, on which is deposited a pattern of metallization essentially as shown. The metallization includes two bus bars 122 and 124 for the transmission of electrical energy to four launch transducers 126, 128, 130 and 132. These launch transducers are staggered, with respect to each other, with their leading edges separated by distances X, Y and Z, respectively, as shown. The distances X and Z are identical; however, the distance Y is larger than X and Z in order to provide temporal separation of the received signals corresponding to the respective signal paths. Further metallization includes four parallel rows of delay pads 134, 136, 138 and 140 and four parallel rows of reflectors 142, 144, 146, and 148. The two rows of reflectors 144 and 146 which are closest to the transducers are called the "front rows" whereas the more distant rows 142 and 148 are called the "back rows" of the transponder. The bus bars 122 and 124 includes contact pads 150 and 152, respectively, to which are connected the associated poles 154 and 156 of a dipole antenna. These two poles are connected to the contact pads by contact elements or wires 158 and 160, represented in dashed lines.
The provision of four transducers 126, 128, 130 and 132 and two rows of reflectors 142, 144, 146, and 148 on each side of the transducers results in a total of sixteen SAW pathways of different lengths and, therefore, sixteen "taps". These sixteen pathways (taps) are numbered 0, 1, 2 . . . D, E, F, as indicated by the reference number (letter) associated with the individual reflectors. Thus, pathway 0 extends from transducer 126 to reflector 0 and back again to transducer 126. Pathway 1 extends from transducer 128 to reflector 1 and back again to transducer 128. The spatial difference in length between pathway 0 and pathway 1 is twice the distance X (the offset distance between transducers 126 and 128). This results in a temporal difference of ΔT in the propagation time of surface acoustic waves. Similarly, pathways 2 extends from transducer 126 to reflector 2 and back again to transducer 126. Pathway 3 extends from transducer 128 to reflector 3 and back to transducer 128. The distance X is chosen such that the temporal differences in the length of the pathway 2 with respect to that of pathway 1, and the length of the pathway 3 with respect to that of pathway 2 are also both equal to ΔT. The remaining pathways 4, 5, 6, 7 . . . . E, D, F are defined by the distances from the respective transducers launching the surface acoustic waves to the associated reflectors and back again. The distance Y is equal to substantially three times the distance X so that the differences in propagation times between pathway 3 and pathway 4 on one side of the device, and pathway B and pathway C on the opposite side are both equal to ΔT. With one exception, all of the temporal differences, from one pathway to the next successive pathway are equal to the same ΔT. The SAW device is dimensioned so that ΔT nominally equals 100 nanoseconds. In order to avoid the possibility that multiple back and forth propagations along a shorter pathway (one of the pathways on the left side of the SAW device as seen in FIG. 1) appear as a single back and forth propagation along a longer pathway (on the right side of the device), the difference in propagation times along pathways 7 and 8 is made nominally equal to 150 nanoseconds.
FIG. 8B shows, for a single transducer 125, connected between bus bars 122 and 124, a set o acoustic wave paths reflecting off encoding elements and returning to the transducers 125.


A method is provided for detecting whether a test composition can inhibit cell fusion mediated by paramyxovirus infection. A method is also provided for identifying an antiviral peptide capable of inhibiting paramyxovirus-induced cell fusion. The methods are based on the discovery that the hemagglutination and fusion-inducing proteins of paramyxoviruses interact to form a complex termed HN:F, that is prerequisite to cell fusion. Antibodies to HN:F provide a novel means of providing improved immunity against paramyxovirus infection.

The paramyxoviruses infect both humans and animals. The group includes well-known human pathogens such as parainfluenza viruses of which several types are known, measles virus, mumps virus,and respiratory syncytial virus. Among the animal-pathogenic paramyxoviruses are canine distemper virus, rinderpest virus, and Newcastle disease virus. The paramyxoviruses are single-stranded RNA viruses. The virus particles have an outer membrane with short projecting spikes, and an inner nucleocapsid. The spikes include two glycoproteins of relevance, a hemagglutinin with neuraminidase activity (HN) and a cell fusion-inducing protein (F).
Pathogenesis and virus spread through the infected host are both facilitated by virus-mediated cell fusion. Recent work has demonstrated that both HN and F are required for cell fusion to occur [Hu, X. et al. (1992) J. Virol. 66:1528-1534; Morrison et al. (1991), J. Virol. 65:813-822; Katzir, Z. et al. (1989) Biochemistry 28:6400-6405; Ebata, S. N. et al. (1991) Virology 183:437-441]. The genes encoding HN and F, respectively, have been cloned from several virus types, including Sendai (SN), simian virus 5 (SV5) parainfluenza type 1 (PI1), parainfluenza type 2 (PI2), and parainfluenza type 3 (PI3). The respective proteins have been purified and various polyclonal and monoclonal antibodies against them have been obtained. Substantial sequence differences exist between the respective HN and F proteins of SN, PI1, PI2, and PI3, although the greatest sequence similarities exist between SN and PI1 [Matsuoka, Y. et al. (1990) Virus Res. 16:107-114; Blumberg, B. et al. (1985) Cell 41:269-278; Elango, N. et al. (1986) J. Virol. 57:481-489; Merson, J. R. et al. (1988) Virology 167:97-105; Millar, N. et al. (1986) J. Gen. Virol. 67:1917-1927]. HN and F can be expressed either individually or together in HeLa-T4 cells using a vaccinia-T7 recombinant virus vector system [Fuerst, et al. (1986) Proc. Nat'l. Acad. Sci. USA 83:8122-8126].
Various experimental approaches have been taken in an attempt to further define the respective roles of HN and F in cell fusion. The possibility of direct association of HN and F has been considered, however, the evidence to date has been negative. Initial studies by Markwell, M. A. K. and Fox, C. F. (1980) J. Virol. 33:152-166 demonstrated that HN of Sendai and Newcastle disease viruses exists as a disulfide linked homodimer. The authors also studied spatial relationships of viral proteins by observing the effects of chemical cross-linking reagents on proteins of the mature virus. Interactions up to 1.1 nm distance were expected to be detectable. The NP (nucleoprotein) and M (membrane) proteins were cross-linked by the reagents. Also the F protein could be cross-linked to itself, suggesting that F exists in the native virus as noncovalent homo-oligomers. No close association between HN and F was observed in the cross-linking studies. Sechoy, O. et al. (1987) J. Biol. Chem. 262:11519-11523 extended the previous study with respect to F, using a system of purified F in reconstituted lipid vesicles. The authors concluded that native F exists as a noncovalent association of homo-tetramers, each consisting of two peptides, F.sub.1 and F.sub.2, linked by a disulfide bond. (F.sub.1 and F.sub.2 are formed by posttranslational cleavage of a precursor, F.sub.0 [Scheid, A. and Choppin, P. W. (1974) Virology 57:475-490; Homma, M. and Ohuchi, M. (1973) J. Virol. 12:1457-1465]. Citovsky, V. et al. (1986) J. Biol. Chem. 261:2235-2239 studied HN and F in reconstituted lipid vesicles by circular dichroism. HN and F were purified from detergent-extracted virus particles. The circular dichroism spectrum of vesicles reconstituted with both HN and F was shifted with respect to the spectra of vesicles reconstituted with HN or F alone. The authors inferred a changed conformation of the glycoproteins of the HN-F vesicles, possibly as the result of action of HN upon F. Nakanishi, M. et al. (1982) Exp. Cell Res. 142:95-101 also studied purified HN and F in reconstituted vesicles, using diphtheria toxin A fragment to confer cytotoxicity on the vesicles and varying proportions of HN and F. Optimum cytotoxicity of the vesicles was observed when the ratio of F to HN was two. Roux, L. (1990) Virology 175:161-166 studied the interaction of HN and F with the immunoglobulin heavy chain binding protein (BiP) by measuring the amount of HN or F precipitated by a monoclonal antibody to BiP. BiP belongs to the heat shock family. It is proposed to function as a "chaperon" protein to aid in maintaining proper folding of other nascent proteins during maturation. Anti-BiP antibody was found to precipitate HN in infected cells, along with BiP, however, only one-fifth as much F was coprecipitated as HN. Katzir, Z. et al. (1989) Biochemistry 28:6400-6405 set out specifically to test whether F-HN complexes are formed in infected cell membranes. The authors measured fluorescence photobleaching recovery as a measure of each component's lateral mobility in the membrane. One component was immobilized with an antigen and the effect on mobility of the other, labeled with a fluorophore, was investigated. The authors did not find any effect of HN on lateral mobility of F, or vice-versa, suggesting that no HN-F complexes formed. For a general review, see Morrison T. and Portner, A. (1991) Structure, function and processing of the glycoproteins of Paramvxoviridae, In The Paramyxoviruses (D. W. Kingsbury ed.) Plenum Publishing Co., New York, pp. 347-382.

Anti-paramyxovirus screening method and vaccine


A concentrated photovoltaic cell comprises a semiconductor stack comprising an upper surface and a lower surface opposite to the upper surface, wherein the upper surface is operable to absorb a light which comprises a light intensity distribution on the upper surface; and an upper electrode formed on the upper surface of the semiconductor stack and comprising an electrode pattern approximately corresponding to the light intensity distribution, wherein the light intensity distribution comprises a high light-concentrated area having a first light intensity and a low light-concentrated area having a second light intensity, wherein the second light intensity is lower than the first light intensity.

Due to the shortage of fossil energy, countries around the world are aware of the importance of the environmental protection. In recent years, alternative energy and renewable energy technologies are developed, wherein the photovoltaic cell gets much attention. The photovoltaic cell directly converts solar energy into electrical energy. When the amount of the sunlight is greater and the concentration magnification of the concentrator module is higher, the electricity per unit area of the photovoltaic cell is higher and the cost of power generation of the photovoltaic cell is cheaper.
The conversion efficiency of the photovoltaic cells differs when that material of the photovoltaic cells changes. For example, the conversion efficiency of a silicon-based photovoltaic cell is about 12%˜20%, and the conversion efficiency of a group III-V based photovoltaic cell is about 31%˜41%. The silicon material absorbs energy having wavelength between 400 nm and 1100 nm, and the group III-V material absorbs energy having wavelength between 300 nm and 1900 nm. The conversion efficiency of the group III-V based photoelectric cell is higher compared with the silicon-based photovoltaic cell.
The concentrated photovoltaic cell generates power by focusing the sunlight on the group III-V based photovoltaic cell through optical concentrator so the power generation efficiency of the photovoltaic cell is increased and the cost of the power generation is reduced. Taking a photovoltaic cell having a size of 4 inches wafer and comprising group III-V based material for example, it can produce 2.4 W power under one sun without the optical concentrator and 650 W power under 500 suns with the optical concentrator. In this case, the concentration magnification of the optical concentrator is expressed with multiples of sun. For example, 500 suns expressed that the light intensity received by the photovoltaic cell with the optical concentrator is 500 times of that without the optical concentrator under the same unit area.
FIG. 1 illustrates a diagram of a conventional concentrator module 1. The concentrator module 1 comprises a first optical concentrator 13, a second optical concentrator 11 and a photovoltaic cell 10. FIG. 2 illustrates a top-viewed diagram of the photovoltaic cell 10. The photovoltaic cell 10 comprises a plurality of collector electrodes 102 and a plurality of grid electrodes 103 formed on an upper surface 101. FIG. 2A illustrates a partial enlargement of a top-viewed diagram of the plurality of grid electrodes 103. Each of the plurality of grid electrodes 103 comprises a same width w, and a spacing s between adjacent grid electrodes 103 is the same. A pitch d between the first grid electrode 103a and the second grid electrode 103b is the sum of the width w and the spacing s. As shown in FIG. 2A, the pitch d between the plurality of grid electrodes 103 is the same.
The first optical concentrator 13 and the second optical concentrator 11 focuses a sunlight 12 on the upper surface 101 of the photovoltaic cell 10 with high concentration magnification, which achieves higher photoelectric conversion efficiency, provides higher power generation and reduces the costs of power generation. However, the light concentration of the conventional concentrator module 1 is uneven. When the sunlight 12 is incident on the upper surface 101 of the photovoltaic cell 10, the light intensity distribution of the sunlight 12 is uneven on the upper surface 101, which leads to higher resistance of the photovoltaic cell 10 and reduces the power generation efficiency of the photovoltaic cell 10 as a whole. FIG. 3 illustrates an example of the conventional photovoltaic cell 10 with the conventional concentrator module 1. FIG. 3 shows an example of the photovoltaic cell 10 having a size 5 mm×5 mm. The photovoltaic cell 10 receives the sunlight 12 from the concentrator module 1. Within a radius of 1 mm from the center of the upper surface 101 of the photovoltaic cell 10, the concentration magnification of the sunlight 12 incident on the photovoltaic cell 10 by the first optical concentrator 13 and the second optical concentrator 11 is more than 1000 suns. Beyond a radius of 1 mm from the center of the upper surface 101 of the photovoltaic cell 10, the concentration magnification of the sunlight 12 incident on the photovoltaic cell 10 by the first optical concentrator 13 and the second optical concentrator 11 is less than 200 suns.


An areal security element for security papers, documents of value and the like, with at least two different areal security features (20, 22) disposed one above the other, which each are recognizable upon viewing the security element from opposite sides, wherein the first security feature (20) has a thin-layer element (42) provided with an areal diffraction structure (46) with color shift effect, which contains a first reflection layer (40) with a first color effect, the second security feature (22) has a second reflection layer (44) provided with an areal diffraction structure (46) for producing a diffractive image, which shows a second color effect different from the first color effect, and wherein the thin-layer element (42) of the first security feature (20) and the reflection layer (44) of the second security feature (22) are provided with the same areal diffraction structure (46).

1. Field of the Invention
The invention relates to an areal security element for security papers, documents of value and the like, a security paper for producing security documents and a document of value with such a security element. The invention in particular relates to a security element in the form of a security stripe or a security thread for embedding in a security document, or in the form of a transfer element for applying onto a security paper, document of value and the like.
2. Description of the Background Art
From the print WO 95/10420 a document of value is known, into which a through opening is punched subsequently to its manufacturing, which then on one side is closed with a cover foil that projects beyond the opening on all sides. Since the cover foil at least in partial areas is transparent, when copying the document of value the background will be visible and thus renders the copy recognizable as such. The cover foil can also have a security feature, such as a hologram.
A general problem with the manufacturing of security elements is that standard holograms with an aluminum reflection layer become more and more freely available. For that reason the security standard of such standard holograms and of the documents of value equipped therewith diminishes more and more.

Areal security element and method for producing it


A fixed weight/permanent weight-lifting apparatus including dumbbells and barbells is provided that have an elongated shaft with a centrally located handle and ends having a smaller diameter than the handle, a pre-coated head welded to each of these ends with a centrally located channel connecting its inner face and its outer face and a counter bore machined into its outer face of the head forming an inner collar. The channel has a diameter sufficient to receive the portion of the end of said shaft that extends beyond the inner collar. Each of the inner collars are joined to that portion of the end that extends beyond the collar.

1. Field of Invention
The present invention relates generally to weight-lifting apparatus and a method of assembling the apparatus, and more particularly to fixed dumbbells and barbells and their assembly for use in light duty environments.
2. Description of the Prior Art
Fixed dumbbells and barbells are those in which the individual weights or heads cannot be added or subtracted by the user, but are permanently fixed to each end of a handle or bar. Fixed dumbbells and barbells have long been used in heavy duty environments such as health clubs, gyms, schools, and YMCAs. In recent years, a fast growing segment of the fitness industry has seen dumbbells and barbells used in light duty environments such as hotels, corporate fitness centers, personal training centers, homes and the like. The goal in assembling such weight-lifting equipment is to employ assembly techniques that coat the heads with a durable, pliable coating and still maintain the overall cost of the equipment used in light duty environments significantly less than the cost for equipment used in heavy environments.
Wilkerson, U.S. Pat. No. 4,538,806 discloses and claims a method of assembling barbells having heads covered with a pliable coating and filled with, for example, metal shot.
There is a need for more efficient and less costly methods for assembling dumbbells and barbells having coated heads than those presently taught in the prior art.

Weight-lifting apparatus and method of assembling same


Various embodiments of child-resistant containers are described which have a release that requires two portions to be simultaneously depressed in order to release the latches and allow the container to be opened. The release may be formed of two pieces coupled to each other and the housing by a fastener, where one of the pieces acts as a spring to bias the release or its actuators in a first position.

The following description includes information that may be useful in understanding the present invention. It is not an admission that any of the information provided herein is prior art or relevant to the presently claimed invention, or that any publication specifically or implicitly referenced is prior art.
Various types of containers exist in the prior art. While certain of the containers have been designed to be child-resistant, such containers typically comprise a pill bottle for storing medications. These containers can be problematic for storing non-pill products, and the child proof functionality often does not lend itself to other shapes of containers.
Furthermore, many existing child-resistant containers comprise complex mechanisms to release and latch the container that are composed of many parts. This increases the overall cost to manufacture and assemble and can lead to failure of the mechanism over time.
All publications identified herein are incorporated by reference to the same extent as if each individual publication or patent application were specifically and individually indicated to be incorporated by reference. Where a definition or use of a term in an incorporated reference is inconsistent or contrary to the definition of that term provided herein, the definition of that term provided herein applies and the definition of that term in the reference does not apply.
Thus, there is still a need for simplified child-resistant containers having a non-cylindrical shape.


An analog discrete current mode negative feedback amplifier circuit for use with a micro-fused strain gauge is disclosed. The amplifier circuit includes a Wheatstone bridge coupled to a first power supply and a second power supply. The first power supply and the second power supply can be configured such that the periodically alternate between two voltage levels. The Wheatstone bridge can be coupled to a negative feedback amplifier circuit with common mode detection. The amplifier circuit can comprise a differential amplifier with a negative feedback configuration coupled to a common mode amplifier. In addition, the output of each of the amplifiers can be coupled to a common-mode amplifier. In a pressure sensing application, the output of the common mode amplifier serves to output the temperature while the differential amplifiers serve to output the pressure.

A traditional electronic pressure sensor is a transducer that generates a signal as a function of the pressure imposed on the transducer. A pressure sensor is used in many different everyday applications to measure pressure of a gas or a liquid. The measurement of pressure can be of great importance in many different areas.


An optical device includes a substrate. The substrate includes a first optical waveguide component having a first optical waveguide, a pair of first projections in which a gap between side faces thereof varies in a direction along an optical axis of the first optical waveguide, and a first pattern. The optical device includes a second optical waveguide component. The second optical waveguide component includes a second optical waveguide, at least one pair of second projections, and a second pattern. The second pattern and the first pattern are soldered to each other and side faces of the at least one pair of second projections are in contact with the side faces of the pair of first projections, respectively.

In recent years, hybrid mounting has been drawing attention in the optical transmission field, which enables a plurality of components each including an optical waveguide to be mounted on one substrate. For an optical device employing this hybrid mounting, a method called self-alignment is sometimes used, in which optical waveguides are optically coupled to each other using surface tension of solder.
That is, in the self-alignment, a substrate on which a first optical waveguide component having an optical waveguide is mounted and a second optical waveguide component having an optical waveguide are prepared, an electrode pattern of the substrate and an electrode pattern of the second optical waveguide component are arranged to be opposed to each other, and the opposed electrode patterns are connected to each other with solder. When the solder between the opposed electrode patterns is melted, a force based on surface tension of the solder between the opposed electrode patterns is applied from the solder to the electrode pattern of the second optical waveguide component, and the second optical waveguide component moves on the substrate in accordance with the force applied to the electrode pattern. As a result, an end face of the optical waveguide of the second optical waveguide component approaches an end face of the optical waveguide of the first optical waveguide component, so that the optical waveguide of the second optical waveguide component is optically coupled to the optical waveguide of the first optical waveguide component.
In this self-alignment, when relative positions of the optical waveguides are displaced from each other, accuracy of optical coupling is lowered and coupling loss increases. Therefore, it is desirable to accurately position a component that is to move on the substrate (that is, the second optical waveguide component) when soldering is performed. Hence, it has been studied that, when two optical waveguides are optically coupled to each other in association with soldering, a side face of a projection provided on the second optical waveguide component is brought into contact with a side face of a projection provided on the substrate, thereby aligning the second optical waveguide component with an appropriate position on the substrate.
However, in a case of simply bringing the side faces of the projections into contact with each other, the second optical waveguide component may be displaced from the appropriate position because there may be individual differences in size of the projections for some manufacturing accuracies of the projections. If the second optical waveguide component is displaced from the appropriate position, the relative positions of the optical waveguide of the second optical waveguide component and the optical waveguide of the first optical waveguide component are displaced from each other, which may result in lowering of accuracy of optical coupling.

Optical device and manufacturing method of optical device


A technique for generating a phase modulated, constant envelope transmit signal having continuous phase transitions between adjacent symbols involves generating a stream of symbols having a sequence of symbol states and identifying antipodal phase transitions between adjacent symbols representing different states. The phase rotation polarity of the continuous phase transitions between adjacent symbols having antipodal phase transitions is controlled such that a phase rotation polarity pattern of the antipodal phase transitions in the stream of symbols is uncorrelated with the sequence of symbol states. The constant envelope transmit signal is then generated from the stream of symbols. The spectral profile of the constant envelope signal can be tailored by selection of the phase trajectory shape and the portion of the symbol over which the continuous phase transition occurs.

Many wireless applications utilize saturated radio RF amplifiers because of their high efficiency. Constant envelope modulations are used with saturated amplifiers because the transmitted information resides in the phase domain, which is relatively unaffected by the nonlinearity of the saturated amplifier. Minimum Shift Keying (MSK), Gaussian MSK (GMSK), and other forms of continuous phase modulation (CPM) can be applied to navigational signals to obtain reduced bandwidth while maintaining a constant envelope. For example, in MSK, a linear ramp phase transition is used instead of an instantaneous phase transition.
Two conventional phase shift keying cases to consider are: binary antipodal schemes, e.g., Binary Phase Shift Keying (BPSK); and M-ary PSK. With BPSK, if a single pseudo-noise (PN) code is to be transmitted on a single quadrature channel, then a binary form of MSK can be used to transmit the code. In this case, two phases that are 180° apart (i.e., "antipodal" phase states) can be used to respectively represent the binary signal states. When using continuous phase transitions between symbols instead of instantaneous phase transitions, there are two equivalent phase trajectory option available to perform the 180° phase transition between the two antipodal phases: proceed in the counterclockwise direction for a total phase change of 180° (a positive phase shift ramp); or proceed in the clockwise direction for a total phase change of −180° (a negative phase shift ramp).
These two options are illustrated in FIG. 1 for the case where the current symbol is a logical '1' at a phase of 0° and the next symbol is a logical '0' at a phase of 180°. Note that an instantaneous phase change would proceed directly back and forth along the I-axis in the example shown in FIG. 1 (i.e., there is no rotational polarity); thus, the issue of phase rotation polarity exists in the context of phase modulated signals that employ continuous phase transitions between adjacent symbols. Either rotational polarity will result in the same end phases and both are identical in terms of the PN code and the CPM. Thus, the continuous phase modulation leaves an ambiguity with respect to the polarity of the phase ramps when transitioning between antipodal phase states.
A CPM BPSK phase trajectory with a fixed, positive phase transition rotational polarity is illustrated in FIG. 2. In this case, the symbol vector proceeds with a counterclockwise phase rotation around the unit circle for all phase transitions, i.e., from a logical '1' at 0° to a logical '0' at 180° and from a logical '0' at 180° to a logical '1' at 0°.
Alternating phase transition rotational polarity per transition is shown in FIG. 3. In this case, the symbol vector rotates back and forth from a logical '1' at 0° to a logical '0' at 180°. That is, a counterclockwise (positive) phase rotation is used to transition from the '1' phase state to the '0' phase state, while a clockwise (negative) phase rotation is used to transition from the '0' phase state to the '1' phase state. Note that in this scheme, the phase trajectory always stays in the upper half plane (as illustrated) or, alternatively, in the lower half plane if the opposite polarity convention is used (not shown). The quadrature (Q) component of the signal will therefore always be either positive (as shown in FIG. 3) or negative (in the convention not shown) for all phase transitions.
The problem with a fixed rotation polarity such as that shown in FIG. 2 is that, if the same direction of rotation is used for all phase transitions, there is a net frequency shift which skews the frequency spectrum of the signal. There is also spreading of the PN symbol information into the next symbol which results in a widening of the autocorrelation peak and a filling in of the side nulls. This degrades the time acquisition performance of the user receiver, which is particularly troublesome for receivers required to accurately determine the arrival time of signals, such as in the case of navigation signals. These effects are shown in FIGS. 4 and 5 for a CPM version of a BOC(10,5) (Binary Offset Carrier) waveform with a fixed phase rotation direction. Specifically, FIG. 4 illustrates the correlation function degradation of a BOC(10,5) CPM signal with a fixed phase rotation direction (solid line) relative to a conventional BOC(10,5) correlation function with instantaneous phase transitions (dashed line). FIG. 5 illustrates the power spectrum of a CPM BOC(10,5) with a fixed phase rotation direction (solid line) relative to the conventional BOC(10,5) power spectrum with instantaneous phase transitions (dashed line). Note that, as expected the CPM achieves a significantly narrower spectrum than the BOC(10,5) waveform with instantaneous phase transitions. However, the fixed phase rotation of the CPM signal results in the overall spectrum being asymmetric, shifted to the right (positive frequency direction).
The problem with alternating rotation with each transition, as shown in FIG. 3, is that, since the Q component has the same polarity for all transitions, the Q component can be looked at as a stream of pulses with a fixed polarity pulse occurring during each phase transition. This generates unwanted clock spurs and also degrades the null depth of the autocorrelation function.
Thus, when conventional CPM methods (including MSK, GMSK, etc.) are used, the waveform may be bandwidth efficient but will have a distorted correlation function when detected in the user receiver. The correlations nulls that occur at integer chip spacings will not be as deep as for a conventional signal, as shown in FIG. 4. As a result of the decreased null depths, time accuracy estimates in the receiver (Early/Late) detector will be degraded. This happens because some of the energy in each navigational chip is spread into the adjacent chip so that the cross-correlation of the waveform with the reference code is no longer zero at +/−half chip offset.
Where there is PN data on both the in-phase (I) and quadrature (Q) channels, such as with quadrature PSK (QPSK) or M-ary phase constellations, CPM can also be used to maintain a constant envelope. However, the possibility of +180° or −180° phase transitions between adjacent symbols still exists. Thus, M-ary PSK schemes still present the issue of ambiguity with respect to antipodal transitions and, consequently, the same problems that occur in the antipodal case. Further, in cases where there are multiple codes such that one code is on the I channel and another code is on the Q channel, or where multiple codes are combined to form an M-ary signal constellation (e.g., Phase-Optimized Constant-Envelope Transmission (POCET), Interplex, Majority Vote, or Intervote combining methods), the same issues described above for the antipodal case will arise when there are 180° phase transitions between adjacent symbols.
Conventional MSK and GMSK are defined to have the phase transition occurring over the entire symbol interval. In the case of GMSK, the waveform is further tailored through varying the BT product where B is the Gaussian filter bandwidth and T is the symbol time. However this approach offers only a limited amount of optimization of bandwidth efficiency versus correlation loss.
In conventional CPM systems, the spectrum is controlled only by the modulation parameters. In the case of GMSK, for example, this would be the BT product. This does not always offer a system designer enough options for designing a waveform that meets desired spectral requirements.

Bandwidth efficient continuous phase modulation


A projectile for an armored target is characterized in that the body consists of a block of dense material with a frustoconical axial bore, its large end opening in the plane of face of the body. The transmission element consists of a rod extending from the rear face of the head and comprises a seating engaged in the bore to leave a clearance between the rear face of the head and the front face, and the body and the nose are locked in a position where the rod is engaged in the bore. Application to armor-piercing fragmentation shells.

The present invention relates to projectiles of the subcaliber type and relates more specifically to such projectiles designed to possess high firing accuracy and substantial terminal efficiency.
Destruction of a target at a relatively long distance may be attempted from a projectile that explodes on impact.
While such projectiles have a substantial terminal efficiency, on the other hand, they have relative accuracy due to their design.
Moreover, such projectiles are expensive and require very strict storage and handling precautions to prevent their unintended explosion.

Armor-piercing fragmentation subcaliber projectile