A power converting apparatus and a power converting method for driving a high voltage AC motor at a variable speed. Conventional invertor systems cannot solve technical subjects such as energy conservation, resource conservation, miniaturization, efficiency promotion and voltage and current waveform distortion suppression for improvement in environment needed by the market, and cannot solve another technical subject of improvement in redundancy such that, upon failure, operation is performed with a normal part. In the present invention, a power converting apparatus of a multiple three-phase pulse width modulation cycloconverter system for driving a high voltage AC motor at a variable speed is used, and bidirectional semiconductor switches are controlled so that voltages of AC outputs to be outputted to single-phase AC terminals of three-phase/single-phase pulse width modulation cycloconverters may have a same phase in each of units but electric angles of basic wave voltage phases may be different by 120 degrees from each other among the three units to drive a high voltage AC motor. If one of the cycloconverters fails, then the single-phase AC terminals of the failed cycloconverter are short-circuited and three sets of switches each consisting of two bidirectional semiconductor switches connected to three-phase AC terminals of the cycloconverters of the other two units in the same group as the failed cycloconverter are successively rendered conducting one by one set at equal time intervals to short-circuit the three sets of the bidirectional semiconductor switches to drive the high voltage AC motor at a variable speed.

1. Field of the Invention
This invention relates to a power converting apparatus and a power converting method for driving a medium to high voltage AC motor at a variable speed, and particularly to a power converting apparatus and a power converting method of a pulse width modulation (PWM) controlling system.
2. Discussion of the Background
Conventionally, for variable speed driving of a high voltage AC motor, a system which employs a high voltage invertor or another system wherein a step-down transformer and a step-up transformer are connected to the input side and the output side of a low voltage invertor to drive the high voltage AC motor is employed.
FIG. 6 is a circuit diagram of a driving circuit which employs a high voltage invertor of a conventional example, and FIG. 7 is a concept diagram illustrating four quadrature operation based on the relationship between the torque and the speed of a motor. In FIG. 6, reference symbol 10 denotes a high voltage AC motor of an object of driving, 101 an invertor unit, 102 a smoothing capacitor unit, 103 a regenerative converter unit, 104A and 104B each denotes an AC reactor, and 105 denotes a three-phase transformer.
The invertor unit 101 includes three-level invertors of the neutral clamping type and employs, for power elements, a GTO (Gate Turn Off Thyristor, hereinafter referred to simply as GTO) to assure a high withstanding voltage for the elements. The power elements are connected in series to achieve divisional sharing of a voltage, and variable voltage variable frequency (VVVF) power is supplied from a high voltage DC power supply formed from the smoothing capacitor unit 102 to the invertor unit 101. In order to keep divisional sharing of a voltage of the GTOS, well known snubber circuits must be installed individually. In the converter unit which supplies a DC voltage to the smoothing capacitor unit 102, the capacity of the high voltage invertors is generally as high as several hundreds kW or more, and the construction of the regenerative converter unit 103 is used for damping energy processing upon deceleration or for four quadrature operation (forward driving, reverse driving, forward regeneration and reverse regeneration) illustrated in FIG. 7. In FIG. 6, two circuits each composed of a combination of thyristors and GTOs are used in series connection, and control between driving and regeneration is performed depending upon the direction of DC power. The regenerative converter unit 103 is connected to secondary windings of the three-phase transformer 105 through the AC reactors 104A and 104B while primary windings of the three-phase transformer 105 are connected to a high voltage commercial power supply so as to receive supply of power.
FIG. 8 is a circuit diagram showing a driving circuit which employs a low voltage invertor of a conventional example. In FIG. 8, reference numeral 10 denotes a high voltage AC motor of an object of driving, 106 an invertor unit, 107 a smoothing capacitor unit, 108 a regenerative converter unit, 109 an AC reactor, 110 a step-down transformer, and 111 a step-up transformer.
The invertor unit 106 includes IGBTs (Insulated Gate Bipolar Transistors, hereinafter referred to simply as IGBTs) and diodes connected in a three-phase bridge circuit and is pulse width modulation (hereinafter referred to simply as PWM) controlled so that it may output a voltage and a frequency necessary to drive the motor 10 through the step-up transformer 111. Since the invertor unit 106 is a low voltage invertor, it is connected to the high voltage AC motor 10 through the step-up transformer 111. Also the regenerative converter unit 108 is composed of IGBTs and diodes connected in a three-phase bridge circuit similarly as in the invertor unit 106, and is connected to secondary windings of the step-down transformer 110 through the AC reactor 109 while primary windings of the step-down transformer 110 are connected to a high voltage commercial power supply so as to receive supply of power. Meanwhile, also DC buses of the regenerative converter unit 108 and the invertor unit 106 are connected to each other through the smoothing capacitor unit 107. Both of the invertor unit 106 and the regenerative converter unit 108 are PWM controlled.
As other motor driving systems, for example, a multiple cycloconverter recited in "Cycloconverter Apparatus" disclosed in Japanese Patent Laid-Open Application No. Heisei 6-245511 and a PWM cycloconverter recited in "Power Converting Apparatus of a Pulse Width Controlling System" disclosed in Japanese Patent Publication Application No. Heisel 7-44834 are known. However, they are not directed to driving of a high voltage AC motor described above.
Meanwhile, the trend of the world is directed to energy conservation, resource conservation, minimum size, high efficiency and low-distortion voltage and current waveform for improvement in environment, and due to complication of application systems, improvement in operation reliability such as improvement in regard to redundancy is required. Also the motor driving systems of the prior art described above naturally become an object of such improvement.
However, from the point of view of energy conservation, resource conservation, minimum size, high efficiency and low-distortion voltage and current waveform for improvement in environment, both of the high voltage invertor system and the low voltage inverter system of the prior art examples described above have the following problems.
In the case of the high voltage invertor system of FIG. 6, a GTO is employed for main circuit elements in order to achieve a high voltage withstanding property. Since a GTO is not a high speed switching element, it is difficult to use a high carrier frequency, and reduction in noise in inverter driving or suppression of waveform distortion cannot be anticipated. Further, since a snubber circuit of a GTO repeats charging and discharging each time switching is performed, also the loss is high, and since it has a circuit construction which employs a high voltage element, assurance of insulation for a main circuit, a bus bar and so forth is required and the snubber circuit is not suitable for minimizing the inverter package. Furthermore, since a GTO driving power supply is required for each GTO and besides a high voltage is applied between control power supplies, it is not easy to generate the control power supplies, and this is a bottle neck to minimize the inverter package.
Meanwhile, in the case of the step-up system of the low voltage inverter of FIG. 8, since it is an IGBT invertor of a low voltage, while high fr equency PWM control is possible and reduction in noise can be anticipated, in order to achieve a large capacity, parallel connection of IGBTs is required, and a countermeasure for parallel balancing and a snubber circuit are required and minimum size is difficult. Further, also an increase in loss of IGBTs, bus bars and snubber circuits arising from high current is estimated, and minimization is difficult also from the phase of cooling. Furthermore, where step-up is performed by a transformer as seen in FIG. 8, since the switching speed of IGBTs is high, that is, since dV/dt upon switching is large, also there is another drawback that, by inductances of wiring lines, floating capacitances of the wiring lines, inductances of a transformer and so forth, a resonance voltage is generated in synchronism with switching of PWM control of an invertor, causing dielectric breakdown of the motor. As a countermeasure against the drawback, it has been proposed to insert a filter between the invertor unit 106 and the step-up transformer 111 of FIG. 8 as recited in "Output Filter Circuit of Voltage Type PWM Invertor" disclosed in Japanese Patent Laid-Open Application No. Heisei 1-72144. In addition, since, upon low frequency operation, the voltage/frequency ratio to be provided to the transformer is set to 1.5 to 2 times that in the proximity of a rated frequency by an invertor in order to assure starting torque, there is another problem that a larger transformer than a transformer for a commercial frequency is required so that magnetic saturation may not occur. Further, if the invertor 106 generates an offset voltage due to a dispersion in switching characteristic of the IGBTs and so forth, then since a DC voltage is applied to the step-up transformer 111, magnetic saturation occurs. Consequently, there is a problem also that excessive current flows.
As a countermeasure against harmonic distortion of an output voltage or current, while the high voltage invertor is controlled by 3-level control and both of PWM control and amplitude control are used, the low voltage invertor system employs only PWM control and exhibits large harmonic distortion. Also for the power supply voltage, since the regenerative converter unit 103 of the high voltage invertor system of FIG. 6 uses 120-degree energization waveforms, low order harmonic distortion remains, and with the low voltage invertor system of FIG. 8, since the regenerative converter unit 108 performs PWM control, although low order harmonics of power supply current are suppressed, high order harmonics remain.
As described above, the conventional invertor systems cannot solve the technical subjects such as energy conservation, resource conservation, minimum size, high efficiency and low-distortion voltage and current waveform for improvement in environment needed by the market. Further, any of the systems cannot solve the technical subject of improvement in redundancy such that, upon failure, operation is performed with a normal part.
Further, of the systems other than the invertor systems, the cycloconverter recited in "Cycloconverter Apparatus" disclosed in Japanese Patent Laid-Open Application No. Heisei 6-245511 cannot raise, since it is of the power supply commutation system, the output frequency only up to 1/3 to 1/2 the power supply frequency. Consequently, the cycloconverter is not suitable for motor driving.
An improvement of the cycloconverter just described is a PWM cycloconverter recited in "Power Converting Apparatus of the Pulse Width Controlling System" disclosed in Japanese Patent Laid-Open Application No. Heisei 7-44834. The PWM cycloconverter has the following characteristics.
1) Miniaturization is easy because it does not require such a DC circuit as is required by an invertor system.
2) The element loss is low and the efficiency is high because the number of elements inserted in series in a route from a power supply to a load is smaller than that of an invertor system.
3) Four quadrature operation is easy because direct AC-AC conversion is used.
However, since also this system is a PWM control power converting system of three-phase inputs and three-phase outputs, although low order harmonics of power supply current are suppressed, high order harmonics remain, and the technical subject of voltage and current waveform distortion suppression cannot be solved for both of the input and the output. Further, in order to drive a high voltage AC motor, a system wherein a power element is so formed as to have a high voltage withstanding property to make a high voltage PWM cycloconverter or a voltage is raised by a transformer is adopted, and the same subjects as those of the high voltage inverter system or the transformer step-up system of a low voltage invertor occur. Furthermore, in the conventional examples described above, all of the systems have a problem that, if some function is damaged, then operation cannot be continued.

Power converting apparatus and method using a multiple three-phase PWM cycloconverter system


A gas powered, pressureless steam cooker has a connector tube for ingress of steam into the cooking compartment so that entry of a low temperature, wet steam can be minimized. The connector tube has a small diameter to constrain the entry of steam into the cooker compartment until it has reached a minimum velocity. A water reservoir is located alongside the oven compartment to the rear of the cooker and an integrated heat exchange tube and firebox are located in the tank. This tube is generally U-shaped, a lower horizontal leg opening at a mouth outside the water reservoir. An inshot burner fires into the lower horizontal leg and combustion gas flow through the U tube giving up heat to water in the reservoir. Additional heat exchange surface may be provided by channels through an upper leg of the U-tube from side to side. Flow of combustion gases may be optimized by the shape of the U-tube and at least a deflecting surface between a vertical web of the U and the upper leg of the U.

1. Field of the Invention
This invention relates to a gas powered pressureless steam cooker, especially to a gas powered steamer for commercial or restaurant use. The term "pressureless steam cooker" is intended to mean a steam cooker in which no great pressure of steam is developed although minor pressure of say 1/2 lb. per square inch may be permissible.
2. Acknowledgement of Prior Art
Steam cookers conventionally cook foods by the use of hot steam and may be powered by a steam generator which is either gas fired or electrical fired. No electric heating elements are present in the cooking compartment itself. For gas fired units, there has always been an ongoing problem in providing steam at a suitable temperature inside the cooking compartment without the loss of efficiency, e.g. heat in delivering the steam to the cooking compartment. For pressureless steam cookers steam is generally supplied into the cooking compartment through an open vent thus allowing ingress of wet steam at low temperatures.
The common design of a steam generator, or heat exchanger, up to today utilizes an atmospheric burner heating the outside of a steam generator. Atmospheric style burners are generally larger and have low efficiency due to their operating characteristics of burning in the open atmosphere. Usually such steam generators are located below the oven compartment but in some cases oven compartments are side-by-side with steam generators.
Examples of steam ovens are shown in:
______________________________________ U.S. Pat. No. 4,281,636 Vegh et al. Aug. 4, 1981 U.S. Pat. No. 4,552,123 Birkner Nov. 12, 1985 U.S. Pat. No. 3,820,524 Buckell June 28, 1974 U.S. Pat. No. 4,460,822 Alden et al. July 17, 1984 U.S. Pat. No. 5,549,038 Kolvites Aug. 27, 1996 U.S. Pat. No. 5,732,614 Oslin Mar. 31, 1998 U.S. Pat. No. 762,590 Lawrence June 14, 1904 U.S. Pat. No. 1,226,044 Wilson May 15, 1917 U.S. Pat. No. 113,241 Ashcroft April 4, 1871 U.S. Pat. No. 2,444,587 Young et al. July 6, 1948 U.S. Pat. No. 5,178,125 Kuen Jan. 12, 1993 ______________________________________


Thermoplastic polymers, rubbery polymeric components and reinforcing fillers are mixed and heated to a range of 380° F. to 440° F. The resulting mixture is then fed into a Banbury mixer, with the resulting mixture fed into a diverter feeding two molds. The velocity of the piston of each mold being filled is controlled to thereby control the density of the molded product along its length. As each mold is filled, it is deposited in a cool water bath (50° F.-60° F.) while the next mold is being filled. The molds are sequentially transferred to an air cooling rack to complete the process. A texture, generally comprising indentations perpendicular to the longitudinal axis on the molded object, provides a frictional surface between the railway crosstie and the ballast beneath the crosstie.

This invention relates, generally, to molded products made from thermo plastic materials, and specifically, to composite railroad ties and methods for making same. The invention relates more specifically to new and improved railroad ties which will last longer in hot, humid environments and to methods for making such railroad ties.


Methods and apparatuses for communicating in a cellular communications network, including provision of a user equipment comprising processing circuitry to: receive from a first cell a control signal comprising a physical Time Division Duplex Uplink-Downlink configuration indicator channel (PTCICH); decode from the PTCICH a Time Division Duplex Uplink-Downlink (TDD UL-DL) configuration; and communicate with the first cell having applied the TDD UL-DL configuration in respect of subframes associated with the first cell, wherein: the PTCICH spans less than or equal to 32 Resource Elements. Also user equipment comprising processing circuitry to: receive a control signal from a first cell; decode the control signal received from the first cell thereby to determine a second cell TDD UL-DL configuration; and communicate with the second cell having applied the second cell TDD UL-DL configuration in respect of subframes associated with the second cell.

Ever greater demand is placed on telecommunication services, which are to accommodate increasingly more efficient and effective communication for increasing numbers of fixed and mobile devices.
Conventional indication of a Time Division Duplex Uplink-Downlink (TDD UL-DL) configuration by employing the Physical Downlink Control Channel (PDCCH) lacks flexibility and can result in unnecessary User Equipment (UE) power consumption. It is therefore desirable to provide more flexible and efficient indication of the TDD UL-DL configuration.

Indication of TDD uplink and downlink configurations


An energy efficient, low distortion amplification apparatus includes: a differential amplifier having a pair of differential inputs, a pair of outputs, and a bias current control input; a digital delay having an n-bit word input and an m-bit word output; a digital-to-analog converter (DAC) having an input coupled to the output of the digital delay and a pair of differential outputs coupled to the pair of differential inputs of the differential amplifier; and a bias current controller having an input coupled to the digital delay and having an output coupled to the bias current control input of the differential amplifier.

Data Over Cable Service Interface Specification (DOCSIS) is an international telecommunications standard that permits the addition of high-speed data transfer over an existing cable TV (CATV) system. It is employed by many cable television operators to provide Internet access over their existing hybrid fiber-coaxial (HFC) infrastructure.
DOCSIS was developed by Cable Television Laboratories, Inc. (a/k/a "CableLabs") and a number of contributing companies in the late 1990's. The first specification, version 1.0, was issued in March 1997. The latest specification, DOCSIS 3.0, was released in August 2006 and significantly increased transmission speeds, both upstream and downstream, while introducing support for Internet Protocol version 6 (IPv6).
The next specification, DOCSIS 3.1, is in development and is expected to be field tested sometime in 2014. The DOCSIS 3.1 standard uses orthogonal frequency-division multiplexing (OFDM) to achieve at least 10 Gbit/s downstream and 1 Gbit/s upstream transmission rates.
With each successive specification the upstream bandwidth has increased. This requires ever-greater power to be produced by the cable modems to drive the DOCSIS upstream amplifiers which must amplify multiple digitally modulated channels simultaneously while keeping spurious distortion products very low, typically more than 50 dB below the carrier level.
To achieve the low distortion levels, power-hungry Class A amplification is typically used for DOCSIS upstream amplifiers. This is because combining multiple digitally modulated carriers creates signals with high peak-to-average ratios. Since the peaks must be faithfully reproduced to minimize bit error rate ("BER"), the amplifier must be biased to accommodate the peaks, resulting in high power dissipation. This is not desirable from a number of perspectives including power consumption, heating of the cable box, and degrading the circuitry of the cable box over time.
These and other limitations of the prior art will become apparent to those of skill in the art upon a reading of the following descriptions and a study of the several figures of the drawing.

Energy efficient, low distortion amplification apparatus


A resonator element includes: at least one resonating arm which performs flexural vibration; a base portion connected to an end of the resonating arm; and a tapered portion which is axisymmetrical with respect to a centerline which bisects the width of the resonating arm, and which has a width increasing toward a portion of the tapered portion connected to the base portion from a portion of the tapered portion connected to the resonating arm, wherein assuming that the length and width of the resonating arm are L and W and the length and width of the tapered portion are Lt and Wt, the shape of the tapered portion is controlled to satisfy a taper length occupancy η=Lt/L and a taper width occupancy ξ=2 Wt/W.

1. Technical Field
The present invention relates to a resonator element and a resonator having the resonator element.
2. Related Art
There is known a resonator element which has a resonating arm and a base portion connected to the resonating arm and in which a tapered portion symmetrical with respect to the vibration center of the resonating arm is provided in a connection portion between the resonating arm and the base portion (for example, refer to JP-A-2005-5896).
In the resonator element disclosed in JP-A-2005-5896, by providing a tapered portion in a connection portion between the resonating arm and the base portion, leakage of fundamental vibration is suppressed, the Q factor is increased accordingly so that a vibration mode is not affected, and it is also possible to prevent the resonator element from being broken by an impact or prevent deterioration of a vibration mode.
In the case where a tapered portion is provided in practice, however, the Q factor is often decreased depending on the shape of the tapered portion. Presumably, this is because a thermoelastic loss resulting from the shape of the tapered portion exceeds an increase in the Q factor resulting from a suppression effect of vibration leakage by providing the tapered portion.

This patent is subject to a terminal disclaimer.

Resonator element and resonator having a tapered arm next to the base


A plurality of flash lamps that irradiate a semiconductor wafer with flash light are arrayed in a plane. The array of the plurality of flash lamps is divided into two zones: a central zone including a region opposed to a central portion of the semiconductor wafer to be treated, and a peripheral zone outside the central zone. During flash light irradiation, an emission time of a flash lamp belonging to the peripheral zone is set to be longer than an emission time of a flash lamp belonging to the central zone. Thus, a greater amount of flash light is applied to the peripheral portion of the semiconductor wafer, where a temperature drop is relatively likely to occur, than to the central portion thereof, thus preventing a relative temperature drop in the peripheral portion of the semiconductor wafer during flash heating.

Field of the Invention
The present invention relates to a heat treatment method and a heat treatment apparatus for irradiating a thin plate-shaped precision electronic substrate (hereinafter referred to simply as a "substrate"), such as a semiconductor wafer, with flash light to heat the substrate.
Description of the Background Art
In the process of manufacturing a semiconductor device, impurity doping is an essential step for forming a pn junction in a semiconductor wafer. At present, impurity doping is typically performed by ion implantation and subsequent annealing. Ion implantation is a technique of causing ions of impurity elements such as boron (B), arsenic (As), or phosphorus (P) to collide against the semiconductor wafer with high acceleration voltage, thereby physically implanting the impurities into the semiconductor wafer. The implanted impurities are activated by annealing. At this time, if the annealing time becomes about several seconds or longer, the implanted impurities are deeply diffused by heat. This results in a junction depth much greater than an intended depth, which may constitute a hindrance to good device formation.
In recent years, attention has been given to flash lamp annealing (FLA) that is an annealing technique of heating a semiconductor wafer in an extremely short time. The flash lamp annealing is a heat treatment technique in which xenon flash lamps (the term "flash lamps" as used hereinafter refers to "xenon flash lamps") are used to irradiate a surface of a semiconductor wafer with flash light, thereby raising the temperature of only the surface of the semiconductor wafer implanted with impurities in an extremely short time (several milliseconds or less).
The xenon flash lamps have a spectral distribution of radiation ranging from ultraviolet to near-infrared regions. The wavelength of the light emitted from the xenon flash lamps is shorter than that of the light emitted from conventional halogen lamps, and approximately coincides with a fundamental absorption band of a silicon semiconductor wafer. Thus, when flash light is applied to the semiconductor wafer from the xenon flash lamps, the temperature of the semiconductor wafer can be raised rapidly, with only a small amount of light transmitted through the semiconductor wafer. It has also been turned out that flash light irradiation, that is, the irradiation of a semiconductor wafer with flash light in an extremely short time of several milliseconds or less allows a selective temperature rise only the surface of the semiconductor wafer and its vicinity. Therefore, the temperature rise in an extremely short time with the xenon flash lamps enables only the activation of impurities without deep diffusion of the impurities.
As examples of a heat treatment apparatus including such xenon flash lamps, U.S. Pat. No. 4,649,261 and US 2003/0183612 A1 disclose heat treatment apparatuses that perform desirable heat treatment with a combination of pulsed light-emitting lamps such as flash lamps that are arranged on the front side of a semiconductor wafer and continuous lighting lamps such as halogen lamps that are arranged on the rear side of the semiconductor wafer. In the heat treatment apparatuses disclosed in U.S. Pat. No. 4,649,261 and US 2003/0183612 A1, a semiconductor wafer is preheated to a certain degree of temperature with, for example, halogen lamps, and then, the temperature of the semiconductor wafer is raised to a desired treatment temperature by pulse heating with the flash lamps.
Although preheating with the halogen lamps, as disclosed in U.S. Pat. No. 4,649,261 and US 2003/0183612 A1, has a processing advantage that the temperature of a semiconductor wafer will be raised to a relatively high preheating temperature in a short time, it may cause the temperature of the peripheral portion of the wafer to be relatively lower than the temperature of the central portion of the wafer. Conceivable causes of such an uneven temperature distribution include heat radiation from the peripheral portion of the semiconductor wafer and heat conduction from the peripheral portion of the semiconductor wafer to a relatively low-temperature quartz susceptor.
If flash light is applied from flash lamps while the temperature distribution in the surface of the semiconductor wafer has been uneven during preheating, a maximum surface temperature of the semiconductor wafer also results in an uneven temperature distribution in the surface thereof. This leads to variations in the extent of the activation of impurities, also resulting in an uneven sheet resistance value.

Light-irradiation heat treatment method and heat treatment apparatus


Interleaving circuit for a multiband OFDM (orthogonal frequency division multiplexing) transceiver of a ultra wide band wireless personal access network transmitting OFDM modulated symbols, wherein each OFDM symbol consists of a predetermined number (NCBPS) of encoded bits,    said interleaving circuit comprising    a symbol interleaving unit which receives an input bitstream of encoded bits and permutes adjacent bits of said input bitstream across different OFDM symbols;    an intra-symbol tone interleaving unit which receives the bits permuted by said symbol interleaving unit and permutes adjacent bits of each OFDM symbol across uncorrelated data sub-carriers; and    an intra-symbol cyclic shift unit which shifts cyclically NCBPS bits of each OFDM symbol in response to a shift value (K) which is changed between adjacent OFDM symbols.

FIG. 1 shows the transmission of data in a wireless system according to the state of the art. Several transceivers belonging to the same wireless local area network (WLAN) use the same data transmission channel by means of time sharing. At any specific time only one transceiver is transmitting. Accordingly the transmissions from each transceiver are burst like. For helping the receiving transceiver to identify a data transmission burst and for extracting the delivered information data the transmitting transceiver sends a predefined preamble signal preceding the data portion of the data transmission burst. The transceiver that receives the data transmission burst comprises a preamble detection unit that identifies the preamble and thus identifies the data transmission burst. The transceiver uses further the preamble for estimating data transmission and channel parameters such as channel response and carrier and timing offsets that are needed for the data information extraction.
Commonly several communication networks share the same data transmission media. Specifically collocated wireless networks utilize the same frequency spectrum.
FIG. 2 shows two collocated wireless networks according to the state of the art.
Wireless local areas networks (WLAN) represent a new form of communications among personal computers or other devices that wish to deliver digital data. A wireless network is one that does not rely on cable as the communications medium. Whether twisted pair, coax, or optical fibers, hard wiring for data communication systems within a building environment is expensive and troublesome to install, maintain and to change. To avoid these disadvantages wireless networks transmit data over the air using signals that cover a broad frequency range from few MHz to a few terahertz. Depending on the frequency involved wireless networks comprise radio wireless networks, microwave wireless networks and infrared wireless networks.
Wireless networks are used mainly for connecting devices within a building or connecting portable or mobile devices to a network. Further applications are keeping mobile devices in contact with a data base and ad hoc networks for example in committee or business meetings.
Wireless local area networks (WLAN) and wireless personal area networks (WPAN) are used to convey information over relatively short ranges. A wireless personal area network (WPAN) is defined in the IEEE 802.15.3 standard.
In many situations and scenarios several wireless local area networks (WLANs) are operated simultaneously with each other in the same local area. A typical situation would be a big office wherein many office cubicles are located belonging to different divisions of the same company, e.g. search division, accounting division, marketing division. The computers of each division are connected in such a situation by means of separate wireless local area networks (WLANs). A wireless local area network (WLAN) comprising several transceivers is referred to as a Piconet.
FIG. 2 shows typical scenario where two wireless local area networks (WLANs) are operated in the same local area.
In the shown example a first transmitting transceiver A2 transmits data to a receiving transceiver A4 of the first wireless local area network WLANA on the data transmission channel of the wireless local area network WLANA. Further a transmitting transceiver B3 of the second wireless local area network WLANB transmits data to a receiving transceiver B1 of the same wireless local network WLANB on the data transmission channel of this wireless local area network. The data exchange between transceivers is performed half duplex, i.e. a transceiver can either send or receive data over a data link to another transceiver of the same wireless local area network. The data are exchanged via data packets.
Each Piconet WLANi has its respective data transmission channel, i.e. the data transmission channel is used by all transceivers of the corresponding Piconet WLANi.
In most cases the frequency resources available for a wireless local area network WLAN are bounded by regulations. Usually a certain frequency band is allocated for the wireless local networks. Within this frequency band each transceiver is required to radiate no more than a specified average power spectral density (PSD).
To operate several wireless local area networks simultaneously several proposals have been made.
In frequency division multiplexing (FDM) systems according to the state of the art the allocated frequency band is divided into several sub-frequency bands. In FDM-system each data transmission channel and consequently each Piconet is using a different frequency sub-band. Thus, data transmission in different Piconets (WLANs) can simultaneously be performed without interference.
The disadvantage of FDM-systems is that the available capacity for each Piconet is reduced compared to the case where any Piconet is allowed to use the entire allocated frequency band.
The channel capacity is given by the following formula:
  cap  =      ∫                  log                          (                      1            +                                          PSD                                                  (                  f                  )                                                            N                                                  (                  f                  )                                                              )                                  ⅆ        f
The capacity of each Piconet is larger if it will be allowed to use the full frequency band instead of just the allocated frequency sub-band The reduction in the capacity in FDM-systems translates directly to throughput reduction. Consequently the achievable data bit rate for any specific transmitter-receiver distance is reduced in FDM-systems.
In a CDMA-DSSS (Code Division Multiple Access—Direct Sequence Spread Spectrum) system according to the state of the art a direct sequence spread spectrum is used as a modulation scheme. In DSSS a sequence of many short data symbols is transmitted for each information symbol. In order to support several data transmission channels or Piconets different data sequences with low cross correlation between them are used for different data transmission channels.
In a CDMA-DSSS-system each channel can use the entire frequency band until the maximum possible throughput can be achieved. If some Piconets are working in the same area then the transmission of one Piconet is seen as additional noise by the other Piconets.
The disadvantage of the CDMA-DSSS-System is that there exists a so called near-far problem. When a transceiver in one Piconet is transmitting this transmission will be seen as additional noise by other Piconets. The level of the additional noise is proportional to the cross correlation between the spreading sequences and the received power level of the interferer's signal. For example if the interfering transceiver of Piconet A is close to a receiving transceiver of Piconet B, i.e. closer than a transmitting receiver of Piconet B then the added noise level that the receiving transceiver of Piconet B sees causes a significant reduction in the achievable bit rate for the receiver, so that even a complete blocking of the data transmission channel can occur.
A further proposal according to the state of the art to operate several wireless local area networks (WLANs) simultaneously is to use a CDMA-PH(Code Division Multiple Access—Frequency Hopping)—System. In this CDMA-PH-System the original frequency band is divided into several sub-frequency bands. Any transmitting transceiver uses a certain frequency sub-band for a certain time interval and moves then to the next frequency band. A predefined frequency hopping sequence controls the order of sub-frequency bands such that both the transmitting and receiving transceiver has the information when to switch to the next frequency and to what sub-frequency band.
In a conventional CDMA-PH-System the different data transmission channels are assigned with different frequency hopping sequences.
FIG. 3A shows a CDMA-PH-System according to the state of the art with data transmission channels. A CDMA-PH-System with four data transmission channels can operate four Piconets or wireless local area networks (WLANs) simultaneously at the same local area. In the shown example any transceiver uses a certain frequency band for a transmission interval for 242 ns, remains idle for a predetermined guard time of 70 ns and uses the next frequency band within the next transmission interval etc.
The frequency hopping sequence is fixed for any data transmission channel A, B, C, D. In the given example data transmission channel A has the frequency hopping sequence abc, channel B has the frequency hopping sequence acb, channel C has the frequency hopping sequence aabbcc and channel D has the frequency hopping sequence aaccbb.
A collision is a situation when two transceivers use the same frequency band at the same time. For example a collision between data transmission channel A and data transmission channel B occurs during the first transmission interval when both channels A, B use frequency fa and during the fourth transmission interval when both channels A, B use again frequency fa. A further collision is for example between channel B and channel D during the first transmission interval when both channels B, D use frequency a and the sixth transmission interval when both channels B, D use frequency fb.
When the frequency hopping order of two wireless networks differs two transceivers that belong to different wireless local area networks can transmit at the same time. It may happen that both transceivers use the same carrier frequency at the same time.
One possible CDMA-FH solution is based on OFDM and is called Multiband OFDM. In this case the transceiver transmits a single OFDM in one band and then hops to the next band for transmitting the next OFDM symbol. FIG. 3A depicts 6 OFDM symbols for each channel.
As shown in FIG. 3A the Multiband OFDM Transceiver performs in a time frequency interleaving (TFI) mode band-hopping wherein in each frequency band an OFDM symbol is transmitted. The band-hopping sequence is defined by a TFC code (Time frequency code) stored in a memory. Different collocated networks use different TFC codes. This enables simultaneous transmission of different networks OFDM symbols from collocated networks collide. In common scenarios the collision level enables efficient communication. Yet in some cases the collisions situation is severe and the communication is not efficient. To overcome severe collisions between transmission of different networks frequency domain separation (known as FDM) between the wireless networks can be implemented. This is achieved by adding TFC codes with constant band usage (fixed frequency bands). Accordingly a Multiband OFDM Transceiver according to the state of the art is switchable between a time frequency interleaving mode (TFI mode) and a fixed frequency interleaving mode (FFI mode). FIG. 3B shows 7 channels (7 TFC) where 4 channels are of TFI type and 3 channels are of FFI type.
As can be seen in FIG. 3B the transceiver occupies in the TFI mode three frequency bands, wherein each frequency band has a predetermined frequency bandwidth.
According to the evolving multiband OFDM standard the period of one OFDM symbol is 312.5 nSec, i.e. a data length of 242.5 nSec (128 samples at 528 Msps) and a silence time of 70 nSec (37 samples at 528 Msps) between two transmissions. Consequently the OFDM symbol rate RS=3.2 MHz=1/312.5 nSec. When using three frequency bands there are seven possible time frequency codes (TFC). The first four TFC codes define the frequency band hopping sequence when the transceiver is in the TFI mode. When the transceiver is switched to the FFI mode the transceiver transmits the signal in a fixed frequency band. As shown in the following table and in FIG. 3B, the fifth TFC code indicates that transceiver transmits a signal in a first frequency band, the sixth TFC code indicates that the transceiver transmits the signal in a second frequency band and the seventh TFC code indicates that the transceiver transmits a signal in a third frequency band.
The following TFC code have three frequency bands as summarized in the following table:
TABLE 1TFC IndexCodeType1[1, 2, 3]TFI2[1, 3, 2]TFI3[1, 1, 2, 2, 3, 3]TFI4[1, 1, 3, 3, 2, 2]TFI5[1]FFI6[2]FFI7[3]FFITFC indices 1-7 in the table corresponds to channels A-G in FIG. 3b.
A single burst of transmission is called a PLCP frame. FIG. 4 shows the data format of a PLPC frame used by a multiband OFDM transceiver. Each frame consists of a preamble, a header and a payload data section. The PLPC header is transmitted with a constant data rate of 39.4 Mbit per second whereas the payload data is transmitted with different data rates varying between 53.3 Mbit per seconds and 480 Mbit per second depending on the selected operation mode of the OFDM transceiver. The PLCP frame as shown in FIG. 4 consists of a plurality of OFDM symbols, wherein each OFDM symbol consists of a predetermined number (NCBPS) of encoded data bits. Each OFDM symbol comprises for instance 100 or 200 encoded data bits depending on the selected data rate. As can be seen from FIG. 3B each OFDM symbol is transmitted within different frequency bands fa, fb, fc according to a predetermined frequency hopping pattern. For example three frequency bands fa, fb, fc, are employed by the OFDM transceiver so that seven different frequency hopping patterns are possible as shown in FIG. 3B via a corresponding number of data transmission channels A, B, C, D, E, F, G. Each frequency band fa, fb, fc employed by the OFDM transceiver comprises a center frequency around which a predetermined number of sub-carriers or tones are provided. A frequency comprises for instance 122 sub-carriers consisting of pilot sub-carriers, guard sub-carriers and data sub-carriers. Each sub-carrier is equidistant to its neighboring sub-carrier and can be modulated separately.
For increasing the performance of the data transmission interleaving is employed by an OFDM transceiver.
In a conventional OFDM transceiver according to the state of the art the encoded bitstream is interleaved prior to modulation by a bit interleaving circuit comprising two stages. The conventional bit interleaving unit comprises a symbol interleaving unit and a tone interleaving unit.
The symbol interleaving unit permutes the bits of the received bitstream across different OFDM symbols to exploit frequency diversity across the sub-frequency bands fa, fb, fc. The tone interleaving stage of the bit interleaving circuit according to the state of the art permutes the bits received by the symbol interleaving stage across different data tones (data sub-carriers) within an OFDM symbol to exploit frequency diversity across tones so that more robustness against narrow band interference signals and against frequency-selective channels is provided.
However, the bit interleaving circuit according to the state of the art as employed in a conventional multiband OFDM transceiver has several disadvantages.
In some cases there exists a correlation between the qualities of tones, i.e. sub-carriers, from two different OFDM symbols which have the same index within the OFDM sequence.
A first case where such a correlation exists is when, as in the FFI mode, i.e. channel E, F, G as shown in FIG. 3b, and TFC index 5, 6, 7, and the table above, no frequency band hopping is performed.
A second case is as in channel C, D shown in FIG. 3b where two adjacent OFDM symbols might be transmitted in the same frequency band. For instance the first two OFDM symbols are transmitted in a frequency band fa as shown in FIG. 3B. The next two OFDM symbols are transmitted in the frequency fb as shown in FIG. 3B. The quality of sub-carriers or tones within such a frequency band are correlated to each other so that the quality of interleaving is degraded.
Even in a TFI mode where the frequency band is changed with each OFDM symbol such as in channel A, channel B shown in FIG. 3B there might be a correlation between different tones within one OFDM symbol because the same tones or sub-carriers within one frequency band can be degraded in the baseband in a similar manner. The OFDM symbol is transmitted within a frequency band which comprises for example 122 sub-carriers. The tones or sub-carriers which are located at the edges of the frequency band may suffer more from imperfect filtering than tones which are located closer to a center frequency of said frequency band. The sub-carriers or tones which are located at the edges of the frequency band might even be cut off by misadjusted filters within the baseband transmission. Another possibility is when zero-IF conversion is used so that the DC offset effects the near-DC tones within the frequency band.
Since conventional interleaving with a bit interleaver circuit according to the state of the art is not optimal in all the above mentioned cases the performance of the data transmission of an OFDM transceiver which employs a conventional bit interleaving circuit having only a symbol interleaving stage and a tone interleaving stage is degraded.

Interleaving circuit for a multiband OFDM transceiver


A system includes a red light emitting diode (LED), a blue LED, and a green LED. The system also includes a red laser, a first filter, a second filter, and a lens. The system includes a first optical path that includes the red LED, the red laser, the first filter, the second filter, and the lens, where the first filter has a filter response to transmit red light from the red laser and to reflect red light from the red LED. The system also includes a second optical path that includes the blue LED, the green LED, the second filter, and the lens, where the second filter has a filter response to transmit blue light from the blue LED, to transmit green light from the green LED, to reflect red light from the red laser, and to reflect red light from the red LED.

Some optical projection systems use light emitting diodes (LEDs), such as blue, green, and red LEDs, as illumination sources. The output image quality of the projection system may be improved by achieving higher brightness from the illumination sources. One way to achieve higher brightness is to increase the output from the green LED. Some green LEDs have a phosphor on the output side of a blue LED. This type of green LED is called a converted green LED. An internal blue LED within the green LED shines on the phosphor, which causes the phosphor to emit green light. The green LED light output can be improved by shining blue light onto the top surface of the phosphor as well as the bottom surface. Utilizing this extra blue light to excite the phosphor from the top as well as the bottom sides is called a top side pump (TSP) green LED. With a TSP architecture, it is common to use an external blue LED to illuminate the front side of the converted green LED. This external illumination causes the green LED to emit more green light. The green light output may increase by up to 50%.


A projector includes a projection optical apparatus that projects an image and a projection position adjustment apparatus that moves the projection optical apparatus in orthogonal directions orthogonal to a central axis of the projection optical apparatus. The projection position adjustment apparatus includes a first movable member that supports the projection optical apparatus, a support member that supports the first movable member movably in the orthogonal directions, and a first transmission unit that moves in the orthogonal directions with the first transmission unit engaging with the first movable member to move the first movable member in the orthogonal directions. The first transmission unit is disposed in a position where the first transmission unit overlaps with part of the first movable member when the projection position adjustment apparatus is viewed along the central axis of the projection optical apparatus.

1. Technical Field
The present invention relates to a projector.
2. Related Art
There is a known projector including a light source apparatus, a light modulation apparatus that modulates light outputted from the light source apparatus to form an image according to image information, and a projection optical apparatus that enlarges and projects the formed image on a screen or any other projection receiving surface. As a projector of this type, there is a known projector including a projection position adjustment apparatus that moves the projection optical apparatus in the upward/downward direction and the rightward/leftward direction orthogonal to the central axis of the projection optical apparatus to adjust the position where the image is projected (see JP-A-2009-175352, for example).
The projection position adjustment apparatus provided in the projector described in JP-A-2009-175352 includes a lens connection section, a first movable plate to which the lens connection section is fixed, a second movable plate, a fixing member, first and second adjustment drivers, and four linkage members.
The lens connection section is connected to a projection lens that forms the projection optical apparatus and fixed to the first movable plate. The first movable plate is configured to be movable along the fixing member in the upward/downward direction and the rightward/leftward direction. The second movable plate is configured to be movable in the rightward/leftward direction between the first movable plate and the fixing member and moves the first movable plate, when the second movable plate moves in the rightward/leftward direction, in the same direction. The fixing member is fixed in the projector and supports the entire projection position adjustment apparatus. The first adjustment driver moves the first movable plate in the upward/downward direction in response to user's operation, and the second adjustment driver moves the second movable plate in the rightward/leftward direction in response to user's operation, whereby the position where the image is projected is changed in the upward/downward direction and the rightward/leftward direction. The linkage members, which are inserted through the fixing member, attach the first movable plate to the fixing member.
In recent years, an electronic apparatus is desired to be compact in size, and a projector is also required to be compact in size. In particular, a projector is not only used after installed where it is used but also used after carried to the location where it is used.
It has been therefore desired to provide a configuration that allows a projector to be more compact.


A sealed electric motor assembly suitable for use in a hostile environment, such as high temperatures, high pressures or in atmospheres containing toxic or corrosive gases. The motor assembly is especially adapted for use with a blower assembly. The stator windings are protected from the hostile environment by hermetically sealing them between the cylindrical housing for the motor and a thin metal canister secured tightly against the inner wall of the housing. Also disclosed are arrangements for removing heat from the motor during its operation. Service life of the electric motor of this invention is greatly increased as compared to conventional motors when operated in a hostile environment.

This invention relates to electric motors and more particularly to an electric motor blower assembly used for circulating gases in a hostile environment. Conventional electric motor blower assemblies usually perform well when operated in a non-hostile environment such as blowing atmospheric air but have been found to have a short performance life if used in many hostile environments, such as the blowing or movement of gases at high temperatures or high pressures or gases containing considerable water vapor or gases that are in themselves toxic or corrosive. It has been discovered that such hostile environments usually attack the stator windings of the electric motor used in the blower assembly and cause a premature failure of the motor.


Parasitic current at the control voltage node of a phase-locked loop (PLL) can significantly reduce performance of the PLL. Off-state transistors in either the charge pump or the filter can cause this parasitic current. A method of canceling a parasitic current generated by the charge pump in the PLL is described. In this method, a leakage current associated with leaky circuits in the charge pump can be determined. An opposing current can be injected to the control voltage node. This opposing current is equal, but opposite, to the leakage current. A method of eliminating a parasitic current generated by the filter in the PLL is also described. In this method, for each programmable capacitor in an unused state, a unity gain buffer can charge the capacitor to the same potential as the control voltage node, thereby providing the same potential on both sides of the switch.

1. Field of the Invention
The present invention relates to a phase-locked loop and in particular to canceling or eliminating a parasitic current that leaks to the control voltage node of this phase-locked loop.
2. Description of the Related Art
Phase-locked loops (PLLs) are frequently used in communication systems to facilitate signal synchronization between a receiver and a transmitter. Specifically, a PLL can be used to synchronize a local oscillator in the receiver with a remote oscillator in the transmitter. This synchronization ensures that the local and remote oscillators have the same or a related frequency and, in some cases, phase.
FIG. 1A illustrates an exemplary phase-locked loop (PLL) 100. PLL 100 includes a phase detector 101, a charge pump 103, a filter 105, and a voltage-controlled oscillator (VCO) 106 (i.e. the local oscillator). Phase detector 101 detects the difference in frequency and phase between a reference signal R (which is extracted from, for example, a transmitted signal or a reference oscillator) and a feedback signal FB (which is an output of VCO 106). If there is a frequency/phase difference, then phase detector 101 generates output signals 102 whose amplitudes are related to that difference (called error signals).
In one embodiment, these error signals are "up" and "down" signals that charge pump 103 can use to source and sink current to and from filter 105 (e.g. a low-pass loop filter). For example, if the VCO frequency is too low, then phase detector 101 sets the "up" signal high, which triggers charge pump 103 to source current into filter 105, thereby increasing the voltage provided to VCO 106. On the other hand, if the VCO frequency is too high, then phase detector 101 sets the "down" signal high, which triggers charge pump 103 to sink current from filter 105, thereby reducing the voltage provided to VCO 106. The effect of the resulting control voltage Vc at control voltage node 104 is to adjust VCO 106 to the same frequency and phase as signal R.
The adjustment made in VCO 106 can be significantly distorted by parasitic current, which can be generated by certain transistors. In general, transistors of current sub-micron CMOS technologies tend to be leaky in their off (i.e. non-conducting) state. That is, a small (but undesirable) parasitic current flows from the drain to the source of a transistor when its gate-to-source voltage is zero (which defines an off state).
Certain transistors in charge pump 103, specifically those serving as switches to source or sink current, can cause significant performance issues if leakage current is present. For example, these sourcing/sinking transistors are ideally turned on for a tiny fraction of a reference period and are turned off (thereby ideally contributing zero current) for the rest of the period. Because these sourcing/sinking transistors actually generate leakage currents in their off state, certain capacitors (shown in FIG. 3A and described below) of filter 105 can be partially charged or discharged within the reference period.
This partial discharge causes an ideal control voltage Vc(ideal) to vary in a periodic manner, thereby resulting in an actual control voltage Vc(actual) shown by the sawtooth waveform in FIG. 1B. This variation in the control voltage Vc at control voltage node 104, in turn, can cause undesirable reference spurs in VCO 106. These reference spurs can detract from the desired signal purity expected of VCO 106.
Certain conditions, such as high temperature when the CMOS device threshold voltage is low, can increase transistor leakage in charge pump 103, thereby exacerbating the control voltage variation and associated spur generation. To minimize this variation in control voltage Vc, longer than minimum length transistors can be used for the sourcing/sinking transistors, thereby resulting in a higher threshold voltage for those transistors. Unfortunately, these larger transistors, although having less leakage, are slower than minimum length devices.
Therefore, a need arises for methods of and circuits for substantially reducing the parasitic current that leaks to the control voltage node of a PLL.

Mitigating parasitic current that leaks to the control voltage node of a phase-locked loop


A stitch control mechanism for a flat knitting machine, for controlling loop size in knitting a knit fabric on a flat knitting machine. A spiral cam plate is attached to one surface of a stitch cam control cam, the spiral cam plate is held between a pair of cam rollers, and the pair of cam rollers are supported on a guide plate combined with a stitch cam and having a portion slidably fitted in a guide slot formed in a base plate.

1. Field of the Invention
The present invention relates to a stitch control mechanism for a flat knitting machine, for controlling stitch size in knitting a knit fabric on a flat knitting machine.
2. Related Art Statement
In knitting a knit fabric, stitch size is dependent on the down stroke of the needle forming a needle loop lowered by a stitch cam. Generally, two stitch cams are disposed respectively on the opposite sides of a raising cam so as to lower the needle. The stitch cams are supported for vertical movement along the raising cam on a base plate and are moved vertically for positioning by stitch cam moving means. As shown in FIG. 5, a stitch cam 51 is supported on a guide plate 54 having a portion slidably fitted in a slot 53 formed in a base plate 52. A stitch cam positioning cam 57 is supported on the output shaft 56 of a motor 55. A cam roller 58 supported on the guide plate 54 at the upper end of the same is biased toward the stitch cam positioning cam 57 by the contractive force of a spring 59. Another cam roller 60 is supported on the lower end of the guide plate 54. The stitch cam 51 is raised when a stitch cam raising cam plate 61 engages the cam roller 60, and the stitch cam 51 is lowered and the cam roller 58 is brought into contact with the stitch cam positioning cam 57 when the stitch cam raising plate 61 separates from the cam roller 60. The stitch cam positioning cam 57 is turned to raise or lower the stitch cam 51 for positioning through the cam roller 58 and the guide plate 54.
Since the cam roller, which moves together with the stitch cam, and the stitch cam positioning cam are engaged by the contractive force of the spring, the displacement of the cam roller is liable to fall behind the turning motion of the stitch cam positioning cam when the knitting machine operates at a high knitting speed.
A stitch control mechanism shown in FIG. 6 incorporates improvements to overcome such a disadvantage. Similarly to the stitch control mechanism shown in FIG. 5, this stitch control mechanism supports a stitch cam 71 on a guide plate 74 having a portion slidably fitted in a slot 73 formed in a guide plate 74. A cam roller 75 pivotally supported on the guide plate 74 engages a stitch cam control lever 76 supported for swing motion on a shaft 77. A cam roller 78 is supported pivotally on the stitch cam control lever 76 in the middle portion of the same and the cam roller 78 is fitted in the spiral cam groove 79 of a stitch cam control cam 80. The stitch cam control cam 80 is fixed to the output shaft 82 of a motor 81. The stitch cam control cam 80 is turned by the motor 81 and, consequently, the stitch cam control lever 76 swings to move the guide plate 74 supporting the cam roller 75 so that the stitch cam 71 is raised or lowered. A small clearance must be formed between the cam roller 78 supported on the stitch cam control lever 76 and the side surface of the spiral cam groove 79 of the stitch cam control cam 80 to allow the cam roller 78 to move along the spiral cam groove 79. Accordingly, the rotation of the motor 81, hence the rotation of the stitch cam control cam 80, may not accurately be transmitted in a linear motion to the stitch cam 71.
Since the spring of the former conventional stitch control mechanism biases the stitch cam continuously downward as stated above, the stitch cam control cam and the cam roller supported on the guide plate supporting the stitch cam must be separated from each other for every loop size adjustment or the motor must be of a large output capacity. Accordingly, the carriage is formed inevitably in a comparatively large size and hence a comparatively large driving mechanism is required.
Furthermore, the spring for biasing the stitch cam downward, the cam plate for separating the cam roller provided on the guide plate supporting the stitch cam from the stitch cam control cam, and the mechanism for operating those components, which are essential to this conventional stitch control mechanism, increase the size and weight of the stitch control mechanism, which is an impediment to the enhancement of the knitting speed of the knitting machine. Still further, since a clearance is formed between the stitch cam control cam and the cam roller, i.e., a stitch cam moving member, the cam roller is unable to follow the movement of the stitch cam control cam without delay.

Stitch control mechanism for a flat knitting machine


An automatic busy message sending system for use in telephone communication to inform the calling subscriber, when the called subscriber's line is busy, of the duration of the busy status by voice. As the calling subscriber can receive, when the called subscriber's line is busy, a busy status message corresponding to the duration of the called subscriber's busy status instead of a busy tone, the calling subscriber can more accurately anticipate the ending time of the called subscriber's communication, and thereby minimize the repetition of the call initiating procedure. This service can be realized without having to add extra equipment to the electronic switching system currently in operation.

1. Field of the Invention
The present invention relates to a telephone switching system, and more particularly to an electronic telephone switching system capable of automatically transmitting, when a calling subscriber encounters any destination that is busy, a voice message indicating the length of time from the beginning of that busy status till the current point of time, i.e. the duration of the busy status, instead of a busy tone to the calling subscriber.
2. Description of the Prior Art
A telephone switching system, upon receiving dial number information from a calling subscriber, identifies the telephone number of the destination, sends out a ringing signal to the telephone set of the destination, and sends back a ring back tone to the calling subscriber. When the subscriber at the destination takes up the handset of his telephone, the switching system detects the off-hook status, stops sending the ringing signal and the ring back tone, and operates a communication path switch to connect, and to keep connected, the calling and the called subscribers.
Meanwhile, if the called subscriber is communicating with another subscriber when a call is initiated by said calling subscriber, the switching system will send out a busy tone to the calling subscriber. The calling subscriber knows by this busy tone that the destination of his call is busy, and puts his handset on the hook, in response to which the switching system is released.
In order for the calling subscriber to achieve connection with his desired destination, he should repeat the call initiating procedure some time after he puts his handset on the hook. If the destination is still busy, he should repeat the procedure until the connection is finally achieved.
To dispense with this troublesome repetition of the call initiating procedure, an electronic messaging service is proposed in the U.S. Pat. No. 4,932,042, "Spontaneous Voice and Data Messaging". The service thereby provided is to send an announcement signal regarding the availability of a messaging service, either simultaneously with or instead of a busy tone, to the calling subscriber. A calling subscriber wishing to use this messaging service dials a prescribed three-digit number. Upon detection of this three-digit number by the switching system, the calling subscriber is connected to a voice message operation center (VMOC), which is supplied by the switching system with the telephone numbers of the calling and called subscribers and billing information.
Meanwhile, in order to verify the telephone number of the destination of the message, the VMOC sends an announcement signal to the calling subscriber. As the calling subscriber responds to that signal by operating a dial key, the VMOC receives and records a voice message from the calling subscriber. Upon completion of the reception and recording of this voice message, call initiation to convey that message is accomplished by the prior art.
The use of this messaging service would require the payment of a subscription fee for this service in addition to the usual fixed monthly due for subscription to the telephone system, because the installation and operation of said VMOC costs a large sum of money. The collection of this additional fee is preventing the messaging service according to said patented invention from being used more extensively.

Automatic busy message sending system for telephone communication


An apparatus and a system, as well as a method and article, may operate to adjust the rate of a clock to be coupled to a processor, such as a video stream processor, according to a previously communicated data rate associated with a data stream selected from a number of data streams.

Electronic devices may be designed to operate at speeds selected for processing a maximum anticipated data rate. Such operation may be suboptimal when data rates vary over a wide range, such as when changes from tuner to tuner occur in a video system (e.g., during channel surfing activity). In such devices, system power usage may be greater than necessary when relatively low-speed data rates are encountered.

Power management apparatus, systems, and methods


A packed tower for the absorption of water or SO3 in sulfuric acid includes a jacket and a domed grid disposed within the jacket. The domed grid is held on an L-shaped ring such that a gap is formed between an outside diameter of the domed grid and the jacket. A bed of packings is held on the domed grid through which the sulfuric acid charged from above trickles. A gas supply tube is disposed in a lower region of the tower. A gas outlet is disposed above the bed. An acid outlet is disposed in the lower region of the tower.

The acid part of a sulfuric acid plant comprises the drying of the sulfur-dioxide (SO2)-containing gases and the air as well as the absorption of the sulfur trioxide (SO3) formed in the contact plant for forming the end products sulfuric acid and oleum. The standard configuration for drier and absorber is the packed tower constructed as counterflow apparatus. This upright cylindrical container is divided into three zones. The lower part is formed by the sump with the acid outlet arranged directly above the bottom or also in the bottom and the gas inlet welded into the sheet metal jacket above the acid level. The middle part contains the packed bed on a grid through which the acid, which is uniformly distributed by a sprinkling system located on the bed, trickles downwards. Above the packed bed the acid inlet to the sprinkling system and the gas outlet are located. Packed towers for the most part consist of a cylindrical sheet metal jacket with a multilayer acid-resistant brick lining and with an arched bottom and a grid of acid-resistant ceramic material for accommodating the packed bed. As grid a domed grid with at least 40% free flow cross-section is incorporated, which is suitable for tower diameters up to 15 m. For the packed bed, saddle packings of ceramic material chiefly are chosen, which have large specific surfaces, whereby an intensive mass transfer between acid and gas is given. (cf. Winnacker/Küchler, Chemische Technik: Prozesse und Produkte, edited by Roland Dittmeyer et al, Vol. 3: Anorganische Grundstoffe, Zwischenprodukte, p. 100 ff, Wiley-VCH Verlag Weinheim, 2005.) A corresponding packed tower also is shown for example in DE 33 20 527 C2.
Because of the deteriorating mounting quality of the brick lining, non-brick-lined stainless steel towers of silicon (Si)-containing stainless steel have increasingly been used for many years. It was found that at several points of the packed tower corrosion problems occur due to waste acid (drier) or oleum formed (absorber). Particularly problematic zones are found in the region of the gas inlet or opposite the gas inlet, where the acid film on the tower wall is only thin. Corrosion also occurs at vertical props for the stainless steel grid or below the packed bed on horizontal surfaces on which the acid film also is only thin. Due to the moisture of the gas in the drying tower waste acid is produced, which is highly corrosive and attacks the steel surfaces. In the absorber towers oleum is produced by the SO3 in gas, which likewise is highly corrosive. In the past, this problem has been countered for example in that around the gas inlet a covering plate of silicon-free austenitic stainless steel is provided, which due to the strong corrosion must however regularly be replaced.


An ultrasound image system has a plurality of channels. At least one of the plurality of channels has a capacitive T/R switch.

Medical or nondestructive testing (NDT) ultrasound imaging applications have a growing demand for more sophisticated transmit-pulse/echo-signal switching in the receiver frontend circuit. This may be necessary to generate high resolution acoustic images. The conventional ultrasound T/R echo switching circuits usually have an array of high-voltage diode switches or MOSFET-based voltage-controlled switch devices in between a transducer probe PZT array and Tx/Rx channel circuitry
These circuits have several limitations. These circuits ether are require to have the forward bias current from two low-noise, very clean power supplies for the diode-bridge type of T/R switch, or have a quite larger size MOSFET analog switch with control circuit for the MOSFET-based type of T/R switch, which normally has higher turn-off trig voltage or current, and higher off-capacitance intrinsically. Second, the transmitting pulses are generally high voltage and the receiving echo signals are generally low voltage. However, both signals have to apply to the same T/R switch. This requires that every T/R switch must have a low on-impedance when the T/R switch is turned on, and that it can withstand high voltage when it is turned off. In order to have low on-resistance in the diode-bridge type circuit normally requires quite large forward bias current (e.g., 5 to 10 mA). Further, in a 128-channel system total power of 6.4 to 12.8 Watts of clean power supplies and 256 pieces of decoupling capacitors are also required. Additional PCB area is also required too for all these circuitry. These requirements make the cost of the T/R switch very high.
Therefore, it would be desirable to provide a circuit and method that overcomes the above problems. The circuit and method would be able to switch the echo signal to overcome the above problems.

Ultrasound capacitive T/R switch device, circuit