A method and a device for measuring interference in a communication system are provided. The method includes measuring interference in a base station of a communication system, in which one or more antenna groups are arranged at different positions in a single cell, includes the steps of determining a reception antenna group which is one of the one or more antenna groups that transmits a signal other than an interference signal to a terminal, determining a reference signal in order to measure the strength of the signal transmitted by the reception antenna group, a step of determining a wireless resource so as to measure the interference in each of the one or more antenna groups, and notifying the terminal with the strength of the signal transmitted by the reception antenna group and information for measuring the interference in each of the one or more antenna groups.

A mobile communication system has been developed as a high-speed and high-quality wireless packet data communication system in order to provide data services and multimedia services as well as voice-based services. In recent years, a variety of mobile communication standards such as High Speed Downlink Packet Access (HSDPA), High Speed Uplink Packet Access (HSUPA), Long Term Evolution (LTE), and Long Term Evolution Advanced (LTE-A) of the 3rd Generation Partnership Project (3GPP), High Rate Packet Data (HRPD) of the 3GPP2, and Institute of Electrical and Electronics Engineers (IEEE) 802.16 have been developed to support a high-speed and high-quality wireless packet data transmission service. In particular, the LTE system is a system that has been developed to efficiently support high-speed wireless packet data transmission, and may maximize a capacity of a wireless system using a variety of wireless access technologies. In addition, an LTE-A system, which evolved from the LTE system, has an improved data transmission capability when compared to the LTE system.
In general, the LTE system refers to a base station and a terminal corresponding to Release 8 or 9 of the 3GPP standard group, and the LTE-A system refers to a base station and a terminal corresponding to Release 10 of the 3GPP standard group. In the 3GPP standard group, the standardization of the subsequent Release having improved performance based on the standardization of the LTE-A system is ongoing even after the standardization of the LTE-A system.
The 3G and 4G wireless packet data communication system such as HSDPA, HSUPA, HRPD, LTE/LTE-A, etc. may use techniques such as an Adaptive Modulation and Coding (hereinafter, referred to as "Adaptive Modulation and Coding (AMC)") method, a channel-sensitive scheduling method, and the like in order to improve the transmission efficiency. When using the AMC method, a transmitter may adjust an amount of data to be transmitted according to a channel status. By way of example, when the channel status is poor, the transmitter may adjust a reception error probability to a desired level by reducing the amount of data to be transmitted, and when the channel status is good, the transmitter may efficiently transmit a large amount of information while adjusting the reception error probability to the desired level by increasing the amount of data to be transmitted. When using a channel-sensitive scheduling resource management method, the transmitter may selectively serve a user having an excellent channel status among multiple users, and therefore the system capacity is increased when compared to allocating a channel to one user and serving the user. This capacity increase is often referred to as "Multi-user Diversity gain". The AMC method and the channel-sensitive scheduling method may receive feedbacks of partial channel status information from a receiver, and apply an appropriate modulation and coding scheme at the time that is determined to be the most efficient.
When used together with a Multiple Input Multiple Output (MIMO) transmission method, the AMC method may include a function of determining the number of spatial layers of signals to be transmitted or a rank thereof. In this case, the AMC method may consider the number of layers to which data is transmitted using MIMO, without simply considering only the coding rate and the modulation scheme in determining the optimum data rate.
In recent years, research has been actively conducted to switch Code Division Multiple Access (CDMA), i.e., a multiple access scheme that has been used in the 2G and 3G mobile communication systems, to Orthogonal Frequency Division Multiple Access (OFDMA) in the next-generation communication system, and, in 3GPP and 3GPP2, the standardization on the evolved systems that use OFDMA is ongoing. It is well known that OFDMA may contribute to an increase in capacity, when compared to CDMA. By way of example, OFDMA can perform frequency domain scheduling on a frequency axis to thereby increase the capacity. As the capacity gain was obtained by the channel-sensitive scheduling method based on the time-varying characteristics of channels, additional capacity gain can be obtained by utilizing the frequency-dependent characteristics of channels.
FIG. 1 is a diagram illustrating time and frequency resources which are used in an LTE/LTE-A system according to the related art.
Referring to FIG. 1, wireless resources, which are allocated to user equipment (UE) by an enhanced Node B (eNB), are divided in a resource block (RB) unit on a frequency axis, and in a subframe unit on a time axis. Generally, one RB includes 12 subcarriers in the LTE/LTE-A system, and occupies a band of 180 kHz. Generally, one subframe includes 14 OFDM symbol sections in the LTE/LET-A system, and occupies a time section of 1 msec. When performing scheduling, the LTE/LTE-A system may allocate resources in the subframe unit on the time axis, and in the RB unit on the frequency axis.
FIG. 2 is a diagram illustrating wireless resources of one subframe and one RB which is a minimum unit for downlink scheduling in an LTE/LTE-A system according to the related art.
Referring to FIG. 2, the wireless resources may be configured with one subframe on the time axis, and one RB on the frequency axis. In this instance, one RB is constituted of 12 subcarriers in a frequency domain, and one subframe is constituted of 14 OFDM symbols in a time domain, so that the wireless resources have a total of 168 unique frequencies and time positions. In LTE/LTE-A, each unique frequency and time position of FIG. 2 is called a resource element (RE).
A plurality of different kinds of signals can be transmitted to the wireless resources illustrated in FIG. 2.
1) Cell Specific Reference Signal (CRS): Reference signal transmitted for all UEs belonging to one cell.
2) Demodulation Reference Signal (DMRS): Reference signal transmitted for a specific UE.
3) Physical Downlink Shared Channel (PDSCH): Data channel transmitted by a downlink, used in order for an eNB to transmit traffic to a UE, and transmitted using an RE in which a reference signal is not transmitted in data region of FIG. 2.
4) Channel Status Information Reference Signal (CSI-RS): Reference signal transmitted for UEs belonging to one cell, and used to measure channel status. Here, a plurality of CSI-RSs can be transmitted to one cell.
5) Other control channels (PHICH, PCFICH, and PDCCH): Provide control information required for receiving PDSCH by UE, or transmitting Acknowledgement/Negative Acknowledgement (ACK/NACK) for operating hybrid automatic repeat request (HARQ) concerning uplink data transmission.
Other than the above-described signals, the LTE-A system may set muting so that a CSI-RS transmitted by another eNB can be transmitted to UEs of the corresponding cell without interference. Muting may be applied in a position in which the CSI-RS can be transmitted, and the UE may receive a traffic signal over the corresponding wireless resources. Due to characteristics of muting, muting may be applied in the position of the CSI-RS and transmission power may not be transmitted, and therefore muting in the LTE-A system may be also referred to as zero-power CSI-RS.
In FIG. 2, the CSI-RS may be transmitted using a part of positions indicated by A, B, C, D, E, E, F, G, H, I, and J depending on the number of remote radio heads (RRHs) that transmit the CSI-RS, and muting may be also applied to the part of the positions indicated by A, B, C, D, E, E, F, G, H, I, and J. In particular, the CSI-RS may be transmitted to 2, 4, and 8 REs depending on the number of CSI-RS ports that transmit the CSI-RS. When the number of CSI-RS ports is 2, the CSI-RS may be transmitted to half of a specific pattern in FIG. 2, when the number of CSI-RS ports is 4, the CSI-RS may be transmitted to the whole of the specific pattern, and when the number of CSI-RS ports is 8, the CSI-RS may be transmitted using two patterns. On the other hand, muting may be always performed in a unit of one pattern. That is, muting may be applied to a plurality of patterns, but may not be applied to only a part of one pattern when a position of muting is not overlapped with the position of the CSI-RS. However, only when the position of the CSI-RS and the position of muting are overlapped with each other, muting may be applied to only a part of one pattern.
In order to measure a downlink channel status in a cellular system, a reference signal should be transmitted. In the LTE-A system of 3GPP, a UE measures a channel status between an eNB and the UE using a CSI-RS transmitted by the eNB. For the channel status, several factors such as an amount of interference in a downlink may be basically considered. The amount of interference in the downlink may include interference signals, thermal noise, and the like generated by RRHs belonging to a neighboring eNB, and may be important when the UE determines the downlink channel status. By way of example, when an eNB with one transmission RRH transmits data to a UE with one reception RRH, the UE should decide energy per symbol to interference density ratio (Es/Io) by determining energy per symbol that can be received by a downlink and an amount of interference that can be simultaneously received in a section that receives the corresponding symbol, from the reference signal received from the eNB. The decided Es/Io may be notified to the eNB, and the eNB may decide a data transmission rate of the downlink that transmits data to the UE using the Es/Io.
In the case of a general mobile communication system, an eNB equipment is disposed in the midpoint of each cell, and the corresponding eNB equipment performs mobile communication with the UE using one or a plurality of RRHs as located in a limited position. A mobile communication system in which RRHs belonging to a single cell are arranged in the same location in this manner is referred to as a centralized antenna system (CAS). On the other hand, a mobile communication system in which RRHs belonging to a single cell are arranged in distributed locations within the cell is referred to as a distributed antenna system (DAS).
FIG. 3 is a diagram illustrating distributed locations of RRHs arranged in a general DAS according to the related art.
Referring to FIG. 3, a DAS includes two cells 300 and 310, and each of the cells 300 and 310 includes a single high-output RRH 320 and four low-output RRHs 340. In this instance, the high-output RRH 320 may provide minimal service throughout the entire region included in a cell region, and the low-output RRHs 340 may provide a high data rate-based service to limited UEs in a limited region within the cell. In addition, the low-output RRHs 340 and the high-output RRH 320 are all connected to a central controller (not illustrated) as shown by dotted lines 330, and thereby operated according to scheduling and wireless resource allocation of the central controller. In a single RRH location geographically separated from the DAS, one or a plurality of RRHs may be arranged. One or a plurality of RRHs arranged in the same location in the DAS may be referred to as an RRH group.
In the DAS shown in FIG. 3, a UE may receive signals from a single geographically separated RRH group, but signals transmitted from the remaining RRH groups may act as interference to the UE.
FIG. 4 is a diagram illustrating an interference phenomenon that occurs when data is transmitted to other UEs for each RRH group in a DAS according to the related art.
Referring to FIG. 4, a UE 1 (400), a UE 2 (420), a UE 3 (440), and a UE 4 (460) receive traffic signals from RRH groups 410, 430, 450, and 470, respectively. The UE 1 (400) may receive interference from other RRH groups that transmit the traffic signals to other UEs while receiving the traffic signals from the RRH group 410, and by way of example, in FIG. 4, signals transmitted from the RRH groups 430, 450, and 470 may generate an interference effect in the UE 1 (400).
In general, there are the following two kinds of interference that occur by another RRH group in a DAS.
1) Inter-cell interference: Interference that occurs in RRH group of other cell
2) Intra-cell interference: Interference that occurs in RRH group of the same cell
In FIG. 4, as intra-cell interference to the UE 1 (400), there is interference that occurs in the RRH group 430 belonging to the same cell, and as inter-cell interference, there is interference that occurs in the RRH groups 450 and 470 of a neighboring cell. The inter-cell interference and intra-cell interference may be received simultaneously with data channel signals of a UE to disturb reception of the data channel signals of the UE.
In order for a UE of the DAS to receive signals at an optimum data transmission rate using a downlink, it is necessary to accurately measure inter-cell interference and intra-cell interference that cause interference to the UE itself, and request a data transmission rate from an eNB based on a compared result between the measured interference and the strength of the received signals.
In the case of a general CAS rather than the DAS, there is only one RRH group for each cell. In this case, intra-cell interference that occurs between other RRH groups within the same cell as shown in FIG. 4 does not occur, and only inter-cell interference that occurs in other cells occurs. When the LTE/LTE-A system is configured as a CAS, it is possible to measure inter-cell interference using the CRS described in FIG. 2. In general, in the CAS, a UE receives the CRS, and then converts the CRS having periodic characteristics in a frequency domain into a delay domain signal using Inverse Fast Fourier Transform (IFFT).
The above information is presented as background information only to assist with an understanding of the present disclosure. No determination has been made, and no assertion is made, as to whether any of the above might be applicable as prior art with regard to the present disclosure.

Method and device for measuring interference in communication system


A conveyor system for transporting at least one magnetically attractable article. The conveyor system includes an elongate track extending along a path including at least one arcuate section; a chain conveyor mounted for movement through the track; at least two grids attached to the chain conveyor, a portion of each of said grids extending laterally relative to said elongate track; at least one magnet mounted on each grid for coupling by magnetic force at least one magnetically attractable article to at least one of the grids; and a connector apparatus for allowing limited movement of the article coupled by the magnet, relative to the grid, while retaining the article in engagement with the grid.

1. Field of the Invention
This invention relates in general to a continuous proofing and baking apparatus for use in a bakery. In particular, the invention relates to a continuous proofing and baking apparatus having an improved conveyor system for carrying product pans through the proofing and baking apparatus.
2. Description of Prior Art
In the commercial baking industry, dough products are exposed to a predetermined temperature and humidity environment to cause the dough to rise. Following this proofing operation, the dough products are transferred to an oven for baking.
The dough products may be conveyed from the proofing apparatus to the oven on a continuous conveyor system. The conveyor system has an elongated track which extends through and between the proofing and baking sections of the apparatus. A conveyor chain, having a plurality of links, moves through the elongated track and carries a plurality of product supporting grids. Each product pan is supported by one or more product supporting grids.
As the conveyor chain moves through the proofing and baking apparatus, the track has some straight portions and some curved portions. As the conveyor chain travels from a straight portion to a curved portion of the track, the relative positions of two consecutive product supporting grids will change. The conveyor system must be able to support the product pans, even though the relative positions of the product supporting grids change.


Provided is an operating device with which the erroneous detection of depression operations can be prevented when tactile sensation is presented, without requiring a sophisticated control. This operating device (1) substantially includes a touch panel (2) that detects tracing on an operation screen; a depression mechanism unit (3) capable of a depression operation in the direction downward from a standard position of the touch panel (2) a push switch (5) that detects a depression operation via the depression mechanism (3), and a drive mechanism unit (7) capable of driving between the standard position of the touch panel (2) and a position thereabove.

As a conventional technique, an input device is known which is provided with a touch sensor for detecting a sliding operation, a tactile presentation portion for vibrating the touch sensor to provide tactile sensation and a load sensing portion sensing a load applied to the touch sensor and thereby detecting a push operation performed on the touch sensor (see, e.g., PTL 1).
This input device can provide a tactile feedback to a user according to sliding and push operations performed on the touch sensor by the user.

Operating device for providing tactile feedback on a depressible touch screen

KABUSHIKI KAISHA TOKAI RIKA DENKI SEISAKUSHO


An amine is prepared by cleaving a carbamate of the formula I

1. Field of the Invention
The present invention relates to a process for preparing amines by catalytic cleaving of carbamates.
2. Discussion of the Background
The cleaving, i.e. the decarboxylation, of carbamates is a common process for preparing amines. The resulting amines are in turn starting compounds for numerous industrial intermediates and active pharmaceutical ingredients.
The general reaction mechanism of the cleaving of carbamates is described by Ewing et al. in their publication J. Amer. Chem. Soc. (1980) 102(9), 3072-3084:

The cleaving of carbamates in the presence of tetra-n-butylammonium fluoride (Bu4NF) is described by Jacquemard et al. in Tetrahedron (2004) 60(44), 10039-10047:

where                R=methyl, ethyl, tert-butyl, benzyl, allyl or phenyl,        R′ is alkyl or aryl,        R″ is hydrogen, alkyl or aryl.
The general structure of the carbamates is described by Gaines in J. Org. Chem. (1985) 50, 410-411 as a mono-molecular zwitterion or disalt.
Japanese publication JP 2006-069941 A describes, inter alia, the thermal decomposition of carbamates to prepare isocyanates. In contrast, JP 2004-262892 A describes the thermal decomposition of carbamates in the presence of a tin catalyst and of a solid catalyst selected from silicates, silica gel and/or metal.
The decomposition of ammonium carbamate to ammonia and carbon dioxide is described by WO 2006/094541 A and EP 1 195 194 A.
The cleaving of carbamates in the presence of a tin catalyst is also described in JP 2004-262892 A. The use of iodine to cleave carbamates is described by Vatele et al. in Tetrahedron Letters (2003), 44(51), 9127-9129.

Abstract: wherein R1=hydrogen, alkyl, aryl or phenyl group, R2=alkyl, aryl or phenyl group, wherein R1 and R2 are independently substituted or unsubstituted, wherein the cleaving of the carbamate is performed in the presence of an acid of the formula II

Abstract: wherein R3, R4 and R5 are independently alkyl or phenyl group, wherein R1, R2, R3, R4 and R5 are the same or different.


A current controlled field emission display includes a current control circuit coupled to respective emitters to control current flow through the emitters. The control circuit includes a limiting resistor to limit the current flow to the emitters. To compensate for temperature or other variations in the resistance of the limiting resistor, the control circuit also includes a compensating resistor that produces an offsetting change in a driving voltage within the control circuit. As the resistance of the limiting resistance drops, which would allow the current through the emitter to increase, the compensating resistor produces an offsetting voltage drop that increases the voltage across the limiting resistance to maintain the current through the limiting resistance constant. The offsetting current reduction thus reduces the overall effect of temperature on current flowing through the emitter set.

Flat panel, matrix addressable displays are widely used in a variety of applications, including computer displays. One type of matrix addressable display well-suited for such applications is the field emission display. Field emission displays typically include a generally planar substrate having an array of projecting emitters. In many cases, the emitters are conical projections integral to the substrate. Typically, the emitters are grouped into emitter sets where the bases of the emitters in each set are commonly connected.
A conductive extraction grid is positioned above the emitters and driven with a voltage of about 30-120 V. The emitter sets are then selectively activated by providing a current path from the bases to the ground. Providing a current path to ground allows electrons to flow from the emitters in response to the extraction grid voltage. If the voltage differential between the emitters and extraction grid is sufficiently high, the resulting electric field extracts electrons from the emitters.
Field emission displays also include display screens mounted adjacent the substrates. The display screens are formed from glass plates coated with a transparent conductive material to form an anode biased to about 1-2 kV. A cathodoluminescent layer covers the exposed surface of the anode. The emitted electrons are attracted by the anode and strike the cathodoluminescent layer, causing the cathodoluminescent layer to emit light at the impact site. The emitted light then passes through the anode and the glass plate where it is visible to a viewer.
The brightness of the light produced in response to the emitted electrons depends, in part, upon the rate at which electrons strike the cathodoluminescent layer, which in turn depends upon the magnitude of current flow to the emitters. The brightness of each area can thus be controlled by controlling the current flow to the respective emitter set. By selectively controlling the current flow to the emitter sets, the light from each area of the display can be controlled and an image can be produced. The light emitted from each of the areas thus becomes all or part of a picture element or "pixel."
One problem with the above-described approach is that the response of the emitters sets and control circuitry vary. One cause of such variation may be environmental factors such as temperature. For example, the components of the control circuitry can have temperature-dependent electrical characteristics such as resistance and current leakage. For a given electrical input, the brightness of the emitted light may then vary according to the temperature in and around the display.

Temperature compensated matrix addressable display


A method for transmitting an analog datastream includes coding data input into a data input device. The resulting coded data is assembled into coded data blocks. These blocks are transformed into at least one discrete multitone symbol. An initial filter value extracted from the discrete multitone symbol is forwarded to a stored-value determining device, to which a frame alignment signal has been applied. A stored value for a first filtering device is determined as a function of the at least one initial filter value and the frame alignment signal. This value, and the multitone symbol, are applied to the first filtering device, thereby generating a filtered discrete multitone symbol. The filtered symbol is forwarded to a second filtering device to which the stored value determined in the stored-value determining device is applied, thereby generating a transient-compensated discrete multitone symbol. This is then converted into an analog signal for transmission.

Usually, a multitone method (DMT—discrete multitone) is used for asymmetric datastream transmission via normal telephone lines, normal telephone lines usually being constructed as asymmetric digital subscriber lines (ADSL).
An essential advantage of ADSL transmission techniques consists in being able to use conventional cable networks for a transmission, twisted copper pairs normally being used.
High-speed digital subscriber lines of the prior art are described, for example, in the publication "High-speed digital subscriber lines, IEEE Journal Sel. Ar. In Comm., Vol. 9, No. 6, August 1991".
Among the transmission methods with a high data rate, which are based on digital subscriber lines (DSL), a number of VDSL (Very High Data Rate DSL) arrangements are known and, for example, methods such as carrierless amplitude/phase (CAP), discrete wavelet multitone (DWMT), single line code (SLC) and discrete multitone (DMT) can be used for these. In the DMT method, the transmit signal is provided from multiple sinusoidal or cosinusoidal signals, where both the amplitude and the phase can be modulated of each individual sinusoidal or cosinusoidal signal. The multiple modulated signals thus obtained are provided as quadrature-amplitude modulated (QAM) signals.
FIG. 4 shows a conventional datastream transmitter in which data 123 to be transmitted are input via a data input device 201. The data 123 to be transmitted are supplied to a coding device 202 in which the data are first coded and then assembled to form coded data blocks 125, a predeterminable number of bits to be transmitted being allocated to a complex number depending on the scaling. Finally, the coded data blocks 125 output by the coding device 202 are supplied to a retransformation device 203.
Conventionally, the retransformation device 203 transforms the data present in the frequency domain into the time domain by means of an inverse fast Fourier transform (IFFT), N samples of a transmitter signal being generated directly from N/2 complex numbers, where all N samples will be designated as a discrete multitone (DMT) symbol in the text which follows. The complex numbers can be provided as amplitude values of cosinusoidal and sinusoidal oscillations (real component and imaginary component) to be sent out within a data block, the frequencies being distributed equidistantly in accordance with the relation:
            f      i        =                            i          ·                      1            T                                                                    i            =      1        ,  2  ,      …                                N      /      2      where T is a period for a transmission of a discrete multitone symbol and N is a number of samples for a discrete multitone symbol. For example, conventional ADSL DMT methods use 256 tones, which can be modulated in amount and phase in each case as sinusoidal tones, in a downstream mode, i.e. in a data transmission from at least one switching center to at least one subscriber. The fundamental frequency is 4.3 kHz and the frequency spacing between successive tones is also 4.3 kHz. Thus, a frequency spectrum from 4.3 kHz (fundamental frequency) to (4.3 kHz+256×4.3 kHz)=1.1 MHz is transmitted. Each DMT symbol is thus represented by a sinusoidal tone which can be modulated in amount and phase, a maximum of 15 bits per symbol usually being represented as complex number. During the transmission of a multitone signal of this type, the problem occurs, however, that transient effects are produced by the transmission channel which, for example, can be constructed as a twisted copper pair, which effects have decayed after, for example, M samples.
In the transmitter device, the last M samples of a DMT symbol are appended to a block start after an inverse fast Fourier transform (IFFT), where the following relation applies: M<N. Due to this cyclic extension (cyclic prefix), a periodic signal can be simulated for the datastream receiver when the transient effect caused by the transmission channel has decayed after M samples and mutual interference between different DMT symbols, i.e. inter-symbol interference (ISI), can be avoided.
As a result, the equalization effort in an equalization device arranged in the datastream receiver can be considerably reduced in conventional methods since after demodulation of the received analog datastream 101 in the datastream receiver, only a simple correction with the inverse frequency response of the transmission channel must be performed in the correction device 112.
A significant disadvantage of a data transmission according to the ADSL method over copper lines in which multitone signals are transmitted consists in that long transient effects occur. The cyclic prefix is thus normally extended in order to supply a periodic signal to the datastream receiver. However, the cyclic prefix must be kept small in relation to the DMT symbol length N, i.e. the following relation must apply:M<<N,since otherwise a reduction in transmission capacity disadvantageously occurs.
In the ADSL standard, a DMT symbol length of N=64 and a value of a cyclic prefix of M=4, for example, is provided for a data transmission from a subscriber to a switch. To limit a transient effect to the cyclic prefix, a special time domain equalizer (TDEQ) in the form of an adaptive transversal filter which operates at a sampling rate Fs (for example 276 kHz in the switching center with ADSL) is provided in the preprocessing device arranged in the datastream receiver in the known method.
Due to the necessary restriction in the length of the cyclic prefix to, for example, M=4, as mentioned above, the quality of transmission is disadvantageously impaired in conventional methods for transmitting an analog datastream 101 since there is still considerable inter-symbol interference (ISI) even when an equalizer is used in the datastream receiver.
A normal transmission channel also disadvantageously contains high-pass and low-pass filters in order to limit the bandwidth of the analog datastream to be transmitted and in order to suppress out-of-band noise in analog-digital and digital-analog converters which can be constructed, for example, as sigma-delta converters.
In particular, it is disadvantageous that when low-pass filters are excited with DMT signals, transient effects occur which have considerable spectral components in a frequency range above the transmission signal band provided. With a sampling rate Fs of, for example, 276 kHz, convolutional products produce spectral components in the transmission signal band which cannot be eliminated by the equalizer arranged in the datastream receiver. These convolutional products are disadvantageously contained as interference signals in the transmission signal band which impairs the quality of transmission.
A multitone signal generated in the time domain is then transmitted in the form of DMT symbols according to FIG. 4. To provide an analog transmitter signal 211, an analog-digital converter is provided for conversion from a digital multitone signal 303 into the analog transmitter signal 211.
A further known datastream transmitter is shown in FIG. 5 and, in addition to the components illustrated in FIG. 4, a first filtering device 131′ and a second filtering device 132′ are here arranged between the retransformation device 203 and the digital-analog converter 204.
FIG. 5 illustrates that a typical transmission channel contains high-pass and low-pass filters for limiting the bandwidth of channel transmission signals. As shown in FIG. 5, the discrete multitone (DMT) symbol 208 is high-pass filtered in the first filtering device 131′ in order to obtain a filtered discrete multitone symbol 209′. This filtered discrete multitone symbol 209′ is low-pass filtered in the second filtering device 132′. The filtering devices 131′ and 132′ used for band limiting have the disadvantage that in the case of excitation with DMT symbols, transient effects occur which limit a data transmission rate. It is particularly if low-pass filters are used as filtering devices 131′ and 132′, respectively, that significant spectral components which, in particular, have an effect with a short cyclic prefix, occur in the frequency range of the signal band. After digital-analog conversion, the time domain signal output by the second filtering device 132′ is finally transferred in a digital-analog converter where transient effects unsuitably occur.

Datastream transmitters for discrete multitone systems


The cook top has a transparent, colorless glass ceramic or glass panel providing a cooking surface, which is made from pre-stressed special glass, and an IR-permeable undercoat on the underside of the panel, which is a heat-resistant inorganic enamel paint. In order to provide a heat-resistant cook top, through which the internal parts of the cooking unit cannot be observed, without reducing the strength of the cook top, the inorganic enamel paint includes 70 to 99, preferably 80 to 95, percent by weight of inorganic pigment and from 1 to 30, preferably 5 to 20, percent by weight of glass flux. The glass flux preferably is a borosilicate glass with a thermal expansion coefficient less than or equal to 4*10−6 K−1. The coating may be applied by screen printing techniques.

1. Field of the Invention
The present invention relates to glass ceramic cook top or glass cook top comprising a glass ceramic or glass panel made of pre-stressed special glass, which provides a cooking surface, and an IR-permeable coating made of a heat-resistant inorganic enamel paint, which comprises an inorganic pigment and a glass flux, on the underside of the glass ceramic or glass panel.
2. Description of the Related Art
Cooking ranges with glass or glass ceramic panels acting as cook tops are commercially available in the art. Glass cook tops are typically transparent for visible light, however glass ceramic cook tops are usually translucent. When illuminated from above, e.g. by the cooking unit lighting, or from below, by heat radiation, the heating elements, cables and other structural components, which are hidden below the glass ceramic cooking area, can be observed and show through it. The appearance of these structural elements is considered to be troublesome for the user.
This sort of glass/glass ceramic cook top is typically, as described in EP 0 220 333, colored with color-imparting ions to reduce transmission, so that the operating components under the glass/glass ceramic cook top are not observable from above. These cook tops are thus practically non-transparent for radiation in the visible range and appear to be black.
It is also known to solve this problem by an observation-preventing coating on the underside of the transparent and/or translucent glass and/or glass ceramic cook top. Thus JPH 7-17409 and JP 51-89517 describe glass ceramic cook tops made from a transparent, colorless glass ceramic whose underside is printed with a temperature-resistant paint. This colored paint is thus formed so that the required non-transparency is provided, i.e. it replaces the otherwise conventional coloring, so that the cook top providing the cooking surface appears black when observed.
EP 0 746 179 A2 discloses the use of a covering mask for making a partially non-transparent cook top. However the application of a observation-preventing coating is costly, especially when it occurs by means of screen printing, as in the case of DE 199 06 737 A1.
The observation preventing coating usually consists of a colored lacquer, which contains organic or inorganic pigments. The lacquer adheres to the glass ceramic because of an organic additive provided in the lacquer. Thus the above-described DE 199 06 737 A1 discloses an alkyd resin lacquer for the coating. Luster-imparting paints, sol-gel coatings and noble metal paints are known as the colored lacquers for the observation-preventing coating from DE 100 14 373 A1 and DE 200 19 210 U1. However foils glued to the glass ceramic with silicone (e.g. aluminum foil) are also known. In addition, coatings, which contain no organic ingredients, are known from DE 200 19 210 U1, e.g. comprising a paint made from borosilicate glass as glass flux and titanium or cerium oxide as pigment.
Organic based paints are used almost exclusively for observation-preventing coatings of the underside of glass/glass ceramic cook tops in practice, glass ceramic cook top. The disadvantage of these organic paints is that they have only limited heat-resistance because of their organic composition. Typical ingredients, such as silicones, polyesters or resins, decompose at temperatures above 400° C. However in operation the underside of the cook top of cooking ranges reaches up to 600° C. The organic paints decompose generally under heat load and the adherence of the coating is lost over time.
Of course the paint made from borosilicate glass flux and titanium or cerium oxide as pigments disclosed in the above-described DE 200 19 210 U1 has the required temperature-resistance. However the strength of the glass ceramic cook top is clearly reduced because of differing thermal expansion coefficients and the mechanical stresses thereby produced. Furthermore the resulting coating is not observation preventing or view blocking.

Glass ceramic or glass cook top with an IR-permeable undercoat


An airway pressure support system (2) includes a housing (4) having an air inlet opening (54), and a filter assembly (50) coupled to a plurality of receiving portions of the housing. The filter assembly is in fluid communication with the air inlet opening. The filter assembly includes a housing portion (62), a first filter media portion (64) attached to the housing portion, and first and second spring members (84, 86) attached to the housing portion, wherein the first and second spring members each have a floating portion and engage the plurality of receiving portions and cause a sealing force to be exerted against the filter assembly.

EP1537903 discloses a respirator comprises a housing enclosing a fan or air pump, protected by a coarse filter, and a fine (breathing air quality) filter. The support frame of the fine filter forms an integral component of the locating frame for the coarse filter, so that its absence prevents installation of the coarse screen, and renders the housing visibly incomplete.
DE102010047565 discloses a respiration device has an air intake aperture for intake of ambient air. A hygiene automatic backwashable cartridge (ABC) filter is positioned in the region of air intake aperture for filtering the virus and bacteria in air flow to 99%.
Many individuals suffer from disordered breathing during sleep. Sleep apnea is a common example of such sleep disordered breathing suffered by millions of people throughout the world. One type of sleep apnea is obstructive sleep apnea (OSA), which is a condition in which sleep is repeatedly interrupted by an inability to breathe due to an obstruction of the airway; typically the upper airway or pharyngeal area. Obstruction of the airway is generally believed to be due, at least in part, to a general relaxation of the muscles which stabilize the upper airway segment, thereby allowing the tissues to collapse the airway. Another type of sleep apnea syndrome is a central apnea, which is a cessation of respiration due to the absence of respiratory signals from the brain's respiratory center. An apnea condition, whether OSA, central, or mixed, which is a combination of OSA and central, is defined as the complete or near cessation of breathing, for example a 90% or greater reduction in peak respiratory air-flow.
Those afflicted with sleep apnea experience sleep fragmentation and complete or nearly complete cessation of ventilation intermittently during sleep with potentially severe degrees of oxyhemoglobin desaturation. These symptoms may be translated clinically into extreme daytime sleepiness, cardiac arrhythmias, pulmonary-artery hypertension, congestive heart failure and/or cognitive dysfunction. Other consequences of sleep apnea include right ventricular dysfunction, carbon dioxide retention during wakefulness, as well as during sleep, and continuous reduced arterial oxygen tension. Sleep apnea sufferers may be at risk for excessive mortality from these factors as well as by an elevated risk for accidents while driving and/or operating potentially dangerous equipment.
Even if a patient does not suffer from a complete or nearly complete obstruction of the airway, it is also known that adverse effects, such as arousals from sleep, can occur where there is only a partial obstruction of the airway. Partial obstruction of the airway typically results in shallow breathing referred to as a hypopnea. A hypopnea is typically defined as a 50% or greater reduction in the peak respiratory air-flow. Other types of sleep disordered breathing include, without limitation, upper airway resistance syndrome (UARS) and vibration of the airway, such as vibration of the pharyngeal wall, commonly referred to as snoring. Thus, in diagnosing a patient with a breathing disorder, such as OSA, central apneas, or UARS, it is important to detect accurately the occurrence of apneas and hypopneas of the patient.
It is well known to treat sleep disordered breathing by applying a positive airway pressure (PAP) to the patient's airway using an airway pressure support system that typically includes a mask, a pressure generating device, and a conduit to deliver positive pressure breathing gas from the pressure generating device to the patient through the mask. This positive pressure effectively "splints" the airway, thereby maintaining an open passage to the lungs. In one type of PAP therapy, known as continuous positive airway pressure (CPAP), the pressure of gas delivered to the patient is constant throughout the patient's breathing cycle. It is also known to provide a positive pressure therapy in which the pressure of gas delivered to the patient varies with the patient's breathing cycle, or varies with the patient's effort, to increase the comfort to the patient. This pressure support technique is referred to as bi-level pressure support, in which the inspiratory positive airway pressure (IPAP) delivered to the patient is higher than the expiratory positive airway pressure (EPAP).
Air filters, specifically air inlet filters, are an important part of airway pressure support systems. Not only do they protect the inner workings of the device by preventing foreign matter from entering the unit, but they also protect the patient from airborne contaminants. In the current airway pressure support system market, air filters are typically die cut pieces of filter media that sit at the air inlet of the device.
There are two types of air filters that are commonly used in airway pressure support systems. The first type of filter, referred to as a coarse particle filter, is structured to trap and filter relatively large pieces of gross particulate matter from the air before it enters the airway pressure support system. The second type of filter, referred to as a fine particle filter, is designed to be employed in combination with a coarse particle filter and is structured to trap and filter smaller pieces of particulate matter and airborne contaminants that would not otherwise be filtered by the coarse particle filter. Use of a fine particle filter in an airway pressure support system is typically optional. Thus, in practice, an airway pressure support system may be used with a coarse particle filter alone or with a combination of a coarse particle filter and a fine particle filter. When used in combination, the coarse particle filter and fine particle filter are placed in series with one another. To ensure adequate filtration, these filters should create an airtight seal between each other and with the device itself. However, in current practice, the coarse particle filter and fine particle filter media are simply placed on top of each other without any mechanism for applying a force to provide a secure seal.
In addition, coarse particle filters and fine particle filters are designed to be replaced and reimbursed by most insurance companies at different time intervals. Most commonly, the coarse particle filter is replaced every six months and the fine particle filter is replaced every thirty days.

Filter assembly and airway pressure support system employing same


Embodiments of the present invention disclose a method and an apparatus for processing micro web log messages in a micro web log system, where a unified rich media control is configured in each client side in the micro web log system, and the rich media control is obtained by encapsulating rich media information processing logics according to an operation type and a media type of rich media information. The method includes: receiving, by the client side in the micro web log system, an operation request for processing rich media information in a micro web log message; and invoking, by the client side in the micro web log system, a corresponding rich media control according to the operation type and the media type of the requested rich media information, and running the control to process the rich media information properly.

Nowadays the online instant messaging (Instant Messaging, IM) tool has been accepted by a vast majority of web users and has become an indispensable software tool of users. It is widely applied not only in the everyday entertainment but also in the users' work. Therefore, the users impose high requirements on IM software in terms of easy-to-use, stability, security, and so on. The IM software primarily enables one-to-one chat between two friends and one-to-many message chat in a group or a discussion group. With development of Internet applications, a micro blog (the word "blog" derives from and is also known as web log) applications similar to twitter are booming (twitter is a variation of an instant messaging mode used in a twitter website, and allows a user to send his/her latest news and ideas to a wireless terminal or a personalized website group by means of a short message service). Such a micro blog ("microblog" for short) service allows a user to publish about 140 Chinese characters. Such a means leads to very rapid propagation and transmission of information, and enhances the user's message chat mode from one-to-one and one-to-many to one-to-infinite. The one-to-infinite mode means that an individual can propagate a message to an infinite number of persons, and can also receive information published by tens of thousands of users. Therefore, such a technology imposes new requirements on applications used by a large quantity of users.
The IM occurs in the form of a dialog. It is characterized by the most original and the most natural interaction mode but low efficiency. It generally takes many dialogs to express a complete meaning. A blog occurs in the form of a document, and is intended to no definite audience. Its meaning is relatively complete, but initiating a blog is relatively difficult. A microblog occurs in the form of a short message, and is intended to no definite audience, but its meaning is relatively complete. Initiating a micro blog is easy. For example, a user may publish a sentence or an emoticon. Therefore, compared with IM messages and blogs, a microblog is characterized by higher efficiency of transmitting information and easier initiation. As a new product and new application extending the IM message propagation mechanism, the microblog grows rapidly.
Information presented by microblog client side software is primarily messages. In existing microblog client side products, information is primarily presented in the form of messages, but little rich media information such as images, music and video is presented. In the existing microblog client side products, the microblog content in the form of various pages such as home page are presented to users for reading in a variety of dimensions. Other types of pages such as list page (including "Search Microblog Users" and "My Audience List Message", and so on) also need to present the microblog message content.
Various pages such as message page (such as various pages or home page mentioned above) and list page in an existing microblog system present the microblog message content in different manners, and there is no reuse between different pages. In this way, to process a microblog message that carries rich media information on various pages, it is necessary to execute corresponding logics on each page respectively, which unavoidably brings many problems such as code logic redundancy, lower microblog system efficiency, and maintenance difficulty.
With development of microblog client side software products, it is necessary to combine microblog applications with various services (such as download service). The microblog system provides a common platform area for presenting various service logics. On the common platform area, control policies need to be exercised for various service logics so that various services can boost further development of the microblog.

Method and apparatus for processing micro web log messages

TENCENT TECHNOLOGY (SHENZHEN) COMPANY LIMITED


A monitor filter screen mounting device has a mounting block, a slide block slidably connected to the locating block by an adjusting screw to hold a filter screen in front of a monitor by a suspension arm, the suspension arm having one end pivoted to a cross bar on the filter screen at the top and an opposite end pivotably mounted on a round head pivot bolt and releasably retained to the slide block at a desired angular position by toothed joints. The position of the filter screen is adjusted horizontally and vertically by means of controlling the adjusting screw and the round head pivot bolt.

1. Field of the Invention
The present invention relates to a filter screen mounting device for mounting a filter screen on a monitor which allows the filter screen to be adjusted vertically as well as horizontally.
2. Description of the Prior Art
A variety of monitor filter screens are known, and widely used to protect the eyes from the glare of the monitor. These monitor filter screens commonly have a pair of hinged angle bars for mounting on the housing of a monitor in the front of the display screen thereof. Because the hinged angle bars can not bear a heavy load, a monitor filter screen of this type may drop from the monitor easily. Further, these monitor filter screens can not be adjusted neither horizontally nor vertically when installed.


A data driving circuit for a display panel. The data driving circuit includes a driving integrated circuit including a plurality of data driving units configured to supply image data to data lines of the display panel; and a package body including: a mount region in which the driving integrated circuit is mounted, logic power input patterns configured to transmit logic power to the plurality of data driving units, and at least two external auxiliary transmission lines formed in the mount region and connected to at least one logic power input pattern. Further, the at least two external auxiliary transmission lines are configured to transmit power received from the at least one connected logic power input pattern to data driving units located farthest from said at least one logic power input pattern.

1. Field of the Invention
An embodiment of the present invention relates to a driving circuit for a display device that prevents a block dim phenomenon and a noise phenomenon from occurring at edges of a screen by supplying power to data driving units located at edges of a driving integrated circuit.
2. Discussion of the Related Art
A driving integrated circuit includes a plurality of data driving units for supplying image data to data lines of a panel. The data driving units are driven by supplying logic power to the data driving units. However, the data driving units are not always driven in an appropriate manner, which results in uneven or distorted images.


A circuit synchronizes parallel data of different timing for transfer. The synchronous data transfer circuit includes a plurality of first flip-flop circuits in which the parallel data are set by a data strobe signal, a plurality of delay circuits, and a plurality of second flip-flop circuits. By configuring the second flip-flop circuits to share generation of a delay amount, the second flip-flop circuits are utilized for data synchronization by the synchronous data transfer circuit. Thus, it becomes possible to configure the delay circuits with a remarkably reduced amount of delay elements.

1. Field of the Invention
The present invention relates to a synchronous data transfer circuit, a computer system and a memory system, for reading and transferring data from a circuit chip disposed on a substrate, and more particularly a synchronous data transfer circuit, a computer system and a memory system for transferring data at the same timing even a delay amount of the data from a circuit chip differs.
2. Description of the Related Art
With the development of the semiconductor technology and a chip mounting technique, there has been provided an apparatus on which a plurality of CPUs and a large capacity main memory device are mounted on a single substrate. For example, an apparatus termed blade server is listed. In such an apparatus, as a matter of mounting, it is difficult to dispose a plurality of modules (chips) with the same distance from other modules (chips). Accordingly, the time an IC chip requesting data (or IC chip of data request source) acquires data from an IC chip of which data is requested (or IC chip of data request target) disperses. This dispersion mainly depends on the line length and the performance of the IC chip.
With the improvement of data processing speed in recent years, the range of the above dispersion becomes hard to ignore. In order to reduce the dispersion, it becomes necessary to provide a data transfer circuit. For example, in a memory device, provision of a DLL (delay locked loop) in a register has been proposed, for example, in the official gazette of the Japanese Unexamined Patent Publication No. 2003-044350, and the official gazette of the Japanese Unexamined Patent Publication No. Hei-11-086545.
The above method using such clock control only can be applied only within an IC chip. However, since fine tuning is further required in an IC chip for transfer to another IC chip being connected, the above method is not applicable without modification.
FIG. 8 shows a block diagram of the conventional synchronous data transfer circuit; FIG. 9 shows a configuration diagram of the conventional delay circuit; and FIGS. 10, 11 show explanation diagrams of the conventional transfer operation. As shown in FIG. 8, a synchronous data transfer circuit (for example, a memory controller) 100 includes a clock control circuit 110 having a frequency dividing circuit 112 for frequency dividing a clock CLK0 of data request source; a read control circuit 120 for reading data from a chip 200 (here, memory) of data request source; and a data assembly circuit 130.
The clock CLK0 is issued to provide timing at which the data request side fetches data. The frequency dividing circuit 112 in the clock control circuit 110 frequency divides the clock CLK0 synchronously with the operation speed of the chip 200 of data request target, and transmits an operation clock CLK1 to the chip 200 of data request target.
In the chip 200 of data request target, in synchronization with the clock CLK1, a data strobe signal DQS [N:0] and data DQ [0]-DQ [N] are transmitted to the read control circuit 120 in response to a read request received. These data DQ [0]-DQ [N] are serial signals.
As shown in FIG. 10, in the read control circuit 120, the data strobe signal DQS [N:0] is input into a DQS control circuit 122. The read control circuit 120 fetches the data DQ [0]-[N] into flip-flop circuits (FF00-0N) 124-0 to 124-N, using the rise up of the DQS [0]-[N] as the clocks for FFs 124-0 to 124-N.
Meanwhile, in the data assembly circuit 130, data assembly timing is specified by the clock CLK0. Therefore, conventionally, it has been configured so that the output flip-flop circuits (FF10-1N) 128-0 128-N in the read control circuit 120 fetch (synchronize) data with the clock CLK0.
As shown in FIG. 10, since the phase of the data strobe signal DQS is not consistent with that of the clock CLK0, a delay amount determined by TAP [N:0] of delay circuits (DL (b0)) 126-0 to 126-N has been added to the outputs of FFs 124-0 to 124-N, and FFs 128-0 to 128-N is applied the clock CLK0, thereby synchronizing at the fetch timing of the data assembly circuit 130.
The data assembly circuit 130 fetches the outputs of the FFs 128-0 to 128-N into flip-flop circuits (FF20-2N) 132-0 to 132-N at the timing of the clock CLK0, and performs data assembly. As such, using the delay circuits 126-0 to 126-N, the synchronization of the data DQ has been achieved.
These delay circuits 126-0 to 126-N are constituted of eight paths having delay elements 140 of 1, 2, 3, 4, 5, 6, 7 and 8 stages, respectively, and a path selector 142, as shown in FIG. 9. In this figure, reference symbols are shown only for the path of delay elements 140 depicted by the triangles, being serially connected into 8 stages. To simplify the diagram, the reference symbols are omitted for other paths of the delay elements depicted by the triangles.
Each delay element 140 is constituted of, for example, transistor having an identical delay amount. Depending on a necessary delay amount, a path is selected by tap selection TAP0 [0] of the selector 142. The outputs of FFs 124-0 to 124-N are thus delayed for the selected delay amount (DLb0 shown in FIG. 10), and then input to FFs 128-0 to 128-N.
When the data DQ [0]-[N] are, for example, 4 bits (N=4) in parallel, the necessary number of these delay circuits 126-0 to 126-N becomes 4. Meanwhile, as shown in FIG. 8, when the chip 200 of data request target outputs a 64-bit parallel signal, dispersion in each signal is substantially large, which makes it difficult to cope with the dispersion by one data strobe signal DQ [N:0].
To solve this problem, as shown in FIG. 11, data strobe signals [N:0]-[N:15] are output with different phases, for example, on a basis of 4 bits. This necessitates the provision of the read control circuits 120 in the corresponding number, namely, 16 in an exemplary case of the aforementioned 64 bits in parallel. Correspondingly, the delay amount TAP of delay circuits 126-0 to 126-N is set. The data assembly circuit 130 synchronizes these 4-bit parallel signals, assembles into a 64-bit parallel signal and transfers.
Further, when a plurality (m) of the IC chips 200 of the request target are existent on a substrate, the signal delay amount for each IC chip 200 differs depending on the line lengths and the performance of the chips 200.
For example, also as shown in FIG. 11, when the phase of the data strobe signal DQS for each IC chip 200 differs from the phase of the signal DQS shown in FIG. 10, the delay amount becomes DLbm, which is different from the delay amount of the aforementioned delay circuit shown in FIG. 9. As a result, as shown in FIG. 8, the necessary number of the read control circuits 120 is 16×m, and further, the necessary number of the delay circuits shown in FIG. 9 is 4×16×m.
As such, in the prior art, it has been necessary to provide delay circuits of which number is determined corresponding to the number of data strobe signals and the data strobe signals corresponding to the number of parallel data. Since the delay circuits are configured so as to be adjustable to arbitrary delay amount individually, the wide range of the delay amount is required. This necessitates that a large number of delay elements have been required in each delay circuit.
For example, in the aforementioned 64-bit parallel transfer, when data strobe signals are issued for every 4 bits, 16 read control circuits and 64 delay circuits are requested. In each delay circuit, because of the wide range of the delay amount, 8 delay paths and 36 delay elements (transistors) have been required, as shown in FIG. 9.
Namely, when viewed from a single read control circuit, 144 (=4×36) delay elements are required. Further, when viewed from one channel (=64 bit parallel), 16 times thereof, i.e. 2,304 delay elements are required. This necessitates a large mounting area on the circuit (chip), which impedes miniaturization and low cost. Also since the power consumption becomes large, it has been difficult to produce the chip with low power consumption. In addition, the delay element using transistors has a large dispersion of the delay amount, producing a deduced delay accuracy, and impeding high-speed synchronous transfer.

Synchronous data transfer circuit, computer system and memory system


A scan line conversion circuit is provided for simultaneously carrying out three-dimensional motion adaptive sequential scan conversion and scan line conversion. The scan line conversion circuit comprises an FIFO memory into which the video signal is input; a sequential scan conversion circuit section for converting an interlace signal sent from the FIFO memory to a non-interlace signal; an address generator into which a vertical enlargement ratio and a synchronous signal are input to generate an address as a spatial position after scan line conversion; a memory control unit for generating a memory control signal based on the address sent from the address generator; a coefficient generator for generating a coefficient for performing scan line conversion; a plurality of first multipliers for multiplying sequentially scan converted signals, sent from the sequential scan conversion circuit section, by respective coefficients; and an adder for adding signals output from the multipliers together.

A scan line conversion circuit is used, particularly in display devices such as liquid crystal displays or plasma displays, for converting input video signals to the number of scan lines in the display device.
In the scan line conversion circuit, in general, when an interlace signal has been input, the number of scan lines is converted by a method wherein, independently of whether the input signal is a static image or a moving image, the scan line is interpolated within the field and, in addition, is converted to a non-interlace signal, or by a method wherein, after conversion to a non-interlace signal by three-dimensional motion adaptive scan line interpolation is once performed, the number of scan lines is converted by resolution conversion processing (Japanese Patent Laid-Open No. 253365/2000).
FIG. 1 is a block diagram showing the construction of a conventional system, FIG. 2 a circuit diagram showing a conventional three-dimensional motion adaptive sequential scan conversion circuit, and FIG. 3 a circuit diagram showing a conventional scan line conversion circuit. As shown in FIG. 1, a video signal is input through an input terminal 301 into a three-dimensional motion adaptive sequential scan conversion circuit 302, is converted to a non-interlace signal by three-dimensional motion adaptive sequential scan conversion, and is then input into a resolution conversion circuit 304 where the interlace signal is converted to the resolution of the display device. Thus, when the three-dimensional motion adaptive sequential scan conversion is used in combination with the resolution conversion, after the three-dimensional motion adaptive sequential scan conversion is performed as shown in FIG. 2, the resolution conversion is performed as shown in FIG. 3. In this case, as shown in FIG. 1, SDRAM 303 for sequential scan conversion should be provided separately from SDRAM 305 for resolution conversion.
In the conventional three-dimensional motion adaptive sequential scan conversion circuit 302 shown in FIG. 2, the interlace signal input into the signal terminal 1 is supplied to a field memory 3, a motion detector 6, and an adder 5. Further, the output of the field memory 3 is input into a next-stage field memory 4.
The signal output from the field memory 4 is input into the adder 5 and is further input into a multiplier 12 for multiplying the signal, output from the adder 5, by ½. In the adder 5 and the multiplier 12, a signal not sent to the current field due to the interlace signal is generated from a signal of one field before the contemplated signal and a signal of one field after the contemplated signal, and interpolation processing, in the case where the input signal is a static image, is performed.
The signal output from the field memory 3 is sent to a 1H delay memory 8, an adder 10, and a multiplier 13, for one-line delaying. The signal output from the 1H delay memory 8 is sent to the adder 10. Further, the output of the adder 10 is sent to a multiplier 13 for multiplying the output signal by ½.
In the adder 10, in the input interlace signal, the upper scan line is added to the lower scan line. Therefore, a signal not to sent to the current field is generated from the upper and lower scan lines, and interpolation processing, in the case where the input signal is a moving image, is performed.
On the other hand, in the motion detector 6, the signal from the input terminal 1 and a signal, which has been delayed by two fields by the field memories 3, 4, that is, a one-frame-before signal, are input, and the motion level is detected by determining the difference between these signals. In the motion detector 6, processing is performed to transform the motion quantity into a coefficient from the motion level. The coefficient output from the motion detector 6 is sent to multipliers 19, 20 where the mixing ratio of the signal, which has been interpolated for the static image, to the signal, which has been interpolated for the moving image, is adaptively varied according to the motion quantity of the input signal.
The output of the multipliers 19 is added to the output of the multiplier 20 in an adder 21, followed by sending from the adder 21 to a line memory 102. Further, the output of the field memory 3 is input into a line memory 101. An output is selected by a switch 103 from the output of the line memory 101 and the output of the line memory 102, and the selected output is sent to a scan line conversion circuit 304.
The conventional scan line conversion circuit 304 shown in FIG. 3 comprises: an FIFO (first in first out) memory 207 for scan line conversion; an address generator 205 for generating an address; a memory control unit 202 for controlling the memory; a coefficient generator 206 for generating a coefficient from a signal obtained from the address generator 205; multipliers 211 to 214 for successively multiplying signals, produced by the scan conversion processing, by the coefficient; and an adder 215 for adding the outputs of the multipliers 211 to 214 together. The signal output from the FIFO memory 207 is sent to a 1H delay memory 208 for one-line delaying and a multiplier 214. The signal output from the 1H delay memory 208 is sent to a 1H delay memory 209 and an adder 213 which have been provided in tandem. The signal output from the 1H delay memory 209 is sent to a 1H delay memory 210 and an adder 212. The signal output from the 1H delay memory 210 is sent to the adder 211.
In the address generator 205, a spatial position after scan line interpolation necessary for scan line conversion, that is, an address, is generated from a synchronous signal of the signal input into a terminal 204 and an enlargement ratio, which is input into a terminal 203 and sets the enlargement ratio in the vertical direction. Based on the address output from the address generator 205, the memory control unit 202 decides a line to be used for resolution conversion. That is, the memory control unit 202 generates a memory control signal for the read/write control of the FIFO memory 207 and the one-line delay memories 208, 209, 210 so as to send line data necessary for resolution conversion to the multipliers 211 to 214 provided in the final stage. Further, in the coefficient generator 206, a coefficient to be sent to the multipliers 211 to 214 is generated based on the signal output from the address generator 205.
Signals equivalent to sequentially scan converted signals are input into the multipliers 211 to 214, and, in the multipliers 211 to 214, the signals are multiplied by the coefficient sent from the coefficient generator 206, and the obtained values are added together in the adder 215, whereby scan line conversion can be performed.
Japanese Patent Laid-Open No. 253365/2000 further proposes a scan line conversion circuit comprising conventional three-dimensional scan line interpolation circuit and resolution conversion circuit.
Resolution conversion circuits are classified into two types. In one type, a frame memory using DRAM is provided. In the other type, no frame memory is provided, and a line memory for several lines is provided within LSI. The provision of the frame memory is advantageous in that the conversion of a field frequency can be carried out or in that partial enlargement or reduction is possible.
Further, at the present time, in many cases, three-dimensional motion adaptive sequential scan conversion is not carried out, and, instead, at the time of resolution conversion processing in the vertical direction, offsetting the initial value for each field serves also as sequential scan conversion. In this case, however, the interlace signal is subjected to intra-field interpolation for sequential scan conversion. Also in the static image, line flicker occurs. Accordingly, the adoption of a method, wherein, in order to prevent the line flicker, resolution conversion is performed after three-dimensional motion adaptive sequential scan conversion, would be led to improved image quality.
When scan line interpolation is performed only within the field, the circuit scale may be small. In this case, however, the line flicker disadvantageously occurs also in the static image. On the other hand, conversion to an interlace signal by three-dimensional motion adaptive scan line interpolation followed by scan line interpolation by resolution conversion poses problems such as large circuit scale, high cost, and required large mounting area.
Further, as shown in FIG. 1, SDRAM 303 for sequential scan conversion should be provided separately from SDRAM 305 for resolution conversion. This poses a problem of the necessity of increasing the number of external SDRAMs. Furthermore, when the scan line conversion circuit disclosed in Japanese Patent Laid-Open No. 253365/2000 has a frame memory, the frame memory should be provided in the three-dimensional scan line interpolation and the resolution conversion circuit. As with the conventional technique, disadvantageously, the number of external SDRAMs should be increased.
Further, in the conventional circuit, when an HDTV signal is input into a display device wherein the number of scan lines is, for example, 768, a method should be used wherein, after a digital HDTV signal input at a clock rate of 74.25 MHz is converted by three-dimensional motion adaptive sequential scan conversion to a signal of the number of scan lines 1080 and clock rate 148.5 MHz, the number of scan lines is converted to 768. In this case, in the course of signal processing, operation should be performed at a higher frequency than the clock rate required for resolution in the display device.
Further, in recent years, an increase in size of liquid crystal displays or the development of large-screen plasma displays has led to a trend toward the practical use of large-screen televisions. In these large-screen televisions, there is a strong demand for an improvement in image quality in the case of the input of interlace signals, and, in addition, a reduction in cost is indispensable. Therefore, the development of a scan line conversion circuit satisfying these demands has been desired in the art.

This patent is subject to a terminal disclaimer.

Scan line conversion circuit for simultaneously carrying out three-dimensional motion adaptive sequential scan conversion and scan line conversion


Disclosed is cable restraint including a flexible and resilient cable retainer including a cable retaining body and a plurality of cable grips, the body and grips defining a cable cavity that is contoured to partially house a cable, a tie down portion extending from a tie down end of the cable retainer, the tie down portion being configured to allow association of the tie down portion with an electronic enclosure, a portability handle extending from the tie down portion, a plug enclosure including a first enclosure portion and a second enclosure portion that is hinged to the first portion, and the second portion being rotatable into enclosing association with the first portion to form and define a plug cavity for disposal of a plug and a plug opening that allows the plug to associate with the electronic enclosure, and a pull handle extending from the first end portion.

1. Field of the Invention
The disclosure relates generally to a cable restraint, and more particularly to a cable restraint that is associable with an electronic enclosure.
2. Description of the Background
Currently, users of power supplies for information technology (IT) applications encounter issues regarding cable retention, plug release, and electronic enclosure portability. Cable retention pertains to prevention of inadvertent unplugging of a cable plug. Cable retention may conflict with safety requirements that allow for a quick disconnect of a power cord from an electronic enclosure, such as a power supply, in the event of an emergency (smoke, electrical arcing, fire). One known solution for cable retention is a cable tie down. This solution has some drawbacks however, in that the user must know, and care, to allow for presence of a service loop (extra cable length) between the tie down and the electrical outlet. Often, tie downs are installed without enough (or any) service loop, making it difficult for the cable to be unplugged in the event of an emergency. Another known solution for cable retention is a "key ring." This solution also includes some drawbacks, in that during installation of the cable, the customer may sometimes forget to feed the cable though the key ring. After the power supply is turned on, one cannot feed the cable through the round ring without unplugging the cable, which requires a power down and/or risks a fault indication that must be cleared.
Referring to plug release difficulties in known plug-outlet associations, tightly packed electronic enclosure components can lead to small clearance access for grasping a plug. If a user grabs the cable by the cable portion to pull the plug out of association with the outlet, this pulling can result in damage to the power cable. In addition, with reference to electronic enclosure portability, tightly packed components can also lead to provision of little surface space for externally attached features such as cable tie downs and/or removable handles for transporting electronic enclosures. As such a cable application that can improve cable retention, plug release, and electronic enclosure portability is desirable.

International Business Machines Corporation


An apparatus for producing a silicon single crystal grown by the Czochralski method includes a main chamber having a round soulder interconnecting the upper end of a side wall and the lower end of a neck of the main chamber. The round shoulder has an inside surface so profiled as to form a portion of the periphery of an ellipse drawn about two foci which are composed of the upper end of a heater and a point of the longitudinal axis of a silicon single crystal being grown. The inside surface has a low emissivity. With the apparatus thus constructed, a silicon single crystal having a high dielectric breakdown strength of oxide film (SiO.sub.2) can be produced in a stable manner with high yield and productivity.

1. Field of the Invention
The present invention relates to an apparatus for producing a silicon single crystal grown by the Czochralski method also known as pulling method.
2. Description of the Prior Art and Related Art
Conventionally, in the production of a silicon single crystal grown by the Czochralski method (hereinafter referred to as "CZ" method), a silicon single crystal producing apparatus such as shown here in FIG. 2 of the accompanying drawings is used. The apparatus includes a main chamber 6 composed of a side wall 6a and an upper wall 7 having at its central portion a neck 4. Hot melt parts, which are used to heat the raw material to melt the same and keep the raw material in the molten or liquid state, are disposed at the central portion of the main chamber 6. The hot melt parts are composed of a quartz crucible 12 protected by a graphite crucible 11, and a heater 16 and a thermal shield or insulator 18 that are disposed around the crucibles 11, 12. The apparatus further includes a gate valve 8 disposed above the neck 4 of the main chamber 6, a pull chamber (not shown) and a crystal lift device including a lift mechanism (neither shown) that are disposed above the gate valve 8. An exhaust port 19 is connected with a vacuum pump (not shown) for controlling the pressure of an inert gas within the main chamber 6 and discharging the inert gas from the main chamber 6.
In operation, in an inert gas atmosphere held under reduced pressure, a seed crystal 26 fixed to the front end of a pull shaft 24, such as a wire, is dipped into a silicon melt 14 in the quartz crucible 12 while the quartz crucible 12 is rotating about its own axis. The seed crystal 26 while being rotated is then pulled up from the silicon melt 14 so that a silicon single crystal 28 is produced.
The single crystal produced by the CZ method is used to fabricate silicon wafers (hereinafter referred to as "CZ wafers"). It has been known that the CZ wafers have a dielectric breakdown strength of oxide film (SiO.sub.2) which is considerably lower than that of those wafers fabricated from a single crystal produced by the floating zone method (FZ method), or that of epitaxial wafers each composed of a thin silicon film epitaxially grown on a CZ wafer. However, due to various advantageous features, the CZ wafers are widely used as a material for fabricating semiconductor devices.
More particularly, in recent years, with the advance of high integration of semiconductor circuits and accompanying microminiaturization of devices, the thickness of a dielectric oxide film at the gate electrode of a MOS-LSI chip has been reduced. However, even in the case of a thin dielectric oxide film, high qualities, such as a high dielectric breakdown strength and a small leakage current during operation of a device, are required. The oxide-film dielectric strength can be deteriorated by, for example, a defect induced by a thermal history during the growth of a silicon single crystal. It has been known that the dielectric breakdown strength characteristic is improved when the crystal growth rate is slowed down to change the thermal history of a silicon single crystal (see, Japanese Patent Laid-open Publication No. 2-267195, for example). The known method, however, requires a slowing down of the crystal growth rate in order to obtain a silicon single crystal with a high dielectric breakdown strength (of oxide film). The productivity of the single crystal growth process is reduced correspondingly.
In order to overcome the foregoing problem of the prior art, the behavior of heat treatments toward the CZ wafers was investigated by the present inventors. Through the investigation, the present inventors found that the dielectric breakdown strength characteristic can be improved by a high-temperature heat treatment conducted at temperatures above 1150.degree. C. Based on that founding, the present inventors proposed a method which is capable of stably producing a silicon single crystal with high productivity by heat-treating a growing single crystal rod at temperatures above 1150.degree. C. for a certain period of time within a growing furnace, and an apparatus suitable for reducing the method into practice (see, Japanese Patent Laid-open Publication No. 5-70283).
Through a further investigation, the present inventors found the proposed apparatus not to be fully satisfactory in that due to the flat and highly emissive top wall of a main chamber such as shown in FIG. 2, the heat generated from a growing single crystal is mostly scattered and lost.

Apparatus for producing silicon single crystal


Disclosed are a method and a system for providing a mobile communication terminal with network-based location measurement by using a G-pCell database. The method and system are advantageous in that, when a mobile communication system employs a network-based location measurement scheme, the influence of repeaters is reduced to improve the positioning stability and measurement accuracy and provide more stable location-based services.

Various types of wireless communication services using mobile communication terminals are being developed in line with the remarkable development of electronic and communication technologies. Conventional services include wireless voice communication services providing mobile communication terminal users with wireless voice communication regardless of time and space, and text message services supplementing the voice communication services.
Wireless Internet services have recently commenced, which provide mobile communication service subscribers with Internet communication services via wireless communication networks, in line with the developing wireless Internet, and various technologies are being developed in connection with the wireless Internet.
Among various wireless Internet services using mobile communication terminals, the LBS (Location-Based Service) is recently drawing much attention due to the wide range of usefulness and convenience. The LBS refers to a communication service for positioning a mobile communication terminal and providing additional information based on the positioning result. The LBS is used in various fields and circumstances, including emergency aid requests, responses to crime reports, GIS (Geographical Information System) for providing information regarding adjacent regions, differentiation of mobile communication fees based on location, traffic information, vehicle navigation, circulation control, location-based CRM (Customer Relationship Management), etc.
Location measurement schemes for providing mobile communication terminals with the LBS include network-based schemes relying on the propagation environment (i.e. cell radius of a base station in a mobile communication network) to measure the location of mobile communication terminals and confirm the location on a software basis, handset-based schemes using GPS (Global Positioning System) receivers mounted in mobile communication terminals, and hybrid schemes combining both schemes.
The A-GPS scheme is one of the handset-based schemes, and is compatible with both European GSM (Global System for Mobile Communication)-based networks using the TDMA (Time Division Multiple Access) wireless access scheme and IS-95-based networks using the CDMA (Code Division Multiple Access) wireless access scheme. According to the GSM wireless scheme, mobile communication terminals are positioned by transmitting/receiving messages through an OMA SUPL (Secure User Plane Location) interface between the mobile communication terminals incorporating GPS receivers and the SPC (SUPL Positioning Center) within the GSM network and through an SUPL POS for A-GPS location measurement within the OMA SUPL (i.e. GSM A-PGS protocol incorporating an RRLP (Radio Resource Location Protocol)). This type of positioning is very accurate because satellite signals are received from at least four GPS satellites to measure the location. The A-GPS system includes an SPC for receiving satellite signals received by mobile communication terminals and calculating the location, and an SLC (SUPL Location Center) for processing the calculation based on information regarding base stations within the GSM mobile communication network or associating the information with other systems.
The E-OTD (Enhanced Observed Time Difference) scheme is a representative network-based location measurement scheme, and has been standardized through LCS Release 98 and 99 by the GSM standardization committee of the European TDMA-based GSM scheme using the TDMA wireless access standard. According to the E-OTD, signals received from at least three base stations by a mobile communication terminal are used to calculate the difference in time of arrival and distance and determine the location. In other words, the E-OTD scheme combines various time difference concepts, including OTD (Observed Time Difference), RTD (Relative Time Difference), GTD (Geometric Time Difference), etc. for network-type location calculation.
The OTD refers to the difference in time spent by signals from two base stations to reach a mobile communication terminal, and can be obtained by measuring the UE Rx-Tx time difference type 2 parameter by a GSM-based mobile communication terminal.
The RTD refers to a parameter used to obtain the difference in starting time of signals transmitted by two base stations, and can only be measured by equipping the base stations with a separate measuring device, i.e. LMU (Location Measurement Unit). This means that, in order to obtain the key parameter necessary for network-type location calculation recommended by the GSM standard, i.e. "GTD=OTD-RTD", not only the OTD, but also the RTD must be obtained for the E-OTD-type network location calculation.
According to network-based location measurement technology, data (PPM, OTD, RTD, etc.) measured by a mobile communication terminal and an LMU is transmitted to a location measurement server according to a protocol (IS-801, RRLP, RRC, etc.) agreed upon between the mobile communication terminal and the server, and the data (PPM, OTD, RTD, etc.) measured by the mobile communication terminal is used by the location measurement server to measure the location of the corresponding mobile communication terminal. The location measurement server conducts network-type location measurement (i.e. the server side measures the location of the terminal that has requested location measurement, location measurement schemes using GPS satellites being excluded), and transmits the result to the requesting party (SLC, CP (Contents Provider), mobile communication terminal that has requested the service, etc).
The network-based location measurement technology includes a cell ID scheme using the base station radius cell, an AOA (Angle of Arrival) scheme according to which the base station receives signals from a mobile communication terminal and calculates the LOB (Line of Bearing) to calculate the location, a TOM (Time of Arrival) scheme according to which the location of a mobile communication terminal is calculated based on the time of arrival of radio waves from at least three base stations, and a TDOA (Time Difference of Arrival) scheme according to which the difference in time of arrival of pilot signals received by a mobile communication terminal from three base stations is measured to calculate the difference in distance between the base stations so that the point of intersection of two resulting hyperbolas is determined as the location of the mobile communication terminal.
However, the above-mentioned conventional network-based location measurement schemes have the following problems.
Firstly, when data regarding parameters (i.e. time and distance) measured by a mobile communication terminal or the mobile communication network is used for triangulation or to calculate the point of intersection of hyperbolas, repeaters heavily affect the result. This means that, if repeaters are used, data measured by a mobile communication terminal regarding the time and distance between the base station and the mobile communication terminal is delayed relative to the original data. This degrades the location measurement accuracy.
Secondly, in the case of triangulation using time and distance measurement parameters in an asynchronous mobile communication network (GSM or W-CDMA), not only the OTD measured by a mobile communication terminal, but also the RTD value measured by an additional LMU equipped with separate GPS equipment must be measured to obtain the location measurement result from the formula. Considering that it has little merit per investment to additionally install LMUs in the entire mobile communication network for the network-type location measurement, triangulation-based network location measurement cannot be used in a region having no LMU installed therein.
Thirdly, when a base station is rearranged, the latitude and longitude data of the rearranged base station is not instantly reflected. This means that there is no confirming whether or not the latitude and longitude data referred to for location measurement is identical to the latitude and longitude data of the rearranged base station.
Fourthly, since each network-based location measurement technology has different characteristics regarding the mobile communication base stations and sectors, excessive human and material resources are used to optimize parameters, which are differently used by base stations or sectors, for the purpose of improving the location measurement accuracy. This slows down the commercialization.

Method and system for providing location measurement of network based to mobile communication terminal by using G-pCell database


An oligonucleotide probe is disclosed, the probe including an oligonucleotide, a fluorescer molecule attached to a first end of the oligonucleotide and a quencher molecule attached to the opposite end of the oligonucleotide. The probe is rendered impervious to digestion by the 5′→3′ exonuclease activity of a polymerase and the 5′→3′ extension of by a polymerase. The invention also includes methods for performing combined PCR amplification and hybridization probing, one such method including the steps of contacting a target nucleic acid sequence with PCR reagents and an oligonucleotide probe as described above, and subjecting these reagents to thermal cycling. One preferred refinement of the above method further includes the addition of a strand displacer to facilitate amplification. Additional similar combined PCR hybridization methods are disclosed, such methods not requiring probes having their 5′ ends protected, wherein (i) the polymerase lacks 5′→3′ exonuclease activity, (ii) a 5′→3′ exonuclease inhibitor is included, and (iii) an exonuclease deactivation step is performed.

This invention relates generally to the field of nucleic amplification and probing, and more particularly, to methods and compositions for performing PCR and probe hybridization using a single reagent mixture.
Nucleic acid amplification techniques provide powerful tools for the study of genetic material. The polymerase chain reaction (PCR) in particular has become a tool of major importance finding applications in cloning, analysis of genetic expression, DNA sequencing, genetic mapping, drug discovery, criminal forensics, and the like, e.g., Innis et al. in PCR Protocols A guide to Methods and Applications, Academic Press, San Diego (1990); and U.S. Pat. Nos. 4,683,195, 4,683,202.
For many important applications, in addition to amplifying a target nucleic acid sequence, it is desirable to further characterize such sequence by treatment with a nucleic acid hybridization probe, i.e., a labeled single stranded polynucleotide which is complementary to all or part of the target sequence, e.g., Nucleic Acid Hybridization, Hames et al. Eds., IRL Press, Oxford (1985). Probe hybridization may provide additional sequence selectivity over simple PCR amplification as well as allowing for the characterization of multiple sequence sites within the target nucleic acid sequence in an independent manner.
Traditionally, PCR and probe hybridization processes have been performed as separate operations. However, it is highly desirable to perform both the PCR and the probe hybridization reactions in a combined manner using a single reagent mixture containing both PCR reagents and probing reagents. There are several advantages realized by combining the PCR and the probing process: (i) only a single reagent addition step is required, thereby allowing the combined reactions to proceed without having to open up the reaction tube, thereby reducing the opportunity for sample mix-up and sample contamination; (ii) fewer reagents are needed; (iii) fewer reagent addition steps are required, making automation more straight forward; and, (iv) in the case of in situ methods, there is no requirement to take apart a sample containment assembly used to protect the integrity of the cellular sample during thermocycling.
One existing method which combines PCR with hybridization probing in a single reaction is the technique know as "Taqman", e.g., Holland et al, Proc. Natl. Acad. Sci. USA 88: 7276-7280 (1991). In the Taqman assay, an oligonucleotide probe, nonextendable at the 3′ end, labeled at the 5′ end, and designed to hybridize within the target sequence, is introduced into the PCR reaction. Hybridization of the probe to a PCR reaction product strand during amplification generates a substrate suitable for the exonuclease activity of the PCR polymerase. Thus, during amplification, the 5′→3′ exonuclease activity of the polymerase enzyme degrades the probe into smaller fragments that can be differentiated from undegraded probe. While a significant improvement over prior methods, the Taqman assay has a number of important drawbacks that limit its utility including (i) the requirement that the polymerase enzyme used for the PCR must include a 5′→3′ exonuclease activity, (ii) the 5′→3′ exonuclease activity must be able to efficiently digest a dye-labeled nucleotide, and (iii) the detectable product of the digestion is a small, rapidly diffusable species which may impact the ability to spatially locate the target sequence when applied to in situ methods.
A second existing method which combines PCR with probing in a single reaction is that disclosed by Higuchi et al. in Biotechnology, 10: 413-417 (1992). In this method, ethidium bromide is added to the PCR reaction and, since the fluorescence of the ethidium bromide increases in the presence of double stranded DNA, an increase in fluorescence can be correlated with the accumulation of double stranded PCR product. However, this method does not provide any sequence specificity beyond the PCR reaction and is limited to the detection of a single sequence site within the target nucleic acid sequence.
A third method which allows for combined amplification and probing steps is that of Bagwell in EP 0601889A2. The probe in Bagwell's method includes a nucleotide sequence which has (i) a segment complementary to the target nucleic acid and (ii) a segment capable of forming one or more hairpins. The probe also includes covalently attached fluorescer and a quencher molecules located such that when a hairpin is formed, the fluorescer and quencher are in close enough proximity to allow resonance energy transfer between them. This method has the significant short coming that it limits the possible probe sequences to those capable of forming a hairpin structure. Moreover, the kinetics and thermodynamics of probe-target binding will be unfavorably affected by the presence of the hairpin structure.

Methods and reagents for combined PCR amplification