A modular integrated rail system for a firearm includes an upper hand guard having a dovetail rail extending longitudinally between the forward end and the rearward end thereof. The rearward end of the upper hand guard includes a clamp that removably secures the upper hand guard to the barrel nut of the firearm. The rearward end of the upper hand guard and the clamp include clamping surfaces configured to cooperatively engage the outer surface of the barrel nut. The upper hand guard further includes mounting channels that are used to mount accessories, such as a lower hand guard or a grenade launcher, to the upper hand guard. The rear end of the upper hand guard includes alignment tabs for automatically aligning the dovetail rail of the upper hand guard with the dovetail rail of the upper receiver during mounting. The tabs also prevent rotation of the rail system relative to the upper receiver during use.

The present invention relates generally to modular interface assemblies for weapons. More specifically, the present invention relates to a modular interface assembly that includes a rail system, which is clamped around and supported by the barrel nut of a firearm and further includes an optional sleeve element that can be installed to extend from the upper receiver rail over the top of the clamped rail system.
As the field of combat and commercial weaponry expands, numerous add-on enhancements have become available for attachment to standard firearms, thereby significantly upgrading the capability of the firearm. Of particular interest in the area of combat weapons is the well-known M16/M4 weapon system generally indicated at 10 in FIG. 1 (M16 and M4 are trademarks of Colt Defense, Inc.). The M16 has been in service for a number of years and will continue to be a popular rifle both in U.S. and foreign militaries for the foreseeable future. Generally, the M16/M4 weapon 10 includes a lower receiver 12, upper receiver 14, butt stock 16, and barrel 18. Referring to FIGS. 1 and 2, the barrel 18 is attached to the front of the upper receiver 14 by inserting the rear end of the barrel into a barrel-receiving receptacle 20 at the front end of the upper receiver 14. A pin 22 on the barrel 18 aligns with a notch 24 in the barrel-receiving receptacle 20 to insure that the barrel 18 is properly aligned with the upper receiver 14 when the barrel is installed into the barrel-receiving receptacle 20. The barrel 18 is held in assembled relation with the upper receiver 14 by a barrel nut 26 that is threaded onto the outside surface of the barrel-receiving receptacle 20. In this manner, the barrel nut 26 is rigidly engaged with the barrel receiving receptacle 20 and the upper receiver 14 of the weapon 10, while also serving to retain the barrel 18 in its installed position. A "delta ring" 28 (FIG. 1) encircles the barrel nut 26 and provides a spring loaded ring for attachment and support of the M16/M4 standard hand guards (not shown) between the delta ring 28 and a forward receptor cap 32 mounted at the front end of the barrel 18. A gas tube 29 extends from the upper receiver 14 to the receptor cap 32 at the front end of the barrel 18. The standard hand guards, when installed, encircle and protect both the barrel 18 and gas tube 29.
The newer models of the M16/M4 weapons 10 further include a mil-std 1913 dovetail rail 30 extending along the top of the upper receiver 14. This integrated receiver rail 30 provides a convenient mounting point for many types of enhancement devices such as scopes and other sighting devices. However, space on the upper receiver rail is limited, and many military personnel often have multiple sighting devices that are each tailored to perform in different combat situations. In addition, there are a variety of lighting devices, handgrips, etc. that could also be attached to the weapon 10 for enhanced use of the weapon 10. The difficulty is that there is simply not enough space on the integrated rail 30 provided on the upper receiver 14 to accommodate all of the desired accessories. Accordingly, the increasing development and refinement of laser sights, infrared lighting, visible lighting, night vision, and specialized scopes and magnifiers, and other accessories continues to drive the need for versatile and reliable integration systems that include additional mil-std 1913 dovetail rails positioned above or around the barrel 18 of the weapon 10 that can support this important equipment and yet stand the test of rugged military use and abuse.
As can be appreciated, the problem in attempting to meet this requirement by integrating a variety of modular attachments onto firearms is the inherent conflict between the unimpeded function of the gun barrel 18 and the desirability to be able to use the barrel 18 and/or other parts of the weapon 10 as a mounting platform for the desired modular attachments. The conflict arises from the fact that any additional weight and/or shocks that may be applied to the barrel from external attachments and accessories can adversely affect the alignment and reliable function of the weapon. It is also a requirement of most military contract specifications that the existing weapon not be modified for the purpose of mounting accessories to the weapon, thus insuring that such integration systems can be used universally on all existing weapons.
Accordingly, it has been realized that to obtain the best and most reliable performance of a rifle, the gun barrel 18 should be physically isolated relative to any other accessories or mounting systems, i.e., "free floating". It is most preferred that nothing be attached to the gun barrel 18, thereby isolating the gun barrel 18 physically from outside elements and eliminating any bending and "droop" along the longitudinal axis of the barrel 18 that may be caused by excess weight near the front end of the barrel. The ideal mounting arrangement for modular attachments is one where the attachment is completely isolated from the gun barrel. This isolation serves two functional purposes. First, in operation, the temperature of a gun barrel can quickly rise to 900° F. This type of heat, as well as the physical shock transferred through the gun barrel during firing, can damage or destroy any attachments that are mounted in direct contact with the barrel. Additionally, the heat generated by the gun barrel is transferred to any devices mounted thereon, such as hand guards, thereby resulting in the direct transfer of enough heat to burn a user's hands and to potentially interfere with the operation of other attachments. The second issue compounding this problem is that if accessories are to be supported by the gun barrel, the gun barrel may need to be enlarged to support the added weight and shock forces. For example, a grenade launcher attachment not only introduces additional weight to the barrel, it also introduces substantial recoil in operation that is transferred into the firearm through the barrel. This in turn means more cantilevered stress on the barrel where it is joined with the upper receiver. The combination of heat and force applied to the barrel tends to pull the barrel chamber out of alignment with the bolt lead, thereby causing bolt lug and extractor failure, ultimately jamming the firearm.
Attempts have been made in the prior art to minimize the amount of weight that is transferred into the barrel by accessory mounts, wherein the accessory mounts are attached to both the upper receiver and the barrel. For example, U.S. Pat. No. 5,826,262 discloses a tubular rail received around the gun barrel. The rear end of the rail system is supported on the delta ring, which is secured around the barrel nut, i.e. supported on the receiver body. However, the front end of the rail system is supported on the receptor cap 32 mounted to the front end of the barrel 18 adjacent the front sight 34. Accordingly, the weight of any accessories mounted on the rail system is still partially carried by the front end of the barrel. In the case of a grenade launcher, the weight is considerable and could affect performance of the weapon.
There are also rail interface systems in the prior art that are supported directly from the upper receiver of the weapon and avoid attachment of any accessories to the gun barrel. For example, in U.S. Pat. No. 5,343,650 an extended rigid interface frame with upper and lower rails is shown joined to a firearm receiver and extends forward about the firearm's barrel to a head assembly replacing the firearm's normal front sight. A weaver type interface return portion is provided below the barrel from the head assembly to the receiver. A yoke braces the extended rigid frame receiver sleeve to the forward portion of the firearm's receiver. The distal end of the extended rigid frame receiver sleeve terminates in a front sight housing that connects the upper and lower rails and provides a housing for advanced laser and sensor components, and the standard front sight bead. The front sight housing is self-supported by the connection of the upper and lower rails running back to the yoke and secured to the top of the receiver. The barrel of the rifle is free floating in that it does not touch the extended rigid frame receiver sleeve in any manner. This permits greater shooting accuracy and protects sensitive electrical components within the front sight housing by isolating the front sight housing from the heat generated from the barrel.
In another example, the ARMS SIR system as disclosed in U.S. Pat. No. 6,499,245 also derives its support by extending a dovetail sleeve rearwardly over the dovetail rail 30 on the upper receiver 14 of the firearm and supporting the rail system in a cantilevered arrangement around the barrel (ARMS and SIR are registered trademarks of Atlantic Research Marketing Systems, Inc.). The ARMS SIR system has been well received by the military and has become popular with many military branches. However, it has been noted in field use that the dovetail sleeve introduces an added height to the existing mil-std 1913 rail of the receiver, and that this added height is not always desirable, particularly for some sighting devices that are popular with the military.
Accordingly, there is perceived to be a need for a unique modular interface rail design for mounting accessories to a firearm that supports the accessory without introducing loads or additional stresses into the barrel of the firearm while also providing an extended longitudinal rail for mounting accessories that is co-planar and in linear alignment with the existing interface rail 30 on the upper receiver 14 over the entire length of the top of the firearm, and in addition providing an optional sleeve over the upper receiver 14 for added height if desired.

Modular integrated rail assembly for firearms


A method and data processing system for accessing an entry in a memory array by placing a tag memory unit (114) in parallel with an operand adder circuit (112) to enable tag lookup and generation of speculative way hit/miss information (126) directly from the operands (111, 113) without using the output sum of the operand adder. PGZ-encoded address bits (0:51) from the operands (111, 113) are applied with a carry-out value (Cout48) to a content-addressable memory array (114) having compact bitcells with embedded partial A+B=K logic to generate two speculative hit/miss signals under control of a delayed evaluate signal. A sum value (EA51) computed from the least significant base and offset address bits determines which of the speculative hit/miss signals is selected for output (126).

Field of the Invention
The present invention is directed in general to the field of electronic circuits. In one aspect, the present invention relates to a memory access apparatus and method for accessing memory by generating way hit/miss information from base and offset address components.
Description of the Related Art
Data is stored or read from memory at a memory address that is typically computed by adding a base address to an offset address in order to arrive at an effective address for the data. For example, base+offset addressing is used to address memory within data caches, instruction caches, and table-lookaside-buffers (TLBs) as well as data or instructions within other CPU memory units. With such memories, the base and offset values of the memory address are used to determine if the addressed information is stored in the memory. Typically a TAG memory makes this determination by storing addresses of stored information and comparing a TAG portion of the address to the stored addresses to determine if the stored information is present in the memory. A determination that the stored information is present is typically called a hit, and the processing required to make this determination is generally time-consuming because the memory storing the TAG portions of the addresses must be accessed and then compared. In addition, the addition of the base and offset values typically performed to arrive at the effective address usually takes at least two cycles to access the memory. In the first cycle, the base and offset addresses are added, and in the second cycle, the memory is accessed. Since at least two cycles are used to access the memory in a traditional processor, the cycle immediately following a load instruction cannot use the result of the load operation. This delay is referred to as "load latency." Load latency is a performance limitation factor in traditional processors. Load latency often manifests itself in a pipelined processor as a load-use penalty with the load results being unavailable for two machine cycles.
Accordingly, a need exists for an improved memory access circuit, system and method that addresses various problems in the art that have been discovered by the above-named inventors where various limitations and disadvantages of conventional solutions and technologies will become apparent to one of skill in the art after reviewing the remainder of the present application with reference to the drawings and detailed description which follow, though it should be understood that this description of the related art section is not intended to serve as an admission that the described subject matter is prior art.

Memory management unit TAG memory with CAM evaluate signal


Methods, systems, and computer-readable media for analyzing a domain are provided. The method includes defining a mask plane that includes a first dimension of a first number of voxels and a second dimension of a second number of voxels, and selecting a plurality of first angles for orientating the mask plane in the domain with respect to a first axis. The method also includes for each one of the plurality of first angles selected populating, using one or more processors, sum cubes associated with each one of a first plurality of subject voxels. The method also includes selecting a plurality of second angles, and for each one of the plurality of second angles selected, calculating a planar sum for each one of a second plurality of subject voxels selected.

Seismic modeling and simulation of subterranean formations may employ a process in which collected seismic data may be used to construct a two-dimensional or three-dimensional model of the formation. In such processes, the seismic data may be in the form of properties of signals that are received after being transmitted through an area of interest, for example, around a borehole, vertically and/or angled from the surface (e.g., land, the ocean floor, etc.), or in other locations. The signal characteristics may include direction, time taken to propagate through the area, amplitude, among a variety of others possible. This information may be combined with knowledge of rock properties and/or the subterranean formation to link these properties to the structure of the formation. Accordingly, the properties of a number of signals may be combined to construct an approximation of the structure of the formation.
The recognition of certain features, e.g., faults, horizons, etc., may be of particular interest in such modeling. However, in some cases, the area of interest, number of signals, number of features, or a combination thereof may be large, such that it would be useful to employ computers to automatically detect, enhance, and extract information about the features. One way this is done is by using a windowed Radon transform. Briefly, a system may discretize the area of interest into pixels (two dimensions) or voxels (three dimensions). The system may then sum the signal property (e.g., intensity) of the pixel or voxel with those of a certain number of its neighbors in a range of directions. If the intensity sum for the subject pixel or voxel is higher in one direction, it may be flagged as part of a possible feature. Next, the system may look for the next pixel or voxel that is part of the same feature or may simply move on to the next pixel or voxel in sequence. Such techniques may employ a "brute force" method, such that a line or plane extending from the subject pixel or voxel sweeps across a range of angles, calculating the intensity sum for each angle, and then moving on to the next pixel or voxel and repeating the process.


A current-detection coil for a current transformer operating by the Rogowski principle has an insulating annular body made of a fiber-reinforced thermosetting plastic and manufactured as a single-piece solid body in a divisible mold. The parting line of a mold serving for manufacturing the annular body is in a flattened region on the outer periphery of the annular body and on a flattened region on the inner periphery. The annular body allows current transformers operating by the Rogowski principle to be manufactured that are capable of measuring in the entire range of currents that may occur and are resistant to mechanical and thermal stresses.

A previously proposed current-detection coil is discussed in the British EPE Journal, vol. 3, No. 1, March 1993, pp. 51-59, where simple solid toroids or flexible cut-open coil bodies are proposed as annular body shapes. Furthermore, the use of circular cut-open plastic rings with a rectangular cross section, which can be bent for placement on a conductor and then return to their annular shape, is discussed therein. Each of the various annular bodies used in the current-detection coils discussed in the EPE Journal are used to form a Rogowski current transformer.
Although all of the current-detection coil embodiments of the EPE Journal can be made to exhibit good transfer characteristics, when they are mass-produced, however, virtually each unit will exhibit different transfer characteristics and therefore individual adjustments or electronic correction circuits must be used to achieve uniform results. The effect of small deviations of the geometry of Rogowski current transformer coils from the ideal shape on their transfer characteristics is known. One solution proposed is to use polished straight ceramic rods as winding supports. See German Journal Technische Mitteilungen AEG-Telefunken 60 (1970) 7, pp. 444-448. Because, however, a current-detection coil operating according to the Rogowski principle must surround a conductor on a closed path, the use of straight ceramic rods requires that a closed current-detection coil be made of four partial coils. In this case, the rectangular junction of the component parts causes considerable interference in the electromagnetic field. In order to remove this influence, the junction points are made field-free through the use of a shielding. The desired accuracy of the Rogowski current transformer is therefore associated with considerable expense for the manufacture of the current-detection coils.

Current-detection coil for a current transformer


When multiple motors (40) share a common DC power supply (10) via a bus (20), a system and method is provided for managing the effect of regenerative energy caused by a motor. The regenerative energy may be sensed, e.g., by a power electronics controller (50, 50′). In response, one or more of the other motors currently in operation may be controlled to operate according to a reduced power factor. Thus, the excess energy is consumed without adding any new components or increasing the amount of work performed in the system.

A simplified configuration of a system in which multiple motors share a common DC power supply is illustrated in FIG. 1. As shown in this figure, each of the motors is connected to the DC power supply via a DC bus. FIG. 1 further shows that the DC bus includes a capacitive element, e.g., one or more capacitors connected across the DC output terminals of the power supply.
In such systems, deceleration of a motor causes energy stored in the motor and/or load to return to the DC bus power supply. This energy, which is referred to as "regenerative energy," is typically collected at the bus capacitor(s). In turn, the regenerative energy causes the voltage in the capacitors to increase, potentially to destructive levels.
An existing method for handling excess energy at the DC bus is to add a "slave axis," as illustrated in FIG. 2. In such a system, the slave axis is used for storing energy or performing useful work, as necessary, to maintain a balance of power at the DC bus. Thus, the slave axis performs additional work, either by contributing to the overall effort of the system or by storing energy, e.g., using a mechanical fly-wheel.
However, adding this slave axis to the system creates additional weight for the system. In many types of systems, any additional weight becomes problematic. For example, in the aerospace industry, it is important to keep system weight to a minimum.
Other existing methods for handling excess energy require the addition of electronic and/or mechanical components specifically for the task of energy dissipation. One example, which is commonly referred to as "dynamic braking," uses an additional power resistor and switch to dissipate energy. Although such methods may solve the problem of destructive voltage levels at the DC bus caused by regenerative energy, they impose additional costs on the system in the form of added components, increased weight, and other thermal concerns.
Thus, it would be advantageous to control the regenerative energy in the DC bus without adding any non-critical system components, such as a slave axis or dynamic braking components.

System and method for regenerative energy control when multiple motors share a common power supply


An optical fiber cable includes a plurality of loose tubes, each of which contain at least one optical fiber therein. The loose tubes are constructed of a polymer with the tubes arranged in at least two layers, each of which are rotated in the same uni-helical direction and are ranged having substantially the same lay length. A jacket encases the loose tubes, where any one of the optical fibers in one of the loose tubes is accessible when the jacket is opened and the at least two layers are untwisted.

High fiber count cables are employed for installations that require a large number of fiber optic connections such as data centers or other computer/data transfer intensive applications. In order to address this need, high fiber count optical fiber cables are produced having increased fiber counts, typically in the range of 216 through 432 fibers.
A typical high fiber count cable design includes an outer cable jacket, within which are multiple loose tube fiber cables arranged in layers around a central strength member. For example, a typical outer cable jacket may include eighteen loose tubes (one six tube layer and one twelve tube layer) to thirty six loose tubes (with an additional eighteen tube layer) therein. Within each loose tube fiber cable, another twelve UV coated optical fibers are loosely arranged. Loose tube fiber cables refer to an arrangement where the multiple individual optical fibers inside the loose tubes are substantially un-connected to the tube. Such an arrangement effectively bundles a large number of optical fibers into a relatively condensed outer jacket.
However, the current arrangement for such cables maintains significant drawbacks. Typically, there are two competing interests in the designs of such high fiber count optical fiber cables. The first concern is that users of such cables would like to have mid-span access to a fiber contained within one of the loose tubes within the larger high fiber cable. This is a desirable quality to allow for more complex installations where a particular fiber may need to be accessed mid-span of the larger cable, rather than at the end of the cable where the fibers are more easily accessed. But, because of the general high fiber count construction, mid-span access proves to be difficult.
For example, the lower (inner) layers of the loose tube cables are typically inaccessible due to the stranding of the internal elements. To access a lower layer, obviously a user would need to move the outer layers first. However, as with many cables having internal components, the internal components need to be stranded within the outer jacket for a number of reasons, such as allowing for easy spooling and un-spooling during installation and preventing vast length differentials between internal components when the larger cable is spooled or bent extensively during installation.
A typical stranding for such high fiber count optical fiber cables having a single layer is for a helical stranding arrangement such that all the internal components of that layer rotate helically in a single direction along the length of the cable. In larger high fiber count cables each of the layers are wound in opposite directions. However, as noted above, such a helical stranding makes it difficult and at times impossible to achieve mid-span access of many of the fibers within the lose tubes inside the cable. For example, with high fiber count optical fiber cables having multiple layers of loose tube optical fiber cables therein, mid-span access to fiber requires an un-twisting of the stranded cable, which is unworkable when there are multiple layers of loose tubes, each of different lay length and direction. Additionally, the problem of un-twisting is exacerbated because the loose tubes employ plastics having a high Young's modulus which are particularly stiff.
Prior art attempts to address this problem use an alternative S-Z stranding of the loose tubes within the cable, where the loose tubes reverse stranding direction periodically along the length of the cable. This allows easier mid-span access to the fibers because it is easier to unwind the tubes in the upper layers allowing the user to reach the fibers contained in the inner layers (center of the outer jacket). However, S-Z designs are less robust stranding designs that tend to unravel or un-rotate over time as a result of temperature expansion, and by excessive handling of the cable during installation/relocation. Use of S-Z may incorporate binders to hold the S-Z rotation, but such binders add yet another barrier to mid-span access of a fiber, add cost weight and additional stranding steps, and also results in undesirable crushing stresses on the fibers inside the tubes.
Another related drawback associated with the prior art in high fiber count optical fiber cables is that prior art cables typically use relatively high Young's modulus plastics for the loose tube construction. High Young's modulus plastics provide a greater level of protection to the fibers contained therein. However, in addition to being difficult to unwind for mid-span access due to their inherent stiffness, the use of such high modulus plastics exhibit higher temperature contraction reactions requiring, the tighter lay lengths in response. However, such designs, in addition to making it more difficult to achieve mid-span access in helical stranded cables because of the tighter winding that needs to be unwound, such high modulus plastics impart some inherent fiber bend stresses which is locked in during manufacture by the twisting of the tubes, and may be augmented by the use of water blocking gels that assist in locking in this strain. Such strain increases attenuation in the fibers and further hampers efforts for mid-span access.

Optical fiber cable with system and method for mid-span access


Apparatus and methods for picking three dimensional objects from images depicted on a video display. The displayed objects are selectively rerendered. During such rerendering the object pixels are compared in depth to the data in a Z buffer for determining visibility. The number and size of the objects subject to rerendering by the rasterization processor is constrained by using the front end graphics processor to define object extents and by rerendering only extents which have been clipped to the boundaries of the pick window. The rerendering operation does not alter the three dimensional graphics image stored in and repetitively scanned from the frame buffer. Selection between multiple objects within the pick window can include a weighted comparison using a pick plane memory to store visibility data by object.

The present invention relates in general to an interface between a user of a graphics display system and the information portrayed on a video display of the system. More particularly, the invention defines apparatus and methods for selecting among visible objects displayed in a three dimensional rendering on a frame buffer based graphics video display system.
The rendering of three dimensional graphic images on a two dimensional raster scan display screen conventionally involves the use of a frame buffer to store the final image for raster display synchronized scanning. This graphics display architecture includes a number of memories which individually store window information, general masking information, and Z buffer information. A representative system is described in U.S. Pat. No. 4,609,917.
In the context of such technology, the frame buffer stores by pixel position image information which is synchronously scanned, converted and rendered on the video display. The information in the frame buffer is the end product of numerous data manipulations involving, among other things, the generation of three dimensional objects by a processor, depth comparisons referenced to a Z buffer and used to render into the frame buffer only the objects or portions thereof which are visible, clipping of the objects to windows, and clipping of the objects to general masks. Consequently, the two dimensional images resident in the frame buffer cannot in most cases be directly related back to the three dimensional objects from which they originated.
This lack of direct correspondence between the original objects and the pixel data in the frame buffer is attributable to the architecture of representative graphics systems. This architecture conventionally includes a front end graphics processor, which defines each object by boundary parameters, and a rasterization processor, which receives such object parameters and transforms those parameters into data by pixel. Thus, three dimensional objects are transformed into pixel data along raster scan referenced lines. The rasterization processor incorporates the clipping effects of window planes, the boundaries defined by masking planes, and the visibility by pixel of three dimensional objects in relation to other objects, as last defined in the Z buffer. The rasterization processor writes the pixel information into the frame buffer only after the masking conditions are satisfied and the pixel of the object subject to writing has a Z axis depth greater than the previous information at that pixel position.
This convention of three dimensional graphics display system architecture and practice makes it very difficult to perform a "pick" operation. The pick operation involves a selection, using a mouse controlled cursor or comparable pointing apparatus, by a human user of an object as visibly rendered on the video display. In such a context, the graphics display system must be able to identify which of the numerous objects rendered by the rasterization processor is the visible object portrayed on the display screen. Of course it is possible to rerender the whole image in the frame buffer and in doing so identify the correspondence of the cursor to the object. Unfortunately, that is not a practical solution in most three dimensional renderings given the extensive computational effort required to regenerate relatively complex images and the user's expectation of immediacy in the pick process.
An additional problem which exists in the contemporary practice of implementing a pick function in graphics system hardware, given the unacceptable slow speed of software implemented pick operations, is the inability of the pick hardware to discern visibility of the object as rendered into the frame buffer and depicted on the video display. Conventionally implemented pick hardware will identify position correspondence between a cursor and an object subject to rendering, but not whether that object is visible or hidden. Though new three dimensional graphics standards now specify that the pick operation should be capable of identifying the visible object at the position of the pick, an efficient implementation of such capability has yet to be defined.
Accordingly, there exists a need for graphics system apparatus and methods by which the pick operation can be implemented to identify visible objects in a three dimensional image generated on a display at a speed consistent with user expectations.

Apparatus and method for relating a point of selection to an object in a graphics display system

International Business Machines Corporation


Pressurized liquid solvent is dispensed on specific areas of a semiconductor wafer. A vented accumulator removes bubbles introduced by an upstream needle valve. The accumulator outlet leads directly to a dispensing tip, without further pressure drop.

The present invention is directed to the application of pressurized liquid to a semiconductor wafer or similar workpiece, and more particularly concerns the removal therefrom of vapor introduced during regulation of the liquid flow.
During the manufacture of integrated circuits on a semiconductor wafer, a photoresist applied to the top surface of the wafer. The wafer is placed in a chuck of a stepper tool for exposing the photoresist in a pattern which defines the devices and interconnection of the circuits to be formed on the wafer. To avoid contamination of the chuck, the photoresist must be kept off the back side of the wafer, and preferably also away from a thin bead around the periphery of the front side.
Conventionally, a very small amount of solvent liquid is dispensed at low pressure for a short time onto carefully determined areas of the wafer while the photoresist coating is being applied, so as to dissolve any photoresist in those areas. However, regulating the flow of this pressurized solvent introduces vapor bubbles into the liquid. A conventional needle valve, for example, regulates the flow very well, but it also generates a stream of tiny bubbles in the highly volatile solvent. These bubbles cause the dispensed solvent to splash into undesired wafer areas, producing pinholes in the photoresist, which ultimately result in defects in the integrated circuits on the wafer. Turning the flow on and off during the process also tends to introduce bubbles, and renders conventional bubble-extraction methods ineffective.
Liquids used for such solvents are highly volatile and flammable, thus compounding the problems in handling them.

Method and apparatus for removing vapor from a pressurized sprayed liquid in the manufacture of semiconductor integrated circuits


An engine that re-circulates its exhaust gas suffers decreased accuracy in estimating an EGR rate real-time especially while the operating state of the engine is in a transitional state, which often results in torque fluctuations and deteriorated exhaust gas.

1. Field of the Invention
The present invention relates to a control technology and method of an engine equipped with an exhaust gas recirculation (hereinafter referred to as "EGR") apparatus that returns part of exhaust gas to an intake pipe.
2. Description of the Related Art
In the field of diesel engines, and the like, there is known a technology for concurrently reducing soot and NOx by using the effect of reducing the combustion temperature based on the application of the exhaust gas recirculation (EGR), and the effect of uniformalizing a mixture gas by increasing an ignition delay period. The combustion temperature and the ignition delay period are influenced by the in-pipe inert gas (CO2) concentration and the in-cylinder oxygen concentration that change by the introduction of EGR. Therefore, in order to carry out the engine control, it is important to correctly detect/estimate not only the EGR rate but also the inert gas concentration or the in-cylinder oxygen concentration. For example, as an estimation method for estimating the in-cylinder oxygen concentration, JP-A-2003-27997 discloses a technology for determining the control amount relating to fuel injection. According to this technology, the control amount is determined by detecting the exhaust gas oxygen concentration by an O2 sensor, and then by estimating the in-cylinder oxygen concentration by use of the exhaust gas oxygen concentration.

Engine control system and control method thereof

Abstract: A sensor for directly detecting an EGR flow rate is disposed in an EGR path. An EGR rate and in-cylinder oxygen concentration are calculated from the output value of that sensor. In addition, when this EGR rate calculation method is used, the calculation is properly switched between a steady operation state characterized by a low load and small rotational fluctuations and a transitional operating state including the acceleration and deceleration. This makes it possible to correctly estimate the EGR rate and the in-cylinder oxygen concentration under a wide range of engine operation conditions, and thereby to avoid the fluctuation of torque and the deterioration of exhaust gas.


The present invention relates to a compound of the general formula (I) wherein Het is an optionally substituted 4-, 5-, 6- or 7-membered heterocyclic ring having at least one nitrogen atom; R1 is hydrogen, hydroxy, C1-C4 alkyl, C3-C4 cycloalkyl, C2-C4 alkenyl or C2-C4 alkynyl; R2 and R3 is each and independently selected from hydrogen, C1-C4 alkyl, C3-C4 cycloalkyl, C2-C4 alkenyl, C2-C4 alkynyl, C1-C4 alkoxy, halogen and cyano, provided that R2 and R3 may not both be hydrogen; R4 is C1-C4 alkyl, C3-C4 cycloalkyl, C2-C4 alkenyl or C2-C4 alkynyl; Ar is an optionally substituted aromatic ring system selected from pyridinyl; 1-naphthyl; 5,6,7,8-tetrahydro-1-naphthyl; quinolinyl; 2,3-dihydro-1,4-benzodioxinyl; 1,3-benzodioxolyl; 5,6,7,8-tetrahydroquinolinyl; 5,6,7,8-tetrahydroisoquinolinyl; 5,6,7,8-tetrahydroquinazolin-4-yl; 1-benzo[b]thiophen-7-yl; 1-benzo[b]thiophen-4-yl; 1-benzo[b]thiophen-3-yl; isoquinolinyl; quinazolinyl; and indan-4-yl; or Ar is substituted phenyl; or an enantiomer thereof or any salt thereof; to a pharmaceutical composition containing said compounds and to the use of said compounds in therapy. The present invention further relates to processes for the preparation of compounds of formula I and to new intermediates used in the preparation thereof.

The neurokinins, also known as the tachykinins, comprise a class of peptide neurotransmitters which are found in the peripheral and central nervous systems. The three principal tachykinins are Substance P (SP), Neurokinin A (NKA) and Neurokinin B (NKB). At least three receptor types are known for the three principal tachykinins. Based upon their relative selectivities favouring the agonists SP, NKA and NKB, the receptors are classified as neurokinin 1 (NK1), neurokinin 2 (NK2) and neurokinin 3 (NK3) receptors, respectively.
There is a need for an orally active and blood brain barrier crossing dual NK1/NK2 receptor antagonist for the treatment of e.g. respiratory, cardiovascular, neuro, pain, oncology, inflammatory and/or gastrointestinal disorders. In order to increase the therapeutic index of such therapy it is desirable to obtain such a compound possessing no or minimal toxicity as well as being selective to said NK receptors. Furthermore, it is considered necessary that said medicament has favourable pharmacokinetic and metabolic properties thus providing an improved therapeutic and safety profile such as lower liver enzyme inhibiting properties.
It is well known that severe problems such as toxicity may occur if plasma levels of one medication are altered by the co-administration of another drug. This phenomenon—which is named drug-drug interactions—could happen if there is a change in the metabolism of one drug caused by the co-administration of another substance possessing liver enzyme inhibiting properties. CYP (cytochrome P450) 3A4 is the most important enzyme in the human liver as a majority of oxidised drugs have been biotransformed by this enzyme. Accordingly, it is undesirable to employ a medication having a significant degree of such liver enzyme inhibiting properties. It has now been found that many NK receptor antagonists known in the art inhibit the CYP3A4 enzyme to a certain level and consequently there is a possible risk if high doses of those compounds are being used in therapy. Thus, there is a need for a novel dual NK1/NK2 receptor antagonist with improved pharmacokinetic properties. The present invention provides compounds with CYP3A4 enzyme inhibiting properties at a low level, as comparatively high IC50 values are obtained in a CYP3A4 inhibiting assay. Said method for determining CYP3A4 inhibition is described in Bapiro et al; Drug Metab. Dispos. 29, 30-35 (2001).
EP 0625509, EP 0630887, WO 95/05377, WO 95/12577, WO 95/15961, WO 96/24582, WO 00/02859, WO 00/20003, WO 00/20389, WO 00/25766, WO 00/34243, WO 02/51807 and WO 03/037889 disclose piperidinylbutylamide derivatives, which are tachykinin antagonists.
"4-Amino-2-(aryl)-butylbenzamides and Their Conformationally Constrained Analogues. Potent Antagonists of the Human Neurokinin-2 (NK2) Receptor", Roderick MacKenzie, A., et al, Bioorganic & Medicinal Chemistry Letters, In Press, available online 15 May 2003, discloses the compound N-[2-(3,4-dichlorophenyl)-4-(3-morpholin-4-ylazetidin-1-yl)butyl]-N-methylbenzamide which was found to possess functional NK2 receptor antagonistic properties.
WO 96/05193, WO 97/27185 and EP 0962457 disclose azetidinylalkyllactam derivatives with tachykinin antagonist activity.
EP 0790248 discloses azetidinylalkylazapiperidones and azetidinylalkyloxapiperidones, which are stated to be tachykinin antagonists.
WO 99/01451 and WO 97/25322 disclose azetidinylalkylpiperidine derivatives claimed to be tachykinin antagonists.
EP 0791592 discloses azetidinylalkylglutarimides with tachykinin antagonistic properties.
An object of the present invention was to provide novel tachykinin antagonists useful in therapy. A further object was to provide novel compounds having blood brain barrier penetrating properties, improved pharmacokinetic and metabolic properties and/or improved selectivity for the NK1/NK2 receptors.


A roofing tile including a tile body having an upper surface covered with a fire resistant coating including an acrylic or asphaltic adhesive, usually the former, containing a wire mesh and intermixed with fire retardant particles formed of a material or materials selected from the group consisting of sodium carbonate, ammonium phosphate and a light weight portland cement concrete, with decorative and heat resistant particles adhered to the upper surface of the adhesive, and desirably including a top layer of sodium silicate covering the last mentioned particles.

This invention relates to improved fire resistant tiles for use in roof structures or the like.
In recent years, building codes in many areas have become increasingly more rigid in requiring that roofs of homes or other buildings be resistant to fire. One way of satisfying this requirement is by treating wooden shingles or shakes with a chemical, such as sodium silicate, intended to be fire resistant. Another approach is to utilize shingles or tiles formed of a material other than wood and which is fire resistant. In some instances, tiles stamped from sheet metal are utilized. However, such metal tiles may tend to conduct heat too readily between the interior and exterior of a building and therefore adversely affect the insulation characteristics of the building.


Method for producing containers which are filled with a liquid filling substance from parisons made from a thermoplastic material, wherein the respective parison is conditioned thermally and is subsequently shaped into the container during a shaping and filling phase in a mold by way of the filling substance as pressure medium, wherein, during the shaping into the container, the parison is preferably guided at least temporarily through a stretching rod and is stretched in the axial direction.

The producing of containers by blow forming from parisons that are made from a thermoplastic material, for example from parisons made from PET (polyethylene terephthalate), with the parisons being fed to different processing stations within a blower machine (DE-OS 43 40 291), is known. A blower machine typically exhibits a heating device for the temperature-regulating or preheating (thermal conditioning) of the parisons and a blowing apparatus with at least one blowing station within whose region the previously thermally conditioned parison is expanded biaxially or multiaxially to form a container. The expansion takes place with the aid of a compressed gas (compressed air) as pressure medium which is introduced with a shaping pressure into the parison that is to be expanded. The process sequence with such an expansion of the parison is explained in DE-OS 43 40 291.
The fundamental configuration of a blowing station is described in DE-OS 42 12 583, while possibilities of thermally conditioning the parisons are explained in DE-OS 23 52 926.
According to a typical further-processing method the containers that are produced by blow forming are fed to a downstream filling apparatus where they are filled with the product or filling substance provided. A separate blower machine and a separate filling machine are therefore used. It is also known for the separate blower machine and the separate filling machine to be combined into a machine block, i.e. into a unitized blower/filler apparatus, with the blow forming and the filling still taking place at separate machine components and in chronological sequence.
There have also already been suggestions for producing containers, in particular including in the shape of bottles, from thermally conditioned or preheated parisons and simultaneously filling them with a liquid filling substance which is fed as a hydraulic pressure medium for expanding the parison and for shaping the container with a shaping and filling pressure such that the respective parison is shaped into the container simultaneously as the filling takes place. Such methods in which a simultaneous shaping and filling of the respective container take place can also be referred to as hydraulic shaping methods or hydraulic container shaping.
When the containers are shaped from the parisons by the filling substance itself, i.e. using the filling substance as hydraulic pressure medium, then the shaping and filling of the containers requires only one machine which however exhibits an increased complexity for this purpose. However initial experimental results with such devices indicate that the quality of the produced containers is still significantly below the quality of conventionally produced blow-formed containers. One reason for this is that a large number of process parameters which are available during the execution of normal blow forming are either absent during hydraulic container forming or could not yet be developed.
A further problem with hydraulic container forming is that contamination of the respective shaping and filling station or of the mold which forms this station and which is executed similar to a blow-mold of a blow-forming machine for producing containers from thermally conditioned parisons by blowing with a pressurized gas, must be avoided. Specifically when the filling substance is totally or partially carbonated there is a significant risk of contamination of the respective shaping and filling station due to losses of the filling substance, in particular when the pressure inside the container is lowered, i.e. during the pressure let-down of the container from the very high shaping and filling pressure to ambient pressure. Such losses of filling substance are due in particular to massive foaming during pressure let-down such that it has not so far been possible to use the simultaneous shaping and filling of containers using parisons and using the filling substance as pressure medium (hydraulic forming technology), in particular for carbonated products.

Method and device for producing containers which are filled with a liquid filling substance


Techniques are provided for improving cardiac output and also suppressing certain forms of apnea/hypopnea within a patient using an implantable medical device, such as a pacemaker or ICD. In one example, a selected pacing parameter—usually the pacing rate—is temporarily altered by an amount sufficient to elevate cardiac output, the elevation in cardiac output being eventually reduced by intrinsic compensatory mechanisms within the patient. The pacing parameter is then temporarily reset for a duration sufficient to allow the compensatory mechanisms to return toward a previous state so as to permit a subsequent alteration in the pacing parameter to again elevate cardiac output. The pacing parameter is repeatedly altered and reset so as to achieve an overall increase in cardiac output despite the intrinsic compensatory mechanisms. The increase in cardiac output is often sufficient to suppress certain forms of apnea/hypopnea, particularly apnea/hypopnea arising from Cheyne-Stokes Respiration (CSR).

Apnea and hypopnea are forms of disordered breathing characterized by periods of significantly reduced respiration. With hypopnea, respiration is reduced but still present. With apnea, however, respiration may cease completely for a minute or longer. Apnea most commonly occurs while the patient is asleep. Indeed, individual episodes of sleep apnea can occur hundreds of times during a single night. Accordingly, patients with apnea often experience excessive fatigue during the day. In addition, apnea can exacerbate various medical conditions, particularly congestive heart failure (CHF) wherein the patient suffers from poor cardiac function. Other medical conditions that can be adversely affected by apnea include: high blood pressure, risk for heart attack and stroke, memory problems, impotency and sexual dysfunction, migraine headaches, depression and anxiety, polycythemia (increase in the number of red blood cells), cor pulmonale (an alteration in the structure and function of the right ventricle caused by a primary disorder of the respiratory system), bradycardia (excessively slow heart rate), tachycardia (excessively fast heart rate), pulmonary hypertension hypoxemia (chronic daytime low blood oxygen) and hypercapnia (increased blood carbon dioxide (CO2)).
Apnea is often associated with Cheyne-Stokes Respiration (CSR), which is an abnormal respiratory pattern occurring in some patients with congestive heart failure (CHF). CSR is characterized by alternating periods of hypopnea and hyperpnea (i.e. fast, deep breathing.) Briefly, respiration is regulated by groups of nerve cells in the brain in response to changing blood chemistry levels, particularly blood CO2 levels. When blood CO2 levels exceed a certain threshold, the groups of nerve cells generate a burst of nerve signals for triggering inspiration. The inspiration nerve signals are relayed via phrenic nerves to the diaphragm and via other nerves to chest wall muscles, which collectively contract to expand the lungs. CSR arises partly due to a time lag between blood CO2 levels sensed by the respiratory control nerve centers of the brain and the blood CO2 levels. With CHF, poor cardiac function results in poor blood flow to the brain such that the nerve centers respond to blood CO2 levels that are no longer properly representative of the overall blood CO2 levels in the body. Hence, the nerve centers trigger an increase in the depth and frequency of breathing in an attempt to compensate for perceived high blood CO2 levels—although the blood CO2 levels will have already dropped. By the time the respiratory control nerve centers detect the drop in blood CO2 levels and act to slow respiration, the blood CO2 levels will have already increased. This cycle becomes increasingly unbalanced until respiration alternates between hypopnea and hyperpnea. The periods of hypopnea often become sufficiently severe that no breathing occurs between the periods of hyperpnea, i.e. periods of frank apnea occur between the periods of hyperpnea. The fluctuating blood chemistry levels caused by alternating between hyperpnea and apnea/hypopnea can significantly exacerbate CHF and other medical conditions. When CHF is still mild, CSR occurs, if at all, only while the patient is sleeping. When it becomes more severe, CSR can occur while the patient is awake. Accordingly, CSR is one mechanism by which apnea can occur while a patient is awake. Hence, apnea is not limited to occurring only while a patient is asleep.
Herein, apnea arising due to CSR is referred to as "CSR-induced apnea." Hypopnea arising due to CSR is referred to as "CSR-induced hypopnea." For the sake of generality, the term "CSR-induced apnea/hypopnea" is used to encompass both conditions. Note that, in some of the medical literature, apnea arising due to CSR is referred to as "CSA-CSR," where CSA stands for "Central Sleep Apnea." However, as noted, CSR-induced apnea can potentially occur while the patient is awake and hence is not necessarily a form of sleep apnea. Moreover, the term CSA also refers to a fundamentally different type of apnea, which is believed to be the result of a neurological condition within the central nervous system. With CSA, phrenic nerve signals are simply not generated during extended periods of time while the patient is asleep or are of insufficient magnitude to trigger sufficient muscle contraction to achieve inhalation. This occurs regardless of the blood CO2 levels and hence appears to be unrelated to CSR-induced forms of apnea. Herein, to avoid any possible confusion, the term CSR-CSA is not used. As noted, the term CSR-induced apnea/hypopnea is instead used. Note, also, that there is yet another form of apnea—obstructive sleep apnea (OSA). OSA arises when the respiration airway is temporarily blocked. OSA is not particularly relevant to the techniques described herein but is mentioned for the sake of completeness.
In view of the adverse consequences of CSR-induced apnea/hypopnea, it is highly desirable to provide techniques for suppressing episodes of the condition. A variety of techniques have been proposed, particularly for use by pacemakers, ICDs or other implantable medical devices. With many such techniques, the implantable device first detects an episode of apnea/hypopnea and then delivers therapy to alleviate the episode. Examples of techniques for detecting episodes of apnea/hypopnea are discussed in U.S. patent application Ser. No. 10/883,857, of Koh et al., entitled "System and Method for Real-Time Apnea/Hypopnea Detection Using an Implantable Medical System," filed Jun. 30, 2004. See, also, U.S. patent application Ser. No. 10/795,009, of Koh, entitled "System and Method for Distinguishing among Obstructive Sleep Apnea, Central Sleep Apnea and Normal Sleep Using an Implantable Medical System," filed Mar. 4, 2004 and U.S. patent application Ser. No. 10/844,023, of Koh, filed May 11, 2004, entitled "System and Method for Providing Demand-Based Cheyne-Stokes Respiration Therapy Using an Implantable Medical Device". Once an episode of apnea/hypopnea is detected, therapy is delivered to terminate the episode and restore respiration that is more normal. Exemplary techniques include applying electrical stimulation directly to the phrenic nerves via implantable nerve stimulators so as to cause the diaphragm to contract. These and other therapeutic techniques are discussed in the aforementioned patent applications as well. If therapy is ineffective, warning signals can be generated to awaken the patient, which is typically sufficient to restore normal respiration.
It would be preferable, however, to provide techniques for suppressing CSR-induced apnea/hypopnea that do not require detecting individual episodes of apnea or hypopnea, as reliable detection can sometimes be problematic. It would also be desirable to provide suppression therapies that do not require phrenic nerve stimulators or other potentially elaborate or intrusive therapeutic measures. One technique that has been proposed is simply to increase a cardiac pacing rate so as to increase cardiac output. So long as the stroke volume of the heart remains constant, an increase in the cardiac pacing rate produces an increase in cardiac output. The increase in cardiac output tends to suppress CSR by increasing the blood flow to the brain thus avoiding the blood CO2 imbalances described above. Suppression of CSR, in turn, suppresses CSR-induced episodes of apnea/hypopnea. However, it is believed by the inventor of the present invention that the increase in cardiac output resulting from an increase in pacing rate is only temporary. This is illustrated in FIG. 1. Briefly, a cardiac pacing rate 2 is increased at time 3 resulting in an increase in cardiac output 4. The increase in cardiac output suppresses CSR, which is illustrated in the figure by way of a respiratory pattern 5 having crescendo/decrescendo patterns alternating between hyperpnea 5 and apnea 6. The increase in cardiac output is sufficient to temporarily diminish the severity of CSR thus permitting more normal respiration 7 to resume. However, as illustrated in the figure, the increase in cardiac output does not last, even though the higher pacing rate is sustained for an extended period of time. The cardiac output soon begins to drop, apparently due to intrinsic hemodynamic compensatory mechanisms within the patient. In this regard, hemodynamic systems of the patient appear to operate to reduce the stroke volume to compensate for the artificially increased heart rate. As a result, CSR eventually resumes (or it again becomes more severe), thus triggering further episodes of apnea. Note that the graphs of FIG. 1 should not be construed as depicting actual clinically-obtained data. The graphs set forth hypothetical data provided to clearly illustrate the affect of the intrinsic compensatory mechanisms on cardiac output and respiration. Actual variations in cardiac output and respiration may differ. Also note that, during actual CSR, the intervals of apnea/hypopnea are often longer in duration than the intervening intervals of hyperpnea. FIG. 1 illustrates relatively short intervals of apnea/hypopnea so as to permit many complete cycles of CSR to be illustrated within the timeline of the figure. The vertical scales of the graphs are in arbitrary units and the features illustrated therein are not necessarily to scale.
It would be highly desirable to provide techniques for use by an implantable medical device for achieving a sustained increase in cardiac output sufficient to suppress apnea/hypopnea, particularly CSR-induced apnea/hypopnea. It is to this end that certain aspects of the invention are directed. Although a sustained increase in cardiac output is helpful in suppressing apnea/hypopnea, the increase is beneficial in and of itself, since increased cardiac output tends to mitigate CHF, pulmonary edema, and other conditions. Accordingly, other aspects of the invention are directed to the more general goal of improving cardiac output within a patient using an implantable medical device.

Implantable medical device with cardiac output- based apnea suppression


A method for adhesively assembling plastic products which comprises placing adjacent to the surfaces of the pieces to be joined a continuous, non-perforated metallic foil strip, which absorbs electromagnetic waves. The strip, which is contiguous with a heat activatable adhesive material, is positioned between the pieces and exposed to electromagnetic waves while the pieces are held together to form a bonded relationship between the plastic pieces.

This disclosure relates to the assembly of a variety of plastic products using an adhesive device that is activated by electromagnetic waves. The plastic products include assemblies from such materials such as synthetic fabrics such as vinyls, rayon, nylon; and items such as plastic toys, picture frames, plastic auto and RV parts and components and the like.
The assembly of plastics is typically effected by adhesives, stitching, stapling; and, in the case of plastic fabrics, heat staking and ultrasonic welding. Each method has its strengths and weaknesses depending on the application.
However, these traditional adhesives are rapidly falling into disfavor as it becomes desirable and even necessary to work quicker and neater and to develop even more secure adhesive bonds. It is apparent, then, that inventions are waiting to be made which address the placement of adhesive material in a neat, clean, safe and effective manner especially with regard to the assembly of plastic products.
Not surprisingly then, others have experimented with alternatives to traditional fastening devices for adhesively joining the components of an assembled product.

Method of adhesively assembling plastic products


A device for linear transmitter (150) with improved loop gain stabilization. The linear transmitter (150) includes a main amplifier loop (178) and an auxiliary loop (176). The device includes an adjustable device (159), connected within the main amplifier feedback loop (178) and further connected to the auxiliary loop (176). The adjustable driver (159) receives an input signal from main amplifier feedback loop (178), amplifies it in accordance with a gain control signal received from the auxiliary loop (176) and provides the amplified signal to main amplifier feedback loop (178).

Radio communication devices use antennas to provide efficient transmission of radio frequency communication signals. The transmission portion of a communication device includes a power amplifier for amplifying the radio frequency signals before they are coupled to the antenna for transmission. The power amplifier design often relies on constant load impedance which is directed at maximizing gain, efficiency, power output level, and the like. The behavior of a transmitter may be affected by its operating environment. For example, a transmitter operating near an electromagnetically reflective structure may be susceptible to energy reflected back through the antenna into the transmitter. Reflective energy may be detrimental to transmitter performance, particularly to the performance of the power amplifier. An isolator or circulator is often inserted between the antenna and the power amplifier to protect against changes in load impedance as a result of reflected energy.
The isolator protects the power amplifier by absorbing the reflected energy and preventing it from reaching the amplifier. The isolator directs the reflected energy to an absorptive load termination. An isolator typically adds significant cost, size and weight to the design of a radio communication device.
U.S. Pat. No. 5,564,087 to Cygan et al., entitled "Method and apparatus for a linear transmitter" discloses another solution to the problem of reflected energy. The solution incorporates a directional coupler to detect the reflected energy and provides a means of adjusting the gain of the power amplifier accordingly. Generally, the gain to the power amplifier is reduced when high levels of reflected energy are present. In this approach, the circuitry for detection of the reflected energy must operate at the transmission frequency. This adds significant cost and complexity to the radio design.
U.S. Pat. No. 5,675,286 to Baker et al., entitled "Method and apparatus for an improved linear transmission", is directed to a method and apparatus for isolator elimination, operative at the baseband frequencies, which is described in detail, herein below. Reference is made to FIG. 1, which is a schematic illustration of a linear transmitter block of a radio communication device, generally referenced 50, which is known in the art. Transmitter block 50 includes an attenuator 52, three summators 54, 66 and 70, a baseband loop filter unit 56, an up-mixer and radio-frequency (RF) filter unit 58, an RF power amplifier 60, a down-mixer and baseband filter unit 64, a phase shifter unit 62, an AGC 68, an adaptor unit 72 and an antenna 74. Summator 54 is connected to attenuator 52, to baseband filter unit 56 and to phase shifter unit 62. Summator 66 is connected to baseband filter unit 56, to up-mixer and RF filter unit 58, to adaptor unit 72 and to AGC 68. Summator 70 is connected to AGC 68, to attenuator 52 and to adaptor unit 72. Adaptor 72 is further connected to AGC 68, to attenuator 52 and to phase shifter unit 62. Up-mixer and RF filter unit 58 is connected to baseband loop filter unit 56 and to power amplifier 60. Down-mixer and baseband filter unit 64 is further connected to power amplifier 60 and to phase shifter unit 62. Antenna 74 is connected to power amplifier 60 and to down-mixer and baseband filter unit 64.
A signal 80 is provided as input to amplifier feedback loop 78 and to isolator elimination circuit 76. Amplifier feedback loop 78 and isolator elimination circuit 76 represent the main amplification lop and the auxiliary loop, respectively. Amplifier feedback loop 78 is a closed loop amplifier structure. Typically, this structure can be considered a Cartesian feedback loop amplifier. The input signal 80 is generally a complex digital baseband signal, having quadrature components, i.e. in-phase (I) component and quadrature (Q) component. Signal 80 is provided to attenuator 52. Attenuator 52 provides an attenuated signal to summator 54. Summator 54 combines this signal with a feedback loop output signal 82 and provides a resulting error signal to baseband filter unit 56. Baseband filter unit 56 provides the filtered error signal to up-mixer and RF filter unit 58. Up-mixer and RF filter unit 58 up-converts the signal to RF and provides it to power amplifier 60. Power amplifier 60 amplifies the signal and provides the amplified signal to antenna 74 for transmission. Antenna 74 forms a load for power amplifier 60. It is noted that this load is susceptible to impedance variations due to its operating environment. Power amplifier 60 provides a portion of the output signal to summator 54, via down-mixer and baseband circuit 64 and phase shifter unit 62, thereby generating feedback loop output signal 82. Feedback loop output signal 82 constitutes a feedback signal for controlling the gain of power amplifier 60 and maintaining transmitter block 50 in the linear mode of operation.
Baseband filter unit 56 also provides a filtered error signal 90 to summator 66. Summator 66 combines error signal 90 with signal 84 from adaptor 72 and provides the result to AGC 68. AGC 68 constitutes a linear gain control circuit of isolator elimination circuit 76. Adaptor 72 controls the gain of AGC 68 by altering the output signal of AGC 68. AGC 68 provides the output signal to summator 70, where it is combined with input signal 80. Summator 70 provides the resulting error signal 94 to adaptor 72, which produces two output control signals 86 and 88. Control signal 86 adjusts the gain of attenuator 52, and control signal 88 adjusts phase shifter 62. Adaptor 72 produces control signals 84, 86 and 88 based on input signal 80 and error signal 94.
To avoid the undesirable effects, it is necessary to design a transmitter with sufficient gain and/or power reserve. Such a design will result in increased complexity, cost, power consumption and more. There is a trade-off between the desired design simplicity and cost-effectiveness on one hand, and necessary dynamic range of system gain and/or output power, on the other hand. It is desirable to provide a linear transmitter, which is cost-effective and simple, yet assuring linear performance and high signal-to-noise ratio.

Apparatus for linear transmitter with improved loop gain stabilization


A system includes a sampling circuit, an analog filter, an analog input, a modulator, and a modulating source. The modulating source includes an output that generates an impulse response encoded as a sigma-delta sequence. The output of the modulating source is coupled to the modulator, which is coupled to the analog input. The analog filter is coupled to the modulator and the sampling circuit. When the system is configured as a filter, an analog signal is modulated with the sigma-delta sequence. The modulated signal is filtered and then sampled.

Many applications require the filtering and subsequent sampling of analog signals. For example, in communications receivers, an analog signal comprising a desired signal, interfering signals, and noise may first be filtered to select the desired signal from among the other undesired signals so that subsequent signal processing may be performed without distortion. Examples of some conventional filters include L-C filters, surface acoustic wave filters, and ceramic filters. Many of these filters are costly, bulky, or difficult to miniaturize. The filters can further lack programmability, which is often desirable. There are other examples of conventional filters that lend themselves to miniaturization in integrated circuits. These include Gm-C filters, MOSFET-C filters, Sallen & Key filters, and others. Many of these consume an unacceptable amount of power or lack the performance required for many applications. Therefore, it would be useful to identify a filter technique that overcomes these conventional limitations.
One technique for filtering and sampling an analog signal is based on sampled correlators as found, for example, in "Principles of Communications Engineering," by Wozencraft and Jacobs. The system, illustrated in FIG. 1, accepts a received signal "x(t)" 102 that is combined with a correlating signal, "h(t−kT)" 108 using a bilinear analog multiplier 104, where "t" is an increment in time, "k" is an integer, and "T" is a desired sampling interval. The output of the multiplier 104 is filtered and sampled at a periodic interval "kT," where "k" is an integer and "T" is the desired sampling interval The resulting output signal, "y[k]," 106 is a discrete-time representation of the filtered analog signal.
An advantage of the system illustrated in FIG. 1 is that filtering and sampling is performed computationally with components that are readily miniaturized. However, the system is not without disadvantages. For example, it is sometimes difficult to build an accurate analog multiplier 104. Without an accurate analog multiplier 104, the performance of this system is often limited. One technique for building analog multipliers is taught by Gilbert in "A Precise Four-Quadrant Multiplier with Subnanosecond Response." However, Gilbert's technique lacks the necessary accuracy for many applications.
The difficulty of building analog multipliers is also recognized in other publications and other United States and European Patents. U.S. Pat. No. 3,953,795 to Brunner, for example, discloses an electric power meter implementing an analog multiplier using a commutating mixer driven by a two-level pulse width modulator. Eynde, in "A Power Metering ASIC with a Sigma-Delta-Based Multiplying ADC," discloses an electric power meter with a commutating mixer which uses a special type of pulse width modulator. Eynde's structure is substantially disclosed in European Patent Off. No. 589,090. U.S. Pat. No. 5,862,069 to Nestler, eliminates analog multiplication problems by performing multiplication in the digital domain after modulating both input signals. Accordingly, there is a need for an apparatus and method that can produce accurate correlation and, hence, filtering of an analog signal in a manner that lends itself to miniaturization within integrated circuits.

Method and apparatus for a programmable filter


A method for channel response estimation for an OFDM communication system based on a certain normal training symbols plus one virtual training symbol begins with receiving at least one preamble symbol of a transmitted data frame. A first channel state information is estimated from the at least one preamble symbol. At least one signalling symbol is then received and equalized and a rate factor and a length factor are extracted from the at least one signalling symbol. A virtual training symbol is formed from the rate factor and the length factor. A second channel state information is estimated from the at least one preamble symbol and the virtual training symbol. The data symbols are equalized by performing channel correction on data symbols with the second channel estimation and pilot sub-carriers in each of the plurality of data symbols.

1. Field of the Invention
This invention relates generally to communication systems. More particularly, this invention relates to estimation of channel responses as involved in Orthogonal Frequency Division Multiplexing (OFDM) communication systems.
2. Description of Related Art
Orthogonal Frequency Division Multiplexing (OFDM) is a multi-carrier modulation scheme resistant to multipath interference and frequency selective fading in communication systems such as wireless local area network (WLAN) and digital audio/video broadcasting. The technique divides the channel bandwidth into multiple narrow band sub-channels or sub-carriers, which are used for transmitting data in parallel with high efficiency spectrum usage, as described in "OFDM for Wireless Multimedia Communications", Van Nee and Prasad, Artech House Publishers, 2000.
In 1999, OFDM was selected by the IEEE 802.11 standardization committee as the technology for a 5 GHz WLAN standard—WLAN Medium Access Control (MAC) and Physical Layer (PHY) Specification: High-speed Physical Layer in the 5 GHz Band, as described in the "Information Technology—Telecommunications and Information Exchange Between Systems—Local and metropolitan area networks—specific requirements—Part 11: "Wireless LAN Medium Access Control (MAC) and Physical Layer (PHY) Specifications", sponsored by the LAN MAN Standards Committee of the IEEE Computer Society, ANSI/IEEE Std 802.11(a), September 1999. The IEEE 802.11(a) standard divides the 5150 MHz to 5350 MHz frequency band and the 5725 MHz to 5825 MHz frequency band into 12 20-MHz communication channels. Each of these 20-MHz channels is composed of 52 narrow band sub-carriers, which are modulated using binary phase shift keying (BPSK), quadrature phase shift keying (QPSK), 16-quadrature amplitude modulation (16-QAM), or 64-QAM to support data rates up to 54 Mbps in 16.6 MHz occupied bandwidth on 20 MHz channelization. A typical block diagram of the baseband processing of an OFDM transceiver in the IEEE 802.11(a) WLAN is shown in FIG. 1. In the transmitter path 5, binary input data 10 are encoded by an industry standard ½ rate convolutional encoder 15. The coding rate can be increased to ⅔ or ¾ by puncturing the coded output bits to accommodate the desired data rate which is one of 6 Mbps, 9 Mbps, 12 Mbps, 18 Mbps, 24 Mbps, 36 Mbps, 48 Mbps or 54 Mbps. After interleaving with an interleaver circuit 20, the binary values are mapped with a mapping circuit 25 onto BPSK, QPSK, 16-QAM or 64-QAM constellations according to the chosen data rate. A time-domain OFDM symbol is thus obtained from a Fast Fourier Transform circuit 35 by applying a length N(N=64) inverse fast Fourier transform (IFFT) to those modulation values 30, YT(0), YT(1), . . . , YT(N−1). The N complex values output from the IFFT 35 are then applied to the sub-carrier multiplexer 40 to form the baseband discrete time samples. The resulting digital samples in the time domain are then converted by the Digital-to-Analog (D/A) converter 45 to analog signals, which are further up-converted to the 5 GHz band, amplified and transmitted to the over-the-air channel 50 through an antenna.
The transmitted signal suffers from some distortions in the channel 50, which usually can be represented by a channel impulse response, h(t) 55, and an additive noise 60, ν(t) 65, in the time-domain. The resulting corrupted signal from the channel 50 is the input to the OFDM receiver 70.
The OFDM receiver 70 basically performs the reverse operations of the transmitter 5. The received RF signal is down-converted to a baseband signal and sent to an Analog-to-Digital (A/D) converter 75. The digital baseband samples are then demultiplexed by the demultiplexing circuit 80 to reconstruct the transmitted time-domain OFDM symbol structure. The digital samples of the demultiplexed signal are applied to a Fast Fourier Transform circuit 85, which creates a frequency-domain OFDM symbol with N complex values, YR(0), YR(1), . . . , YR(N−1) 90. The N complex values, YR(0), YR(1), . . . , YR(N−1) 90 are applied to a the Modulation De-Mapping circuit 95, which converts the complex values to a binary sequence. The binary sequence is the input to the De-Interleaver circuit 100, which correctly orders the sequence to reflect the original structure of the encoded transmitted data. The de-interleaved data sequence is applied to the Viterbi Decoder 105 for recovery of the transmitted data to form the receiver's output data 110.
To properly receive a data frame in burst-mode transmission, the receiver 70 first has to detect the arrival of a frame, find the unknown beginning sample instant of the first and subsequent OFDM data symbols, and, estimate/correct for any carrier frequency offset imparted to the sub-carriers due to variation in the nominal values of the oscillator frequencies in both the transmitter 5 (remote device) and receiver 70 (local device), using structured training symbols 117, 118, and 119 contained in a frame preamble 115 as shown in FIG. 2. After that, the receiver 70 can perform an FFT to convert a time-domain OFDM data symbol into N frequency-domain complex values, YR(0), YR(1), . . . , YR(N−1) 90. Ideally, these values are expected to be same as YT(0), YT(1), . . . , YT(N−1) 30, but they are usually distorted by the channel. Thus, before they are used as the input to the de-mapping 95, the values, YR(0), YR(1), . . . , YR(N−1) 90, should be compensated first so that the equalized values, ŶR(0), ŶR(1), . . . , ŶR(N−1), are the good estimations of YT(0), YT(1), . . . , YT(N−1) 30.
When a wireless channel is typically modelled in the time domain as the joint effort of multipath effects with a channel impulse response, h(t) 55 and an additive noise, ν(t) 65, the frequency-domain relation between the YT(k) and YR(k) can be expressed asYR(k)=H(k)·YT(k)+V(k), k=0 to N−1  (1)                where:                    V(k) is the frequency-domain additive noise.            H(k) is the channel transfer function which is usually called as channel state information.The channel estimation thus is simply to find the estimation of channel state information, Ĥ(k), so that ŶR(k)=YR(k)/Ĥ(k) approximates YT(k) as closely as possible.
Different algorithms have been proposed for channel estimations. The Minimum Mean Square Error (MMSE) based algorithms give low mean square estimation errors, but may be either impractical or not robust because they are of high computational complexity and usually require an assumption of the channel statistics. "OFDM channel estimation by singular value decomposition", Edfors et al., IEEE Trans. On Communications, vol. 46, no. 7, pp.931-939, July 1998 describes such a technique for channel estimation. On the other hand, the Least Square (LS) estimation, which is given byĤ(k)=YR(k)/YT(k), k=0 to N−1  (2)is much simpler and easier to be implemented. However, when compared with the MMSE estimation, the LS estimation yields a higher mean square estimation error, which will translate to degradation in system performance in terms of higher bit error rate (BER) and packet error rate (PER) and should be further reduced to an acceptable level in practice.
U.S. Patent Application Publication 2001/0036235 A1 (Kadous), teaches a method and apparatus for improving LS channel estimate in OFDM communication systems. The method and apparatus allows a channel estimate to be determined independent of having knowledge on channel statistics. Channel estimation is performed by determining and then utilizing an LS estimate and an interpolation coefficient for each transmitting antenna. The interpolation coefficient is determined independently from the statistics of the channel, i.e., without needing the channel multipath power profile (CMPP). The interpolator coefficient is multiplied by an LS estimate for each transmitting antenna to determine the channel estimate for each channel.
U.S. Pat. No. 6,487,253 (Jones, IV, et al.) illustrates another method for achieving improved channel response estimation in an OFDM system in the presence of interference. The interference and/or noise present on the received training symbols are estimated first. Based on the measured noise and/or interference, a weighting among training symbols is developed. Channel response is then estimated based on a weighted least square procedure.
When these two techniques can be employed to enhance the LS channel estimation in an OFDM system, the derivation of robust interpolation coefficients or weighting factors may require multiple transmitter or receiver antennas.
In a wireless LAN system, which performs burst-mode transmission of data frames with limited length, the channel state information can be assumed to be constant within the transmission duration of a frame, and, therefore, the channel estimation is commonly performed by the use of one or more known OFDM symbols, which are usually called training symbols. In the IEEE 802.11a standard, the two long training symbols 118 and 119 contained in the preamble 115 of each data frame can be used for estimating the channel state information. It is easy to see that, by averaging the LS estimations of channel state information obtained from the prescribed training symbols, the effects of the channel noise can be reduced to some extent. By averaging the two identical long training symbols in the IEEE 802.11a WLAN, for example, the reference amplitudes and phases for doing coherent demodulation can be obtained with a noise level that is 3 dB lower than the noise level of data symbols. "An integrated 802.11(a) baseband and MAC processor", Thomson et al., Proc. IEEE Int. Solid State Circuits Conf. (ISSCC), February 2002, pp.126-127 describes an 0.25 μm CMOS baseband and MAC processor for the IEEE 802.11(a) WLAN standard that averages the two long training symbols 118 and 119 of FIG. 2 for channel estimation/correction. U.S. Pat. No. 5,903,610 (Skold, et al.) illustrates a receiver of a digital radio communication system including a combined channel estimate that is formed by averaging a long channel estimate with a short channel estimate. Also, "Effect of channel estimation error in OFDM-based WLAN", Cheon et al., IEEE Communication Letters, vol.6, pp.190-192, May 2002 examines the performance degradation due to the channel estimation error in an OFDM-based WLAN. The average effective Signal to Noise ratio and average bit error probabilities (BEPs) are derived in a Rayleigh fading channel. Cheon et al. have demonstrated the advantages of averaging the two long training symbols 118 and 119 of FIG. 2 for channel estimation over using a single training symbol.
The choice of the number of training symbols that can be used for channel estimation is a trade-off between a good channel estimation performance and a short training time, because the training symbols contain no actual data information. In the training structure (preamble) 115 of FIG. 2, the total training length is 16 μs. The periods t1 to t10 contain short training symbols 117 and T1 and T2 denote long training symbols 118 and 119. The dashed boundaries in the figure denote repetitions due to the periodicity of the inverse Fourier transform. A short OFDM training symbol 117 consists of 12 sub-carriers, which are modulated by the elements of a fixed spectral sequence as defined in paragraph 17.3.3 of IEEE 802.11(a) standard. A long OFDM training symbol 118 and 119 consists of 53 sub-carriers (including a zero value at dc), which are modulated by the elements of the spectral sequence as defined in paragraph 17.3.3 of IEEE 802.11(a) standard.
The long OFDM training symbols are followed by the OFDM SIGNAL symbol 120, which contains the RATE 121 and the LENGTH 123 fields of the TXVECTOR as defined in paragraph 17.2.2 of IEEE 802.11(a) standard. The encoding of the SIGNAL field 126 into an OFDM symbol 120 proceeds with convolutional encoding (R=½), interleaving, less noise sensitive BPSK mapping, pilot insertion, and OFDM modulation. The contents of the SIGNAL field 126 are not scrambled. The SIGNAL field 126 is composed of 24 bits, as illustrated in FIG. 2. The four bits 0 to 3 121 shall encode the RATE. Bit 4 122 is reserved for future use. Bits 5-16 shall encode the LENGTH field 123 of the TXVECTOR, with the least significant bit (LSB) being transmitted first. Bit 17 124 is a positive parity (even parity) bit for bits 0-16. The bits 18-23 constitute the SIGNAL TAIL field 125, and all 6 bits are set to zero. The RATE field 121 conveys information about the type of modulation and the coding rate as used for the rest of the frame. The 4 bits of RATE field 121 are set according to the values in Table 1. The LENGTH field 123 is an unsigned 12-bit integer that indicates the number of octets in the protocol service data unit that the MAC is currently requesting the PHY to transmit. This value is used by the PHY to determine the number of octet transfers that will occur between the MAC and the PHY after receiving a request to start transmission. This value is also used by the receiver 70 to determine the number of data octets contained in a received data frame.
TABLE 1RATEData Rate(4 bits)(Mbps)Modulation TypeCoding Rate11016BPSK½11119BPSK¾010112QPSK½011118QPSK¾10012416-QAM½10113616-QAM¾00014864-QAM⅔00115464-QAM¾
The OFDM SIGNAL symbol 120 is followed by variable number of OFDM DATA symbols 127 which contain actual data information. It can be seen that the long training time will reduce the efficiency of spectrum usage. It is thus useful to have a technique for achieving enhanced training capability with no extra training time required.

Method for reducing channel estimation error in an OFDM system

Agency for Science, Technology, and Research


A method including the steps of: modifying at least one part of a sapphire substrate by dry etching to thereby form any one of a dot shape, a stripe shape, a lattice shape, etc. as an island shape on the sapphire substrate; forming an AlN buffer layer on the sapphire substrate; and epitaxially growing a desired Group III nitride compound semiconductor vertically and laterally so that the AlN layer formed on a modified portion of the surface of the sapphire substrate is covered with the desirably Group III nitride compound semiconductor without any gap while the AlN layer formed on a non-modified portion of the surface of the sapphire substrate is used as a seed, wherein the AlN buffer layer is formed by means of reactive sputtering with Al as a target in an nitrogen atmosphere.

For example, a Group III nitride compound semiconductor used in a light-emitting device is a direct transition type semiconductor exhibiting an emission spectrum in a wide range of from ultraviolet to red. The Group III nitride compound semiconductor is applied to a light-emitting device such as a light-emitting diode (LED), a laser diode (LD), or the like. Application of the Group III nitride compound semiconductor to a transistor such as an FET or the like has been developed actively because the band gap of the Group III nitride compound semiconductor is so wide that the device using the Group III nitride compound semiconductor can be expected to operate more stably at a high temperature than a device using another semiconductor. Development of the Group III nitride compound semiconductor into various general semiconductor devices has been expected from the environmental viewpoint because arsenic is not used as the main component. Generally, the Group III nitride compound semiconductor is formed on sapphire used as a substrate.
If the Group III nitride compound semiconductor is formed on such a sapphire substrate, dislocation is caused by misfit in lattice constant between sapphire and the Group III nitride compound semiconductor. For this reason, there is a problem that device characteristic becomes poor. The dislocation caused by the misfit is threading dislocation that passes through a semiconductor layer vertically (in a direction perpendicular to a substrate surface). There is a problem that dislocation of about 109 cm−2 propagates in the Group III nitride compound semiconductor. When Group III nitride compound semiconductor layers different in composition are laminated, it propagates up to the uppermost layer. For example, in the case of a light-emitting device, there is therefore a problem that device characteristic such as threshold current in LD, device life in LD and LED, etc. cannot be improved. Also in the case of another semiconductor device, no device but a semiconductor device low in mobility can be formed because electrons are scattered by defects. These problems occur also in the case where another substrate is used.
For example, there has been accordingly proposed a method in which: a Group III nitride compound semiconductor layer is once formed on a substrate with or without interposition of a buffer layer; a mask is formed on part of an upper surface of the Group III nitride compound semiconductor layer so that the Group III nitride compound semiconductor cannot be epitaxially grown vertically from the masked portion; and the Group III nitride compound semiconductor is epitaxially grown vertically and laterally from the non-masked portion to thereby form a Group III nitride compound semiconductor layer low in threading dislocation above the masked portion.
In the technique described in the official gazette, it is however necessary to perform Group III nitride compound semiconductor growth twice before and after masking because different apparatuses are used for Group III nitride compound semiconductor growth and masking respectively. Moreover, in the technique described in the official gazette, a space is apt to be formed between the Group III nitride compound semiconductor layer formed above the masked portion and the mask because the Group III nitride compound semiconductor cannot be formed on them asked portion with good adhesion. As a result, when separation into devices is performed in an after-process, peeling, cracking or chipping occurs in the portion where the space is formed.
The invention has been made to solve the aforementioned problems and an object of the invention is to provide a semiconductor device in which a Group III nitride compound semiconductor layer suppressed in terms of threading dislocation is formed by a smaller number of steps as a whole and in which peeling, cracking or chipping hardly occurs.

Process for producing group III nitride compound semiconductor


A nonvolatile semiconductor memory device according to an embodiment comprises a control unit, in a data write, determining at least one of: a first requirement that the number of times of a data read on first through n1-th pages (where n1 is an integer of 1 to N−1) of a target block executed after the most recent data erase on the target block, is less than a reference number of times; and a second requirement that the number of memory cells whose threshold voltage is higher than a reference voltage, of a plurality of memory cells of a reference page of n1+1-th through N-th pages of the target block, is less than a reference number, and when the determined requirement is satisfied, writing additional data to the n1+1-th through N-th pages of the target block.

1. Field
Embodiments relate to a nonvolatile semiconductor memory device.
2. Description of the Related Art
In the case of a NAND type nonvolatile semiconductor memory device, sometimes, when performing data read on a selected page, a threshold voltage of a memory cell of another page of a block to which the selected page belongs ends up varying due to the influence of a read voltage applied to a word line at this time.


An energy storing exterior cover for a user electronic device including an integrated energy store and an energy transfer interface for transferring energy electrically to a user electronic device when covered by the cover.

There is a trend for electronic devices to increase the number or type of functions they perform and therefore increase their energy consumption. There is also a trend to decrease the size of electronic devices.
As the amount of energy stored at an electronic device is limited by its size it can be difficult to balance the need for energy consumption with the need for a small compact device.
One approach to this problem has been to make electronic devices more energy efficient.
Another approach to this problem has been to develop more efficient energy stores. This has involved the development of batteries, such as Li-ion batteries, that store greater energy per unit volume and then the replacement of Li-ion batteries by fuel cells.
It would be desirable to increase the energy available for use at an electronic device.


Determining voltage level validity for a power-on reset condition is described. A supply voltage is applied to an integrated circuit. An oscillating signal is generated responsive to the supply voltage applied. A counting occurs responsive to oscillations of the oscillating signal. A triggering occurs responsive to reaching a first voltage level of the supply voltage for the power-on reset condition. A first count of the counting occurs responsive to the triggering. A second count is selected responsive to the first count. A second level is accepted as having at least met a threshold for the supply voltage responsive to the counting reaching the second count for the power-on reset condition.

Programmable logic devices ("PLDs") are a well-known type of integrated circuit that can be programmed to perform specified logic functions. One type of PLD, the field programmable gate array ("FPGA"), typically includes an array of programmable tiles. These programmable tiles can include, for example, input/output blocks ("IOBs"), configurable logic blocks ("CLBs"), dedicated random access memory blocks ("BRAMs"), multipliers, digital signal processing blocks ("DSPs"), processors, clock managers, delay lock loops ("DLLs"), and so forth. Notably, as used herein, "include" and "including" mean including without limitation.
One such FPGA is the Xilinx Virtex® FPGA available from Xilinx, Inc., 2100 Logic Drive, San Jose, Calif. 95124. Another type of PLD is the Complex Programmable Logic Device ("CPLD"). A CPLD includes two or more "function blocks" connected together and to input/output ("I/O") resources by an interconnect switch matrix. Each function block of the CPLD includes a two-level AND/OR structure similar to those used in Programmable Logic Arrays ("PLAs") and Programmable Array Logic ("PAL") devices. Other PLDs are programmed by applying a processing layer, such as a metal layer, that programmably interconnects the various elements on the device. These PLDs are known as mask programmable devices. PLDs can also be implemented in other ways, for example, using fuse or antifuse technology. The terms "PLD" and "programmable logic device" include but are not limited to these exemplary devices, as well as encompassing devices that are only partially programmable.
For purposes of clarity, FPGAs are described below though other types of PLDs may be used. FPGAs may include one or more embedded microprocessors. For example, a microprocessor may be located in an area reserved for it, generally referred to as a "processor block."
A certain amount of supply current during a power-up of an integrated circuit is used to ensure proper initialization. The actual current consumed depends on the power-up ramp rate of the power supply used. Additionally, one or more supply voltages may first have to pass through their power-on reset threshold voltage level or levels for an integrated circuit to properly power up. An example of such a threshold voltage level may be a supply voltage rise-time limit, such as from initiation of a power-on reset condition, including a power-up condition. From the time of initiation of a power-on reset, power-on reset threshold voltage level or levels are to be at least met for a supply voltage or supply voltages, respectively, within a rise-time window.
A manufacturer of an integrated circuit may specify particular minimum supply currents and a maximum supply voltage level rise time in order for a customer to select a power supply meeting those specifications. To reduce cost of power supplies needed for use of an integrated circuit, power-on reset circuits may trip relatively late within a supply voltage rise-time window and still properly initialize the integrated circuit. However, with the scaling down in size of integrated circuits, there has been a commensurate reduction in supply voltage level. This has meant that a variation in a supply voltage level has a narrower voltage difference window than previously allowed. This scaling down has also meant less predictability of transistor-threshold voltage, meaning that variation in power-on reset circuit trip points may be large compared with a supply voltage level, which presently is generally one volt or less. Building power-on reset circuits having sufficient precision may be too costly and may add significant complexity.
Accordingly, it would both desirable and useful to provide means for determining supply voltage level validity using a conventional power-on reset circuit.