A bipolar junction transistor (BJT) used as a controlled rectifier (CR) that is driven and controlled by an electronic circuit, such that when (in a first mode of operation) the collector-emitter voltage (VCE) is above an offset voltage (VCEOS) the circuit drives the base of the SR BJT with a base current proportional to the difference between VCE and VCEOS, turning the CR BJT on with a base current proportional to the collector current. When the VCE of the CR BJT falls below VCEOS the drive circuit turns on a second transistor coupled between the base and collector of the CR BJT to turn the CR BJT rapidly off and maintains it in the off state as the collector voltage reverses. When the VCE of the CR BJT exceeds a threshold voltage above the normal conduction VCE range, the control circuit increases the base drive current to accelerate the CR BJT turn-on process. A second mode of operation occurs in the presence of an xe2x80x9cOFFxe2x80x9d drive command to the CR, which turns the CR off when the collector voltage is above the emitter voltage. Turn off of the CR is now accomplished with a third transistor coupled between the base and emitter of the CR BJT.

The present invention relates to controlled rectifiers and, more specifically, to the efficient, economical and optimal driving and control of a bipolar junction transistor (BJT) used as a controlled rectifier (CR).
To facilitate a better understanding of the present invention, the following information on the development of synchronous and controlled rectification and the operation of PN junction and bipolar semiconductor devices is presented.
DC to DC power converters are typically used to stabilize or isolate a power supply signal from upstream irregularities (i.e., voltage/power surges, momentary power outages, etc.). Various transformer and non-transformer based power converters are known in the art. These power converters typically employ a rectifying device to convert either a transformed AC signal, a chopped DC or a similar signal (depending on the power converter arrangement) into a DC output signal. This output DC signal constitutes a relatively stable power supply signal. Depending on the range of voltage (and current) for which the power converter is designed, the power converter may be used, for example, in power supplies for personal electronic devices, laptop or personal computers, engineering workstations and Internet servers. While the present invention is particularly concerned with electronic/digital logic circuits, it should be recognized that the teachings of the present invention are applicable to rectifying device operation in any voltage/current range and for any purpose.
For many years the standard power supply voltage level for electronic logic circuits was 5V. Recently, this voltage level has dropped in many instances to 3.3V and 2.5V, and there are plans within the industry to further reduce this voltage level. As this voltage level drops, however, the forward conduction voltage drop of the rectifying device becomes the dominant source of power loss and inefficiency. For example, a Schottky diode is typically used when a low voltage drop is desired, and a typical Schottky diode has a 500 mV forward voltage drop. This limits the theoretical efficiency of a DC to DC power converter to 80% at two volts output (before other power conversion losses are taken into account). This efficiency limit further drops to less than 67% at one volt output, and 50% at 500 mV output. These efficiency limits are deemed unacceptable.
In addition to concerns about forward voltage drop and other power inefficiencies, power converters and rectifying devices therein are expected to have high power densities. This mandates a higher switching frequency such that less energy is processed in each switching cycle, which in turn permits smaller component sizes. Switching frequencies have risen from 5 to 20 KHz thirty years ago (where the push was to get above the audible range) up to 100 KHz to 1 MHz at present. Thus, technology that does not support rapid switching is not preferred for most rectification applications.
With respect to known semiconductor rectifying devices, these include rectifying diodes (PN and Schottky junction in Si, GaAs, etc.) and rectifying transistors (bipolar and field effect). The forward voltage drop of a rectifying diode can be reduced by design, but only to around 300 mV to 200 mV before a point of diminishing returns is reached where increasing reverse leakage current losses outweigh the decreasing conduction losses. This is due to an inherent physical limit of rectifying diodes and does not depend on semiconductor material or whether the construction is that of a conventional P-N junction diode or a Schottky junction diode. For this reason, amongst others, diodes are not desirable as rectifying devices for low voltage level applications.
Rectifying transistors in which transistor driving is in xe2x80x9csynchronismxe2x80x9d with the direction of current flow across the transistor have increased in popularity due to their favorable forward voltage drops relative to diodes. Typically, the synchronous rectifying transistor is driven xe2x80x9conxe2x80x9d to provide a low forward voltage drop when current flow across the rectifying transistor is in a designated forward direction, and is driven xe2x80x9coffxe2x80x9d to block conduction when current flow across the rectifying transistor would be in the opposite direction.
A controlled rectifier (CR) is typically achieved when a synchronous rectifier may be driven into an off state regardless of the direction of potential current flow across the rectifying transistor. Thus the xe2x80x9ccontrolxe2x80x9d of a CR puts the rectifier into one of two states: a xe2x80x9cnormalxe2x80x9d state where the device behaves as a synchronous (or conventional) rectifier, always conducting when current flow would be in a designated forward direction, and always blocking current flow in the reverse or opposite direction, and a second xe2x80x9cforced offxe2x80x9d state where the rectifier blocks current flow in either direction. Turning a CR off when a rectifier would normally conduct allows the average output voltage (or current) to be reduced, or controlled, by the CR. If the CR is allowed to conduct for only 50% of the time that a conventional or SR would conduct, e.g., the output voltage is reduced by half. The additional control achieved by replacing conventional or synchronous rectifiers with controlled rectifiers can be used to significant advantage in several applications.
In a DC to DC power converter with transformer isolation, control of output voltage is achieved by adjusting the switching times of the input side transistor switch or switches, which requires that the isolated output voltage information be sensed and fed back to the input side. Controlled rectification allows the input transistors to be switched at fixed times, while the conduction times of the CR on the output side are adjusted to regulate the output voltage, avoiding the need to send control information to the isolated input, which can reduce costs and/or improve performance.
Further advantages accrue with two or more output voltages from the same DC to DC converter. Conventional or SR allows only one output voltage to be fully regulated and current limited by controlling the input side transistor switching times; other outputs are at best xe2x80x9csemi-regulatedxe2x80x9d, and often require post-regulation by additional circuitry which increases costs and reduces efficiency. Furthermore, a separate transformer winding is required for each distinct output voltage. Controlled rectification allows each output voltage to be completely and independently controlled with maximum efficiency, and a single transformer output winding can be used for several similar output voltages, reducing costs. For example, a single winding might be used to simultaneously provide 5V, 3.3V and 2.5V outputs with individual CR.
Hybrid approaches are also possible. In the previous example, the 5V output may be regulated by control of the input side transistors, while the 3.3V and 2.5V outputs are derived from the same transformer output winding by CR.
Both the Bipolar Junction Transistor (BJT) and the Metal-Oxide-Semiconductor Field Effect Transistor (MOSFET) have been used as a synchronous rectifier transistor, also termed a xe2x80x9csynchronous rectifierxe2x80x9d (SR). Although the BJT has a longer history of use as an SR, the MOSFET is used almost exclusively at present due to its fast switching speed and perceived ease of driving. BJTs are little used at present due to slow switching speeds in general, and a slow turn-off in particular. However, the BJT has the ability to block current flow in either direction when in an off state, and thus may also be used as a controlled rectifier transistor, also termed a xe2x80x9ccontrolled rectifierxe2x80x9d (CR). The conventional power MOSFET can only block current flow in one direction, and thus cannot be used as a true CR.
The present invention recognizes that the BJT is a conductivity modulated device whereas the MOSFET is not. As a result of this distinction, the BJT can achieve a lower forward voltage drop for a given forward current density and reverse voltage blocking capability. Major technical costs of the lower voltage drop, however, are associated with the requirement to inject, maintain and remove the conductivity modulating stored charge. Nonetheless, the lower conduction voltage of the CR BJT could be used to advantage at lower output voltages, if the BJT switching speed (e.g., turn-off and turn-on) could be improved in a cost-effective and efficient manner (which as discussed below is a purpose, amongst others, of the present invention).
Furthermore, the present invention recognizes that an advantage of using a BJT as a controlled rectifier is that the average output voltage or current of an isolated power converter can be regulated from the output side by modulation of the CR conduction duty cycle. With multiple outputs, the voltage or current of each output can be independently controlled. Since the construction of conventional MOSFETs prevents their use as CRs, and BJTs are considered an obsolete technology for switching mode applications, the technique of controlled rectification to regulate output voltage or current is not widely known at present in the field of high frequency switching mode power conversion.
Though MOSFETs and BJTs have certain similarities in design and construction, they also have substantial differences that impact their behavior and the type of circuits that are suitable for driving them. In order to appreciate the drive requirements of a BJT, it is helpful to first consider the basic semiconductor structures from which most other semiconductor devices are built: the PN junction diode.
P-N Diode Construction and Operation
Semiconductor materials are nearly insulating in the pure state, but they may be doped with impurities to create mobile electric charges which improve their conductivity markedly. These impurities may be either xe2x80x9cNxe2x80x9d type (for xe2x80x9cnegativexe2x80x9d) which produces free electrons in the semiconductor material or xe2x80x9cPxe2x80x9d type (for xe2x80x9cpositivexe2x80x9d) which produces holes in the semiconductor material. The negatively charged free electrons are mobile, and will flow towards a positive charge and away from a negative charge. In P-type semiconductors, the xe2x80x9cpositivexe2x80x9d charge carrier is the local deficiency of an electron, often referred to as a xe2x80x9chole,xe2x80x9d and holes are also considered to be mobile (much as an air bubble is mobile in water). Holes will flow in a direction opposite to that of electrons in the presence of an applied electric field. When no electric field is applied, mobile electrons and holes diffuse with no net flow in any direction.
FIG. 1A illustrates a representative P-N junction diode 1 formed by adjacent regions of equally doped P-type semiconductor 2 and N-type semiconductor 3. Conductive contacts for an anode (A) 4 and a cathode (K) 5 are made to regions 2 and 3, respectively. Diffusion of electrons and holes near the junction causes some to meet, recombine and neutralize each other; conceptually, the electrons fall into the holes. Fixed oppositely charged atoms are left behind, which produce a voltage field opposing further electron and hole flow towards the junction. A small region near the junction, often termed the xe2x80x9cspace charge layer,xe2x80x9d is left depleted of mobile charge carriers and the diode will carry essentially no current if small voltages are applied across the device.
FIG. 1B illustrates the P-N junction of FIG. 1A to which is applied a reverse biased voltage by voltage source 6. The negative terminal of the voltage source is connected to anode 4 and the positive terminal to cathode 5. This causes holes and electrons to move towards the oppositely charged electrodes and away from the junction, creating a larger depletion region. Only a small xe2x80x9cleakagexe2x80x9d current flows across this region and it is due to thermal (or radiation) generated electron-hole pairs.
FIG. 1C illustrates the same P-N junction exposed to a forward biased voltage by voltage source 7 and current limiting resistor 8. Both the holes and electrons can be thought of as being pushed towards the junction. When the forward junction voltage is greater than about 600 mV (for silicon), the depletion region is overcome and the holes and electrons cross over the junction in significant numbers. The electrodes replenish the supply of holes and electrons, and thus create a continuous forward current flow IF. For historical reasons, xe2x80x9ccurrentxe2x80x9d is said to flow in the direction opposite to electron flow. Thus, the P-N junction diode allows current to flow in one direction but not in the other, and may be used to xe2x80x9crectifyxe2x80x9d and alternating current (AC) into a direct current (DC) which flows in only one direction.
FIGS. 2A-2B illustrate P-N junction diodes having unequal doping levels to achieve a higher reverse voltage capability. Diode 11 in FIG. 2A has a heavily doped xe2x80x9cP+xe2x80x9d region 12 and a lightly doped xe2x80x9cNxe2x88x92xe2x80x9d region 13. The lightly doped region has a significant electrical resistance, indicated by resistance symbol 14. The depletion region is much thicker in the Nxe2x88x92 than the P+ material, as equal charges must be drawn from both sides of the junction for recombination. This holds true when a reverse voltage is applied, as shown in FIG. 2B, and it is the much wider Nxe2x88x92 depletion region which sustains a higher reverse voltage before breakdown occurs. (Note that relative depletion region thickness is not shown to scale, typically differing by two to three orders of magnitude.) In FIG. 2C, diode 11 of FIG. 2A is shown with an applied forward biased voltage. This diode initially has a higher xe2x80x9cforward recoveryxe2x80x9d conduction voltage than that of diode 1 (FIG. 1C) due to the voltage drop through resistance 14. The relative hole and electron currents crossing the junction are proportional to the relative doping levels, however, so there are far more holes flowing from the P+ region into the Nxe2x88x92 region than electrons flowing the other way. This heavy injection of holes into the lightly doped Nxe2x88x92 region attracts an equal number of electrons to maintain overall charge neutrality, creating a highly conductive electron-hole plasma which lowers the resistance of the lightly doped region dramatically. This conductivity modulation of resistance 14 causes the forward voltage drop to quickly approach that of diode 1 in FIG. 1C. However, the stored charge in the Nxe2x88x92 region must be removed by electron-hole recombination or a momentary reverse current flow, before reverse voltage blocking with low leakage currents can again be achieved.
BJT Construction and Operation
The semiconductor structure of an NPN BJT is that of two back-to-back PN junction as shown in FIG. 3A. The power BJT 31 of FIG. 3A has an N+ emitter region 32, a central P doped base 33, and an Nxe2x88x92 collector region 34. Conductive electrodes 35-37 connect to the emitter, collector and base regions, respectively.
In FIGS. 3B and 3C, BJT 31 is driven to conduct a collector current (IC) with a base current (IB) from an adjustable current source 38. A current limited voltage source 39 provides a forward bias from collector to emitter. Various regions of operation of BJT 31 are illustrated in FIG. 3D, where the collector-emitter voltage (VCE) and IC are plotted as a function of IB.
In the xe2x80x9clinearxe2x80x9d region of operation, where the collector voltage is well above the base voltage, the base current largely consists of holes (H) flowing from the base region into the emitter (FIG. 3B). The emitter is far more heavily doped than the base, causing a much larger current of electrons (E) to flow from the emitter back into the base. The base region is made very thin, however, and most of these electrons pass through the base region and into the collector, causing a collector current to flow, which is much larger than the base current. Some of the electrons from the emitter recombine with holes in the base, and a small extra base current flows to resupply these holes. An electron-hole plasma also forms in the base region and results in a small stored charge, but these and related effects are generally of minor importance in switching applications and will be ignored. In the linear region of operation, IC is essentially proportional to IB until the current limit point of voltage source 39 is reached, where VCE then falls very quickly for a small increase in IB, as shown in the left portion of FIG. 3D. The current gain xe2x80x9cxcex2xe2x80x9d of a BJT is the ratio of IC/IB in the linear region of operation.
For the BJT used in this illustrative example, the internal collector-emitter junction voltage is approaching zero when the external VCE falls to about 450 mV, due to the voltage drop on the internal collector resistance 30. The internal base-collector junction is now forward biased, and a further increase in base current (known as xe2x80x9cbase overdrivexe2x80x9d) causes holes to be injected into the collector as well as the emitter, as shown in FIG. 3C. Note the change from a linear to a logarithmic IB scale at this point in FIG. 3D. In this xe2x80x9cquasi-saturationxe2x80x9d region the holes injected by the base cause a conductive electron-hole plasma to form in the collector which lowers the resistance of the collector region by conductivity modulation, as occurred in the Nxe2x88x92 region of the diode of FIG. 2C. Higher base currents cause more holes to be injected into the collector, resulting in a denser plasma and a proportional reduction in VCE (the middle region of FIG. 3D) until the collector resistance becomes lower than other parasitic resistances in the BJT. At least initially, the decrease in conduction power loss outweighs the increase in base drive power loss. The xe2x80x9chard,xe2x80x9d xe2x80x9cdeepxe2x80x9d or xe2x80x9cfullxe2x80x9d saturation region is now entered, with VCE an order of magnitude lower than when quasi-saturation was entered. In hard saturation, a higher base current results in little change in VCE while increasing the stored charge in the collector plasma. VCE actually increases somewhat as the base current approaches the collector current in magnitude, due to increased emitter current resistive voltage drops, as shown in the right portion of FIG. 3D. The power loss benefits of base overdrive are lost as deep saturation is approached and entered.
The stored charge must be removed from the collector before the BJT can turn-off and sustain voltage without high leakage currents. Removal of this charge entails a turn-off storage time, which results in an inherently slower turn-off than is achievable with MOSFETs which do not have such a stored charge. Excessive base overdrive thus also slows BJT turn off and increases switching losses.
The foregoing simplified review of semiconductor device operation illustrates that driving BJTs to minimize total losses is somewhat complex. First, a high base drive current pulse must be supplied at turn-on to drive a BJT into quasi-saturation as quickly as possible to minimize turn-on switching losses. Second, the xe2x80x9conxe2x80x9d base drive current must be controlled to provide a near optimal degree of BJT quasi-saturation to minimize the sum of drive and conduction losses over a range of conduction currents. Third, the quasi-saturation voltage v. drive current relationship must take switching frequency into account at higher frequencies, i.e., higher quasi-saturation voltages reduce stored charge, leading to faster turn-off and lower switching transition losses. Finally, the conductivity modulating stored charge must be removed as quickly as possible through a high reverse base drive current, further minimizing turn-off losses. This latter aspect is discussed in more detail in the above-referenced U.S. patent application having Ser. No. 09/971,047 and attorney docket no. CARS0001.
It has long been known that, as a general rule, the xe2x80x9coptimalxe2x80x9d base drive to keep a BJT in a desired state of quasi-saturation to minimize drive and conduction power losses is to operate at a constant forced current gain xcex2f is, defined as:
xcex2f=IC/IBxe2x80x83xe2x80x83(1)
where xcex2f is less than xcex2 in the linear region of operation. Based on both simple theory and empirical measurements, the inventor herein has found that the collector-emitter resistance (Rce) of high gain BJTs (for VCE less than a few 100 mV) may be closely approximated by:
Rce=Kr/IB+Roxe2x80x83xe2x80x83(2)
where Kr is a constant, which depends on the BJT design and construction. Thus, the resistance Rce is the sum of a conductivity modulated term inversely proportional to the base current (Kr/IB) and a fixed resistance (Ro) which is not affected by the base drive current. The collector-emitter voltage, VCE, is the product of IC and Rce, which yields:
VCE=Kr(IC/IB)+(IC)(Ro)xe2x80x83xe2x80x83(3)
If the BJT is to be operated at the forced gain of equation (1), it can be calculated from equations (1) and (3) that:
IB=Kb(VCExe2x88x92VCEOS)xe2x80x83xe2x80x83(4)
where xe2x80x9cVCEOSxe2x80x9d is a collector-emitter xe2x80x9coffsetxe2x80x9d voltage given by:
VCEOS=(xcex2f)(Kr)xe2x80x83xe2x80x83(5)
and xe2x80x9cKbxe2x80x9d is a constant of proportionality:
Kb=1/(xcex2f)(Ro).xe2x80x83xe2x80x83(6)
Thus, the present invention recognizes that the BJT will be driven at an essentially constant forced current gain if the base drive current is made proportional to the difference between the collector-emitter voltage and an offset voltage.
Various prior art circuits for driving a BJT are discussed below after the following definitions and notes. These prior art arrangements include those that drive a BJT used as a conventional transistor and those that drive a BJT used as a synchronous or controlled rectifier.
Definitions and Notes
In the following discussion, and for the remainder of this document, the following definitions and subsequent notes generally apply to circuits using BJTs as controlled rectifiers, unless otherwise stated or intrinsically implied.
1) A xe2x80x9ctransistorxe2x80x9d is an active (controllable) semiconductor device with at least three electrodes, such that the signal present at one electrode controls the state of conduction between the other two electrodes.
2) The generic term Field Effect Transistor (FET) is used to include the Junction Field Effect Transistor (JFET) as well as the MOSFET, which in turn is used generically for any Insulated Gate FET (or IGFET), including the less commonly used Metal-Insulator-Semiconductor FET (MISFET) device.
3) A xe2x80x9cpositivexe2x80x9d voltage will be that which is normally applied to the collector of a BJT relative to the emitter, or to the drain of a FET relative to the source, when the device is operating as a conventional transistor; a xe2x80x9cnegativexe2x80x9d voltage will be one of reverse polarity.
4) The collector voltage of a BJT will be said to be xe2x80x9cabovexe2x80x9d the emitter voltage when it is of a positive polarity (as defined above), and will be said to be xe2x80x9cbelowxe2x80x9d the emitter voltage when it is of a negative polarity, hence xe2x80x9cabovexe2x80x9d is used generally as synonymous with xe2x80x9cpositivexe2x80x9d and xe2x80x9cbelowxe2x80x9d is used generally as synonymous with xe2x80x9cnegativexe2x80x9d.
5) A xe2x80x9cpositivexe2x80x9d current flow is one which occurs in normally conducting BJTs and FETs, i.e., from collector to emitter or drain to source respectively, regardless of device polarity; a xe2x80x9cnegativexe2x80x9d or xe2x80x9creversexe2x80x9d current flow is in the opposite direction. Likewise, a xe2x80x9cpositivexe2x80x9d base current is in a direction to turn a BJT on, or cause it to conduct current, while a xe2x80x9cnegativexe2x80x9d or xe2x80x9creversexe2x80x9d base current is in the opposite direction, which will turn a conducting BJT off, again regardless of device polarity.
6) A BJT xe2x80x9canti-saturationxe2x80x9d circuit is one which decreases the base drive current when the collector voltage falls below some low level, and prevents the BJT from entering deep saturation.
7) A xe2x80x9cpowerxe2x80x9d transistor is a relatively high current (typically greater than one ampere) BJT or FET serving as a conventional transistor or, in the present application, as a controlled rectifier.
8) The essentially symmetrical structure of a BJT allows it to be operated in an xe2x80x9cinvertedxe2x80x9d mode, where the function of the collector and emitter are interchanged. The terms xe2x80x9cemitterxe2x80x9d and xe2x80x9ccollectorxe2x80x9d will be used in their functional context, wherein the magnitude of the emitter current of a conducting BJT is the sum of the base and collector currents of the same polarity, and is therefore larger in absolute value than either the base or collector current.
9) The structure of a power MOSFET is normally asymmetrical, as are some integrated circuit (IC) FETs, with the body of the FET shorted to the source terminal. Low voltage FETs used in ICs may be essentially symmetrical, with the FET body connected to the substrate or to a supply voltage. In the case of a symmetrical FET construction, the terms xe2x80x9csourcexe2x80x9d and xe2x80x9cdrainxe2x80x9d are used in their functional context in a circuit.
10) An NPN BJT and an N-channel FET are considered to be of the same xe2x80x9cpolarityxe2x80x9d whether the FET is a MOSFET or a JFET.
11) A PNP BJT and a P-channel FET are considered to be of the same polarity and of opposite polarity to an NPN BJT or an N-channel FET.
12) Unless otherwise defined, diodes may consist of: a PN semiconductor junction; a metal-semiconductor junction (i.e., a Schottky diode); or a diode connected transistor, wherein the base of a BJT is connected to the collector or the gate of a MOSFET is connected to the drain, or a like device. For a given semiconductor material, the forward voltage drop of a Schottky diode is less than that of a P-N junction diode or a diode connected BJT. The forward conduction voltage drop of a diode connected MOSFET is determined by the drain current vs. gate voltage relationship, which depends significantly on design and construction, but is typically greater than that of a P-N junction diode or diode connected BJT.
13) The terms xe2x80x9crectifierxe2x80x9d and xe2x80x9cdiodexe2x80x9d are often used synonymously, although there are distinctions: a rectifier is a device that allows current to pass in only one direction, whereas a diode is (at least functionally) a two terminal device that may be used as a rectifier.
14) A xe2x80x9ctransconductance amplifierxe2x80x9d is an amplifying device or circuit that produces an output current proportional to an input voltage or to the voltage difference between two inputs. In some cases, the proportionality constant of this amplifier may be defined by a control voltage or current on another input.
The symbols used for transconductance amplifiers herein are those of a conventional voltage input/voltage output operational amplifier with a current source symbol added to the output. A xe2x80x9cdouble circlexe2x80x9d current source symbol is used when the output current may be of either polarity, as first used in FIG. 15A, while a xe2x80x9csingle circle with enclosed directional arrowxe2x80x9d current source symbol is used when the amplifier can only source current (as in FIG. 16A) or only sink current (FIG. 18).
Note that definitions (3), (4) and (5) above are equivalent to conventional usage when applied to NPN BJTs and N-channel FETs, but are the opposite of conventional usage when applied to PNP BJTs and P-channel FETs. This is done to achieve terms in the claims that are independent of the polarity of device used. Also note that:
1) The polarity of an entire circuit may be changed by changing the polarity of all transistors, and reversing the polarity of all diodes, voltage sources and current sources, without changing the essential behavior of the circuit. It is also recognized that FETs may often be substituted for BJTs and vice versa, particularly in a control or logic circuit, without changing the essential nature, function or behavior of the circuit. For this substitution the collector, base and emitter of a BJT are equivalent to the drain, gate and source of a FET, respectively. The principle limitation of device substitution is that devices must remain of the same type when a matching of characteristics is required.
2) BJTs are conventionally considered to be current driven devices, due to the roughly constant ratio between the base drive and collector currents over several decades of current, during which the base-emitter voltage changes by only a few hundred mV. FETs on the other hand are considered to be voltage driven devices, as the gate-source voltage controls the drain current with essentially no gate current flow under steady state conditions.
3) Base current drive for BJTs (e.g., a turn-on or turn-off drive command as discussed below) is usually shown herein as derived from a voltage source and a current determining resistor, but various well known current source circuits may be used instead.
4) The concept of an xe2x80x9coptimalxe2x80x9d BJT base drive current is utilized herein as that drive current which minimizes (or xe2x80x9coptimizesxe2x80x9d) the sum of BJT conduction and base drive power losses for a given collector current. The optimal base drive current may also include the effects on switching losses, as well as BJT operating temperature. However, such xe2x80x9coptimaxe2x80x9d are (as here) typically very xe2x80x9cbroadxe2x80x9d, or insensitive to moderate deviations from optimum conditions. For example, if the base drive current (and drive power loss) is 10% above or below the theoretical optimum, the change in conduction loss will be 10% lower or higher, respectively, and the sum of the two losses will only rise by about 0.5%, and is obviously well within the scope of xe2x80x9coptimalxe2x80x9d drive operation in practice. Operation at twice (or half) the theoretical optimum base drive, on the other hand, will raise the sum of drive and conduction losses about 25%, which begins to represent a significant loss in performance. The extent to which the losses are minimized is a design choice, and operation removed from a theoretical xe2x80x9coptimumxe2x80x9d may be necessary or unavoidable. Circumstances leading to deviations from an optimum BJT base drive include but are not limited to: fabrication and process variations in component values or properties; a requirement to use standard components and/or component values in the drive circuit; the desire or need to xe2x80x9cstandardizexe2x80x9d a drive circuit for a range of BJT SR devices; or the need to operate over a range of frequencies in a given application, such as in a resonant converter.
Prior Art
While there are several prior art techniques for driving a BJT, not all of these are applicable to driving a BJT used as a controlled rectifier (CR). A reason for this inapplicability is that in conventional BJT implemented circuits the applied voltage does not reverse polarity, while in controlled rectification the applied collector emitter voltage does reverse direction. In addition, in controlled rectification a BJT CR must be able to be turned-off rapidly with either polarity of applied voltage; both when the applied voltage reverses (as with a conventional or synchronous rectifier) and when on-demand turnoff is desired with forward voltage applied to achieve controlled rectification. Several prior art approaches or techniques are now discussed. Note that many disclose use of a synchronous rectifier.
FIG. 4 illustrates a transformer-coupled proportional base drive technique, also known as regenerative drive. Turn-on of power BJT 40 is accomplished by turning transistor 41 on momentarily, which draws a pulse of current from voltage source 46 through current limiting resistor 45 and winding 43 of transformer 47. This current is transformed into a pulse of current in the base of BJT 40 through winding 49, turning BJT 40 on. Collector current in BJT 40 is then transformed into a proportional base drive current by the turns ratio of windings 48 and 49. Turn-off is accomplished by turning 42 on momentarily, which pulses sufficient current through winding 44 to overcome the collector current in 48, reversing the base drive current and turning BJT 40 off.
This drive method has numerous advantages, including an optimal base drive during conduction through direct xe2x80x9csensingxe2x80x9d of the collector current, relatively fast turn-on and turnoff, a near minimum drive power requirement, transformer isolation between the drive circuitry and the power transistor and operation from a single drive supply voltage. A disadvantage is an inability to turn the BJT off under xe2x80x9cfault currentxe2x80x9d conditions, when resistor 45 limits the turn-off drive to the point where the BJT 40 base drive current cannot be reversed to initiate the turn-off process. Other disadvantages relate to the requirement for a transformer which tends to be relatively large and expensive and which places limits on the maximum conduction duty cycle due to the need to reset the core magnetization current during the xe2x80x9coffxe2x80x9d interval. A principal disadvantage is that the drive method cannot be directly adapted to CR; any connection of base drive winding 49 to either the collector or emitter of BJT 40 would allow uncontrolled conduction of BJT 40 in one direction or the other.
FIG. 5 illustrates another adaptation of a regenerative drive current transformer to a BJT rectifier. The opposite end of the base drive winding 59 has been moved from the emitter to the collector of BJT 40 to allow automatic turn-on when the VCE of BJT 40 becomes positive, and again there is an optimal base drive current during conduction. Turn-off is also automatic because a reverse current in winding 58 reverses the base drive current in winding 59. Other advantages of the embodiment of FIG. 5 are an absolute minimum of base drive power consumption and no need for base drive voltage sources. Versions of this BJT SR drive method have been utilized in U.S. Pat. No. 3,582,758, granted to Gunn on Jun. 1, 1971, for a Rectifier using low Saturation Voltage Transistors, and U.S. Pat. No. 3,940,682, granted to Park et. al., on Feb. 24, 1976, for Rectifier Circuits using Transistors as Rectifying Elements, among others.
Disadvantages are a relatively slow turn-on and turn-off unless additional circuitry is employed, as discussed by Eiji Sakai and Koosuke Harada in xe2x80x9cA New Synchronous Rectifier Using Bipolar Transistor Driven by Current Transformer,xe2x80x9d published in the proceedings of the IEEE INTELEC 1992 conference, pp. 424-429. The principal drawback of the circuits illustrated in FIGS. 4 and 5 is the requirement for the drive transformer itself, which is relatively large and expensive, and thus unsuitable for large volume, low cost, high power density circuits. Use of a drive transformer also places restrictions on the relative xe2x80x9conxe2x80x9d and xe2x80x9coffxe2x80x9d periods of a driven BJT, as noted above. The drive method also has the same limitation as that of FIG. 4, requiring additional circuitry to enable blocking of current flow in either direction, as required for CR.
Various transformerless BJT drive circuits are illustrated in FIGS. 6 through 10. In the circuit of FIG. 6, switch 61 connects a base driving resistor 62 to a positive voltage source 63 for turn-on of BJT 40, while turn-off is accomplished by switching the resistor to a negative voltage source 64. A faster turn-on and turn-off is provided by surge currents through a xe2x80x9cspeed-upxe2x80x9d capacitor 65, which may have a series resistance (not shown) to limit the surge current. Besides the requirement for a second drive voltage of opposite polarity, the principle disadvantage of this circuit is a constant base drive current regardless of collector current. This causes BJT 40 to be overdriven into deep saturation at collector currents somewhat less than the design maximum, increasing power losses through slower switching and excess drive currents. The xe2x80x9conxe2x80x9d and xe2x80x9coffxe2x80x9d base drive currents, and thus the saturation voltages and switching times are also dependent on (and will vary with) the drive supply voltages. This can become problematic, particularly when the drive supply voltages are derived from a winding on a power converter transformer or inductor.
The need for the second drive voltage can be avoided by using the circuits of FIGS. 7 and 8. In FIG. 7, the negative voltage source 64 is simply eliminated and switch 61 now connects to the circuit xe2x80x9ccommonxe2x80x9d to turn BJT 40 off. There is still a turn-off base current pulse due to the xe2x80x9conxe2x80x9d state voltage on capacitor 65, but turn-on and turn-off will be a little slower unless the size of capacitor 65 is increased to compensate for the lower voltage swing on 62 and 65. An alternative approach to speed up turn-off, which requires more drive charge (product of current and time) than for a fast turn-on, is shown in FIG. 8. Turn-on of BJT 40 is essentially similar to FIG. 6 or 7, but turn-off is accomplished by switching the drive voltage 63 to a second (smaller) BJT 69, through drive resistor 67 and speed-up capacitor 68 which xe2x80x9cshorts outxe2x80x9d the base of BJT 40 to the emitter. Since the turn-off base current can be very high due to the current gain of BJT 69, the turn-off can be faster than with previous approaches if the internal base impedance of BJT 40 is sufficiently low. While the drive circuit of FIG. 6 could be used for synchronous or controlled rectification as long as the negative drive voltage 64 is greater than the reverse collector voltage on BJT 40, the drive circuits of FIGS. 7 and 8 cannot be so used. This is because when the collector voltage on BJT 40 becomes negative during the intended xe2x80x9coffxe2x80x9d period, a positive base current will flow through resistor 62 (in FIG. 7) or BJT 69 (in FIG. 8), turning BJT 40 xe2x80x9conxe2x80x9d in the inverted mode.
The problem of a constant base drive current xe2x80x9coverdrivingxe2x80x9d the BJT at lower collector current in FIGS. 6-8 can be addressed with a Baker clamp anti-saturation circuit. An example is shown in FIG. 9, with diodes 70, 71 and 72 added to the drive circuit of FIG. 6. Diode 71 biases the anode of diode 70 a xe2x80x9cdiode dropxe2x80x9d (about 0.7 V) higher than the base of BJT 40 during the xe2x80x9conxe2x80x9d drive period. When the collector voltage of BJT 40 falls below the base voltage, diode 70 conducts to divert excess base drive current to the collector, reducing the base drive until the collector voltage ceases to fall. This keeps BJT 40 out of deep saturation, but the collector-emitter xe2x80x9conxe2x80x9d voltage is now nearly constant with the collector current; the VCE could be lower at lower currents to reduce conduction loss. Diode 72 allows reversal of the BJT 40 base current for driving turn-off of that transistor.
As shown in FIG. 10, the Baker clamp performance can be improved by the addition of a resistor 73 in series with diode 70. At lower collector currents there will be more base drive current diverted through diode 70 and the increased voltage drop on resistor 73 allows the VCE of BJT 40 to fall to a lower level. Suitable selection of component values allows an essentially constant forced gain of BJT 40 to be achieved. The Baker clamp could also be applied to the drive circuits of FIGS. 7 and 8, but they would still not be suitable for synchronous rectification driving due to SR BJT turn-on with a negative collector voltage applied.
Although the modified Baker clamp circuit of FIG. 10 could be made xe2x80x9cself drivingxe2x80x9d with additional circuitry to determine when the BJT SR should be turned xe2x80x9conxe2x80x9d and xe2x80x9coffxe2x80x9d, it would still have several severe limitations in addition to the requirement for two drive supply voltages. First, the drive current is always at the high level required for the maximum collector current, with the excess drive current simply diverted to the BJT collector at lower collector currents. Thus drive power savings at lower collector currents are not realized.
Furthermore, unlike the circuit of FIG. 5, the circuit of FIG. 10 is not self driving for CR applications. It would require additional circuitry to determine when the CR BJT is to be driven xe2x80x9conxe2x80x9d and xe2x80x9coffxe2x80x9d.
A greater problem is the insufficiently accurate control of IB vs. VCE for CR applications.
One reason for using a rectifying transistor is to achieve a lower voltage drop than obtainable with conventional P-N or Schottky diodes, which can be as low as 300 mV or less for applications below 10V DC output. Thus, BJT CR quasi-saturation voltages must be in the range of a few tens of mV to about 200 mV maximum for there to be any efficiency advantage in using BJTs as controlled rectifiers. This requires a VCE sensing and control accuracy of perhaps 5-20 mV or better over a wide range of ambient and junction temperatures (from xe2x88x9265xc2x0 C. to +125xc2x0 C. ambient in severe applications), with BJT CR junction temperatures perhaps 5-50xc2x0 C. higher than ambient. This is difficult if not impossible to achieve with Baker clamp circuitry. A Baker clamp (and other BJT anti-saturation circuits) controls IB based on the collector-base voltage VCB of a BJT, rather than the collector-emitter voltage VCE, which differs from VCE by the base-emitter xe2x80x9cdiodexe2x80x9d voltage VBE. The problem is that the forward voltage drop of diodes (including the VBE of BJTs) can vary by 50-100 mV or more from one device to another (unless part of the same integrated circuit) and the forward drop changes with temperature at a typical rate of about 2 mV/xc2x0C. Even if diodes 70 and 71 of FIG. 10 are matched and thermally coupled, problems are still encountered with variations in the VBE xe2x80x9cdiodexe2x80x9d voltage of BJT 40, both from device-to-device and with temperature changes. Thus, the Baker clamp can control the IB vs. VCE relationship only to within about 200 mV or so, which is inadequate for low voltage SR applications. Other prior art BJT anti-saturation circuits utilize only a diode coupled between the base and collector of a BJT, without a diode in series with the base drive (such as diode 70 in FIGS. 10 and 11). These might appear to overcome the 2 mV/xc2x0 C. temperature coefficient (TC) problem, with the TC of the anti-saturation diode compensating for that of the VBE of the BJT, but this is not the case. The junction temperature of the BJT, which is dissipating significant power, will not match that of the anti-saturation diode. Even a modest 25xc2x0 C. junction temperature difference will result in about a 50 mV error in the VCE quasi-saturation voltage, which is unacceptable for low voltage SR applications.
Other prior art circuits have also been developed to provide some form of xe2x80x9cself drivingxe2x80x9d for a BJT used as a SR or CR. These circuits are now discussed.
FIG. 11 is a reproduction of FIG. 1 of U.S. Pat. No. 4,716,514, issued to Patel on Dec. 29, 1987, for a Synchronous Power Rectifier. When primary side transistors Q1 and Q2 are on and SR Q3 is to conduct, it is turned on by the voltage on the N2 winding of transformer T1 producing a current flow in R1. When Q1 and Q2 turn-off, the voltage on N2 reverses and turns Q3 off (in a manner similar to FIG. 6, without the speedup capacitor), while Q4 is turned on by the voltage on the N2 winding of inductor L1 causing a current to flow in resistor R2. A problem arises when Q1 and Q2 turn back on as the voltage on N2 of L1 tries to keep Q4 on when it must be turned off. This problem is solved by the addition of winding N4 to transformer T1 which forces Q4 to turn-off by reversing the base current flow of Q4 through diode D3.
Advantages of this circuit are that no drive voltage supplies are required for turn-on or turn-off, and no additional magnetic devices are required. Disadvantages of this circuit, on the other hand, are several. Extra windings are generally required on the existing magnetics and the turn-off drive delivery may be delayed which results in higher losses than with a timely delivered turn-off drive command. Perhaps more significant, are the problems previously discussed in regard to the prior art circuit of FIG. 6. First, the available drive currents vary with changes in operating voltages. A severe example occurs when the output (Vo) is overloaded and the voltage falls to near zero, providing no drive voltage to turn-on Q4 (when Q3 is off) until the voltage on Q4 becomes relatively high, causing large conduction losses. Second, the rectifier base drive currents are xe2x80x9cfixedxe2x80x9d and do not change with collector current, driving the rectifiers into deep saturation at lower current levels, slowing switching and increasing power losses. This circuit as taught lacks the switching speed-up capacitors and Baker clamp circuits discussed previously, though these could be added by a skilled practitioner. Nonetheless, the problem of a lack of precise control of quasi-saturation voltages would remain. This self-driving SR circuit also does not readily lend itself to controlled rectification.
FIGS. 12A and 12B are reproductions of FIGS. 5 and 6 from a paper by W. E. Rippel entitled xe2x80x9cA New Closed Loop Adaptive Base Drive Scheme Minimizes Transistor Drive and Saturation Losses,xe2x80x9d published in the proceedings of Powercon 11 (Section G1, pp. 1-13), April 1984, Dallas, Tex. The designators xe2x80x9cUxe2x80x9d and xe2x80x9cQxe2x80x9d have been added to the operational amplifier (op-amp) and BJT of FIG. 12A, respectively. This paper provides a tutorial on prior art BJT driving techniques and is hereby incorporated herein by reference.
When the switch SW-1 of FIG. 12A is in the ON position, the op-amp U generates a voltage across resistor RS (and thus a base drive current IB) which is proportional to the difference between voltages Vxe2x80x2CE and V1. Since the voltage Vxe2x80x2CE is the sum of the voltage drop on diode D and the collector-emitter voltage (VCE) of BJT Q, this circuit can, over a range of VCE, produce a base drive current proportional to the difference between VCE and an offset voltage as required by equation (4) and as shown qualitatively in FIG. 12B for the curve labeled xe2x80x9cON-STATE.xe2x80x9d Note that equation (4) herein is equivalent to equation (8) in the Rippel paper, although derived by a different procedure. When switch SW-1 is in the OFF position, IB is depressed to the point where it is always negative regardless of VCE, as shown in the FIG. 12B curve labeled xe2x80x9cOFF-STATExe2x80x9d, and BJT Q is turned off.
This circuit achieves proportional base drive without the use of drive transformers, has high base overdrive capability for fast turn-on and turn-off, and has fewer sources of error in controlling IB vs. VCE than that of FIG. 10. Nonetheless, temperature and device-to-device variations in the forward voltage drop of diode D would still produce errors in VCE control that are perhaps ten times those that are tolerable in CR applications. It should be noted, however, that the control accuracy is quite acceptable in its intended application, which (in the paper) is the driving of 450V Darlington BJTs. The stated VCE xe2x80x9csaturationxe2x80x9d voltage of this high voltage BJT is given as 2.0 V maximum under rated conditions. Although not explicitly noted, quasi-saturation voltages seem to be in the range of 1.3V at lower currents to possibly 4V at maximum current, based on the marginally readable oscilloscope photographs of FIGS. 11-16 in the paper.
Although the circuit of FIG. 12A was not designed to drive a BJT as a SR or CR, it could perform the CR function. If SW1 were left in the ON position, the BJT Q would be turned off whenever the current IC tried to reverse, performing as an SR (within the base drive voltage limits of amplifier U). Placing SW1 in the OFF position turns Q off, allowing controlled rectification in principle. The IB vs. VCE control accuracy is inadequate for low voltage SR or CR applications, however, due to the uncompensated voltage drop on diode D, which is added to VCE of Q to produce the voltage Vxe2x80x2CE actually sensed by amplifier U.
The circuit of FIG. 13A is reproduced from FIG. 7 of a paper by the inventor herein entitled xe2x80x9cIsolation of Faulted Power Modules in Low Voltage DC Distributed/Redundant Power Systemsxe2x80x9d published in the proceedings of the PCIMxe2x80x291 conference, pp. 52-64, September 1991, in Universal City, Calif. (resistor designators have been added). This paper proposed the use of a BJT as a reverse current xe2x80x9cblocking diodexe2x80x9d in the output of paralleled low voltage power supplies. The BJT drive circuit of FIG. 13 senses the VCE of Q1 with BJTs Q2 and Q3 coupled to the collector and emitter of Q1, respectively. Feedback through Q5 keeps the currents in matched BJTs Q2 and Q3 equal when Q1 is conducting so that the voltage on resistor R1 is equal to the VCE of Q1, and thus the current in R1 is accurately proportional to VCE. When the VCE of Q1 rises to about 30 mV, the 200 uA current in R1 (and R3) produces a sufficient voltage across R3 to begin turning Q4 on. The base drive current to Q1 from Q4 is then proportional to the increase in the Q1 VCE voltage, satisfying the relationship of equation (4) with a forced gain xcex2f of 167. The base current IB of Q1 is plotted as a function of VCE in FIG. 13B.
An advantage of this circuit is that the accuracy of VCE sensing is many times better than that of the other prior art discussed. In addition, turn-on is automatic and reasonably fast, requiring a momentary high positive VCE on Q1 to generate the turn-on base drive current pulse. No provision was made for turning Q1 off, however, and any VCE below 30 mV simply causes base drive to Q1 to cease. The Q1 self turn-off time is on the order of one to several microseconds which is very slow by fast rectifier standards, but would be adequate for the intended isolation of a faulted power module, particularly compared to the alternative use of fuses with xe2x80x9cblowing timesxe2x80x9d on the order of milliseconds. Adapting this BJT drive method, with its accurate sensing of VCE and control of BJT base current, to perform CR at high frequencies is an object of the present invention.
Another technique for controlling the drive current of a BJT SR to avoid deep saturation is taught in two related U.S. Pat. Nos. 5,721,483 and 5,905,368 granted to Kolluri et. al. on Feb. 24, 1998, and May 18, 1999, respectively, both for a Method and Apparatus for Enabling a Step-up or Step-down Operation Using a Synchronous Rectifier Circuit. FIG. 5 from both of these patents is reproduced as FIG. 14 herein. FIG. 14 illustrates a synchronous rectifying BJT Q1 of the PNP type (as opposed to the NPN type used for illustration of the present invention further below).
Kolluri et al. states in the ""368 patent that xe2x80x9cIn the preferred embodiment, the base drive to the PNP [pass] transistor is controlled such that the collector current is about 0.9 times the collector current value if the device were operating with VCB of 0 V . . . xe2x80x9d for which xe2x80x9c. . . the VCE across the device is approximately 200 mV.xe2x80x9d (Col. 6, lines 32-39.)
In essence, this is accomplished by using Q2 as a xe2x80x9creplicaxe2x80x9d device which is xe2x80x9cscaled 1/N of PNP pass transistor Q1xe2x80x9d (Col. 6, line 50). Transistors Q6 and Q5 force Q3 to operate with a VCB of approximately zero volts, but transistor Q2 (matched to Q3) is made to conduct about 90% of the Q3 collector current by the 1:0.9 scaled current mirror consisting of Q11 and Q12 with emitter resistors R3 and R4, respectively. Since Q2 and Q3 have the same VBE, Q2 conducts only 90% of the collector current that would flow if VCB were zero, and is thus on xe2x80x9cthe edge of saturation.xe2x80x9d Furthermore, the control circuit also maintains a close relationship between the VCE of replica device Q2 and that of PNP pass transistor Q1, through the action of transistors Q7, Q8 and associated circuitry (Col. 6, lines 54-56). Since Q1 and Q2 also have the same VBE, Q1 likewise operates on xe2x80x9cthe edge of saturationxe2x80x9d with a forced gain about 10% lower than the IC/IB ratio at VCB=0.
The principal drawback of this approach is that the drive control circuit requires transistors which are accurate xe2x80x9cscale modelsxe2x80x9d of the SR transistor, and is thus only suitable for applications where the SR transistor and drive control are integrated on the same semiconductor chip. Another limitation of the circuit is the apparent use of resistor R1 to turn Q1 off, a common technique for conventional transistors. Schottky diode D3 in series with R1 would prevent turn-on of Q1 in the inverted mode when VCE reverses, but under those conditions R1 would not provide turn-off drive in the form of a reverse base drive current, and turn-off would be relatively slow.
In view of this and related prior art, a need exists for a low cost BJT controlled rectifier drive circuit and method that achieves:
Automatic turn-on of the BJT CR when current flow would be in a forward direction, and turn-off of the BJT when current flow would be in a reverse direction;
Fast turn-on of a BJT CR through a high initial base drive current;
Fast turn-off of a BJT CR through a high reverse base drive current when the collector-emitter voltage is near zero or negative;
On demand turn-off of a BJT CR regardless of conduction state across the BJT.
Base drive proportional to VCE (above an offset voltage) during the xe2x80x9conxe2x80x9d conduction period without the need for current transformers;
Precise control of the BJT quasi-saturation voltage during the xe2x80x9conxe2x80x9d period, without VCE sensing errors due to mismatched diode drops, uncompensated diode drop TCs, or TC [what is TC?] errors due to differing junction temperatures; and
Operation over a range of drive supply voltages without significant change in current control and drive characteristics.
It is further desirable for reduced cost that the circuit and method be capable of operation with a single drive supply voltage and be suitable for realization as an integrated circuit, with or without the power BJT CR integrated on the same semiconductor chip.
The last requirement mandates that the drive circuit consist almost exclusively of diodes, transistors and resistors, with capacitor requirements limited to small value integratable capacitors or external xe2x80x9csupply voltage bypassxe2x80x9d capacitors or the like. The use of transformers or inductors would also not meet these requirements.
Accordingly, it is an object of the present invention to provide a BJT CR drive circuit that achieves controlled rectification in an energy-efficient and a cost-efficient manner.
It is another object of the present invention to provide such a drive circuit that achieves rapid turn-on and turn-off of a BJT used as a controlled rectifier.
It is another object of the present invention to provide a BJT CR drive circuit that operates over a wide range of drive supply voltages.
It is also an object of the present invention to provide a BJT CR that overcomes the shortcomings of the prior art and achieves the desired features set forth above.
These and related objects of the present invention are achieved by use of an apparatus and method for control and driving BJT used as controlled rectifier as described herein.
The attainment of the foregoing and related advantages and features of the invention should be more readily apparent to those skilled in the art, after review of the following more detailed description of the invention taken together with the drawings.

Apparatus and method for control and driving BJT used as controlled rectifier


Systems and methods are provided for simulating fraction power and control in transportation systems under design conditions and/or utilizing real-time data.

Management of complex electrical systems such as power delivery and management in the transportation sector requires analysis of a wide array of variables. Some variables may include physical properties unique to power delivery lines, stopping and starting power required to move large vehicles such as trolleys and buses, weather, line interruptions, and many others. Use of a discrete resource, namely a specific number of tracks, rails, etc. on which vehicles may move also requires management of complex timetables and budgeting for expected and unexpected delays in the system. Because physical movement of vehicles in the system constantly impacts and influences the electrical load being felt by different parts of the system, analysis may become quite complex and burdensome. To this point an integrated system which is able to catalog and utilize the vast number of variables used in complex transportation systems has not existed in a way that makes it convenient for users to model real world scenarios, run effective simulations, and predict future scenarios in an effective and time efficient manner.


A secondary battery including an electrode assembly; a pouch case accommodating the electrode assembly and including an unsealed part and a sealed part at a wing portion of the pouch case; and an electrode tab including a first side electrically connected to the electrode assembly, and a second side drawn out from the pouch case through the wing portion, and the unsealed part includes a first unsealed part at an outer periphery of the wing portion which is directed toward the second side of the electrode tab, and a second unsealed part at an inner periphery of the wing portion which is directed toward the electrode assembly, and the sealed part is between the first unsealed part and the second unsealed part.

1. Field
Aspects of embodiments of the present invention relate to a secondary battery.
2. Description of the Related Art
Generally, a secondary battery is chargeable and dischargeable and is widely used as a power source for various portable small electronic devices, such as cellular phones, personal digital assistants (PDAs), notebook computers, and the like. In particular, lithium (Li) secondary batteries having an operation voltage of 3.6V, which is about three times the operation voltage of comparable Ni—Cd or Ni-MH batteries, and exhibiting a relatively high energy density per unit weight, have been increasingly widely used.
In a typical Li secondary battery, a lithium-based oxide may be used as a positive electrode active material, and a carbon-based material may be used as a negative electrode active material. The Li secondary battery may be manufactured in various shapes, e.g., cylindrical can types, rectangular or prismatic can types, pouch types, etc.
In addition, Li secondary batteries may be classified as Li ion batteries using liquid electrolyte and polymer electrolyte batteries using polymer solid electrolyte, depending on the electrolyte used therein. Liquid polymer batteries may be classified as pure solid-type Li polymer batteries without electrolyte and gel-type polymer batteries with electrolyte, depending on the solid polymeric electrolyte used therein.
The Li polymer battery using polymeric solid electrolyte includes a pouch generally formed of a metal thin film as a receptacle of an electrode assembly, and peripheral portions of the pouch are sealed in a state in which the electrode assembly is placed into the pouch. Ends of positive and negative electrode tabs electrically connected to positive and negative electrodes of the electrode assembly are drawn out from the pouch to then be electrically connected to a protection circuit board.
In the aforementioned conventional pouch-type secondary battery, edge portions of three sides of the pouch are sealed. Specifically, the sealing of a terminal portion, that is, a top portion, of the pouch is performed linearly in an oblong shape. Here, the sealing may have both advantages and disadvantages depending on whether the sealing is performed closer to a cell space of the pouch accommodating the electrode assembly or to the drawn electrode tabs.
In the former case, an inner layer lifting phenomenon, in which an inner layer of the pouch is melted and lifted as a result of the sealing, is not exhibited outside the outer line of the pouch, which is advantageous in attaining a desired total height margin of the pouch. In this case, however, there is a risk of a separator being partially lifted and fused by the electrode tabs. In addition, several problems, including a reduction in electrolyte impregnating capability, degassing, or the like, may be undesirably caused.
By contrast, in the latter case, occurrence of fusion of the separator is avoidable. However, the inner layer lifting phenomenon may be exhibited, which is disadvantageous in attaining a desired total height margin of the pouch.


Ultrasound imaging is synchronized with measurements. Generating three-dimensional representations is synchronized with measuring one or more parameters. For example, measurements are preformed based on navigating through a volume. The measurements are linked to the corresponding three-dimensional representations. As another example, the user selects a measurement from a graph. A three-dimensional representation of the volume associated with the selected measurement is presented.

The present embodiments relate to three-dimensional (3D) or four-dimensional (4D) imaging. In particular, measurements are provided for 3D or 4D imaging.
3D and 4D ultrasound imaging may show a baby face to the parents or provide medical diagnostic information. Two-dimensional or matrix arrays allowing real-time 3D (i.e., 4D) imaging provide diagnostic information for cardiologists. One alternative is to use one-dimensional arrays, slices of two-dimensional (2D) images created by a mechanically or electronically rotating probe (e.g., wobbler) to form a volume. For orthogonal 3D rendering, parallel rays extend through the volume. Data is rendered to a display as a function of the rays. To obtain an aesthetically pleasing volume image, various filtering methods, opacity curves, tint maps and smoothing filtering are provided. Perspective rendering may alternatively be used. The three-dimensional representations are displayed to a user.
Area, distance, volume or other quantitative data may be obtained automatically or manually.

Synchronized three or four-dimensional medical ultrasound imaging and measurements


A perovskite photoelectric conversion element includes a light transmitting substrate 11, on a front surface of which light is made incident, an oxide porous layer 13, formed on a rear surface of the light transmitting substrate 11 and with metal oxide particles 12 connected in a network, a metal porous layer 15, formed on a rear surface of the oxide porous layer 13 and with metal particles 14 connected in a network, a porous insulating layer 17, formed on a rear surface of the metal porous layer 15, a first electrode layer 18, formed on and across an entirety of a rear surface of the porous insulating layer 17, a second electrode layer 19, connected to the metal porous layer 15 and formed at a portion different from the first electrode layer 18 in a state of being insulated from the first electrode layer 18, and perovskite 20.

Perovskite photoelectric conversion elements (perovskite solar cells) have such advantages as being high in power generation efficiency in comparison to dye-sensitized solar cells, being high in visible light utilization efficiency in comparison to silicon solar cells, being excellent in deformability due to being a thin film, being low in manufacturing cost due to enabling the thin film to be formed by solution coating and drying, etc., and are being actively researched in recent years (see, for example, Non-Patent Documents 1 to 4).

Back contact type perovskite photoelectric conversion element


A semiautomatic control system or control method ensures a driving method which is as energy-saving as possible. In the system, there is both a) manual inputting of control data, for example a definition of a vehicle speed specified value, by the driver of the vehicle and b) automatic determination of the most energy-saving vehicle speed as a function of predefined peripheral conditions, for example a general travel schedule, the topology of a route, vehicle-specific peripheral conditions and the like. An inference system determines which of these two values is smaller, and the smaller value is used to make possible safe and at the same time energy-saving travel (or journey). A safety check of the manual travel speed input value and of the automatically determined travel speed specified value can also be performed to reliably prevent a maximum permissible travel speed value from being exceeded.

The present invention relates to a semiautomatic control system and a method for vehicles.
When journeys are made with vehicles, in particular rail vehicles, numerous peripheral conditions must be taken into account in order to determine control commands such as accelerating, braking or the definition of setpoint speeds. The peripheral conditions are, for example, a general travel schedule, temporary characteristics (for example owing to roadworks or engineering works), signalling devices, specified values set by a control centre, the topology of the route, weather influences and vehicle-specific peripheral conditions such as the mass of the vehicle or traction mass, maximum power and maximum speed. Furthermore, remaining degrees of freedom can be utilized in order to optimize the driving method, for example with respect to minimum energy consumption or minimum wear.
For this reason, it is suitable to use an automatic control system to provide support for a human driver of the vehicle.
Two different categories of control systems are conventionally known for this:
1. Manual control system with auxiliary means
An example of such a control sequence is shown in FIG. 4. In such control systems, on the one hand automatic control actions or control data are determined automatically in a control device by reference to predefined peripheral conditions (step S41a) and displayed to the driver of the vehicle as decision suggestions on a display device (step S42a). In step S41m, the driver of the vehicle inputs control data to an input device, either taking into account the displayed decision suggestions for the control data or not taking them into account. In step S42m, the vehicle is then controlled in accordance with the control data which had been input.
2. Fully automatic control system with manual intervention feature
The control sequence in such a control system is shown in FIG. 5. In such control systems, automatic control actions are applied directly. In step S51, control data for the vehicle are determined automatically with reference to predefined peripheral conditions. This automatically determined control data is output directly to the vehicle control device in step S53, without the driver of the vehicle having access to this control data. However, it is still possible for the driver of the vehicle to intervene manually (step S52) in situations relating to safety (emergencies) by inputting control data.
The first category of control systems, i.e. manual control systems with auxiliary means, is mostly found in contemporary implementations of a driving method which is optimized in terms of energy.
For example, in xe2x80x9cDevelopment of an on-board energy saving train operation system for the shinkansen electric railcarsxe2x80x9d by Shinobu Yasukawa, Shinichiro Fujita, Takehisa Hasebe and Koichi Sato in Quarterly Reports of RTRI, Tokyo, 28(2-4):54-62, 1987, and in xe2x80x9cRechnergestxc3xctzte energiesparende Fahrweise im Regelbetriebxe2x80x9d [Computer-assisted, energy-saving driving method in closed-loop control mode] by Gerhard Voss and Dirk Sanftleben in xe2x80x9cETRxe2x80x94Eisenbahntechnische Rundschauxe2x80x9d [Railway periodical], 47(1):25-31, 1998 a manual control system is described in which times for starting roll-out phases are suggested to the driver of the vehicle by means of an LED. The publication mentioned first additionally discloses that speed suggestions should be displayed, making it possible for this information to be output on a display device, for example a screen.
The second category of control systems, i.e. fully automatic control systems, has hitherto been applied principally in experimental trials. For example, in xe2x80x9cEinfluss der Fahrtechnik auf den Energieverbrauch eines Personenzugsxe2x80x9d [Influence of driving technique on the energy consumption of a passenger train] by Bozetech Sula in Elektrische Bahnen [Electric railways], 88(4):192-198, 1990 a comparison is made between manual and automatic operation, i.e. manual and fully automatic control systems. Here, it is determined by means of trials that the most energy saving driving method can be achieved with completely automatic control. A travel optimizer with a speed controller, a device for target breaking and a device for programmed control of the journey are used to perform control. Within the scope of these investigations it has been recognized that travelling using xe2x80x9cauxiliary meansxe2x80x9d, i.e. suggestions given by the travel optimizer, is relatively demanding for the driver of the vehicle because it is necessary to follow the readings of the measuring devices and the kilometer markings very strictly. In addition, when performing manual operation in response to suggestions it is not possible to achieve such fine gradation corresponding to the situation in which it has occurred.
In European patent application EP 0 755 840 A1, a method and a device for optimizing the driving method of a vehicle, preferably a rail vehicle, is disclosed. Here, both aforementioned categories of control systems are applied. Route data are permanently stored in a computing device before and/or during the journey. During the journey, dynamic status data relating to the distance and/or speed of the vehicle are recorded and fed to the computing device. A recommendation for a driving method which is optimized in terms of energy and/or as wear-free as possible is computed from said data. The computing result is displayed to the driver of the vehicle on a display device. The possibility of using these computing results directly for actuating the vehicle drive is also disclosed, so that a fully automatic control system is obtained in which the vehicle driver can intervene for safety reasons, but is not intended to do so during normal operation.
As a rule, the journey is monitored by safety devices in all control systems. For example, in the publication mentioned above xe2x80x9cDevelopment of an On-Board Energy-Saving Train Operation System for the Shinkansen Electric Railcarsxe2x80x9d a method of monitoring the speed is described in which when a permitted maximum speed (threshold value) is exceeded, automatic braking to a speed below this threshold can be initiated.
In the conventional manual control systems with auxiliary means described above it has, however, proven problematic that the human driver of the vehicle, who must in any case take into account numerous peripheral conditions and display instruments, must absorb and process additional information. This leads to additional loading. In addition, a computed optimized driving method may only be unsatisfactorily implemented owing to reading errors and time delays which inevitably occur. Furthermore, the driver of the vehicle may feel that these auxiliary means of the manual of control system are demotivating in that the decision suggestions remove the need for him to exercise his own initiative.
Such disadvantages no longer occur in a fully automatic control system such as is proposed in EP 0 755 840 A1, for example. In the publication cited above xe2x80x9cEinfluss der Fahrtechnik auf den Energieverbrauch eines Personenzugsxe2x80x9d [Influence of the driving technique on the energy consumption of a passenger train] it has even been shown that a driving method which is optimized in terms of energy can be implemented better in an automatic mode by means of a fully automatic control system.
However, the direct application of automatic control actions which are determined has the main disadvantage that they may be incorrect owing to peripheral conditions which are possibly only unsatisfactorily taken into account by the automatic control system. For this reason, it is currently not yet possible to dispense with the driver, and the driver has the task of monitoring, and if appropriate, intervening in the fully automatic control system, as proposed in EP 0 755 840 A1. However, this can lead to reduced attentiveness of the driver during a relatively long period of automatic operation, and thus entails, inter alia, the risk of in particular situations in which the driver should in fact intervene being overlooked. The direct application of automatic control actions which are determined is also a practical disadvantage because existing safety concepts for the automatic operation which is carried out with them has to be revised.
For this reason, the object of the present invention is to design a semiautomatic control system and control method with which a driving method which is optimized in terms of energy can easily be achieved without losses in attentiveness of the driver of the vehicle inevitably occurring, and without existing safety concepts having to be revised.
It is thus ensured in a simple manner that both a computed optimum control means is precisely implemented and nevertheless the driver of the vehicle remains attentive during the entire journey. Furthermore, possible errors by the driver of the vehicle can be corrected by the semiautomatic control system or control method.
These and further objects, advantages and features of the invention are apparent from the detailed description below of a preferred exemplary embodiment of the semiautomatic control system according to the invention and control method according to the invention in conjunction with the drawing.

Semiautomatic control system and method for vehicles


An object is to provide a showcase in which an attaching height of a guard for preventing falling of goods can be easily changed without changing a shelf guard member for preventing the falling of the goods depending on a height of the goods. The showcase is constituted by bridging shelves for displaying the goods to front and rear supports comprising a plurality of vertically ranging engagement holes, and comprises the guard for preventing falling of goods, disposed above an edge portion of each shelf. This guard for preventing the falling of the goods comprises: a guard main body constituted of a stria or the like; and a guard receiver having a guard holding portion, and the guard main body is disengageably attached to the engagement holes formed in the front support via the guard receiver.

The present invention relates to a showcase in which goods such as canned beverages or plastic-bottled beverages are laid on shelves for displaying the goods.
Heretofore, this type of showcase has been installed in a store such as a supermarket or a convenience store, and used for displaying and dispensing goods. Especially, as a hot dispensing showcase for warming and dispensing a canned beverage or a plastic-bottled beverage, for example, there is a showcase 100 shown in FIG. 9. The showcase 100 comprises a display chamber 102 whose periphery is surrounded with transparent glass 101. In an upper part of the display chamber 102, there are disposed a plurality of stages of shelves 103 each including a heater (not shown) on a lower surface thereof. In a lower part of the display chamber 102, there are disposed a plurality of stages of shelves 104 constituted of racks, and goods are displayed on these shelves 103, 104. A front surface and/or a rear surface of the display chamber 102 is openably closed by a door 105 made of transparent glass. The shelves 103, 104 are supported by shelf supports 106 vertically disposed in four corners of the display chamber 102.
Here, each of the shelves 103 bridged to the upper part of the display chamber 102 comprises: a metal plate 103A; and a frame member 103B disposed on a peripheral edge of the metal plate, and a shelf guard 107 fixed to the frame member 103B by spot welding or the like is disposed on a front edge of each shelf 103. Accordingly, the goods displayed on the shelves 103 are prevented from falling in advance.
On the other hand, the shelves 104 bridged to the lower part of the display chamber 102 comprise racks. As shown in FIG. 9, peripheral edges of the shelves are slightly bent upwards, and prevent the falling of the goods displayed on the shelves 104 in advance.
Moreover, in addition, a wire extending in a horizontal direction above a front end portion of each shelf plate is fixed to the shelf plate by the spot welding or the like. Accordingly, the falling of the goods displayed on the shelves is prevented in advance (see, e.g., Japanese Patent Application Laid-Open No. 8-196395).
However, since goods falling preventing means such as the shelf guard disposed on the conventional shelf is positioned only in a predetermined height from a shelf upper surface, there has been a problem that it is not possible to cope with goods height dimensions diversified nowadays. Therefore, it is necessary to prepare shelves provided with shelf guards in height dimensions corresponding to those of goods such as a 500 ml plastic-bottled beverage, a 350 ml plastic-bottled beverage, and a 50 ml bottled beverage for use in nutrition supplement drink. In this case, every time the beverages to be laid are changed, they have to be disadvantageously changed together with the shelves.
Therefore, it is proposed that the shelf guards be detachably inserted and fixed to the shelves, and the shelf guard having a height corresponding to the height dimension of the goods be selected and used. However, there is a problem that a plurality of shelf guards have to be prepared in order to adjust the heights. When the shelf guard is inserted and fixed to the shelf upper surface, a force applied to a shelf front part is concentrated on a portion into which the shelf guard is inserted. Therefore, especially strengths of the shelf guard and the shelf itself need to be enhanced, and the shelf itself might break as the case may be.


In typical commercial/aerospace applications, synchronous generators provide power to both the high voltage and the low voltage loads. This disclosure describes a method of regulating the output voltage for all of the loads.

Voltage regulation for a synchronous generator includes a voltage regulator for both a high voltage supply and a low voltage supply.
In modern electrical systems, there are various types of loads that require different voltages. These voltages are preferably more or less constant over wide load variations.
In the prior art, different methods of voltage regulation have been identified for a given type of voltage. This is true for both 3-phase AC voltage and a rectified DC voltage.
In the prior art, standard conventional DC/DC converter for the high voltage to low voltage conversion are used. This requires a transformation ratio of 20:1 or higher, and results in large expensive components and a very narrow range of voltage regulation. As a consequence, it is difficult to maintain output voltage constant with desired dynamic response.
The prior art voltage regulation could be characterized as providing adequate voltage control over either a high voltage supply or a low voltage supply. However, the prior art has been deficient in providing adequate control over both.

Multi output voltage regulation of a synchronous generator in a power system


A resource configuration method for switching, comprising: a first network access point determining configuration information for a switching resource pool with a second network access point; the first network access point sending the configuration information for the switching resource pool to a mobile station; the first network access point sending first instruction information to the mobile station, the first instruction information instructing the mobile station to communicate, using a switching resource from the switching resource pool, with at least one of the first network access point, the second network access point, and another mobile station.

An application field of a network system in the future will be extended to Vehicle to X (V2X) communication, including Vehicle to Vehicle (V2V) communication, Vehicle to Infrastructure (V2I) communication, Vehicle to Pedestrian (V2P) communication, Pedestrian to Vehicle (P2V) communication and the like.
V2X communication may effectively improve traffic efficiency, avoid traffic accidents and reduce operation risks. Researches show that V2V communication may solve about 82% of collisions without casualties and about 40% of collisions occurring at crossroads in the United States.
In a V2X communication process, a vehicle is required to keep connected with a network all the time to effectively receive information of a network side, ensure resource allocation reliability and improve data transmission efficiency. However, in some scenarios, for example, when there is no network coverage, vehicles usually communicate according to preconfigured resources, so that it is impossible to ensure ordered information interaction of a large number of connected vehicles, and resource utilization efficiency of a wireless network is reduced to a great extent. Therefore, V2X communication in the future will exist on the basis of vehicle-to-network communication in a biggest range, and it is necessary to ensure smooth connections between vehicles and networks to ensure ordered scheduling and resource allocation of the networks for the vehicles.
In such a process, a vehicle keeps moving and may be faster in some scenarios, so that how to ensure smooth and successful handover of the vehicle between different network APs becomes a key problem urgent to be solved.
A mechanism capable of ensuring handover exists in a conventional cellular network. However, there exists a longer transmission interruption delay in an existing handover flow, and a transient channel change problem for a mobile station which moves fast and the like may not be effectively solved.

Resource configuration method for switching, network access point, and mobile station

GUANGDONG OPPO MOBILE TELECOMMUNICATIONS CORP., LTD.


A spa with a pop-up waterfall apparatus constructed according to the invention includes a container for a reservoir of water and a water-sheeting manifold assembly mounted on the container The water-sheeting manifold assembly includes a water-sheeting manifold that is adapted for movement between a lowered first position and an elevated second position. A waterfall apparatus constructed according to one aspect of the invention includes a housing defining an upwardly opening chamber that extends from a bottom portion of the housing to a top portion of the housing, and it also includes a water-sheeting manifold that fits at least partially within the upwardly opening chamber. The housing includes a water inlet in fluid communication with the upwardly opening chamber that functions as means for coupling water from a separate source of pressurized water to the upwardly opening chamber. The water-sheeting manifold fits in the upwardly opening chamber moveably for upward movement under influence of water pressure in the upwardly opening chamber from the lowered first position to the elevated second position in which the water-sheeting manifold presents a higher profile and discharges a sheet of water simulating a waterfall.

1. Technical Field
This invention relates generally to pools, spas, hot tubs, and the like, and more particularly to a waterfall apparatus for recreational and therapeutic water reservoir containers.
2. Description of Related Art
A water-sheeting manifold assembly is sometimes referred to as a waterfall apparatus. It includes a water-sheeting manifold in the form of an enclosure that receives a flow of water form a pressurized source and discharges a sheet of water that simulates a waterfall. The enclosure defines a chamber into which the water flows. The enclosure also defines an outlet in the form of a horizontally extending slot out of which the sheet of water is discharged. The waterfall is visually appealing and provides a soothing sound.
Some existing recreational and therapeutic water reservoir containers, such as pools, spas, hot tubs, and the like, include some form of waterfall apparatus connected to the reservoir water supply. The waterfall apparatus receives water at an inlet on the waterfall apparatus and discharges a sheet of water through a horizontally extending slot to produce a waterfall that falls to the surface of the water below. Various examples and details of construction are described in U.S. Pat. Nos. 6,450,418; 5,537,696; 5,249,744; and 4,881,280.
Although commonly used, some waterfall apparatuses have certain problems. A typical spa waterfall apparatus, for example, may be mounted on a sidewall of a spa. It extends upwardly from the upper edge of the sidewall (the peripheral edge of the spa) where it pours a sheet of water inwardly and downwardly toward the surface of the water in the spa. One problem with such a waterfall apparatus is that it presents a profile that remains visible and in the way when not in use. Thus, a need exists for a waterfall apparatus that overcomes this concern.
It is an object of this invention to overcome the forgoing and other disadvantages of prior art waterfall apparatuses. This object is achieved by providing what may be referred to as a pop-up waterfall apparatus. It includes a water-sheeting manifold that moves (preferably under influence of water pressure) from a lowered position that presents a low profile to an elevated position in which the waterfall is produced. When not in use, the waterfall apparatus is less visible and out of the way.
To paraphrase some of the more precise language appearing in the claims in order to develop the nomenclature employed, a spa with a waterfall apparatus constructed according to the invention includes a container for a reservoir of water and a water-sheeting manifold assembly mounted on the container. As used in this description and the following claims, the word xe2x80x9cspaxe2x80x9d includes pools, tubs, and any other such containers for a reservoir of water, including those used for recreational or therapeutic purposes.
According to a major aspect of the invention, the water-sheeting manifold assembly on the spa container includes a water-sheeting manifold that is adapted for movement between a lowered first position and an elevated second position. Preferably, the water-sheeting manifold assembly is adapted to raise the water-sheeting manifold from the first position to the second position under influence of water pressure, and a control panel is provided for enabling a user to control a flow of water from a separate source of pressurized water to the water-sheeting manifold assembly in order to thereby selectively raise and lower the water-sheeting manifold.
A water-sheeting manifold assembly constructed according to one aspect of the invention includes a housing defining an upwardly opening chamber that extends from a bottom portion of the housing to a top portion of the housing, and a water-sheeting manifold that fits at least partially within the upwardly opening chamber. The housing includes a water inlet in fluid communication with the upwardly opening chamber that functions as means for coupling water from a separate source of pressurized water to the upwardly opening chamber. The water-sheeting manifold fits in the upwardly opening chamber moveably for upward movement under influence of water pressure in the upwardly opening chamber from a lowered first position to an elevated second position. In the first position, the water-sheeting manifold presents a low profile. In the second position, it presents a higher profile and discharges a sheet of water simulating a waterfall.


An electronic throttle control system with a plastic composite housing member and injection molded motor assembly. The motor assembly is preassembled with a gear shaft member to a metal mounting plate which in turn is injection molded into the composite housing member. A portion of the motor is secured to the electronic throttle control housing during the injection molding process. The distance between the gear shaft member and motor remains constant.

This invention relates to electronic throttle control mechanisms for internal combustion engines.
Valve assemblies for engines and related systems typically utilize rotatable valve members in fluid flow passageways to assist in regulating fluid flow through them. For example, throttle valve members are positioned in air induction passageways for internal combustion engines. The valve assemblies are controlled either mechanically or electronically and utilize a mechanism which directly operates the valve member.
For consideration such as weight, cost, and ease of manufacturing, it is preferred to have the body, cover, and some of the component members or mechanisms made from plastic materials, such as plastic composite materials. However, these materials often can expand or contract in the temperature range extremes typically experienced in vehicle engine environments. Moreover, certain operational components, such as gear members and rotatable shaft members, should be accurately positioned at all times for optimum operations. Also, it is preferable for the motors for electronic throttle control systems to not be enclosed in composite housings due to the need to dissipate heat generated in the use of the systems.
It would be desirable to have an electronic valve control system with a plastic composite housing and cover member and which accurately maintains the operating components in position at all times. It also would be desirable to have an electronic valve control system with an injection molded housing which incorporates the motor in the molding process and still allows sufficient dissipation of generated heat.
It is an object of the present invention to provide an improved electronic valve control system which utilizes molded plastic composite materials and accurately maintains the position of operating components. It is a further object of the present invention to provide an improved electronic throttle control assembly in which the motor is injection molded as part of the housing.
In accordance with the present invention, a metal motor mounting plate is provided which is insert injection molded into the plastic composite material forming the housing. A gear shaft is mounted in the metal mounting plate so that it will be accurately positioned for use with the operational mechanism of the electronic throttle control system. The motor is attached to the mounting plate prior to the injection molding procedure, and the mold and injection molded materials only covers a portion of the motor. Shutoff areas in the mold prevent the injected plastic material from covering certain areas of the motor and gear shaft.
Other features and advantages of the present invention will become apparent from the following description of the invention, particularly when viewed in accordance with the accompanying drawings and appended claims.

Electronic throttle body with insert molded actuator motor


The present invention provides ultralow carbon thin gauge steel sheet and a method for producing the same where coalescence and growth of inclusions in the molten steel are prevented and the inclusions are finely dispersed in the steel sheet, whereby surface defects and cracks at the time of press forming are prevented, growth of recrystallized grains at the time of continuous annealing is promoted, and a high r value (r value≧2.0) and elongation (total elongation≧50%) are exhibited, that is, ultralow carbon thin gauge steel sheet excellent in surface conditions, formability, and workability comprised of, by mass %, 0.00030.003%≦C≦0.003%, Si≦0.01%, Mn≦0.1%, P≦0.02%, S≦0.01%, 0.0005%≦N≦0.0025%, 0.01%≦acid soluble Ti≦0.07%, acid soluble Al≦0.003%, and 0.002%≦La+Ce+Nd≦0.02% and a balance of iron and unavoidable impurities, said steel sheet characterized by containing at least cerium oxysulfite, lanthanum oxysulfite, and neodymium oxysulfite.

In general, for automobiles, household electrical appliances, and other applications requiring excellent workability, for example, as disclosed in Japanese Patent Publication (B) No. 42-12348 and Japanese Patent Publication (B) No. 54-12883, ultralow carbon steel having a C concentration of 0.015 mass % or less and including Ti, Nb, and other strong carbide forming elements are being broadly used. Attempts have been made to further improve workability up to now by improving the method of production. Further, Japanese Patent Publication (A) No. 3-170618 and Japanese Patent Publication (A) No. 4-52229 propose steel sheet excellent in deep drawability, stretch formability, and other aspects of workability by increasing the sheet thickness in the final hot rolling or raising the hot rolled sheet coiling temperature. However, the problem has arisen that the increasing harshness of the hot rolling conditions increases the load on the heating furnace and hot rolling machine.
In the above ultralow carbon steel including Ti or Nb, fine carbides are present in the steel, so recrystallization is remarkably suppressed. For this reason, annealing at a high temperature becomes necessary. There are also issues such as the occurrence of heat buckling or sheet breakage during rolling and the increase in the amount of energy consumption. As opposed to this, as shown in Japanese Patent Publication (A) No. 6-212354 and Japanese Patent Publication (A) No. 6-271978, steel sheet with a low recrystallization temperature has been developed by setting suitable amounts of Mn and P in ultralow carbon steel not containing Nb or Ti and changing the hot rolling conditions. However, in these inventions, Mn or P is added in large amounts, so the alloy cost rises and therefore obtaining steel sheet for ultradeep drawing of a total elongation or 50% or more and a Lankford value (r value) of 2.0 or more is difficult. Further, ultralow carbon steel sheet usually is produced by deoxidizing by Al not yet deoxidized molten steel decarburized to the ultralow carbon range in a vacuum degassing system (RH) etc., that is, is "Al killed steel", so the molten steel contains a large amount of alumina inclusions. These alumina inclusions easily coalesce and join together in the molten steel and remain in the cast slab as large alumina clusters, so at the time of hot rolling and cold rolling, the alumina clusters become exposed at the steel sheet surface and cause surface defects. Further, when the alumina clusters remain inside the steel sheet, they become the cause of cracks, defects, and other flaws at the time of press forming. The formability also sharply falls.
In particular, in ultralow carbon steel, if the workability becomes better, the susceptibility to surface defects or cracks rises and even if going to the trouble of developing steel sheet with excellent workability, the yield obtained as a product is low and a large cost increase is incurred. To deal with these problems accompanying Al deoxidation, for example, as shown in
Japanese Patent Publication (A) No. 61-276756 and Japanese Patent Publication (A) No. 58-185752, the method has been proposed of treating molten steel by Ca to convert the alumina clusters to low melting point calcium aluminate for quick removal by floatation. However, conversion of alumina clusters requires a large amount of Ca. It is known that Ca reacts with the S in the steel to form CaS and becomes a cause of rusting. Further, as shown in Japanese Patent Publication (A) No. 10-226843, the method has also been developed of adding fine amounts of Al and Ti for deoxidation and controlling the inclusions in the molten steel to inclusion compositions with good crushability mainly comprised of Ti oxides, Mn oxides, Si oxides, and alumina.
However, molten steel contains dissolved Al, so if the molten steel is reoxidized by the slag or air, the composition of titania-based inclusions caused by Ti deoxidation changes to the high alumina side and results in aggregation and coarsening, so this is not a fundamental resolution of the problems of surface defects and press defects. Further, the Mn oxides, Si oxides, and Ti oxides have to be made complex, but the upper limit value of the amount of addition of Ti is low, so there was the problem that a high workability material could not necessarily be obtained.

Thin gauge steel sheet excellent in surface conditions, formability, and workability and method for producing the same

Nippon Steel & Sumitomo Metal Corporation