A transmission circuit includes: a modulated signal generator for outputting a modulated signal; an envelope detector for outputting an amplitude modulated voltage; an amplitude roll-off LPF for attenuating the amplitude modulated voltage by using a damping property represented by an exponential function in which an argument is exponentially proportional to a frequency; a power-supply-voltage generator for generating a power supply voltage in accordance with an amplitude value of the amplitude modulated voltage; a phase detector for outputting a phase modulated signal; a phase roll-off LPF for attenuating the phase modulated signal by using a damping property represented by an exponential function in which an argument is exponentially proportional to an offset frequency with respect to a center frequency; and an RF power amplifier.

The present invention relates to circuits for transmitters of radio communications, below denoted as "transmission circuits".
In general, to process modulated signals with amplitude modulation, especially multiple modulation such as Quadrature Amplitude Modulation (QAM), radio-frequency power amplifiers in transmission circuits for transmitting power to antennas need linear amplification. Therefore, class A or AB had been adopted as operation classes of the RF power amplifiers.
However, with the progress of broadband communication, communication methods using subcarrier modulation such as Orthogonal Frequency Division Multiplex (OFDM) have come to be used and high efficiency is not expected any more with the conventional class-A or AB radio frequency amplifiers. That is, in the OFDM modulation, subcarriers are overlapped so that a large amount of power is instantaneously generated at random time and the ratio between the average power and the peak power, i.e., a Peak to Average Power Ratio (PAPR), is high. Therefore, a large amount of DC power always needs to be held in order to allow linear amplification of peak power which is much greater than the average power. With respect to class-A operation, the efficiency is only 50% at the maximum. In particular, in the case of the OFDM modulation, since the PAPR is high, DC power obtained by multiplying current by the difference between the peak voltage for the peak power and the instantaneous voltage for the instantaneous power is lost in the form of heat in almost all the time except for the period in which the peak power is output. Accordingly, the resultant efficiency decreases largely.
As a result, portable wireless equipments using batteries as their power supplies, for example, have shorter continuous operation time, and thus inconveniences arise in actual application.
To solve this problem, a conventional Envelope Elimination and Restoration (EER) technique known as Kahn Technique was disclosed in FIG. 6 of U.S. Pat. No. 6,256,482B1, for example.
In the configuration disclosed in the above patent (see FIG. 6), an input RF modulated signal is detected and divided into two components. One of the components is an amplitude component corresponding to the envelope of the modulated signal. This amplitude component is subjected to an amplitude modulation by an amplitude modulator constituted by, for example, a switching regulator, and is supplied to a power-supply-voltage terminal of an RF power amplifier. The other component is controlled by an amplitude control amplifier (limiter) to have a constant amplitude, and serves a phase modulated signal (phase component) whose phase only is modulated. This phase component is supplied to an RF input terminal of the RF power amplifier.
According to the EER technique, a switching amplifier with high efficiency can be used as the RF power amplifier so that a minimum power supply voltage necessary for power amplification is supplied to the power-supply-voltage terminal of the RF power amplifier, thereby enhancing the efficiency.
As disclosed in Japanese Laid-Open Publication No. 3-34709 (see FIG. 1), for example, another EER technique is proposed. This EER technique is suitable for digital signal processing and obtains a phase modulated signal by a quadrature modulation of a complex envelope signal. In the configuration disclosed in this publication, a modulated signal with a remaining modulated amplitude is supplied as a phase modulated signal to an RF power amplifier.
FIG. 13 is a block circuit diagram schematically showing a conventional transmission circuit using an EER technique. This transmission circuit includes: a modulated signal generator 101 for outputting a modulated signal to two branched lines; an envelope detector 102 for receiving one of the branched modulated signals, detecting the envelope of the received signal and outputting an amplitude component thereof; a power-supply-voltage generator 103 (a DC-to-DC converter) for receiving the amplitude component from the envelope detector 102 and generating a power supply voltage in accordance with the amplitude value; a phase detector 104 (an amplitude control amplifier) for receiving the other one of the branched modulated signals and outputting a phase modulated signal as a phase component; a quadrature modulator 105 for receiving the phase modulated signal from the phase detector 104 and performing a quadrature modulation thereon; and an RF power amplifier 106 for receiving the output from the power-supply-voltage generator 103 at its power-supply-voltage terminal and receiving the output from the quadrature modulator 105 at its RF input terminal.
The modulated signal generator 101 performs a modulation such as QAM or OFDM modulation based on data generated inside the circuit or supplied from the outside and outputs a modulated signal for transmission represented by a complex envelope. The envelope detector 102 obtains the absolute value of the complex envelope representing the modulated signal and outputs an amplitude component. The power-supply-voltage generator 103 performs DC-to-DC conversion, thereby generating a power supply voltage in accordance with the amplitude component. The phase detector 104 keeps the absolute value of the complex envelope at a constant value while maintaining the phase of the complex envelope representing the modulated signal, and outputs a phase component represented by the complex envelope. The quadrature modulator 105 performs a quadrature modulation on the phase component represented by the complex envelope and converts the component into an RF signal, thereby generating a phase modulated signal. The RF power amplifier 106 amplifies the phase modulated signal (phase component) to obtain an RF signal with an amplitude according to the amplitude modulated voltage (amplitude component), thereby outputting a modulated signal whose amplitude and phase are variable.

Matsushita Electric Industrial Co., Ltd.


The present invention relates to a method for diagnosing Q-fever in a subject, the method comprising the steps of: (a) obtaining a sample from said subject, (b) contacting said sample with a source of a Coxiella burnetii antigen and (c) determining the expression level of a pro-inflammatory cytokine such as IFN-γ in said sample at the end of step (b).

Q-fever is a systemic infection caused by the intracellular bacterium Coxiella burnetii. The infection is common in animals, especially livestock, and transmitted to humans mainly through the airborne route. The disease presentation varies widely, ranging from asymptomatic infection, acute febrile illness most often with pneumonia, chronic complicated Q-fever (mainly with endocarditis or vascular infection) and post-Q-fever fatigue.
Currently The Netherlands is faced with the most extensive outbreak of Q-fever, related to goat farms. Since 2007, an increasing number of human cases have occurred. The diagnosis is based on a history of exposure, clinical examination and on PCR and serology. Due to the difficulty in culturing Coxiella burnetii in the laboratory, the variation in the antibody response, and the difficulties to standardize serology, the laboratory diagnosis is often not easy. Quite some weight is being given to the phase dependent antibodies, where phase 1 antibodies are seen in chronic infection and the phase 2 antibodies are related to acute infection. With accumulating experience in complicated cases in the current Dutch epidemic, we have experienced that the results of the serology may be equivocal.
If diagnosed in the acute stage, the disease can be cured with a relatively short course of the antibiotic doxycyclin, but chronic Q-fever is a much more difficult-to-treat infection. Chronic Q-fever has a high mortality rate. A special situation is Q-fever in pregnancy, in which the risk for the unborn and the management is not entirely clear.
A special area where the diagnostic tool is critical is the Q-fever vaccination of humans. Currently, patients are screened with serology and a skin test, and only if both tests are negative, vaccination is regarded a safe procedure. The skin test is not easy to perform and laborious. Its sensitivity and specificity are unknown.
There is a need for a specific and sensitive method for diagnosing Q-fever in a subject, circumventing all the drawbacks of existing methods. In addition, the current tests (with the possible exception of the skin test) do not assess the state of specific cellular immunity, which is needed for cure of the infection.

Method for diagnosing Q-fever using a cellular immunological test


A radar system includes a controller equipped with memory for storing data. The controller is configured to receive a time-domain signal representative of a reflected signal detected by an antenna, and transform the time-domain signal into a plurality of range datasets. Each range dataset corresponds to one of the plurality of chirps, each range dataset is represented by a series of values assigned to a plurality of range bins, and each of the values includes a sign bit. The controller is also configured to compress the plurality of range datasets by storing in the memory a portion of each of the values assigned to at least one of the plurality of range bins, wherein the portion is defined to exclude a first number of redundant sign bits of each value. The controller may further compress the portion by retaining a second number of bits of the data by excluding some of the least significant bits of each value.

It is known that radar systems using a Fast Chirp Waveform to emit a radar signal often accumulate large amounts of processed data in memory as arrays of data produced by a Range Fast Fourier Transform (FFT) and a Doppler FFT. The memory size depends on a variety of system parameters, but it is not uncommon for up 10,000,000 bytes (10 MB) to be temporarily stored. To minimize system cost, it is desirable to reduce the amount of memory necessary to implement a Fast Chirp Waveform.

Radar data compression system and method


In a fixing apparatus using a fixing belt having an elastic layer as a fixing member and an image forming apparatus mounting such a fixing apparatus thereto, before a temperature detecting unit detects temperature change caused upon recording material conveying into a nip region, an electric power required to be applied to a heating element to heat the heating element is corrected to a predetermined electric power, and, when the fixing apparatus is started-up upon starting of print, at a predetermined timing, the electric power required to be applied to the heating element to heat the heating element is corrected to the predetermined electric power and a value of the predetermined electric power is substantially the same as an electric power value required for reaching the fixing apparatus to the predetermined power. The value of the predetermined power is determined in accordance with heat capacity of the recording material or a heat accumulating degree of the fixing apparatus on demand.

1. Field of the Invention
The present invention relates to a fixing apparatus and an image forming apparatus.
More specifically, the present invention relates to a fixing apparatus comprising, at least, a heating element, an electric power supplying portion (heating means) for supplying an electric power to the heating element, at least one or more temperature detecting means, a first rotary member shifted together with a recording material and a second rotary member for cooperating with the recording material to define an abutment portion therebetween and wherein a temperature of the first rotary member is controlled by performing feedback control of the electric power supplied from the electric power supplying portion to the heating element on the basis of a temperature detected by the temperature detecting means and the recording material bearing an image is nipped and conveyed at the abutment portion thereby to heat the recording material, and an image forming apparatus, for example, such as an electro-photographic copier, printer, facsimile and the like, having such a fixing apparatus.
2. Related Background Art
In recent years, colorization in an image forming apparatus such as a printer, a copier and the like has been progressed. As a fixing apparatus used in such an image forming apparatus, a fixing apparatus of heat roller type including a fixing member having an elastic layer is well known.
However, in the fixing apparatus of heat roller type having the elastic layer, there was a problem that a heat capacity of the heat roller itself becomes great and a time (warm-up time) required for increasing a temperature of a fixing roller up to a temperature suitable for toner image fixing becomes long, and, further, the cost of the fixing apparatus was increased.
As a fixing apparatus having short warm-up time, a fixing apparatus of belt fixing time widely used in a monochromatic image forming apparatus is well known. A schematic constructional view showing one example of such a fixing apparatus is illustrated in FIG. 24.
The fixing apparatus is generally designated by the reference numeral 201. A fixing belt unit 202 is an assembly comprising a heat folder 207 having a half circular bucket-shaped cross-section, a fixing heater 204 secured to a lower surface of the heater holder 207 along a longitudinal direction of the heater holder (perpendicular to the plane of FIG. 24) and a fixing belt 203 having an endless thin layer and loosely mounted around the heater holder 207 having the fixing heater.
An elastic pressurizing roller 205 has metal core both ends of which are rotatably supported between side plates of the fixing apparatus.
The fixing belt unit 202 is disposed on the elastic pressurizing roller 205 in parallel with the pressurizing roller 205 with the fixing heater 204 facing downwardly, and both ends of the heater holder, 207 are pushed upwardly with a predetermined urging force by means of biasing means (not shown). With this arrangement, a fixing nip portion 206 having a predetermined width is formed by abutting a lower surface of the fixing heater 204 against an upper surface of the elastic pressurizing roller 205 with the interposition of the fixing belt 203 in opposition to elasticity of the pressurizing roller.
The elastic pressurizing roller 205 is rotatingly driven at a predetermined peripheral speed by means of a driving mechanism (not shown) in a direction shown by the arrow. By such rotation of the elastic pressurizing roller 205, a rotational force is applied to the fixing belt 203 at the fixing nip portion 206 by a friction force between the elastic pressurizing roller 205 and an outer peripheral surface of the fixing belt 203, with the fixing belt 203 is rotatingly driven around the heater holder 207 at a peripheral speed substantially corresponding to the peripheral speed of the elastic pressurizing roller 205 in a direction shown by the arrow in a condition that an inner surface of the fixing belt is closely contacted with and slid on the lower surface of the fixing heater 204 at the fixing nip portion 206.
The fixing belt 203 is constituted by a heat-resistive resin endless belt having a thickness of about 50 μm and a mold releasing layer (coating layer made of fluoro-resin or the like) formed on the endless belt and having a thickness of 10 μm. Further, in order to reduce a heat capacity of the fixing belt 203, any elastic layer is not provided on the fixing belt 203.
The fixing heater 204 is constituted by a resistive heating element formed on a ceramic substrate. Temperature detecting means 209 is contacted with a rear surface of the fixing heater 204 to detect a temperature of the fixing heater 204, and temperature control is performed by means of control means (not shown) by controlling an electric power supplied to the fixing heater 204 so that the temperature of the fixing heater 204 becomes a desired temperature.
In a condition that the elastic pressurizing roller 205 is rotatingly driven and the fixing belt 203 is rotatingly driven and the fixing heater 204 is started-up to the predetermined temperature to be subjected to the temperature control, a recording material P bearing a unfixed toner image t is introduced between the fixing belt 203 and the elastic pressurizing roller 205 at the fixing nip portion 206. The recording material P together with the fixing belt 203 is pinched and conveyed at the fixing nip portion 206 while closely contacting a surface of the recording material bearing the unfixed toner image with the outer surface of the fixing belt 203. During such pinching and conveyance, heat from the fixing heater 204 is applied to the recording material P via the fixing belt 203 and the recording material is subjected to pressure from the fixing nip portion 206, with the result that the unfixed toner image t is fixed onto the recording material P by heat and pressure as a permanent fixed image. After passed through the fixing nip portion 206, the recording material P is self-stripped from the surface of the fixing belt 203 by curvature and then is discharged.
In the fixing apparatus 201 having the above-mentioned construction, since the heat capacity of the fixing belt 203 is very small, after the electric power is applied to the fixing heater 204, the temperature of the fixing nip portion 206 can be increased, for a short time, up to a temperature capable of fixing the toner image.
However, in a case where the belt type fixing apparatus 201 using the fixing belt 203 having no elastic layer is used as a fixing apparatus of a color image forming apparatus, since the fixing belt 203 as the fixing member has no elastic layer, the surface of the fixing belt 203 cannot follow unevenness of the surface of the recording material, unevenness caused by presence/absence of the toner layer and unevenness of the toner layer itself, thereby causing difference in heat amount supplied from the fixing belt 203 between the recessed portions and the protruded portions. In the protruded portions adequately contacted with the fixing belt 203, since the heat from the fixing belt 203 is well transferred, the great heat amount is given; while, in the recessed portions which are not contacted with the fixing belt 203 so well, since the heat from the fixing belt 203 is hard to be transferred in comparison with the protruded portions, given heat amount is small. As such, since the heat amount is differentiated due to the unevenness of the toner layer, a fusing condition of the toner becomes uneven, thereby generating gloss irregularity, which affects an influence upon the fixed image.
Particularly, in the color image, since plural color toner images are overlapped and color-mixed, the unevenness of the toner layer is greater in comparison with the monochromatic image, and, thus, in the case where the fixing belt 203 has no elastic layer, gloss irregularity of the fixed image becomes greater, thereby worsening image quality. Further, in a case where the recording material P is an OHP sheet, when the fixed image is projected, light scattering is generated due to the fact that the surface of the fixed image is not microscopically uniform, thereby resulting in reduction in permeability.
Although it can be considered that the fixing belt 203 having no elastic layer is used and silicone oil is coated on the fixing belt 203 so that the heat is well transferred onto the unevenness of the recording material P and the unfixed toner image t, there were problems that the cost is increased and that the fixed image and the recording material P are contaminated by oil.
To avoid this, a fixing apparatus constituting a low cost color on demand fixing apparatus by using a fixing belt having an elastic layer in a belt fixing apparatus has been proposed (for example, refer to Japanese Patent Application Laid-Open No. 11-15303).
FIG. 25 is a schematic constructional view of a belt fixing apparatus using a fixing belt 203 having an elastic layer as a fixing member. Constructional members and parts common to those in FIG. 24 are designated by the same reference numerals and explanation thereof will be omitted.
In a case where this fixing apparatus is used, since heat transfer characteristics of silicone rubber used in the elastic layer of the fixing belt 203 is small, temperature response of the fixing belt 203 is bad and an increasing speed of a temperature of a fixing sleeve is very slow in comparison with the temperature increasing speed of the fixing heater 204. Further, a difference between the temperature of the fixing heater 204 and the temperature of the fixing belt 203 reaches several tens of ° C. at the maximum, and, such temperature difference is greatly differentiated between idle-rotation and sheet passing. Thus, as is in conventional cases, in a system in which the temperature detecting element 209 is disposed on the rear surface of the fixing heater 204, it was very difficult to control the temperature of the fixing belt 203.
To avoid this, in the apparatus shown in FIG. 25, the temperature detecting element 209 is disposed on the inner surface or front surface of the fixing belt 203 rather than the fixing heater 204 to detect the temperature of the fixing belt 203 itself and the temperature control of the fixing belt 203 is performed by controlling the temperature of the fixing heater 204 by feedback control such as PID control.
By using such a construction, the temperature of the fixing belt 203 can be controlled more accurately.
However, as a result that a fixing operation was performed by using this fixing apparatus, the following two problems occurred.
(1) Problem 1
From a condition that constant temperature control is performed to stabilize the temperature, when the recording material P is passed, immediately after the recording material P rushes into the fixing nip portion 206, the temperature of the fixing belt 203 is decreased abruptly, and, thereafter, although the temperature is increased, overshoot is generated, thereby resulting in great temperature fluctuation.
This phenomenon will now be explained.
In the conventional fixing apparatus, after PID control is continued in an idle rotation condition, the recording material P is passed. In this case, as shown in FIG. 5, immediately after the recording material P conveys into the fixing nip portion 206, the temperature of the inner surface of the fixing belt 203 is decreased abruptly, and, thereafter, although the temperature is increased, overshoot is generated. Concretely, as shown in FIG. 5, in the idle rotation before the sheet passing, although the temperature of the inner surface of the fixing sleeve 203 is stabilized, immediately after the recording material P conveys into the fixing nip portion 206, it takes about 10 seconds until the temperature of the inner surface of the fixing belt 203 is decreased up to a temperature lower than a target temperature by about 10° C. and then is increased up to a temperature greater than the target temperature by about 7° C. and then is stabilized. During a period from immediately after the recording material P conveys into the fixing nip portion 206 to when the temperature of the inner surface of the fixing sleeve 203 is stabilized, temperature ripple (difference between a maximum temperature value and a minimum temperature value within a given time) was about 17° C. In subsequent continuous sheet passing, stable temperature control (temperature ripple of about 60° C.) was achieved.
In the condition that the temperature of the inner surface of the fixing sleeve 203 is stabilized during the idle rotation, the electric power being applied to the fixing heater 204 is about 80 W, and in the condition that the temperature of the inner surface of the fixing sleeve 203 is stabilized in the subsequent continuous sheet passing, the electric power being applied to the fixing heater 204 is about 300 W which is substantially constant.
On the other hand, immediately after the recording material P conveys into the fixing nip portion 206, behavior that the electric power is increased up to about 500 W due to the temperature reduction and fluctuates by about 20 seconds and then is stabilized at about 300 W was indicated. This phenomenon could not be solved even when each gain in the PID control is adjusted, and it was difficult suppress the temperature fluctuation for more.
In this way, when the temperature fluctuation of about 17° C. is generated, in a color image forming apparatus of electro-photographic type used in the test, the fixing abilities of the outputted prints were greatly dispersed and gloss properties were greatly changed thereby to worsen image quality. Furthermore, depending upon the recording material and/or printing pattern, poor fixing occurred due to great temperature fluctuation.
Further, this phenomenon became more and more conspicuous as a process speed was increased, and, in the measurement of FIG. , although the process speed was 87 mm/sec, when the process speed was 190 mm/sec, the temperature of the inner surface of the fixing belt 203 was decreased to a value lower than the target temperature by about 20° C. immediately after the recording material P conveys into the fixing nip portion 206, and thereafter, the temperature was increased to a value greater than the target temperature by about 8° C. to provide temperature nipple of about 28° C. In this case, the fluctuation range of the gloss property is further increased and partial poor fixing is also worsened.
(2) Problem 2
In the start-up, the overshoot is great. Further, in the start-up, the temperature nipple is great, and, after the start-up and upon the recording material rushing, dispersion of temperature becomes great.
This phenomenon will now be explained.
In this fixing apparatus, start-up temperature control sequence is constituted by the following two steps a and b.    a. "start-up ((fix) electric power control"; and    b. "PID control"
The "start-up electric power control" in the step a is control in which a constant electric power is applied in order to quickly start-up the temperature of the fixing apparatus and to ensure on-demand, and, in this example, the electric power of 1000 W is applied to the fixing heater 204. In this case, the fixing belt 203 is heated by the fixing heater 204 while being driven by the rotation of the pressurizing roller 205. When the detection temperature of the temperature detecting means 209 reaches a predetermined temperature (target temperature of −20° C.: thus, for example, when the target temperature is 190° C., 190° C.−20° C.=170° C.), the sequence goes to the "PID control" in the step b, where the electric power applied to the fixing heater 204 is controlled the temperature of the rear surface of the fixing belt 203 approaches the target temperature by means of the PID control.
When the above controls are used, as shown in FIG. 21, the overshoot is generated and the temperature ripple due to the overshoot becomes great.
If the overshoot and the temperature ripple become great in the start-up, the following two problems will occur.
I. If operations at the high temperature due to the overshoot in the start-up are repeated, damage on various parts of the fixing apparatus becomes great, thereby shortening a service life of the fixing apparatus.
II. Since the temperature nipple is great in the start-up, at the moment when the recording material P enters into the nip portion, the temperature is not stabilized, and, since the temperature fluctuation becomes great while the recording material P is passing through the fixing nip portion 206, gloss fluctuation in each outputted print becomes great, which results in an unfavorable image. Further, in dependence upon the kind of the recording material P and/or printing pattern, poor fixing will occur at areas where the temperature is decreased.
On the other hand, it is possible to suppress the overshoot more or less by starting-up the temperature gently with a small electric power. In this case, however, it takes a long term until the fixing apparatus is started-up to the predetermined temperature, thereby worsening the on-demand property.
In this way, when the conventional fixing apparatus controlling method is used, there is a trade-off relationship between the on-demand property and the stability of the temperature control.
The Inventors zealously investigated such a phenomenon and, it was found that these problems are generated by the following two reasons:
1) Since heat transfer characteristics of the silicone rubber used in the elastic layer of the fixing belt 203 is small and the heat capacity is great, response is worsened during a period from when the electric power is applied to the fixing heater 204 and to when the temperature of the fixing belt 203 is increased.
2) Since the position of the temperature detecting element 209 for detecting the temperature of the rear surface of the fixing belt 203 is spaced apart from the fixing heater 204 as the heating member, detection timing is delayed.
That is to say, in the feedback control as represented by the PID control, since change in the control amount is detected and an operating amount corresponding thereto is added, if the temperature control tries to be performed on the basis of the PID control, time lag due to the above two reasons 1) and 2) becomes great, and, if the change in temperature is great as is in the start-up and the recording material P rushing, even when the detection result of the temperature detecting element 209 is reflected upon the electric power control, since the actual temperatures of the fixing heater 204 and the fixing nip portion 206 have already different values, correct temperature control cannot be achieved, with the result that the overshoot and hunting (temperature ripple) are apt to occur.

Fixing apparatus and image forming apparatus with temperature controller increasing electric power substantially at a timing with temperature degradation


The present invention relates to a method for providing control information in a mobile communication system. More specifically, the invention relates to a method for providing control information through a network in mobile communication. The method comprises: a step for configuring the information associated with plural frequency resource groups and the power levels thereof in the control information of preset data format, wherein the plural frequency resource groups are for application of FFR (Fractional Frequency Reuse), and a step for transmitting the control information to at least one terminal.

Frequency reuse is one of methods for increasing the number of channels per a unit area in a cellular system. Because a radio wave gets weaker with distance, radio waves interfere less with each other in places apart from each other by a specific distance and thus the same frequency channel can be used in the places. Use of the same frequency in a plurality of areas based on this principle may lead to a great increase in subscriber capacity. The efficient frequency utilization is called frequency reuse. An identifiable geographical unit is called a cell (mobile communication cell) and frequency channel switching between cells for seamless communication is called handoff. In an analog cellular communication scheme, frequency reuse is essential. A frequency reuse ratio is one of parameters representing frequency efficiency in a cellular system. The frequency reuse ratio is calculated by dividing the total number of cells (sectors) using the same frequency by the total number of cells (sectors) in a multi-cell structure.
A 1G system (e.g. AMPS) has a frequency reuse ratio less than 1. For instance, in a 7-cell frequency reuse scheme, the frequency reuse ratio is 1/7. Compared to the 1G system, a 2G system (e.g. CDMA and TDMA) has an improved frequency reuse ratio. For instance, the frequency reuse ratio may reach 1/4 or 1/3 in GSM being a combination of FDMA and TDMA. In 2G CDMA and 3G WCDMA systems, the frequency reuse ratio may reach 1, thus increasing spectral efficiency and reducing network deployment cost.
When all sectors of one cell and all sells of one network use the same frequency, a frequency reuse ratio of 1 can be achieved. However, the frequency reuse ratio of 1 in a cellular network means that users at a cell edge have degraded signal reception performance due to interference from a neighbor cell.
In OFDMA, a channel is divided into subchannels and thus a signal is transmitted on a subchannel, without using the entire channel as is done in 3G (CDMA2000 or WCDMA). This feature of OFDMA may simultaneously improve the throughputs of users at the center of a cell and users at the boundary of the cell (the edge of the cell). Specifically, the cell center is near to a BS and thus safe from co-channel interference of a neighbor cell. Accordingly, inner users at the cell center may use all available subchannels. However, cell-edge users may use only a part of all available subchannels. Frequencies are allocated to users at the boundary between adjacent cells such that each cell may use a different subchannel. This scheme is called FFR (fractional frequency reuse).

Method for providing control information associated with fractional frequency reuse


A portable communication apparatus having a camera module with a camera lens. The apparatus comprises a correction lens arranged to facilitate for said camera module to register a sharp image of an object which is in an immediate vicinity of said correction lens. The apparatus also comprises a controller coupled to said camera module and being arranged to perform an analysis of the registered image of said object and provide an output which is indicative of a result of said analysis.

To input data in a portable communication apparatus, for example a mobile phone, there are besides pushing keys on the apparatus many different types of man-machine interface solutions.
One example is a space-saving optical mouse for a mobile phone developed by Dutch researchers at the Philips Center for Industrial Technology and described on the Internet page [http://optics.org/articles/news/8/6/23/1]. The optical mouse is made up of laser diodes and a detector placed beneath a lens. The light from the laser is focused on an object, such as a fingertip, and is reflected back and thereby mixes within the internal laser cavity. A disadvantage with this technique is that it is implemented using expensive, dedicated laser technology.
The use of fingerprint is well known for controlling access to different types of systems. In WO 01/11577 A1 a system for checking the right to access sensitive information using biometric data is described.
When sharing a mobile phone between users today the users also have to share the memory content of these mobile phones, that is phonebook and calendar and the only possibility for secrecy and privacy is exchanging SIM cards when switching user. This way the different users at least get their own phonebook and inbox for SMS-messages.

Mobile communications equipment with built-in camera


An interface device between an electrical network and energy consumer systems, including an input converter with a first winding connected on one side with a reverse blocking module and on the other with a first semiconductor chopper device, second and third windings having an end connected to output converters via first and second reverse blocking modules, and another end connected to the output converters, via a switching cell, a fourth winding having an end to be connected to the network, via a second semiconductor chopper device; and a control module for the semiconductor chopper devices. The interface device also includes a buffer capacitor having a first end connected to the output converters, and another end, to which a reference voltage is applied, isolated from the network; and a power reserve having an end to be connected to the network and another end connected to the fourth winding.

The invention relates to an interface device between an electrical network and electrical energy consumer systems, notably on board an aircraft, for example an aeroplane.
Current electronic systems require an interface device between an electrical network and various consumer systems which accomplish functions required of them. This interface device enables a voltage delivered by a direct current electrical network or, by extension, resulting from the rectification of an alternating voltage originating from an alternating electrical network, to the different consumer systems, but also enables them to avoid the disturbances to which the electrical network is subject. These disturbances may be network power dropouts or a power voltage below a threshold. Below this threshold the interface device no longer operates satisfactorily. In the remainder of the description the term dropout will be used, but this also encompasses a network supply voltage below the threshold.
A power reserve function, for example a capacitor, enables electrical energy to be stored, and protection to be provided against dropouts. A galvanic isolation function enables the current between two electrical circuits electrically connected to one another to be prevented from flowing.
FIG. 1 illustrates a functional diagram which places an interface device 10 between a direct current electrical network RE and a consumer system 11. At the output of the interface device 10 a modified, isolated electrical network 12, which is protected against dropouts, is obtained in this manner. The main existing solutions in terms of architecture to produce such an interface device are as follows:                use of two converter stages: the first to produce a galvanic isolation function, and the second to manage a power reserve function;        use of a converter stage allowing management of a power reserve function, and possibly a galvanic isolation function.
In a first solution, two converter stages are used to obtain a regulated output voltage and a power reserve. An existing topology is illustrated in FIG. 2. This FIG. 2 shows several DC-DC output converters C1 to CN, for example of the "Buck" type. In the remainder of the description the term "several" means at least two. A converter of the "Buck" type, also called a serial voltage chopper, is a switched mode power supply which converts a direct current voltage into another direct voltage of lower value. As a variant, these output converters C1 to CN could be "Boost" type converters. A converter of the "Boost" type, also called a parallel chopper, is a switched mode power supply which converts a direct current voltage into another direct voltage of higher value.
Each output converter C1 to CN has two inputs; these two inputs are intended to be connected to an electrical network RE; for one of them the connection is made through a first switch 20. A capacitor Ctampon ["Cbuffer"] is connected between the two inputs of the output converters C1 to CN. These converters C1 to CN deliver regulated output voltages Vout1 to VoutN.
There is also a "Boost" type input converter 21 having two inputs connected to the electrical network RE and an output connected to an input of each of the output converters C1 to CN through a second switch 22, wherein this output is also connected to a terminal of a power reserve Cres, the other terminal of which is connected to a ground. The input converter 21 enables the power reserve Cres to be charged to a voltage higher than that delivered by the electrical network RE. The voltage delivered by the electrical network RE is called Vin.
The output converters C1 to CN enable galvanic isolation between the electrical network RE and the output voltages Vout1 to VoutN to be accomplished. They must adapt to the variations of voltage Vin of the electrical network RE, and to those of voltage Vres at the terminals of the power reserve Cres. To choose the power source of the output converters C1 to CN which accomplish the galvanic isolation it is necessary to put first and second switches 20 and 22 in place.
The first switch 20 enables the output converters C1 to CN to be disconnected from the electrical network RE, and second switch 22 enables the power reserve Cres to be connected at the input of the output converters C1 to CN. These switches 20 and 21 work in opposition. When the first switch 20 is closed, the output converters C1 to CN are powered directly by the electrical network RE. When the second switch 22 is closed, the output converters C1 to CN are powered by the electrical energy previously stored in the power reserve Cres.
The purpose of the capacitor Ctampon is to ensure that a voltage exists at the input of the output converters C1 to CN during transition phases. It is the case, indeed, that the switches 20 and 22 cannot be controlled in a strictly synchronous manner. These two switches 20, 22 therefore remain open at the same time, approximately 1 μs. Capacitor Ctampon, the value of which is much lower than that of the power reserve Cres, enables the output converters C1 to CN to be powered for several milliseconds.
This first solution has the following advantages:                It is simple, and includes two separate converter stages.        It enables a useful compromise to be found between the voltage at the terminals of the power reserve Cres and the volume it occupies.        The establishment time of output voltage Vout1 to VoutN of the output converters C1 to CN does not depend on the charge of the power reserve Cres.        The input converter 21 is dimensioned to provide only the power required to charge the power reserve Cres.
Conversely, this first solution has the following disadvantages:                It is bulky. It comprises, indeed, n+1 converters.        It is impossible to control the current draw at start-up due to the structure of the input converter 21.        The input voltage of the output converters C1 to CN, which accomplish the galvanic isolation, may be greater than the variation of the input voltage Vin of the electrical network RE.        The losses during the switching of switch 22 are substantial; they are proportional to the capacity of capacitor Ctampon and switch 22 is subject to substantial stress during these phases.
In a second solution, a converter is used to obtain a regulated voltage. A widely used topology is illustrated in FIG. 3. In this figure an input converter of the "forward" or "flyback" type 30 has two inputs connected to the electrical network RE and two outputs connected at the input of each of the output converters C1 to CN. The "forward" type converter is also known by the name "convertisseur direct" [direct converter] in French. The "flyback" type converter is also known by the name "convertisseur a accumulation" [accumulation converter] in French.
The output converters C1 to CN are of the same nature as in the previous example. The power reserve Cres is connected between the two outputs of the input converter 30. The input converter 30 enables the voltages at the terminals of the power reserve Cres to be regulated, and may possibly accomplish the galvanic isolation. The power reserve Cres is placed at the output of this input converter 30, requiring a substantial capacitive volume, such that a minimum voltage Vres is guaranteed, at the terminals of the power reserve Cres, during a dropout. During a dropout phase in electrical network RE, the input converter 30 is no longer functional. The output power is supplied by the power reserve Cres, which is discharging. A bypass filtering capacitor Cin is connected to the terminals of the inputs of the input converter 30.
This second solution has the following advantages:                It is simple. Indeed, it comprises a single input converter 30 having an operating range over the input voltage equal to the dynamic range of the electrical network.
Conversely, this second solution has the following disadvantages:                The start-up time is greater than in the first solution, since the power reserve Cres must be charged for the interface to be functional.        The input converter 30 must be dimensioned to guarantee the output power, but also charging of the power reserve Cres over the allowed time.        In a steady state all the power supplied to the energy consumer systems cause losses in all converters 30 and C1 to CN.
In conclusion, there is the same number of converters in both solutions; in the first solution the input converter 21 does not provide galvanic isolation, and in the second solution input the converter 30 can provide galvanic isolation.

Interface device between an electrical network and consumer systems


An exhaust gas sensor system for use with an internal combustion engine having an exhaust conduit and a catalytic converter. The system includes an exhaust gas oxygen sensor, temperature sensor, and signal conditioner. The exhaust gas oxygen sensor is positioned on the conduit, downstream of the catalytic converter, and provides an oxygen level signal. The temperature sensor is also downstream of the catalytic converter, sensing the temperature of the oxygen sensor. A signal conditioner receives outputs from both the exhaust gas oxygen sensor and the temperature sensor. The oxygen level signal from the oxygen sensor is adjusted, according to the temperature sensed by the temperature sensor, to provide a more accurate oxygen level signal to other components of the engine such as, for example, an air-fuel controller.

The present invention relates generally to electronic engine controls and to feedback controls for engine operation using exhaust gas oxygen sensors. More particularly, the present invention relates to a sensor system having an exhaust gas oxygen ("EGO") sensor interconnected to an exhaust system downstream of a catalytic converter.
Many automotive vehicles include an internal combustion engine and an exhaust system that provides a conduit for heated combustion gas to move away from the engine. The temperature of the exhaust gas ranges from ambient temperature, when the engine has not been in operation recently, to 400.degree. Celsius or more.
A typical exhaust system may include an EGO sensor assembly and a catalytic converter. The catalytic converter promotes the conversion of hydrocarbons, carbon monoxide, and oxides of nitrogen into less noxious compounds. An EGO sensor is often placed "upstream" of the catalytic converter. The terms "downstream" and "upstream" are relative terms used to denote relative positions along the exhaust conduit, or pipe, of the vehicle. The term "downstream" refers to positions along the exhaust conduit that are reached by a particle in the exhaust gas later in time than positions that are "upstream."
Many air-fuel control systems in presently available vehicles, with the EGO sensor located upstream of the catalyst, provide an air-fuel feedback signal for a closed-loop air-fuel delivery system in the engine. The upstream EGO sensor, however, can be "poisoned" by certain compounds, such as lead or silicone. Such components may be present in the raw exhaust gas. This may occur, for example, if a motorist improperly uses "leaded" gasoline in an engine designed only for "unleaded" gasoline. Such poisoning may render the EGO sensor ineffective in accurately ascertaining the level of the oxygen concentration in the exhaust gas.
Also, the output characteristics of an upstream EGO sensor may change over time. Moreover, under some operating conditions, the upstream EGO sensor may be unable to bring the exhaust gas flowing nearby it to a substantial equilibrium. Such conditions may be dependent on, for example, the engine load and cylinder-to-cylinder air-fuel maldistribution in the engine. As a result, the EGO sensor will exhibit "offset errors."
Further, many EGO sensors only operate effectively if the temperature of the sensor is within a particular range. The temperature of the sensor is, of course, influenced by the temperature of the adjacent exhaust gas. To assist an EGO sensor to make accurate measurements over a wide range of exhaust gas temperatures, the EGO sensor assembly often includes an electric heater physically adjacent, or near, the EGO sensor. Such a heated exhaust gas oxygen sensor is a type of EGO sensor and is often referred to as a HEGO sensor. When actuated, the heater warms the sensor to enable it make more accurate measurements and, thus, reduce the effect of temperature variations of the exhaust gas passing through the exhaust pipe of the vehicle.
Prior art systems exist for controlling the air-fuel ratio of an internal combustion engine. For example, U.S. Pat. No. 4,708,777, issued to Kuraoka, discloses an air/fuel ratio feedback control system that is responsive to an EGO sensor. The EGO sensor is maintained at a predetermined temperature by feedback from the sensor heater.
Thus, some prior systems have attempted to maintain a constant air-fuel ratio operating point, which is independent of the exhaust gas temperature. In addition to maintaining a constant, closed-loop air-fuel ratio operating point independent of exhaust gas temperature or engine operating conditions, however, it is also desirable to have an EGO sensor that may more accurately detect oxygen levels, regardless of the exhaust gas constituencies and poisoning effects. In this way, the feedback control enables the controller to more precisely regulate the operation of the internal combustion engine.
Further, since the EGO sensor assemblies are generally mass-produced and put on many cars, even a small savings on one part of the assembly can accumulate to a substantial annual savings. Thus, an EGO sensor system should not have an excessive number of parts nor high manufacturing costs. Moreover, it is important that the sensor assembly be reliable.

Oxygen sensor system with signal correction


A method of rapid non-data aided carrier signal acquisition for a low symbol rate carrier signal, comprising receiving and converting an analog intermediate frequency carrier to a digital carrier signal, down-converting the signal to substantially baseband, reducing a sampling rate of the digital carrier signal using a decimation filter, determining a highest Fast Fourier Transform (FFT) based on a result of one or more FFT's generated by an FFT module using a peak finder, selecting an input source for the FFT module using a multiplexer, generating a frequency estimate of the digital carrier signal using a walking coarse detector, tuning a carrier recovery loop (CRL) based on the frequency estimate generated by the walking coarse detector, determining a final carrier frequency offset estimate using a result of the FFT module, modulation removal, and the peak finder, and programming an oscillator within the CRL to the final carrier frequency offset estimate.

1. Technical Field
Aspects of this document relate generally to telecommunication systems and techniques for transmitting data across a telecommunication channel.
2. Background Art
In digital communication receivers employing M-ary PSK or M-ary QAM, carrier frequency estimation is commonly accomplished using Fast Fourier Transform (FFT) together with modulation removal. As the symbol rate becomes low, that is, less than 32 kilosymbols/second, the frequency offset becomes larger relative to the symbol rate. Hence the acquisition range of the receiver becomes larger than the symbol rate as well. When the acquisition range of the receiver is 40 times the symbol rate, for example, frequency offset estimation known in the prior art will take 160 sequential FFT operations together with modulation removal for QPSK.
To reduce the length and complexity of the detailed description and to establish a current state of the art, Applicant hereby incorporates by reference in their entirety each reference listed in the numbered paragraphs below:
U.S. Pat. No. 4,904,930—Method of Carrier Frequency Estimation, David C. Nicholas, Feb. 27, 1990.
U.S. Pat. No. 4,912,422—Demodulation System for PSK Signals with Low Carrier to Noise Ratio and Large Frequency Offset, Kobayahi et al., Mar. 27, 1990.
U.S. Pat. No. 5,272,446—Digitally Implemented Fast Frequency Estimator/Demodulator for Low Bit Rte Maritime and Mobile Data Communications Without the Use of an Acquisition Preamble, Chalmers et al., Dec. 21, 1993.
U.S. Pat. No. 5,233,632—Communication system receiver apparatus and method for fast carrier acquisition, Baum et al., Aug. 3, 1993.
U.S. Pat. No. 7,151,807—Fast acquisition of timing an carrier frequency from received signal, Ernest C. Chen, Dec. 19, 2006.
U.S. Pat. No. 6,771,699—Method of rapid carrier-frequency offset acquisition using a periodic training sequence, Karaoquz, et al., Aug. 3, 2004.
Heinrich Meyr, Marc Moeneclaey, and Stefan A. Fechtel. Digital Communication Receivers: Synchronization, Channel Estimation, and Signal Processing. John Wiley & Sons, Inc. 1998. pages 457-462.
Bernard Sklar. Digital Communications: Fundamentals and Applications. Second Edition. Prentice Hall PTR. 2001. Pages 63-75 (Sampling Theorem) and Pages 598-643 (Synchronization).
Floyd M. Gardner. Phaselock Techniques. John Wiley & Sons, Inc. 1979.
Alain Blanchard. Phase-Locked Loops: Application to Coherent Receiver Design. John Wiley & Sons, Inc. 1976.
Roland E. Best. Phase-Locked Loops: Design, Simulation, and Applications. Fifth Edition. The McGraw-Hill Companies, Inc. 2003.
Ferdinand Classen and Heinrich Meyr. Two Frequency Estimation Schemes Operating Independently of Timing Information. IEEE 1993. Pages 1996-2000.

Low symbol rate rapid carrier acquisition with extremely large frequency offset for digital communication receiver


A reel for a cutting system for harvesting machines has a central carrier body (1) defining a longitudinal axis (2). The reel central carrier body has an outer face (7) between a first axial end A and a second axial end B. Several support elements (20) are arranged along the longitudinal axis (2) on the carrier body (1) distanced from each other and supporting tine carriers (21) arranged on a circle around the longitudinal axis (2). Clamping sleeves (3) have inner faces (6) adapted to the outer face (7) of the carrier body (1). The clamping sleeves enclose the outer face (7) in a circumferential direction and leave an axially extending slot (8) between a first circumferential end (9) and a second circumferential end (10), which ends delimit the slot (8). The support elements (20) are mounted on the clamping sleeves. A clamping mechanism (18) is arranged on each of the clamping sleeves (3) close to the first circumferential end (9) and the second circumferential end (10) of the clamping sleeve (3). The clamping mechanism enables the clamping sleeves (3) to be tightened against the outer face (7) of the carrier body (1).

The reels usually include a multiplicity of support elements. The support elements radially extend to a central carrier body in the form of a tube, and are circumferentially distributed around and connected to the central carrier body. The central carrier body is rotationally accommodated in bearing arms. Depending on the width of the cutting table of a cutting system of a harvesting machine, the reel can have a length of 10 m and more. The object of the reel is to guide the harvested goods to the cutter bar, to pull the cut harvested goods away from the cutter bar and to guide the same to the further devices of the machine for further processing.
A reel for a cutter system of a harvesting machine is, for example, disclosed in U.S. Pat. No. 5,007,235. Support discs are non-rotationally fixed along the longitudinal axis of the tubular central carrier body in a distributed manner. Each support disc is provided with five support arms. The arms are distributedly arranged around the circumference and extend essentially radially to the outside away from the longitudinal axis. The arms are connected to the outer circumference of the discs. The arms on the support discs are aligned in an axial direction parallel to the longitudinal axis and support tine carriers in bearings. These tine carriers are provided, respectively, with tines.
The reel described in U.S. Pat. No. 3,771,299 includes triangular support elements rigidly attached on the tubular carrier body. The elements are distanced from each other along the longitudinal axis.
DE 199 56 431 A1 describes a reel where the central carrier body is constructed of a multitude of tubular segments, formed, respectively, of halves of tube segments. Tine carriers are mounted star-like in the area of the abutment points of the tube segments on the arms. The tine carriers are eccentrically supported to the longitudinal axis.
The rigid welded connection of the support elements to the central carrier body has a disadvantageous effect on the strength of the carrier body. The welded connection causes a structural change in the tubular carrier body, which carrier body is due to weight reasons, thin-walled. The stability of the carrier body is, because of this, negatively influenced. A further disadvantage is that expensive devices need to be provided during the manufacturing process to hold the support elements in the correct position to connect to the carrier body. Furthermore, since the length of the carrier tube has to be accurately adapted to the reel over the complete length of the reel, which is long, tight tolerances need to be maintained.
DD 111 768 describes a reel for harvesting machines with a central carrier body. The carrier body is made depending on the working width of one or a multitude of tube portions. The individual tube portions can be connected to each other by flanges mounted on their ends and by screws. Also, flanges attach the reel star to the tube portions of the central carrier body.


A container device that has an outer shell with an opening that is sealed by a closure mechanism. The closure mechanism can include magnetic strips that are configured to partially or wholly seal the opening.

Containers may be designed to store a user's personal belongings in order to provide a degree of protection from incidental impact (e.g. drops), as well as from liquids and dirt. Containers may be composed of rigid materials such as metal or plastics or flexible materials such as fabric or foams. Containers may be designed with an opening/aperture that allows access to the interior contents of the container. The opening may also be provided with a closure mechanism.


The present invention is directed to compounds having the formula:

It is known that many antimicrobial drugs are becoming less effective against increasingly virulent and drug-resistant organisms. Hence, the benefits of new, more effective antimicrobial agents are well recognized.
Triclosan, (2,4,4′-trichloro-2′-hydroxydiphenyl ether), has been widely used for more than thirty years as an antibacterial agent in numerous consumer products, including toothpastes, mouthwashes, soaps, children's toys, and kitchen equipment. The widespread use of triclosan was predicated on the belief that triclosan acts through a non-specific mechanism involving bacterial membrane disruption. Triclosan was, therefore, assumed not to induce resistant strains of bacteria.
Recent findings, however, show that triclosan does not act non-specifically as originally thought, but rather, specifically, by inhibiting the fatty acid biosynthesis (FAS) pathway in certain bacteria and other organisms. In particular, it has been found that in numerous organisms, including Escherichia coli, Staphylococcus aureus, Bacillus subtilis, and the malarial parasite Plasmodium falciparum, triclosan targets an NADH-dependent trans-2-enoyl-ACP reductase known as FabI. The foregoing organisms are believed to overcome the action of triclosan by mutating the gene that encodes FabI, also known as the fabI gene. Not surprisingly, then, it has also been found that triclosan is not immune to resistance, and that, in fact, bacteria are becoming increasingly triclosan-resistant.
A notable example of anti-bacterial resistance is found in the case of tuberculosis. Tuberculosis is a debilitating disease responsible for the deaths of three million people per year. Isoniazid, also known as isonicotinic acid hydrazide, or INH, is currently the most relied upon drug for the treatment of tuberculosis. Due to the increasing prevalence of resistance in Mycobacterium tuberculosis, current tuberculosis treatment regimens typically include the use of multiple antibiotics over an extended period of time. A typical regimen for treating tuberculosis is the administration of isoniazid, rifampicin, and pyrazinamide in combination with ethanbutol or streptomycin for two months, followed by the administration of isoniazid and rifampicin for four months.
Such multiple drug treatments, even if effective, have significant disadvantages, such as increased risk of side-effects by patients, prolonged treatment time, and high expense. In addition, multiple drug treatment is being severely compromised by the emergence of multi-drug resistant M. tuberculosis (MDR-TB).
As with triclosan, isoniazid targets the fatty acid synthase pathway of M. tuberculosis. In order to be effective, isoniazid requires activation by the mycobacterial catalase-peroxidase enzyme KatG.
At least one mode by which M. tuberculosis resists isoniazid is a genetic mutation of the KatG enzyme. The alteration of the KatG enzyme disables the activation of isoniazid, thereby significantly reducing the efficacy of the drug.
The diazaborines are another class of compounds known to target FabI in E. Coli. The diazaborines form a covalent adduct with the NAD(H) cofactor. Diazaborine-resistant organisms arise from genetic mutations that alter the residues that form the NAD-diazaborine binding pocket, thereby reducing the affinity of the drug for the enzyme.
Accordingly, there is a need for new and improved antimicrobial compounds that are effective against increasingly virulent and drug-resistant organisms, and that overcome the limitations of the drugs described thus far.

Abstract: In the formula, A2 represents N or C—X1a—R1 and A4 represents N or C—X2b—R2, provided that A2 and A4 are not both N. R1 and R2 independently represent H or a hydrocarbon group, provided that R1 and R2 are not both H. The hydrocarbon groups of R1 and R2 are preferably alkyl or alkenyl groups. More preferably, the hydrocarbon groups are alkyl groups. A1, A3, A6, and A8 independently represent CH or N. A5 and A9 independently represent CR3 or N, wherein R3 independently represents H, methyl, ethyl, or halo. A7 represents CR4 or N, wherein R4 represents H, methyl, ethyl, halo, nitro, hydroxy, amino, amido, or a methyl or ethyl group substituted independently with halo, nitro, hydroxy, amino, or amido. Not more than three of A5, A6, A7, A8, and A9 are N.

Research Foundation of State University of New York


An implantable cardiac stimulation device provides bichamber pacing and dynamic bichamber and single chamber sensing. The device includes a sensing circuit that senses activity of a heart, a lead system coupled to a plurality of chambers of the heart, and a cardiac rate circuit that determines a cardiac rate of the heart. A control circuit causes the lead system to couple the sensing circuit to corresponding chambers of the heart to enable bichamber trigger pacing when the cardiac rate is below a given rate and to a single chamber of the heart when the cardiac rate is above the given rate to enable enhanced tachycardia sensing.

Implantable cardiac devices are well known in the art. They may take the form of implantable defibrillators or cardioverters which treat accelerated rhythms of the heart such as fibrillation or implantable pacemakers which maintain the heart rate above a prescribed limit, such as, for example, to treat a bradycardia. Implantable cardiac devices are also known which incorporate both a pacemaker and a defibrillator.
A pacemaker may be considered to be comprised of two major components. One component is a pulse generator which generates the pacing stimulation pulses and includes the electronic circuitry and the power cell or battery. The other component is the lead, or leads, having electrodes which electrically couple the pacemaker to the heart. A lead may provide both unipolar and bipolar pacing and/or sensing electrode configurations. In the unipolar configuration, the pacing stimulation pulses are applied or intrinsic responses are sensed between a single electrode carried by the lead, in electrical contact with the desired heart chamber, and the pulse generator case. The electrode serves as the cathode (negative pole) and the case serves as the anode (positive pole). In the bipolar configuration, the pacing stimulation pulses are applied or intrinsic responses are sensed between a pair of closely spaced electrodes carried by the lead, in electrical contact with the desired heart chamber, with the most proximal electrode serving as the anode and the most distal electrode serving as the cathode.
Pacemakers deliver pacing pulses to the heart to induce a depolarization and a mechanical contraction of that chamber when the patient's own intrinsic rhythm fails. To this end, pacemakers include sensing circuits that sense cardiac activity for the detection of intrinsic cardiac events such as intrinsic atrial events (P waves) and intrinsic ventricular events (R waves). By monitoring such P waves and/or R waves, the pacemaker circuits are able to determine the intrinsic rhythm of the heart and provide stimulation pacing pulses that force atrial and/or ventricular depolarizations at appropriate times in the cardiac cycle when required to help stabilize the electrical rhythm of the heart.
Pacemakers are described as single-chamber or dual-chamber systems. A single-chamber system stimulates and senses in one chamber of the heart (atrium or ventricle). A dual-chamber system stimulates and/or senses in both chambers of the heart (atrium and ventricle). Dual-chamber systems may typically be programmed to operate in either a dual-chamber mode or a single-chamber mode.
Recently, there has been the introduction of pacing systems that stimulate in corresponding chambers of the heart as, for example, the right ventricle (RV) and left ventricle (LV). These are termed biventricular stimulation devices.
Biventricular pacing has been shown to coordinate contractions of the left and right ventricles, reduce the amount of blood flow that leaks through the mitral valve, and decreases the motion of the septal wall that separates the chambers of the heart. Such motion can affect the quantity of blood that the ventricle can pump out in a single beat.
Biventricular pacing has been found to be particularly advantageous in patients suffering from congestive heart disease because of the improved ability of the left ventricle to fully pump blood from the heart. As a result, patients are able to tolerate greater exertion, have a longer life span, and experience a higher quality of life.
Biatrial pacing has also been suggested to also lend in coordinating contractions of the right and left atria. As used herein, the term corresponding chambers is meant to refer to either the combination of the right and left atria or the combination of the right and left ventricles.
One form of biventricular pacing is referred to as Cardiac Resynchronization Therapy (CRT). It has been shown to have a particular positive effect on patients with heart failure (HF). There are a number of ways CRT may be performed. For example, during an atrial tracking mode (DDD), the AV/PV intervals may be set to be short, a negative hysteresis value can be used to ensure pacing, or a trigger mode may be established to provide a pacing stimulation pulse upon a sensed event. For non-tracking modes (DDI, VVI) a high ventricular pacing rate or triggered pacing may be used.
Triggered pacing offers the patient a more physiologic AV/PV delay (as it is their own intrinsic rate) and ensures high percentage of pacing during times of atrial tachycardia (AT) or atrial fibrillation (AF), a critical time for HF patients to receive CRT therapy. However, for triggered pacing to be truly effective in synchronization, the sensing should come from the left and right ventricles. Additionally, sensing from both of these corresponding chambers will ensure that long conduction delays or premature ventricular contractions (PVCs) do not cause extra harm from pacing into a vulnerable period that may come from sensing from one chamber only. For example, a PVC may occur on the left side and be sensed over 100 ms later in the right side to which the device would elicit a biventricular pace pulse into a potential vulnerable period.
Using a combined sensing configuration (left and right ventricle) however poses a problem with accurate ventricular tachycardia and fibrillation detection. For example, if the patient has a very wide QRS complex (which is very likely with an HF patient), then combined right and left sensing may cause the device to double count, thus delivering a more aggressive therapy than may be necessary, i.e., fibrillation therapy instead of antitachycardia pacing (ATP) therapy, or even an inappropriate therapy.
For devices with independent right and left sensing, this may not pose such a problem. But for devices that have limited sense channels, an approach that could encompass right and left sensing while pacing and right sensing while in a tachycardia would be preferred. The present invention addresses this and other issues.

Implantable cardiac stimulation device and method providing dynamic sensing configurations for bichamber stimulation and tachyarrhythmia detection


A method, a system and a computer program product for re-using verification results associated with a circuit design to eliminate a formal verification re-run associated with a subsequent verification of the circuit design. A Verification Data Re-use (VDR) utility initiates the creation of a first netlist data structure and a first set of target signatures for the circuit design. The VDR utility initiates an initial functional verification run of the circuit design and stores the results of the verification run. When a subsequent verification of the initial design is initiated, the VDR utility compares the first set of target signatures with a second set of target signatures for the subsequent verification run. A match of target signatures indicates that corresponding design targets have an identical functionality and the VDR utility re-uses verification results from the initial verification run to eliminate an extensive formal verification re-run for the circuit design.

1. Technical Field
The present invention generally relates to electrical circuit design verification systems and in particular to the application of historical verification data in electrical circuit design verification systems.
2. Description of the Related Art
The increasing complexity of modem hardware designs dictates that extremely rigorous functional verification practices are necessary to flush out the majority of circuit design flaws. These verification practices take on many forms, from the running of extensive simulation "regression buckets" that test the design behavior under specific conditions, to more exhaustive semi-formal or formal verification techniques that are executed to determine complete design correctness. Overall, the verification process requires exponential resources with respect to design size. Formal verification therefore tends to be applicable only to smaller designs, and even then run times easily may consume hours, days, or months. Incomplete verification techniques such as simulation and semi-formal approaches consume lesser computational resources and are, thus, scalable to much larger designs. Nevertheless, in order to obtain as much coverage as possible from these incomplete techniques, these verification techniques are often extensively deployed, collectively requiring hours, days or even months of run-time.
One intrinsic design characteristic that complicates the verification process is that the verification process may have to be performed more than once on a given design. The verification process may be performed multiple times because the design evolves many times during the development phase when new functionality is entered, design flaws are rectified, synthesis optimizations are performed on the design to meet timing/size constraints, and pervasive functions are added such as test logic. Despite how small a change may be, unless the change is demonstrated to be very trivial, it is often necessary to re-perform the verification process anew every time the design changes. Multiple runs of the verification process may be a particular bottleneck in the case of a very late design change as in a change that occurs immediately before design fabrication.

Enhancing formal design verification by reusing previous results

International Business Machines Corporation


A data encoding method includes receiving a sequence of N scrambled blocks produced by scrambling a sequence of N preliminary blocks comprising one of a data block and a control block, the control block being one of K types and including a block-type field, each scrambled block having a block header indicating a scrambled data block or a scrambled control block; encoding the sequence of N scrambled blocks into an encoded block by deleting the block headers, and in the event the sequence contains any scrambled control blocks, deleting a set of scrambled bits corresponding to respective block-type field bits of at least one control block in the sequence of preliminary blocks such that the other block-type field bits are sufficient to indicate the type of control block, and adding position indicator bits indicating position of each scrambled control block in the received sequence of scrambled blocks.

This invention relates generally to encoding and decoding of data transmitted over data communications networks.
Data to be transmitted over a communications network is commonly encoded in order to improve transmission characteristics, e.g., to improve data recovery rates and/or compress data for higher data transfer rates. In high-speed interconnect technologies such as 10 Gb/s, 40 Gb/s and 100 Gb/s Ethernet, InfiniBand, and 10-, 16- and 20-gigabit fiber channel, data is formatted in 64-bit blocks which undergo various stages of encoding and other processing prior to transmission. For example, an encoding stage employs a rate-64/66 modulation code. This code is a block code with the redundancy of two header bits per 64-bit payload. The 64-bit payload of each 66-bit block is either a control block, containing control information for the transmission process, or a data block containing actual data (i.e. user data, CRC (cyclic redundancy check) data and other "non-control" data). The 2-bit header of the 66-bit block indicates whether the attached payload is a data block or a control block.
The particular format of data and control blocks is defined by the appropriate transmission protocol. In general, different types of control block are defined for use in the system, and each control block includes a dedicated field which indicates the type of that control block. This field, referred to herein as a "block-type field", may contain one of a predefined set of K bit-patterns each corresponding to a respective one of K different types of control block used in the transmission system. In 100 Gb/s Ethernet, for example, the block format is defined in Ethernet standard IEEE 802.3ba-2010 and is shown in FIG. 1 of the accompanying drawings. The left-hand column of this table indicates the eight data or control "characters" (represented by letters D, C, O, S or T with suffix 0 to 7 denoting position in the block format) of data and control blocks. The rest of the table indicates the 66-bit output block format for the rate 64/66 modulation code. The 66-bit format for data blocks is shown at the top of the table. This starts with the 2-bit sync header 01 indicating that the 64-bit payload is a data block. There are eleven different types of control block in this instance and the 66-bit format for these is as indicated beneath the data block format. In each case this starts with the 2-bit sync header 10 indicating that the 64-bit payload is a control block. The control block commences with an 8-bit block-type field. This contains a bit-pattern representing one of eleven different values, indicated in hexadecimal in the column headed "Block Type Field", for the eleven different types of control block. The set of eleven 8-bit patterns constitute a rate-4/8 code, with Hamming distance 4, for indicating the eleven different types of control block.
Modulation encoded blocks are typically subject to various additional processing stages, such as scrambling, compression encoding and error correction processing, before transmission. Scrambling is a common process for improving transmission characteristics whereby each bit in an input block is combined with one or more other bits (whose values are generated continuously in a time-dependent manner, e.g., from previously scrambled bits) to produce a corresponding bit in the resulting scrambled block. Compression encoding (often referred to as "transcoding") is performed in order to increase data rates. In the systems described above, for example, rate-64/66 modulation coding is performed in the Physical Coding Sublayer (PCS) of the physical layer of the OSI (Open Systems Interconnection) reference model. Within this PCS layer, modulation encoded blocks are subject to a scrambling process illustrated in FIG. 2 of the accompanying drawings. The sequence of data or control blocks forming the payloads of successive 66-bit encoded blocks provides the input to the scrambler, the 2-bit sync headers being removed before scrambling. The resulting preliminary blocks at the scrambler input have 64 bits, p0, p1, . . . , p63. The input sequence of preliminary blocks is scrambled to produce a corresponding sequence of scrambled blocks, each with 64 bits s0, s1, . . . , s63. The scrambler is a self-synchronizing scrambler defined in the IEEE 802.3ba standard and shown in FIG. 3 of the accompanying drawings. This scrambler, which runs continuously, includes a 58-bit shift register S0 to S57 in a feedback arrangement with two modulo-2 adders (labeled "+" in the figure). After scrambling, the 2-bit sync header is added to each scrambled block to produce the transmit block. The resulting stream of transmit blocks is then further processed in the PCS sublayer before forwarding to the Reed-Solomon (RS) forward error correction (FEC) sublayer RS-FEC. For example, the blocks are distributed over multiple channels, or "lanes", after insertion of alignment markers for block recovery at the receiver.
Transcoding in the above systems is performed in the RS-FEC sublayer. The transcoding process transforms a group of N data or control blocks into a single encoded output block. For instance, N 66-bit blocks generated with the aforementioned rate-64/66 modulation code may be converted into a single (N*64+L)-bit block. Note that if L<2N, this transcoding process always results in compression. Such a transcoding process is described in our copending U.S. patent application Ser. No. 13/765,382, filed Feb. 12, 2013. One embodiment provides a 64b66b to 256b257b transcoding scheme which is used in the RS-FEC sublayer defined by IEEE Ethernet task force 802.3bj for full-duplex 100 Gb/s data transmission. This scheme encodes a sequence of N=4 rate-64/66 modulation coded blocks into a single 257-bit output block. The 2-bit sync headers of the 66-bit input blocks are deleted in the encoding process, along with the second 4-bit nibble of the block-type field of the first control block (if any) in the four-block sequence. Due to redundancy in the block-type field, the remaining block-type field bits are sufficient in themselves to indicate control block type. This redundancy in the block-type field permits recovery of the entire block-type field by mapping the remaining block-type field bits to the missing bits during the corresponding decoding process (referred to herein as "inverse transcoding") at the receiver. A 5-bit header is then added to the sequence. This includes a 4-bit position indicator to indicate the position of any control blocks in the block order of the four-block sequence, and a single bit to indicate whether or not the encoded output block contains any control blocks.
The transcoding schemes described in co-pending U.S. application Ser. No. 13/765,382, filed Feb. 12, 2013, preserve the order of data and control blocks in the input block sequence. Other known transcoding schemes reshuffle (rearrange) the order of the incoming blocks. For example, a 64b66b to 512b514b transcoding scheme is described in "Bit-Error-Tolerant (512*N)B/(513*N+1)B Code for 40 Gb/s and 100 Gb/s Ethernet Transport", Teshima et al., IEEE Infocom Workshops 2008. Control blocks in a sequence of N=8 rate-64/66 modulation coded input blocks are grouped together at the start of the encoded output block, after a 2-bit header. The 2-bit sync headers of the 66-bit input blocks are deleted, and the entire block-type field of each control block is replaced by a new 8-bit field. This includes a 4-bit encoding indicating control block type, and a 3-bit position field indicating the original position of the control block in the 8-block input sequence. The position field allows block order to be restored during inverse transcoding, and the 4-bit encoding permits identification of control block type and hence recovery of the block type field. A related 64b66b to 1024b1027b transcoding scheme is also described. Similar transcoding schemes based on reshuffling control blocks to the start of the encoded block have been described for 64b66b to 256b257b transcoding and 64b66b to 512b513b transcoding.
The transcoding processes described above exploit redundancy in the block-type field of control blocks, permitting deletion in the transcoder of certain block-type field bits which can be regenerated from the encoded blocks at the receiver. In order to do this, however, the input block sequence, which is received in scrambled form from the PCS sublayer, is first descrambled in the RS-FEC sublayer before supply to the transcoder. This results in a sublayer architecture which is shown in FIG. 4A, for the transmitter (TX) side, and in FIG. 4b for receiver (RX) side. (These figures show the architecture for non-return-to-zero signaling, a binary pulse amplitude modulation scheme, but the same basic steps are required in the alternative architecture for other pulse amplitude modulation signaling schemes). At the transmitter, scrambled blocks distributed over lanes from the PCS sublayer are received by the RS-FEC sublayer. The blocks are first aligned and alignment markers removed. The scrambled blocks are then descrambled for supply to the transcoder. After transcoding the blocks must be rescrambled, and alignment markers (mapped from the alignment function at the input) are reinserted. Error correction processing is then performed by an RS FEC encoder, and the resulting output words are distributed over lanes for onward transmission. This structure necessitates a corresponding structure at the receiver as shown in FIG. 4B. In addition to the resulting complex sub-layer structure, the use of a self-synchronizing scrambler as described above requires use of such a scrambler following inverse transcoding in the receiver. This is undesirable due to the problem of error multiplication: a single bit error at the input of the RX self-synchronizing scrambler multiplies to infinitely many errors. Although the descrambler in the RX PCS sublayer regenerates the possibly-erroneous original sequence at the RX scrambler input, massive error multiplication within a sublayer is undesirable in an architecture that cleanly separates the sublayers.