A method and device delivers microwave power to an antenna through a coaxial cable utilizing air or other gases as its dielectric core. The cable includes supports made of low-loss materials to keep the inner conductor centered in the cable, and defining spaces therebetween for the air or gas. Channels in the supports allow the air or gas to circulate in the cable. The gas may be chilled or cooled to provide an addition level of heat dissipation. The device enables delivery of large amounts of power to tissue without undue heating of the feed cable or peripheral tissues, and without increasing the diameter of the feeding cable or antenna, keeping the antenna safe for percutaneous use.

Use of energy to ablate, resect or otherwise cause necrosis in diseased tissue has proven beneficial both to human and to animal health. Microwave ablation and hyperthermia are well-established techniques to heat tumors to the point of necrosis. Larger zones of necrosis and shorter treatment times may be realized by applying larger powers to the antenna. Antennas used to deliver energy at microwave frequencies (300 MHz–300 GHz) to tissue typically require a coaxial cable to feed energy to the antenna. A coaxial antenna is an antenna created from a coaxial transmission line—an electromagnetic structure whereby an inner conductor wire, a dielectric core and outer conductor wire share a common axis. Current coaxial antenna designs use a polymer [e.g., polytetrafluoroethylene (PTFE)] as the dielectric core. Small cable and antenna diameters are required to ensure the procedure is minimally-invasive and safe.
Limitations of the above techniques center on the power rating and diameter of the coaxial cable used to feed the antenna, as well as microwave losses inside the coaxial cable dielectric core. An approximately exponential relationship between cable diameter and power rating exists; that is, as cable diameter decreases, the amount of power that cable may handle without failure decreases exponentially. Losses inside the coaxial cable dielectric core cause heat to be generated when large microwave powers are applied. This causes undue heating of the feeding cable, which causes unwanted necrosis of tissue near the feed cable and is undesirable for patient safety. Thus, the antenna input power is limited by the amount of power the feeding cable may handle without failure and by peripheral heating caused by the feed cable. This, in turn, limits the size of the zone of necrosis obtained in a given time. For this reason, current microwave ablation and hyperthermia antennas are limited in their ability to be operated at high powers and still be safe for percutaneous use.
Therefore, there is a need for a method and device for the delivery of microwave power to tissue which overcomes the above identified disadvantages and limitations of, and which represents an improvement over current coaxial antenna designs. The present disclosure fulfills this need.


The invention provides platinum-based probe compounds having the structure: ##STR1## wherein: Pt is a platinum atom, PROBE is a probe biomolecule for associating to a target biomolecule, M is a detectable marker moiety, and X and Y are stabilizing substituents. Also provided are platinum-based labeling compounds having the structure: ##STR2## wherein: Pt is a platinum atom, M is a detectable marker moiety, A is a displaceable leaving group, and X and Y are stabilizing substituents. The invention further provides platinum-based linker compounds having the structure: ##STR3## wherein: Pt is a platinum atom, A and B are the same or different reactive moieties, and X and Y are stabilizing substituents. Other Pt.sup.II and Pt.sup.IV compounds are also provided. Moreover, the invention provides methods for the preparation and use of these compounds, as well as diagnostic kits which contain the compounds.

The invention generally relates to platinum compounds, which may be used for detectably labeling biological target molecules. More particularly, the invention relates to platinum-based linker compounds, labeling compounds, and detectably labeled molecular probes and methods of using such compounds for the detection and localization of biological substances of interest.
Platinum (coordination) compounds have been considered interesting molecules for a very long time. For a review of some of these compounds and their uses we refer to Reedijk et al., J. Struct. Bonding, 67: 53-72 (1987). This article describes the anti-tumor compound cis-Pt(NH.sub.3).sub.2 Cl.sub.2, which compound has a high affinity for certain biological or bio-organic molecules, including amongst others, proteins and DNA molecules. In particular, it appears that such compounds have a marked affinity for the N.sup.7 -nitrogen atom in the purine bases guanine and adenine, as well as for sulfur groups in macromolecules.
By dissociation of the two chlorine ligands in these compounds, two reactive sites arise, with which such platinum compounds can cross-link within or between molecules. For example, such compounds are known to cross-link between two neighboring guanine and/or adenine bases in the same or opposite DNA strands, thereby inhibiting the replication of the DNA molecules. The application of cis-platinum (cisplatin) as an anti-tumor drug (cytostaticum) is based on this mechanism.
Other applications of the cross-linking capacity of such compounds have been explored. For example, U.S. Pat. No. 4,490,543 discloses platinum-based radiation sensitizers which are said to be derivatives of cisplatin. These sensitizers contain two reactive groups and, accordingly, cross-link biological molecules such as DNA.
In contrast, monochlorinated platinum compounds like Pt(dien)Cl appear to keep their DNA affinity, but they do not form cross-links and interfere only slightly with the base pairing of complementary DNA strands. As such, these compounds have no anti-tumor activity.
According to U.S. Pat. No. 4,711,955, non-radioactive nucleic acid labeling techniques are preferred in present medical-biological practice, especially diagnostic practice. The presently applied known non-radioactive labeling techniques for DNA and RNA are globally to be divided in two categories:
1. Labeling which proceeds via enzymatic or organic synthetic routes; for instance biotin, bromodeoxyuridine (BrdU), digoxigenin, fluorescein and peroxidase; PA1 2. Labeling by direct chemical coupling, like photobiotin, AAF, mercury, and sulfone groups. The '955 document discloses one such labeling method, but does not provide any information concerning use of platinum-based compounds for this purpose. PA1 PROBE is a probe biomolecule for associating to a target biomolecule, PA1 M is a detectable marker moiety, and PA1 X and Y are stabilizing substituents. PA1 M is a detectable marker moiety, PA1 A is a displaceable leaving group, and PA1 X and Y are stabilizing substituents. PA1 PROBE is a biomolecule as defined previously for associating to a target biomolecule, PA1 SUBSTRATE is a substantially solid material, and PA1 X and Y are stabilizing substituents as defined previously. PA1 A and B are the same or different reactive moieties, and PA1 X and Y are stabilizing substituents. PA1 providing to a testable system a platinum-based detectably labeled probe as described above, and PA1 detecting a target biomolecule by determining the extent of association or binding between the detectably labeled probe and the target biomolecule. In this method, the detectably labeled probe binds at least selectively or preferentially to the target biomolecule. In preferred embodiments, however, the detectably labeled probe binds specifically, more preferably uniquely, to the target biomolecule. A highly preferred embodiment involves the induction and detection of in situ hybridization between a detectably labeled nucleic acid of the invention and a target nucleic acid. PA1 contacting a sample, which contains a target biomolecule, with a platinum-based binding substance as described elsewhere herein. In this embodiment, the platinum-based binding substance, which is linked to a substantially solid substrate or surface, possesses a probe molecule associates with or binds to a target biomolecule. Thus, the invention provides, in a preferred situation, a method for specifically binding a target molecule in a quantitative or semi-quantitative assay such as an ELISA. Alternatively, the invention provides a method for separating a target molecule from a sample, such as for purification of the target substance. PA1 PROBE is a probe biomolecule for associating to a target biomolecule, PA1 M is a detectable marker moiety, and PA1 X and Y are stabilizing substituents; and PA1 A and B are the same or different reactive moieties, and PA1 X and Y are stabilizing substituents; and PA1 A is a reactive moiety, PA1 M is a detectable marker moiety, and PA1 X and Y are stabilizing substituents; and PA1 modifying a starting material, which includes a compound of the structure: ##STR11## wherein C represents an electronegative reactive moiety, to produce a platinum-based linker compound of the structure: ##STR12## wherein X represents any stabilizing bridge and wherein A and B represent the same or different reactive moieties. Preferably, X represents a polyamine, more preferably, an aliphatic diamine, still more preferably, an aliphatic diamine having 2-6 carbon atoms, and most preferably, an ethylenediamine group. The reactive moieties A and B are preferably the same. Moreover, A and B are preferably selected from among halogens, NO.sub.3.sup.--, and SO.sub.3.sup.--. Preferably the C moiety is selected from among halogens, NO.sub.3.sup.--, and SO.sub.3.sup.--. PA1 reacting a starting material of the structure: ##STR13## wherein C represents a halogen, with ethylenediamine, and reacting the resulting compound with AgNO.sub.3, to provide a linker comprises a platinum compound of the formula: ##STR14## wherein X represents an ethylenediamine group and wherein A and B represent NO.sub.3.sup.--. PA1 modifying a compound of the structure: ##STR15## wherein X represents any stabilizing bridge and wherein A and B represent the same or different reactive moieties, to produce a platinum-based labeling compound of the structure: ##STR16## wherein M is a detectable marker moiety. PA1 reacting a platinum-based labeling compound of the formula: ##STR17## with a biomolecule, whereby, as a result of said reacting, the A group of the platinum-based labeling compound is replaced by the biomolecule. Preferably, the biomolecule is selected from among proteins, peptides, DNA molecules, and RNA molecules. PA1 a platinum-based linker compound of the formula ##STR18## a detectable marker moiety; and a container. The diagnostic is kit preferably further includes additional detection means such as other marker moieties. PA1 PROBE is a probe biomolecule for associating to a target biomolecule, PA1 M is a detectable marker moiety, and PA1 X and Y are stabilizing substituents. PA1 M is a detectable marker moiety as defined previously, PA1 A is a displaceable leaving group, and PA1 X and Y are stabilizing substituents as defined previously. PA1 PROBE is a biomolecule as defined previously for associating to a target biomolecule, PA1 SUBSTRATE is a substantially solid material, and PA1 X and Y are stabilizing substituents as defined previously. PA1 A and B are the same or different reactive moieties, and PA1 X and Y are stabilizing substituents as defined previously. PA1 1. Direct and almost instantaneous labeling of macromolecules without necessity of enzymatic or organo-synthetic procedures; PA1 2. One-step purification of labeled molecules by means of a simple routine technique; PA1 3. Direct and/or indirect detection of labeled molecules by way of almost all known (microscopic) techniques. PA1 (a) hybridizing a nucleic acid probe labeled with a compound of Formula 1 or 2 with a nucleic acid to be detected; and PA1 (b) detecting the nucleic acid probe. PA1 (a) reacting a compound having the structure: ##STR27## with potassium iodide (KI) in a suitable solvent under suitable conditions so as to form a iodated platinum compound having the structure: ##STR28## (b) reacting the iodated platinum compound obtained in step (a) with ethylenediamine in a suitable solvent so as to form a diethyleneamine iodated platinum compound and represented by the formula Pt(en)I.sub.2 and having the structure: ##STR29## (c) reacting the compound obtained in step (b) with silver nitrate (AgNO.sub.3), the reaction being carried out in a suitable solvent, under suitable conditions so as to form a compound having the structure: ##STR30## (d) reacting the compound obtained in step (c) with potassium chloride (KCl) in a suitable solvent under suitable conditions so as to form a compound having the structure: ##STR31## (e) reacting the compound obtained from step (d) with AgNO.sub.3 in a suitable solvent, under suitable conditions so as to form a compound having the structure: ##STR32## (f) recovering the compound obtained from step (e) as a modified platinum starting compound for the synthesis of marker-bound Pt-containing compounds for use in detectably labeling biological target molecules.
Application of detectable labels to biomolecules has been typically accompanied by a number of problems. Such problems are a result of several factors, including the complexity of previous labeling procedures, limitations on the length of the synthetic oligonucleotides which have been desired to be labeled, the necessity of using health-injuring compounds, and a lack of stability of the label when it has been bound to the nucleic acid.
Given these limitations in the conventional methods of labeling bio-organic target molecules, it is desirable to provide labeling method which overcomes these problems. Probe compounds are desired which can accommodate ready attachment of a large variety of detectable markers. It is also desirable to provide detectably labeled probe compounds which can quickly and easily identify specific bio-organic target molecules such as nucleic acids and proteins. In addition, it would be desirable to provide detectably-labeled probe compounds which would specifically interact with particular bio-organic target molecules, but which do not substantially interfere with their function.
The present invention now provides platinum-containing compounds, in the application of which the above-mentioned disadvantages are effectively removed.

Platinum-containing compounds, methods for their preparation and applications thereof


In a method for diagnosis of a sensor in a motor vehicle having an internal combustion engine, during operation of the motor vehicle, an output signal of the sensor is monitored for whether a maximum value of the output signal undershoots a first threshold value and/or a minimum value of the output signal overshoots a second threshold value. In that case a signal is forwarded to a controller that signals to the controller that the sensor, upon restarting of the motor vehicle, might be furnishing incorrect data.

The present invention relates to a method for diagnosis of a sensor, for instance in a motor vehicle having an internal combustion engine. A sensor of this kind that is to be diagnosed may for instance be a phase sensor on the camshaft of the motor vehicle. Depending on whether a detected variable is above or below a defined threshold value, such a sensor furnishes a corresponding output signal.
Such a sensor is for instance a sensor that reacts to magnetic fields, by which the speed of rotation and/or the position of a transducer wheel that has teeth can be ascertained. A sensor of this kind is constructed and arranged such that the transducer wheel, whose position or rotary speed is to be ascertained, passes between the sensor and a magnet, causing the sensor to record a weak magnetic field when one tooth of the transducer wheel is just then diametrically opposite it, and to record a strong magnetic field when no tooth of the transducer wheel (that is, a gap) is just then diametrically opposite it (or vice versa). Such a sensor can operate inductively or be based on Hall or XMR technology.
FIG. 2 schematically shows one such arrangement with a transducer. In it, R indicates the transducer wheel, G indicates the magnet and the transducer containing the sensor, and W indicates the element on which the transducer wheel R is mounted and whose rotary speed and/or position is to be ascertained; the element W is for instance the crankshaft or the camshaft of an internal combustion engine.
For the sake of completeness it should be noted that the arrangement shown in FIG. 2 is shown highly schematically. In particular, in practice, the transducer wheel R may have more teeth, or differently shaped teeth.
The magnetic field recorded by the sensor is converted in the sensor into a current or a voltage, whose magnitude is directly or indirectly proportional to the magnitude of the magnetic field.
The sensor on which the method for diagnosis according to the invention is based outputs a digital signal. To that end, it compares the electrical variable, into which the recorded magnetic field was converted, with a threshold value, and it outputs a signal at a high level if and as long as the electrical variable is higher than the threshold value, and it outputs a signal at a low level if and as long as the electrical variable is less than the threshold value (or vice versa).
It should be clear and requires no further explanation that such a sensor outputs the output signal expected from it only if the threshold value has been correctly defined.
In practice, however, it is known that the magnitude of the magnetic field recorded by the sensor and the electrical variable into which it is converted is dependent on various factors, such as the temperature, the location of the sensor, the degree of soiling, aging, and so forth so that an originally optimally defined threshold value is suddenly no longer optimal or is even entirely unusable.
For this reason, self-calibrating sensors are often used, which are capable of automatically adapting the threshold value to given conditions. This can be done for instance in that during normal operations the sensors ascertain the range within which the variable to be compared with the threshold value varies, and then change the threshold value in such a way that it is located for instance precisely in the middle of this range.
However, this kind of self-calibration is not always successful. Specifically, it can be done only with the transducer wheel rotating, since only then can the range within which the variable to be compared with the threshold value varies be ascertained.
On the other hand, it is sometimes important, immediately after the sensor and/or the arrangement containing it has been put into operation, that is, while the transducer wheel is still at a standstill, to obtain information about the position or the rotary speed of the element to be monitored.
This is the case for instance if the sensor is used to monitor the position and/or the rotary speed of the camshaft of an internal combustion engine. In that case, it is desirable to obtain information about the position of the camshaft even before the engine is started. This information, or more precisely the information as to whether the sensor is just then diametrically opposite a transducer wheel tooth or a gap, is needed to enable optimal starting of the engine.
Since the sensor cannot calibrate if and as long as the camshaft is at a standstill, however, it cannot be assumed with certainty that the information that the sensor provides about the camshaft position is correct.
Equivalent problems exist in all other sensors whose output signal depends on whether a variable detected is above or below a threshold value.


A system can produce a suitable voltage for powering the memory modules plugged into the memory module slots of a motherboard. A power-good signal is issued when the motherboard is powered up. A power safety device on the motherboard then issues a 2.5V to the memory module slot. If DDR DRAM type of memory modules are not detected after a while, the power safety device will turn off the 2,5V supply and provide a 3.3V, which is suitable for SDRAM type memory modules.This invention avoids sending a 3.3V to DDR DRAM modules, thereby burning the memory chip. The presence of DDR DRAM modules can be detected by a general-purpose purpose input/output port through accessing the recorded data in the EEPROM of the memory module. Alternatively, memory module type can be determined by sending out a low-current pulse signal to the memory module slot. Hence, a suitable voltage source is automatically provided to power the memory modules in the slots.

1. Field of Invention
The present invention relates to a type of motherboard. More particularly, the present invention relates to a motherboard system capable of producing a suitable voltage source to power synchronous dynamic random access memory (SDRAM) and double-data-rate dynamic random access memory (DDR DRAM).
2. Description of Related Art
At present, most personal computers use synchronous dynamic access memory (SDRAM) for program and data storage. Since SDRAM responds to the rising edge of a system clock, operations are triggered by the rising edge of a clock cycle only. Following recent advances in semiconductor technology, another type of memory known as double-data-rate DRAM (DDR DRAM) is introduced into the market. DDR DRAM responds to both the rising and the falling edge of a system clock, and hence the operating speed of memory is almost doubled.
The operating modes of SDRAM and DDR DRAM are different in many aspects, including; (1) SDRAM uses normal clock pulse signals while DDR DRAM uses differential clock signals; (2) SDRAM uses VDD=3.3V while DDR DRAM uses VDD=2.5V and VDDQ=2.5V; (3) SDRAM does not require a reference voltage, but DDR DRAM requires a reference voltage whose value is about xc2xd VDDQ; (4) SDRAM connects to a data bus that operates on CMOS logic while DDR DRAM connects to a data bus that operates on series-stub-terminated logic 2 (STTLxe2x80x942); (5) there is no need for the SDRAM connected data bus connected to use a terminated voltage (VTT), but DDR DRAM connected to data bus must use a terminated voltage (VTT) to absorb reflected waves; and (6) there is no need for the SDRAM connected to data bus to use a pull-up resistor, but the DDR DRAM connected to data bus must use a pull-up resistor. However, a DDR DRAM is able to operate at a speed roughly double that of the SDRAM.
Due to the aforementioned differences between SDRAM and DDR DRAM, particularly to the power requirements, inappropriate provision of power supply can pin to severe problems. For example, if DDR DRAM modules are plugged into the slots on a motherboard, the memory modules may be burnt if the power supply provides a 3.3 V. Alternatively, if SDRAM modules are plugged into the slots on a motherboard, the memory modules may not operate normally if the power supply only provides a 2.5V voltage source.
Accordingly, one object of the present invention is to provide a system on a motherboard capable of automatically providing a 2.5V voltage source to power DDR DRAM modules and a 3.3V voltage source to power SDRAM. Consequently, consumers do not have to worry about burning the memory modules or memory malfunction due to a difference in operating voltages between the two types of memory modules.
To achieve these and other advantages and in accordance with the purpose of the invention, as embodied arid broadly described herein, the invention provides an automatic voltage generation system on a motherboard. The system includes a general-purpose input/output port, a memory module slot and a power safety device. The memory module slot is used for accommodating a memory module. The power safety device is coupled to the general-purpose input/output port and the memory module slot. According to the output from the general-purpose input/output port of a chipset, the power source pin of the power safety device outputs a 3.3V when SDRAM is plugged into the memory module slot. Alternatively, the power source pin of the power safety device outputs a 2.5V when DDR DRAM is plugged into the memory module slot.
The motherboard further includes a central processing unit (CPU) and the memory module further includes an electrical erasable programmable read-only-memory (EEPROM) for recording related data indicating the memory module When the system is boot up, the CPU is able to read the related data recorded in the EEPROM through a chipset and recognize the type of memory module plugged into the slots and control the output from the general-purpose input/output port. Therefore, the power safety device is able to determine whether to supply a 3.3V or a 2.5V to the voltage source pin automatically.
The power safety device further includes a first flip-flop, a second flip-flop, a first field effect transistor and a second field effect transistor. The first flip-flop has a first input terminal, a second input terminal and a positive phase output terminal. The first input terminal of the first flip-flop receives output from the general-purpose input/output port and the second input terminal of the first flip-flop receives signal from a power-good signal supplier The second flip-flop has a first input terminal, a second input terminal and a negative phase output terminal. The first input terminal of the second flip-flop receives output from the general-purpose input/output port and the second input terminal of the second flip-flop receives signal from a power-good signal supplier. The first field effect transistor has a gate electrode, a first source/drain electrode and a second source/drain electrode. The gate electrode of the first field effect transistor is coupled to the positive phase output terminal of the first flip-flop and a +12V voltage source via a first resistor. The first source/drain electrode of the first field effect transistor is coupled to a 3.3V voltage source and the second source/drain is electrode of the first field effect transistor is coupled to the voltage source pin. The second field effect transistor has a gate electrode, a first source/drain electrode and a second source/drain electrode, The gate electrode of the second field effect transistor is coupled to the negative phase output terminal of the second flip-flop and a +12V voltage source via a second resistor. The first source/drain electrode of the second field effect transistor is coupled to a 2.5V voltage source and the second source/drain electrode of the second field effect transistor is coupled to the voltage source pin.
On starting the computer system, the power-good signal causes the second source/drain electrode of the second field effect transistor to output 2.5V to the voltage source pin. If the memory module slot contains a SDRAM module, output from the general-purpose input/output port causes the second source/drain electrode of the first field effect transistor to output 3.3V to the voltage source pin. On the other hand, if the memory module slot contains a DDR DRAM module, the 2.5V output to the voltage source pin is maintained. In brief, the power safety device is able to output 3.3V or 2.5V to the voltage source pin automatically according to the type of memory module plugged into the memory slot.
The power safety device can also provide a low-current pulse to the reference voltage source pin of the memory module slot. By checking if any fluctuation in the low-current pulse exceeds a pre-defined range, a current one of SDRAM or DDR DRAM being plugged into the memory module slot can be determined. According to the type of memory modules plugged into the slots, a 3.3V or a 2.5V is automatically put on the voltage source pin.
It is to be understood that both the foregoing general description and the following detailed description are exemplary, and are intended to provide further explanation of the invention as claimed.

System for automatic generation of suitable voltage source on motherboard


The present invention provides for a method and system for determining the concentration of a surfactant in a transparent hydrocarbon by performing the steps of adding a solid to the hydrocarbon; adding a second masking reagent to the hydrocarbon; mixing to form a dispersion; separating the dispersion from the mixture; and measuring the optical absorption of the dispersion.

Hydrocarbons are susceptible to chemical reactions due to aging, heat and oxidation. One effect of oxidation, for example, is to produce soluble and insoluble materials of higher molecular weight and boiling point than the original hydrocarbon. These materials can have deleterious effects on the ultimate use of the hydrocarbons.
In particular, turbine combustion fuel oils, such as JP-4, JP-5, JP-7, JP-8, Jet A, Jet A-1, and Jet B are subject to harsh conditions. Fuel burned in an aircraft engine is often used as a cooling medium or heat sink to cool aircraft subsystems and engine lubricating systems prior to its combustion. As the fuel passes through the heat exchangers, it is subjected to temperature increases of several hundred degrees in a matter of seconds. Any unstable constituents in the fuel will quickly react under these severe conditions forming gums, varnishes and coke deposits. These deposits will plug-up the components leading to operational problems including reduced thrust and performance anomalies.
Treatment of these problems in hydrocarbons has often been accomplished with additives such as antioxidants, metal deactivators and corrosion inhibitors. Surfactants in the form of dispersants are often added to assist in retarding fouling of the hydrocarbons. However, hydrocarbons contain a number of naturally occurring dispersants as well as those of the additive variety. Consequently, this makes analytical determination of any one dispersant present in the hydrocarbon difficult due to contributions from the other dispersants present. The present inventors have surprisingly discovered an analytical method for determining the concentration of selected dispersants in transparent hydrocarbon fuel that avoids interferences from other dispersants present in the hydrocarbon.
Analytical tests based on turbidity measurement are often encountered in clinical chemistry and molecular biology laboratories. In these tests, turbidity measurement is used as a means of detecting and quantifying the concentration of a species which is made to precipitate from solution or absorb onto dispersed latex particles. A related analytical method utilizes turbidity measurements to detect suspended solids. Nephelometry is a photometric analytical technique for measuring the light scattered by finely divided particles of a substance in suspension. These techniques are used in industries as diverse as brewing and smart appliance manufacture (e.g., washing machines). The focus application of the turbidity measurement in these examples is to quantitate the solids present in the fluid, not to quantitate the concentration of dispersant in the fluid. However, none of these techniques account for methods to counteract the interference of other dispersants in the tested fluids.

Methods for determining the concentration of surfactants in hydrocarbons


A data aggregation method and data aggregation apparatus are provided. More particularly the data aggregation method includes: generating a plurality of data aggregation trees according to connection relationships between clusters that constitute a sensor network using a geographical code (GGC), selecting a single activation zone for each of the clusters which configures each of the plurality of data aggregation trees, selecting a maximum energy tree having a maximum total residual energy from the plurality of data aggregation trees, based on the single activation zone, and generating a tree list including activation zone information which corresponds to the maximum energy tree, and information about the maximum energy tree.

1. Field of the Invention
The present invention relates to a wireless sensor network of sensor nodes and zones. More particularly, the present invention relates to a data aggregation apparatus and method using the apparatus which can easily recognize location information of all sensor nodes, and which can apply the location information to expand a network by aggregating data using zone scheduling on a wireless sensor network.
2. Description of Related Art
A wireless sensor network refers to a network (e.g., a wireless mesh network) including tens to several thousands of sensor nodes capable of communicating with each other using a radio frequency (RF) to sense physical space, the network being spread over a wide area. Due to development of a sensor technique, technology for a microelectromechanical system (MEMS) can be connected via a wireless network and is embracing low power electronics engineering technology, low power RF planning technology, and the like. As a result, miniature sensor nodes are being developed that operate at low power levels and have low costs.
A large number of sensor nodes are formed within the same domain or a similar domain of interest to form a wireless sensor network, and may be applied to various applications such as health management, military, home networking, environmental monitoring, factory management, and disaster management. The locations of the sensor nodes are not required to be predetermined in the wireless sensor network, therefore the sensor nodes may be arbitrarily located and may be applied to areas difficult to access, such as in disaster rescue areas.
When a security network is configured in a wireless sensor network, a sensor surveilling burglar's movement and a sensor sensing open/close status of an entrance door are used as the sensor nodes, and the sensor nodes sense the burglar's movement or the open/close status of the entrance door, and the sensor nodes wirelessly transmit the burglar's movement or the open/close status of the entrance door to a control apparatus.
Protocols in the sensor network have an ability of self-organizing, and sensor nodes operate by cooperating with each other.
In the wireless sensor network, a large number of sensor nodes may be located in the region of interest without a predetermined arrangement. The wireless sensor network has an advantage in that overall operation of a network is not affected when a predetermined sensor node fails and no longer functions since neighboring sensor nodes sense similar information.
However, the wireless sensor network has drawbacks of low speed wireless media transmission, frequent transmission errors, a limited power supply, and difficulty in replacing the sensor nodes due to arbitrary distribution of the sensor nodes. Since a lifetime of the wireless sensor network is determined by a lifetime of a battery of the sensor nodes, an operation lifetime is required to be extended as long as possible to extend the lifetime of the the wireless sensor network. Thus, protocols for the wireless sensor network are needed to improve the lifetime of an entire system and to cope with dynamic changes of the wireless sensor network by distributing energy consumption within the entire network.
Accordingly, there is a need for an apparatus and method which can effectively improve the lifetime of the wireless sensor network.

Apparatus for data aggregation using zone scheduling in wireless sensor network and method thereof

Korea University Industrial & Academic Collaboration Foundation


The invention includes a method for placement of sidelobe inhibitors on a radiation-patterning tool. Elements of the tool are represented by design features in a modeling domain. The modeling domain is utilized to generate vectors spanning between edges of design features within a threshold spatial distance of one another. Locations of vector midpoints are identified. The locations are utilized in identifying areas where sidelobe overlap is likely. The areas are shifted from the modeling domain to a real domain, and in the real domain correspond to regions of the tool where sidelobe overlap is likely to occur. Sidelobe inhibitors are formed across at least some of the identified regions of the tool. The invention includes computer readable code that can enable a computer to determine locations for placement of the sidelobe inhibitors.

Photolithography is commonly used during formation of integrated circuits on semiconductor wafers. More specifically, a form of radiant energy (such as, for example, ultraviolet light) is passed through a radiation-patterning tool and onto a radiation-sensitive material (such as, for example, photoresist) associated with a semiconductor wafer. The radiation-patterning tool can be referred to as a photomask or a reticle. The term "photomask" traditionally is understood to refer to masks which define a pattern for an entirety of a wafer, and the term "reticle" is traditionally understood to refer to a patterning tool which defines a pattern for only a portion of a wafer. However, the terms "photomask" (or more generally "mask") and "reticle" are frequently used interchangeably in modern parlance, so that either term can refer to a radiation-patterning tool that encompasses either a portion or an entirety of a wafer. For purposes of interpreting this disclosure and the claims that follow, the terms "reticle" and "photomask" are utilized with their traditional meanings.
Radiation-patterning tools contain light restrictive regions (for example, totally opaque or attenuated/half-toned regions) and light transmissive regions (for example, totally transparent regions) formed in a desired pattern. A grating pattern, for example, can be used to define parallel-spaced conductive lines on a semiconductor wafer. As discussed previously, the wafer is provided with a layer of radiation-sensitive material (such as, for example, photosensitive resist material, which is commonly referred to as photoresist). Radiation passes through the radiation-patterning tool onto the layer of photoresist and transfers a pattern defined by the radiation-patterning tool onto the photoresist. The photoresist is then developed to remove either the exposed portions of photoresist for a positive photoresist or the unexposed portions of the photoresist for a negative photoresist. The remaining patterned photoresist can then be used as a mask on the wafer during a subsequent semiconductor fabrication step, such as, for example, ion implantation or etching relative to materials on the wafer proximate the photoresist.
Advances in semiconductor integrated circuit performance have typically been accompanied by a simultaneous decrease in integrated circuit device dimensions and a decrease in the dimensions of conductor elements which connect those integrated circuit devices. The demand for ever smaller integrated circuit devices brings with it demands for ever-decreasing dimensions of structural elements, and ever-increasing requirements for precision and accuracy in radiation patterning.
FIG. 1 shows a flow chart illustrating a typical process utilized for creating a pattern for a radiation-patterning tool. At an initial step 10, a preliminary design is created for the radiation-patterning tool and verified. The creation of the design begins with provision of a desired pattern which is ultimately to be formed in photoresist. Subsequently, elements are developed for the radiation-patterning tool to roughly produce the desired pattern on photoresist from radiation passed through the radiation-patterning tool. The elements form a rough correspondence to the desired pattern in that the first approximation of the elements largely ignores effects of interference on radiation passing through the radiation-patterning tool.
After the design is believed to be complete, (i.e., once it is believed that all patterned features which are to be patterned in photoresist with the radiation-patterning tool are represented by elements in the design) the design is submitted to a verification process to confirm that the design is complete.
After the design has been created and verified, it is subjected to optical proximity correction (shown as step 20 in FIG. 1). The optical proximity correction takes into account various interference factors that influence radiation passing through a radiation-patterning tool (i.e., constructive and destructive interference effects that result from passing radiation through patterns having dimensions on the same order as the wavelength of the radiation, or smaller). The optical proximity correction can be utilized to correct all parts of the design, or only some parts of the design. In other words, the optical proximity correction can be applied to only some portions of a design, while other portions are not optical proximity corrected. Typically there will be a verification step following the optical proximity correction.
The steps of generating a design from a desired pattern which is to be provided in photoresist, verification of the design, optical proximity correction, and verification of the correction, are typically accomplished primarily through the use of software. A suitable software package which can be utilized for one or more of the steps is HERCULES™/TAURUS OPC™, which is available from Synopsys Corporation™.
The optical proximity correction creates a dataset which is subsequently translated into a pattern formed on a radiation-patterning tool. The process of translating the dataset into a pattern on the radiation-patterning tool is frequently referred to as taping the pattern onto the radiation-patterning tool. In such context, the terms "tape" and "tape out" refer to a process of transferring the dataset to appropriate hardware which writes a pattern represented by the dataset onto the radiation-patterning tool. The process of writing onto the radiation-patterning tool can be accomplished by, for example, laser writing and/or electron-beam writing methodologies. The step of taping the pattern onto the radiation-patterning tool is shown in FIG. 1 as step 30.
After the pattern has been formed on the radiation-patterning tool, the tool can be utilized for patterning radiation in semiconductor fabrication processes. FIG. 2 illustrates an exemplary apparatus 40 in which a radiation-patterning tool is utilized for patterning radiation. Apparatus 40 comprises a lamp 42 which generates radiation 44. Apparatus 40 further comprises a radiation-patterning tool 46 through which radiation 44 is passed. A semiconductor substrate 48 having a radiation-sensitive material 50 thereover is illustrated associated with apparatus 40. The radiation passing through radiation-patterning tool 46 impacts radiation-sensitive material 50 to form a pattern within the radiation-sensitive material. The process of forming a pattern in a radiation-sensitive material with a radiation-patterning tool can be referred to as a printing operation.
Radiation-patterning tool 46 typically comprises an opaque material (such as chrome) over a transparent material (such as a glass). Radiation-patterning tool 46 has a front side where the pattern is formed as features (or windows) extending through the opaque material, and has a back side in opposing relation to the front side. The shown radiation-patterning tool has two opposing sides 45 and 47, and in practice one of the two sides would be the front side (typically side 45) and the other would be the back side. In some applications features can be printed on both the front side and back side of the radiation-patterning tool.
As discussed above, radiation-patterning tool 46 will typically have a pattern with dimensions on the order of the wavelength of the radiation passing through the radiation-patterning tool, or smaller. Accordingly, various interference effects can occur as the radiation passes through the radiation-patterning tool so that the radiation exiting the radiation-patterning tool will transfer a pattern somewhat different than the pattern of the radiation-patterning tool. Such is illustrated diagrammatically in FIG. 3. Specifically, FIG. 3 illustrates an exemplary pattern 60, which can be desired to be formed in a radiation-sensitive material, and illustrates an approximation of a pattern 70 which would be formed in a radiation-patterning tool to generate the pattern 60. Pattern 70 is referred to as an approximation because the pattern is a qualitative representation of the type of pattern utilized in the radiation-patterning tool for generating pattern 60, rather than a quantitative representation.
The FIG. 1 process can, for example, start with a pattern identical to pattern 60 being provided at the design step (10) of the radiation-patterning tool fabrication process, and such design would then be converted to the shape 70 during the optical proximity correction (20) step.
FIG. 4 illustrates an exemplary design which can be desired to be formed in a radiation-sensitive material 80, and illustrates elements in a radiation-patterning tool 84 utilized to create such design.
Radiation-sensitive material 80 is illustrated in top view, and the design formed within the material comprises a plurality of repeating units 82. The shown units 82 are circular in patterned dimension, and can be utilized, for example, in forming contact openings. One of the shown units is centered around a location 83, and another of the units is centered around a location 85.
Radiation-patterning tool 84 comprises a plurality of repeating elements 86. The elements 86 are in a one-to-one correspondence with the units 82 formed in the radiation-sensitive material. Further, each of elements 86 is approximately square in shape. In operation, radiation is passed through radiation-patterning tool 84 to form the pattern of printed images 82 on radiation-sensitive material 80. Regions 86 of the radiation-patterning tool can be either more transparent to radiation than surrounding regions of the radiation-patterning tool, or can be less transparent, depending on whether the radiation-sensitive material corresponds to a positive or negative material. If elements 86 are more transmissive to radiation than surrounding regions, the elements 86 can effectively be windows which allow radiation to pass through those specific regions of the radiation-patterning tool.
In particular aspects of the prior art, printed images 82 correspond to regions where light has passed through windows 86 of radiation-patterning tool 84. If windows 86 have dimensions on the order of the wavelength of the light passing through the windows, there can be sidelobes of energy 88 extending around each of features 82. The sidelobes are commonly referred to as sombreros. The energy within the sombreros is generally too low to form a printed feature within the radiation-sensitive material 80. However, if two or more sidelobes converge near one another, they can form a printed feature at the location of the convergence. Regions 90 and 92 illustrate locations where four sidelobes converge near one another, and accordingly illustrate locations where printed features can undesirably occur due to the convergence of the sidelobes.
Sidelobe convergence is described in additional detail with reference to FIGS. 5-7. Referring initially to FIG. 5, electric field strength of light passing through a window 86 to form a feature 82 centered around location 83 is illustrated (the figure is for diagrammatic purposes only, and not quantitative). A large positive field strength occurs at location 83, and negative sidelobes occur at locations 88. The large positive field strength can be referred to as a primary lobe.
The effect induced by radiation impacting radiation-sensitive material 80 is actually proportional to the intensity of the radiation, rather than the field strength, and the intensity is a function of the square of the field strength. FIG. 6 is a diagrammatic graph of the intensity of the radiation utilized to pattern the feature centered around location 83. Since the intensity is the square of the field strength values, the sidelobes have a positive value, as does the primary lobe at location 83. Accordingly, if the magnitude of sidelobes 88 is sufficient, the sidelobes can generate printed features in the radiation-sensitive material 80. However, the magnitude of the sidelobes generated from light passing through a single window is typically too small to form a printed feature.
FIG. 7 diagrammatically illustrates the additive effect of a sidelobe formed from radiation centered around location 83, and a sidelobe formed from radiation center on location 85. The radiation formed around location 85 is identical to that formed around location 83, and accordingly a main lobe of radiation occurs at location 85 which is about identical in magnitude to the main lobe occurring around location 83. However, since a sidelobe from the radiation around location 85 overlaps a sidelobe from around location 83, the two sidelobes combined to form a lobe 94 having increased intensity relative to the main lobes at locations 83 and 85. Lobe 94 results from combination of sidelobes from two adjacent patterns of radiation, and the concept illustrated in FIG. 7 can be extended to combinations of three or more proximate patterns of radiation. Accordingly, a lobe formed from the combination of the sidelobes can grow significantly in intensity relative to the main lobes occurring at the centered regions of the patterns, and eventually the lobe formed from the combined sidelobes can have sufficient intensity to generate a printed feature.
Various techniques have been developed for identifying locations where sidelobes may combine to form a printed feature. Once such locations are identified, a radiation-patterning tool can be modified to prevent the undesired combination of sidelobes. For instance, FIG. 8 illustrates a portion of a mathematical construct 98 utilized to form radiation-patterning tool 84 (FIG. 4). Construct 98 comprises comprising four design features 101 which correspond to the elements 86 of radiation-patterning tool 84, and illustrates a prior art method for identifying a location where sidelobes may combine during utilization of the radiation-patterning tool. Construct 98 may correspond to a mathematical model formed at step 10 of the FIG. 1 process.
A calculation is performed on the spatial characteristics of design features 101, and such calculation can be considered to create a polygon 100 extending between the features 101. It is to be understood that the calculation is occurring in a mathematical domain during development of a pattern for a radiation-patterning tool, and accordingly the polygon 100 is not a real feature. The shown polygon is a rectangle extending from vertices of adjacent elements. The rectangle is subsequently utilized to determine the location of a sidelobe inhibitor 102. The location is initially determined as part of the mathematical model, but eventually the location is shifted to the real domain and an actual sidelobe inhibitor is formed at a real location of a radiation-patterning tool corresponding to the location 102 of the mathematical domain. The sidelobe inhibitor is utilized to prevent formation of an undesired printed feature from occurring at the location where sidelobes from radiation passing through the elements 86 (FIG. 4) converge.
FIG. 9 shows a portion of a radiation-patterning tool 84 identical to the tool described with reference to FIG. 4, except that a sidelobe inhibitor 103 is formed to prevent sidelobes of radiation passing through windows 86 from combining. Inhibitor 103 has length and width dimensions of "X" and "Y". Such dimensions will typically be about ½ of the wavelength of radiation passed through tool 84 to form the pattern in the radiation-sensitive material. Inhibitor 103 can be formed by etching into an opaque material associated with tool 84 to form a region where radiation will be in phase with the main lobe of FIG. 5, and thus out of phase relative to other portions of sidelobe radiation. Such can cause destructive interference which ultimately cancels a significant amount of intensity from the combined sidelobes. The inhibitor 103 can thus correspond to a phasing region.
FIG. 10 is a color representation of a construct 97 showing actual results obtained from a prior art routine utilized to calculate placement locations for sidelobe inhibitors. Construct 97 corresponds to a mathematical model describing locations for placements of elements (shown as blue squares) and sidewall inhibitors (red blocks). Construct 97 also contains the polygons generated by the sidelobe placement routine (brown lines). The calculations utilized for determining the location of the sidelobe inhibitors would typically occur between steps 10 and 20 of the FIG. 1 process. In other words, the calculations would occur after a rough mathematical model is created for placement of elements in a radiation-patterning tool, and before optical proximity correction. The elements are actually represented as spatial information (also referred to herein as design features) in the mathematical model.
The placement routine described with reference to FIG. 10 will occasionally misidentify a location where sidelobe interference occurs. This can lead to placement of a sidelobe inhibitor at the wrong location. Also, the methodology described with reference to FIG. 10 will occasionally not recognize multiple discrete locations where sidelobe interference will occur if the locations are too close to one another. Such can result in a single sidelobe inhibitor being placed across an average of several locations where sidelobe combinations occur, rather than being placed at each discrete location where the combinations occur. This can result in some locations not having an appropriate sidelobe inhibitor provided.
It would be desirable to develop new methodology for identifying locations of radiation-patterning tools where sidelobe inhibitors are to be provided.

Methods of forming radiation-patterning tools; carrier waves and computer readable media


An inspiration model device according to the invention is provided with a spontaneous emotion unit, a knowledge database, and a conception unit. The spontaneous emotion unit prepares, as data, plural emotion states in advance, which are obtained by modeling human emotions, and causes state transitions to occur repeatedly between the emotion states according a stochastic model of the Schrödinger equation. The knowledge database simulates a human inspiration source that is influenced by sensibility by classifying externally collected knowledge data depending on the degrees of correlation with the emotion states, and accumulates the knowledge data. When receiving an external input, the conception unit simulates human conception by combining the external input with an emotion state of the spontaneous emotion unit and searching the knowledge database for related knowledge data using, as a search key, the combination or the like.

1. Field of the Invention
The present invention relates to an inspiration model device for simulating human conception and a related method and program.
The invention also relates to a spontaneous emotion model device that is the main part of the above inspiration model device, and a related method and program.
In particular, the invention relates to a technique that is applied to the following product fields etc.:
(1) Question and answer systems that use a computer or a robot and are improved in naturalness.
(2) Idea search systems capable of reflecting sensibility as humans possess.
(3) Experimental models of a human psychological state.
(4) Game systems capable of expressing emotions and motions in a more human-like manner.
(5) Agent systems.
(6) Information search methods capable of retrieving an appropriate answer at high speed from an information source having an enormous amount of information utilizing conception (what is called intuition) by a spontaneous emotion.
2. Description of the Related Art
One conventional technique relating to the invention is "Emotion Generation Device and Emotion Generation Method" that is disclosed in Patent document 1 (Japanese Unexamined Patent Application Publication No. 11-265239). The emotion representing the internal state of a human, for example, varies in a variety of manners depending on the situation. Patent document 1 discloses a technique for realizing generation of an emotion in an unpredictable situation. More specifically, an emotion of a device itself is generated by evaluating a situation in the light of a predictable situation. Further, emotions that occurred actually in the past and associated situations are analyzed, whereby unpredictable incidental conditions specific to the respective situations and corresponding emotions are learned. If a newly input situation satisfies a learned incidental condition, the emotion corresponding to that incidental condition is output.
The present inventor disclosed "Sensibility Generation Method, Sensibility Generation Device, and Software" in Patent document 2 (Japanese Unexamined Patent Application Publication No. 2002-215183). In this conventional technique, instinct parameters representing the degrees of comfort, danger, and achievement are generated as motivation information on the basis of input information such as emotion states of a dialog partner, and a fundamental emotion such as joy or anger is generated and controlled on the basis of the generated instinctive motivation information. In particular, this conventional technique succeeded in generating an emotion state that is closer to one human will have by allowing the control rules (tendency in terms of humans) of the generation and control of a fundamental emotion to reflect personalities such as reason and will.
Incidentally, the human brain is considered a big parallel processing machine consisting of a very large number of neurons. As a result, a problem solution by the brain is recognized as a check-everything-basis information search process by parallel processing.
However, realization of such a check-everything-basis information search process by a computer or the like is associated with various problems; for example, an enormous search dictionary needs to be prepared and a very long search time is necessary. Further, a check-everything-basis information search merely finds, from a search range, an average solution that satisfies prescribed search conditions and cannot output an unexpected solution (brainstorm).
The human brain has a rare ability of producing a brainstorm, which is of a different nature than the abilities of the check-everything-basis information search process.

Inspirational model device, spontaneous emotion model device, and related methods and programs


A storage capacitor includes a first capacitor portion and a second capacitor portion, the second capacitor portion being disposed above the first capacitor portion, thereby defining a first direction. The first and the second portions each include a hollow body made of a conductive material, respectively, thereby forming a first capacitor electrode. An upper diameter of each of the hollow bodies is larger than a lower diameter of the hollow body, the diameter being measured perpendicularly with respect to the first direction. The storage capacitor also includes a second capacitor electrode and a dielectric material disposed between the first and the second capacitor electrodes. The storage capacitor also includes an insulating material disposed outside the hollow bodies, and a layer of an insulating material. A lower side of the insulating layer is disposed at a height of an upper side of the first capacitor portion.

Memory cells of a dynamic random access memory (DRAM) generally comprise a storage capacitor for storing an electrical charge which represents information to be stored, and an access transistor which is connected with the storage capacitor. The access transistor comprises a first and a second source/drain region, a channel connecting the first and second source/drain regions as well as a gate electrode controlling an electrical current flow between the first and second source/drain regions. The transistor usually is at least partially formed in a semiconductor substrate. The gate electrode forms part of a word line, and it is electrically isolated from the channel via a gate dielectric. By addressing the access transistor via the corresponding word line, the information stored in the storage capacitor is read out. In addition, by addressing the access transistor and transmitting an information signal via a bit line, data is stored in the corresponding memory cell, which is assigned to the specific word line and bit line.
According to an implementation of the DRAM memory cell, the electrical charge is stored in a stacked capacitor, which is formed above the surface of the substrate.
For future DRAM technologies, an increased cell capacitance for high performance and low power applications is desirable. For example, the cell capacitance can be increased by increasing the height of the stacked capacitor.
A memory device further comprises a support portion which is usually disposed at the edge of the memory cell array. Circuits such as: decoders, sense amplifiers and word line drivers for activating a word line are located in the support portion. Generally, the peripheral portion of a memory device includes circuitry for addressing memory cells and for sensing and processing the signals received from the individual memory cells.
Usually, the peripheral portion is formed in the same semiconductor substrate as the individual memory cells. Hence, a manufacturing process by which the components of the memory cell and the peripheral portion can be formed simultaneously is desirable.

Storage capacitor, a memory device and a method of manufacturing the same


A method for producing a white powder, which includes forming on a surface of a base particle a titanium oxide film and a metallic silver film contiguously one upon another in this order, wherein the base particle includes a magnetic material, and wherein said step of forming on the surface of the base particle the titanium oxide film includes dispersing the base particle in a buffer solution and adding a titanium salt aqueous solution while agitating the dispersion.

In order to use powder in various applications, there is known a technology of covering the powder with another material to add a new function thereto. For instance, in existing one-component color magnetic toner or color magnetic ink, on magnetic base particles having the magnetism such as iron powder, a coloring layer is disposed to form. In order to obtain a clear color image with the one-component color magnetic toner or color magnetic ink, the magnetic toner or magnetic ink itself has to be colored with a brilliant color. However, since particles of a magnetic material that becomes a base material thereof are generally colored black, when a coloring layer is directly disposed on a surface thereof, as a whole, a dark color results.
On the other hand, the inventors have proposed a method of rendering powder white, in which on a surface of a base particle, under application of ultrasonic, a metallic silver film is formed, and, owing to the reflectivity of the film, powder is rendered white (for instance, patent document 1). The powder provided with the above-cited metallic silver film can reflect all incident light to be able to render the powder white. It is suggested that thus obtained white powder can be made into magnetic toner or ink, or when a surface of the white powder is provided with a coloring layer, brilliantly colored color magnetic toner or ink can be produced.

Method for producing white powder comprising titanium oxide film and metallic silver film


An adjustable protecting guard apparatus for a table saw having a guard positionable over the saw blade. The adjustable protection guard apparatus has a telescoping arm extending from one end of the saw table which supports a guard superstructure. A hinged connection on an edge of the guard superstructure permits the guard superstructure and the attached guard to pivot up and away from the saw blade. A control module simultaneously deactivates the motor rotating the saw blade with the pivoting of the guard superstructure. The control module has a multi-position key switch having a by-pass position permitting pivoting of the guard without de-energizing the motor. This permits difficult cuts to be made which would not be feasible with the guard in place. An anti-kickback pawl is also provided within the guard and prevents the saw from rejecting a workpiece in a direction opposite to its feed direction.

I. Field of the Invention
The present invention relates to a saw guard, and more particularly to an adjustable protecting guard for a table saw.
II. Description of the Prior Art
Protective guards for table saw blades are known in the art. The concept of the guard which is pivotably movable away from a blade is known, and is illustrated in U.S. Pat. No. 302,041 issued to Sill. Also known in the art are saw guards with a transparent structure as is shown in U.S. Pat. No. 3,949,636 issued to Ball et al and U.S. Pat. No. 4,625,604 issued to Handler et al.
These prior art however do not show a saw guard which is pivotal relative to a telescoping supporting bracket extending across a saw table and over a workpiece to be cut. Also not shown is a table saw guard with a multi-position key-switch and a bypass which allows the saw to continue operating when the guard is in a retracted and away position so as to permit difficult sawing operations to be undertaken.
The prior art further does not disclose a device for preventing kickback of the workpiece by the saw blade during sawing which is integrally built into a protecting saw guard.

Adjustable protecting guard apparatus for a blade of a table saw


A test circuit and method for a semiconductor memory array such as a dynamic random access memory (DRAM) or static random access memory (SRAM) array that reduces the required testing time. A row of memory cells is concurrently written to a logic level, then read. Any faulty memory cells will discharge both true and complementary data lines through a diode or a diode-connected FET. The resulting voltage on the data line is less than its precharged high logic level, allowing detection of any faulty memory cell in the row of memory cells.

Semiconductor memory manufacturing technology allows the creation of high density memory arrays on an integrated circuit chip. Such memory arrays consist of many memory cells, each cell capable of storing one data bit. Dynamic random access memories (DRAMs) are manufactured with a storage capacity as high as 64 megabits. Static random access memories (SRAMs) are manufactured with a storage capacity as high as 4 megabits. Future technological advances are expected to further increase memory storage capacity.
By limiting the number of data bits in the memory cell array which can be accessed simultaneously (the bit width), the memory cell array density is increased, thereby increasing the storage capacity of the memory cell array. However, a smaller bit width increases the time required to test the memory array. For example, only one bit can be written into or read from a memory cell array having a bit width of one. To read and write both a "0" state and a "1" state, a 16 megabit memory having a single bit width must be accessed 64 million times. It is desirable to reduce the time required to test large memory arrays by reducing the number of times the array is accessed.


A level shifting circuit includes a first stage and a second stage. The first stage and second stage are operatively coupled to a first and second power supply. The first stage translates a differential input voltage into an intermediate differential voltage. The second stage translates the intermediate differential voltage into a differential output voltage and provides feedback to the first stage in response to translating the intermediate differential voltage. The first stage reduces current flow between the first and second power supply through the second stage in response to the feedback.

Many modern systems include multiple circuits that interact with each other. Some circuits operate at a low voltage level in order to reduce power consumption. However, many systems also have circuits that operate at a higher voltage level. Level shifting circuits are generally interposed between circuits operating at different voltage levels in order to facilitate communication. For example, a leveling shifting circuit can be employed for transferring a signal generated by a circuit having a low voltage level to a circuit having a higher voltage level and vice versa.
Conventional level shifting circuits tend to leak current between reference voltages. This leakage current increases power consumption of the circuit, which is undesirable.
It is therefore desirable, among other things, to provide a level shifting circuit that reduces leakage current between references voltages and thus consumes less power than conventional level shifting circuits.

Reduced leakage voltage level shifting circuit


A system for recirculating a hot gas flowing through a gas turbine generally includes a transition piece having a downstream surface, a stationary nozzle having a leading edge surface adjacent to the transition piece downstream surface, a gap defined between the transition piece downstream surface and the stationary nozzle leading edge surface, and a projection having an inner arcuate surface radially separated from an outer surface. The projection generally extends from the stationary nozzle leading edge surface towards the transition piece downstream surface and at least partially decreases the gap. A seal extends across the gap and may be in contact with the transition piece and the stationary nozzle leading edge surface. A recirculation zone may be at least partially defined between the projection inner arcuate surface, the stationary nozzle leading edge surface and the transition piece downstream surface.

Industrial gas turbines generally include a plurality of combustors disposed in an annular array about an axial centerline of the gas turbine. Hot gases of combustion flow from each combustor through a transition piece and across a first-stage of stationary nozzles. Because the transition piece and the stationary nozzles are formed of different materials and are subjected to different temperatures during operation, each may experience different degrees of thermal growth as the gas turbine cycles through various operating modes. As a result, the transition piece and the first-stage of stationary nozzles may move radially, circumferentially and axially relative to one another. Also, similar relative movement may occur as a result of dynamic pulsing of the combustion process.
Typically, thermal growth variations are addressed by providing a gap between the transition piece and the first-stage of the stationary nozzles. In addition, one or more seals may be provided to seal the gap and thereby reduce cold air leakage into the hot gas path. However, due to the high pressure and temperature of the hot gas flowing across the gap, at least a portion of the hot gas may be ingested into the gap and flow against the seal, thereby degrading the seal over time.
One method for cooling the seal and for purging the hot gas from the gap includes flowing a purge medium such as compressed air into the gap at a pressure sufficient to purge the gap and/or cool the seal. Although this method is generally effective, larger gaps require a greater volume of the purging medium to affectively purge the gap. As a result, the greater volume of unburned and/or unmixed purging medium may increase the levels of undesirable combustion emissions such as, but not limiting of, nitrogen oxide (NOx) and/or carbon monoxide (CO). Thus, an improved system and method for recirculating the hot gas ingested into the gap would be useful.

System and method for recirculating a hot gas flowing through a gas turbine


A direct conversion receiver includes a mixer for converting an RF signal into a baseband signal, a dynamic DC offset compensator for compensating for a dynamic DC offset caused by the effect of second order nonlinear distortion of the mixer on an interfering wave that enters the input terminals of the mixer, and a static DC offset compensator for compensating for a static DC offset caused by self-mixing of a local signal and a leakage component of the local signal that appears at the input terminals of the mixer. The static DC offset compensation starts at a time t1 after a DC offset compensation operation has started. The static DC offset compensation is finished at the next time t2, and then the dynamic DC offset compensation starts. The dynamic DC offset compensation is finished at the next time t3. With this configuration, only a change in DC offset due to the dynamic DC offset is compensated after compensating for the static DC offset component. The dynamic DC offset compensator does not have an element for storing the magnitude of the static DC offset.

Technologies using a direct conversion system have been proposed with recent progress in reducing the size and cost of a radio receiver. This system directly converts an RF input signal into a low-frequency baseband signal, and therefore is advantageous, e.g., in eliminating an intermediate frequency filter as compared to a conventional system that requires the intermediate frequency.
A frequency conversion is performed by mixing an RF input signal and a local signal whose frequency is equal to that of the RF input signal. In the direct conversion system, however, if there is second order nonlinear distortion in a mixer, and the input signal level is high, a DC offset occurs in the output baseband signal. This case will be described in detail by referring to FIGS. 10 and 11. FIG. 10 shows the spectrum of the RF input signal. Reference numeral 61 denotes a weak-level desired signal that has a center frequency equal to a local signal frequency fLO. Reference numeral 62 denotes a high-level interfering signal that has a frequency fINT higher than the local signal frequency fLO.
When the RF input signal including such a high-level interfering signal is input to the mixer, the output signal of the mixer has a spectrum as shown in FIG. 11. Reference numerals 63, 64 denote components of the mixer output after the frequency conversion of the desired signal 61 and the interfering signal 62 of the RF input signal, respectively. Reference numeral 65 denotes a DC offset caused by the high-level interfering signal in the presence of second order nonlinear distortion of the mixer.
In the direct conversion system, the DC offset 65 occurs within the band of the desired signal 63 of the mixer output, and thus reduces the reception sensitivity. If the mixer is composed of a differential circuit, and the differential balance is perfectly symmetrical, the second order nonlinear distortion is not present. Actually, however, the elements constituting the differential circuit cannot have perfect symmetry due to manufacturing variations. Therefore, it is not possible to eliminate the second order nonlinear distortion.
On the other hand, in the direct conversion system, leakage of the local signal frequency fLO into the RF input terminal may result in self-mixing of the mixer. The self-mixing also causes a DC offset in the mixer output, even if the input signal level is low. This case will be described in detail by referring to FIGS. 12 and 13. FIG. 12 shows the spectrum of the RF input signal. Reference numeral 66 denotes a spectrum that indicates the leakage of the local signal frequency fLO into the RF input terminal. When the local signal 66 that has leaked into the RF input terminal is input to the mixer, the output signal of the mixer has a spectrum as shown in FIG. 13. Reference numerals 67, 68 denote spectrums obtained by the frequency conversion using the local signal frequency fLO: 67 is a DC offset, and 68 is a signal having a frequency of 2×fLO.
In the direct conversion system, the DC offset 67 occurs due to self-mixing within the band of the desired signal of the mixer output, and thus reduces the reception sensitivity. Because of the asymmetry of a layout of the mixer or the parasitic components of devices constituting the mixer, the self-mixing cannot be removed completely. Therefore, it is not possible to make the DC offset zero in the mixer output.
The low-frequency baseband signal output from the mixer enters a baseband block. The baseband block includes a filter that attenuates the interfering wave other than the desired signal at the RF input terminal, and an amplifier that can vary the gain for maintaining the amplitude of the baseband signal constant at the output terminal of the baseband block, regardless of the magnitude of the input signal from the RF input terminal. However, neither the filter nor the differential circuit of the amplifier can have perfect symmetry due to manufacturing variations. Therefore, a DC offset also occurs in the baseband block. Consequently, the DC offset that has occurred in the mixer output by self-mixing and has been multiplied by the gain of the amplifier is combined with the DC offset that has occurred in the baseband block, and then is output from the output terminal of the baseband block.
As described above, two types of DC offsets are problems for a direct conversion receiver and are categorized as: a DC offset that varies depending on the RF input signal; and a DC offset that is kept constant regardless of the RF input signal. The former can be referred to as "dynamic DC offset", and the latter can be referred to as "static DC offset".
Conventionally, techniques of compensating both for the dynamic DC offset and the static DC offset have been proposed.
U.S. Pat. No. 6,535,725 discloses a method for compensating for a dynamic DC offset that occurs in the output of a mixer by detecting an interfering signal included in an RF input signal. First, this method will be described by referring to FIG. 14.
In FIG. 14, a mixer 73 includes a switching cell 71 and an RF input cell 72. In the RF input cell 72, an RF input signal is input from RF input terminals 83, 84 and amplified. In the switching cell 71, the amplified RF signal is mixed with a local signal that is input from local input terminals 81, 82, and thus is converted into an IF signal. The IF signal is output from output terminals 79, 80. The direct conversion system also is called "zero IF system", since the center frequency of the IF signal is at DC.
The switching cell 71 includes bipolar transistors Q1, Q2, Q3, and Q4. If all the transistors have exactly the same properties, the balance of a differential circuit is perfectly symmetrical. However, the properties of the individual transistors Q1, Q2, Q3 and Q4 deviate from the ideal properties due to manufacturing variations. Therefore, second order nonlinear distortion may occur during the conversion of the RF input signal into the IF signal. This leads to a DC offset in the mixer output, as shown in FIG. 11. As is well known, the DC offset is proportional to the square of the input signal strength. Accordingly, the DC offset of the mixer output is increased as the level of the interfering wave included in the input signal becomes higher.
A DC offset compensator 78 includes a detector 76, a controller 75, a correction generator 74, and a user interface 77. The detector 76 detects the RF input signal and outputs a detection signal. The controller 75 generates a control signal in accordance with the detection signal. The correction generator 74 generates a correction signal in accordance with the control signal of the controller 75 so as to reduce the DC offset at the output terminals 79, 80 of the mixer 73. The operation of the DC offset compensator 78 allows the correction signal of the correction generator 74 to change with the strength of the RF signal input to the mixer 73, thereby compensating for the DC offset of the mixer output. The second order nonlinear distortion of the mixer 73 is caused by manufacturing variations and differs in characteristics individually. Thus, the DC offset compensator 78 further has the function of adjusting the control signal of the controller 75 with the user interface 77.
Next, a method for compensating for the static DC offset that occurs at the output terminal of a baseband block will be described by referring to FIG. 15. FIG. 15 shows the basic configuration of a conventional DC offset calibration system.
In FIG. 15, reference numeral 1 denotes an RF amplifier composed of a LNA (low noise amplifier) for amplifying a high-frequency signal, 2 denotes a mixer for converting an RF input signal into a low-frequency baseband signal, and 3 denotes a local signal generator. A local signal that has leaked into RF input terminals 16, 17 is amplified by the RF amplifier 1, combined with a local signal that has leaked into mixer input terminals 20, 21, and appears at the mixer input terminals 20, 21 as represented by 66 in FIG. 12. This local signal appearing at the mixer input terminals 20, 21 is mixed with a local signal that has a frequency of fLO and is generated from output terminals 22, 23 of the local signal generator 3, so that the frequency is converted, and the DC offset 67 as shown in FIG. 13 occurs at output terminals 24, 25 of the mixer 2.
In FIG. 15, reference numeral 4 denotes a baseband block that includes a LPF (low-pass filter) 5 for attenuating an interfering signal and an AMP 6. The AMP 6 has a variable gain function for maintaining the magnitude of the baseband signal constant, regardless of the magnitude of the RF input signal from the RF input terminals 16, 17. The DC offset 67 and the 2×fLO component 68 caused by self-mixing (FIG. 13) are output from the output terminals 24, 25 of the mixer 2 and input to the LPF 5. Then, the 2×fLO component 68 is attenuated fully. The DC offset 67 appears at output terminals 26, 27 of the LPF 5, is amplified by the AMP 6, and appears at baseband output terminals 28, 29. Because of the asymmetry of a differential circuit of the baseband block 4, a DC offset also occurs in the baseband block 4. Consequently, the DC offset caused by self-mixing and the DC offset that occurred in the baseband block 4 are combined and output from the baseband output terminals 28, 29.
In FIG. 15, a static DC offset compensator 7 includes a comparator 8, a successive approximation register (referred to as "SAR" in the following) 9, and a D/A converter (referred to as "DAC" in the following) 10a. The comparator 8 is used to discriminate the polarity of the DC offset at the baseband output terminals 28, 29. The SAR 9 compares the output signal of the comparator 8 with the polarity of the initial discrimination and outputs zero if the polarity is changed, and outputs 1 if the polarity is unchanged. The DAC 10a converts the digital signal output from the SAR 9 into an analog signal.
The comparator 8 and the SAR 9 constitute a successive approximation A/D converter. The DC offset that occurs in the baseband output is an analog signal, and therefore is converted into a digital signal by the successive approximation A/D converter. The converted digital signal is returned to an analog signal by the DAC 10a. The signal of the baseband output terminals 28, 29 is supplied to the input of the static DC offset compensator 7, and the output of the static DC offset compensator 7 is supplied to the output terminals 24, 25 of the mixer 2, thus forming a feedback loop. The static DC offset compensator 7 supplies a compensation current to the output terminals 24, 25 of the mixer 2 so as to compensate for the DC offset appearing at the baseband output terminals 28, 29.
Referring to FIGS. 16 to 20, the operation of the static DC offset compensator 7 and the process of compensating for the static DC offset will be described in detail, while taking the DAC 10a with 5 bits as an example.
FIG. 16 is a flowchart for explaining the operation of the static DC offset compensator 7.
FIG. 17 shows the relationship between a compensation current flowing into the output terminals 24, 25 (FIG. 15) of the mixer and the amount of a DC offset that occurs for the compensation current. The horizontal axis indicates the magnitude of the compensation current, and the vertical axis indicates the amount of the DC offset appearing at the baseband output terminals 28, 29. In FIG. 17, a line 71 represents the relationship between the amounts of the compensation current and the DC offset when an initial DC offset is output in the positive direction, and a line 72 represents the relationship between the amounts of the compensation current and the DC offset when an initial DC offset is output in the negative direction. The static DC offset compensator 7 switches the polarity of the compensation current in accordance with the polarity of the initial DC offset. For both the lines 71 and 72, the DC offset at the baseband output terminals 28, 29 becomes smaller as the absolute value of the compensation current is increased from the origin on the horizontal axis. Thus, the static DC offset has the characteristics that the amount of the DC offset that occurs in the baseband output is reduced with an increase in the absolute value of the compensation current. Therefore, the DC offset at the baseband output terminals 28, 29 can be adjusted by adjusting the absolute value of the compensation current in accordance with the polarity of the DC offset occurred.
FIGS. 18 and 19 show the process of adjusting the DC offset and correspond to FIG. 17. The vertical axis indicates the amount of the DC offset appearing at the baseband output terminals 28, 29, and the horizontal axis indicates time. FIG. 20 is a read table for each bit of the DAC 10a. The output of the DAC 10a is set so that the compensation current is increased monotonically from LSB to MSB.
The process of adjusting the DC offset when the initial DC offset is output in the positive direction as represented by the line 71 in FIG. 17 will be described below.
In FIG. 16, the static DC offset compensator 7 starts operating in the step S1. In the step S2, all the bits of the DAC 10a are set to zero, so that the read value of the DAC 10a is 00000, and no compensation current is output from the DAC 10a. Therefore, a DC offset in the initial state that corresponds to a time t10 of the line 71 in FIG. 17 occurs at the baseband output terminals 28, 29. The adjustment is performed to reduce this value. Subsequently, in the step S3, the comparator 8 detects the initial DC offset. In the step S4, MSB of the DAC 10a is set to 1 by the SAR 9, and the DAC 10a outputs a compensation current in accordance with 10000. Thus, a DC offset corresponding to a time till of the line 71 in FIG. 17 occurs.
Next, in the step S5, the comparator 8 detects the present DC offset. In the step S6, the SAR 9 compares the initial DC offset with the present DC offset, i.e., the DC offset at the time t10 with the DC offset at the time t11 of the line 71 in FIG. 17. Both the output values at t10 and t11 are positive, and thus not inverted. Then, in the step S8, the SAR 9 determines whether the bit is LSB or not. Since the result is NO, the SAR 9 sets the next bit to 1 in the step S9, and the DAC 10a is set at 11000. The DAC 10a outputs a compensation current in accordance with 11000, and a DC offset corresponding to a time t12 of the line 71 in FIG. 17 occurs.
Again, in the step S5, the comparator 8 detects the present DC offset. In the step S6, the SAR 9 compares the initial DC offset with the present DC offset, i.e., the DC offset at the time t10 with the DC offset at the time t12. As shown in FIG. 17, both the output values at t10 and t12 are positive and thus not inverted. Then, in the step S8, the SAR 9 determines whether the bit is LSB or not. Since the result is NO, the SAR 9 sets the next bit to 1 in the step S9, and the DAC 10a is set at 11100. The DAC 10a outputs a compensation current in accordance with 11100, and a DC offset corresponding to a time t13 of the line 71 in FIG. 17 occurs.
Again, in the step S5, the comparator 8 detects the present DC offset. In the step S6, the SAR 9 compares the initial DC offset with the present DC offset, i.e., the DC offset at the time t10 with the DC offset at the time t13. As shown in FIG. 17, the output values at t10 and t13 are inverted. Therefore, in the step S7, the present bit is returned to zero, and 11000 is read from the SAR 9. Then, in the step S8, the SAR 9 determines whether the bit is LSB or not. Since the result is NO, the SAR 9 sets the next bit to 1 in the step S9, and the DAC 10a is set at 11010. The DAC 10a outputs a compensation current in accordance with 11010, and a DC offset corresponding to a time t14 of the line 71 in FIG. 17 occurs.
Again, in the step S5, the comparator 8 detects the present DC offset. In the step S6, the SAR 9 compares the initial DC offset with the present DC offset, i.e., the DC offset at the time t10 with the DC offset at the time t14. As shown in FIG. 17, the output values at t10 and t14 are inverted. Therefore, in the step S7, the present bit is returned to zero, and 11000 is read from the SAR 9. Then, in the step S8, the SAR 9 determines whether the bit is LSB or not. Since the result is NO, the SAR 9 sets the next bit to 1 in the step S9, and the DAC 10a is set at 11001. The DAC 10a outputs a compensation current in accordance with 11001, and a DC offset corresponding to a time t15 of the line 71 in FIG. 17 occurs.
Again, in the step S5, the comparator 8 detects the present DC offset. In the step S6, the SAR 9 compares the initial DC offset with the present DC offset, i.e., the DC offset at the time t10 with the DC offset at the time t15. As shown in FIG. 17, both the output values at t10 and t15 are positive and thus not inverted. Then, in the step S8, the SAR 9 determines whether the bit is LSB or not. Since the result is YES, 11001 is stored in the DAC 10a. In the step S10, the operation is END, and the DC offset compensation is finished.
The above operation can be summarized in FIG. 18. The initial DC offset at t10 is approximated successively during the time from t11 to t15 and compensated in the direction of decreasing the DC offset. At t16, the DC offset compensation is finished with the storage of 11001 in the DAC 10a, and the static DC offset that is output in the positive direction at the baseband output terminals 28, 29 can be compensated.
Next, the process of adjusting the DC offset when the initial DC offset is output in the negative direction as represented by the line 72 in FIG. 17 will be described below.
In FIG. 16, the static DC offset compensator 7 starts operating in the step S1. In the step S2, all the bits of the DAC 10a are set to zero, so that the read value of the DAC 10a is 00000, and no compensation current is output from the DAC 10a. Therefore, a DC offset in the initial state that corresponds to a time t20 of the line 72 in FIG. 17 occurs at the baseband output terminals 28, 29. The adjustment is performed to reduce this value. Subsequently, in the step S3, the comparator 8 detects the initial DC offset. In the step S4, MSB of the DAC 10a is set to 1 by the SAR 9, and the DAC 10a outputs a compensation current in accordance with 10000. Thus, a DC offset corresponding to a time t21 of the line 72 in FIG. 17 occurs.
Next, in the step S5, the comparator 8 detects the present DC offset. In the step S6, the SAR 9 compares the initial DC offset with the present DC offset, i.e., the DC offset at the time t20 with the DC offset at the time t21. As shown in FIG. 17, both the output values at t20 and t21 are negative and thus not inverted. Then, in the step S8, the SAR 9 determines whether the bit is LSB or not. Since the result is NO, the SAR 9 sets the next bit to 1 in the step S9, and the DAC 10a is set at 11000. The DAC 10a outputs a compensation current in accordance with 11000, and a DC offset corresponding to a time t22 of the line 72 in FIG. 17 occurs.
Again, in the step S5, the comparator 8 detects the present DC offset. In the step S6, the SAR 9 compares the initial DC offset with the present DC offset, i.e., the DC offset at the time t20 with the DC offset at the time t22. As shown in FIG. 17, the output values at t20 and t22 are inverted. Therefore, in the step S7, the present bit is returned to zero, and 10000 is read from the SAR 9. Then, in the step S8, the SAR 9 determines whether the bit is LSB or not. Since the result is NO, the SAR 9 sets the next bit to 1 in the step S9, and the DAC 10a is set at 10100. The DAC 10a outputs a compensation current in accordance with 10100, and a DC offset corresponding to a time t23 of the line 72 in FIG. 17 occurs.
Again, in the step S5, the comparator 8 detects the present DC offset. In the step S6, the SAR 9 compares the initial DC offset with the present DC offset, i.e., the DC offset at the time t20 with the DC offset at the time t23. As shown in FIG. 17, both the output values at t20 and t23 are negative and thus not inverted. Then, in the step S8, the SAR 9 determines whether the bit is LSB or not. Since the result is NO, the SAR 9 sets the next bit to 1 in the step S9, and the DAC 10a is set at 10110. The DAC 10a outputs a compensation current in accordance with 10110, and a DC offset corresponding to a time t24 of the line 72 in FIG. 17 occurs.
Again, in the step S5, the comparator 8 detects the present DC offset. In the step S6, the SAR 9 compares the initial DC offset with the present DC offset, i.e., the DC offset at the time t20 with the DC offset at the time t24. As shown in FIG. 17, both the output values at t20 and t24 are negative and thus not inverted. Then, in the step S8, the SAR 9 determines whether the bit is LSB or not. Since the result is NO, the SAR 9 sets the next bit to 1 in the step S9, and the DAC 10a is set at 10111. The DAC 10a outputs a compensation current in accordance with 10111, and a DC offset corresponding to a time t25 of the line 72 in FIG. 17 occurs.
Again, in the step S5, the comparator 8 detects the present DC offset. In the step S6, the SAR 9 compares the initial DC offset with the present DC offset, i.e., the DC offset at the time t20 with the DC offset at the time t25. As shown in FIG. 17, the output values at t20 and t25 are inverted. Therefore, in the step S7, the present bit is returned to zero, and 10110 is read from the SAR 9. Then, in the step S8, the SAR 9 determines whether the bit is LSB or not. Since the result is YES, 10110 is stored in the DAC 10a. In the step S10, the operation is END, and the DC offset compensation is finished.
The above operation for adjusting the DC offset as represented by the line 72 in FIG. 17 can be summarized in FIG. 19. The initial DC offset at t20 is approximated successively during the time from t21 to t25 and compensated in the direction of decreasing the DC offset. At t26, the DC offset compensation is finished with the storage of 10110 in the DAC 10a, and the static DC offset that is output in the negative direction at the baseband output terminals 28, 29 can be compensated.
However, when the RF input signal in FIG. 10 is input to the mixer, the actual output of the mixer includes, as shown in FIG. 21, a desired signal 63 converted by the mixer, an interfering signal 64 resulting from the conversion of a high-level interfering signal by the mixer, a dynamic DC offset 65, a static DC offset 67, and a 2×fLO signal 68 converted by the mixer due to self-mixing. Therefore, if the configuration in FIG. 14 is used, the dynamic DC offset 65 has to be compensated in the following manner. First, the magnitude of the static DC offset 67 is stored, and then the magnitude of the dynamic DC offset 65 is detected as the amount by which the DC offset is changed from the static DC offset 67. Therefore, this configuration requires a means for temporally storing the magnitude of the static DC offset 67. Moreover, a radio receiver employing a direct conversion system also should have a means for compensating for the static DC offset, so that the system becomes larger and more complicated.

Direct conversion receiver with DC offset compensation

Matsushita Electric Industrial Co., Ltd.


A round producing apparatus for a spinning reel includes a sound producing piece in the form of a leaf spring attached to a disk mounted on a spool shaft, and an annular toothed element supported by a spool for acting on a distal end of the sound producing piece to produce a clicking sound. The sound producing piece is cantilevered and extends from the disk along directions of rotation of the spool. A coil spring acts as a one-way clutch for allowing the annular toothed element to rotate only in a line unwinding direction of the spool.

1. Field of the Invention
This invention relates to sound producing apparatus for use in spinning reels. More particularly, the invention relates to a sound producing apparatus for a spinning reel having a sound producing mechanism between a spool and a spool shaft for producing a clicking sound from rotation of the spool relative to the spool shaft, the sound producing mechanism including a sound producing piece and an engaging element for acting on the sound producing piece to generate the clicking sound.
2. Description of the Related Art
In a conventional sound producing apparatus for a spinning reel of this type, a sound producing piece attached to the spool shaft is extended to act on an engaging element attached to the spool. A clicking sound is generated with rotation of the spool in either of line winding and unwinding directions.
To generate the clicking sound with rotation in either direction of the spool, it is necessary for the sound producing piece to extend straight to the engaging element and also to have a sufficient thickness for strength.
Thus, although sufficient strength is secured, the clicking sound generated lacks in lilt. In this sense, there is room for improvement.
In order to produce a lilting sound, the sound producing piece needs to have an increased length. However, this is undesirable from the viewpoint of compactness of the spinning reel in that a large gap is formed between the engaging element and a mounting section of the sound producing piece.

Sound producing apparatus for spinning reel