A communication system for sending a sequence of symbols on a communication link. The system includes a transmitter for placing information indicative of the sequence of symbols on the communication link and a receiver for receiving the information placed on the communication link by the transmitter. The transmitter includes a clock for defining successive frames, each of the frames including M time intervals, where M is an integer greater than 1. A modulator modulates each of M carrier signals with a signal related to the value of one of the symbols thereby generating a modulated carrier signal corresponding to each of the carrier signals. The modulated carriers are combined into a sum signal which is transmitted on the communication link. The carrier signals include first and second carriers, the first carrier having a different bandwidth than the second carrier. In one embodiment, the modulator includes a tree-structured array of filter banks having M leaf nodes, each of the values related to the symbols forming an input to a corresponding one of the leaf nodes. Each of the nodes includes one of the filter banks. Similarly, the receiver can be constructed of a tree-structured array of sub-band filter banks for converting M time-domain samples received on the communication link to M symbol values. A stereophonic audio signal decompression method that includes decoding, using a decoder, a compressed stereophonic audio signal. A de-quantizer de-quantizes the compressed stereophonic audio signal to generate sets of frequency components for synthesizing left and right audio signals. A controller switches to constructing a single set of frequency components by averaging corresponding frequency components in the left and right audio signals when a computational workload exceeds a capacity of a decompression system and a synthesizer synthesizes a monaural audio time domain signal.

While digital audio recordings provide many advantages over analog systems, the data storage requirements for high-fidelity recordings are substantial. A high fidelity recording typically requires more than one million bits per second of playback time. The total storage needed for even a short recording is too high for many computer applications. In addition, the digital bit rates inherent in non-compressed high fidelity audio recordings makes the transmission of such audio tracks over limited bandwidth transmission systems difficult. Hence, systems for compressing audio sound tracks to reduce the storage and bandwidth requirements are in great demand.
One class of prior an audio compression systems divide the sound track into a series of segments. Over the time interval represented by each segment, the sound track is analyzed to determine the signal components in each of a plurality of frequency bands. The measured components are then replaced by approximations requiring fewer bits to represent, but which preserve features of the sound track that are important to a human listener. At the receiver, an approximation to the original sound track is generated by reversing the analysis process with the approximations in place of the original signal components.
The analysis and synthesis operations are normally carried out with the aid of perfect, or near perfect, reconstruction filter banks. The systems in question include an analysis filter bank which generates a set of decimated subband outputs from a segment of the sound track. Each decimated subband output represents the signal in a predetermined frequency range. The inverse operation is carried out by a synthesis filter bank which accepts a set of decimated subband outputs and generates therefrom a segment of audio sound track. In practice, the synthesis and analysis filter banks are implemented on digital computers which may be general purpose computers or special computers designed to more efficiently carry out the operations. If the analysis and synthesis operations are carried out with sufficient precision, the segment of audio sound track generated by the synthesis filter bank will match the original segment of audio sound track that was inputted to the analysis filter bank. The differences between the reconstructed audio sound track and the original sound track can be made arbitrarily small. In this case, the specific filter bank characteristics such as the length of the segment analyzed, the number of filters in the filter bank, and the location and shape of filter response characteristics would be of little interest, since any set of filter banks satisfying the perfect, or near-perfect, reconstruction condition would exactly regenerate the audio segment.
Unfortunately, the replacement of the frequency components generated by the analysis filter bank with a quantized approximation thereto results in artifacts that do depend on the detail characteristics of the filter banks. There is no single segment length for which the artifacts in the reconstructed audio track can be minimized. Hence, the length of the segments analyzed in prior art systems is chosen to be a compromise. When the frequency components are replaced by approximations, an error is introduced in each component. An error in a given frequency component produces an acoustical effect which is equivalent to the introduction of a noise signal with frequency characteristics that depend on filter characteristics of the corresponding filter in the filter bank. The noise signal will be present over the entire segment of the reconstructed sound track. Hence, the length of the segments is reflected in the types of artifacts introduced by the approximations. If the segment is short, the artifacts are less noticeable. Hence, short segments are preferred. However, if the segment is too short, there is insufficient spectral resolution to acquire information needed to properly determine the minimum number of bits needed to represent each frequency component. On the other hand, if the segment is too long, temporal resolution of the human auditory system will detect artifacts.
Prior art systems also utilize filter banks in which the frequency bands are uniform in size. Systems with a few (16-32) sub-bands in a 0-22 kHz frequency range are generally called "subband coders" while those with a large number of sub-bands (.gtoreq.64) are called "transform coders". It is known from psychophysical studies of the human auditory system that there are critical bandwidths which vary with frequency. The information in a critical band may be approximated by a component representing the time averaged signal amplitude in the critical band.
In addition, the ear's sensitivity to a noise source in the presence of a localized frequency component such as a sine tone depends on the relative levels of the signals and on the relation of the noise spectral components to the tone. The errors introduced by approximating the frequency components may be viewed as "noise". The noise becomes significantly less audible if its spectral energy is within one critical bandwidth of the tone. Hence, it is advantageous to use frequency decompositions which approximate the critical band structure of the auditory system.
Systems which utilize uniform frequency bands are poorly suited for systems designed to take advantage of this type of approximation. In principle, each audio segment can be analyzed to generate a large number of uniform frequency bands, and then, several bands at the higher frequencies could be merged to provide a decomposition into critical bands. This approach imposes the same temporal constraints on all frequency bands. That is, the time window over which the low frequency data is generated for each band is the same as the time window over which each high-frequency band is generated. To provide accuracy in the low frequency ranges, the time window must be very long. This leads to temporal artifacts that become audible at higher frequencies. Hence, systems in which the audio segment is decomposed into uniform sub-bands with adequate low-frequency resolution cannot take full advantage of the critical band properties of the auditory system.
Prior art systems that recognize this limitation have attempted to solve the problem by utilizing analysis and synthesis filter banks based on QMF filter banks that analyze a segment of an audio sound track to generate frequency components in two frequency bands. To obtain a decomposition of the segment into frequency components representing the amplitudes of the signal in critical bands, these two frequency band QMF filters are arranged in a tree-structured configuration. That is, each of the outputs of the first level filter becomes the input to another filter bank at least one of whose two outputs is fed to yet another level, and so on. The leaf nodes of this tree provide an approximation to a critical band analysis of the input audio track. It can be shown that this type of filter bank used different length audio segments to generate the different frequency components. That is, a low frequency component represents the signal amplitude in an audio segment that is much longer than a high-frequency component. Hence, the need to choose a single compromise audio segment length is eliminated.
While tree structured filter banks having many layers may be used to decompose the frequency spectrum into critical bands, such filter banks introduce significant aliasing artifacts that limit their utility. In a multilevel filter bank, the aliasing artifacts are expected to increase exponentially with the number of levels. Hence, filter banes with large numbers of levels are to be avoided. Unfortunately, filter banks based on QMF filters which divide the signal into two bandlimited signals require large numbers of levels.
Prior art audio compression systems are also poorly suited to applications in which the playback of the material is to be carried out on a digital computer. The use of audio for computer applications is increasingly in demand. Audio is being integrated into multimedia applications such as computer based entertainment, training, and demonstration systems. Over the course of the next few years, many new personal computers will be outfitted with audio playback and recording capability. In addition, existing computers will be upgraded for audio with the addition of plug-in peripherals.
Computer based audio and video systems have been limited to the use of costly outboard equipment such as an analog laser disc player for playback of audio and video. This has limited the usefulness and applicability of such systems. With such systems it is necessary to provide a user with a highly specialized playback configuration, and there is no possibility of distributing the media electronically. However, personal computer based systems using compressed audio and video data promise to provide inexpensive playback solutions and allow distribution of program material on digital disks or over a computer network.
Until recently, the use of high quality audio on computer platforms has been limited due to the enormous data rate required tier storage and playback. Quality has been compromised in order to store the audio data conveniently on disk. Although some increase in performance and some reduction in bandwidth has been gained using conventional audio compression methods, these improvements have not been sufficient to allow playback of high fidelity recordings on the commonly used computer platforms without the addition of expensive special purpose hardware.
One solution to this problem-would be to use lower quality playback on computer platforms that lack the computational resources to decode compressed audio material at high fidelity quality levels. Unfortunately, this solution requires that the audio material be coded at various quality levels. Hence, each audio program would need to be stored in a plurality of formats. Different types of users would then be sent the format suited to their application. The cost and complexity of maintaining such multi-format libraries makes this solution unattractive. In addition, the storage requirements of the multiple formats partially defeats the basic goal of reducing the amount of storage needed to store the audio material.
Furthermore, the above discussion assumes that the computational resources of a particular playback platform are fixed. This assumption is not always true in practice. The computational resources of a computing system are often shared among a plurality of applications that are running in a time-shared environment. Similarly, communication links between the playback platform and shared storage facilities also may be shared. As the playback resources change, the format of the audio material must change in systems utilizing a multi-format compression approach. This problem has not been adequately solved in prior art systems.
In prior art multi-carrier systems, a communication path having a fixed bandwidth is divided into a number of sub-bands having different frequencies. The width of the sub-bands is chosen to be the same for all sub-bands and small enough to allow the distortion in each sub-band to be modeled by a single attenuation and phase shift for the band. If the noise level in each band is known, the volume of data sent in each band may be maximized for any given bit error rate by choosing a symbol set for each channel having the maximum number of symbols consistent with the available signal-to-noise ratio of the channel. By using each sub-band at its maximum capacity, the amount of data that can be transmitted in the communication path for a given error rate is maximized.
For example, consider a system in which one of the sub-channels has a signal-to-noise ratio which allows at least 16 digital levels to be distinguished from one another with an acceptable error rate. In this case, a symbol set having 16 possible signal values is chosen. If the incoming data stream is binary, each consecutive group of 4 bits is used to compute the corresponding symbol value which is then sent on the communication channel in the sub-band in question.
In digitally implemented multi-carrier systems, the actual synthesis of the signal representing the sum of the various modulated carriers is carried out via a mathematical transformation that generates a sequence of numbers that represents the amplitude of the signal as function of time. For example, a sum signal may be generated by applying an inverse Fourier transformation to a data vector generated from the symbols to be transmitted in the next time interval. Similarly, the symbols are recovered at the receiver using the corresponding inverse transformation.
The computational workload inherent in synthesizing and analyzing the multi-carrier signal is related to the number of sub-bands. For example, if Fourier transforms are utilized, the workload is of order NlogN where N is the number of sub-bands. Similar relationships exist for other transforms. Hence, it is advantageous to minimize the number of sub-bands.
There are two factors that determine the number of sub-bands in prior art systems. First, the prior art systems utilize a uniform bandwidth. Hence, the number of sub-bands is at least as great as the total bandwidth available for transmission divided by the bandwidth of the smallest sub-band. The size of the smallest sub-band is determined by need to characterize each channel by a single attenuation and phase shift. Thus, the sub-band having the most rapidly varying distortion sets the number of sub-bands and the computational workload in the case in which white noise is the primary contributor to the signal-to-noise ratio.
In systems in which the major source of interference is narrow band interference, the minimum sub-band is set with reference to the narrowest sub-band that must be removed from the communication channel to avoid the interference. Consider a communication channel consisting of a twisted pair of wires which is operated at a total communication band which overlaps with the AM broadcast band in frequency. Because of the imperfect shielding of the wires, interference from strong radio stations will be picked up by the twisted pair. Hence, the sub-bands that correspond to these radio signals are not usable. In this case, prior art systems break the communication band into a series of uniform sub-bands in which certain sub-bands are not used. Ideally, the sub-bands are sufficiently narrow that only the portion of the spectrum that is blocked by a radio signal is lost when a sub-band is marked as being unusable.
Broadly, it is the object of the present invention to provide an improved multi-carrier transmission system.
It is a further object of the present invention to provide a multi-carrier transmission system having a lower computational workload than imposed by systems having bands of equal band-width.
These and other objects of the present invention will become apparent to those skilled in the art from the following detailed description of the invention and the accompanying drawings.

Stereophonic audio signal decompression switching to monaural audio signal


A (meth)allylsilane compound that a functional group of a (meth)allylsilyl group or a halogenosilyl group bonded to the (meth)allylsilyl group via a spacer group is bonded directly or through a divergent spacer group to a dehydrogenated residue of an amino group of an amino group-containing compound; a carbaminic acid ester group or an amide group derived from a dehydrogenated residue of the amino group; an aromatic compound; a polymerizable unsaturated groups; perfluoro group; a dehydrogenated residue of saccharide or a carbohydrate polyol (excluding when the divergent spacer group is an alkylene group, or an alkylene group and an arylene group); a dehydrogenated residue of an amino acid; a halogenosilyl group; or a substituted silyl group in which a halogen of the halogenosilyl group is substituted.

Organic/inorganic hybrid materials such as organic siloxane compounds exhibit both characteristics of organic properties such as hydrophobicity derived from the organic group and inorganic properties such as hydrophilicity or high reactivity to water and condensational reactivity derived from a siloxy group. A functional material having various functions such as a refractive index control function, a light absorption function, a light emitting function, a charge transport function, etc. can be prepared through polymerization under sol-gel reaction conditions from such organic/inorganic hybrid materials such as especially the organic siloxanes containing a trialkoxysilyl group.
However, the organic/inorganic hybrid materials may not perform a desired silane coupling reaction with other functional groups due to the fact that tri-alkoxy silyl groups thereof tend to preferentially perform exchange reactions with alkoxy groups or silanol groups of other molecules or may be hard to perform sol-gel reactions due to the fact that tri-alkoxy silyl groups tend to perform hydrolysis reactions, when compared to polymerization reaction in which chemical products or reagents which contain no siloxane group are performed through condensation or polymerization. Furthermore the organic/inorganic hybrid materials is difficult to be purified by silica-gel chromatography and also difficult to be stocked for a long period of time, due to the fact that the hybrid materials being unstable to moisture and water. Accordingly, functional materials prepared from the conventional organic/inorganic hybrid materials are poor in yield, in production efficiency and purity.
In Non Patent Document 1, the inventors of the present invention disclose preparation of functional aryl (diallyl)ethoxy silanes which could be precursors in sol-gel reaction and disclose their palladium-catalyzed coupling reaction. Further, in Patent Document 1, the inventors of the present invention disclose a method for preparing an organic silica composite material such as a mesoporous body, etc. through hydrolysis and polycondensation reactions of an organic silane compound having allyl groups in a solvent.
Compounds containing a perfluoro group have a strong bond between a carbon atom and a fluorine atom, so that it has an excellent heat resistance and has further small coefficient of friction or low refraction index, due to the existence of fluorine atoms. Therefore, such compound containing the perfluoro group is used to give water- or oil-repellency, to exhibit a moderate lipophilicity, anti-reflective property and also to protect the surface of a substrate by forming a coating onto a surface of a base material or applying paint thereof. For example, in Patent Document 2, optical members that are made of a hardened material prepared from fluorine-substituted-alicyclic group-containing (meth)acrylic acid ester through heating or an active energy ray irradiation, are disclosed. In Patent Document 3, fine particles which are cross-linked with each other and are made of fluorine containing acrylic polymer which is modified by a crosslinkable functional group and hardened by the crosslinkable functional group, and an optical material containing the bonded fine particles are disclosed. In Patent Documents 2 and 3, a fluorine containing polymer is not chemically bonded directly to the various base materials such as silica particles, glass plates, metal plates, resin films, etc. Fluorine-containing polymers have poor scratch resistance and are easily scratched when contacted.
An organic silane compound which can be a silane coupling agent developing various functionalities has long been desired. Such organic silane compound has more reactivity than the conventional organic/inorganic hybrid materials; has reaction-specific, charge-specific, or stereo-specific property; can be induced from various starting compounds such as amino group-containing compounds; has moderate hydrophilic and hydrophobic properties and three-dimensional characteristics; and further has a moderate condensational or polycondensational reactivity to give functions to various raw materials. Further, a fluorine-containing organic silane compound, which can develop much better scratch resistance, can form a stronger bonding derived from a direct chemical bonding, and can develop moderate water repellency and oil repellency, has been desired. The layer of the fluorine-containing organic silane compound can be more excellent rather than a merely formed coating layer which is attached physically, through roughness of the surface and electrostatic attraction, to the surface of the base material such as particles, plates or film-like ones.

(Meth)allylsilane compound, silane coupling agent thereof, and functional material using the same


An oral appliance includes: 1) a body defining a channel to accommodate an upper dentition; and 2) a motion sensor. The body includes a front portion defining a recess, and the motion sensor is affixed to the front portion.

Concussion and other brain injuries resulting from severe head impacts are a leading cause of death or disability among children and young adults. Sub-concussive head impacts were originally thought to be relatively harmless due to the lack of immediate neurological symptoms; however, a growing body of evidence shows that the cumulative effects of multiple sub-concussive and concussive head impacts can accelerate the onset of Alzheimer's, Parkinson's and other neurodegenerative diseases.
There are many unanswered questions behind the neuroscience of concussion and these questions will remain so without a validated measurement technique. Efforts to model brain biomechanics have been made. However, questions remain about what force levels could lead to a concussive event or whether this threshold changes with age, gender, or type of activity. One promising hypothesis in the field is that white matter injury explains the typical transient symptoms of concussion that last for a few weeks. The hypothesis is that rotation of the head is transmitted into elongation of axons, which varies by the direction in which the head is rotated. As axons are rapidly elongated, microtubules slide past one another by stretching the tau protein that connects filaments. Excessive elongation causes rupture of the tau protein connections, cargo trafficking deficiency, and rupture of the plasma membrane. The massive ion flux caused by this disruption leads to a neurochemical and metabolic cascade, which explains both the increased vulnerability after concussion as well as the two-week time course of recovery stress. This hypothesis is based primarily on in vitro and animal experiments, with a small amount of human data from clinically diagnosed concussions that were reconstructed in the laboratory and simulated in finite element models.
Further, important questions in the field relate to neurodegeneration that may appear later in life due to repetitive concussive or sub-concussive blows. For example, it would be beneficial to know how many sub-concussive or concussive blows significantly increase the risk of Alzheimer's, chronic traumatic encephalopathy, or other neurodegenerative diseases.
Additionally, although it has been documented that vulnerability to concussion increases during the recovery after an initial injury, it is unclear if the force level to cause concussion is actually reduced if a patient is less able to avoid dangerous situations, or simply if the likelihood to report symptoms of concussion is increased.
Thus, there is a need to identify the underlying biomechanical and neurobiological mechanism of head impacts to properly treat, diagnose, and prevent short-term brain injury and long-term neurodegeneration. To understand the underlying biomechanical and neurobiological mechanism of head impacts, a relationship between head impact acceleration and neurological deficits should first be characterized.
To measure head impact acceleration, studies have relied on data from sensors that are constrained in their ability to fully characterize head motions, or on reconstructions of concussive football impacts using non-biofidelic dummy tests. The reliance on imperfect data has led to conflicting conclusions on the mechanism of injury and unreliable brain injury metrics.
Once the biomechanical and neurobiological mechanism of head impacts is understood, preventive measures may be evaluated. Once a relationship between head impact acceleration and neurological deficits is determined, detection of injury may be improved. Real-time measurements of head impacts on the field would provide an ability to quantify the severity of head impacts and count the occurrence of various types of head impacts. Quantifying the severity of head impacts is useful for real-time diagnosis of concussions, which is important because an initial concussion can increase the vulnerability to a second concussion. Counting the occurrence of head impacts is important because repetitive concussive and sub-concussive head impacts can accelerate the onset of long-term neurodegeneration. The current protocol for monitoring brain health involves on field observations and neuropsychological sideline tests to quantify head impact severity and diagnose concussions, and video analysis to count the occurrence of head impacts. Unfortunately, these techniques are subjective, costly, and cannot be implemented in real-time.
It is against this background that a need arose to develop the embodiments described in this disclosure.

Oral appliance for measuring head motions by isolating sensors from jaw perturbance

The Board of Trustees of the Leland Stanford Junior University


A DC offset cancellation circuit that is capable of canceling a DC offset voltage occurring between a pair of differential output signals of a differential amplification circuit, while preventing a signal waveform from being distorted due to accumulation of AC components and a photo-electric pulse conversion circuit that is capable of generating an electrical pulse signal that accurately reproduces a rise timing and a fall timing of an optical pulse signal by canceling the DC offset voltage are provided. A photo-electric pulse conversion circuit is provided with a photodiode, an I–V conversion circuit, a first differential amplification circuit having a DC offset cancellation circuit, a second differential amplification circuit, a reference voltage generation circuit, and a comparison circuit. The DC offset cancellation circuit uses a changeover circuit to change a state of a low-pass filter with a hold function in synchronization with an inversion electrical pulse signal, and performs a negative feedback of a filtered signal which is generated by subjecting third differential signals to low-pass filtration or a hold filtered signal which is a filtered signal held during changeover.

1. Field of the Invention
The present invention relates to a DC offset cancellation circuit that cancels a DC offset voltage occurring between a pair of complementary differential output signals outputted from a differential amplification circuit, a differential amplification circuit with a DC offset cancellation circuit, and a photo-electric pulse conversion circuit that uses the differential amplification circuit capable of DC offset cancellation to convert an optical pulse signal to a corresponding electrical pulse.
Alternatively, it relates to a pulse shaping circuit that generates a shaped pulse signal whose logic changes in a manner similar to a rise and a fall of a base square-wave pulse signal, a pulse generation circuit that uses this pulse shaping circuit, and a photo-electric pulse conversion circuit that uses the pulse shaping circuit to convert an optical pulse signal to a corresponding electrical pulse.
2. Description of Related Art
(Related Art 1)
In a differential amplification circuit that amplifies an input signal and outputs a pair of differential output signals, a difference in reference voltages (hereinafter referred to also as a DC offset voltage) occurring between a pair of complementary differential output signals outputted from the differential amplification circuit, namely, between a non-inversion output signal and an inversion output signal presents at times a problem. Therefore, a differential amplification circuit provided with a DC offset cancellation circuit that cancels the DC offset voltage is proposed.
A photo-electric pulse conversion circuit 10 shown in FIG. 35 will be explained as an example. The photo-electric pulse conversion circuit 10 converts an optical pulse signal LT to an electrical pulse signal xRX. For example, it is used as a receiving circuit in IrDA communications and transmits an inversion electrical pulse signal xRX to a demodulator circuit at a later stage.
When it is used in such optical communications, the distance from a transmitter circuit (a light source) to a receiving circuit (photodiode PD) is not constant, and therefore there are various conditions. In some cases, the received optical pulse signal LT is very feeble due to a long distance, and in other cases, the received signal LT is extremely strong due to a short distance. As a result, a current input signal fluctuates from scores of nA to several mA which is several hundred times as large as the scores of nA. Even in such cases, it is necessary to reliably receive the optical pulse signal, to shape the waveform while accurately maintaining a pulse width thereof, and to send the resultant signal to the demodulator circuit at a later stage.
In the photo-electric pulse conversion circuit 10, when the photodiode PD receives the optical pulse signal LT that rises at a second timing t2 and falls at a first timing t1, a pulsating current signal Iin flows according to the intensity of the light. An I–V conversion circuit IV converts this current signal Iin to a pair of complementary differential voltage signals, namely, a non-inversion voltage signal V1P that is in the same phase as the optical pulse signal LT and the current signal Iin and an inversion voltage signal V1M that is complementary thereto and outputs these signals. The waveforms of the differential voltage signals V1P, V1M when a large signal is inputted are slightly different from those when a small signal is inputted as shown in FIG. 36. When a small signal is inputted, the current signal Iin of the photodiode PD having a pulse width tpw that nearly corresponds to the optical pulse signal LT is obtained. When a large signal is inputted, however, the waveform has a dull rising edge and a dull falling edge though it has a generally square shape. This is because the electrical signal fails to accurately follow changes in the optical input. Particularly, since the falling edge after the first timing t1 falls slowly, the non-inversion voltage signal V1P also falls slowly as shown in FIG. 36.
The differential voltage signals V1P, V1M are then amplified by a first differential amplification circuit AMP1 provided with a DC offset cancellation circuit OFC indicated by dashed lines in FIG. 35 and a second differential amplification circuit AMP2. Then, as shown in FIG. 37, a reference voltage VREF according to an output VO of the amplifier is generated by a reference voltage generation circuit REFG and both signals are compared with each other by a comparison circuit CMP to obtain an inversion electrical pulse signal xRX which has the pulse width tpw corresponding to the optical pulse signal LT and which falls at the second timing t2 and rises at the first timing t1.
More specifically, an offset adding circuit (mixing circuit) OFP is used to mix the offset cancellation voltage VOC into the differential voltage signals V1P, V1M such that a negative feedback is performed, thereby generating second differential signals V2P, V2M which are amplified by the first differential amplification circuit AMP1 to output third differential signals V3P, V3M. In the DC offset cancellation circuit OFC, the third differential signals V3P, V3M are filtered by a low-pass filter LPF having characteristics of a cutoff frequency fc1 and a through rate SR1 to obtain the offset cancellation voltage VOC. Since the DC offset voltage occurring between the third differential signals V3P and V3M is negatively fed back in this manner, the DC offset voltage between the differential output terminals of the differential amplification circuit AMP1 can be canceled. If a DC offset voltage exists, an output VO from the second differential amplification circuit AMP2 fluctuates causing the pulse width obtained in the comparison circuit CMP to fluctuate. Thus, the pulse width of the inversion electrical pulse signal xRX obtained may become different from the optical pulse signal. By canceling the DC offset voltage, however, the inversion electrical pulse signal xRX having the pulse width tpw which accurately corresponds to the optical pulse signal can be obtained.
To obtain the inversion electrical pulse signal xRX having the accurate pulse width tpw, it is necessary to give the reference voltage VREF an appropriate time constant according to the magnitude of the output VO.
(Related Art 2)
It is possible to employ a differential amplification circuit provided with a DC offset cancellation circuit in the same manner also in a photo-electric pulse conversion circuit 20 with another configuration (see FIG. 39).
The photodiode PD receives the optical pulse signal LT that rises at the second timing t2 and falls at the first timing t1 to provide the current signal Iin also in this photo-electric pulse conversion circuit 20. However, the photo-electric pulse conversion circuit 20 uses, instead of the I–V conversion circuit IV, a differentiating I–V conversion circuit DIV to convert a waveform of the current signal Iin to a pair of complementary differential voltage signals VD1P, VD1M whose waveforms are similar to a differentiated waveform of the current signal Iin. The differential voltage signals VD1P, VD1M are then amplified by the differential amplification circuit AMP provided with the offset cancellation circuit OFC to output third differential signals VD3P, VD3M. The third differential signals VD3P and VD3M are compared with each other by the comparison circuit CMP and the inversion electrical pulse signal xRX is obtained.
In the photo-electric pulse conversion circuit 20, the differential voltage signals VD1P, VD1M whose waveforms are similar to a differentiated waveform of the current signal Iin are obtained, and are then amplified. A third differential signal VD3P and a third differential signal VD3M that sharply fall or rise at the first or the second timing t1, t2 are compared. It is therefore possible to accurately reproduce the pulse width tpw of the optical pulse signal LT in the obtained inversion electrical pulse signal xRX. In addition, the circuit has the advantage that there is no need of separately using the reference voltage generation circuit REFG to generate the reference voltage VREF according to the output VO as in related art 1 (see FIG. 35).
A DC offset voltage VOS of a small value may be added in the comparison circuit CMP to prevent a malfunction caused by noise.
(Related Art 3)
In the circuit shown in related art 2 (see FIG. 39), the pulse signal is once differentiated to obtain the differential signals, and using these differential signals, a pulse signal having the same pulse width as the original pulse signal is obtained. As a circuit of the same type, a photo-electric pulse conversion circuit 30 shown in FIG. 41 may be configured.
Namely, in the photo-electric pulse conversion circuit 30, the optical pulse signal LT that rises at the second timing t2 and falls at the first timing t1 is received by the photodiode PD and the current signal Iin is obtained. Then, the current signal Iin is converted to the corresponding voltage signal V1 and the resultant voltage signal is outputted by the I–V conversion circuit IV. The voltage signal V1 is then amplified by the differential amplification circuit AMP. Thereafter, a differentiating differential amplification circuit DAMP is used to differentiate and amplify the second differential signals V2P, V2M to output third differential signals VD3P, VD3M. In addition, an offset voltage VOS of a small value is added so that the reference voltage of the third non-inversion signal VD3P is relatively lower than the reference voltage of the third inversion signal VD3M. These third differential signals VD3P, VD3M are then compared with each other by the comparison circuit CMP to obtain the inversion electrical pulse signal xRX that falls at the second timing t2 and rises at the first timing t1. As mentioned above, the purpose of adding the offset voltage VOS is to prevent a malfunction caused by noise.
The current signal Iin and the voltage signal V1 have slightly dull waveforms that gradually fall after the first timing t1 (see FIG. 36) also in the photo-electric pulse conversion circuit 30. However, since the third differential signals VD3P, VD3M that sharply rise or fall at the first or the second timing t1, t2 are compared to obtain the inversion electrical pulse signal xRX, it is possible to accurately reproduce the pulse width tpw of the optical pulse signal LT in the inversion electrical pulse signal xRX. Furthermore, the circuit has the advantage that there is no need of separately using the reference voltage generation circuit REFG to generate the reference voltage VREF according to the output VO as in the circuit 10 shown in related art 1 (FIG. 35).
In the photo-electric pulse conversion circuit 10 of related art 1, however, the low-pass filter LPF with the cutoff frequency fc1 is used to provide a negative feedback control of the DC offset voltage. Therefore, not only DC components, but also low-frequency components of AC components contained in the pulse signal waveform are fed back. Namely, as shown in FIG. 38(a), if the DC offset voltage DCO exists between the third differential signals V3P and V3M of the first differential amplification circuit AMP1 and is negatively fed back, DC cancellation components are contained in the offset cancellation voltage VOC outputted from the low-pass filter LPF as shown in FIG. 38(b) and they function to cancel the DC offset voltage DCO. However, since the low-frequency AC components also pass through the low-pass filter LPF as mentioned above, the low-frequency AC components are also superposed on the offset cancellation voltage VOC as shown in FIG. 38(b).
The magnitude of these AC components contained in the offset cancellation voltage VOC gradually increases during a second period d2 from the second timing t2 to the first timing t1. During a first period d1 from the first timing t1 to the second timing t2, it gradually decreases to return to an original zero level which is maintained. This is because the AC components are contained in the second period d2 as can be easily understood from FIG. 38(a). The gradient of the graph showing the offset cancellation voltage VOC corresponds to the characteristics of the low-pass filter LPF (the cutoff frequency and the through rate), and the increasing gradient and the decreasing gradient become almost the same.
If the second period d2 is longer than the first period d1 as shown in FIG. 38(c), the AC components contained in the offset cancellation voltage VOC cannot decrease in the first period by the amount increased in the second period. As a result, the AC components contained in the offset cancellation voltage VOC gradually accumulate as shown in FIG. 38(d) (in this example, they gradually increase). Therefore, as a result of accumulated AC components being negatively fed back, the waveforms of the third differential signals V3P and V3M as the output from the first differential amplification circuit AMP1 are distorted. This could result in a malfunction or other problem when the inversion electrical pulse signal xRX is obtained in the comparison circuit CMP. Moreover, as AC components accumulate, the waveforms are distorted so as to gradually shift downward, and approach an upper limit value or a lower limit value of the third differential signals V3P, V3M. As a result, the dynamic range may become small and the signal amplitude may become small, and in extreme cases, the third differential signals V3P, V3M may disappear.
On the other hand, in the photo-electric pulse conversion circuit 20 according to related art 2, the current signal Iin as shown in FIG. 40(b) flows through the photodiode PD when receiving the optical pulse signal LT with the pulse width tpw as shown in FIG. 40(a). FIG. 40(b) shows a case in which a large optical pulse signal LT with a high intensity is inputted. In the first period d1, the current signal Iin forms a gradually decreasing long tail. The signal is then subjected to differentiation and I–V conversion performed by the differentiating I–V conversion circuit DIV to obtain the non-inversion voltage signal VD1P shown in FIG. 40(c) and the inversion voltage signal VD1M. These signals are then amplified by the differential amplification circuit AMP to obtain the third differential signals VD3P, VD3M [see FIG. 40(d)]. In this example, the amplitudes of the amplified third differential signals VD3P, VD3M are limited by the upper limit value or the lower limit value in the second period d2 and the first half of the first period d1. As a result, their waveforms are not similar to the waveform of the non-inversion voltage signal VD1P shown in FIG. 40(c). Unlike the photo-electric pulse conversion circuit 10 [see FIGS. 37(a) and (c)], this circuit employs a differentiated waveform and therefore the non-inversion voltage signal VD1P swings to the positive and negative directions (upward and downward) with respect to the reference voltage.
The low-pass filter LPF with the cutoff frequency fc1 (through rate SR1) is used also in the photo-electric pulse conversion circuit 20 according to related art 2. Therefore, not only the DC components but also the low-frequency components of the AC components are fed back. That is, the AC components are superposed on the offset cancellation voltage VOC.
The magnitude of the AC components contained in the offset cancellation voltage VOC gradually increases in the second period d2 from the second timing t2 to the first timing t1 as shown in FIG. 40(e) and gradually decreases in the first period d1 from the first timing t1 to the second timing t2. Unlike the photo-electric pulse conversion circuit 10 [see FIG. 38(b)], however, it continues decreasing in the first period d1. The gradient of the graph showing the offset cancellation voltage VOC corresponds to the characteristics of the low-pass filter LPF (the cutoff frequency and the through rate), and the increasing gradient and decreasing gradients become almost the same.
If the first period d1 is not equal to the second period d2 (if the duty ratio of the pulse is not 50%), the offset cancellation voltage VOC gradually fluctuates. If d1>d2 as shown in FIG. 40, for example, the AC components contained in the offset cancellation voltage VOC gradually accumulate, causing the offset cancellation voltage VOC to gradually diminish as shown in FIG. 40(e).
As a result, the third differential signals VD3P, VD3M of the differential amplification circuit AMP are distorted and the third non-inversion signal VD3P approaches the upper limit value as shown in FIG. 40(f), for example. Therefore, a malfunction may occur when obtaining the inversion electrical pulse signal xRX in the comparison circuit CMP. Moreover, the waveforms approach the upper limit value or the lower limit value of the third differential signals VD3P, VD3M. As a result, the dynamic range may become small and the signal amplitude may become small and, in extreme cases, the third differential signals VD3P, VD3M may become disappear.
In the photo-electric pulse conversion circuit 30 according to related art 3, when the pulse width tpw of the base pulse signal, that is, the optical pulse signal LT becomes long, the gradually downward-going third non-inversion signal VD3P and the gradually upward-going third inversion signal VD3M can cross at a time tx after the second timing t2 as shown in FIG. 42. Then, the inversion electrical pulse signal xRX which is the output of the comparison circuit CMP is inverted. Therefore, as shown in the lower part of FIG. 42, there arises a problem that the pulse width of the inversion electrical pulse signal xRX, which should rise at the first timing t1, becomes shorter. Particularly, the pulse width tends to become shorter when the offset voltage VOS is made greater in an attempt to prevent a malfunction caused by noise.
If the comparison circuit CMP is given hysteresis characteristics (hysteresis voltage Vh) so as to satisfy Vh>VOS as shown in FIG. 43, the third non-inversion signal VD3P and the third inversion signal VD3M do not cross between the second timing t2 and the first timing t1, and thus the correct pulse width tpw is obtained in the inversion electrical pulse signal xRX.
In the case where an arrangement is made to satisfy Vh>VOS as described above and once the third inversion signal VD3M becomes smaller than the third non-inversion signal VD3P when the circuit is started or noise intrudes, the inversion electrical pulse xRX which is the output of the comparison circuit CMP is inverted, that is, the level of the inversion electrical pulse xRX becomes LOW level as shown in FIG. 44. This also causes the same effect as relatively decreasing the third non-inversion signal VD3P by the amount equal to hysteresis voltage Vh. Since the third inversion signal VD3M becomes smaller than the third non-inversion signal VD3P, the inversion electrical pulse signal xRX is fixed to LOW level even after the noise has disappeared and the inversion electrical pulse signal xRX does not fall at the second timing t2. It thereafter returns to HIGH level at the first timing t1. In this case, therefore, the optical pulse signal has not been properly received.
Thus, in the photo-electric pulse conversion circuits 20, 30 according to related arts 2, 3, it is difficult to prevent a malfunction due to noise or the like by setting the offset voltage VOS and the hysteresis voltage Vh to adequate values simultaneously.

DC offset cancellation circuit, differential amplification circuit with DC offset cancellation circuit, photo-electric pulse conversion circuit, pulse shaping circuit, and pulse generation circuit


An improved method and process is provided for verifying a digital logic design complies with certain manufacturing test rules or guidelines. A replacement is created for any portion of a design to make it usable by the manufacturing test tool set, without requiring the contents of that portion of the design to be implemented. The inputs and outputs of a portion of the design are examined for violations of the manufacturing test rules or guidelines. If there are no violations, the contents of this portion of the design are replaced with some basic contents which satisfy the manufacturing structure rules. The interconnections between logic blocks can then be tested using test generation tools to ensure the design does not violate manufacturing test rules or guidelines The compliance verification can thus be done much earlier in the design process than typically occurs.

During the design of any digital component, tests are required to determine the manufacturability of the component. FIG. 1 illustrates a conventional flow diagram of a typical digital design. Rough sizing and capabilities are created in the Design Concept stage, as represented by blocks 1, 2, and 3, along with some high level data or control flow paths, and some high level interconnections between major functions. During the high level design (HLD) phase, represented by blocks 4, 5, and 6, the design is broken down into its smallest pieces. These design pieces are coded in a design language, for example, Very high speed integrated circuit Hardware Description Language (VHDL). Possible design languages are well known in the art. At this point, the design represents a close approximation of the final design, where the coded pieces do not necessarily achieve the final timing, area, or power budgets, nor do they implement all the necessary functions. During the low level design (LLD) phase, represented by blocks 7, 8, and 9, all levels of logic are implemented. Through successive iterations, the design eventually meets all budgets and performs all functions.
The verification of manufacturability, i.e., that the complete design complies with test structure rules, cannot begin until all low level blocks have been coded, represented in box 6. The impact of block interconnections on manufacturability also cannot be verified until box 6, or perhaps as late as chip completion, represented by box 9. This can cause schedule slippage and/or lower quality manufacturing tests to be developed if problems are found late in the design process.
Accordingly, there exists a need for an improved method and process to verify a digital logic design complies with certain manufacturing test rules or guidelines. The improved method and process should allow verification of manufacturability before all low level logics are implemented. The present invention addresses such a need.

Method and apparatus for manufacturing test generation

International Business Machines Corporation


A process is provided in which an alcohol having X carbon atoms is reacted over an L-type zeolite catalyst to produce a higher molecular weight alcohol. In some embodiments, a first alcohol having X carbon atoms is condensed with a second alcohol having Y carbon atoms to produce a branched-chain alcohol having X+Y carbon atoms. Processes for making ethers useful as fuel oxygenates which incorporate the foregoing process steps also are disclosed.

The production of higher molecular weight alcohols from lower molecular weight feedstocks has long interested organic chemists. Such reactions are of much greater interest following the 1991 amendments to the Clean Air Act. These amendments require the blending of fuel oxygenates into gasoline in non-attainment regions of the United States. The required oxygenates are intended to lower carbon monoxide and ozone pollution contributions from internal combustion engine exhausts. Two compounds particularly suited for use as gasoline oxygenates are tertiary amyl methyl alcohol ether (TAME) and methyl tertiary butyl ether (MTBE).
Certain branched-chain alcohols such as isobutanol and isoamyl alcohol are useful as precursors for TAME and MTBE. Isobutanol is useful because it can be dehydrated to form isobutylene, which subsequently can be etherified with methanol to yield MTBE. Isoamyl alcohol, a common name for an isomeric mixture of 2-methyl-1-butanol and 3-methyl-1-butanol, can be reacted with methanol to yield TAME. The expected widespread use of fuel oxygenates requires that refiners develop efficient processes for producing these oxygenates and their precursors.
Guerbert syntheses typically have been used to prepare higher molecular weight branched-chain alcohols from lower molecular weight starting materials. A representative Guerbert synthesis is disclosed in U.S. Pat. No. 4,518,810. In this example, a single alcohol feedstock is batch-reacted over a copper-nickel alkaline catalyst at temperatures from about 200 to 250 degrees Centigrade to produce a dimerized alcohol product. The product alcohol is a branched-chain dimer alcohol in which a dehydration reaction has removed the hydroxyl group from the alpha carbon of a first alcohol molecule and a hydrogen atom from a beta methylene carbon of a second alcohol molecule. The resultant alcohol has a carbon--carbon bond located between the alpha carbon of the first alcohol's carbon chain and the beta carbon of the second alcohol's carbon chain.
In addition to the dimerization reaction discussed above, Guerbert syntheses also have been used to condense together two different alcohols. In this case, at least one of the two alcohols must contain a dehydratable carbon atom at the beta position. An example of this type of reaction is that disclosed in U.S. Pat. No. 2,050,788, in which a mixture of ethyl and methyl alcohol and a stoichiometric excess of hydrogen is condensed over a MgO/CuO catalyst at temperatures between about 200 and 350 degrees Centigrade to provide a mixture of higher molecular weight alcohols.
While the foregoing Guerbert synthesis reactions can, under certain conditions, provide useful yields of certain higher molecular weight alcohols, the catalysts employed are believed to be subject to scintering under regeneration conditions and therefore not well-suited to the cyclic operation common in catalytic refinery operations. Refiners therefore continue to seek improved processes for producing fuel oxygenates and oxygenate precursors. Preferably, these processes should employ modern, commercially-available, easily-regenerated catalysts.


A medical information server receives a signal from a client device over a network, representing a first user interaction of a user with respect to first medical information displayed to a user. A user interaction analyzer invokes a first set of ECCD rules associated with the user based on the first user interaction to determine medical data categories that the user is likely interested in. The first set of ECCD rules was generated by an ECCD engine based on prior user interactions of the user. A data retrieval module accesses medical data servers corresponding to the medical data categories to retrieve medical data of the medical data categories. A view generator integrates the retrieved medical data to generate one or more views of second medical information and transmits the views of second medical information to a client device to be displayed on a display of the client device.

As medical software becomes more sophisticated and universal, intelligent integration of information and data from different data sources becomes more important and more challenging. For example, electronic medical records (EMRs) or electronic health records (EHRs) are becoming increasingly common as is clinical trials software, research software as well as others. But medical data resides in several data sources that do not communicate well together, including a radiology information system (RIS), a hospital information system (HIS), a picture archiving and communication system (PACS), a laboratory information system (LIS), an enterprise content management (ECM) system, EMR data, various directories as well as other data sources. These data sources may be provided in various servers operated by different organizations or entities.
In addition to the lack of communication among various data sources, medical software, such as EMR, HER, clinical trial software etc., even where they do connect with one or more of these data sources, do not present the information in an intelligent nor useful manner. A user, such as a physician, technician, patient etc., needs to access all of the information that is relevant easily and without having to actively open several different software packages, search for the patient, and then search for the information that is pertinent.
For example, if a patient is being treated for lung cancer, his physician needs to view the patient's medications, relevant procedures he/she has had, results of the procedures, results of relevant lab tests, demographic data, relevant notes, reports, dictation, etc. The physician does not need to view an x-ray of a broken toe, for example.
Currently this is not practical within any medical software packages. Some EMR software allows a user to view a static image, such as an X-ray, along with the demographic data, but the user is not able to easily view all relevant information together. Currently, the user must know where to find the relevant information and back out of screens, directories, search windows etc. to view it, or locate the information on a different system with a different login etc. Although some connectors to the various data sources exist, there does not exist an engine, or system, which has the knowledge to display relevant data at the same time, in a way that is useful to the end user.
There is a need for an intelligent system which can not only connect to varied and different data sources, but display relevant information, in context, to the end user in a way that is useful. Ideally such an intelligent system will display the relevant information from different data sources to the user, in context, within a viewer which is organized in a logical way to display the data.

Evolving contextual clinical data engine for medical information


A pre-tensioned bearing assembly, including: a first rolling bearing including a first outer ring, first inner ring including a first cylindrical surface, and a plurality of rolling elements radially disposed between the first inner and outer rings; a second rolling bearing including a second outer ring, a second inner ring including a second cylindrical surface, and a plurality of rolling elements radially disposed between the second inner and outer rings; a shaft including a shoulder in contact with the second rolling bearing; a first conically-shaped sleeve in contact with the first cylindrical surface and the second cylindrical surface, and including a conical inner surface; a second conically-shaped sleeve in contact with the conical inner surface; and a shaft nut including an internal thread in cooperation with the shaft, and fixing the first inner ring and the second inner ring axially onto the shaft.

A pre-tensioned rolling element bearing is understood to mean an arrangement in which a pre-tensioning acts on the rolling bearings used for support of, for example, a shaft, which restores an existing play between the rolling elements and the raceways provided by the bearing rings, thus ensuring a smooth and precise running of the rolling bearing
To set this pre-tensioning, so-called shrink fitting has become established. For this purpose, the rolling bearing to be mounted is heated and, in this state, fitted onto the shaft with the bore thereof in the inner ring. If the rolling bearing then cools down, an interference fit is created between the inner ring of the rolling bearing and the shaft, which is responsible for the slight expansion of the inner ring, and any play in the unmounted state of the rolling bearing is reset or even completely eliminated. If the shaft is to be protected, it is not uncommon for an intermediate ring to be provided before the rolling bearing is shrunk onto the shaft, which is also shrunk onto the shaft before the rolling bearing is connected to the shaft. If an intermediate ring is used, the interference fit between the intermediate ring and the rolling bearing results in the pre-tensioning of the rolling bearing.
As is easy to see, the production of the desired pre-tensioning of the rolling bearing by shrink fitting is very complex. The same applies, of course, to the disassembly of pre-tensioned rolling bearings, because in contrast to assembly, the bore in the inner ring cannot be enlarged versus the shaft by heating, because heat sources acting on the rolling bearing always heat the shaft as well. For this reason, pre-tensioned rolling bearings can generally only be disassembled by using mechanically acting measures that destroy the rolling bearing.


Systems, methods, and other embodiments associated with secure service discovery in a neighbor awareness network are described. According to one embodiment, a device includes service logic configured to generate a discovery communication that is associated with a service provided in a neighbor awareness network (NAN). The NAN includes a plurality of remote devices. Encoding logic configured to generate a secure identifier from a unique identifier of the service by, (i) encoding the unique identifier, and (ii) truncating the encoded unique identifier to form the secure identifier. The secure identifier is inserted in the discovery communication.

As wireless devices become more pervasive and technologically capable, services shared between the wireless devices also become more pervasive. For example, the services may include functionality that is provided between devices in an ad-hoc manner or functionality that relates to interacting with other devices within a close proximity when a wireless device is moving between locations. A wireless device that moves between locations will not initially be aware of other devices in different locations and not aware of services provided by the other devices. Discovery communications are a type of wireless communication that the wireless device may use to discover devices and services provided by the devices.
A discovery communication is an open communication that is not encrypted or otherwise secured in order to permit other devices to easily determine availability of services from the discovery communication. However, because the communications are open they are vulnerable to eavesdropping and attacks by malicious users. Accordingly, sharing services between wireless devices presents security difficulties.

Method and apparatus for securely discovering services in a wireless network


A heat-transfer recording medium is disclosed, which produces no lowering in transfer density even when the medium is used under such higher-temperature conditions as in a facsimile equipment and is capable of providing an easily readable transferred image having a low surface gloss, as well as having an excellent keep-secrecy and producing few voids. The recording medium comprises a support; a heat-transfer layer, provided on the support, comprising a thermo-fusible substance and a colorant; and a non-transfer layer which is interposed between the support and the transfer layer and comprises a colorant, a thermoplastic polymer selected from the following Group A and a thermoplastic polymer selected from the following Group B:. Group A: an .alpha.-olefin polymer, an .alpha.-olefin-maleic anhydride copolymer, an .alpha.-olefin-alkyl acrylate and methacrylate copolymer, an .alpha.-olefin-vinyl acetate copolymer, an .alpha.-olefin-acrylic or methacrylic acid copolymer, a vinyl acetate polymer, an alkyl acrylate or methacrylate polymer, a acrylic or methacrylic acid polymer and a styrene-containing polymer; PA0 Group B: A polyester resin, a polyurethane resin, a polyvinyl chloride resin and an epoxy resin.

In recent years, the attempts have been made by adopting heat-transfer image recording systems to facsimile equipment so as to improve the preservability of recorded images to make them better than in the conventional heat-sensitive image forming systems applied to facsimile equipments, as well as to make it possible to print the images on plain paper sheets.
There has, however, been a problem that the printed images obtained from the conventional heat-transfer image recording media are hardly readable because the surfaces thereof are high in gloss.
There have been proposed, for example, a heat-transfer image recording medium interposed a mat layer containing a resin and a mat pigment between a support and an ink layer so as to diminish the gloss of transferred images, or another heat-transfer image recording medium interposed an anchor layer capable of forming fine unevenness on the pealable surface of an ink layer so as to diminish the gloss of transferred images.
Though the heat-transfer image recording media mentioned above may be able to diminish the gloss of transferred images, when using a heat-transfer image recording medium such as shown in FIG. 2, for example, with a facsimile equipment easily become regenerative in high-temperature conditions because the power is being switched ON all day long, there have been the following problems; Ink layer A containing a heat-fusible substance is liable to be peeled off from the inside of the ink layer so as to hinder ink layer A from peeling off from interface C between the ink layer A and a mat or anchor layer that is expected to be the peelable surface of ink layer A and, therefore, the thickness of ink layer A which is to be resultingly transferred onto a subject transferred medium becomes thinner so as to lower a transferred image density. In FIG. 2, D is a support member.
In the conventional heat-transfer image recording media have a problem that the keep-secrecy is liable to leak out, because readable portions so called `white clearness portions` are so produced as to correspond to the configurations of transferred images after the ink layer is transferred to a transferred subject medium.
For trying to improve the quality of printed images formed on heat-transfer image recording media so as to obtain sharp and clear printed images, it is desirable to make void productions a few and resolving power higher.


A method and device are described for the production of foams from two-component reactive mixtures using carbon dioxide as expanding agent, by mixing at least one of the reactive components with carbon dioxide under pressure. That mixture is then mixed with the other reactive component(s) to form a foamable reactive mixture, which is expanded and cured to form the final product. The foamable reactive mixture is expanded to ambient pressure through at least one fine-meshed screen (21a, 21b, 21c).

The present invention relates to a method and a device for the production of foams using as an expanding agent carbon dioxide dissolved under pressure, whereby the substance to be expanded is mixed under pressure with preferably liquid carbon dioxide and subsequently expanded, with the formation of foam. The expandable substances used are in particular liquid starting products for plastics, which cure to form foamed plastics due to a polyaddition reaction or polycondensation reaction which commences after the expansion. The invention relates specifically to polyurethane foams.
In the production of polyurethane foams, a liquid or gaseous expanding agent is added to at least one of the reactive components (polyisocyanate and compounds containing hydrogen atoms reactive with isocyanate groups, in particular polyols). The other component is then added with mixing and the mixture obtained is conveyed either batchwise into a mold or continuously on a conveyor belt, where the mixture foams and cures.
For the production of the foam a number of methods have found wide application in practice. Liquids (such as low-molecular chlorofluoro-carbons, methylene chloride, pentane and the like) which evaporate at low temperature are often used. These materials evaporate out of the liquid reaction mixture and form bubbles. It is also possible to force air into the reaction mixture or into one of the components (mechanical foam production).
Finally, in the case of polyurethane foams, water is often added to the polyol component as the expanding agent. After mixture with the isocyanate component, the water liberates carbon dioxide as the expanding gas by reacting with the isocyanate (chemical foam production).
Liquid carbon dioxide has already often been proposed as an expanding agent on the grounds of environmental acceptability and industrial hygiene and owing to the comparatively high solubility of liquid carbon dioxide in the polyol component. See, for example, British patent 803,771 and U.S. Pat. Nos. 3,184,419, 4,337,318 and 5,120,770. However, liquid carbon dioxide has not gained acceptance in practice, apparently due to the difficulties of producing uniform foams during the necessary expansion of the reaction mixture from pressures of between 10 and 20 bar. The problem is, on one hand, that the carbon dioxide vaporizes relatively suddenly so that a very large increase in volume takes place in the reaction mixture, for example, by a factor of approximately 10, which is difficult to control. On the other hand, the reaction mixture tends to inhibit release of the carbon dioxide, which can be from 3 to 6 bar below the equilibrium vapor pressure of CO.sub.2 at the relevant temperature, so that a sudden explosive release of carbon dioxide occurs, with the result that large voids or bubbles are enclosed within the foam.
According to company literature from the Cannon Group, the problem of expansion by means of liquid carbon dioxide can be overcome by adding oxygen to the reaction mixture to form bubble nuclei, by carrying out the expansion in stages and by creating a special device for depositing the foam. Details of the method have not as yet been published and are not known.

Method and device for the production of foam using carbon dioxide dissolved under pressure


A system of measuring hemoglobin and bilirubin parameters in a whole blood sample using optical absorbance. The system includes an optical-sample module, a spectrometer module, an optical fiber module optically connecting the optical-sample module to the spectrometer module, and a processor module. The optical-sample module has a light-emitting module having a LED light source, a cuvette and a calibrating-light module. The processor module receives and processes an electrical signal from the spectrometer module and transforms the electrical signal into an output signal useable for displaying and reporting hemoglobin parameter values and/or total bilirubin parameter values for the whole blood sample.

1. Field of the Invention
The present invention relates generally to spectroscopic systems and methods for the identification and characterization of hemoglobin parameters in blood.
2. Description of the Prior Art
An ultraviolet-visible light spectroscopic system involves absorption spectroscopy or reflectance spectroscopy. As the name implies, such systems use light in the visible and near ultraviolet ranges for analyzing a sample. The wavelength range is typically from about 400 nm to about 700 nm. The absorption or reflectance of the visible light directly affects the perceived color of the chemicals involved. UV/Vis spectroscopy is routinely used in analytical chemistry for the quantitative determination of different analytes, such as transition metal ions, highly conjugated organic compounds, and biological macromolecules. Spectroscopic analysis is commonly carried out in solutions but solids and gases may also be studied.
A near-infrared spectroscopic system also involves absorption spectroscopy or reflectance spectroscopy. Such systems use light in the near-infrared range for analyzing a sample. The wavelength range is typically from about 700 nm to less than 2,500 nm. Typical applications include pharmaceutical, medical diagnostics (including blood sugar and pulse oximetry), food and agrochemical quality control, and combustion research, as well as research in functional neuroimaging, sports medicine & science, elite sports training, ergonomics, rehabilitation, neonatal research, brain computer interface, urology (bladder contraction), and neurology (neurovascular coupling).
Instrumentation for near-IR (NIR) spectroscopy is similar to instruments for the UV-visible and mid-IR ranges. The basic parts of a spectrophotometer are a light source, a holder for the sample, a diffraction grating in a monochromator or a prism to separate the different wavelengths of light, and a detector. The radiation source is often a Tungsten filament (300-2500 nm), a deuterium arc lamp, which is continuous over the ultraviolet region (190-400 nm), Xenon arc lamp, which is continuous from 160-2,000 nm, or more recently, light emitting diodes (LED) for the visible wavelengths. The detector is typically a photomultiplier tube, a photodiode, a photodiode array or a charge-coupled device (CCD). Single photodiode detectors and photomultiplier tubes are used with scanning monochromators, which filter the light so that only light of a single wavelength reaches the detector at one time. The scanning monochromator moves the diffraction grating to "step-through" each wavelength so that its intensity may be measured as a function of wavelength. Fixed monochromators are used with CCDs and photodiode arrays. As both of these devices consist of many detectors grouped into one or two dimensional arrays, they are able to collect light of different wavelengths on different pixels or groups of pixels simultaneously. Common incandescent or quartz halogen light bulbs are most often used as broadband sources of near-infrared radiation for analytical applications. Light-emitting diodes (LEDs) are also used. The type of detector used depends primarily on the range of wavelengths to be measured.
The primary application of NIR spectroscopy to the human body uses the fact that the transmission and absorption of NIR light in human body tissues contains information about hemoglobin concentration changes. By employing several wavelengths and time resolved (frequency or time domain) method and/or spatially resolved methods, blood flow, volume and absolute tissue saturation (StO2 or Tissue Saturation Index (TSI)) can be quantified. Applications of oximetry by NIRS methods include neuroscience, ergonomics, rehabilitation, brain computer interface, urology, the detection of illnesses that affect the blood circulation (e.g., peripheral vascular disease), the detection and assessment of breast tumors, and the optimization of training in sports medicine.
With respect to absorption spectroscopy, the Beer-Lambert law states that the absorbance of a solution is directly proportional to the concentration of the absorbing species in the solution and the path length. Thus, for a fixed path length, UV/Vis and NIR spectroscopy can be used to determine the concentration of the absorber in a solution. The method is most often used in a quantitative way to determine concentrations of an absorbing species in solution, using the Beer-Lambert law: A=log10(I0/I)=εcL
where A is the measured absorbance, in Absorbance Units (AU),
I0 is the intensity of the incident light at a given wavelength,
I is the transmitted intensity,
L the path length through the sample, and
c the concentration of the absorbing species.
For each species and wavelength, ε is a constant known as the molar absorptivity or extinction coefficient. This constant is a fundamental molecular property in a given solvent, at a particular temperature and pressure, and has units of 1/M*cm or often AU/M*cm. The absorbance and extinction ε are sometimes defined in terms of the natural logarithm instead of the base-10 logarithm.
The Beer-Lambert Law is useful for characterizing many compounds but does not hold as a universal relationship for the concentration and absorption of all substances.
It is recognized by those skilled in the art that various factors affect these spectroscopic systems. These factors include spectral bandwidth, wavelength error, stray light, deviations from the Beer-Lambert law, and measurement uncertainty sources.
Stray light is an important factor that affects spectroscopic systems. Stray light causes an instrument to report an incorrectly low absorbance.
Deviations from the Beer-Lambert law arise based on concentrations. At sufficiently high concentrations, the absorption bands will saturate and show absorption flattening. The absorption peak appears to flatten because close to 100% of the light is already being absorbed. The concentration at which this occurs depends on the particular compound being measured.
Measurement uncertainty arises in quantitative chemical analysis where the results are additionally affected by uncertainty sources from the nature of the compounds and/or solutions that are measured. These include spectral interferences caused by absorption band overlap, fading of the color of the absorbing species (caused by decomposition or reaction) and possible composition mismatch between the sample and the calibration solution.

Analyte system and method for determining hemoglobin parameters in whole blood


A circuit with variation correction function is capable of obtaining an output characteristic near a desired value by suppressing variation of the output characteristic regardless of manufacturing characteristic variations of a component. An output signal different in phase from a reference signal is obtained by a dummy circuit having a capacitor of a same structure as a correction object capacitor in an operation circuit. The difference in phase between the reference signal and the output signal reflects manufacturing characteristic variation of the correction object capacitor. The difference in phase is detected by a phase comparator circuit and control signals are created by a control signal conversion circuit. Switches in the operation circuit are changed over with the control signal to adjust the capacitance of the correction object capacitor. The correction object capacitor is actuated at a capacitance near design capacitance regardless of manufacturing characteristic variation of the correction object capacitor.

1. Field of the Invention
The present invention relates to an integrated circuit (hereinafter referred to as "IC") and more particularly to a circuit with variation correction function, which suppresses variation of output characteristic regardless of variation of characteristic of components due to manufacturing reason so as to obtain an output near a desired characteristic value. For example, preferably, the present invention is applied to such mobile communication device as a portable phone.
2. Description of Related Art
Recently, reduction of the quantity of required components has been demanded in many electronic device fields, which is represented by such a mobile communication device as a portable phone. For example, in case of the aforementioned mobile communication device, if its oscillation circuit is incorporated in an IC, this contributes to the reduction of the quantity of its components. FIG. 16 shows a voltage control oscillation circuit (hereinafter referred to as "VCO circuit") as an example of such an oscillation circuit. The VCO circuit of FIG. 16 contains a resonant circuit comprising an inductor section 101 and a capacitor section 102. Its oscillation frequency f is given by a following equation with inductance L of the inductor section 101 and capacitance C of the capacitor 102.
[Expression 1]
  f  =      1          2            π                    LC
The capacity of a variable capacitor in the capacitor section 102 can be changed by control voltage VT. Thus, the oscillation frequency f can be adjusted according to the control voltage VT. A phase lock loop (hereinafter referred to as "PLL") is constituted of such a VCO circuit. Upon use, the oscillation frequency f of the VCO circuit is made to coincide with a setting frequency of PLL.
However, if the aforementioned VCO circuit is incorporated in an IC, following problems occur. That is, characteristic values of incorporated elements vary unavoidably on manufacturing process of the IC. Consequently, the oscillation characteristic of the VCO circuit varies. This will be explained with reference to a graph of FIG. 17. This graph indicates an oscillation frequency f while the control voltage VT is raised from 0.5 V to 2.5 V. This graph further indicates five results that the capacitance C of the capacitor section 102 is finished to −20%, −10%, ±0%, +10%, +20%.
Here, it is assumed that the setting frequency of the PLL is set to 620 MHz. Then, a following matter becomes evident from the graph of FIG. 17. If this circuit is finished as its specification indicates or to ±0%, the control voltage VT for acquiring an output of the setting frequency is 1.6 V. Contrary to this, if the circuit is finished to −20%, the control voltage VT is 1.0 V. On the other hand, if it is finished to +20%, the control voltage VT in this case is 2.2 V, which is different from the preceding case by 1.2 V. The variation of the oscillation characteristic appears as a variation of the control voltage VT when the PLL is locked.
If the capacitance C of the capacitor section 102 is larger or smaller, the oscillation frequency band of the VCO circuit may be deflected completely from the lock range of the PLL. In this case, the PLL locking fails, so that the system cannot be used as desired. Further, even if the oscillation frequency band exists within the lock range, if the control voltage VT is extremely higher or lower than its normal value, the loop characteristic of the PLL is different from its ordinary characteristic. Thus, the phase noise characteristic may deteriorate or lock-up time until the oscillation frequency is locked to the setting frequency may take longer. In the meantime, a similar problem, which may be generated by manufacturing characteristic variation, exists not only in the capacitance of the capacitor device but also in other kinds of devices.

Circuit with variation correction function


Various pathophysiological states are determined by evaluating the rate of 17O-containing water production oxygen using proton MRI. The method can be used to identify viable but threatened myocardium, injured by viable muscle, penumbra tissue in a stroke patient, and the like.

One of the key items of information that physicians need to know for the selection and timing of medical treatment is the state of diseased tissue and the likelihood it will return to normal viability in response to therapy. Many methods are used to assess tissue health including the history of the patient's symptoms, physical examination, and laboratory and imaging studies of tissue biochemistry and anatomy. Abnormal anatomy, signal changes reflecting the state of tissue water and basic physiologic measurements such as water diffusion and blood flow are among the methods used. These methods are useful but are themselves secondary signs of more fundamental properties of tissue metabolism and potential. Magnetic resonance imagery (MRI) is one of the most frequently used imaging methods to assess tissue viability and although it is a highly sensitive and specific tool, it also depends primarily on secondary signs of tissue health to assess tissue viability.
The ability of MRI procedures to distinguish between subareas within a given tissue area is often limited. One example of this concerns the determination of penumbra in a stroke patient. The importance of this determination is apparent from the fact that there are an estimated 731,000 strokes and 4 million stroke survivors annually in the United States alone, making stroke a major cause of long-term disability. Considering also the mortality and morbidity caused from cardiac arrest, traumatic brain injury and perinatal asphyxia, it is easily understood that the economic and social burden of CNS injury is huge. Beyond the direct hospitalization and treatment costs, the indirect costs of lost productive years is of importance because ischemic stroke is prevalent in the older population and traumatic brain injury and perinatal asphyxia are diseases mainly of the young and newborns, respectively. Thus, indirect costs are even higher than direct costs for traumatic brain injury and perinatal asphyxia.
In the assessment of acute cerebral ischemic stroke (and cerebral tissue at risk for ischemic injury in chronic vascular stenosis), a key item of information needed for therapy decisions is an accurate differential identification of normal tissue, tissue injured beyond recovery (the ischemic infarct "core") and tissue at risk for permanent injury but potentially salvageable with therapy (the ischemic infarct "penumbra"). The infarct core is the zone of absent cerebral blood flow (CBF) or low CBF with tissue metabolism below a viable threshold (CMRO2), usually identified on MRI as a zone of severely restricted diffusion indicating the failure of cellular energy metabolism. The penumbra, tissue that remains viable but has reduced blood flow and is at high risk of energy failure and cell death, has two zones. The region of the penumbra called the "oligemic" zone is where prolonged reduced blood flow may be compatible with tissue survival and therapy may only be needed to prevent further flow reduction. Thus, the compensation for early or mild reduced blood flow in stroke (Stage 1) takes the form of vasodilation and increased coronary blood volume which maintains blood flow sufficiently to allow normal cellular metabolism to be maintained at a normal oxygen extraction fraction (OEF). With more severe blood flow reduction (Stage II), the vasodilation capacity is exceeded, normal flow cannot be maintained and OEF is increased because more oxygen must be extracted per volume of blood to maintain normal cellular metabolism. This is the true ischemic penumbra and is also called the zone of "misery perfusion". This tissue will not survive without treatment to improve blood flow. The end point of this trend is extreme or prolonged blood flow reduction leading to failure of cellular energy metabolism, apoptosis, cell death and reduction of OEF below normal in response to the absence of oxygen demand. "Necrotic" cell death occurs because the mitochondria are incapable of maintaining ATP production with inadequate oxygen delivery. This "oxidative stress" also activates a mitochondrial trigger for "apopotic" cell death, or programmed cell death mediated by a cascade of protein activation and DNA injury. Similar principles apply to the assessment of tissue at risk in chronic vascular stenosis where elevated OEF identifies tissue at the highest risk of developing infarction.
When normal perfusion pressure is not maintained, reflex vasodilation occurs to maintain normal blood flow. This response, as well as the reflex vasoconstriction observed with increased perfusion pressure, is known as autoregulation or Stage 1 hemodynamic compromise. It maintains normal flow by reducing the vascular resistance to arterial inflow. With further reductions in perfusion pressure, the capacity of autoregulatory vasodilation to maintain normal blood flow is overcome and blood flow begins to decrease. Although the delivery of oxygen falls, the brain can increase the amount of oxygen it extracts from the blood (the oxygen extraction fraction or OEF) to maintain normal cerebral oxygen metabolism and function. This phenomenon of reduced blood flow and increased oxygen extraction is the reason for "misery perfusion" or Stage 2 hemodynamic failure. Once oxygen extraction becomes maximal, further decreases in perfusion pressure (and consequently blood flow) will lead to disruption in normal oxygen metabolism and ultimately to infarction.
At present, regional measurements of OEF can be made only with positron emission tomography (PET) using O-15 labeled radiotracers. Both absolute values and side-to-side ratios of quantitative and relative OEF have been used for the determination of abnormal from normal. MRI measurements using pulse sequences sensitive to deoxy-hemoglobin, which is increased in regions with increased oxygen extraction, are just beginning to be be developed to provide similar information.
The present invention is based on a recognition that tissue blood flow, molecular oxygen delivery, oxygen metabolic rate and oxygen extraction fraction can all be measured with a form of MRI which employs 17O2. While all of these parameters are important for the assessment of the metabolic state of tissue, the oxygen extraction fraction (OEF) is the variable which is the most sensitive and specific for tissue in a state of "oxidative stress" and so is the key predictor of tissue viability because total oxygen consumption may appear to be constant due to the tissue's ability to compensate for locally reduced oxygen delivery. Reductions of blood flow, oxygen delivery and oxygen metabolic rate below normal levels do not specifically indicate whether the oxygen supply is deficient to tissue with normal metabolic demand (ischemia or hypoxemia) or appropriate for tissue with a reduced metabolic demand (metabolically suppressed, stunned or hibernating or dead tissue). Elevation of the OEF specifically indicates that there is a tissue metabolic demand for oxygen that is not being adequately met by the oxygen supply for the blood. More precisely, there is an increased gradient along which oxygen is diffusing between the higher concentration in blood and the lower concentration in tissue, produced by the rate of oxygen consumption in tissue mitochondria. This increased concentration gradient induces a greater percentage of blood oxygen to be released from hemoglobin and transported to the tissue. OEF is not elevated, but is reduced in metabolically suppressed or dead tissue because the blood-tissue oxygen concentration gradient is minimized. The prediction of tissue viability by OEF is based on the frequent observation that substantial, prolonged OEF elevation directly precedes cell death in a variety of tissues. The oxidative stress indicated by elevated OEF leads to cell death by necrosis from mitochondrial energy depletion (failure of ATP production) and/or mitochondrial triggering of the molecular cascade leading to apoptosis, or "programmed cell death".
The quantitative measure of oxygen removal from blood, the oxygen extraction fraction (OEF), is the percent of oxygen removed per unit volume of blood, caused by oxygen diffusion along the gradient from the high concentration in the blood to the low concentration in the metabolizing tissue. Normal brain tissue OEF is approximately 40% and may be as high as 60-70% in the subarea of penumbra misery perfusion. In acute ischemia, tissue with elevated OEF is likely to progress to cell death unless normal blood flow is restored (e.g. by thrombolysis) or metabolic demand is reduced (e.g. by hypothermia). In transient ischemic attack (TIA) or chronic vascular stenosis, tissue with elevated OEF is at risk for permanent injury if blood flow is reduced below the marginal level maintaining tissue viability.
It is important to initiate acute therapy promptly to prevent penumbra tissue from progressing to infarction and thereby preserve, at least to some degree, the normal functioning of this brain tissue. These acute therapies have no effect on the ischemic core or infarction but involves a high degree of risk. For example, anti-coagulant therapy can be dangerous to the patient in that normal coagulation is suppressed. Because of the possible adverse effects, the treating physician must evaluate the risks and benefits of the therapy, an evaluation which includes consideration of the likelihood of converting the penumbra tissue into "normal" tissue. This requires estimating how much true ischemic penumbra tissue exists.
In the absence of other factors, one of the principle items of information on which the treating physician makes an evaluation of the risk and benefits is the amount of time since the stroke apparently occurred. Since the progression from penumbra to infarction is relatively rapid, usually measured in terms of hours, and there is a lack of precession about when the cerebral schema event commenced, the treating physician is essentially forced to make a risk/benefit evaluation based on the limited information which may be available. A diagnostic procedure which will assist the physician in making an evaluation is clearly desirable. Positron emission tomography (PET) has been found to provide detailed insight into the changes which occur after a cerebral ischemia. PET detects viable but hyperperfused tissue that is characterized by an elevated OEF and accessible to acute therapy. It is thus capable of providing a relatively accurate identification of the true ischemic penumbra tissue. However, PET involves the use of a radioactive isotope of oxygen, 15O, and a substantial radiation dose. Consequently, the substitution of data from stroke magnetic resonance imagery induced markers has been recognized as desirable. But stroke MRI induced markers for tissue integrity or cerebral profusion do not satisfactorily identify the penumbra.
Currently, the most frequently used MRI methods for evaluating the health of brain tissue and the risk of infarction are bolus-contrast perfusion-weighted imaging (PWI) and diffusion-weighted imaging (DWI). PWI is performed by injecting a rapid bolus of gadolinium-chelate contrast agent intravenously while monitoring its passage through the brain circulation with rapid MR imaging, at approximately 1 image per second. Spatial maps of cerebral blood volume (CBV), tissue circulation mean transit time (MTT) and cerebral blood flow (CBF) are calculated using the central volume principle. These maps are compared to the diffusion weighted images which show infarcted tissue as high signal, representing restricted diffusion in cells that have undergone complete failure of energy metabolism. When this comparison shows areas of reduced perfusion outside the restricted diffusion infarction, a PWI-DWI "mismatch" is present and is considered a surrogate marker of the metabolic penumbra.
The major limitation of the PWI-DWI mismatch assessment of tissue at risk is that it is based on indirect measures of the tissue metabolic status. The practical effect of this limitation was studied in order to evaluate the accuracy of using mismatch to indicate penumbra and the study found that mismatch overestimated the volume of penumbra, and therefore the tissue at risk which was subject to acute therapy. See Sobesky et. al., Does the Mismatch Match the Penumbra?, Stroke, 2005; 36:980-985. Accordingly, mismatch as determined by MRI procedures provides inaccurate information to the treating physician. But an MRI procedure has the advantages over PET of an absence of ionizing radiation exposure to the patient and medical personnel, and the absence of a need for an expensive, complex, on-site cyclotron and radiation chemistry lab. An MRI procedure which more accurately identifies penumbra information for the treating physician that mismatch is clearly desirable.
The present invention is based, in part, on the recognition that the mismatch zone includes a mix of tissue with reduced flow and normal OEF ("oligemic" perfusion) and tissue with elevated OEF ("misery" perfusion). In the oligemic perfusion zone, cellular oxygen demand is close to the oxygen level supplied by blood flow and the tissue is likely to remain viable with continued low perfusion. In the misery perfusion zone, cellular oxygen demand is well above the oxygen level supplied by blood and the tissue is likely to progress to infarction unless perfusion is improved or returned to normal. It is also based, in part, on the recognition that the PET analysis is not effected by the oxygen uptake compensation by the tissue in the same way that MRI mismatch analysis is effected. It also takes advantage of the stable nature of the 17O2 isotope compared to the 2 minute half-life decay of 15O2 which will allow distribution of an 17O2 viability agent in an "off the shelf" form to the large installed base of MRI scanners now in operation.
It is the object of this invention to provide a new method for differentiating or monitoring tissue response to stress, such as for instance, determining penumbra brain tissue in a stroke patient, using proton magnetic resonance imaging. This and other objects of the invention will become apparent from the following detailed description.