An ultrasound inspection roller provided with a wheel, a sensing system and a support for holding the sensing system inside the wheel, a wedge connected to the ultrasound sensing system at one end and provided with a curved profile at its other end facing the wheel, adapted to the curvilinear shape of the wheel, and a liquid of a density higher than 9.9*102 kg/mm3 placed inside the wheel such that the sensory system, the wheel and the wedge are acoustically coupled. The roller allows the early detection of problems during manufacturing of composites and the performance of corrective measures in real time, and assures a good coupling between the transducers and the material to be inspected in dry conditions.

The present invention is related to non-destructive inspection techniques to perform real time quality control of, in particular, but not limited to, fiber composite material manufacturing processes, out-of-autoclave curing processes and conventional processes such as autoclave. More in particular, the invention relates to a new technique based on an ultrasonic inspection system.
Most composite material parts manufactured today for the aeronautics industry are traditionally made according to autoclave curing processes. The main weaknesses of this technology include:
The inevitable heavy investment not only due to all the necessary equipment but also due to all the infrastructure associated with the autoclave process (space necessary, installation, tools, auxiliary materials, power consumption, maintenance, etc.);
The need of having a large enough autoclave to house the part to be manufactured, which clearly imposes a limitation on the design of the parts to be manufactured, reducing manufacturing flexibility; and
The reduced process energy yield making the composite material parts manufactured according to this curing technology more expensive.
Among the options considered to overcome these drawbacks would be the use of alternative technologies for curing/reticulating the resin without the need of autoclave. Some of these technologies are electron beam curing, microwave curing, radiofrequency curing and induction curing. Each of these technologies has its advantages and drawbacks as well as its degree of maturity for different applications, and all of them have in common the elimination of the autoclave for the final curing of the part. Based on the need for a non-destructive inspection of the composite parts, and particularly parts manufactured according to the new processes, it is obvious that these new processes must be provided with a solution/an inspection technology which is capable of being adapted to the specific requirements of these processes, but which in turn maintain at least the capacities of the current systems for inspecting autoclave manufactured parts. The current non-destructive ultrasound testing techniques (NDT) perform quality control on the part once the part has been made and the material is completely cured.
NDTs have been used for decades to detect material and structural defects. The basic idea is simple: an electronic system excites an ultrasonic transducer generating a high frequency mechanical wave pulse which is introduced and propagates in the material to be inspected. When the ultrasonic pulse finds a heterogeneity inside the material, part of the incident energy is reflected in the form of an echo which returns to the transducer, where the vibrations are converted again into electrical signals. After amplification, the echoes indicate the presence of a defect, its position and size estimation.
These techniques have been greatly developed and have many applications in different industrial areas. They are particularly indispensable in the fields of aeronautics, energy generation, oil prospecting, rail transport and public works in which the verification of structure and part safety margins is an obligation and is performed with NDT techniques. Likewise, most manufacturing industries (metallurgy, automobile, machine-tool, etc.) use NDT techniques to assure the quality of their products.
The NDT systems have often been automated by associating them with robots that mechanically scan the parts to assure the structural integrity throughout their dimension.
Technology advancements over the past few years in the field of electronics and ultrasound transduction technologies have allowed performing inspections with a never before seen precision and speed. Currently the state of the art of the technology allows constructing very complex electronic systems from tens to several hundreds of channels, this is possible as a result of the mass integration of hardware processing functions provided by the technology.
The ultrasonic array-based technology facilitates arbitrary deflection and focusing of the ultrasonic beam in emission and reception without the need of moving or altering the transducer by means of an electronic control. The composition of the traces received allows constructing 2D and 3D images of the inside of the inspected materials. In fact, ultrasound has been used in medicine for almost three centuries with remarkable success.
Since the market appearance of the first successful system of NDT with arrays [FOCUS-TomoScan©, by Rd Tech, Canada] in 1996, this technology is being demanded by a growing number of users in different industrial sectors since the advantages it provides compensate its high cost.
One of the problems limiting the applicability of the inspection by means of ultrasounds on carbon fiber parts is the need of coupling the transducers to the materials. Due to the large acoustic impedance differences, great insertion losses occur between emission and reception if a suitable coupling means is not available. This is solved by, for instance, performing complete or partial immersions using water jets or micro-troughs, which is a cumbersome procedure, as in EP1914544A2, or need water films in contact with the material to be inspected to ultrasonically couple the transducer with such material.

Ultrasound inspection system and ultrasonic quality control method


The present invention discloses an overlay alignment measurement apparatus and method. The overlay target is periodic and is illuminated by coherent radiation; a Fourier transform lens optically computes the Fourier transform of the target. Analysis of the spatial frequencies at the Fourier plane yields overlay alignment information.

1. Field of the Invention
This invention relates to a method and apparatus for overlay alignment measurement during various stages of integrated circuit processing for microelectronic fabrication.
2. Description of Related Art Including Information Disclosed Under 37 CFR 1.97 and 1.98
Planar technology that is conventionally used in high volume production of integrated circuits used in computers, communication equipment, cellular phones and other consumer electronic devices, relies on successive processing of materials deposited in the form of thin films on top of silicon or other substrates. If two successive layers are misaligned with respect to each other, the device performance degrades and it may fail. For example, without any limitations, a misalignment can lead to an increase in the resistance of via's and contacts and hence an increase in the RC time delay of the circuit, leading to a reduction in the operable speed of the device. It is of great interest to measure the alignment accuracy of one layer's placement or registration with respect to a preceding layer's position with high precision and repeatability and with good matching between metrology tools. "Perfect alignment" rarely happens; every integrated circuit design allows for some misalignment between two succeeding layers; it is of great importance to know how well two layers are aligned with regard to each other and to know whether or not the "misalignment" is outside a prescribed tolerance band. Overlay error is the mis-registration in placement of a layer above a previously processed layer. As used herein "overlay error" or "alignment error" is defined as the deviation from perfect alignment in the x and the y direction as noted by Δx or Δy; when rotational misalignment is present, Δx or Δy is a function of the exact location of the measurement.
Overlay error measurement is normally done with specially designed targets (or marks) placed at pre-selected locations on the wafer. These targets are typically several tens of microns large and fit within the scribe or kerf area between rows and columns of die. Traditionally overlay targets consisted of box-in-box or bar-in-bar structures. However, recently, grating-based overlay metrology techniques have emerged which require targets periodic in nature. These gratings can be one-dimensional (i.e. rows of lines) or two-dimensional, for example, rows or columns of squares, rectangles, triangles, parallelograms, circles, trapezoids or any other shape that is lithographically printable. U.S. Pat. No. 6,985,618 describes a wide variety of overlay targets. Typically, a grating is placed on the previous layer and another grating on the current layer that is being processed. Normally a grating placed on a previous layer is fully processed (i.e. etched or filled, etc.) and the grating on the current layer is in a photo-resist layer. An overlay metrology tool measures a deviation in placement of a grating in a photo-resist layer with respect to a grating in the processed layer. "Overlay error" is defined as the deviation from ideal, rotational and translational, of the layer being processed to a layer previously processed.
In an overlay metrology tool described in U.S. Pat. No. 6,023,338, issued to Bareket, the system requires mechanical scanning of a focused spot on the gratings. In U.S. Pat. No. 6,710,876, issued to Nikoonahad et al, optical phase of the first order diffracted beams is used to extract the overlay error. In U.S. Pat. No. 7,009,704 issued to Nikoonahad et al the overlay target is imaged with a CCD using a mid numerical aperture (NA) lens and the shift between the adjacent gratings are computed by software. Mechanical systems suffer from wear and tear, friction, instability and drift over time leading to unreliable overlay error measurement. With reference to the optical phase techniques as described in U.S. Pat. No. 6,710,876, the information content in the first diffracted orders is insufficient and it does not adequately address the overlay measurement needs. Also, measuring the optical phase from first order diffraction alone suffers from complications in separating large phase shift arising from the height of the resist layer and the periodic phase shifts embedded in the phase of the reflection coefficient of the previously processed layer. The use of linear mechanical translation of a Wallaston prism or the use of an acousto-optic deflector has been proposed but these add to the complexity of the system. The overlay error control necessary for sub-100 nm integrated circuit devices is less than about five percent of the minimum feature size with a 3σ precision of less than one percent. This resolution is certainly below the imaging capability of any optical imaging system and, as a result, current imaging systems, as described in 7,009,704, resort to a significant amount of algorithmic post processing to attain meaningful overlay information. None of the techniques described above are fully satisfactory. Clearly additional sensitivity to "overlay-only" errors is of great value. It is, therefore, desirable to develop a system with better performance and a simpler characteristic.

Overlay error measurement using fourier optics


A linear drive recumbent cycle (10) having linearly reciprocating pedals (68a,b) located approximately above the front wheel (14) in order to achieve a short wheel base recumbent bicycle design with the crank (42) located low between the wheels and with the seat located near or behind the center of gravity of the bike. The top extremities of a pair of pedal sliders (32a,b) are journalled for independent pendulum-type oscillation in a pedal-slider bracket (28) which is connected to a frame (12) through a front tube (30). The linearly reciprocating pedals (68a,b) are attached to the lower extremity of the pedal-sliders (32a,b) and are rotateably linked to the crank (42) through a pair of curved links (34a,b) so as to transfer power from the pedals (68a,b) to the crank (42). The pedal-sliders are short and straight whereby saving weight. The axis of reciprocative movement of the pedal-sliders (32a,b) cross above the axis of the crank (42) giving it an inherent tendency to rotate forward whereby eliminating crank kick back or crank getting stuck during use.

1. Field of Invention
This invention relates to cycles, such as a bicycle, tricycle, motorcycle. More specifically, this invention relates to a recumbent bicycle. Even more specifically, this invention relates to a recumbent bicycle that uses a linearly drive to convert rotary pedal motion to linearly reciprocating pedal motion and with pedals positioned approximately above and on either side of the front wheel to create a short wheel base recumbent bicycle with the seat at or behind the center of gravity for stability. Even more particularly, this invention relates to a recumbent bicycle with an uniquely aligned linear drive that will eliminate crank kickbacks or crank getting stuck during use for a smooth operating linear drive recumbent bicycle. 2. Description of Prior Art
Heretofore, recumbent bicycles are known in prior art and one typical embodiment of bicycles of this type is disclosed in U.S. Pat. Nos. 4,878,684 (E. W. Lemle, November 1989). As is clear from the prior art, it is desirable to construct a recumbent bicycle with an adjustable seat whose position is low with respect to the ground and with pedals located in front of the seat. Both seat and pedals are positioned along the frame of the bicycle. However, recumbent bicycle as illustrated in the foregoing reference have a wheel base that is substantially longer than conventional upright bicycles. Its subsequent weight and length make them slow and hard to maneuverable in tight places. Therefore, it is desirable that the wheel base of a recumbent bicycle be much shorter. To achieve a short wheel base, it is desirable for recumbent bicycles to have linearly reciprocating pedals of my invention so that the crank can remain low between the two wheels and not have to be moved very high up above the front wheel or moved in front of the front wheel.
The closest series of prior arts are disclosed in the SAE Technical paper Series 840021 called the "Evolution of Recumbent Bicycles and the Design of the Avatar Bluebell" by Dr. David Gordon Wilson of the Massachusetts Institute of Technology. The MacMillan Velocipede of about 1839 shown in FIG. 1 on page 1 uses a linear drive but the pedal-sliders interfere with the higher amplitudes of front-wheel steering movements. Also, its crank was connected directly to the axle giving it only one speed, thus making it too strenuous to ride. So, my invention modernizes the structure of the MacMillan Velocipede and made it work.
The second diagram of FIG. 15 on page 6 of the aforementioned technical paper shows a linear drive mechanism very similar to both my invention and the MacMillan Velocipede but differs in structure and have many disadvantages. And the author mentioned the disadvantages of his invention in the paragraph above and below the diagram. First, his pedal-sliders had to be long and curved to fulfill its intended purpose of shortening the wheel base. Since the pedal-sliders have to be strong enough to withstand high pedaling forces, his pedal-sliders design "weight far more . . . than the pedal that it replaced". This is due to the way that the pedal-sliders are attached to the frame. The advantage of my invention is that it allows the use of straight and much shorter pedal-sliders to cut weight. Further, the pedal-sliders of my invention can also be moved forward without extending the length of the pedal-sliders as in this prior art. This allows the accommodation of very long legged people.
Although the first paragraph on page 7 of the prior art had mentioned that the pedal's linear motion axis does not pass through the crank axis giving it a "quick-return motion". But it did not disclose in detail what it means or perhaps did not further investigate its unobvious advantage. The unobvious advantage of not having the pedal's linear motion axis pass through the axis of the crank had been known in piston engine design. So, this prior art did not invent this. And no other prior art explains this principle which prevents the bicycle crank from kicking back or getting stuck when used with a linear drive mechanism.
Therefore, the prior art of the above two paragraph had a structure that is not practical. In addition, it did not describe how and why will the advantage of not having the pedal's linear motion axis passing through the axis of the crank benefit it.


A raw material is dispersed around a rotating separator main body having a plurality of classifying blades annularly arranged on the main body and is exposed on a rising flow of classifying air so as to sift off crude powder which cannot be entrained on the flow of the classifying air. The remaining raw material entrained on the classifying air is peripherally introduced into the separator main body for classification into fine and crude powders, the fine power being the powder which is allowed to pass through the classifying blades while the crude powder is the powder which is not allowed to pass. A dispersion plate is arranged between the separator main body and a raw material inlet above the separator main body to receive and disperse the raw material from the inlet. The dispersion plate is rotatable at number of revolutions different from that of the separator main body. Thus, the dispersion plate can be set at proper number of revolutions independently of number of revolutions of the separator main body required for classification.

The present invention relates to a classifier used in cement manufacturing facilities and the like.
FIGS. 1 and 2 show a conventional classifier. In the figures, reference numeral 1 denotes a classifier casing which comprises an upper cylindrical casing section 4 having a raw material inlet 3 for receiving a raw material 2 in a position on an upper surface of the section 4 and a lower funnel-shaped casing section 5 contiguous with a lower portion of the section 4 and having its diameter gradually reduced downwardly.
A vortex chamber 7 with a plurality of louvers 6 annularly arranged on an inner periphery of the chamber 7 is formed on and is protruded outwardly from a lower portion of the lower casing section 5. A classifying air introduction pipe 8 opened to the chamber 7 is arranged tangentially on an outer periphery of the chamber 7 (see FIG. 2).
Under the vortex chamber 7, the lower casing section 5 has a lower crude powder outlet 10 for discharging crude powder 9 falling along an inner surface of the lower casing section 5.
Reference numeral 11 designates a separator main body which is driven by a drive 12 on the upper surface of the upper casing section 4 and which is supported from above by a rotary shaft 13 rotatably supported substantially at a center of the section 4. The separator main body 11 comprises a mounting plate 15 fixed to a lower end of the rotary shaft 13, a dispersion plate 16 fixed to an upper portion of the shaft 13 to receive the raw material 2 from the inlet 3 and classifying blades 17 arranged peripherally along an outer periphery of the plate 15 and fixed between the plates 15 and 16.
Reference numeral 18 represents a fine powder hopper arranged under the separator main body 11. The fine powder hopper 18 comprises a funnel-shaped hopper main body 19 having its diameter gradually reduced downwardly and an exhaust outlet 22 extending from a lower end of the hopper main body 19 through the lower casing section 5 to outside of the section 5 so that fine powder 20 introduced via the classifying blades 17 into the separator main body 11 can be discharged outside together with classifying air 21.
When the drive 12 is actuated to rotate the separator main body 11 via the rotary shaft 13 and at the same time the classifying air 21 is introduced from the introduction pipe 8 into the vortex chamber 7 and the raw material 2 is charged by a predetermined quantity into the casing 1 through the inlet 3 on the upper surface of the casing 1, the raw material 2 thus charged falls onto the rotating dispersion plate 16 so that, with centrifugal force being given to the raw material 2, the raw material 2 is dispersed peripherally outwardly of the plate 16.
In this case, the classifying air 21 from the introduction pipe 8 into the vortex chamber 7 is given swirling force by the chamber 7 and flows into the casing 1 through the louvers 6. Thus, a flow of classifying air 21 is provided which swirls up along the inner surface of the lower casing section 5 and flows into the rotating separator main body 11 through the classifying blades 17, so that the raw material 2 falling from above onto the dispersion plate 16 is dispersed into the flow of the classifying air 21.
The raw material 2 entrained on the flow of the classifying air 21 is classified into fine and crude powders 20 and 9, the fine powder 20 being the powder which is allowed to pass through the classifying blades 17 while the crude powder 9 is the powder which is not allowed to pass. Only the fine powder 20 allowed to pass through the classifying blades 17 is guided into the separator main body 11, flows down into the hopper main body 19 of the fine powder hopper 18 and is discharged outside together with the classifying air 21 through the exhaust outlet 22.
On the other hand, the crude powder 9 which is not allowed to pass through the classifying blades 17 and is flapped onto the inner surface of the casing 1 as well as the crude powder 9 which cannot be entrained on the flow of the classifying air 21 from the beginning are away from the flow of the classifying air 21, fall down along the inner surface of the casing 1 and are discharged outside through the lower outlet 10.
In such classifier, number of revolutions of the separator main body 11 required for classification of the raw material 2 will depend on particle size distribution of the raw material 2 to be classified and is usually in a range of 100 to 1000 r.p.m. Number of revolutions of the dispersion plate 16 required for dispersion of the raw material 2 will depend on diameter of the dispersion plate 16 and amount of raw material to be supplied and is usually on the order of 10 to 100 r.p.m.
However, in the conventional classifier as shown in FIGS. 1 and 2, the dispersion plate 16, which is integrated with the separator main body 1, is rotated at the rotating speed of the separator main body 11 required for classification. As a result, the raw material which falls onto the dispersion plate 16 is scattered with centrifugal force higher than required; therefore, most of the raw material 2, which is to be entrained on the flow of the classifying air 21 and transported to the classifying blades 17 of the separator main body 11, will violently strike on and fall along the inner surface of the casing 1 without fully entrained on the flow of the classifying air 21, resulting in discharge of most of the powder as crude powder 9 without having an enough chance for classification, which disadvantageously leads to lowering of classifying efficiency. Because more quantity of powder strikes on and falls along the inner surface of the casing 1, the inner surface of the casing 1 tends to be worn out.
The present invention was made in view of the above and has its object to provide a classifier capable of preventing a centrifugal force more than required from being given to raw material to be dispersed by a dispersion plate.

Classifier having a rotatable dispersion plate

Ishikawajima-Harima Jukogyo Kabushiki Kaisha


A state machine representation is provided that includes a plurality of nodes that are assigned a unique node identifier and an output value, and that are grouped into color segments. The nodes are interconnected by transitional vectors, with each transitional vector being assigned a unique value. Further, at least one of the nodes has a termination vector that leads to a termination point. To encrypt the set of data values, a start node is identified and any path of transitional vectors is be traversed from node to node. As each transitional vector is traversed, its corresponding unique value will be recorded in sequence. When a node is reached that has an output value that matches one of the set of data values, an invalid unique value is designated and recorded in sequence with the unique values.

As the use of computer technology grows, security in message transmission is becoming an increasing concern. Specifically, today many businesses and governments transmit information over computer networks such as the Internet. In many cases, the information transmitted is of a secretive nature. For example, a business might need to transmit valuable trade secret information from one facility to another. Further, governments might need to transmit sensitive military or strategic information to allies and forces abroad. In any of these cases, the secretive nature of the information must never be compromised.
To date, many systems for encrypting or otherwise securing data have been developed. One common methodology is to use a security key that is a pseudo-random number. The security key is used as an exclusive-OR (XOR) value against bytes in the data. Typically, the pseudo-random number is generated using a method known to both the sender and the intended recipient of the data. Once encrypted, the data is transmitted to the recipient who can then decrypt the data using a "seed" value and the pseudo-random number. Generally, a pseudo-random number generator on the sender side generates a new random number for each sequence of bytes to be encrypted. A pseudo-random number generator on the recipient side will generate the same sequence of pseudo-random numbers to decrypt the data. In methods such as these, there is a one to one correspondence of encrypted bytes to source bytes.
Typically, methods such as these creates various security risks. For example, the encryption can be broken through trial and error. Specifically, security under these methods relies upon the complexity of the pseudo-random number (i.e., the "mask"). The more bits the mask has, the longer it will take to decrypt the message. However, given the advanced nature of today's computer technology, any encrypted data set (regardless of bit quantity in the mask) can be decrypted through trial and error in a finite amount of time. In an attempt to address these issues, the above-incorporated patent application describes a state machine representation for encrypting and decrypting a message. The described state machine representation typically includes a set of nodes that are interconnected by transitional vectors, with each transitional vector being assigned a unique value. Some or all of the nodes in the state machine representation are assigned data values from the message, with each data value being assigned to at least one of the nodes. To encrypt the message, a start node will be identified and any path of transitional vectors will be traversed from node to node. As each transitional vector is traversed, its corresponding unique number will be recorded in sequence. When one of the data values in the message to be encrypted is encountered/reached at a node, a random number that is not representative of any of the transitional vectors is generated and recorded (in sequence with the unique values). This process continues until all of the data values in the message have been encountered/reached. Once complete, the recorded string of unique values and random values is outputted to a recipient who has access to a copy of the state machine representation. Using the string of values, the recipient can follow the sequence/path of transitional vectors traversed in encrypting the message to recover the data values.
As effective as the state machine representation of the incorporated patent application can be, various security risks can still be posed. Specifically, since both the encrypting party and the decrypting party must have access to the state machine representation, the state machine representation is typically distributed to both parties. During such distribution, it is possible for an intruder or other unauthorized party to intercept the state machine representation. If the intruder is also able to intercept the string of values, he/she could decrypt the message.
In view of the foregoing, there exists a need for an improved state machine representation for encrypting and decrypting a set of data values, and a method for protecting the same. In addition, a need exists for a method, system and program product for encrypting and decrypting a set of data values using the improved state machine representation. To this extent, a need exists for nodes of the state machine representation to be grouped into distinct color segments, with no two directly connected nodes being in the same color segment. A further need exists for one or more of the nodes to have a termination (transitional) vector that leads to a termination point instead of to another node.

State machine representation for encrypting and decrypting a set of data values and method for protecting the same

International Business Machines Corporation


An antenna assembly including a resonator element which is operatively connected to the ground plane of a wireless communication device (WCD) is described. The resonator element comprises a flexible resonator support substrate of dielectric material which supports a conductive element or portion. The flexible nature of the substrate facilitates coupling to the ground plane of a WCD at a variety of locations including the interior, the exterior, or within a portion of the housing of the WCD itself. The resonator element may be provided with a dielectric support element which operatively connects the resonator element to the ground plane of a printed wiring board of a WCD. The support element includes a predetermined support edge portion which helps maintain the resonator element in a preferred predetermined shape, and a bottom edge with a conductive strip portion which is used to operatively connect the support element to a ground plane. The resonator element includes a conductive portion which may take several embodiments. In one preferred embodiment, the conductive portion is a wire member which is coiled about the flexible resonator support substrate. In another preferred embodiment, the conductive portion includes a conductive channel-shaped layer and at least one trace or line spanning the sides of the channel-shaped layer. In another preferred embodiment, the conductive portion includes a conductive layer deposited on the flexible resonator support substrate in an array pattern. All of the preferred resonator embodiments include a discrete electrical connection location which is operatively connected to separate signal and ground lines, and whose position may be varied depending upon the frequency ranges and performance requirements.

The present invention finds primary utility in the field of wireless communications. More particularly, the present invention relates to an antenna assembly suitable for wireless transmission of analog and/or digital data in a single or multiple frequency band antenna system.
A variety of prior art antennas are currently used in wireless communication devices. One type of antenna is an external half wave single or multi-band dipole. This antenna typically extends or is extensible from the body of a wireless communication device (WCD) in a linear fashion. While this type of antenna is acceptable for use in conjunction with some WCDs, several drawbacks impede greater acceptance and use of such external half wave single or multi-band dipole antennas. One significant drawback is that the antenna is typically mounted at least partially external to the body of a WCD which places the antenna in an exposed position where it may be accidentally or deliberately damaged, bent, broken, or contaminated.
Furthermore, due to the physical configuration of this class of omni-directional antenna, optimizing performance for a particular polarization and/or directional signal is not an option. That is, these types of prior art antennas are relatively insensitive to directional signal optimization or, said another way, these types of prior art antennas can operate in a variety of positions relative to a source signal without substantial signal degradation. This performance characteristic is often known as an xe2x80x9comni-directionalxe2x80x9d quality, or characteristic, of signal receipt and transmission. This means that electromagnetic waves radiate substantially equally in all directions during transmitting operations. Such prior art antennas also are substantially equally sensitive to receiving signals from any given direction (assuming adequate signal strength). Unfortunately, for a hand held WCD utilizing such a prior art antenna, the antenna radiates electromagnetic radiation toward a human user of the WCD equipped with such an antenna as there is essentially no front-to-back ratio. For reference, the applicant notes that for multi-band versions of prior art types of antenna, the external half wave single or multi-band dipole antenna (i.e., where resonances are achieved through the use of inductor-capacitor (LC) traps), signal gain on the order of approximately a positive two decibels (+2 dBi) are common and expected.
In addition, due mainly to the inherent shape of such prior art antennas, when operating they are typically primarily sensitive to receiving (and sending) vertical polarization communication signals and may not adequately respond to communication signals that suffer from polarization rotation due to the effects of passive reflection of the communication signals between source and receiver equipment. Furthermore, such prior art antennas are inherently inadequate in sensitivity to horizontal polarization communication signals.
Another type of prior art antenna useful with portable wireless communication gear is an external quarter wave single or multi-band asymmetric wire dipole. This type of antenna operates much like the aforementioned external half-wavelength dipole antenna but requires an additional quarter wave conductor to produce additional resonances and, significantly, suffers the same drawbacks as the aforementioned half wave single band, or multi-band, dipole antenna.
Therefore, the inventor recognizes and addresses herein a need in the art of WCD antenna design for an antenna assembly which is compact and lightweight, that is less prone to breakage and has no moving parts (which may fail, become bent, and/or misaligned), and, which utilizes the available interior spaces and structure of a WCD to achieve a more compact final configuration.
There is also a need for a multi-frequency antenna assembly which is able to receive and transmit electromagnetic radiation at one or more preselected operational frequencies.
There is also a need in the art for a deformable antenna resonator which is equally responsive to a variety of different communication signals having a variety of polarization orientations.
There also exists a need in the art for an antenna assembly which is compact and lightweight and which can receive and transmit electromagnetic signals at one or more discrete frequencies and which antenna assembly can be tuned to one or more frequencies.
The invention herein taught, fully enabled, described and illustrated in detail herein is a multiple band antenna assembly for use in a wireless communication device (WCD) which meets the shortcomings of the prior art. The inventive antenna assembly of the present invention includes a deformable resonator element disposed on a dielectric resonator support substrate and operatively electrically connected to both an RF signal line and to a ground plane associated with a WCD. The resonator element comprises a substrate which supports a conductive element or portion.
The deformable substrate of the resonator element is preferably sufficiently flexible to permit fabrication of a variety of antenna shapes and configurations depending on the available space within a WCD. The flexibility of the substrate allows for a variety of shapes for the resonator element to be coupled to WCDs at a variety of locations with respect to the WCD, including discrete single or multiple locations disposed in the interior, the exterior, and/or located at discrete locations along the periphery of electronics disposed within a portion of the housing of the WCD. Preferably, the resonator element is curved or arcuately shaped, however, other configurations are possible and clearly within the purview of those skilled in the art to which the present invention is directed.
The resonator element also includes a conductive portion which may take several forms in different embodiments of the present invention. In one preferred embodiment, the conductive portion is a wire member which is coiled about the flexible resonator support substrate. In another embodiment, the conductive portion includes at least one trace of electrically conducting material spanning the resonator element and contacting a conductive layer. In another preferred embodiment, the conductive portion includes an array of deposited conductive material in contact with a continuous conductive layer. All of the preferred resonator embodiments of the present invention include a discrete electrical connection location which is operatively coupled to separate signal and ground lines. The position of the discrete electrical connection location may be varied depending upon the frequency ranges and performance requirements for a given application or a particular configuration or style of WCD.
The resonator element is preferably provided with a generally planar bridge, or support, element which mechanically supports and electrically couples the resonator element to the ground plane of reduced electrical potential preferably disposed on or in a printed wiring board of a WCD. The bridge or support element is formed of dielectric material and includes a first edge portion which helps support and maintain the resonator element in a desired, preferably arcuate, configuration. In one embodiment, the support element also has an edge portion with a conductive strip portion which is used to operatively connect the support element to a ground plane.
The antenna assembly comprises a dielectric resonator support element and an electrically conducting resonator element and an electrical connector element electrically coupling the resonator element to the ground plane of the WCD. The resonator support element may itself support the resonator element or may include another a preformed resonator support substrate which in combination with a dielectric bridge member supports the resonator element, respectively, or supported directly by a substrate having discrete electrical components coupled thereto (i.e., the printed wiring board, or xe2x80x9cPWBxe2x80x9d) providing function to the overall operation of a WCD.
In one preferred embodiment, the resonator element includes an outwardly facing conductive portion with a plurality of inwardly facing discrete conductive portions electrically coupled thereto. More specifically, the outwardly facing portion of the resonator element may comprise an elongated band or sheet of electrically conductive material, and the inwardly facing conductive portions comprise a plurality of transverse bands of electrically conducting material electrically coupled to and spaced from the outwardly facing portion. Preferably, the flexible resonator support substrate is in a supporting relation to the outwardly and inwardly facing conductive portions and is comprised of a material (such as laminated epoxy, cyanate ester, polyimides, PTFE, etc.) having dielectric properties. The resonator element may be formed into a variety of shapes, for example, a xe2x80x9cC-shapedxe2x80x9d member, and the resonator support member may have a xe2x80x9cD-shapedxe2x80x9d member that when configured as taught herein share a common curvature, or cooperative supporting orientation or configuration. With respect to the xe2x80x9cdeformablexe2x80x9dcharacteristic of the resonator member, said characteristic is useful primarily during manufacture of the antenna assembly of the instant invention and does not contribute meaningfully to any functionality of the resulting antenna assembly.
In one embodiment, an electrically conducting connector element is preferably located adjacent the dielectric bridge member and the connector is bifurcated (e.g., shaped with elongate dual fork features) with the major end configured to operatively connect to the resonator element and the minor ends configured to operatively connect to the ground plane of a WCD and a radio frequency (RF) input/output signal feed, respectively.
The dielectric bridge member may optionally support an electrically conducting area or patch electrically coupled to the ground plane of the WCD thereby extending the effective electrical length of the ground plane. Generally, the dielectric bridge member and the flexible resonator support substrate are comprised of material having sufficient dielectric properties and may vary in thickness, shape, size and composition but generally are intended to provide mechanical support to the electrified components of the WCD, including the antenna assembly. In one embodiment of the present invention an integrated monolithic substrate member performs all the functions of the first and dielectric bridge member described herein in reference to preferred embodiments.
Generally, the dielectric bridge member and resonator support member presents a substantially planar surface including opposing major support surfaces and a support edge designed to conform to support the deformable flexible resonator support substrate. The support edge is provided with a plurality of stand-offs which enable the dielectric bridge member to contact the flexible resonator support substrate in a non-conductive relation. The downwardly facing edge includes a pair of downwardly extending tabs through which the antenna assembly may be attached to the printed wiring board of a WCD.
The resonator support substrate and the dielectric bridge member are connected to each other in a preferably orthogonal configuration and in combination with the resonator element electrically coupled to the printed wiring board and attendant ground plane of a WCD thus forming the antenna assembly.
As will be appreciated by those of skill in the art to which the invention is directed, the size, shape, physical configuration, electrical and frequency performance characteristics of the antenna assembly will depend in part on the particulars of a given WCD design iteration in view of desired operating frequency (or frequencies), interior dimensions, electrical power constraints, composition of WCD components, and the like. Further, the antenna assembly may be coupled to a WCD at a variety of locations, including the interior, the exterior, within a portion of the housing of the WCD itself, and may be coupled via a suitable antenna interface outlet using conventional components.
It is an object of the present invention to provide a compact antenna assembly designed to be incorporated into a variety of WCDs by conforming to diverse geometries within the interior space of such devices.
It is another object of the present invention to reduce the potential for damage and/or breakage of traditional antenna designs by reducing external parts to a minimum and firmly mounting antenna assembly components to pre-existing structure of compact WCDs.
It is another object of the present invention to simplify construction of an antenna assembly according to the present invention through use of known and traditional antenna, semiconductor, and electronic device fabrication techniques and technologies for production of multiple frequency band antennas.
Accordingly, another feature of the present invention is to provide a compact and effective family of designs for an antenna assembly operable in more than one frequency band.
In addition, with respect to said family of multi-frequency antenna designs, a further feature of the present invention is a single dedicated discrete electrical connection location selectively defining each frequency band while commonly electrically coupled at a single contact location to both the RF signal line and the associated ground plane.
Yet another feature and advantage of the present invention relates to a family or class of antenna assembly designs capable of conforming to existing structure of a compact WCD into which it is incorporated, including incorporating all components and electrical connections for the antenna assembly during original manufacture of the WCD on a common dielectric substrate member or members supporting the electrical circuit components of the WCD.
Still another feature of the present invention relates to the several effective antenna assembly embodiments thereof having no portion thereof external to the WCD and having no moving parts subject to breakage, physical degradation or other loss.
It is an additional object and feature of the present invention to provide an antenna assembly which may be incorporated into a WCD and wherein the resonating element portion of the antenna assembly is tunable over a range of discrete frequencies.
These and other objects, features and advantages will become apparent in light of the following detailed description of the preferred embodiments in connection with the drawings. Those skilled in the art of WCD antenna design will readily appreciate that these drawings and embodiments are merely illustrative and not intended to limit as to the true spirit and scope of the invention disclosed, taught and enabled herein.

Flexible substrate wide band, multi-frequency antenna system


A quadrature modulation circuit includes a mixer circuit including an integrated sign modulation control circuit and a plurality of mixer ports. The mixer ports include a first input port, a second input port, an output port and a sign modulation control port. The modulation circuit generates a modulated signal by operation of the mixer circuit multiplying a modulating signal applied to the first input port with a carrier signal applied to the second input port to generate a mixed signal output from the output port, and by operation of the integrated sign modulation control circuit controlling polarity switching of a signal at one of the mixer ports in response to a sign modulation control signal input to the sign modulation control port.

The increasing market demand for wireless connectivity coupled with innovations in integrated circuit technology has resulted in increased research and development of low cost, low power, and compact monolithic integration of radio transmitters, receivers, and transceivers capable of operating at GHz carrier frequencies. For example, advances in high-speed silicon-germanium technology, together with the availability for unlicensed use of 3 GHz-7 GHz of bandwidth in the 60 GHz ISM (industrial-scientific-medical) band, have served as a catalyst for potential widespread implementation of low cost 60 GHz wireless data transmission systems using integrated 60 GHz radio transceivers.
Indeed, the large unlicensed bandwidth available in the 60 GHz ISM band enables wireless systems to be implemented using digital modulation techniques such as QAM (quadrature amplitude modulation), ASK (amplitude-shift key), FSK (frequency-shift key) or PSK (phase-shift key) modulations to achieve gigabit-rate modulations that support high bandwidth applications such as wireless Gigabit Ethernet and HDTV (high definition television) streaming. In comparison to conventional analog modulation techniques, digital modulation schemes such as QAM, ASK, FSK and PSK are more power efficient and are more robust to noise and multipath effects, etc. Moreover, digital modulation provides more information capacity, higher data security, and better quality communication, especially for low power, high data rate applications.
The quadrature modulator is a fundamental component that is commonly used for digital microwave radio communications systems. FIG. 1 is a block diagram illustrating a quadrature modulator (10) having a conventional framework. The quadrature modulator (10) comprises a first mixer (11), a second mixer (12), a phase shifter (13) and summing circuit (14). The inputs to the quadrature modulator (10) include modulating signals I (in-phase signal) and Q (quadrature-phase signal), and an LO (local oscillator) signal. The I and Q modulating signals may be analog signals I(t) and Q(t) generated by a baseband processor which implements complex DSP processing functions for purposes of processing and encoding a data signal (digital data) to generate independent I and Q serialized digital data streams to represent the input data. The baseband processor can implement DAC (digital to analog converters) to covert the I and Q digital streams to analog signals I(t) and Q(t), which are input to respective IQ mixers (11) and (12).
The LO port of the first mixer (11) receives the LO signal as an in-phase LO signal, LO1(0°), while the phase shifter (13) phase shifts the LO signal by 90° to produce a quadrature-phase LO signal, LOQ (90°), which is input to the LO port of the second mixer (12). The first mixer (11) mixes the I-channel input signal, I(t), with the in-phase LO signal, LO1, and the second mixer (12) mixes the Q-channel input signal, Q(t), with the quadrature-phase LO signal, LOQ. The outputs of the mixers (11) and (12) are combined by the summing circuit (14) to generate a QAM modulated output:Vout(t)=[I(t)*cos(2πfLOt)]−[Q(t)*sin(2πfLOt)]  (1)where I(t)=A(t)cos φ(t) and Q(t)=A(t)sin φ(t).
The QAM output signal of the modulator (10) can also be represented as:Vout(t)=A(t)cos [(wLOt)−φ(t)]  (2),where the signal amplitude isA(t)=√{square root over (I2(t)+Q2(t))}{square root over (I2(t)+Q2(t))}  (3)and where the phase modulation (shift) isφ(t)=tan−1[Q(t)/I(t)]  (4)
In this regard, a conventional quadrature modulator such as depicted in FIG. 1 can support a modulation scheme in which one or more parameters of the carrier (LO) frequency (amplitude, phase and/or frequency) can be modulated to represent information. The quadrature modulator (10) of FIG. 1 can readily support various QAM modulation modes and PSK modulation modes, such as BPSK (binary phase-shift key) (which is essentially 2-QAM) and QPSK (quadrature phase-shift key) modulation (which is essentially 4-QAM). In theory, any modulation format can be implemented using quadrature (IQ) modulation techniques, including ASK and FSK, by converting an input stream of data bits to appropriate waveform values on the baseband I and Q inputs to realize the desired modulation, as is understood by those of ordinary skill in the art. When using quadrature modulation schemes, however, it is a non-trivial, challenging and resource-consuming task to create the required baseband waveforms for a desired modulation (e.g., FSK) at the very high speeds that are required to achieve gigabit rate modulation.

Quadrature modulation circuits and systems supporting multiple modulation modes at gigabit data rates


Disclosed herein are methods that include contacting a skin surface with a first liquid composition; and then contacting in the skin surface with a cationic coated article loaded with a second liquid composition, while at least some portion of the first liquid composition remains on the skin surface, wherein one or both of the first liquid composition or the second liquid composition includes acrylate copolymer particles dispersed therein, the acrylate copolymer particles including the reaction product of a reaction mixture, the reaction mixture including monomers, the monomers including from about 5 wt % to about 50 wt % of at least one high Tg monomer where the wt % of the high Tg monomer is with respect to the total weight of the monomers in the reaction mixture; and from about 20 wt % to about 80 wt % of at least one low Tg monomer where the wt % of the low Tg monomer is with respect to the total weight of the monomers in the reaction mixture, wherein the particles have a number average diameter of at least about 100 nm and wherein at least one and only one of the first or the second composition comprises greater than or equal to 60 wt % of at least one alcohol.

There is a great deal of interest and urgency in preventing the spread of C. difficile, particularly in medical settings such as hospitals. Patients in a hospital setting often times develop C. difficile infections during or shortly after a course of antibiotics. While it is relatively easy to kill the vegetative form of C. difficile, the spore form of C. difficile, can be very difficult to kill. New technologies are therefore needed to address the problem of preventing the spread of C. difficile, between patients, health care workers, and the environment.

Methods of removing spores comprising alcohol, acrylate copolymer particles, and a cationic coated article


A digital receiver has a receiving section for receiving a digital modulated signal, a digital demodulation section for judging a logical state of a detection output signal obtained by performing a digital detection operation for the digital modulated signal, for demodulating the digital modulated signal and for generating a data signal. The digital demodulation section has a binary conversion circuit for converting the data signal into a binary signal and a receiving quality detection device for detecting a receiving quality of the data signal based on a wave-form of the binary signal. The receiving quality detection device has a pulse generator for generating pulses having a time width between a positive pulse width of the wave-form of the binary signal based on the digital modulated signal and a time width between a third time width and a time width which is obtained by subtracting a negative pulse width of the wave-form of the binary signal from a second pulse width. The digital demodulation section has a wave-form shifter for shifting the wave-form of the binary signal of the in phase, a data judgement device for judging a state of the detection output signal based on a synchronous clock signal, and an exclusive OR device for performing a logical exclusive OR operation between the wave-form of a wave-form of output from the data judgement device and a wave-form of output from the wave-form shift means.

1. Field of the Invention
The present invention relates to a digital receiver such as a FM multiplex broadcast receiver used for Frequency Modulation (FM) multiplex broadcasting, and more particularly to a digital receiver for receiving a digital modulated signal which is modulated in a digital form using a digital modulation section, for demodulating the digital modulated signal to a data signal by a demodulation section, and for performing desired processes on the data signal.
2. Description of the Prior Art
FIG. 40 is a configuration diagram showing a configuration of a conventional Frequency Modulation (FM) multiplex broadcast receiver as a digital receiver for receiving a digital modulated signal used for FM broadcasting and for displaying the received data.
In FIG. 40, reference numbers 1 and 2 designate antennas for receiving electromagnetic waves propagating in space, reference number 3 denotes a diversity section for selecting one of the antennas based on a control signal (sig7a) transferred from comparator 7, a reference number 4 denotes a tuner section for selecting a desired carrier frequency, for amplifying the selected frequency, and for converting the amplified frequency into an Intermediate Frequency (IF). Reference number 5 designates a Phase Locked Loop (PLL) section for controlling the channel frequencies transmitted from broadcast stations received in the tuner section 4, reference number 6 denotes a Frequency Modulation (FM) detection section for converting the FM signal (sig4a) which has been converted with the Intermediate Frequency (IF) by the tuner section 4 into a base band signal (sig6a). The FM detection section 6 generates a stop signal (sig6b) to stop a scanning for a desired broadcast frequency during an automatic broadcast frequency selection operation and a received electric field intensity signal, namely a S meter signal (sig6b), which is a smoothed output of the intermediate frequency signal, and transmits the S meter signal (sig6b). The receive section in the conventional digital receiver shown in FIG. 40 includes the components 1 to 6 described above.
Reference number 8 denotes an analogue to digital (A/D) conversion device for the voltage value of the S meter signal (sig6b) to a digital signal in a digital form to transmit it to a control section 11. Reference number 7 designates a comparator for comparing the voltage of the S meter signal (sig6b) with a reference voltage VR1 and for transmitting a control signal (sig7a) as the comparison result of the comparison operation to the diversity section 3.
Reference number 9 denotes a digital demodulation section for selecting a multiplex transmission signal which is modulated in a digital form from the base band signal (sig6a) by the FM detection section 6, for digitally demodulating the multiplex transmission signal, and for generating a data signal (sig9a) and a synchronous clock signal (sig9b), reference number 10 designates an error correction section for performing an error correction operation based on an error correction flag in the multiplex transmission signal, and generating a data signal (sig10a) obtained after the error correction operation, and auxiliary information (sig10b) such as a synchronous establishment signal, a receive rate and the like. Reference number 11 designates the control section for performing tuning control and a data processing of received data from the error correction section 10, and for processing an operational signal as a control signal from an operation section 13 which is indicated by an operator. Reference number 12 designates a display section for displaying information based on data from the control section 11, and the reference number 13 is the operation section for receiving required data such as a tuning frequency which is selected by an operator and generating and transmitting the control data based on the selected tuning frequency and the like to the control section 11.
FIG. 41 is a drawing showing an example of an internal circuit of the digital demodulation section 9 in the conventional digital receiver shown in FIG. 40.
In the demodulation section 9 shown in FIG. 41, reference number 16 is a Band Pass Filter section for selecting the multiplex transmission signal which is modulated in a digital form in a base band signal transmitted from the FM detection section 6, reference number 17 is a binary quantization device for converting the digital modulated signal into a binary signal, reference number 18 is a delay device for delaying the binary signal (sig17) by the time period of 1 bit in order to perform a delay detection operation and generating a delayed signal (sig18), reference number 19 is an exclusive OR operation device for performing an exclusive OR operation between the binary signal (sig17) and the delayed signal (sig18), reference number 20 denotes a Low Pass Filter (LPF) for eliminating a high frequency component in the output of the exclusive OR operation device. Reference number 21 designates a data judgement device for judging of a state of the output signal from the LPF 20, and for converting this output signal into a digital wave-form and for transferring a judgement signal (sig9a) as the result of the judgement by the data judgement device 21 which is the data signal (sig9a) which is demodulated.
Reference number 23 designates a binary code quantization phase comparator for detecting a phase difference between the digital detection signal and the synchronous clock signal (sig9b) at a change point of the digital detection signal in value, and for generating a leading phase pulse (sig23a) and a delay phase pulse (sig23b). Reference number 24 denotes a sequential filter, which includes counter circuits capable of executing a digital integration operation, for performing an integration count operation to count output from the binary code quantization phase comparator 23 in order to eliminate an influence of a noise such as jitter, and for generating and transmitting a delay phase control signal (sig24a) and a leading phase control signal (sig24b). Reference number 25 denotes a fixed frequency oscillator, a reference number 26 designates a pulse addition/elimination device for adding a pulse to the output from the fixed frequency oscillator 25 when receiving the delay phase control signal (sig24a) from the sequential filter 24, and for eliminating the pulse when receiving the leading phase control signal (sig24b) from the sequential filter 24. A reference number 27 denotes a divider for dividing the output transmitted from the pulse addition/elimination device 26 into the synchronous clock signal (sig9b). Thus, a synchronous clock reproduction device 22 includes the components designated by the reference number 23-37 described above.
FIG. 42 is an example of an internal circuit of the diversity section 3 in the conventional digital receiver device shown in FIG. 40. In the diversity section 3 shown in FIG. 42, reference number 302 designates a selection device for selecting one of the output signal from the fixed frequency oscillator 301 and a ground signal based on the control signal (sig7a) from the comparator 7 and transmitting the selected one, reference number 303 designates a flip flop circuit for inverting a logical value of the current output signal from the flip flop circuit 303 when receiving a rising edge of the output signal from the selection device 302. Reference number 304 denotes a selection device for selecting one of the antennas 1 and 2 based on the output signal as a control signal from the flip flop circuit 303. Thus, the diversity section 3 includes the components designated by the reference numbers 301 to 304 described above.
Next, the operation of the diversity section 3 described above will now be explained.
Firstly, the diversity section 3 selects one of the antennas 1 and 2 based on the control signal from the comparator 7 which is also controlled based on the S meter signal (sig6b) from the FM detection section 6 and transmits a high frequency signal (sig3) which has received by the selected antenna. Because the selection device 302 in the diversity section 3 selects and transmits an output signal (sig3) from the fixed frequency oscillator 301 to the tuner section 4 when the output signal (sig7a) from the comparator 7 is a positive logical value where the voltage value of the S meter signal having the feature shown in FIG. 44 is not more than an antenna switch voltage level, the antennas 1 and 2 are switched alternately when the output signal (sig303) from the flip flop 303 is periodically changed to the positive logical value and a negative logical value. This antenna switching operation described above is continued until the control signal (sig7a) is changed to the negative logical value as the output signal from the comparator 7 until the voltage value of the S meter signal (sig6b) is more than the antenna switching level voltage and after the selection device 302 selects and transmits the ground signal, and after the output signal from the flip flop 303 is fixed in order to selects one of the antennas 1 and 2 continuously.
FIG. 43 shows an example of changing the value of the S meter signal (sig6a) and the antenna switch operation.
The tuner section 4 tunes the high frequency signal (sig3) from the diversity section 3 with a desired frequency based on the tuning voltage transmitted form the PLL section 5 and converts the desired high frequency signal in the high frequency signal (sig3) to the Intermediate Frequency signal.
The FM detection section 6 receives the Intermediate frequency signal from the tuner section 4 and performs an FM detection operation of the Intermediate frequency (IF) signal received and converts it in to a baseband signal (sig6b), and transmits the baseband signal (sig6b), the S meter signal which is obtained by rectification of the Intermediate Frequency (IF) signal, and the stop signal, indicating that the strength of the received electric magnetic field is more than a desired strength.
FIG. 43 shows an output characteristic of the stop signal to the received electric magnetic field in the conventional digital receiver shown in FIG. 40.
FIG. 44 shows an output characteristic of the S meter signal to the received electric magnetic field in the conventional digital receiver shown in FIG. 40.
The digital demodulation section 9 selects a multiplex transmission signal which is modulated in a digital form in the baseband signal (sig6a) transmitted from the FM detection section 6 and generates a digital demodulated signal (sig9a) and a synchronous clock signal (sig9b) which will be described later and transmits them.
An audio processing section 14 selects an audio signal in the baseband signal (sig6a) transmitted from the FM detection section 6 and amplifies an audio stereo signal obtained by demodulating the audio signal in stereo form and transmits the audio stereo signal to the speaker in order to drive the speaker 15.
The error correction section 10 receives the digital demodulated signal (sig9a) transmitted from the digital demodulation section 9 and performs the error correction operation for the digital demodulated signal (sig9a) by using the error correction flag in the digital demodulated signal (sig9a) and generates a data signal (sig10a) and the auxiliary information (sig10b) such as the synchronous establishing signal and the receive rate and transmits them to the control section 11.
The control section 11 receives the data signal (sig10a) from the error correction section 10 and displays the data signal (sig10a) on the display section 12 and performs several control operation such as the tuning control operation and the audio control operation.
Next, the tuning operation of the control section 11 in the conventional digital receiver shown in FIG. 40 will be explained. The tuning operation of the control section 11 relates to features of a digital receiver according to the present invention.
In the tuning control operation, the data (sig11b) indicating a tuning frequency is transmitted from the control section 11 to the PLL section 5 by using the PLL control signal (sig11b). Then the PLL section 5 generates the tuning voltage based on the data (sig11) indicating the tuning frequency. Then the tuner section 4 tunes the high frequency signal (sig3) from the diversity section 3 based on the tuning voltage.
A fully automatic selection operation for broadcast stations sequentially changes data of the tuning frequency and scans the tuning frequency and stops the scanning operation when the strength of the received electromagnetic field is more than the predetermined value by using the stop signal (sig6c) transmitted from the FM detection section 6.
Prior to explanation of the digital demodulation operation executed by the digital demodulation section 9, we will explain the basic theory about the delay detection method which is used in the conventional digital receiver shown in FIG. 40.
In general, a modulated signal is defined the following equation EQU s(t)=cos 2.pi.f t+.PHI.(t)!,
where f is a carrier frequency of a digital modulated signal, and .PHI.(t) is a digital modulated component.
When the modulated signal is multiplied by a signal which is delayed by the time period of a bit and then the high frequency component in the result of the multiplication is eliminated, the sig20 is EQU sig20=cos 2.pi.f t+.PHI.(t)-.PHI.(t-T!.
Because the term "2.pi. f t" becomes 9.5.pi. in a FM multiplex broadcasting, the above equation becomes EQU sig20=sin .PHI.(t)-.PHI.(t-T)!.
Thus, only modulated component .PHI. (t) in the modulated signal can be selected.
This digital modulation method used for the FM multiplex broadcasting is called the L-MSK modulation method.
This modulation method is one of the frequency modulation methods. In this modulation method, as shown in FIG. 45, the positive logical state is 80 KHz and the negative logical state is 72 KHz.
Next, the operation of the digital demodulation section 9 will be explained.
Because the baseband signal of the FM multiplex broadcasting has a spectrum structure as shown in FIG. 46, a multiplex transmission signal which is modulated in a digital form by the band-pass filter 16 is selected from the baseband signal (sig6a) transmitted from the FM detection section 6. Next, the digital modulated signal is converted into a digital signal by the binary code quantization device 17 in order to delay the digital modulated signal by the time period "T" of a bit by using the delay device including shift registers.
The exclusive OR logical output (sig19) between the digital modulated signal (sig17) which has been converted in a binary form and the signal (sig18) which is delayed by the time period "T" of a bit executed by the delay device 18 is transmitted to the data judgement device 21 through the LPF 20. Thereby the detection output signal in which a high frequency component is eliminated is obtained, as shown in FIG. 47.
The detection output signal (sig20) is judged by the data judgement device 21 in synchronization with the synchronous clock signal transmitted from the synchronous clock reproduction section 22 and the data judgement device generates the data signal (sig9a) based on the result of the data judgement operation.
The synchronous clock reproduction section 22 compares the detection output signal (sig20) with the synchronous clock signal at the rising edge or the falling edge of the detection output signal (sig20). Then, the section 22 generates the leading phase signal (sig23a) when the phase of the synchronous clock signal is ahead of the phase of the detection output signal, and generates the delay phase signal (sig23b) when the phase of the synchronous clock signal is delayed.
The sequential filter 24 in the synchronous clock reproduction section 22 includes counter circuits each of which has a function of the digital integration operation in order to decrease the influence of the noises such as jitter. Here the configuration and the operation of a typical N before M counter circuit will be explained.
The N before M counter circuit generates the delay phase control signal (sig24a) when the total number of the leading phase signals is N (where N is a positive number) and generates the leading phase control signal (sig24b) when the total number of the delay phase signals is N (where N is a positive number), until the total sum of the numbers of the leading phase signals and the delay phase signals reaches M (where M is a positive number). In addition, the N before M counter circuit is reset when the total number of the leading phase signals and the delay phase signal reaches M (where M is a positive number).
The pulse addition/elimination device 26 transmits the output from the oscillator 25 not including a pulse when the device 26 receives the delay control signal (sig24a) from the sequential filter 24. On the other hand, the pulse addition/elimination device 26 transmits the output from the oscillator 25 including a pulse when the device 26 receives the leading control signal (sig24b) from the sequential filter 24.
The divider 27 divides the signal from the pulse addition/elimination device 26 and generates the synchronous clock signal (sig9b). The oscillation frequency from the oscillator 25 is selected as to become a frequency that is k times of the frequency of the synchronous clock signal, and the divided rate of the divider 27 is selected as to become the 1/k of the oscillation frequency from the oscillator 25 (where k is a positive integer).
Japanese Laid-Open Patent Applications Nos. 6-276113, 4-47729, and 4-47730 discuss conventional digital receivers, such as the conventional FM multiplex broadcast receivers described above.
Since the conventional FM multiplex broadcast receiver has the configuration described above, there is no method to detect a data receiving state of the receiver other than the method wherein the data receiving state is detected by receiving the synchronous establishing signal (sig10b) to inform the establishment of the data synchronous operation for the data transmitted from the error correction section 10, or by using the data error rate obtained by the error detection operation or the error correction operation, or by using the S meter value.
There are a block synchronous establishing signal and a frame synchronous establishing signal in the synchronous establishing signal. In the use of the block synchronous establishing signal, there is a problem that there is a great time delay which is counted from the receiving of a electromagnetic wave transmitted from a broadcast station to the detection of the synchronous establishing signal because this block synchronous establishing signal is obtained based on a sequential detection of the synchronous bit by several times. Accordingly, it is difficult to use the block synchronous establishing signal as a control signal for devices such as the diversity section 3 shown in FIG. 40 where a rapid response is required.
FIG. 48 shows a data format structure used for the FM multiplex broadcasting. In the case of the FM multiplex broadcasting, since a block is made up of 272 bits, a time delay of several hundred m sec! is caused in order to detect the establishment of the block synchronous in some case. In addition, because 16 bits in a block is used for the establishment of the block synchronous operation, there is no reliability that the synchronous bits (16 bits) is equal to the other part of the data format in receiving state such as a receiving time, as shown in FIG. 48.
Further, because the frame synchronous establishment signal is generated after receiving of the synchronous bits having a predetermined pattern, it takes more time for detecting the establishment of the frame synchronous rather than that of the block synchronous. In addition, because the receiving rate is calculated based on the number of blocks which have been uncorrected in the error correction operation, it take more times to detect the establishment of the receiving rate rather than that of the frame synchronous establishment signal.
Although, the relationship between a voltage value of the S meter signal and the receiving rate, namely the error rate can be clearly shown by using the S meter signal (sig6b), it is difficult to detect a deterioration of the data receiving state which is caused by a phase difference of multi-path even if the DU rate is increased. Accordingly, in some cases, the data receiving state cannot be detected by using the S meter signal (sig6b).
In addition, all of FM broadcast stations do not always transmit a multiplexed broadcast. Further, all of actual broadcast time periods in a FM multiplex broadcast station do not transmit the multiplexed broadcast. Accordingly, it can not be detected whether or not the multiplexed broadcasting is now ON by using the S meter signal (sig6b) or the stop signal (sig6c). In this case, when automatically selects a multiplex broadcasting, it must be detect the absence of the multiplex broadcasting by using the synchronous establishment signal and the receiving rate after selecting it based on the S meter signal (sig6b) or the stop signal (sig6c). Further, in this case, it must be check the absence of the multiplex broadcasting for a broadcast station which transmits no multiplexed broadcast. Because it takes more times in the case described above, there is a problem.
In addition, there is information to display after accumulating received data in a memory in the conventional FM multiplexed broadcast described above. Therefore, in some cases the received data can not be displayed before the completion of data accumulation even if the data receiving state is good.
Because the conventional FM multiplex broadcast receiver has the configuration described above, it is inconvenient in the conventional receiver that an operator cannot immediately know whether received data will be displayed after the completion of the data accumulation in a good data receiving state or there is no data to display in a poor data receiving state.
Moreover, because the sequential filter 24 for reducing the influence of noises such as jitter by performing an integral count operation of the output from the binary code quantization phase comparator is used in the conventional clock synchronous reproduction section 22, it is difficult to respond to a sharp phase change which is caused in the diversity section 3 by switching the antennas 1 and 2. This is also a problem.


There is disclosed a connector which can be used with either of crimp-type terminals and press-connecting terminals, thereby enhancing an operation efficiency, and also reducing the cost. The connector includes a housing having an opening formed at one side of rear half portions of terminal receiving chambers, and wires are inserted respectively into the terminal receiving chambers through the opening, and the housing has a cover for covering the opening. The cover is integrally connected by molding to that portion of the housing, disposed forwardly of the opening, through a hinge so that the cover can be turned between a closed and an open position. The cover and the housing are provided with engagement means by which the cover can be retained relative to the housing in a two-stage manner, that is, in a provisionally-retained condition in which the cover is provisionally retained relative to the housing to provisionally retain terminals received respectively in the terminal receiving chambers, and a completely-retained condition in which the cover is completely retained relative to the housing to completely retain the terminals and also to close the opening.

1. Field of the Invention
This invention relates to a connector for use with either of press-connecting terminals and crimp-type terminals.
2. Related Art
FIG. 1 shows a connector for press-connecting terminals disclosed in Japanese Utility Model Unexamined Publication No. 4-136869, and a housing 30 of this connector includes terminal receiving chambers 31 which are open as at 30a at one side of a rear half portion of the housing 30, upwardly-projecting cover retaining projections 32 formed respectively on upper edges of partition walls, a wall portion 33 extending forwardly from the opening 30a, and a cover insertion hole 34. Press-connecting terminals (not shown) are inserted respectively into the terminal receiving chambers 31, and then a front end of a separate cover 35 is inserted into the cover insertion hole 34, and the cover 35 is pushed down, so that the cover retaining projections 32 are fitted respectively in retaining grooves 36 formed in the cover 35.
FIG. 2 shows a connector for crimp-type terminals disclosed in Japanese Patent Unexamined Publication No. 5-226025, and rear ends of a pair of covers 42 are connected by hinges 43 to a rear end of a housing 41, having terminal receiving chambers 40, so that each cover 42 can be opened and closed, and each cover 42 is closed from the rear side of the housing 41 toward the front side thereof.
However, when crimp-type terminals are to be used in the press-connecting terminal connector shown in FIG. 1 the plurality of crimp-type terminals are often inserted into the terminal receiving chambers at different times, depending on the kinds of wires and so on, and in this case there is encountered a problem that the terminals are liable to be withdrawn before the separate cover 35 is attached to the housing 30. On the other hand, in order to prevent the withdrawal of the terminals, if the separate cover 35 is attached to the housing 30 each time the crimp-type terminals are inserted into the corresponding terminal receiving chambers 31, there is encountered a problem that the operation is cumbersome.
When press-connecting terminals are to be used in the crimp-type connector shown in FIG. 2, there is encountered a problem that since the housing 41 has walls 41 formed at its upper and lower sides, a wire can not be press-connected to the press-connecting terminal, mounted in each terminal receiving chamber 40, even when the cover 42 is opened. Even if each wall 44 is removed to open rear half portions of the terminal receiving chambers 40, the cover 42 exists above this opening, and it is difficult to press-connect each wire to the associated terminal.


A coil device 10 comprises a first bobbin 40 provided with a first winding part 45 at an outer circumference and a second bobbin 50 provided with a second winding part 55 at an outer circumference. A plurality of partition walls 46, separating portions of the wire 22 which are adjacent to each other along the scroll axis Z of the first wire 22 are formed on the first winding part 45 along the scroll axis at predetermined intervals. A section width w1 of each section 47, which is along the scroll axis Z, separated by the partition walls 46 is determined so that only one wire 22 can pass through. A height of the partition walls 46 is determined so that one or more of the wires 22 can pass through. At least one connecting groove 46a, connecting each section which are adjacent to each other, is formed on each partition wall 46.

1. Field of the Invention
The present invention relates to a coil device preferably used for a resonance transformer and the like.
2. Description of the Related Art
Coil devices are used in various electrical products for various uses. For instance, in a lighting circuit for a backlight of liquid crystal display, a leakage transformer, which is as a resonance transformer for driving a display device with higher voltages, is generally used.
For a leakage transformer, as shown in the following Reference 1 for instance, a horizontal-type coil device, to which a scroll axis of coil is arranged parallel to a mounting substrate surface of the coil device, is known. Such horizontal-type coil device has a problem that a leakage flux toward upward and downward directions with respect to the mounting substrate surface is large.
In order to make the leakage flux small, it is considered that top and bottom of the horizontal-type coil device is covered with aluminum board or aluminum foil. However, with this, heat dissipation may be deteriorated.
Further, for other leakage transformers, as shown in the following Reference 2 for instance, a vertical-type coil device, to which a scroll axis of coil is arranged perpendicular to a mounting substrate surface of the coil device, is known. With its configuration, it enables to make the leakage flux toward upward and downward directions with respect to the mounting substrate surface small.
However, for conventional coil devices, a primary coil and a secondary coil are composed of wires wound by normal regular winding. Particularly, for the secondary coil which produces a high voltage, there is a problem with voltage withstandability since the start of winding wire for a first layer and the end of winding wire for a second layer closely contact with each other, and the voltage differences between them get larger. Further, for the regular winding, it is a way to wind one wire in a spiral way for a first layer and then to wind back on the first layer from end to start of the first layer for a second layer. For the subsequent layers, the same process is applied.
Further, as the frequency of voltage applied to a coil device gets higher, there is a problem with the current flow since wires, which are adjacent to each other, exert effects on each other. Furthermore, for the coil devices used for a leakage transformer, it is important to stabilize leakage characteristics. However, for the conventional coil devices wherein wires are wound in regular winding, there is a problem with the stability of leakage characteristics.    [Reference 1] Japanese Published Unexamined Application No: 2006-108390    [Reference 2] Japanese Published Unexamined Application No: 2005-158927