Elastography can produce quality strain images in vitro and in vivo. Standard elastography uses a coherent cross-correlation technique to estimate tissue displacement and tissue strain using a subsequent gradient operator. While coherent estimation methods generally have the advantage of being highly accurate and precise, even relatively small undesired motions are likely to cause enough signal decorrelation to produce significant degradation of the elastogram. For elastography to become more universally practical in such applications as hand-held, intravascular and abdominal imaging, the limitations associated with coherent strain estimation methods that require tissue and system stability, must be overcome. In this paper, we propose the use of a spectral shift method that uses a centroid shift estimate to measure local strain directly. Furthermore, we also show theoretically that a spectral bandwidth method can also provide a direct strain estimation. We demonstrate that strain estimation using the spectral shift technique is moderately less precise but far more robust than the cross-correlation method. A theoretical analysis as well as simulations and experimental results are used to illustrate the properties associated with this method.

1. Field of the Invention
The present invention relates to a method of measuring strain in a target body, using the transmission, reception, processing and normalization of ultrasound signals.
2. Description of the Prior Art
Imaging of elastic parameters of soft tissue has developed into a new tool for diagnosis of disease. Current estimators of tissue motion, include a time-domain cross-correlation based speckle tracking algorithm, and a Fourier based speckle phase-tracking technique. These techniques are coherent estimation techniques, i.e., these methods are sensitive to phase variations. The coherent estimation techniques generally have the advantage of being highly precise. Strain Filter (SF) analysis has shown, however, that they are not very robust in the presence of even a small amount of decorrelation between the pre- and post-compression signals. The term xe2x80x98robustnessxe2x80x99 has been used in statistical analysis to denote the good performance of statistical tests, i.e., the homogeneity of the variance calculation, even if the data deviates from theoretical requirements. By equivalence in elastography, xe2x80x3robustnessxe2x80x2 denotes the consistently good performance of the estimator even at high decorrelation noise, i.e., keeping the variance of estimation at a relatively constant and low level at a large range of noise levels.
The term xe2x80x9cdecorrelationxe2x80x9d as used herein, is defined as the loss of full correlation between the pre- and postcompressed windowed signal segments. Therefore, decorrelation may be encountered due to many sources, such as intrawindow axial motion undesired lateral or elevational motion, jitter (i.e., any cause of misregistration between the pre- and post-compressed A-line segments), unstable mechanical setup, etc. The main idea in this study is to introduce a new estimator that is more immune to decorrelation compared to other estimators.
The tissue strain estimator is a spectral estimator that estimates strain directly. Since the proposed estimator uses the power spectrum, it is incoherent, i.e., it does not use the phase of the signal. Previously reported incoherent methods include optical flow speckle tracking, envelope cross-correlation, and spectral chirp z-transform techniques. Generally, incoherent methods may be less precise but significantly more robust. For example, we have demonstrated this property for the case of time-delay estimation using the envelope of echo-signals. This may be a significant advantage where elastography is to be practiced in situations involving (1) undesired scanning motion, such as the case of using an unstable handheld transducer and/or (2) undesired tissue motion, such as abdominal or intravascular elastography. This property of the estimator is demonstrated later in this paper in the simulation results section through testing of its immunity to noise caused by jitter.
The main idea behind a spectral strain approach is based on the Fourier scaling property, which implies that a compression or expansion of the time-domain signal should lead to an expansion or compression of its power spectrum, respectively. One of the most well known and thoroughly studied spectral motion effects is the Doppler shift, which typically links the frequency shift to the scatterer velocity between emissions. Velocities towards the transducer result in a positive frequency shift, while the opposite is true for scatterers that move away from the transducer. However, since the scatterers within a given resolution length do not move at the same velocities, a spectrum of Doppler frequencies is observed. Therefore, initially in ultrasound, the methods of velocity estimation for the measurement of blood flow mainly operated in the frequency domain, otherwise known as spectrum analysis techniques and measured the mean velocity of scatterers across the vessel lumen (indicative of the volumetric flow rate) by estimating the mean frequency of the power spectrum. Despite the success of these techniques even in in vivo vessels, detection of the Doppler frequency shift, which is typically on the order of 1 kHz, is not possible for pulsed instruments, since the downshift in frequency due to attenuation (on the order of 10-100 kHz) is expected to dominate over the Doppler shift. Since in elastography the pre- and postcompressed segments are approximately identical depths, the attenuation effect on the two spectra is assumed to be identical and cancelled out when the two spectra are compared.
Strain estimation using spectral methods depends on the subsequent change in the scatterer statistics. Spectral methods typically link one or more signal parameters to the change in mean scatterer spacing. One prior art relates the relative change in the mean scatterer spacing to the strain incurred during a cardiac cycle. This method assumes the presence of underlying scatterer periodicities. Despite the fact that this has also been demonstrated to work in in vivo intravascular applications, the main assumptions of regular spacing or periodicities may not hold for most tissues. In contrast, as shown in the theory section, the spectral methods mentioned in this paper make no assumptions regarding the composition of the tissue scatterers.
Typically in elastography time-domain techniques are used that involve the computation of the time-delay to estimate the displacement following an applied compression, and the estimation of strain by applying gradient operations on the previously obtained time-delay estimates. As mentioned earlier, an important advantage associated with these spectral methods as well as other estimators, such as the adaptive stretching estimator, is that they can be used to estimate strain directly; i.e., without involving the use of noisy gradient operators. In the latter case, the gradient operation introduces additional amplification of the noise into the strain estimation process, thus degrading the strain estimates. Furthermore, similar to the adaptive stretching estimator, only one estimation window is needed, for both the magnitude and the sign of the strain to be estimated.
As shown later in the theory section, spectral estimators can be divided into two main groups: a) the spectral shift methods and b) the spectral bandwidth methods. Despite the fact that we develop expressions that show direct strain estimation in both cases, in this paper we focus primarily on a spectral shift method; we estimate the relative shift in the spectral centroid caused by compressive or tensile tissue strain. Therefore, throughout this paper this new estimator is referred to as the xe2x80x9ccentroid strain estimatorxe2x80x9d, xe2x80x9ccentroid estimatorxe2x80x9d or xe2x80x9ccentroid methodxe2x80x9d. Current investigations that deal with the development of alternative shift estimators as well as bandwidth estimators will not be reported in this study.
The spectral centroid has been widely used in estimating the Doppler shift, attenuation and backscattering. The theory underlying the use of centroid strain estimators is presented in the next section. One-dimensional (1-D), instead of two-dimensional (2-D), motion simulations are used in order to more accurately study the performance of the estimator, i.e., independent of the effect of signal decorrelation in two dimensions that complicates the measurements. Simulation results in 1-D illustrate the insensitivity of the centroid strain estimator to signal decorrelation effects. It is important to note that, as mentioned earlier, decorrelation can be due to several sources. For the purpose of this paper, we consider solely the axial decorrelation effect in this 1-D model. We, thus, assume that the robustness demonstrated by the spectral estimator vis-à-vis this effect is a more general property that can be further applied at other decorrelation scenarios. For example, it is shown in the results section how the spectral method is indeed more immune to jitter, another source of decorrelation. The elastograms obtained using these simulations as well as phantom experiments illustrate the robustness of the spectral centroid strain estimator. The properties of the new estimator are discussed and summarized in the Conclusion section.
In this section, we show analytically that for Gaussian echo spectra, the relative spectral shift is a direct measure of tissue strain. We also show the relative bandwidth variation can also be used as a direct strain estimator.
Signal and noise model.
The pre- and post-compression echo signals are given as follows:
r1(z)=h(z)*e(z)+n1(z)xe2x80x83xe2x80x83(1)
r2(z)=h(z)*e(az)+n2(z)xe2x80x83xe2x80x83(2)
where z is a spatial variable, r1(z) and r1(z) are the received RF signals before and after compression, respectively, h(z) is the impulse response of the ultrasound system or point-spread function (PSF), e(z) is the scattering function, n1(z) and n2(z) are independent zero-mean white noise sources and xcex1 is the compression coefficient (or, strain factor) linked to strain s through                     a        =                              1                          1              -              s                                ≅                      1            +            s                                              (        3        )
The approximation holds for s less than  less than 1, where the strain s for a one-dimensional homogeneous target is typically defined in mechanics by                     s        =                                            L              0                        -            L                                L            0                                              (        4        )
where L0 and L are the pre- and postcompressed axial dimensions of the target. From Eq. (4) the reader should note that positive strain denotes compression (and a greater than 1) while negative strain denotes tension (and a less than 1). The reader should note that throughout this paper the subscripts 1 and 2 denote pre- and postcompression parameters, respectively.
Assuming that h(z) and e(z) in Eqs. (1) and (2) can be described by their autocorrelation functions that may be modeled by modulated Gaussian functions, we obtain                                           h                                      (              z              )                                =                                    1                                                                    2                                        π                                                                                L                  h                                                                                exp                                            (                                                                            -                                              z                        2                                                              /                    2                                                                          L                    h                    2                                                  )                                                  sin                                      xe2x80x83                                                  (                                                k                  h                                                z                            )
                        and                            (        5        )                                          e                                (            z            )                          =                              1                                                            2                                    π                                                                          L                e                                                                    exp                                      (                                                                    -                                          z                      2                                                        /                  2                                                                  L                  e                  2                                            )                                          cos                                xe2x80x83                                          (                                          k                e                                          z                        )                                              (        6        )
where Lh and Le are the resolution lengths of the PSF and of the scattering function, respectively, kh is the central spatial frequency of the PSF, and ke is the central spatial frequency of the scattering function.
The one-sided power spectra of the pre- and post-compression RF signals (positive frequencies) are given respectively by                                           R            1                                          (            k            )                          =                                            1              4                                                  exp                                            (                                  -                                                            1                      2                                                                                [                                                                                (                                                              k                                -                                                                  k                                  h                                                                                )                                                        2                                                                                L                            h                            2                                                                          +                                                                                (                                                              k                                -                                                                  k                                  e                                                                                )                                                        2                                                                                L                            e                            2                                                                                ]                                                                      )                                              +                                    N              l                                                  (              k              )                                                          (        7        )                                                      R            2                                          (            k            )                          =                                            1                              4                                a                                                                exp                                            (                                  -                                                            1                      2                                                                                [                                                                                (                                                              k                                -                                                                  k                                  h                                                                                )                                                        2                                                                                L                            h                            2                                                                          +                                                                                (                                                              k                                -                                                                  ak                                  e                                                                                )                                                        2                                                                                L                              e                              2                                                                                a                              2                                                                                ]                                                                      )                                              +                                    N              2                                                  (              k              )                                                          (        8        )
where N1(k) and N2(k) are independent power spectra of zero-mean white noise processes, i.e.,                     ⟨        N            1              (        k        )              ⟩    =                              ⟨          N                2                  (          k          )                    ⟩        =    0.
A brief observation of Eqs. (7) and (8) reveals the centroid shift in the scattering spectrum resulting from the compression. In other words, if fe1 and fe2 are the center frequencies of the scattering spectrum before and after compression, respectively, and assuming that the speed of sound in the tissue c remains constant, from Eqs. (7) and (8) we have                                                                                f                  e2                                -                                  f                  e1                                            =                                                c                                      2                                        π                                                                                (                                                            ak                      e                                        -                                          k                      e                                                        )                                                                                =                                                      (                                          a                      -                      1                                        )                                                                          f                    e1                                                              ,                                                          (        9        )
or, from Eq. (3)                                                         f              e2                        -                          f              e1                                            f            e1                          ≅        s                            (        10        )
So, the relative centroid shift in the scattering function spectrum constitutes a direct strain estimator. A similar result is found later (Eq. 16) using the spectrum of the received signal. In the section below, we use Eq. (10) as a guide in the formulation of the new estimator.
Effect of strain on the spectrum of the received signal
The centroid of the power spectrum of the received signal is defined as follows                               f          c                =                              c                          2                            π                                                                                ∫                                      -                    ∞                                    ∞                                                                                kR                      1                                                                                (                      k                      )                                                                                ⅆ                    k                                                                                ∫                                      -                    ∞                                    ∞                                                                                R                      1                                                                                (                      k                      )                                                                                ⅆ                    k                                                                        .                                              (        11        )
The centroid estimate for the precompression power spectrum is given by:                               f          e1                =                              c                          2                            π                                                                                k                  h                                                                  L                  h                  2                                            +                                                k                  e                                                                  L                  e                  2                                                                                L                h                2                            +                              L                e                2                                                                        (        12        )
In a similar manner we can derive the expression for the centroid of the post-compression power spectrum by replacing ke, and Le, in Eq. (9) by their corresponding parameters in the post-compression power spectrum, i.e., ake and       L    e    a
(as indicated from Eq. 7) to obtain:                               f          c2                =                              c                          2                            π                                                                                k                  h                                                                  L                  h                  2                                            +                                                k                  e                                                                                L                    e                    2                                    a                                                                                L                h                2                            +                                                L                  e                  2                                                  a                  2                                                                                (        13        )
Note that the PSF parameters remain unchanged. Since both centroids depend on the center frequencies and bandwidths of the scattering function and the PSF and by consulting Eq. (10), we normalize this effect by using the following ratio as a candidate strain estimator:                                                         f              c2                                                  xe2x80x83                        -                          xe2x80x83                                                  f              c1                                            f            c1                                            xe2x80x83                =                  xe2x80x83                                                                                k                    h                                                                          xe2x80x83                                                                          a                    2                                                                          xe2x80x83                                                                          L                    h                    2                                                                                xe2x80x83                                +                                  xe2x80x83                                                                                ak                    e                                                                          xe2x80x83                                                                          L                    e                    2                                                                                a                    2                                                                          xe2x80x83                                                                          L                    h                    2                                                                                xe2x80x83                                +                                  xe2x80x83                                                                  L                  e                  2                                                                                xe2x80x83                        -                          xe2x80x83                                                                                k                    h                                                                          xe2x80x83                                                                          L                    h                    2                                                                                xe2x80x83                                +                                  xe2x80x83                                                                                k                    e                                                                          xe2x80x83                                                                          L                    e                    2                                                                                L                  h                  2                                                                  xe2x80x83                                +                                  xe2x80x83                                                                  L                  e                  2                                                                                k                  h                                                                  xe2x80x83                                                                  L                  h                  2                                                                          xe2x80x83                            +                              xe2x80x83                                                                            k                  e                                                                  xe2x80x83                                                                  L                  e                  2                                                                                L                h                2                                                          xe2x80x83                            +                              xe2x80x83                                                          L                e                2                                                                        (        14        )
The parameters Le and Lh are related to Be and Bh, the equivalent noise spectral bandwidths for the scattering and PSF spectra, through             B      h        =                            1                      2                                      π                                                  L              h                                                        xe2x80x83                        and                          xe2x80x83                                  B          e                    =              1                  2                                π                                          L            e                                ,
respectively.
However, the PSF bandwidth is typically much smaller than the bandwidth of the scattering function, i.e., Be greater than  greater than Bh; thereby, Le less than  less than Lh and, therefore, a2L2h greater than  greater than L2h. After cancellation of common terms in the numerator and denominator of Eq. (14), we obtain                                                                                f                    c2                                    -                                      f                    c1                                                                    f                  c1                                            ≅                              xe2x80x83                                                                                (                                          a                      -                      I                                        )                                                                                k                        e                                                                                L                        e                        2                                                                                L                        h                        2                                            +                                              L                        e                        2                                                                                k                        h                                                                                L                        h                        2                                                              +                                                                  k                        e                                                                                L                        e                        2                                                                                L                      h                      2                                        +                                          L                      e                      2                                                                                =                              xe2x80x83                                                                            (                                      a                    -                    1                                    )                                                                                k                        h                                                                                L                        h                        2                                                                                k                        e                                                                                L                        e                        2                                                                              +                  1                                                                                (        15        )
or, from the small strain approximation case of Eq. (3) (i.e., in mathematical terms, for strains less than 10%),                                                         f              c2                        -                          f              c1                                            f            c1                          ≅        As                            (        16        )
where A is given by                               A                                xe2x80x83                    =                      xe2x80x83                                          1                                                                                k                    h                                                                          xe2x80x83                                                                          L                    h                    2                                                                                k                    e                                                                          xe2x80x83                                                                          L                    e                    2                                                                                xe2x80x83                            +                              xe2x80x83                                          1
                                  or          ,                                    (        17        )                                A                          xe2x80x83                ≅                  xe2x80x83                                                            k              e                                                  xe2x80x83                                                  B              h              2                                                          k              h                                                  xe2x80x83                                                  B              e              2                                                          (        18        )
Inspection of Eq. (16) leads to the following interesting observations:
The relative spectral centroid shift can be used as a direct strain estimator. We can also observe a direct analogy between the classic definition of strain (Eq. (4)) and the estimator of Eq. (16), which establishes this method as a simple and straight-forward way of estimating the strain.
When the strain is positive (or, compressive), a frequency upshift occurs, i.e.,fc2xe2x88x92fc1 greater than 0. Conversely, a tensile (or, negative) strain results in a frequency downshift, i.e.,fc2xe2x88x92fc1 less than 0. Therefore, the estimator of Eq. (16) provides directly not only the magnitude of the strain but also its sign.
Since constant A is independent of the strain, it will introduce a uniform bias on the resulting elastogram. This should not affect the resulting elastogram, since the latter depicts relative values of strain. The reader should note that the effect of local bandwidth variations is ignored.
The scattering spectrum must be a bandpass and bandlimited spectrum in order to estimate the strain using the centroid estimator. Otherwise, if Be, is infinite and/or if ke is zero, constant A in Eq. (16) will always be zero regardless the strain.
For relatively larger strains, using the more general form of Eq. (3), Eq. (16) becomes                     f        c2            -              f        c1                    f      c1        ≅      A              s              1        -        s
or, by solving for the strain,                     s        ≅                                            f              c2                        -                          f              c1                                                                          (                                  A                  -                  1                                )                                                          f                c1                                      +                          f              c2                                                          (        19        )
which is a less straight-forward, but still a direct way of estimating higher strains.
Eq. (16) is reminiscent of the well-known Doppler effect, according to which the ratio of the centroid shift to the center frequency may provide a reliable measure of velocity. According to Eq. (16), in the case of strain, a similar effect also occurs. Also, among others have shown how in broadband Doppler the bandwidth of the resulting spectrum also changes with velocity and that the output RF Doppler spectrum is a frequency-shifted and compressed (or stretched) replica of the transmitted one. Similarly, in the case of strain measurement, that the following expression provides a direct estimation of strain:                                                         B              2                        -                          B              1                                            B            1                          ≅        ks                            (        20        )
where and are the post- and precompression bandwidths, respectively and k is a constant. So, strain, like velocity, introduces these two coupled effects of centroid shift and bandwidth variation in the power spectrum. Spectral broadening (i.e., in the case of compression) or contraction (i.e., in the case of tension) can introduce a bias in the measurement. A general expression is derived, linking the centroids fc1 and fc2 (before and after compression, respectively), the pure frequency shift (i.e., in case the pre and postcompression spectra are identical, only centered at different frequencies separated by a shift), xcex94f, and a bias term xcex2 denoting the spectral broadening (or, compression) due to strain s:
fc2xe2x88x92fc1=xcex94f+xcex2(s)xe2x80x83xe2x80x83(21)
where
xcex2(s)=B2(s)xe2x88x92B1 xe2x80x83xe2x80x83(22)
In order to estimate the strain without the bias associated with spectral broadening, the following equation can be used that results from Eqs. (16), (20), (21) and (22):
xcex94f≅Afc1sxe2x88x92kB1sxe2x80x83xe2x80x83(23)
and solving for strain, the unbiased estimator is given by                     s        ≅                              Δ                                      xe2x80x83                                    f                                              Af                              c                1                                      -                          kB              1                                                          (        24        )
However, Eq. (24) requires a bandwidth estimation and since the bandwidth estimator is not part of this study, we use Eq. (16) as the strain estimator and show a bias with simulations, which is partly due to the previously described bias due to spectral broadening.

Power spectral strain estimators in elastography

The Board of Regents of the University of Texas System


A test socket is provided for use in testing integrated circuits, especially integrated circuits in BGA packages. The test socket comprises a base member and a cover member configured to move vertically between an upper position and a lower position with respect to the base member. Springs are positioned between the base member and the cover member and configured to bias the cover member in the upper position. A lever is coupled to the base member and to the cover member and is configured to pivot to an open position when the cover member is in the lower position and to pivot to a closed position to confine an integrated circuit within the base member when the cover member is in the upper position.

Many of the integrated circuits (ICs) fabricated today are mounted in a ball grid array (BGA) package. The BGA package has a plurality of solder balls (often hundreds of solder balls) on its undersurface that are coupled to device terminals on the integrated circuit chip. In use, the solder balls are soldered to a printed circuit board to achieve the proper interconnection between the circuitry on the printed circuit board and the integrated circuit. Before any IC can be sold it must be tested to insure that the device meets the intended specification. Testing an IC in a BGA package requires holding the plurality of solder balls of the IC package against electrodes (often spring loaded electrical contacts known as POGO pins) that are connected to a testing apparatus. Holding the solder balls against the electrodes usually requires a test socket that precisely aligns the large number of small, closely spaced solder balls to the test electrodes. Present day test sockets are usually useful for only a single package size and type with a single predetermined array of solder balls. Thus if a device is mounted in a package of a different size or having a different array of solder balls, a new test socket must be used. Requiring a new test socket is both expensive and time consuming because of the lead time to design and acquire the new socket. Such sockets are also prone to damaging the solder balls and are not particularly amenable to automated insertion and unloading of the integrated circuits being tested.
Accordingly, it is desirable to provide an integrated circuit test socket that can be readily changed to accommodate different BGA package sizes and arrays of solder balls. In addition, it is desirable to provide an IC test socket into which it is easy to automatically insert devices for test and that does not damage the solder balls on the IC package. Furthermore, other desirable features and characteristics of the present invention will become apparent from the subsequent detailed description and the appended claims, taken in conjunction with the accompanying drawings and the foregoing technical field and background.


A melt-formulated, multiparticulate, oral dosage form containing clavulanic acid and/or at least one physiologically acceptable salt thereof and at least one sucrose fatty acid ester and optionally further physiologically acceptable auxiliary substances; a process for producing such a dosage form, combination dosage forms with β-lactam antibiotics, and the pharmaceutical use thereof, e.g., for treating bacterial infections.

The present invention relates to a melt-formulated, multiparticulate, oral dosage form containing clavulanic acid and/or one or more of the physiologically acceptable salts thereof and one or more sucrose fatty acid esters and optionally further physiologically acceptable auxiliary substances, to a process for the production thereof and to the use thereof.
Clavulanic acid, in particular in the form of potassium clavulanate, is frequently used in combination with a β-lactam antibiotic, such as for example amoxicillin, for the treatment of bacterial infections by Gram negative and Gram positive β-lactam-resistant pathogens. Due to the combination with clavulanic acid, which is capable of cleaving β-lactamases, these pathogens, despite initial resistance, again become sensitive to treatment with amoxicillin. This combination is used inter alia for the treatment of otitis media in children. Treatment involves the administration of different dose combinations of clavulanic acid to amoxicillin, with ratios of 1:4, 1:7 or 1:8 being the most common.
One major problem in the production of pharmaceutical preparations comprising clavulanate, preferably potassium clavulanate, is this compound's high susceptibility to hydrolysis. Even over the course of processing and storage of the compound under normal conditions (ambient temperature of 25° C. and 60% relative atmospheric humidity), increasing hydrolysis of the clavulanic acid occurs within a few hours to days, this being accompanied by an intense discoloration and release of CO2.
Decomposition of the compound is accelerated as temperature and humidity increase and furthermore proceeds autocatalytically, a cascade of different decomposition products possibly arising. This decomposition is described in greater detail in WO 94/16696.
As a consequence of this susceptibility to hydrolysis, when processing potassium clavulanate, which is conventionally already commercially available as a mixture with a grade of microcrystalline cellulose which is in a particularly desiccated form or with dry silicon dioxide, it is necessary both to exclude moisture (relative atmospheric humidity of the direct surroundings of less than 20%) and to keep the room or ambient temperature below 20° C.
Already known powders for suspension and film coated tablets are thus based on powder mixtures with silicon dioxide or on a dry compacted form of the clavulanate, wherein the auxiliary substances used must also be dried before use. When coating the tablets with a film, special low-moisture processes are used, as are for example described in WO 95/28927.
Since pellets are conventionally produced using aqueous solutions or with the assistance of lipophilic matrix materials, such a formulation method is not suitable for formulating mixtures containing clavulanate to yield pellets. Due to clavulanate's susceptibility to moisture, no aqueous solutions can be used and, due to the delaying action on active ingredient release which conventionally occurs, no lipophilic matrix materials may be used either. This is in particular contrary to the therapeutic goal, according to which rapid release and availability in the body are to be achieved.

Melt-formulated, multi-particulate oral dosage form


A mobile communication device for use in a packet switched wireless communications network includes a display, which is able to indicate an amount of data remaining from a subscribed amount. This allows the user to see how much data he can send or receive, within a prepaid subscription. The display is graphical, allowing the user to judge the amounts of data easily, as well as numerical. Advantageously, the mobile communication device calculates the amounts of data sent and received in such a way that it avoids double counting of retransmissions.

In a conventional second generation mobile communications system, for example operated under the Global System for Mobile Communications (GSM), a mobile subscriber is typically charged an amount which depends on the duration of his connection to the network. In the case of a prepaid phone, a subscriber pays in advance, and receives a corresponding credit.
EP-A-1030506 shows a prepaid mobile phone, which has a display, which displays the current credit value numerically on the display of the phone.
By contrast, in a third generation mobile communications system, for example operated under the General Packet Radio Service (GPRS) digital cellular telecommunications system, it is envisaged that subscribers will advantageously be able to maintain a permanent network connection. In this case, it may be inappropriate to charge the subscriber an amount which is dependent on the duration of the connection to the network. Rather, it may be more appropriate to make a charge which is related to the amount of data exchanged between the subscriber and the network.
One possibility is that a network operator may make a basic charge, for example monthly, with the subscriber being allowed to send or receive a specified amount of data within that basic charge.

Method and device for indicating amounts of data in mobile networks


A stop unit stops transmitting data to a plurality of nodes for every predetermined period. An acquisition unit acquires versions of routing tables, which are updated in accordance with movement of a query, from the plurality of nodes when the transmission of the data is stopped. A comparison unit compares the versions of the routing tables of the plurality of nodes that are acquired. When there is a node in which the routing table of an old version is stored as a result of the comparison, an update unit updates the routing table of the node.

Recently, as a technique for processing many sets of data that are collected every moment from various targets in a parallel manner, CEP (complex event processing) has been known. In the complex event processing, an event is detected from received data, and processing relating to the detected event is performed. Although there is when the complex event process is called ESP (event stream processing), here, the complex event processing together with the ESP will be referred to as CEP.
In a CEP system that performs the complex event processing, there is a case where a large amount of received data is temporarily processed, and, in such a case, the processing load increases, and the processing performance may decrease. In addition, since the CEP system is a system that processes data in real time, the CEP system is expected to be always running, and is not allowed to be stopped.
Thus, in the CEP system, by using a system technique, in which resources can be flexibly allocated, used for a cloud or the like, the processing is divided into a plurality of servers or virtual machines (VMs) in accordance with a variation in the processing load. For example, a transaction requesting sentence called a query or data accompanied with a query that is arranged at a server or a virtual machine, which has a high processing load, is moved to another server or another virtual machine as a processing element, whereby the processing is divided.
However, in a CEP system, there is a possibility that the data order changes in a case where the movement of a processing element occurs. For example, it is assumed that a data transmission source server is A, a movement source server of a processing element is B, and a movement destination server of the processing element is C. Here, a case will be considered in which the transmission source server A transmits data to the movement source server B, and a processing element is moved from the movement source server B to the movement destination server C. Data transmitted from the transmission source server A to the movement source server B at timing when the processing element is moved from the movement source server B to the movement destination server C arrives at the movement source server B. Since the processing element has moved to the movement destination server C, the movement source server B transmits the arrived data to the movement destination server C. When the movement of the processing element has been completed, the movement destination server C notifies the transmission source server A of movement completion of the processing element. After the notification of the movement completion, the transmission source server A transmits data directly to the movement destination server C. In such a case, there is a possibility that the data that has been directly transmitted from the transmission source server A to the movement destination server C arrives at the movement destination server C before the data transmitted from the transmission source server A to the movement destination server C through the movement source server B.
When the data order changes as above, there is a case where a represented meaning changes in accordance with the data order. For example, a case will be considered in which a security system, in which a door sensor detecting the unlocking of a front door of a house using a key and a motion sensor detecting a person present indoor are arranged, and data transmitted from the door sensor and the motion sensor is monitored, is realized by a CEP system. In such a case, when the order of data of the detection of opening or closing of the front door using a key, which is acquired by the door sensor, and data of the detection of an indoor motion, which is acquired by the motion sensor, changes, the meaning changes. For example, a case where an object is detected indoor after the detection of unlocking the front door using a key can be regarded as owner's return to the house. On the other hand, a case where the unlocking the front door using a key is detected after detection of an indoor object can be regarded as a thief's flight.
Thus, a technique has been proposed in which the data order is maintained even when the movement of a processing element occurs. In this technique, for example, when a movement source server and a movement destination server of a processing element are determined, a controller that manages a CEP system transmits a movement request to the movement source server and the movement destination server. The movement source server sends pause requests to all the transmission sources that transmit data to the movement source server and waits for ack for the pause requests, thereby locking the transmission of data in the transmission source and buffering data of the processing element as a movement target in the transmission source. When the movement source server receives ack from all the transmission sources, there is no data that is in the process of transmission. The movement source server processes data until ack is received from all the transmission sources. Then, when ack is received from all the transmission sources, the movement source server starts the movement of the processing element. After the movement of the processing element is completed, the movement destination server transmits a restart request to all the transmission sources. When the restart request is received, the transmission source changes the transmission destination of the data to the movement destination server by changing a routing table, restarts the transmission of data, and transmits the buffered data to the movement destination server.
Patent Document 1: Japanese Laid-open Patent Publication No. 2011-039820
Non patent Document 1: Mehul A. Shah, Joseph M. Hellerstein, Sirish Chandrasekaran and Michael J. Franklin, "Flux: An Adaptive Partitioning Operator for Continuous Query Systems", ICDE, 2003
However, according to the related arts, when the processing element is moved, the movement source server stops the transmission of data in the transmission source by transmitting a pause request to all the transmission sources. Therefore, according to the related arts, there is overhead of stopping data processing for every movement of the processing element. More specifically, when many processing elements are moved, data processing is stopped every time when each processing element is moved, and accordingly, the overhead increases.

Server device, computer-readable storage medium, and method of assuring data order


A device is provided for supporting at least part of the weight of a snowboard on a boot of a snowboarder while sitting in a chairlift having no footrest, the snowboard having a boot binding for receiving and securing the boot onto the snowboard, where the device includes an elongated support member having a central portion, and fasteners at both ends, one for joining to the boot, and the other for joining to the boot binding so that the elongated support member is operable to transfer at least a portion of the weight on the boot binding to the boot. Other techniques are described that incorporate the foregoing back foot support device directly into the boot binding, by, for example, the boot binding manufacturer.

A large percentage of chairlifts on ski and snowboard mountains do not have rests for the rider's feet while the rider is riding the chair up the mountain. This creates a major inconvenience for a chairlift rider using a snowboard. When a chairlift rider is wearing a snowboard, the snowboard is only attached to the front foot. The orientation of the foot across the snowboard is close to perpendicular. The front foot is not centered on the board and an uncomfortable twisting load is transferred to the foot and subsequently the ankle and knee of the rider. This is not a major inconvenience for skiers because a skier has one ski on each leg and the weight is evenly distributed.
This problem has been felt since the inception of snowboarding. Known approaches to solve this problem include different types of tethers to the board that are either connected to the rider or attach to the lift, and features added to the snowboard to support the weight of the snowboard with the rear foot. Tethers can be complicated and uncomfortable to attach to the body when that is the design. Tethers can also be difficult and dangerous to hook and unhook while on the chairlift. It is also an extra part to carry around. One known feature that may be added to the snowboard is a base plate that is mounted between the binding and the board. Attached to the base plate is a hook that the rider can use to support the load of the board with their back foot. This design can affect the performance of the snowboard by changing the stiffness due to the added attachment plate. The plate also raises the height of the back binding. Further, this relatively large and complex assembly would not be inexpensive to manufacture and would be difficult to assemble. Some of the other designs require adding holes to the snowboard as part of the design. This is undesirable on many levels and would void the warrantee of the snowboards.
In view of the foregoing, there is a need for improved techniques of snowboard design, which enable a chairlift rider to support the weight of the snowboard with his back foot while not affecting the performance of the snowboard.
Unless otherwise indicated illustrations in the figures are not necessarily drawn to scale.


A capacitor embedded printed circuit board (PCB) includes a multilayer polymer capacitor layer with a plurality of polymer sheets. One or more first inner electrodes and second inner electrodes, separated by one or more of the plurality of polymer sheets, are alternately disposed to form a pair. A plurality of first extended electrodes and second extended electrodes protrude from the first inner electrodes and second inner electrodes, respectively. One or more insulating layers are laminated on one or both surfaces of the multilayer polymer capacitor. A plurality of first via holes for capacitor, and a plurality of second via holes for capacitor, penetrating the multilayer polymer capacitor layer are connected to the first extended electrodes and the second extended electrodes, respectively. The plurality of the first and second extended electrodes are alternately disposed to be opposite to each other.

1. Field of the Invention
The present invention relates to a capacitor embedded printed circuit board (PCB), and more particularly, to a capacitor embedded PCB, capable of decreasing an inductance by controlling an electrode structure of the embedded capacitor and of easily embodying high capacitance for unit area by controlling a number of multilayers and an area of electrodes.
2. Description of the Related Art
In general, a capacitor is a device storing energy as the form of charges and has properties, in which charges are stored and a current does not flows in the case of a direct current (DC) power source and a current flows in proportion to a change in voltage according to a capacitance of the capacitor and time while the capacitor is charged/discharged in the case of an alternating current (AC).
Due to such properties, capacitors are essential passive devices used for coupling and decoupling, filtering, impedance matching, charge pumping, and demodulation in such as digital circuits, analog circuits, and electric electronic circuits such as high frequency circuits and generally manufactured in various forms such as chips and disks and mounted on a surface on a PCB to be used.
However, as electronic devices become miniaturized and complicated, an area for mounting passive elements on a PCB is reduced. Also, as speed of electronic devices becomes high, frequency is increased. Thereupon, parasitic impedances occurring due to various causes such as a conductor or solder between passive devices and an integrated chip (IC) cause various problems. To solve problems, various attempts to embed a capacitor in a PCB are made.
Up to now, on a surface of general PCBs, a general discrete chip resistor or a general discrete chip capacitor is mounted. However, recently, PCBs with a passive device such as a resistor or capacitor, embedded therein, have been developed.
Passive device embedded PCB technology indicates that a passive device such as a resistor or capacitor is inserted into an inner layer of a board by using new materials and processes and replaces a conventional chip resistor or chip capacitor. That is, a PCB with a passive device embedded therein has the passive device, for example, a capacitor embedded in an inner layer or the outside of the board itself. Regardless of a size of the board, when the capacitor that is a passive device is integrated into the PCB as a part, the capacitor is called as an "embedded capacitor" and the PCB is called as an "capacitor embedded PCB". Most important feature of the capacitor embedded PCB is that there is no need to mount a capacitor on a surface of a PCB since the capacitor is provided as a part of the PCB.
To manufacture a wireless terminal miniaturized while having various functions, it is required to use a higher RF bandwidth. According to this, self resonance frequency of a passive device is required to be higher. Also, in the case of a decoupling capacitor used for power source stabilization, it is required to reduce impedance at high frequency.
FIG. 1 is a cross-sectional view illustrating a conventional capacitor embedded PCB.
Referring to FIG. 1, a capacitor includes inner electrodes 14 and 17 and a dielectric layer 11 interposed between the inner electrodes 14 and 17. The inner electrodes 14 and 17 are connected to positive and negative electrodes via via holes 16 and 19, respectively. In the capacitor embedded PCB, directions of currents passing through an anode and cathode of the capacitor embedded in the PCB are allowed to be opposite to each other to compensate parasitic inductance occurring due to the current, thereby reducing entire parasitic inductance.
However, in the case of PCB where a conventional capacitor is embedded, it is advantageous to reduce inductance but a capacitance value thereof is too small to be actually used.

Capacitor embedded printed circuit board


Method and apparatus for sensing the displacements of micromachined devices and sensors. The method is referred to as the enhanced modulated integrative differential optical sensing (EMIDOS). The target micromachined proof-mass, for which displacements are measured, includes a grid of slits. The micromachined device is bonded to a CMOS chip containing a matching photodiodes array and their readout electronics. The grid is aligned with the photociiodes. An illumination source, such as an LED, is then mounted above the micromachined device. A model for the noise equivalent displacement (NED), including mechanical, electrical and optical domains, as well as all noise sources is derived. The model predicts that displacements below 10−3 [√{square root over ( )}Hz] can be measured. The design comprises innovative inertial sensors, an accelerometer and a rategyroscope employing the EMIDOS. Performance models for the noise equivalent acceleration (NEA) and noise equivalent rate (NER) are also derived. The models show that an accelerometer with a very low NEA can be realized.

For several decades microelectronics progressed in decreasing size while simultaneously increasing complexity. Mechanical structures can now be fabricated to comparable dimensions, and can be highly integrated with the electronics into microelectromechanical systems (MEMS).
MEMS can now be realized as sensors, actuators or mechanical structures. Products using MEMS are used in such life-saving devices as airbag accelerometers and disposable blood-pressure transducers to monitor the heart rate. MEMS devices have become more complex, and have incorporated an increasing number of features, progressing from components to electronic systems. The trend is measured in terms of the level of integration of MEMS mechanic and electronics.
The motion of the micro mechanical elements in micro-sensors is generally proportional to a particular physical measurement of interest, such as acceleration or partial pressure. Therefore it is important to sense this motion, in-order to reproduce the physical measure. In devices of more general application, the motion sensing may be used for feedback control, for example, to restore the micro-machined device to its original physical location. In high-grade sensors, such as inertial grade gyroscopes and accelerometers, as well as high-grade microphones, the sensing of motion in the sub-picometer (pm) range is required. Moreover, the motion sensing apparatus, itself, should be small and integrative, in-order to function effectively in the dimensions of the micro-machined devices.
Although the same laws of physics apply in three-dimensional structures, in the micron and sub-micron scales, nonlinear effects that are generally negligible in the macro-environment become more significance. Generally, micro-machined proof-masses can move in six degrees of freedom: three along linear axes: x,y and z; and three angles of rotation about these axes.
The main factor that limits the minimum detectable displacement is the noise sources that sensing systems are susceptible to. Spurious noise sources include mechanical noise, electrical noise, light noise, etc. Noise sources can be either "white" noise, that is frequency independent, or frequency dependent, for example "1/f" noise that is inversely dependent on frequency.
Many motion sensing techniques are known in the art of micro-machined sensors and devices, as described in the following background references:    [1] J. A. Plaza, H. Chen, J. Esteve and E. Lora-Tamayo, "New bulk accelerometer for triaxial detection", Sensors and Actuators A: Physical, Vol. 66, 1998, pp. 105–108.    [2] R. Voss, K. Bauer, W. Ficker, T. Gleissner, W. Kupke, M. Rose, S. Sassen, J. Schalk, H. Seidel and E. Stenzel, "Silicon angular rate sensor for automotive applications with piezoelectric drive and piezoresistive read-out", Proc. of Transducers'97, Chicago, 16–19 Jun. 1997, pp. 879–882.    [3] Y. Nemirovsky, A. Nemirovsky, P. Murlat and N. Setter, "Design of a novel thin-film piezoelectric accelerometer", Sensors and Actuators A: Physical, Vol. 56, 1996, pp. 239–249.    [4] B. E. Boser and R. T. Howe, "Surface micromachined accelerometers", J. of Solid-State Circuits, Vol. 31, No. 3, March 1996, pp. 366–375.    [5] M. Weinberg, J. Connelly, A. Kourepenis and D. Sargent, "Microelectromechanical instrument and systems development at the Charles Stark Draper Laboratory, INC.", Proceedings of the IEEE Digital Avionics Systems Conference, 1997, pp. 8.5–33–8.5–40.    [6] H. K. Rocksatd, T. W. Kenny, J. K. Reynolds, W. J. Kaiser and T. B. Gabrielson, "A miniature high-sensitive broad-band accelerometer based on electron tunneling transducers", Sensors and Actuators A: Physical, Vol. 43, 1994, pp. 107–114.    [7] R. L. Kubena, D. J. Vickers-Kirby, R. J. Joyce and Frederick P. Stratton, "A new tunneling based sensor for inertial rotation rate measurements", JMEMS, Vol. 8, no. 4, December 1999, pp. 439–447.    [8] U. A. Dauderstadt, P. H. S. de Vries, R. Hiratsuka and P. M. Sarro, "Silicon accelerometer based on thermopiles", Sensors and Actuators A: Physical, Vol. 46–47, 1995, pp. 201–204.    [9] O. Degani, "Investigation of Microelectromechanical Systems employing Modulated Integrative Differential Optical Sensing", M. Sc. Thesis, Supervised by Y. Nemirovsky, Technion, 1999.    [10] T. Storgaard-Larsen, S. Bouwstra and O. Leistiko, "Opto-mechanical accelerometer based on strain sensing by bragg grating in a planar waveguide", Sensors and Actuators A: Physical, Vol. 52, 1996, pp. 25–32.    [11] G. Schopfer, W. Elflein, M. de Labachelerie, H. Porte and S. Ballandras, "Lateral optical accelerometer micromachined in (100) silicon with remote readout based on coherence modulation", Sensors and Actuators A: Physical, Vol. 68, 1998, pp. 344–349.    [12] T. B. Gabrielson, "Mechanical-Thermal Noise in Micromachined Acoustic and Vibration Sensors", IEEE Trans. On Elec. Dev., Vol. 40, No. 5, May 1993.    [13] M. Bao, H. Yang, H. Yin and S. Shen, "Effects of electrostatic forces generated by the driving signal on capacitive sensing devices", Sensors and Actuators A: Physical, Vol. 84, 2000, pp. 213–219.    [14] S. P. Timoshenko, I. N. Goodier, Theory of Elasticity, McGraw-Hill, New-York, 1970.    [15] D. D. Lynch, "Coriolis vibratory gyros", Proc. of GYRO technology symposium, Stuttgart, Germany, 15–16 Sep. 1998, pp. 1.0–1.14.
As described in the above references, the known motion sensing techniques using micro-machined sensors and devices include: piezoresistive [1,2], piezoelectric [3], capacitive [4,5], tunneling [6,7], thermal [8] and optical [9-11] sensing. Few methods have shown the capability for sensing motions in the sub-picometer range.
The most sensitive method, so far, is based on the tunneling effect between a sharp tip and a facing electrode. It has been shown theoretically [12] that the Noise Equivalent Displacement (NED) of this method, at the medium frequencies range, is in the order of 10−2–10−3[pm/√Hertz] (10−4–10−5[_/√Hertz]) (Hz). Nevertheless, the tunneling effect suffers from inherent "1/f" noise, up to the range of a few Kilohertz (KHz), and its apparatus is quite difficult to realize. Moreover, due to the close proximity required between the tip and the facing electrode, of the order of a few _ (Angstrom units), the sensing is adversely affected by a relatively high damping coefficient and thus a high thermal-mechanical noise.
Another technique, which is the one most commonly used in micro-machined devices, and which have shown to be sufficiently sensitive, is capacitive sensing. This method is rather simple to realize and is highly integrable. Capacitive sensing does not suffer from inherent noise sources. The noise of the capacitive transducer is mainly contributed by the electronic readout circuits or by thermal-mechanical noise. Therefore, by proper design of the electronic readout, capacitive sensing is limited only by thermal-mechanical noise, and theoretically a NED of 10−2[pm/√Hz] (10−4[_/√Hz]) can be achieved.
Capacitive sensing, on the other hand, suffers from possible cross talk with the electronic readout signals, which may exert a parasitic force on the device [13]. Moreover, without proper shielding it may suffer from Electromagnetic Interference (EMI). Due to the close proximity between the capacitor plates, which is required to achieve high sensitivity, the capacitive transducer is subjected to a rather high damping coefficient, which results in a higher thermal-mechanical noise. To lower the damping, vacuum encapsulation is usually required for high-grade sensors.
More recently, a piezoresistive configuration with a high degree of symmetry was reported to yield a NED in the order of a few [pm/√Hz], which is not quite low enough, but is rather near to the order required [14]. The piezoresistive method is quite easy to realize, and integrate, and was one of the first sensing methods to be used in micro-sensors. Moreover, it does not require close proximity between surfaces, and thus, benefits from a low damping coefficient and low thermal-mechanical noise. Nevertheless, it suffers from a high temperature coefficient, which limits the micro-sensor's performance.
Certain aspects of this problem are addressed in the above-referenced co-pending Israeli Patent Application No. 122,947, which is assigned to the assignees of the present patent application and discloses a viable system to sense sub-angstrom displacements. As with the capacitive method, MIDOS [9] does not suffer from inherent 1/f noise. As an improvement over capacitive methods, MIDOS is neither susceptible to EMI nor to cross talk from readout circuits. The integration of the motion sensing elements, the photodiodes, and the readout electronics on the same chip also reduces the system noise. Moreover, the sensing is based on in-plane motion, and close proximity is not required between the mechanical elements. Thus, the damping coefficient is lower, and the thermal-mechanical noise, also, is at lower vacuum levels than those required for the capacitive transducer. Nevertheless, the NED of the MIDOS technique is still far from achieving the demands of high-grade sensors.
No other sensing technique has as yet shown a NED lower than 1[pm/√Hz].
Thus there is a need for a micro-machined sensor capable of detecting motion in the sub-picometer range, and without the drawbacks of prior art devices.

Method and apparatus for micro-machined sensors using enhanced modulated integrative differential optical sensing

Technion Research & Development Foundation Ltd.

Rafael - Armament Development Authority Ltd.


An FM broadcast transmitter transmits a broadcast signal having a carrier at a broadcast frequency and sidebands, able to be transmitted at full power, within a transmission bandwidth around the carrier. It includes a source of a modulated FM stereo signal having a carrier at the broadcast frequency and having sidebands with a bandwidth less than the transmission bandwidth representing a stereo signal. It also includes a source of a modulated in-band-on-channel (IBOC) signal, having carrier pulses spaced relative to each other to represent the IBOC digital data signal encoded as a variable pulse width encoded signal, and a bandwidth within the transmission bandwidth not overlapping the FM stereo signal sidebands. A signal combiner combines the modulated FM stereo signal and the modulated IBOC signal to form the broadcast signal.

In the United States, FM broadcasters can transmit information in sidebands within 100 kHz of their assigned carrier frequency at full power, and from 100 kHz to 200 kHz around the carrier at 30 dB down from full power. The standard stereo audio signal is placed in a bandwidth within 53 kHz of the carrier. The broadcaster is, thus, able to transmit other information in the remainder of the bandwidth, subject to the constraints described above.
It has become desirable for FM broadcasters to simultaneously broadcast stereo audio and digital data. The digital data could, for example, represent a high quality version of the stereo audio being broadcast. This requires a relatively high data rate channel which is restricted to a relatively narrow bandwidth. For example, a digital data stream carrying high quality audio can have a bit rate of 128 kilobits per second (kbps). A signal carrying such a data stream cannot be transmitted in the bandwidth available in an FM broadcast signal without some form of compression to decrease the bandwidth required for the signal.
It is always desirable to provide data at higher data rates through channels which have limited bandwidth. Many modulation techniques have been developed for increasing the data rate through a channel. For example, M-ary phase shift keyed (PSK) and Quadrature Amplitude Modulation (QAM) techniques permit compression by encoding a plurality of data bits in each transmitted symbol. Such systems have constraints associated with them. First, the hardware associated with such systems is expensive. This is because these techniques require a high level of channel linearity in order to operate properly. Consequently, extensive signal processing must be performed for carrier tracking, symbol recovery, interpolation and signal shaping. Second, such techniques are sensitive to multipath effects. These effects need to be compensated for in the receiver. Third, these systems often require bandwidths beyond those available in some applications (for example in-band on-channel broadcast FM subcarrier service) for the desired data rates.

In-band-on-channel broadcast system for digital data


A painting wagon carries a plurality of components to be painted through a painting facility. The wagon includes a component supporting assembly having a plurality of opposed pairs of component mounting pins fixed thereto. The component supporting assembly is supported by a base structure having wheels which permit the painting wagon to be translated. The component supporting assembly may be selectively adjusted between a contracted configuration in which each of the pairs of opposed mounting pins, and thus the components mounted thereon, are oriented in contracted proximity to one another, so as to minimize the overall size of the wagon with the components mounted thereon to accommodate given dimensional limitations of the painting facility and an expanded configuration in which the pairs of opposed mounting pins, and thus the components mounted thereon, are oriented so as to be expanded apart from one another so that individual components may be mounted and dismounted onto the component mounting pins with reduced danger of bumping an adjacently mounted component and thus damaging the surface finish of the components.

1. Field of the Invention
The present invention relates to a painting wagon for carrying a plurality of components to be painted through a painting facility and more specifically to an expandable painting wagon which is adjustable between an expanded configuration for facilitating convenient loading and unloading of the components thereon and a contracted configuration for increasing the number of components that can be carried per unit length of the wagon. The present invention further relates to a method for simultaneously painting a plurality of such components.
2. Background Information
Certain body panels used in motor vehicles are painted separately from the remainder of the motor vehicle to which the panels are to be attached. Such panels are often painted by mounting a plurality of the panels on a wagon, moving the wagon through a painting facility, and applying paint to the panels as they are moved through the facility. The painting facility may include, for example, a multiple stage washer, booths for color coat application, booths for clear coat application, cure ovens, and a cool down section. Individual wagons may be moved through the paint facility by a wagon propelling mechanism, such as an in-ground chain conveyor which pulls the paint wagons along at a predetermined spacing.
The costs of painting a particular component are significantly impacted by the throughput (i.e., the number of components that can be painted per unit time or per wagon). This is largely affected by the number of components that can be mounted per unit length on the wagon.
It should also be noted that the layout and construction of the paint facility places restrictions on the wagon size. More particularly, physical limitations of the cure oven width and height, tunnel radius, spacing between wagons, etc. define limitations for the wagon size. In addition, the track along which the wagon is translated may be oval or rectangular, thus requiring sharp turns which mandate a compactness for the wagon not necessarily required for the straighter portions of the track. These dimensional limitations further impact throughput. It is thus desirable to maximize the number of components which are mounted per unit length on the painting wagon, especially at certain areas within the painting facility that have such dimensional limitations.
Traditionally, one wagon design has been used for all components. The wagon includes a translating base which carries a removable top portion. The top portion includes multiple mounting pin locations, which enable the components to be mounted on the top portion with desired spacing therebetween. The number of components which can be mounted on the wagon is limited by the component size and by the spacing between components. To maximize the number of components per unit length of the wagon, the components are mounted on the wagon as close as possible to one another.
In certain instances, the conventional method of mounting components as close as possible to one another does not take advantage of all available space on the painting wagon. More particularly, in some instances, only a portion of a body panel is to be painted a given color. For example, plastic fascias for certain automobiles are two toned, and therefore, only one-half of the fascia is painted while the other half is masked. In this situation, it would be desirable for the components to be nested such that the portion of each component to be painted overlaps the portion of the adjacent component that is not to be painted or masked to thereby maximize the number of components per unit length of the wagon. Nesting of components in such fashion, however, poses problems in that with the components mounted so closely to one another, it is difficult to load the components onto the cart before painting and unload the components off the cart after painting without bumping the components into one another and thus damaging the surface finish of the components.
In other situations, where it is desirable for the entire component to be painted so that nesting of components during painting is not practical, it may, nevertheless, be desirable for the components to be nested or overlapped on a shorter length wagon while the components are not being painted to enable the wagon to more easily negotiate tight turns during its travel through the painting facility.
From the above, it can be appreciated that there exists a need for a painting wagon that enables the components to be nested on the wagon during certain periods of a painting cycle and to be spaced further apart during other periods of the painting cycle. It is, therefore, an object of the present invention to satisfy this need. To accomplish this object, the present invention provides a painting wagon for carrying a plurality of components to be painted and translating the plurality of components through a painting facility, the painting wagon comprising: a component supporting assembly having a plurality of component mounting structures each constructed and arranged to permit an associated one of the plurality of components to be mounted thereon, so that said components can be mounted in side by side relation to one another along a given direction, the component supporting assembly being constructed and arranged to be movable between: (1) an expanded configuration in which the component mounting structures are positioned in spaced apart relation with respect to one another along the given direction so that a number of components are mounted per unit length in the given direction; and (2) a contracted configuration in which the component mounting structures are positioned in closer proximity to one another relative to the spaced apart relation so that at least one of the components mounted thereon has a portion thereof disposed in overlying relation with respect to an adjacent one of the components and so that the number of components mounted per unit length in the given direction is greater than when the component supporting assembly is in the expanded configuration; an actuating assembly operatively connected with the component supporting assembly and constructed and arranged to move the component supporting assembly between the contracted configuration and the expanded configuration so that the component supporting assembly is 1) positioned in the expanded configuration to facilitate mounting of the components on the component mounting structures, 2) positioned in the contracted configuration while the painting wagon is translated through at least a portion of the painting facility, and 3) positioned in the expanded configuration to facilitate dismounting of the components from the component mounting structures; and a base structure constructed and arranged to support the component supporting assembly and the actuating assembly and to permit translation of the painting wagon through the paint facility.
It is also an object of the invention to provide a method of painting a portion of each of a plurality of components. The method comprises a method for simultaneously painting a portion of each of a plurality of components comprising: providing a mask to cover a portion of each of the components so as to prevent paint from being applied to the portion; mounting the components in spaced relation to one another on a paint wagon; moving the components into closer proximity to one another relative to the spaced relation so that at least one component has an unmasked portion thereof to be painted disposed in overlying relation with respect to a masked portion of an adjacent one of the components; applying paint to said components so that the unmasked portions thereof are painted; moving the components back to the spaced apart relation to one another; and removing the components from the paint wagon.
Other objects, features and characteristics of the present invention along with the method of operation and assembly will be appreciated from study of the following detailed description and the appended claims and drawings, all of which form part of this application and wherein like reference numerals designate corresponding parts in the various figures.


A four way 2:1 bandwidth RF splitter/combiner is described. When used as a splitter, the splitter/combiner provides equal amplitude output signals while maintaining quadrature phase over the entire bandwidth of the input signal. This splitter/combiner also maintains a one to one VSWR and eliminates back door intermodulation. When used as a combiner, the splitter/combiner losslessly combines four equal amplitude quadrature phase signals.

a. Technical Field
The present invention pertains to electrical power splitters or combiners. More particularly, the invention pertains to splitters or combiners which operate to split or combine 2:1 bandwidth RF signals. Because a splitter of the present invention may be operated as a combiner, in which case the splitter outputs become combiner inputs and the splitter input becomes a combiner output, when the following discussion refers to a splitter of the present invention the discussion also implicitly refers to a combiner.
b. Problems in the Art
Electrical signals often must be divided and/or combined. For example, the power output requirements for an RF signal ma exceed the capability of readily available RF amplifiers; and to produce the required power output, the RF signal is divided and delivered to multiple amplifiers. The amplifiers' outputs are then combined to provide a power output which none of the amplifiers could have produced individually.
Quadrature hybrid couplers (couplers) are often used in power splitters. The couplers have two inputs and two outputs, one of which inputs is connected to a termination resistance matched to the system characteristic impedance (typically 50 ohms for RF signal applications). By terminating this input in this fashion, reflections at the other input are eliminated and a one-to-one VSWR is maintained. Applying a signal to the other input of the coupler produces signals at the two outputs of the coupler, each of which contains approximately half the power from the input signal.
At one output, the 0.degree. or AC-coupled output, the phase relative to the input signal is 0.degree. and at the other output, the -90.degree. or DC-coupled output, the relative phase is 90.degree. , which is inherently characteristic of a 3 db 90.degree. coupler. Additionally, although nominally half the power is delivered to each output, the amplitude response of the coupler varies according to the frequency of the input. That is, the outputs do not each have exactly one-half the power of the input signal. The frequency-dependent amplitude characteristics of the coupler outputs are illustrated in FIG. 3 of the appended drawings. Notice that one output contains more than half the input signal and the other output, complementarily, less than half the input over a frequency range of operation. This imbalance at the outputs is typically no greater than .+-.0.4 db at most. The 0.degree. output will typically have a maximum amplitude output of -2.6 db located at the center frequency, and the -90.degree. output normally has a minimum amplitude output of -3.4 db at the center frequency.
When one coupler drives two other couplers, thereby creating a four-way power divider, the imbalance at the four outputs of the two driven couplers is typically .+-.0.8 db. Two of the outputs are normally balanced at approximately -6.0 db (1/4 the input power), but the other two outputs are unbalanced. One output is typically at -5.2 db and the other output is at -6.8 db (in contrast to the nominal, desired -6.0 db).
If these divided signals were then sent to four amplifiers for amplification, one of the amplifiers would be presented a signal at approximately -5.2 db; that is approximately 30 % of the input signal power, rather than the desired 25 %. Because it is best to use identical amplifiers, each amplifier would have to be sized to handle 30 % of the input signal value. This requirement obviously limits amplifier selection, requires greater amplifier capacity, and reduces reliability due to the fact that one amplifier is carrying an excess burden that should, ideally, be shared among four amplifiers. This amplitude imbalance and its concomitant demands on amplifier capacity, reduced selection, and reduced reliability is the major shortcoming of prior art splitters.
Adding a fourth coupler (see FIG. 4), balances the two unbalanced outputs, thereby solving the amplitude imbalance problem of the prior art four-way splitter. Unfortunately, there are phase errors associated with this solution which, until the present, have never been addressed. These phase errors contribute to amplitude errors which are significant enough to negate the amplitude enhancement when the four amplified signals are recombined.
It is therefore an object of the present invention to provide a new and improved splitter which exhibits no amplitude imbalance at the output of the splitters, while, at the same time, eliminating phase errors which have heretofore cancelled the beneficial effects of a four-coupler splitter. These and other objects, features, and advantages of the present invention will become apparent from the specification and claims.


An improved walking aid tip that is easy to use and provides for better support on smooth or slippery surfaces. The tip combines the features of an enlarged base for better stability on sand and gravel, a combination of convex and flat base to provide support for the user when the walking aid is positioned at a wide variety of angles with the ground surface, where the user is walking or standing still, a base made of a non-flexible rubber material so that the cane does not wobble and become unstable when in use and so that the base does not crack easily and have to be replaced frequently, an enlarged rigid support member that entirely covers and supports the rubber base so that the cane is less apt to wobble and so that the base is less likely to crack and have to be replaced prematurely, and a socket, a threaded screw, and a threaded screw hole to allow for easy installation of the tip. The enlarged base also allows people who would normally have to use a more cumbersome quad cane (e.g., with four heads) to use a more streamlined but yet supportive single head cane tip.

Tips for ambulatory aids such as canes or crutches in use today are conventionally molded from rubber to have a flat or concave bottom surface about one and one-half to two inches in diameter. I have discovered that these conventional cane and crutch tips perform poorly for several reasons. First, because they are made of molded flexible rubber, conventional tips tend to slip on smooth floors, wet surfaces, and loose surfaces such as gravel and sand, resulting in a potentially dangerous situation for the user. Second, because these conventional tips are often flat bottomed, they must be placed precisely flat on the surface of the floor in order to provide stable support to their user. Third, I have found that conventional tips are often too small to provide adequate support on a ground surface covered by a loose material such as snow, sand or gravel. Conventional tips often sink several inches into loose sand and cause the user to lose balance or even fall, and I have noticed that many individuals have foregone the use of a cane in favor of a more stable, albeit cumbersome walker. Fourth, I have found that because conventional cane tips use a molded flexible rubber material, they tend to become unstable and wobble when in use. Fifth, because the conventional tips are made of molded flexible rubber and because there is little or no rigid material supporting these rubber tips, the conventional tips have a very limited life as they tend to crack easily with use. Thus, conventional cane tips often have to be replaced quite frequently.
There have been several efforts to overcome some of the above deficiencies. The Crutch Tip Construction of Urban, U.S. Pat. No. 4,630,626 proposes a tip made of a resilient, elastomeric rubber, an enlarged tip, and a rigid plastic or metallic socket for receiving the cane or crutch. A later effort, Wilkinson, U.S. Pat. No. 4,899,771 for a Walking Aid discloses a rocker shaped tip for a cane having a large rectangular bottom surface area and a pair of curved edges to permit the user to permit limited tilting of the cane or crutch during walking. Another effort, U.S. Pat. No. 4,098,283 for Specialized Crutch Tips to Tritle, Jr. provided a cane tip with a disk having a convex lower surface surrounded by a flat peripheral disk, or ride, to cut into and grab sand to prevent slippage. More recent designs such as the Radial Crutch Tip Assembly shown in U.S. Pat. No. 5,409,029 to Davis and the Walking Cane Usable on Slippery and Icy Surfaces in U.S. Pat. No. 5,301,704 to Brown rely upon a curved lower surface generally configured in the shape of a knob allowing the user to slightly tilt their appliance while walking. I have discovered however, that in use these knob-like tips tend to restrict the motion of the appliance to a direction lying within a plane parallel to the ground, a limitation that tends to make the use of the cane or crutch awkward while changing direction in mid-stride. Both designs purport to use a non-slip bottom surface.


An oscillator tuning system and an oscillator tuning method are provided. The system includes a determination unit which determines whether a power which is used in an RFID tag having an RFID oscillator is greater than a reference value; and a frequency tuner which tunes a driving frequency of the RFID oscillator according to a result of the determination. The method includes determining whether a power which is used in an RFID tag having an RFID oscillator is greater than a reference value; and tuning a driving frequency of the RFID oscillator according to a result of the determination.

1. Field of the Invention
Apparatuses and methods consistent with the present invention relate to an oscillator tuning system and an oscillator tuning method which enables a backscatter link frequency (BLF) of a radio frequency identification (RFID) oscillator to be within a range of a threshold by distinguishing where there is abundant energy, i.e. a near field region, from where there is less abundant energy, i.e. a far field region and operating the RFID oscillator at a driving frequency according to the distinguished regions.
2. Description of Related Art
An RFID tag is a type of a transponder device that can respond to being read by sending contents of its embedded memory by backscatter communication to an interrogator, i.e. a reader. A passive RFID tag has no bakery; instead, it gets all the needed energy to send a signal to the reader from a carrier signal of the reader.
When an oscillator within the RFID tag is operated using a driving frequency of 1.28 MHz, a backscatter link frequency of the oscillator according to 'PC Gen. 2 Spec.' is out of some part of the range of a threshold. Accordingly, power consumption may be reduced. However, it is not easy to stably process a signal of an oscillator within an RFID tag being operated using a driving frequency of 1.28 MHz.
In order to address the above problem, an oscillator operating method, which is performed by increasing the driving frequency, is provided. However, the method has a problem in that power consumption is so great that good performance may not be guaranteed when processing the signal of the RFID tag in a long distance environment, despite the fact a backscatter link frequency (BLF) of the RFID oscillator may be within a range of the threshold in all regions by operating the oscillator using a higher frequency of 1.92 MHz.
Therefore, a new tuning model for an RFID oscillator is needed which can adjustably process the signal of the RFID tag in all regions, and smoothly process the RFID tag in a long distance environment by adaptively tuning the driving frequency of the RFID oscillator.

Oscillator tuning system and oscillator tuning method