A compression ignition internal combustion engine system combusts DME as fuel using high fuel injection pressures (e.g., above 800 bar) and low oxygen concentrations (e.g., below 17%, through high levels of EGR), thereby resulting in simultaneously very low NOx and PM engine-out emissions.

The use of diesel engines in motor vehicles has greatly added to the atmospheric presence of harmful pollutants such as nitrogen oxides (NOx) and particulate matter (PM). Conventional diesel engines emit NOx and/or PM substantially in excess of desired environmental levels. Nevertheless, because of their fuel efficiency, diesel engines remain preferable to gasoline engines for many applications. Attempts to reduce NOx and PM emissions from diesel engines have therefore continued for many years.
Thus far, the prior art has not provided a robust diesel combustion system (i.e. providing commercially acceptable responsiveness and horsepower across diverse speed and load ranges) that is capable of maintaining engine-out emissions levels of both NOx and PM simultaneously within environmentally desired levels. The challenge of trying to maintain diesel engine emission levels of both NOx and PM simultaneously below environmentally acceptable levels has been a long-standing environmental and industry problem that has never fully been overcome despite extensive, concerted efforts by government and industry worldwide.
With the problem of simultaneously satisfactory engine-out NOx and PM reductions unsolved by the prior art for diesel engines, the diesel industry has instead turned primarily to development of NOx and PM aftertreatments (i.e. post-engine, but before the exhaust gas is released to the atmosphere) to meet current and upcoming international PM and NOx environmental regulations. However, such aftertreatment systems can be expensive, create size ("packaging") concerns, and/or retain issues of effectiveness and durability. As an example, an ongoing challenge for NOx traps is maintaining NOx emissions always below desired levels on vehicles with diverse duty cycles, and the effectiveness and durability of NOx absorbers may also be undermined by factors such as high temperatures or by sulfate adsorption and desulfization due to sulfur content in diesel fuel. As a second example, to effectively maintain PM levels within environmentally acceptable emission levels, PM filter traps depend on engine-out PM emissions not being too high. In addition, PM filter traps must be regenerated (such as by combusting the particulate matter trapped therein), with the frequency of such renewal dependent upon the amount of engine-out PM the trap is forced to catch and retain.
Furthermore, global demand for diesel fuel is placing enormous pressure on world petroleum supplies. Competition for middle distillates has accelerated over the past decade due to rapid growth in demand in both Europe and the developing world, most notably in Asia. Indeed, the world economy and the environment rely heavily on ready supplies of low-sulfur diesel fuel. As conventional supplies of petroleum continue to tighten and eventually decline, finding sustainable alternatives to clean diesel fuel may also become a critical international priority.
Non-petroleum-derived diesel fuel alternatives include biodiesel from plant or vegetable oils, and synthetic diesel or DME from the gasification and reforming of natural gas, coal or biomass. In the United States, vast coal reserves and the availability of arable land make coal and biomass prime candidates as alternative fuel feedstocks.
Among the diesel fuel alternatives in the US, biodiesel and, to a somewhat lesser extent thus far, synthetic diesel have been used somewhat transparently as replacements for low-sulfur diesel fuel. DME has not yet seen commercial success in the US, due mainly to a lack of supporting infrastructure and the technical challenges of onboard fuel storage and fuel system reliability. Compared to biodiesel and synthetic diesel, DME has the advantage that its combustion generally produces very little PM, which potentially allows it to be used to meet US heavy-duty PM emissions requirements without the added cost of a PM filter trap. However, combustion of DME has traditionally produced high NOx emissions, and thus commercialization of a DME engine would be expected to require NOx aftertreatment to meet environmentally desired levels.
Most prototype DME injection systems used in engine demonstrations have been limited to peak injection pressures of between 250 and 300 bar. Little effort has been made to increase fuel injection pressure in DME engines, mainly because of the challenges of pumping a highly compressible, low-viscosity fuel efficiently. In addition, since DME is somewhat like propane in that it flashes to a gas at atmospheric pressure, it may have been felt that there would be little or no advantage to, higher pressure injection of DME.

Low emission dimethyl ether (DME) engine

The United States of America as represented by the Administrator of the U.S. Environmental Protection Agency


A method and apparatus for decoding trellis-coded digital data. Digital data samples that have been convolutionally encoded by trellis-coded modulation (TCM) for purposes of forward error correction require a computationally intensive Viterbi decoder for best performance. A technique known as pragmatic trellis-code modulation (PTCM) reduces decoder complexity but at the expense of degraded performance in some situations. This disclosure provides a solution that retains the reduced complexity of PTCM, but uses a full trellis approach that yields significantly better performance in terms of bit error rate (BER).

The present invention generally relates to techniques for decoding of communication signals that have been encoded using convolutional coding. More particularly, the invention pertains to improvements in decoding signals that have been encoded using a coding technique generally known as trellis coded modulation (TCM). Communication of data through digital communication channels is subject to well known limitations. For communication channels involving orbiting satellites, there is an inherent transponder power limitation, given that a satellite must rely on batteries and solar power. In addition, rapid growth in data rate requirements have resulted in significant bandwidth limitations. Every communication channel is, of course, subject to unwanted noise interference. The effects of noise can be mitigated by employing higher transmitted power and greater bandwidth, but in a practical digital communication channel power and bandwidth are limited, so communication engineers have developed various techniques to correct digital data streams affected by noise. One measure of transmission channel performance is its transmission efficiency, measured in bits per second per Hertz, that is, the data transmission rate in bits per second, for each Hertz of bandwidth.
A widely accepted technique for increasing transmission efficiency is known as forward error correcting (FEC) coding. In essence, the principle of FEC coding is to generate redundancy in the data stream to be transmitted, such that each bit of data has an influence on the state of other adjacent bits of transmitted code. Given this inherent redundancy in the transmitted code stream, a receiver with an appropriate decoder can not only decode the received data but can correct errors in the received data based on reconstruction of the data using redundant information from adjacent bits. A detailed explanation, and even a simple example, of FEC coding is much more complicated than this simple statement of the principle.
FEC coding is often effected by means of a convolutional encoder, which, in its simplest form is a shift register and combinatorial logic that combines the binary states of selected stages of the shift register. The logic typically takes the form of exclusive-OR gates, each of which performs a modulo-two addition of its input bits. Data bits from each shift register stage are clocked not only to the next stage but into selected ones of the logic gates. Output data is clocked from selected logic gates. The rate of a convolutional encoder is defined as the ratio of its number of inputs to its number of outputs. For example, a convolutional encoder with two outputs for every input is referred to as a rate 1/2 encoder. Convolutional encoders may be designed to operate at other rates, such as 3/4, 4/5 and so forth. A convolutional encoder is also defined in part by its "constraint length," which is defined as the number of output bits affected by a given input bit (typically this value is one more than the number of shift register stages in the encoder) and by code generator polynomials. These may be expressed as binary or octal numbers that define the shift register connections to the modulo-two adders of the encoder.
A convolutional encoder may also be conveniently thought of as a state machine that changes state in response to a stream of input data bits. Consideration of the various changes of state that can occur in response to a data stream gives rise to a convenient graphical representation of the state machine, known as a trellis diagram. A trellis diagram is a rectangular matrix of dots. Each dot in the same row represents the same state, for example state 00, state 01, state 10, or state 11. Each column of dots represents possible states at a particular time instant. In a trellis diagram, if all possible state transitions from one time instant to the next are represented by diagonal or horizontal lines from one column of dots to the next, the result is a criss-cross pattern that resembles a trellis.
The trellis diagram format may also be used to depict the actual state transitions of a convolutional encoder when encoding a specific data stream. In this case, the diagram less resembles a trellis but shows in graphical form the succession of states assumed by the encoder during the encoding of a series of input data bits. The trellis diagram is key to understanding the decoding technique originally developed by Andrew J. Viterbi and referred to as Viterbi decoding. Encoding digital data convolutionally in the manner briefly described above and mapping the bit values onto an expanded signal constellation so that the free Euclidean distance is maximized is the essence of trellis coded modulation (TCM).
It will be appreciated that, even in a simple case of an encoder using just a few stages, the trellis diagram is computationally complex. A more useful encoder would have a larger constraint length, such as seven, and the encoder would have 26 or 64 states. The corresponding trellis diagram and the Viterbi decoding algorithm are complex and cumbersome in the extreme. Efforts to simplify TCM decoding led to a method known as pragmatic trellis-coded modulation (PTCM), described in a paper by Andrew J. Viterbi et al., entitled "A Pragmatic Approach to Trellis-Coded Modulation," July 1989 IEEE Comm. Magazine, pp. 11-19.
The design of TCM encoders and decoders for optimum performance depends in part on the type of data modulation employed. The PTCM technique proposed in the Viterbi et al. paper is an approach that permits a single basic encoder and decoder using a rate of 1/2 and a constraint length of seven, to be applied to various modulation schemes. Digital modulation schemes using phase and/or amplitude modulation are generally defined by signal constellations, i.e., graphical representations of the possible signal phases and amplitudes used to represent various data states. The PTCM technique is said to be applicable to a wide variety of signal constellations. Prior to the Viterbi et al. paper, G. Ungerboeck showed in a paper entitled "Channel Coding with Multilevel/Phase Signals," IEEE Trans. on Info. Theory, vol IT-28, pp. 55-67, January 1982, that it was possible to achieve Asymptotic Coding Gains (ACGs) of as much as 6 dB within precisely the same signal spectral bandwidth, by doubling the signal constellation set from M=2k−1 symbols to M=2k symbols, employing a rate (k−1)/k convolutional code, and mapping the bits onto the signal constellation in such a way that the free Euclidean distance of the resulting trellis was maximized. The Viterbi et al. paper pointed out that there were a number of drawbacks to the Ungerboeck approach, as subsequently refined by others. First, each signal configuration seemed to require a different convolution code, and no single encoder-decoder could be used for a wide set of parameters. Second, the Ungerboeck approach suffered from the practical limitation that the number of trellis code states used in most application was very low, typically four or eight, and rarely more than sixteen. Finally, performance in the Ungerboeck model was always to be measured by ACG, which is a function of free Euclidean distance. Viterbi et al. believed that using free Euclidean distance to measure performance would lead to misleading results because not only may there be a large number of paths at the same distance, but also the next higher Euclidean distance between unmerged paths may be only a very small amount larger than the free distance.
In summary, Ungerboeck determined the optimal convolutional code for several TCM modulation types. Then, Viterbi et al. showed that in most cases the de facto industry standard encoder (with constraint length 7, rate 1/2 and generator polynomials 1718 and 1338) could be used along with a technique known as "puncturing" to perform the convolutional coding with little or no loss in bit error rate (BER) performance in many situations. In the puncturing technique, selected bits of the encoder output stream are discarded, resulting in a higher effective code rate. For example, if one output bit in every four from a rate 1/2 encoder is discarded, the encoder effectively outputs three bits for every two input bits and effectively functions as a 2/3 encoder. If two bits of each group of six output bits are discarded, the encoder produces four output bits for each set of three input bits and effectively functions as a 3/4 encoder. When the encoded data stream is punctured in this manner, the data stream received and decoded must be "de-punctured" to supply the discarded data. Typically, de-puncturing is effected by inserting median values that do not affect the decoding process.
Although PTCM decoding techniques represent a practical advantage over earlier convolution encoding and decoding methods, puncturing and de-puncturing the code necessarily results in diminished BER performance. A return to the earlier approach of performing a "full-rate" trellis decoding operation, without puncturing and de-puncturing of the code, presents an extraordinary increase in computational complexity, even at the standard constraint length of seven bits. Accordingly, it would be highly advantageous to provide a convolutional encoding and decoding approach that retains the simplifying advantages of the PTCM approach, but does not result in lowering of the BER performance. The present invention is directed to this end.

Efficient decoding of trellis coded modulation waveforms


A system and method for balancing the capacitive charge on touch sensor electrodes so that every two adjacent routes have the same capacitance as any other adjacent two routes, wherein routing electrodes are spaced further and further apart, or graduated, as they get longer, to thereby balance the capacitance on the touch sensor electrodes without having to add or subtract an offset from each touch sensor electrode.

Field of the Invention
This invention relates generally to touch sensors including touchpads and touchscreens. More specifically, the present invention is a method of balancing capacitance on the electrodes to improve performance.
Description of Related Art
There are several designs for capacitance sensitive touch sensors. It is useful to examine the underlying technology to better understand how any capacitance sensitive touch sensor can be modified to work with the present invention.
The CIRQUE® Corporation touchpad is a mutual capacitance-sensing device and an example is illustrated as a block diagram in FIG. 1. In this touchpad 10, a grid of X (12) and Y (14) electrodes and a sense electrode 16 is used to define the touch-sensitive area 18 of the touchpad. Typically, the touchpad 10 is a rectangular grid of approximately 16 by 12 electrodes, or 8 by 6 electrodes when there are space constraints. Interlaced with these X (12) and Y (14) (or row and column) electrodes is a single sense electrode 16. All position measurements are made through the sense electrode 16.
The CIRQUE® Corporation touchpad 10 measures an imbalance in electrical charge on the sense line 16. When no pointing object is on or in proximity to the touchpad 10, the touchpad circuitry 20 is in a balanced state, and there is no charge imbalance on the sense line 16. When a pointing object creates imbalance because of capacitive coupling when the object approaches or touches a touch surface (the sensing area 18 of the touchpad 10), a change in capacitance occurs on the electrodes 12, 14. What is measured is the change in capacitance, but not the absolute capacitance value on the electrodes 12, 14. The touchpad 10 determines the change in capacitance by measuring the amount of charge that must be injected onto the sense line 16 to reestablish or regain balance of charge on the sense line.
The system above is utilized to determine the position of a finger on or in proximity to a touchpad 10 as follows. This example describes row electrodes 12, and is repeated in the same manner for the column electrodes 14. The values obtained from the row and column electrode measurements determine an intersection which is the centroid of the pointing object on or in proximity to the touchpad 10.
In the first step, a first set of row electrodes 12 are driven with a first signal from P, N generator 22, and a different but adjacent second set of row electrodes are driven with a second signal from the P, N generator. The touchpad circuitry 20 obtains a value from the sense line 16 using a mutual capacitance measuring device 26 that indicates which row electrode is closest to the pointing object. However, the touchpad circuitry 20 under the control of some microcontroller 28 cannot yet determine on which side of the row electrode the pointing object is located, nor can the touchpad circuitry 20 determine just how far the pointing object is located away from the electrode. Thus, the system shifts by one electrode the group of electrodes 12 to be driven. In other words, the electrode on one side of the group is added, while the electrode on the opposite side of the group is no longer driven. The new group is then driven by the P, N generator 22 and a second measurement of the sense line 16 is taken.
From these two measurements, it is possible to determine on which side of the row electrode the pointing object is located, and how far away. Pointing object position determination is then performed by using an equation that compares the magnitude of the two signals measured.
The sensitivity or resolution of the CIRQUE® Corporation touchpad is much higher than the 16 by 12 grid of row and column electrodes implies. The resolution is typically on the order of 960 counts per inch, or greater. The exact resolution is determined by the sensitivity of the components, the spacing between the electrodes 12, 14 on the same rows and columns, and other factors that are not material to the present invention.
The process above is repeated for the Y or column electrodes 14 using a P, N generator 24
Although the CIRQUE® touchpad described above uses a grid of X and Y electrodes 12, 14 and a separate and single sense electrode 16, the sense electrode can actually be the X or Y electrodes 12, 14 by using multiplexing.
With this background in touch sensor technology, it is possible to analyze the improvement provided by the present invention.

Graduated routing for electrodes coupled to touch sensor electrodes


A power supply device includes a DC-DC converter configured to regulate an output voltage based on an externally input instruction, and a discharge circuit connected to an output capacitor of the DC-DC converter, wherein the discharge circuit includes a switch element configured to switch between a conductive state and a nonconductive state according to an instruction, and a capacitor-resistor (C-R) circuit configured to charge electric charge accumulated in the output capacitor of the DC-DC converter when the switch element is conductive and to discharge the electric charge to ground when the switch element is nonconductive.

1. Field of the Invention
The present invention relates to a power supply device with a DC-DC converter and a recording apparatus using the power supply device.
2. Description of the Related Art
A step-down DC-DC converter of pulse width modulation (PWM) switching type is used for power supply to a load of a drive source or an electric circuit in electronic apparatuses. The step-down DC-DC converter of PWM switching type generates a predetermined constant output voltage by performing constant value control while comparing a target voltage value for setting an output voltage with a feedback voltage value from the output voltage.
A DC-DC converter that controls an output voltage to be supplied to a load controls the output voltage by changing a target voltage value for feedback constant voltage control according to a signal from an external control unit or an external electronic apparatus. Alternatively, the DC-DC converter changes a feedback voltage value by adding current to a feedback voltage from the output voltage, thereby controlling the output voltage.
Operation of a conventional step-down DC-DC converter of PWM switching type for controlling an output voltage is described below.
FIG. 9 illustrates a basic configuration of the conventional step-down DC-DC converter 50a. The DC-DC converter 50a includes a digital to analog (D-A) converter 201 and sets a value corresponding to an output voltage value targeted by a control unit 30 to the D-A converter 201. The DC-DC converter 50a operates to maintain the output voltage value at the target voltage value so as to be stable at the target voltage. The DC-DC converter 50a changes a value to be set to the D-A converter 201, thereby changing a voltage value to be maintained.
In the step-down DC-DC converter 50a of PWM switching type illustrated in FIG. 9, an input voltage VHin supplied from a power supply unit (not-shown) is input to a switching element Q101. Then, an alternate current output converted by the switching element Q101 and a diode D101 is output via a reactor L101, so that an output voltage VH is supplied to a load 2-1.
A capacitor C001 is connected to a direct current side of the switching element Q101 and a capacitor C002 is connected to an alternate current side of the switching element Q101 via the reactor L101. The reactor L101 and the capacitor C002 constitute a smoothing circuit.
An output voltage VH detected at an output terminal of the smoothing circuit is divided by a resistor R101 and a resistor R102. The divided voltage (feedback voltage) is input to an error amplifier 202 included in a PWM control circuit (PWM control IC) 200. The PWM control circuit 200 performs feedback control to make the output voltage constant.
The circuit for performing constant voltage feedback control includes the PWM control circuit (PWM control IC) 200, which includes the error amplifier 202, a PWM comparator 203, and a triangular-wave signal generator 205, resistors R103, R104, R105, and R106, and a capacitor C003.
A discharge circuit unit H includes a switch element Q01 and a resistor R01.
One side of the switch element Q01 is connected to ground HGND and the other side of the switch element Q01 is connected to a VH output via the resistor R01. A control terminal of the switch element Q01 is connected to a control unit 30.
The switch element Q01 is turned ON or OFF in response to a DCHRG signal from the control unit 30. For example, the switch element Q01 becomes conductive when the DCHRG signal is at level H (high) and becomes nonconductive when the DCHRG signal is at level L (low).
Now, a control operation for regulating an output voltage is described below. The error amplifier 202 receives a reference voltage Vref supplied from the D-A converter 201 and a feedback voltage of the output voltage VH supplied from the resistors R101 and R102.
An output signal from the error amplifier 202 is input to the PWM comparator 203, which determines a PWM duty ratio. The PWM comparator 203 performs a comparison between an output signal from the error amplifier 202 and a triangular-wave signal output from the triangular-wave signal generator 205. The output from the PWM comparator 203, as an output signal from the PWM control circuit (PWM control IC) 200, is used to control the switching element Q101 via a metal-oxide semiconductor (MOS) drive circuit 204.
The resistors R105 and R106 and the capacitor C003, which are connected between the inverting terminal and output terminal of the error amplifier 202, constitute an exemplary phase compensation circuit.
The control unit 30, which is included in an electronic apparatus, outputs a setting signal DA_S to the D-A converter 201. The resistors R103 and R104 divide the reference voltage Vref output from an output terminal Aout of the D-A converter 201 and input the divided reference voltage Vref′ to the inverting terminal of the error amplifier 202.
The D-A converter 201 regulates a voltage of the Vref terminal based on a digital value of the setting signal DA_S and supplies the voltage to the inverting terminal of the error amplifier 202 as a voltage Vref′ divided by the resistors R103 and R104.
If the D-A converter 201 is an 8-bit D-A converter, the reference voltage Vref can be regulated in 28 stages (two to the eighth power stages), namely, in 256 stages.
A non-inverting terminal of the error amplifier 202 is connected to a voltage dividing point at which the voltage between the output voltage VH of the DC-DC converter 50a and the ground is divided by the resistors R101 and R102. The output voltage VH is expressed by equation (1).VH=Vref·(R101+R102)/R102  (1)The PWM comparator 203 performs feedback control to regulate the value of the output voltage VH to a target voltage value. The output voltage VH can be regulated in 256 stages between a maximum voltage VHmax and a minimum voltage VHmin.
An exemplary DC-DC converter in which an output voltage range of 24 V-19 V can be regulated in two to the eighth power (256) stages by using an 8-bit D-A converter is described below. In this case, a change in voltage corresponding to one bit of control data for the D-A converter is expressed by the following equation:(24 V−19 V)/28≈19.5 mV
The PWM control type DC-DC converter, in which the diode D101 is located at a low side between the switching element Q101 and the ground (HGND) as illustrated in FIG. 9, has a rather low-cost configuration. When the DC-DC converter having the above-described configuration raises an output voltage, the DC-DC converter changes the target voltage value (or changes the feedback voltage value). For example, the DC-DC converter increases an on-duty width of a MOS-FET at a high side. Accordingly, electric power is supplied from an input side to allow the output voltage to rise. A time period required for raising the output voltage to a target voltage value is determined depending on a response time of a feedback loop of the DC-DC converter.
On the other hand, in decreasing the output voltage, the output voltage VH of the DC-DC converter cannot drop immediately in response to a change of the reference voltage Vref of the D-A converter. This is because an output capacitor accumulates electric charge at a voltage generated before the output voltage drops and there is only a voltage-dividing resistor that can discharge electric charge accumulated in the output capacitor.
The voltage-dividing resistor, which determines a feedback voltage, generally employs a constant between several kΩ and several tens of kΩ so as not to degrade power conversion efficiency of the DC-DC converter. Accordingly, current flowing in the voltage-dividing resistor is several mA at the most.
Therefore, upon a light load, for example, in lowering the output voltage while a load current is 0 A, there is no path for discharging excessive electric charge having been accumulated in the output capacitor. Accordingly, it is conventionally necessary to configure a discharge circuit unit H, in which the switch element Q01 and the resistor R01 are connected in series with the output terminal as illustrated in FIG. 9, to enable discharge of the excessive electric charge accumulated in the output capacitor to the ground HGND to lower the output voltage.
Japanese Patent Application Laid-Open No. 2005-168235 discusses a configuration of the discharge circuit unit H, which discharges electric charge of a capacitor at the output terminal when lowering the output voltage.
In Japanese Patent Application Laid-Open No. 2005-168235, a signal for driving the discharge circuit unit H has a constant pulse width independent from a setting value of the output voltage so as to step down the output voltage to a target voltage value within a defined time period in throughout a voltage range required as the output voltage of the DC-DC converter.
In the above-described DC-DC converter, which regulates the output voltage, a control unit preliminarily determines a pre-set voltage. For example, the DC-DC converter has a function of regulating the output voltage to perform energy correction with respect to variation of parts at the load side and environmental variation.
The above-described DC-DC converter, which supplies a power supply voltage to an electronic apparatus, is required to change the output voltage in a short time period in response to a command from the control unit while the DC-DC converter is outputting a voltage of a certain value.
Operation of the DC-DC converter is described with reference to waveforms illustrated in FIG. 10 with regard to a case where the control unit outputs a setting signal to change the output voltage to an output voltage V1 (Vo>V1) when the DC-DC converter is outputting a certain output voltage Vo.
Prior to receiving the setting signal (DA_S) for changing the output voltage, if a load current of the DC-DC converter is 0 A, the DC-DC converter can maintain the output voltage level if the DC-DC converter is supplied with an amount of electric power corresponding to that having been lost in the DC-DC converter. Therefore, the switching element Q101 is in a state of OFF operation, namely, duty 0% operation, almost throughout the switching cycle.
If the DC-DC converter receives, from the control unit 30, the setting signal (DA_S) for changing the output voltage from Vo to V1 (Vo>V1) between time t0 and time t1 illustrated in FIG. 10, an output value from the D-A converter 201 becomes smaller (not shown). Accordingly, the control unit 30 changes the target setting voltage of the DC-DC converter.
After the control unit 30 sets the target setting voltage of the DC-DC converter, the control unit 30 transmits a DCHRG signal (discharge command) of a preliminary set constant pulse width for a time interval between time t2 and time t4.
When the switch element Q01 receives the DCHRG signal at level H from the control unit 30 at time t2, the switch element Q01 becomes conductive. Since the load current is not extracted from the output between time t1 and time t4, the VH voltage remains at Vo between time t1 and time t2 before receiving the DCHRG signal. When the switch element Q01 becomes conductive at time t2, discharge current flows to the ground HGND via the resistor R01 to allow a potential of the capacitor C002 (VH) to drop from Vo to the target voltage V1, so that the output voltage reaches the target voltage V1 at time t3.
The output voltage in a time interval between time t2 and time t3 is expressed by equation (2):V1=Vo·exp(−t/(C002·R01))  (2)The current flowing in the resistor R01 becomes I R01=V1/R01 according to the voltage V1.
Since the control unit 30 outputs the pulse width of the DCHRG signal between time t2 and time t4, the output voltage immediately drops to the target voltage V1 before a time interval between time t3 and time t4. At that time, the DC/DC converter is still performing constant voltage control at the target voltage V1, the output voltage V1 is continuously applied to the resistor R01.
The switch element Q01 is turned OFF when the DCHRG signal reaches level L at time t4, thus resulting in terminating a series of VH modulation control. In the above description, it is assumed that an on-resistance of the switch element Q01 is 0Ω.
The output voltage change range (output voltage regulating range) of the DC-DC converter illustrated in FIG. 9 is between the maximum value VHmax and the minimum value VHmin. In stepping down the output voltage VH from a certain initial voltage Vo to the target voltage V1, the pulse width required to step down the output voltage VH to the target voltage V1 can be expressed by equation (3). Here, a capacitance of the output capacitor C002 is C002 [μF], a discharge resistance value is R01 [Ω], and the on-resistance of the switch element Q01 of the discharge circuit unit H is ignored.t=−C002·R01·LN(V1/Vo)  (3)
For example, 220 μF of the capacitance and 100Ω of the discharge resistance value R01 are applied to the above formula, a pulse width of 0.936 ms is required, for example, when the output voltage is lowered by 1.0 V, namely, from 24 V to 23 V. Further, a pulse width of 1.128 ms is required when the output voltage is lowered by 1.0 V, namely, from 20 V to 19 V.
In view of equation (3), it is understood that, as a potential difference between the initial voltage Vo and the target voltage V1 becomes larger, a time period required for stepping down the output voltage VH becomes longer. It is also understood that, as the capacitance of the output capacitor C002 becomes larger, the pulse width for driving the discharge circuit unit H becomes longer.
As described above, a conduction time of the discharge circuit unit H for stepping down a constant voltage varies with an initial setting and a target voltage if the output capacitor C002 is defined.
For example, a DC-DC converter that is capable of changing an output voltage between the maximum value VHmax of 24 V and the minimum value VHmin of 19 V requires the longest discharge time to lower the VH voltage from 24 V to 19 V.
Here, if this condition is applied to the above circuit constant, the DC-DC converter requires a discharge time of 5.132 ms to lower the voltage by 5 V, namely, from 24 V to 19 V. Thus, the DC-DC converter drives the discharge circuit unit H at a constant pulse width of 5.132 ms to change (regulate) the VH voltage output range to the target voltage.
In other words, the DC-DC converter requires a pulse width of 5.132 ms to lower the voltage to the target voltage value within the time period defined by the constant pulse width in the changeable range of the DC-DC converter.
A DC-DC converter discussed in Japanese Patent Application Laid-Open No. 2005-168235 is described below. The DC-DC converter continuously drives the discharge circuit unit H at a constant pulse width to lower the output voltage to a target voltage value within a predetermined time period in the output voltage range of the DC-DC converter.
If a DC-DC converter in which the output voltage range between 24 V and 19 V can be regulated by an 8-bit D-A converter into 28 (256) stages is exemplified as the DC-DC converter configured to regulate the output voltage, a voltage change for one bit of the D-A converter is expressed by the following formula:(24 V−19 V)/28≈19.5 mV
In other words, the DC-DC converter drives the discharge circuit unit H at the constant pulse width of 5.132 ms to step down the output voltage both in the case of stepping down the output voltage of the DC-DC converter from 24 V to 19 V and in the case of stepping down the output voltage of the DC-DC converter from 24 V by 19.5 mV.
Here, a maximum time period t for stepping down the output voltage by driving the discharge circuit unit H with a constant pulse width throughout the whole output voltage range can be expressed by equation (4).t=−C002·R01·LN(VHmin/VHmax)  (4)
Consequently, the pulse width of the DCHRG signal for stepping down the set voltage value is determined according to a conduction time that satisfies a variation width (ΔVHmax) based on the maximum value (VHmax) and the minimum value (VHmin) of the voltage range. Namely, in this example, it is a conduction time of 5.132 ms that satisfies a change of the voltage from 24 V to 19 V.
Electric power to be applied to the discharge resistor is described below. As illustrated in FIG. 9, the output voltage of the DC-DC converter, configured to regulate the output voltage, has an output voltage width between the maximum value VHmax and the minimum value VHmin. Therefore, the voltage value applied to the discharge resistor upon conduction of the switch element Q01 is also a voltage value between the maximum value VHmax and the minimum value VHmin. However, the applied voltage is not always constant because of the DC-DC converter configured to modulate the output voltage. For example, in the case of a DC-DC converter capable of changing the output voltage between 24 V and 19 V, the voltage between 24 V and 19 V may be applied to the discharge resistor.
Here, a case where the discharge resistance is 100Ω is described. When the output voltage is 24 V, electric power continuously applied to the discharge resistor is 5.76 W. When the output voltage is 19 V, electric power continuously applied to the discharge resistor is 3.61 W. The continuously applied electric power in the case where the output voltage is 24 V becomes about 1.6 times as a case where the output voltage is 19 V.
In the discharge circuit unit H, including the switch element and the resistor connected in series, the output voltage VH is continuously applied to the discharge resistor while the switch element of the discharge circuit unit H is conductive even if the output voltage drops and reaches the target value during a step-down process of the output voltage. Accordingly, a constant discharge current continues to flow in the discharge resistor, namely, a constant electric power is continuously applied to the discharge resistor.
In a typical characteristic of the resistor, a limited electric power at one-pulse is defined by pulse limiting electric power curves as illustrated in FIG. 11. FIG. 11 illustrates a mere example of pulse limiting electric power curves of a lead resistor, namely, pulse limiting electric power curves of five types of rated power between 0.17 W and 2 W and resistor sizes. The characteristic is defined for each respective resistor regardless of types of resistors (for example, metal film, carbon, oxidative metal, and fusing resistor) or manufacturers thereof.
In the typical resistor, as the electric power application time becomes longer, the limiting electric power decreases more. Also, as the rated power becomes higher, the resistor size becomes larger and the limiting value of the pulse limiting electric power curve becomes high. However, if the resistor size becomes larger, there is required a larger installation space, thus resulting in higher cost.
If the output capacitance C002 and the output voltage range (VHmax, VHmin) are determined according to formula (4), the time period t required for stepping down the output voltage is determined according to the resistance value of the discharge resistor.
The maximum electric power is applied to the resistor at a time when the VH voltage for one bit of the D-A converter (19.5 mV) is stepped down from the maximum voltage VHmax.
FIG. 12 illustrates a calculating result of a discharge time required for stepping down the voltage in the whole voltage range and electric power applied to the discharge circuit unit H for respective different constants of the discharge resistor R01.
A required discharge time is represented by a pulse width value in FIG. 12 calculated with the proviso that the maximum value VHmax is 24 V, the minimum value VHmin is 19 V, and the capacitor C002 is 220 μF in formula (1), for respective resistance values of the resistor R01 between 10Ω and 220Ω.
For the ease of calculation, the resistor applied electric power is a value of electric power applied to the resistor when the discharge circuit unit H is made conductive while the output voltage remains at 24 V. That is, the applied electric power is calculated by 24 V×24 V/R01 to obtain the maximum electric power to be applied to the resistor.
FIG. 13 is a graph obtained by plotting the result illustrated in FIG. 12 into pulse limiting electric power curves. As apparent from FIGS. 12 and 13, as the resistance value becomes larger, the applied electric power becomes smaller, thus resulting in enabling the use of a resistor having a smaller rated power. However, it is seen that the conduction time of the discharge circuit unit H, namely, a pulse width required for driving the discharge circuit unit H for stepping down the output voltage, is elongated.
Further, as the resistance value becomes smaller, the output voltage drops in a short time period. However, the electric power to be applied to the resistor becomes larger. Therefore, the use of such a resistor having both a large rated power and a large resistor size is required.
As understood from the above description, it is required to make the resistance value of the electric discharge resistor smaller to drive the discharge circuit always at a constant pulse to step down the output voltage to the target output voltage value within a predetermined time period in the output voltage range of the DC-DC converter.
However, in using the resistor within the pulse limiting electric power curve, a resistor having a large rated power and a large resistor size is required. Also, in using a resistor requiring a small installation space, the electric power to be applied to the resistor is required to be limited within the pulse limiting electric power curve, such that the time period for stepping down the output voltage to the target value tends to take time in a conventional configuration.
Considering the above-described point, in order to avoid increase of the sizes of the rated power and, thus, a power source unit, it is required to shorten the discharge time by connecting small-sized rated power resistors in series to reduce the electric power to be applied to the resistors. Thus, a problem of increased cost arises.

Power supply device and recording apparatus


A clip includes: a pin having a head and a shank; and a grommet having a collar and a leg. The leg of the grommet is divided into a plurality of expansion leg pieces by means of slits extending along an axial direction. The expansion leg pieces of the leg are expanded outwardly by the insertion of the pin into an inner side of the grommet, so as to clamp plate-like members by the collar and the expansion leg pieces. The leg of the grommet has resilient beams which project outwardly of the expansion leg pieces on a collar side when the pin is inserted into the inner side of the grommet.

1. Field of the Invention
The present invention relates to a two-piece type clip for fixing a plate-like member, such as an interior or exterior part of an automobile, to a vehicle body panel, for example.
2. Description of the Related Art
Each conventional clip of this type, although not specifically shown, is comprised of two parts including a pin and a grommet, which are integrally molded from a synthetic resin. The former pin has a head and a shank, while the latter grommet has a collar and a leg, the leg being divided into a plurality of expansion leg pieces by means of slits extending along an axial direction (refer to, for example, JP-A-11-270525).
When a plate-like member of an automobile is actually fixed to a body panel, the shank of the pin is first inserted into inner sides of the expansion leg pieces through an opening formed in the collar of the grommet, thereby obtaining a state in which the two members are temporarily retained. In this state, the leg of the grommet is pushed into attaching holes bored in advance in the plate-like member and the body panel, so as to completely push the pin shaft into the inner sides of the expansion leg pieces. Then, the expansion leg pieces of the grommet expand outwardly. Consequently, the plate-like member and the body panel are clamped from inner and outer sides by the collar of the grommet and the expansion leg pieces, thereby fixing the plate-like member to the body panel side in a superposed state.
Accordingly, with the conventional clip, the divided expansion leg pieces of the leg are forcibly expanded outwardly by the insertion of the pin into the inner side of the grommet, so as to fix the plate-like member to the body panel in a superposed state. At this time, in a case where the total plate thickness of the plate-like member and the body panel is large, not much problem is encountered in the fixation since the dip portion side of each expansion leg piece is capable of expanding sufficiently in terms of its structure. However, in a case where the total plate thickness of the plate-like member and the body panel is conversely small, the expanding force naturally becomes insufficient in the light of the fact that the expansion leg pieces must expand on the root side of the collar. Therefore, in the event that a large external force is applied or the effect of vibrations persists for a long time after the fixation of the plate-like member, there has been a possibility that the plate-like member rattles unexpectedly, or unpleasant noise occurs due to the rattling. It should be noted that, in the conventional clip, a construction is adopted in which expanding points of the expansion leg pieces are made different in the axial direction, with this measure alone it has not been impossible to expect a sufficient effect.


A computer input device uses an ordinary sheet material such as a card, letter or envelope to select and point to a symbol. A remote portion of the sheet material covers sensors which sense available ambient light. Uncovered sensors provide background light information. A single sensor is associated with multiple symbols and is associated with symbols in at least two specific groups of symbols. A processor is connected to a demultiplexer and a multiplexer, and the demultiplexer and multiplexer are connected to individual sensors. Biasing resistors are connected to individual sensors. The processor selects a pair of connectors from the demultiplexer and multiplexer, thus sampling a particular sensor. The output of that sensor is converted a digital signal, which is sent to the processor and stored in a random access memory. After sufficient sensors have been interrogated, the processor provides an output which is particularly related to the sensor or group of sensors which are covered by the pointer sheet, and which is specifically related to a symbol pointed to by the pointer sheet. In one embodiment, the processor output is displayed as a particular information. In one embodiment, the processor output illuminates lamps over individual receiver bins to indicate the receiver bin in which the pointer piece should be deposited.

This inventions concerns computer inputs, and particularly object pointer input devices.
Most prior art keyboard devices use the concept of discrete keys. The keys require interconnect wires implemented in a keyboard composed of multiple keys. One prior art device codes each key with multiple switches. The switches may be optical sensors. A group of one or more sensors is dedicated to a particular key. When a key closure occurs, the code of that key can be detected on the interconnect wires. The appearance and use of that prior art keyboard would be like any standard keyboard in use today. The implementation of the key electronics is the only difference. Mechanical keys are replaced with coded optical keys. Multiple sensors are involved in detecting a key. The use of multiple sensors for one key is inefficient. As an example, one device requires 36 sensors for 14 keys.
One prior art device requires a special light source in an enclosure. Multiple sensors make up a key, and each individual key must be uniquely fabricated and interconnected. Keys are not interchangeable.
In another keyboard which is similar because it uses the same discrete key concept, a switch is a single emitter-receiver sensor pair. The light path between the two is broken to produce a key closure.
One touch screen uses a finger or similar object to point to a location in a rectangular area. The rectangular area has optical transmitters and receivers on its boarders. Resolution is determined by the size of the object used for pointing.
Another prior art device uses optical sensors and detects coordinates. There are major differences in the technology and method used for determining coordinates in that device. A major portion is mechanical and uses motors, slotted belts and rotating drums to scan a defined area for an object. Parabolic mirrors concentrate lights that must span the x and y coordinates. Synchronization of the moving parts to electronic counters is required to determine coordinates. The rate of change if determined by the speed of the moving parts. Resolution is determined by the object used for pointing, the quality of the optics, and the mechanical-to-electronic interface tolerances.
One prior art device requires a pointer that breaks a plane of light to the sensors. The pointer is positioned perpendicular to the plane of light. Pointers would typically be thin cylindrical objects, like pencils, fingers, etc. The ability to determine a coordinate is determined by the diameter of the pointer and the quality of the optics.
A need exists to provide simplified electronic inputs which provide versatility and accuracy using common objects as pointers.


A method apparatus for retrieval of data from an index database is disclosed. Each address in an index memory corresponds to an attribute that may be possessed by one or more records in the index database. Index memory datawords located at each index memory address include a number of binary bits equal to the number of records in the index database. If a record possesses an attribute, the value of the bit at a position corresponding to the record's address in that attribute's index memory dataword will be a binary "1". Priority encoder circuitry is provided to locate the positions of each of the 1's in the index memory datawords so that all of the records in the index database that possess an attribute can be determined. Logic circuitry is provided to combine index memory datawords logically to form new datawords that can be used to identify records that possess either all of a plurality of selected attributes, or one or more of a plurality of selected attributes. This retrieval system employs parallel hardware circuitry to increase data retrieval speed.

The present invention relates, in general, to a method and apparatus for locating and retrieving data from an index in a binary digital memory.
Data indexes are frequently employed in applications where specific data is to be located in, and retrieved from, a memory containing a large quantity of data, such as a database. Searching for specific data in a large database is usually a time consuming procedure, even though it is implemented on a computer.
A common problem in indexing systems is that the data is not identified in an efficient manner. For example, consider automobile part numbers. These numbers are often made up of perhaps a dozen or more digits and letters, and there are therefore trillions of possible part numbers. However, perhaps only fifty thousand of those numbers will be assigned to actual parts. To locate the parts in a warehouse or to locate the part inventory records stored in a file, an index is required to map the small subset of assigned part numbers to a set of location numbers. Clearly, it would be very inefficient and time consuming if the part number searching procedure implemented by the index involved searching all possible part numbers, whether the numbers have actually been assigned to parts or not.
Another example of this problem is a book index which maps words to page numbers. Since there are more than 200 billion (26.sup.8) possible combinations of eight letters, but a considerably smaller number of eight letter words, the same situation exists here as with the automobile parts, where a few elements from a large set are mapped onto a second smaller set.
This indexing problem can also be illustrated by the pattern recognition process employed by the human brain. There is an extremely large number of possible patterns of sensory inputs that can be presented to the millions of sensory neurons, but only a relatively small number of those patterns have "meaning" and will be recognized by the brain.
Computers have been utilized to implement data indexes because of their inherent capability of being able to search through a large quantity of data in a relatively short period of time. Sequential searching is the simplest approach, but is usually too slow. Various software techniques, such as hash codes, table look up algorithms, B-trees and other approaches have been developed in an effort to speed up the process.
Hash codes are often utilized to solve the problem discussed above, and are algorithms which transform a large number into a smaller one, usually through use of some type of division algorithm. All of these software techniques still require substantial time for the execution of the algorithms. The present invention seeks to overcome this drawback.

Method and apparatus for representing and interrogating an index in a digital memory


Peripheral surfaces of end portions of individual optical fibers constituting a first optical fiber array are in close contact with an optical fiber fitting surface of a base, while the peripheral surfaces of the end portions of the individual optical fibers are in close contact with one another. Peripheral surfaces of end portions of optical fibers on both sides out of the individual optical fibers constituting the first optical fiber array are in close contact with a pair of stopper members, respectively, disposed on both sides of an end portion of the first optical fiber array. A second optical fiber or optical fiber array is opposed to an end face of the end portion of the first optical fiber array. Between projecting portions of the pair of stopper members projected from the end face of the first optical fiber array, peripheral surfaces of end portions of the individual optical fibers constituting the second optical fiber or optical fiber array are in close contact with the optical fiber fitting surface of the base.

The optical fiber is composed of a circular-in-section core serving as an optical path, and a tubular clad surrounding the core, and its outer periphery is sheathed as required. In an optical transmission system using such optical fibers as a means for information and/or energy transmission, there are provided optical switches for switching the optical path, and optical connectors for connecting or extending the optical path. For the interconnection of two optical fibers at the optical switches or optical connectors, it is important to make the optical axis of one optical fiber coincident with the optical axis of the other optical fiber in order to maintain the optical path at good transmission characteristics, because the core that forms the optical path of the optical fiber has an extremely thin diameter.
Optical switches (switches for switching the transmission path) are commonly considered indispensable for transmission systems using optical fibers, and there have been proposed various types of optical switches. Among others, movable-fiber type optical switches that allow the optical fibers to be directly moved and switched are regarded promising by virtue of their high switching speed, low control voltage, frequency independence of characteristics, as well as low price and good miniaturizability. Further, for such movable-fiber type optical switches, there has been a demand for the realization of those capable of functioning as a two-input, two-output optical switch.
A conventionally known movable-fiber type optical switch that has realized such two-inputs and two-outputs is disclosed in Japanese Patent Laid-Open Publication No. SHO 61-272713. The construction of this prior-art optical switch is explained with reference to FIG. 73. In a casing 302 of an optical switch 301, a pair of blocks 303, 304 are disposed. The first block 303 is fixed to the casing 302 by a leg portion 305. The first and second blocks 303, 304 have through holes 306, 307 (which otherwise may be recesses), respectively, as well as side faces 308, 309 slanted at a specified angle with respect to the center axes of the through holes 306, 307. Further, a pair of plate springs 310 are fixed by screws 311 to the side faces 308, 309 of the first and second blocks 303, 304. Thus, the second block 304 is supported on the first block 303 with the plate springs 310. On both sides of the second block 304 are disposed stoppers 312, respectively. Further, on one side of the second block 304, there is disposed, for example, an electromagnetic actuator 313, which is capable of driving the second block 304 in a direction of arrow 314 against the elasticity of the plate spring 310.
Into the through holes 306, 307 of the first and second blocks 303, 304 of the optical switch 301 as described above, are inserted pin holders 317 and 318 holding three optical fibers 315a, 315b, 315c and 316a, 316b, 316c, respectively, where the pin holders 317, 318 are held so that the end faces of the pin holders 317, 318 come into in contact with each other. In these pin holders 317, 318, each three optical fibers 315a, 315b, 315c and 316a, 316b, 316c extend up to the end faces of the pin holders 317, 318, and are arranged precisely at a specified pitch at the end faces, with the end faces of the optical fibers 315a, 315b, 315c and 316a, 316b, 316c being exposed to the end faces of the pin holders 317, 318. Further, although not shown, the other end of the optical fiber 315c and the other end of the optical fiber 316a are connected to each other via an optical attenuator (not shown).
Thus, when the plate springs 310 are in a generally straight, normal state without any external force acting on the first and second blocks 303, 304, the optical fibers 315a and 316a are in an optical transmission relation, the optical fibers 315b and 316b are in an optical transmission relation, and the optical fibers 315c and 316c are in an optical transmission relation, as schematically shown in FIG. 74A. Accordingly, in this state, the optical fibers 315a and 316c have an optical transmission relation with each other via the optical fibers 316a and 315c.
On the other hand, when the actuator 313 is driven so that an external force along the arrow 314 acts on the second block 304, the plate spring 310 is elastically deformed, causing the second block 304 to be laterally displaced along the arrow 314. As a result, as shown in FIG. 74B, the optical fibers 315a and 316b are positioned in a line so as to come into an optical transmission relation, while the optical fibers 315b and 316c are also positioned in a line. Thus, an inversion switching of the optical switch 301 is effected.
However, such conventional optical switches would involve complex construction of the optical fiber array because each optical fiber array is made up by inserting three optical fibers into a pin holder, and inserting this pin holder into the through hole or recess of the first or second block. This would result in large size of the optical fiber array as well as heavy weight. Further, because of also increased size of optical switches as well as heavy weight of optical fiber arrays, it would be difficult to attain faster switching speeds.
Also, the optical fiber array would be required to have an extremely high precision for the array pitch of the optical fibers. For example, connecting two optical fibers of the graded index type with core diameter 50 .mu.m and clad diameter 125 .mu.m oppositely to each other would involve a tolerance of 3 .mu.m or less, an extremely high precision. As a result, extremely high level of machining would be required for the first or second block with a through hole or recess provided, which in turn would need high molding precision also for the pin holders. This would make a factor of increase in the cost of the optical fiber array, as a disadvantage.
Even if the dimensional precision or the like for the first or second block can be obtained, it would be difficult to attain as high precision of adjustment as, for example, a tolerance of 3 .mu.m because the standstill position of the movable-side block is restricted by adjusting the amount of projection of the stopper. As a result, it has been difficult to attain the positioning of the optical fiber array, and in turn to obtain the precision of optical axis alignment of the optical fibers.
Also, since it needs a large force to displace the second block by overcoming the inertia due to the mass of the block or the reaction force of the plate spring, the actuator for driving the second block would be large sized. As a result, the optical switch itself would be large sized, the power consumption would also be increased, and the manufacturing cost of the actuator would be increased, as further problems. Further, due to the large inertia of the second block and the reaction force of the plate spring, it would be difficult to attain faster switching speed of the switching operation.
Furthermore, since the optical fibers would make contact with jigs or the like in the assembling process of the optical fibers, foreign matters would stick to the end faces of the end portions of the optical fibers. This would cause the optical transmission characteristics to be deteriorated.
Generally, the optical fiber is formed from glass or plastics, and extremely thin diameters are employed to obtain good optical transmission characteristics. Therefore, the optical fiber would be so weak in strength and inadequate to forcedly bend. As a result, there has been a need of taking a large spacing of optical fiber array so that bends will least occur at the connecting portions of optical fibers. This would cause the whole unit to be large sized.
As described above, the disadvantages of the prior art to be solved by the present invention can be summarized as follows:
(1) Because of the difficulty in precisely positioning the optical fiber array on the movable side, it is difficult to attain a precision optical axis alignment between optical fibers; PA1 (2) The optical fiber array is complex in construction and large in weight. Since such a heavy optical fiber array is driven, the actuator is necessarily increased in size, causing the weight to increase, so that the optical switch becomes heavy weight; PA1 (3) Since the optical fiber array is complex in construction and large in size so that the actuator is also large sized, it is difficult to miniature the optical switch; PA1 (4) Because of the large-sized actuator, the power consumption is increased; PA1 (5) Since the optical fiber array of heavy weight is driven against its inertia and the reaction force of the plate spring, the switching speed of switching operation is slow; PA1 (6) Because of the need of high-precision machined parts such as blocks and pin holders as well as the need of a large-sized actuator, the optical switch takes high manufacturing cost; and PA1 (7) Foreign matters stick to the end faces of the optical fibers such that the optical transmission characteristics are deteriorated. PA1 peripheral surfaces of end portions of individual optical fibers constituting a first optical fiber array are in close contact with an optical fiber fitting surface of a base, while the peripheral surfaces of the end portions of the individual optical fibers are in close contact with one another; PA1 peripheral surfaces of end portions of optical fibers on both sides out of the individual optical fibers constituting the first optical fiber array are in close contact with a pair of stopper members, respectively, disposed on both sides of an end portion of the first optical fiber array; PA1 a second optical fiber or optical fiber array is opposed to an end face of the end portion of the first optical fiber array; and that PA1 between projecting portions of the pair of stopper members projected from the end face of the first optical fiber array, peripheral surfaces of end portions of the individual optical fibers constituting the second optical fiber or optical fiber array are in close contact with the optical fiber fitting surface of the base. PA1 holding the optical fiber or optical fiber array so that the end portion thereof is positioned away from the fitting surface of the base; PA1 depressing the optical fiber or optical fiber array toward the fitting surface of the base by using a depressing member, between holding portion and end portion of the optical fiber or optical fiber array, so that the optical fiber or optical fiber array is flexed; PA1 stopping the depressing member in a state that the end portion of the optical fiber or optical fiber array is in close contact with the fitting surface of the base; and PA1 fixing the optical fiber or optical fiber array to the base at at least one place between the holding portion and the end portion of the optical fiber or optical fiber array. PA1 holding the optical fiber array so that the end portions of its individual optical fibers are arrayed in generally parallel with one another and in close contact with the fitting surface of the base; PA1 sandwiching the optical fiber array from both sides between holding portion and end portion of the optical fiber array so that spacings between the optical fibers constituting the optical fiber array are narrowed stepwise, and maintaining the spacings with the end portions of the individual optical fibers kept in close contact with one another; and PA1 bonding the individual optical fibers constituting the optical fiber array, with one another, at at least one place between the holding portion and the end portion of the optical fiber array. PA1 a base having a planar-shaped optical fiber fitting surface; PA1 a holding portion for holding an optical fiber array by sandwiching it against the base; PA1 a first pressing portion which extends from the holding portion and which presses the optical fiber array between the holding portion and an end portion of the optical fiber array so that a peripheral surface of the end portion of the optical fiber array is brought into close contact with the fitting surface of the base; and PA1 a second pressing portion which extends from the holding portion and which presses optical fibers located on both sides in the optical fiber array between the holding portion and the end portion of the optical fiber array so that the peripheral surfaces of the end portion of the optical fiber array are brought into close contact with one another. PA1 a base having a planar-shaped optical fiber fitting surface; PA1 a fixed optical fiber composed of a plurality of optical fibers, peripheral surfaces of end portions of the optical fibers being arrayed in close contact with the fitting surface of the base and in close contact with one another; PA1 a pair of stopper members which are in close contact with peripheral surfaces of end portions of optical fibers located on both sides, respectively, out of the individual optical fibers constituting the fixed optical fiber; PA1 a movable optical fiber composed of optical fiber with the number smaller than that of the fixed optical fiber, peripheral surface of the end portion of the movable optical fiber being in close contact with the fitting surface of the base, end face of end portion of the movable optical fiber being opposed to an end face of an end portion of the fixed optical fiber with a minute spacing; and PA1 drive means for reciprocatingly moving the movable optical fiber along a direction perpendicular to its optical axis within a range restricted by projecting portions of the pair of stopper members that are projected from the end face of the fixed optical fiber. PA1 a holding portion for sandwiching and holding an optical fiber by sandwiching it against the base; PA1 a first pressing portion which extends from the holding portion and which presses the optical fiber between the holding portion and an end portion of the optical fibers so that a peripheral surface of the end portion of the optical fiber is brought into close contact with the fitting surface of the base; and PA1 a second pressing portion which extends from the holding portion and which presses optical fibers located on both sides in the optical fiber, between the holding portion and the end portion of the optical fiber so that peripheral surfaces of the end portion of the optical fiber are brought into close contact with one another. According to this embodiment, when the pressing portion are fitted to the base, the end portion of the optical fiber is put into close contact with the base by the first pressing portion, while their end portions are put into close contact with one another by the second pressing portion. Therefore, the close contact of the optical fibers with the base and the close contact of one optical fiber with another can be easily accomplished, allowing an easy assembly. PA1 a base having a planar-shaped optical fiber fitting surface; PA1 a first optical fiber array composed of a plurality of optical fibers, peripheral surfaces of end portions of the optical fibers being arrayed in close contact with the fitting surface of the base and in close contact with one another; PA1 a second optical fiber array composed of a plurality of optical fibers, end face of end portion of the second optical fiber array being opposed to an end face of an end portion of the first optical fiber array on the fitting surface of the base, peripheral surfaces of end portions of the optical fibers being arrayed in close contact with the fitting surface of the base in close contact with one another; and PA1 a pair of stopper members which are in close contact with peripheral surfaces of end portions of optical fibers located on both sides, out of the individual optical fibers constituting the first optical fiber array and the second optical fiber array.

Optical fiber connecting structure, optical switch and optical connector


There are provided a learning apparatus, a learning method, and a program that enable, by using one type of device data, learning of a plurality of models using different data formats. A learning data acquiring section (36) acquires first data that is first-type device data. A first learning section (42) performs learning of a first model (34(1)) in which an estimation using the first-type device data is executed by using the first data. A learning data generating section (40) generates second data that is second-type device data the format of which differs from the format of the first-type device data on the basis of the first data. A second learning section (44) performs learning of a second model (34(2)) in which an estimation using the second-type device data is executed by using the second data.

In recent years, there has been noticed a technique of an artificial intelligence for performing an estimation of an estimation target by using a machine learning model such as a neural network or a support vector machine (SVM). In the foregoing technique, for example, an estimation of an estimation target is performed on the basis of an output when data of the estimation target is input to a learnt model.

Learning apparatus, estimation apparatus, learning method, and program


A lens-fitted photographic film unit comprises a main body having a photograph-taking function and a rear cover for shielding the rear of the main body in a light-tight manner. The main body has a film supply chamber and a film take-up chamber both of which are provided with respective bottom openings. The film supply chamber is provided with a first side edge limiting plate so as partially to shield the bottom opening. The rear cover is provided with a cap for closing the bottom opening of the film supply chamber and a second side edge limiting plate. A winding shaft is engaged with a film leader of a photographic film, drawn from a photographic film cassette and positioned in a film supply chamber. Similarly, the empty photographic film cassette is positioned in a film take-up chamber. When the rear cover is attached to the main body, an arc-shaped cutout of the first side edge limiting plate faces an arc-shaped cutout of the second side edge limiting plate to form a circular opening through which the winding shaft is inserted and withdrawn. The winding shaft is rotated in the opening to wind up the film thereon. After forming a film roll, the winding shaft is pulled out of the film roll through the opening and the cap is closed to seal the unit light-tightly.

1. Field of the Invention
The present invention relates to a lens-fitted photographic film unit and an assembling method therefor, and more particularly to a lens-fitted photographic film unit whose unexposed film roll contained in a film supply chamber can be formed properly in a factory.
2. Prior Art
There are widely prevalent lens-fitted photographic film units (hereinafter referred to simply as film units) which have a photographic film and a simple photograph-taking function incorporated therein. Such film units are available at shops at low prices so as to take photographs without the need for carrying about heavy and expensive cameras. These film units are designed to attract customers at low prices not much higher than the price of photographic film cassettes, by providing them with simple structures.
The film units comprise a film housing and an outer casing for imparting a decorative external appearance to the film units. The film housing has a main body provided with a shutter mechanism, a taking lens, a film advancing mechanism and so forth, a front cover attached to the front of the main body and a rear cover attached to the rear of the main body. The main body includes a film take-up chamber for loading a photographic film cassette and a film supply chamber for containing a film roll which is formed by coiling an unexposed photographic film strip which has been drawn out of the photographic film cassette.
FIG. 5 illustrates a conventional way of inserting a film roll 51a in a film supply chamber 50. An unexposed photographic film 51 is wound in a roll about a winding shaft 52 so as to form the film roll 51a in a darkroom of a factory, which roll is then inserted in the film supply chamber 50 by the winding shaft 52. When the winding shaft 52 is withdrawn from the film supply chamber 50, loosening of the film roll 52a occurs so that the upper portion of the film roll 51a rises to impart to the film roll 51a an oval shape. When a rear cover (not shown) is attached to a main body in this condition, the film roll 51a is deformed, and this deformation induces a larger film advancing torque or resistance to advance. In order to eliminate this problem, a pressing jig 53 is used to press the upper portion of the film roll 51a so as to prevent the film roll 51a from ballooning to an oval shape.
Furthermore, when the winding shaft 52 is pulled from the film roll 51a, a side edge of the film roll 51a projects outside the film supply chamber 50 in a spiral, because of the movement of the winding shaft 52. In this condition, the rear cover crushes the projecting side portion, which can result in failure of film advance. For preventing this, upon winding a photographic film 51, the side edge of the film roll 51a is brought into contact with a side edge straightening jig 54 which is initially fitted on the winding shaft 52. The film roll 51a is inserted in the film supply chamber 50 in such a manner that the side edge straightening jig 54 fits into a bottom opening of the film supply chamber 50. The side edge straightening jig 54 is removed after the winding shaft 52 is pulled completely from the film roll 51a.
However, the pressing jig 53 must be removed prior to attaching the rear cover, which causes loosening of the film roll 51a as well. Besides, the diameter of the film roll of a photographic film for 24 frames is different from that of a photographic film for 36 frames, so that it is required to use a different size of pressing jig in each case, which is troublesome and time consuming.
Also, it is necessary to remove the side edge straightening jig 54 after the tendency of the film roll 51a to protrude has been corrected. This is another factor that reduces the speed of assembly. Moreover, when the side edge straightening jig 54 as well as the film roll 51a is inserted in the film supply chamber 50, the side edge straightening jig 54 is brought into contact with the edge of the bottom opening of the film supply chamber 50, which may produce foreign particles since the edge of chamber 50 is rubbed by the side edge straightening jig 54. Especially in the case of photographic film of 24 frames, since the diameter of the film roll is smaller than that of a photographic film of 36 frames, contact pressure between the edge of the bottom opening of the film supply chamber 50 and the side edge straightening jig 54 is high, which increases the chance of foreign particles.

Lens-fitted photographic film unit and method for assembling the same


Methods and devices are disclosed for providing additional functionality using multi-function keys. The device includes key assemblies for receiving a contact input or a press input. A processing system is coupled to the key assemblies and configured to determine a number of objects contacting a key assembly. The processing system provides a first response when the key assembly has received the press input from a first number of objects and provides a second response when key assembly has received the press input from a second number of objects. The method includes determining a number of objects contacting a key assembly and whether the key assembly has received a contact input or a press input. A first response is provided when the key assembly has received the press input from a first number of objects and a second function is provided when key assembly has received the press input from a second number of objects.

Pressable touchsurfaces (touch surfaces which can be pressed) are widely used in a variety of input devices, including as the surfaces of keys or buttons for keypads or keyboards, and as the surfaces of touch pads or touch screens. It is desirable to improve the usability of these input systems.

Multi-function keys providing additional functions and previews of functions


An aluminum phosphate composition comprising aluminum phosphate, aluminum polyphosphate, aluminum metaphosphate, or a mixture thereof. The composition may be characterized by, when in powder form, having particles wherein some of the particles have at least one or more voids per particle. In addition, the composition is characterized by exhibiting two endothermic peaks in Differential Scanning Calorimetry between about 90 degrees to about 250 degrees Celsius. The composition is also characterized by, when in powder form, having a dispersibility of at least 0.025 grams per 1.0 gram of water. The composition is made by a process comprising contacting phosphoric acid with aluminum sulfate and an alkaline solution to produce an aluminum phosphate based product; and optionally calcining the aluminum phosphate, polyphosphate or metaphosphate based product at an elevated temperature. The composition is useful in paints and as a substitute for titanium dioxide.

1. Field of the Invention
The invention relates to methods of making particles of aluminum phosphate, aluminum metaphosphate, aluminum orthophosphate and aluminum polyphosphate. This invention further relates to use of such particles as pigments in paints and other applications.
2. Background of the Invention
Titanium dioxide is the most common white pigment due to its strong ability to backscatter visible light, which is in turn dependent on its refractive index. Substitutes for titanium dioxide have been sought, but the refractive indexes of both the anatase and rutile forms of this oxide are much higher than those of any other white powder, due to structural reasons.
Titanium dioxide pigments are insoluble in coating vehicles in which they are dispersed. The performance properties of such titanium dioxide pigments, including its physical and chemical characteristics, are determined by the particle size of the pigment and the chemical composition of its surface. Titanium dioxide is commercially available in two crystal structures: anatase and rutile. Rutile titanium dioxide pigments are preferred as they scatter light more efficiently and are more stable and durable than anatase pigments. Titanium dioxide scatters light very efficiently due to its large refractive index. The decorative and functional attributes of titanium dioxide, due to its scattering capabilities, make it a highly desirable pigment. However, titanium dioxide is known to be an expensive pigment to manufacture. Accordingly, there is a need for a more affordable substitute for titanium dioxide as a pigment.
As mentioned, a desired feature of titanium dioxide is its large capacity of spreading (or scattering) the visible light. This property is the result of its high refraction index, together with the absence of electronic transitions in the visible part of the spectrum. Many attempts have been carried out to replace the titanium dioxide, partially or totally in its applications as pigment. However, the refraction indices of its two forms, anatase and rutile, are difficult to obtain by other white solid substances (Handbook of Chemistry and Physics, CRC Press, 57th ed., 1983). Thus, the search for new white pigments led to the search of systems with other light spreading mechanism. Multiphase media, which present a large variation of the refraction index, may operate as light spreaders.
The current options for manufacturing processes of pigments or paints that result in a film containing "pores" in the internal part of the particles or between the particles and the resin is also quite limited. Some techniques for hollow particle preparation have been described in the literature, however, most techniques involve the manufacturing of spheroidal hollow and polymeric particles by polymerization in emulsion. An example is the study of N. Kawahashi and E. Matijevic (Preparation of Hollow Spherical Particle of Itrium Compounds, J Colloid and Interface Science 143(1), 103, 1991) on the coating of the polystyrene latex with basic itrium carbonate and subsequent calcination in high air temperatures, producing hollow particles of itrium compounds.
The preparation of hollow particles of aluminum metaphosphates by chemical reaction between the sodium metaphosphate and aluminum sulfate, followed by thermal treatment, was described by Galembeck et al. in Brazilian Patent BR 9104581. This study referred to the formation of hollow particles of aluminum phosphate synthesized from sodium phosphate and aluminum nitrate. As mentioned, the two pigments, aluminum phosphate and metaphosphate, can be used to replace a large part of TiO2 in paints based on PVA latex or acrylic emulsions.
Brazilian Patent BR 9500522-6 of Galembeck et al. describes a way of making a white pigment from a double aluminum and calcium metaphosphate, obtained directly by a chemical reaction between the aluminum metaphosphate and calcium carbonate particles in a polymeric latex emulsion type aqueous medium. This patent extended the previous results to calcium salts that, from the environmental point of view, are advantageous due to their full atoxicity.
Several publications discuss the synthesis of aluminum phosphate materials primarily for use as a catalyst support including crystalline and amorphous forms. Many of these methods yield highly porous and crystalline forms and few thermally stable amorphous compositions. Examples of such materials are described in U.S. Pat. Nos. 3,943,231; 4,289,863; 5,030,431; 5,292,701; 5,496,529; 5,552,361; 5,698,758; 5,707,442; 6,022,513; and 6,461,415. There exists a need, however, for aluminum phosphate with hollow particles, particularly for a powder that could be manufactured with relative ease.

This patent is subject to a terminal disclaimer.

Aluminum phosphate, polyphosphate, and metaphosphate particles and their use as pigments in paints and method of making same


A spiral die assembly can obtain a homogeneous pipe body, an inflation film with/without drawing and a functional film by dispersing molten resin homogeneously, and the spiral die assembly is capable of being applied to various molten resins and extruding conditions. The spiral die assembly comprises a main body 1 of said spiral die assembly, a mandrel 2 having a spiral groove 3 around its outer surface inserted on the inside of the main body and a rotatable ring 5 having a desired width fitted along the outer surface of the spiral groove 3 and fitted in a portion of a die cylinder which forms the inner surface of the main body 1 in a range corresponding to both ends of the spiral groove 3.

1. Field of the Invention
The present invention relates to a spiral die assembly suitable for forming tubular body by an extrusion molding.
2. Brief Description of the Related Art
In conventional extrusion molding methods for forming tubular bodies, a spider die which frequently causes spider marks and a straight die which tends to cause uneven thickness in molded products are not employed, but spiral die assemblies as shown in FIGS. 4A and 4B are employed instead. In such a spiral die assembly a spiral groove 3 is formed on the outer surface of a mandrel 2 which is inserted in a main body 1 of the die assembly. Molten resin supplied from an extruder is usually introduced into the mandrel via an inlet 31 and divided its flow path into two to eight flow paths at a dividing inlet 32, and finally moved to a die lip (rightmost end of FIG. 4A or 4B) via the spiral groove 3. Depth of the spiral groove 3 is gradually decreasing toward the die lip. Portion of molten resin is moved between the inner surface of the main body 1 and the mandrel 2 as molten resin being moved in the spiral groove 3 and its flow is adjusted in a reserving portion 33 (see FIG. 4B) and a straight portion (a land portion: a rip gap portion), so that molten resin reaches to the die lip in an adjusted flow. Sometimes, a spiral die assembly having no reserving portions (see FIG. 4A) is used depending on products to be molded.
Except the above-mentioned spiral die assemblies, various dies have been proposed in order to homogenize molten resin at the outlet of the die lip.
1) When a porous pipe is formed out of high viscous material, the material between a cylinder and a mandrel is heated by frictional force in order to disperse strands after passing a breaker plate or in order to adhere and to polymerize powdered material. The frictional heat is obtained by rotating the cylinder. The Japanese laid open patent No. 8-57936 (reference 1) discloses a die arrangement such that a die for extrusion molding and a molded product can be relatively rotated (see FIG. 5). The relative rotation is continuously or intermittently executed and is controlled its rotational mode and degree of the rotation so as to control properties of the molded products.
2) The Japanese laid open patent No. 59-9035 (reference 2) discloses a method to modify a thickness of an extruded pipe by controlling a gap between a die and a rotatable eccentric nipple (mandrel) and also controlling a detected temperature difference in a circumferential direction. In the extruder for forming the pipe, a cross head 62 in which a resin passage between a die 64 and a nipple 65 is eccentrically arranged except at the end of the resin passage (see FIG. 6) and a sensor is arranged to detect temperature of molten resin in the radial direction. The die and the nipple are arranged such that either one of them can be rotated around an axis for preventing uneven thickness in the extruded product.
3) The Japanese laid open patent No. 10-29237 (reference 3) discloses a molding method for forming laminated products out of different materials. Distributed and leaked different materials from spiral grooves can be laminated in a predetermined order by rotating a mandrel equipped with the spirals (see FIG. 7).
Usually in a die assembly, a molten resin flow along the inner surface of the cylinder (hereinafter referred as "land flow") and molten resin flows in spiral grooves (hereinafter referred as "spiral flow") are moved to a die lip as a flow ratio between the two molten resin flows being kept in a fixed balance. This fixed balance is very important in order to keep good appearance in the molded pipes. When the balance breaks, defects such as streak, uneven appearance, uneven thickness, uneven strength and the like appear in the molded pipes.
An appropriate balance between the land flow and the spiral flow is empirically determined according to a width of the spiral groove, a depth of the groove, a pitch of the groove and the like. And the appropriate balance is also varied according to a flow rate, a temperature, a visco-elasticity of molten resin and the like. When the balance is varied, weld marks are frequently formed in the molded products, since proper measures against such varied balance are not established. Sometimes molten resin is kneaded in order to eliminate weld marks by raising the temperature and lowering the visco-elasticity of molten resin, but uneven thickness, contamination (die marks etc.) due to thermal degradation of the molten resin and like are caused.
A round die or a flat die is used in the extruder for forming molten and kneaded resin into a desired shape.
A pipe body such as film, pipe, blown container, net and the like are formed through the round die. Performance of the die for distributing molten resin uniformly around a circumference on an imaginary plane, affects quality of the pipe body, except molding asymmetrical product out of molten resin extruded from the die lip. When the molten resin is not distributed uniformly or homogeneously around the circumference, the following defects are caused: (a) uneven thickness, (b) deteriorated appearance (weld mark, streak, unevenness, wrinkle and so forth), (c) dispersion in strength, (d) dispersion in electrical properties, and the like. In addition, the round die has the following problems: difficulty to substitute remaining resin in the groove with different type of new resin, to require a considerable time and a considerable amount of wasted resin for substituting resins.
Since visco-elasticity of molten resin in the die depends on its temperature and pressure, and varies according to its flow rate, a molten resin flow in the die should be controlled variably. But the molten resin flow in a straight portion (land portion) near the die lip should be controlled by a fixed means in order to attain stable and straight flow.
The weld mark (weld line) mentioned in (b) means a fine line appearing in the extruded pipe, in other words "a streaked state" when molten resin being flowed in the grooves are combined but not mixed homogeneously.
These problems cannot be solved in the above-mentioned prior arts (spiral dies shown in FIGS. 4A and 4B and references 1 to 3).


A relative-dielectric-constant measuring apparatus according to the present invention includes an ellipsometer and a capacitance measuring part. The ellipsometer allows non-contact measurements of the film thickness and optical constants of an insulation film formed on the upper surface of a wafer. The capacitance measuring part, on the other hand, allows non-contact measurements of the gap between the insulation film and an electrode and accumulation capacitance. The relative-dielectric-constant measuring apparatus can calculate the relative dielectric constant of the insulation film based on the measured film thickness, gap, and accumulation capacitance. Thus, the relative dielectric constant of the insulation film can be determined without contact and with high precision.

1. Field of the Invention
The present invention relates to a relative-dielectric-constant measuring method and apparatus for measuring the relative dielectric constant of an insulation film formed on the surface of a substrate.
2. Description of the Background Art
One of the most difficult issues associated with finer interconnections in semiconductor devices is the process of interconnecting electronic circuits. A problem with finer interconnections in semiconductor devices is a time delay of electric signals. Thus, in recent years progress has been made in multilevel interconnect technology.
Between two multilevel interconnect lines, an interlayer insulation film (hereinafter simply referred to as an "insulation film") is buried. The insulation film is evaluated from its various physical properties such as insulating properties, mechanical strength, thermal resistance, and relative dielectric constant. Among these, the most important physical property is the relative dielectric constant. The insulation film is desirable to have a low relative dielectric constant for device operation. However, it is known that damages caused to the insulation film by processes such as etching will increase the relative dielectric constant of the insulation film. Thus, the material for the insulation film is selected also from the viewpoint of easy processing and leakage properties, and on the other hand, a technique for measuring the relative dielectric constant of an insulation film becomes important.
As techniques for measuring the relative dielectric constant of an insulation film, for example, there are known a technique for forming a metal electrode on the surface of an insulation film, and a technique for bringing a metal probe into contact with the surface of an insulation film.
However, the technique for forming a metal electrode on the surface of an insulation film might cause changes in the properties of an insulation film due to heat generated during electrode formation, and the like. Also, this technique is destructive measurement, so the measurement is time-consuming.
On the other hand, the technique for bringing a metal probe into contact with the surface of an insulation film provides contact between the metal probe and the surface of an insulation film by way of a liquid metal such as mercury. Thus, there is no need to form a metal electrode on the surface of the insulation film, thus allowing easy and simple measurement. However, high-precision measurement is difficult because of an unstable contact area of mercury. Besides, the use of mercury that is harmful to the human body is also a problem.
In order to solve the aforementioned problems, non-contact measurement of the relative dielectric constant of an insulation film is desirable. As a technique for non-contact measurement of the relative dielectric constant of an insulation film, there is known a technique for applying charge to the surface of an insulation film by corona charging and then obtaining surface charge and potential at that time, thereby to obtain the relative dielectric constant of the insulation film. However, since the insulation film generally has a porous structure to reduce its relative dielectric constant, a large amount of charge applied to the surface of the insulation film will leak into the film. Thus, it is difficult to accurately calculate the relative dielectric constant of an insulation film through analysis that is based on the assumption that charge exists only on the surface of the insulation film.
That is, using either of the aforementioned measurement techniques involves difficulty in non-contact and accurate measurement of the relative dielectric constant of an insulation film.

Method and apparatus for measuring relative dielectric constant