The present invention provides a transmission equalization system for use with an integrated circuit package employing a substrate. In one embodiment, the transmission equalization system includes a signal transmission subsystem having a pair of transmission line conductors located in the substrate and employing a differential electrical signal. The transmission equalization system also includes an equalization subsystem located proximate the pair of transmission line conductors that employs at least one aperture positioned and oriented to provide a substantially equivalent transmission environment for each of the pair of transmission line conductors.

The present invention is directed, in general to integrated circuit packaging, and, more specifically, to a transmission equalization system and an integrated circuit package employing the same.
Reliable operation of electronic devices, especially semiconductor devices, is of a primary importance today. Often the success of a company may depend on the success at which new technology is reliably established, especially in a developing market. Particularly challenging is the general market demand for devices that perform more functions at increasingly faster operating speeds. This demand for increased functionality usually drives the device design toward more complexity, which typically involves higher component densities that may lead to higher packaging densities. Higher densities alone complicate the environment for signal transmissions within the device. The demand for faster signal speeds further exacerbates this condition causing the environments and associated designs that were acceptable at slower signal speeds to cross into the realm of unacceptability.
Signal environments that exist in an integrated circuit die may require special attention to insure that the ever closer proximity of signals will not generate cross-talk or other interference conditions. Although challenging, these on-chip detrimental conditions may be overcome through appropriate layout and shielding designs. High speed, high density packaging substrate design, however, typically involves a constant trade-off between cost and electrical signal integrity requirements. To reduce costs, signals are often routed in a microstrip construction where the signal only references one power or ground plane.
High-speed, high-density substrate design for multilayer, organic flip chip packages also involves constant trade-off between the substrate design rules and electrical signal integrity requirements. Metal planes are frequently used in the substrate stackup for noise shielding and to minimize signal cross-talk. From a signal integrity standpoint, solid metal planes are preferred since they introduce the fewest discontinuities. However, from a substrate manufacturing and reliability standpoint, voided planes are preferred since they promote adhesion between the substrate layers and provide vents for moisture egress at elevated temperatures without causing packaging delamination. Therefore, there is a trade-off between signal integrity and substrate integrity in the design of a substrate.
When a four-layer package substrate is used in a packaging assembly configuration, the traces routed on the top layer are typically subject to several discontinuities in their electrical environment. One source of discontinuity is the microstrip trace routing from the central area employing a semiconductor die to the extremities of the package over a single power or ground plane. The microstrip conductors are often routed in pairs to accommodate a differential signal employed between the two conductors. In this configuration, if either one of the pair of conductors experiences a different electrical environment from an environmental discontinuity, a signal imbalance condition typically occurs. This signal imbalance provides a distortion that may cause signal errors, enhanced reflected noise and losses during signal transmission. Of course, similar problems may occur with multilayer boards having more than four layers or that employ strip line configurations.
Voids in the voided planes typically have a uniform cross-section and display a uniform pattern with respect to the metal sheet. However, these voids typically impact each of a pair of conductors that reside in a layer above in a random manner. This randomness causes an environmental discontinuity thereby providing a different electrical environment to each of the pair of conductors. This causes a signal imbalance condition having the adverse effects mentioned above.
Accordingly, what is needed in the art is a way to provide a metal layer having voids or apertures that does not differentially distort the electrical environment associated with a pair of conductors.
To address the above-discussed deficiencies of the prior art, the present invention provides a transmission equalization system for use with an integrated circuit package employing a substrate. In one embodiment, the transmission equalization system includes a signal transmission subsystem having a pair of transmission line conductors located in the substrate and employing a differential electrical signal. The transmission equalization system also includes an equalization subsystem located proximate the pair of transmission line conductors and employing at least one aperture positioned and oriented to provide a substantially equivalent transmission environment for each of the pair of transmission line conductors.
In another aspect, the present invention provides a method of manufacturing an integrated circuit package. The method of manufacturing includes providing a substrate. The method further includes forming at least a pair of transmission line conductors in the substrate and positioning and orienting at least one aperture proximate the pair of transmission line conductors to provide a substantially equivalent transmission environment for each of the pair of transmission line conductors.
In yet another aspect, the present invention provides an integrated circuit package that includes a substrate. The integrated circuit package also includes a transmission equalization system with a signal transmission subsystem having a pair of transmission line conductors located in the substrate and employing a differential electrical signal. The transmission equalization system also includes an equalization subsystem located proximate the pair of transmission line conductors that employs at least one aperture positioned and oriented to provide a substantially equivalent transmission environment for each of the pair of transmission line conductors.
The foregoing has outlined, rather broadly, preferred and alternative features of the present invention so that those skilled in the art may better understand the detailed description of the invention that follows. Additional features of the invention will be described hereinafter that form the subject of the claims of the invention. Those skilled in the art should appreciate that they can readily use the disclosed conception and specific embodiment as a basis for designing or modifying other structures for carrying out the same purposes of the present invention. Those skilled in the art should also realize that such equivalent constructions do not depart from the spirit and scope of the invention in its broadest form.

Transmission equalization system and an integrated circuit package employing the same


An improved disc-type ground tillage implement for tilling ground. The implement has a frame connectable to a towing machine and having multiple pairs of gangs of blades connected to the frame. The frame is supported by a plurality of wheels which may be used to raise and lower the implement in relation to the ground. Each of the pairs of blade gangs includes a front and a rear gang, there being a pair of outer right blade gangs, at least one pair of center blade gangs and a pair of outer left blade gangs. The front and the rear gangs of the outer right and the outer left gang pairs converge as they extend out away from the left-right center of the frame and the outer left and outer right gangs are non-continuous with a center gang pair.

1. Field of the Invention
The present invention relates to farm machinery and more particularly to disc-type tillage equipment.
2. Description of the Prior Art
Prior to this device disc-type tillage devices had many disadvantages such as: they could turn in only one direction while working the ground, or could not be turned at all without raising the device; larger disc devices had a large width to depth ratio so they did not follow changes in ground elevation well; they could only be operated in one direction when trying to get close to an obstacle such as a fence or did not till the ground well near an obstacle; when operated in the direction necessary to get close to an object they often left ridges; and an inexperienced or inattentive operator could easily damage the device by turning or crossing a ditch or ridge.
Traditionally disc-type tillage devices have had two different configurations. One type of disc device is the offset disc which has rows of disc gangs arranged in a V shape, and the other type of disc device is known as the conventional or tandem disc, which has rows of disc gangs arranged in an X shape. With either type, the disc gangs are suspended from a frame which may be connected to a farm tractor, bull dozer or the like.
The offset disc has a row of front discs and a rear row of discs. (The individual discs sometimes being referred to as blades). Each disc or blade is cupped and rotates along a central axis. As the device is pulled, the front row of blades cut the ground and throws it out while the rear row of blades re-cuts the ground and moves it back to about its original position. This type of disc device may only be turned in one direction while in the ground without causing the front blades on the inside of the turn to dig in too deeply. If the blades are allowed to dig in, they will create a ridge and may bend or break. Since, by convention, the rows of blades converge on the left side the offset disc will allow a left-hand turn, but requires lifting out of the ground for a right-hand turn.
The other type of disc device is the conventional disc (sometimes referred to as a tandem) which may generally be designed with a greater width, but has many similar disadvantages. As the conventional disc is pulled over the ground the two rows of front blade gangs (the ones nearest the tractor) cut the ground and throw it outwardly away from the center of the device and then the rear blade gangs re-cut the ground and move it inwardly toward the center of the device. Thus, the ground is tilled and left in generally the same location as it was originally.
This arrangement of gangs is suitable if one is working level ground and traveling straight. However, with an implement of suitable width (left to right) for today's large tractors, the depth (front to rear) is such that when one crosses unlevel ground such as a ditch, the wheels and the blades near the center of the device leave the ground. This not only leaves ground in the bottom of the ditch untilled but also causes the weight of the device to be concentrated on the outer most blades so that they dig in leaving ridges, and possibly even damaging the device. The smaller the depth the less this is a problem, but with the arrangement of the gangs on this type of device, the depth increases significantly as the device gets wider thereby greatly intensifying the problem.
Similarly when one tries to turn the conventional disc device without raising it, the device will tend to pivot around the wheel to the inside of the turn causing the blades on the inside front of the device be pushed considerably against their cup shape. As a result, these blades will cut deeply into the ground and leave a ridge. In a turn such as this it is common for the blades to be bent or broken.

Offset/tandom disc-type tillage implement


Methods of treating language, behavioral and social disorders are described, including methods of treating language disorders associated with electrographic abnormalities in the primary or associative language cortex of persons with autism spectrum disorders, pervasive developmental delay or acquired epileptic aphasia. A language, behavioral and social disorder may be treated by detecting epileptiform activity or an electrographic seizure for a subject's brain and applying neurostimulation to a language cortical region of the subject's brain (e.g., a primary or associative language cortical region). Detection of epileptiform activity or an electrographic seizure and stimulation of language cortex may be performed by a sensing and/or stimulation electrode that is inserted into a subject's brain and connected to one or more neurostimulation devices for monitoring and/or stimulating the language cortex.

Language disorders may have psychological and physiological etiologies. For example, the autism spectrum disorders (autism, pervasive developmental disorder, and acquired epileptic aphasia) are associated with language, behavior and social disability. These disabilities are believed to be a manifestation of electrographic and other functional disturbances in one or more language regions of the brain, as well as regions mediating behavior and social interaction skills. Persons with autism and pervasive developmental disorders, as well as acquired epileptic aphasias such as Landau-Kleffner, have a high prevalence of epilepsy and of epileptiform abnormalities.
Autism (sometimes called "classical autism") is the most common condition in a group of developmental disorders known as the autism spectrum disorders. The Diagnostic and Statistical Manual of Mental Disorders (DSM-IV) defines autistic disorder as a syndrome with qualitative impairments in social interaction, communication, and restricted and stereotyped patterns of behavior, interests and activities. Autism varies widely in its severity and symptoms and may go unrecognized, especially in mildly affected children, or when masked by more debilitating handicaps. Symptoms typically begin before age 3 and may include problems using and understanding language; difficulty relating to people, objects, and events; unusual play with toys and other objects; difficulty with changes in routine or familiar surroundings, and repetitive body movements or behavior patterns.
Other autism spectrum disorders are characterized by delays in the development of socialization and communication skills. Asperger syndrome is an autism spectrum disorder characterized by a greater or lesser degree of impairment in language and communication skills, as well as repetitive or restrictive patterns of thought and behavior. Other autism spectrum disorders include Rett syndrome, childhood disintegrative disorder and pervasive developmental disorder not otherwise specified (usually referred to as PDD-NOS).
Landau-Kleffner syndrome is the best-described syndrome of acquired epileptic aphasia. This condition affects children, usually between the ages of 3 and 7, who previously had no developmental, language, or interactional difficulties. Subject's typically experience a rather abrupt loss of language comprehension and expression, usually coincident with the onset of seizures and a profoundly abnormal sleep EEG. Electrographic status epilepticus (prolonged seizures) in sleep, or continuous spike-wave in slow-wave sleep is typical in acquired epileptic aphasia and in the Landau Kleffner Syndrome (Trevathan E., "Seizures and epilepsy among children with language regression and autistic spectrum disorders." J Child Neurol 2004; 19(S1):549-57). The cause for the aphasia in Landau-Kleffner syndrome is uncertain. The seizures and the aphasia are believed to reflect abnormal brain functioning or the aphasia may be a consequence of the seizure discharges.
Clinically evident seizures, as well as subclinical epileptiform discharges and epileptiform electrographic abnormalities, may exacerbate or even cause the cognitive, language and behavior disorders characterizing the autism spectrum disorders and acquired epileptic aphasias. As many as 75% of children with autism have electroencephalogram (EEG) abnormalities and up to 46% have clinical seizures (Hughes J R, Melyn M., "EEG and seizures in autistic children and adolescents: further findings with therapeutic implications." Clin EEG Neurosci 2005; 36(1):15-20; Hrdlicka M., Komarck V., Propper L., Kulisck R., Zumrova A., Faladova L., Havlociocova M., Sedlacek Z., Blainy M., Urbanek T., "Not EEG abnormalities but epilepsy is associated with autistic regression and mental functioning in childhood autism." Eur Child Adolesc Psychiatry 2004; 13(4):209-213; Tharp B R, "Epileptic encephalopathies and their relationship to developmental disorders: do spikes cause autism?" Ment Retard Dev Disabil Res Rev 2004:10(2):132-134). The prevalence of epilepsy and epileptiform abnormalities in persons with autism spectrum disorders is highest in those with moderate to severe retardation, motor deficits, severe receptive language deficits (Tuchman R., Rapin I., "Epilepsy in autism." Lancet Neurol 2002; 1 (6):352-358), early language regression (Trevathan E., "Seizures and epilepsy among children with language regression and autistic spectrum disorders." J Child Neurol 2004; 19(S1):S49-57) or abnormal development during the first year of life. Epileptic seizures are also associated with progressive regression in children with autism (Rossi P. G., Parmeggiani A., Bach V., Santucci M., Visconti P., "EEG features and epilepsy in patients with autism." Brain Dev 1995; 17(3):169-174).
Currently, there is no cure for autism spectrum disorders or acquired epileptic aphasias. Therapies and behavioral interventions are designed to remedy specific symptoms. Treatment plans coordinate therapies and interventions that target social interaction, verbal and nonverbal communication, and obsessive or repetitive routines and interests. Medications are used to address specific behavioral problems. Furthermore, we are not aware of any randomized clinical trials of treatments for autistic language regression. Some persons with autism improve cognitively when treated with antiepileptic drugs (AEDs). However, the potential for cognitive side effects limits AED use.
As an alternative or adjunct to medication, surgical interventions have also been used as treatments. For example, vagus nerve stimulation (VNS) and multiple subpial transections, or cortical resection, have been performed to treat autism spectrum disorders. The impact of VNS on quality of life and alertness in 6 children with Landau-Kleffner Syndrome and 59 persons with autism was examined utilizing a retrospective subject outcome registry (Park Y. D., "The effects of vagus nerve stimulation therapy on patients with intractable seizures and either Landau-Kleffner syndrome or autism." Epilepsy Behav 2003; 4(3):286-290). Fifty-eight percent of the subjects with Landau-Kleffner Syndrome and 78% of the subjects with autism reported improved quality of life and enhanced alertness. It is not possible to differentiate whether these effects were related to reductions in seizures or to an independent mood effect given this study design.
Other surgical treatments include subpial transection. For example, a small number of children with Landau-Kleffner syndrome who have not responded to antiepileptic medications have been treated with subpial transection (Nass R, Neville B. G., Harkness W. F., Cross J. H., Cass H. C., Burch V. C., Lees J. A., Taylor S. C., "Surgical treatment of severe autistic regression in childhood epilepsy." Pediatr Neurol 1997; 16(2): 137-140). This procedure severs interneuronal connections perpendicular to the trajectory of the cortical neuron. The procedure is performed in the language area of the frontal lobe coincident with the maximal electroencephalographic abnormalities. The procedure is thought to inhibit the propensity for abnormal electroencephalographic discharges to propagate to adjacent neurons while preserving fiber tracts subserving motor and sensory function.
Thus, there is a need for treatments and systems for treating language, behavioral and/or social disorders, and particularly those related to autism, pervasive developmental disorders, and acquired epileptic aphasias. A system and method of using such a system could benefit individuals with autism spectrum disorders and acquired epileptic aphasias, for which there is no effective treatment.

Treatment of language, behavior and social disorders


Techniques and screening messages based on tags in an automotive environment, such as, messages communicated via a communication bus, like the CAN bus. Messages can be tagged with either a binary or probabilistic tag indicating whether the message is fraudulent. ECUs coupled to the CAN bus can receive the messages and the message tags and can determine whether to fully consume the message based on the tag.

Modern automobiles include a number of sensors, controllers, and processors. These devices often communicate signals and/or messages via a common bus. For example, messages can be communicated between devices via a controller area network (CAN) bus. Consumption of malicious signals and/or messages can directly impact the behavior of the automobile.


A separator having a separator housing, a separator wheel arranged inside the separator housing and having an axis of rotation (X), and a guide vane assembly arranged in the separator housing, an annular space being provided between the guide vane assembly and the separator housing radially (R) perpendicular to the axis of rotation (X) and a separation zone being provided between the guide vane assembly and the separator wheel, and the guide vane assembly having a plurality of vertical guide vanes. In order to increase separation performance, at least one deflecting element is arranged between at least two adjacent vertical guide vanes and has at least one downward-pointing curved and/or bent portion.

By separation is meant in general the sorting of solids according to certain criteria such as mass density or particle size. Winnowing is a group of separation processes in which a gas stream, the so-called separating air, is used to accomplish this sorting. The active principle is based on that fact that fine or small particles are influenced more strongly and carried along by the gas stream than are large or coarse particles.
Wind separators are used for example for the classifying of coal dust and other grist of a mill. The goal here is to separate particles which have been ground sufficiently small after the grinding process from particles needing further grinding. These two particle groups are also called fines and tailings. Basically, a separator may also be used for the sorting or classifying of solids of different origin.
There are various kinds of wind separators. A major distinguishing criterion is the manner in which the solid substance being separated, or the feedstock, and the separating air are introduced into the separator. Thus, solids and separating air may either be introduced separately from each other or jointly.
A wind separator, in which solids and separating air are introduced jointly, is known from US 2010/0236458 A1. The disclosed wind separator is used for sorting of coal dust. The mixture of coal dust and separating air is admitted to the separator housing from underneath. The inlet volume flow of the gas-solids mixture flows entirely from the outside into the interior of a guide vane assembly. The guide vane assembly has a multitude of deflecting elements, between which the mixture flows. The deflecting elements are tilted relative to the horizontal by 50 to 70° and secured. Inside the guide vane assembly is situated a separator wheel. The separator wheel is driven in rotation and has a multitude of fins, running substantially vertically. Fine particles by virtue of the flow and despite the rotation of the separator wheel can get in between the fins of the separator wheel and are afterwards sucked out at the top. Coarse particles, on the other hand, strike against the fins and are bounced back in this way and finally drop down because of gravity.
Another separator with deflecting elements tilted to the horizontal is the louver separator of type LJKS, known from the article "State of the art of separator technology—separators for bulk goods" by S. Bernotat, appearing in ZKG International 43 (1990) February, No. 2, Wiesbaden, DE.
In other wind separators the guide vanes of the guide vane assembly are arranged vertically, such as in WO 2014/124899 A1. The guide vanes proposed there may be straight or curved. Similar wind separators are also known from the publications EP 1 239 966 B1, EP 2 659988 A1, DE 44 23 815 C2 and EP 1 153 661 A1. In the case of EP 2 659 988 A1, the fins are adjustable.
In EP 1 153 661 A1, both vertical and horizontal fins are used, which on the whole should result in a more uniform flow. Furthermore, the horizontal partitioning of the flow path should have the effect that the flow is directed at the separator wheel along its entire height, which should help improve the separating efficiency.
A mill with an integrated separator is known from US 2012/0138718 A1.
However, the deflecting elements of US 2010/0236458 A1 and the fins of EP 1 153 661 A1 represent a kind of obstacle to the flow. Especially in US 2010/0236458 A1, the mixture of feedstock and separator air flows almost vertically against the deflecting elements. In this way, a backflow or a swirling of the mixture may occur before flowing into the guide vane assembly.

This patent is subject to a terminal disclaimer.


In an echo canceller having PCM interfaces used in a PCM telephone circuit, an echo estimator (6) is coupled with a receive path through a linear code converter (13) for obtaining a linear receive input signal, and a first subtractor (7) is inserted after a linear code converter (13) in a transmit path for cancelling an echo component in a linear transmit input signal and for transmitting a first residual echo to the far-end talker. A non-linear quantization processor (24) is inserted between the first subtracter (7) and the echo estimator (6), and produces a new echo estimate containing quantization noise, based on an echo estimate which is derived from the echo estimator (6). The filter coefficients in the echo estimator (6) are updated by using the second residual echo obtained by the second subtracter (23) separately prepared from the first subtracter (7). The new echo estimate from the non-linear quantization processor (24) can drastically cancel not only the echo but also the quantization noise in the linear transmit input signal without any effect on the performance of the echo estimator (6). The echo canceller having the non-linear quantization processor (24) and subtracters separately prepared for transmitting the first residual echo to the far-end talker and for updating the echo estimator (6) can achieve a high echo return loss enhancement even for a multi-linked PCM channel.

In a telephone channel having a four-wire long distance PCM trunk, hybrid circuits are connected for 2wire-4wire conversion. The hybrid circuits at each end of a call interface two-wire subscriber loops to the four-wire long distance PCM trunk. The near-end echo of a far-end talker on the transmit path through a hybrid circuit is returned to the talker who perceives it as an echo.
FIG. 1 shows the basic structure of a conventional echo canceller with 64 kbit/s PCM interfaces for cancelling the echo in a PCM telephone channel. The input and output signals of the echo canceller (1) are assumed to be digitized with either 8-bit .mu.- or A-law PCM and are processed digitally. The 8-bit PCM receive input signal Rin from the far-end talker is fed to a receive input port (2). The 8-bit PCM transmit input signal Sin at a transmit input port (5) is returned as the near-end echo over the echo path including at least a PCM coder (16) and a PCM decoder (15), which corresponds to a circuit from a receive output port (3) to the transmit input port (5) through the hybrid circuit (4). The 8-bit PCM receive output signal Rout from the port (3) is decoded to an analog signal through the PCM decoder (15). The hybrid circuit (4) transmits the analog signal through a port (17) to the near-end talker. A part of the analog signal at the hybrid circuit (4) is leaked as an echo to the transmit path and is encoded through the PCM coder (16). The conventional echo canceller (1) synthesizes a replica of the near-end echo by an echo estimator (6) and suppresses the echo by subtracting this replica at a subtracter (7) from a linear transmit input signal, which is converted from the 8-bit PCM transmit input signal Sin through a linear code converter (13). A residual echo derived as an output of the subtracter (7) is fed to a center clipper (8) as a non-linear unit to clip the residual echo having a level lower than a given threshold, and/or is converted to a 8-bit PCM code through a non-linear code converter (14) to be transmitted from the transmit output port (9) of the echo canceller (1) having a 64 kbit/s PCM interface.
A double talk detector (10) is provided for detecting the double talk situation between the near- and far-end talkers, and a low level detector (11) is also provided for detecting a linear receive input signal having a level lower than a given threshold derived through a linear code converter (12). In the double talk detector (10), levels of the linear transmit input signal and the linear receive input signal are compared to detect the double talk situation.
In the echo estimator (6) which comprises an adaptive digital transversal filter having a finite impulse response (FIR), filter coefficients of the filter are adaptively updated according to the residual echo and the linear receive input signal stored in the echo estimator (6) so as to minimize the level of the residual echo either at every sample time or at every sample interval. The updating of the filter coefficients is inhibited during double talking or during the period when the low level detector (11) detects a linear receive input signal having a low level.
In the case of an echo canceller having PCM interfaces so as to connect it with the PCM telephone channel, the quantization noise is always induced by the non-linear process of the PCM codecs (15,16) in the echo path, resulting in degraded performance of the echo cancellation. Because the quantization noise in the linear transmit input signal is independent of the linear receive input signal, it cannot be estimated by the echo estimator (6) which uses only the linear processing by the FIR filter. The updating of the filter coefficients in the echo estimator (6) by the residual echo containing the quantization noise also results in a deviation of the impulse response of the echo estimator (6) from the optimum, inducing a degradation of the echo cancellation performance. The echo cancellation performance is limited by the amount of the quantization noise.
In a multi-linked PCM telephone channel, PCM codecs are inserted after the PCM decoder (15) and/or before the PCM coder (16) in the echo path. The tandem connection of the PCM codecs produces a large amount of the quantization noise relative to the number of PCM codecs. When a low rate codec like a 32 kbit/s ADPCM or a codec having a rate lower than 16 kbit/s is applied instead of the PCM codec, a larger amount of quantization noise is produced by the codec. Thus, the echo cancellation performance always degrades due to a PCM tandem connection or a low rate codec connection. This degradation cannot be avoided by using only the linear processes in a conventional echo canceller. It is therefore very difficult for the echo canceller (1) to achieve a high echo cancellation performance under these circumstances.
FIG. 2 shows a conventional acoustic echo canceller (40) used in a tele-conference or a TV conference. A linear receive input signal and a transmit input signal are directly fed to the receive input port (2) and the transmit input port (5), respectively. A linear receive output signal Rout at the port (3) is fed to a speaker at the port (172) after converting it to an analog acoustic signal through a D/A converter (18). An analog acoustic signal at a port (171) from a microphone is also converted to a linear transmit input signal through an A/D converter (19). A part of the analog acoustic signal from the speaker is also fed to the A/D converter (19) in the transmit path through the microphone as an echo in the echo path.
The A/D and D/A converters (18,19) induce a quantization noise which cannot be cancelled by the acoustic echo canceller (40) due to non-linear processing as in the PCM echo canceller mentioned above. If the length of a quantization bit in the A/D and D/A converters (18,19) is short, a large amount of quantization noise is produced, resulting in a degradation of acoustic echo cancellation. A longer bit length requires a huge amount of processing in the acoustic echo canceller having a very high sampling frequency due to a wide transmission bandwidth and a long echo path delay.
FIG. 3 is a basic structure of another echo canceller (50) having a non-linear quantization processor to reduce the quantization noise in the linear transmit output signal for a PCM telephone channel as described in U.S. Pat. No. 5,247,512. The echo canceller (50) can cancel the quantization noise when the echo estimate (6) successfully converges to provide an accurate echo estimate.
The echo canceller (50) comprises an echo estimator (6); a non-linear quantization processor (20) for generating a new echo estimate containing a quantization noise as the output corresponding to the echo estimate from the echo estimator (6); a subtracter (7) for obtaining a residual echo as the linear transmit output signal sent to the far-end; a switch circuit (21) for connecting either the output of the echo estimator (6) or the output of the non-linear quantization processor (20) with the subtracter (7) so as to subtract the output at a first input of the subtracter (7) from the linear transmit input signal at a second input of the subtracter (7); and a switch controller (22). The switch controller (22) controls the switch circuit (21), the non-linear quantization processor (20) and the echo estimator (6).
In the non-linear quantization processor (20), the echo estimate from the echo estimator (6) as an input at the port (201) is converted into a PCM code and then is inverse-converted again to a linear code having a quantization noise as a new echo estimate which is output at a port (202). The non-linear quantization processor (20) can therefore provide a new echo estimate which has the same value as that of the linear transmit input signal including a quantization noise or a value very close to it. In the echo estimator (6), the adaptation speed for the filter coefficient updating is adjusted according to the convergence/divergence state of the echo estimator (6) which is determined by the switch controller (22).
Two processing modes are provided, that is, a linear processing mode and a non-linear processing mode. The processing mode is chosen by the switch controller (22), according to the convergence/divergence state of the echo estimator (6). In the switch controller (22), the power of the linear transmit input signal and the power of the residual echo are calculated for comparison so as to judge the convergence/divergence state in the echo estimator (6) and select either the non-linear or linear modes.
When the switch controller (22) judges the echo estimator (6) to be convergent, the switch controller (22) selects the non-linear processing mode and then controls the switch circuit (21) to connect the non-linear quantization processor (20) to the subtracter (7). The output of the non-linear quantization processor (22) is then supplied through the switch circuit (21) to the first input of the subtracter (7) so as to cancel the echo and the quantization noise in the linear transmit input signal at the second input of the subtracter (7).
In the linear processing mode selected by the switch controller (22) due to divergence in the echo estimator (6), the output of the echo estimator (6) is directly supplied to the subtracter (7) through the switch circuit (21) to obtain the same residual echo as in the conventional echo canceller. The divergent echo estimator (6) cannot provide an echo estimate which is sufficiently accurate for generating an appropriate quantization noise for the linear transmit input signal. The non-linear processor (20) is therefore bypassed by controlling the switch circuit (21) in this mode.
In the initial stage of the echo canceller (50), the linear processing mode is selected due to divergence. After judgement of convergence in the echo estimator (6), the switch controller (22) selects the non-linear processing mode to ultimately cancel the quantization noise in the linear transmit input signal.
During double talking, the switch controller (22) constrains the switch circuit (21) to bypass the non-linear quantization processor (20), as the near-end talker's talkspurt is considerably affected by the non-linear quantization processor (20). The same procedures for the adaptation speed in the echo estimator (6) are executed as those in the above conventional echo canceller in the double talk situation or the low level situation in the linear receive input signal.
FIG. 4 shows a basic configuration of the non-linear quantization processor (20). The non-linear quantization processor (20) comprises a multi-output non-linear quantizer (210) and an optimum echo estimate selector (211). The multi-output non-linear quantizer (210) provides a plurality of quantized echo estimates for the optimum echo estimate selector (211) as candidates for an optimum echo estimate giving a minimum level of the residual echo of the output at the subtracter (7). These candidates of quantized echo estimates include the quantized echo estimate directly corresponding to the echo estimate from the echo estimator (6) through the port (201) and also quantized echo estimates near to it. The optimum echo estimate can be provided as the output of the optimum echo estimate selector (211), giving the minimum level of the residual echo among the candidates.
The non-linear quantization processor (20) can search the optimum echo estimate from a wide range of candidates to reduce the quantization noise in the transmit input signal. The non-linear quantization processor (20) can always suppress the residual echo as much as possible, even if the echo estimate from the echo estimator (6) largely deviates from the linear transmit input signal with incomplete convergence. This means that the deviation of the filter coefficients in the echo estimator (6) cannot be precisely updated by this minimized residual echo so as to make the echo estimator (6) converge rapidly during the incomplete convergence, resulting in incomplete cancellation of large echo and quantization noise. It is thus difficult for the echo canceller (50) to achieve a very rapid convergence in the echo estimator (6) for a large echo and to perform excellent cancellation of echo having a wide dynamic range of levels and quantization noise.
In the above circumstances, no previous echo canceller was able to provide sufficiently high performance with a rapid convergence in the echo estimator (6) and excellent cancellation of the echo having a wide range of levels and the quantization noise. This invention however provides a new echo canceller which can resolve these problems.

Echo canceller using two residual echoes


A method for fitting a hearing aid (1) comprises the steps of recording the audiogram of the hearing aid user (6), selecting a fitting setting, using the hearing aid (1) for a period of time, and logging data representing information on the sound environment in which the hearing aid is being used. The method comprises the steps of providing a computer based questionnaire on a first computer (2), said questionnaire comprising questions directed to the hearing aid users subjective experience with using the hearing aid, correlating questionnaire data with logged data, and selecting and, implementing a new hearing aid setting into the hearing aid. The invention further provides a method for optimizing the function of a hearing aid, and a computer program product.

1. Field of the Invention
The present invention relates to hearing aids. The invention further relates to methods of adjusting hearing aids. The invention more specifically concerns a method for fitting a hearing aid to the needs of a hearing aid user comprising recording the audiogram of the hearing aid user, selecting a fitting setting and implementing the setting in the hearing aid, using the hearing aid by the hearing aid user for a period of time, and logging data representing information on the sound environment in which the hearing aid is being used, said data being logged in a memory in the hearing aid.
Hearing aids comprise a number of different parameters which are adjusted by an audiologist, or the hearing aid fitter, to the needs of the individual hearing aid user before the hearing aid user starts using the hearing aid. This adjustment is based on a measured audiogram for the hearing aid user. The audiogram provides information on the hearing loss in specific frequency ranges. The parameters are on one hand a number of basic parameters for adapting the hearing aid to the specific hearing loss of the hearing aid user, e.g. the amplification in different frequency bands and the compression. On the other hand more complex features may need adjustment or may be activated or deactivated, e.g. the feedback cancellation, the directional characteristic, transposing higher frequencies to a lower frequency range, noise reduction functions, choice of input source etc. These parameters, especially the first mentioned, are usually also adjusted to different types of listening situations relevant to the individual hearing aid user.
Typically a number of programs are defined for a hearing aid. These programs comprise preselected settings of all parameters in order to optimize the listening situation in specific sound environments. These programs are set up in order to improve the listening situation in different situations, e.g. car driving, listening to music, meeting etc. The hearing aid user can then select the appropriate program in a given sound environment.
2. The Prior Art
The hearing aid may be provided with features for automatic program selection, as described in WO-A1-2007/045253, based on an analysis of the sound environment. However, the hearing aid user will have the possibility of overruling such a selection.
The different parameters, or the settings within different programs, will usually need fine tuning or readjustment at least once, and often several times during the lifespan of the hearing aid. The need for fine tuning often arises as the hearing aid user gets some experience with using the hearing aid in different types of sound environment. The need for fine tuning may also arise from changes in the hearing loss.
The fine tuning is often based on a discussion between the hearing aid user and the fitter, e.g. when the hearing aid is in for a service check. The fine tuning may also be based on a paper questionnaire or diary filled out by the hearing aid user before or during visiting the fitter. Such a questionnaire comprises different categories of questions directed to areas such as physical and functional aspects (e.g. the mechanical fitting of the hearing aid to the ear), general sound quality (e.g. experience with feedback), speech intelligibility in quit surroundings, speech intelligibility in noisy surroundings, soft sounds, loud sounds, comfort in noise, own voice etc.
Based on the subjective response on these questionnaires a score can be achieved within each category, and the fitter may suggest a change in adjustment of the hearing aid.
It is known from EP-A1-1376440 to have a system for handling forms for information over the internet. This disclosure is directed to order forms comprising data for manufacturing an ear piece at a central production facility.
U.S. Pat. No. 7,200,237 discloses a method for performing a general upgrade of the software on a hearing aid by sending the software over the internet.
Besides the subjective questionnaires, the fitter may also have access to data on the sound environment in which the hearing aid has been used. It is known from WO 2007/045276 to log, by the hearing aid, data representing mal information on the sound environment in which the hearing aid is being used.


A trailer sidewall structural system comprises a plurality of interconnecting modules, with each module having a dissimilarly configured, pair of opposing elongate, lateral edges. Such pair of lateral edges are configured so as to have a first edge of a first module, engage and interlock with a complemental second edge of an abutting second module, the modules being otherwise identical. The complemental lateral lengths of each module form a pair of opposing keyways, that permit interlocking of the lateral edges of two otherwise identically configurated, planar modules, thus providing for a variable length of modules, as may be dictated by the longitudinal dimensions required of a trailer sidewall, and the transverse dimensions of each such module are being adapted to be secured to the deck and top rails located along the trailer sidewalls.

Over the road trailer-tractors are a major single factor in our industrial economy and they contribute to the prompt and economical transport of goods throughout this nation. The separatable element (usually a multi-wheeled entity) undergoes many hauls that is sometimes switched from tractor to tractor. Efficient and durable fabrication of trailers is a major construction industry. A means to permit fabrication of trailer sidewalls of variable length and durability is an important business goal. There is an existing industry for the building of trailers from rigid-metallic panels, or modules, having standardized dimensions. While the abutting panel heights may be commonly matched, the sidewall lengths depend upon the number of modules arrayed, and interlocked, and needed to produce a given length of a trailer sidewall.
Interlocking, along lateral vertical edges, of the modules is a feature crucial to present smooth, sidewall products. Smooth sidewalls reduce air drag and improve road mileage of tractor-trailers. Fabricating sidewall modules is a chore of continuing concern, to combine reasonable cost, to foster efficient assembly of the abutting panels, and to provide for long-term sidewall durability, given the over-the-road physical stressors that burden trailer longevity.

Trailer external smooth side assembled with structural modules


The present invention relates to a coating composition comprising: a) silanized colloidal silica particles; b) alkyd-containing binder; c) a carrier fluid, wherein the coating composition is essentially free from a cobalt-based drier. The present invention also relates to a method of producing a coating composition and the use of such coating composition for coating of a substrate.

It is commonly known in the art to use siccatives in coating dispersions such as paint dispersions. Typically, such siccatives contain metal soaps (metallic driers). Metal soaps may contain for example cobalt, lead, vanadium, or manganese but also soaps with other metals are available in the art. A previously preferred metal soap was cobalt soap due to its excellent performance with respect to drying time. Such metallic driers also contribute to reduced tackiness. Also, such driers are dissolved in suitable solvents, normally in aliphatic hydrocarbons which introduce up to 1 wt % volatile organic compounds to a coating dispersion or emulsion. Such driers, however, are hazardous to human health and may even be cancerogeneous.
The amount of drier used in waterborne systems is often used in higher concentrations than in solventborne systems to compensate for losses of primary driers (or active driers) such as cobalt, vanadium and manganese based driers.
While the solids portion of a coating dispersion adheres to the coated substrate, the solvent component if solventborne evaporates and causes environmental concern. The solvent materials are mostly volatile organic compounds (VOCS) that contribute to the creation of ozone (smog) in the lower atmosphere and are toxic to human health. Some solvents may also be classified as hazardous air pollutants (HAPs).
Alkyds are known to be used in coatings, in particular in paints. Such paints may be applied by means of solventborne and waterborne formulations. However, in particular waterborne paints may suffer from disadvantages such as low drying speed, drier incompatibility, through-drying, surface defects, poor hardness etc.
The present invention intends to provide an environmentally adapted coating composition which considerably reduces the metallic drier (active drier) content therein, or even completely omits the presence thereof in order to minimize evaporation of volatile organic compounds as a coating composition is drying on a substrate and hazardous metals, in particular cobalt, remaining in the formed coating. A further objective of the invention is to provide reduced yellowing of coatings.
The invention also intends to provide a composition which improves the vapour permeability of the film formed from e.g. a coating dispersion. A further objective is to reduce the drying time subsequent to application of such coating composition to a substrate. A further objective is to provide a reduced tack-free time. Another objective is to provide an improved dirt-pick up resistance to the film formed from the coating composition. A further objective of the invention is to improve the block resistance of the composition. A further objective is to provide an acceptable hardness to the coating. A further important objective of the present invention is to provide a stable coating composition which is not liable to gelling or precipitation. It is thus a requirement that the coating composition can be stored for a certain period of time, for example at least one year or at least two years without detrimental effects caused by separation or gelling. According to one embodiment, yet an objective is to provide an environmentally adapted system which is waterborne. According to yet one embodiment, the invention also intends to provide a composition having an improved hiding power.
The present invention intends to solve the problems referred to above.


A multiplexed fiber optic sensor system including a first optical fiber having a first end arranged to receive light from a light souce, at least two optical fibers having diameters smaller than the first optical fiber, and at least two fiber optic sensors, each of the at least two smaller diameter optical fibers arranged between the first optical fiber and one of the sensors for transmitting light from the first optical fiber to that sensor. The sensors can be static or dynamic pressure sensors, strain sensors, temperature sensors or other environmental sensors.

Active sound control systems often require, in addition to actuator and electronic control components, specialized sensor devices. The requirements associated with such sensors to a large part are determined by the particular active control approach employed, as well as the kind of performance expected of the sound controlling system. One type of sensor is the microphone, which measures sound or the existence of atmospheric pressure waves in a particular area of interest.
These sensors are often used as components of instrumentation packages employed on missiles, satellites or other rocket payloads, aircraft or other mechanisms in which atmospheric pressure waves, acceleration or vibration can be quite severe. When in employed in such environments, weight becomes a critical variable. A 10 gram weight differential at rest becomes a 150 gram weight difference during a 15G acceleration event.
While various electro magnetic microphones offer high levels of sensitivity and good performance in terms of size and efficiency, these electromagnetic devices are often very sensitive to EMI and aren't always physically robust. The hardening of these devices to EMI almost invariably causes an increase in physical size, weight, sensitivity and expense.
The main emphasis has been given to phase modulated interferometric fiber optic sensors for detecting pressure and strain. Though interferometric type sensors offer very high sensitivity, polarization effects and high cost have delayed the implementation of these sensors to commercial applications. Frequency modulated sensors utilizing fiber gratings are very promising for monitoring strain in different location along one fiber.
One solution is to employ microphones that are not electrical or semiconductor in nature. The use of fiber optics for sensors in general, and for sound detection in particular, are one option available to designers which simultaneously solves the EMI sensitivity, size and weight problems inherent in electrical sensors.
Since fiber optics use light rather than electricity, a fiber optic microphone is generally insensitive to EMI, or EMF and is therefore more efficient in an environment that has a large amount of electro magnetic energy. As a result, fiber optic microphones can be located adjacent to or actually attached to electronic devices that generate large electro magnetic fields without negative affects to either the microphone or the area of interest.
Intensity modulated fiber optic sensors, while often less sensitive than comparable interferometric sensors, are simpler and less expensive.
Therefore it is an object of this invention to offer an intensity modulated fiber optic pressure sensor that is small and lightweight.
It is yet a further object to offer a pressure sensor that is EMI insensitive.
It is a further object of this invention to offer a pressure sensor that features high levels of sensitivity.
It is a yet another object offer a pressure sensor that is physically robust.
It is a further object of this invention to offer a pressure sensor that is economical.

The United States of America as represented by the Secretary of the Navy


A method and apparatus for renumbering memory arrays may be used in memory systems having a target memory capacity formed from a number (N) of memory arrays. Upon powerup of the memory system, if a portion (M) of the N memory arrays are identified as defective, the memory arrays are renumbered from 0 to (N-M) to provide a logically continuous memory system having a smaller memory capacity than the target memory capacity.

The present invention relates to memory systems. In particular, the present invention relates to the renumbering of arrays within memory systems.
To reduce manufacturing costs, manufacturers of integrated circuits search for ways to improve yields and reduce the rejection rate for defective individual parts. One method of reducing the rejection rate is to provide redundant or auxiliary circuit components on the integrated circuit. This method is practical where testing can locate the defective component, and the circuit is readily reconfigurable to substitute a redundant equivalent for the defective component. This method is widely used in integrated circuit memory arrays such as random access memories.
Memory circuits are characterized by the regular repetition of multitudes of memory cells. The location of each memory cell is defined by a unique address which typically identifies a particular row and column in the memory matrix arrays. The memory circuit includes row and column decoders that decode different combinations of signals at an address input to the memory circuit. Memory circuits provide for redundancy by including on the same circuit several duplicate rows and/or columns of memory cells to replace any row or column having defective memory cells. Separate decoders are provided for the redundant rows or columns that are programmable using programming elements such as fusible links. Once the integrated circuit is tested and the locations of the defective memory cells are determined, the programmable redundancy decoders are programmed to decode those addresses that correspond to the rows or columns with defective cells. The defective rows or columns are subsequently disabled. This way every time a defective row or column is addressed, a redundant equivalent is selected instead.
Unfortunately, when a memory array has more defective rows or columns than redundant equivalents, the entire memory device must be discarded. Discarding chips having defective bits is wasteful, inefficient and costly.
Accordingly, it would be desirable to provide a memory design which allows a chip with one or more defective memory arrays to be used rather than discarded.

Renumbered array architecture for multi-array memories


A new and distinct strawberry variety, Fragaria×ananassa, cv. ‘Sahara’ is characterized by an early time of flowering, an early time of ripening, an RHS red group (near 41 C to 43 C) fruit flesh color, a terminal leaflet length to width ratio that is longer than broad, and a terminal leaflet base shape that is acute.

The present invention relates to a new and distinct strawberry variety. The varietal denomination of the new variety is 'Sahara'. The new variety was designated by the breeder as 'Sahara 06.43H.79'. The new variety of strawberry was created in a breeding program by crossing two parents in 2006 in Cartaya (Huelva), Spain; in particular, by crossing as seed parent an undistributed strawberry parent designated '02-125' (unpatented) and as pollen parent an strawberry parent designated '03.98' (unpatented). Each parent is a selection from breeder's program and has not been commercialized.
The resulting seedling of the new variety was grown and asexually propagated by runners in Cartaya (Huelva), Spain, 7° W., 37° N., 45 feet elevation. Propagation by runners included propagation runners, separately for each varieties, first into a Screen-House, and after in the fields. Clones of the new variety were further asexually propagated and extensively tested. Each variety was reproduced by stolons in the nurseries. In order to establish and bring to health the initial head clones, mother plants that had developed several several stolons were subjected to a heat treatment, or Thermoterapy, at 36° C.-37° C. for 3 to 4 weeks. After that treatment, apical meristems were cut and developed (1 apical meristem corresponding to 1 rooting plant) in an in vitro culture for 5 to 6 weeks. This propagation and testing has demonstrated that the combination of traits disclosed herein which characterize the new variety are fixed and retained true to type through successive generations of asexual reproduction.
Among the characteristics which appear to distinguish the new variety from its closest variety of which I am aware, 'Sabrosa' (U.S. Plant Pat. No. 16,558) is a combination of traits which include: a terminal leaflet length to width ratio that is longer than broad for the new variety, as compared to a ratio that is as long as broad for 'Sabrosa,' a terminal leaflet base shape that is acute as compared to obtuse for 'Sabrosa,' a fruit flesh color that is (RHS red group near 41 C to 43 C) as compared to RHS red group near 43 B for 'Sabrosa,' a fruit hollow center that is weakly expressed as compared to absent or very weakly expressed for 'Sabrosa,' and an early time of flowering (50% of plants at first flower) and early time of ripening (50% of plants with ripe fruit) as compared to medium times of flowering and ripening for 'Sabrosa.' The new variety is distinguished from its parents by the following characteristics possessed by 'Sahara' which are different than, or not possessed, by the seed parent designated '02-125' (unpatented) and the pollen parent designated '03-98' (unpatented). (1) Seed parent '02-125' (unpatented) is less vigorous than the plant of the new variety 'Sahara'. (2) Habit plant of seed parent '02-125' (unpatented) flat, whereas in the new variety 'Sahara' is globose. (3) In seed parent '02-125' (unpatented) the position of the inflorescence relative to the foliage is above, whereas in the new variety 'Sahara' is level with. (4) In seed parent '02-125' (unpatented) the fruit size is smaller than in the new variety 'Sahara'. (5) In pollen parent '03-98' (unpatented) the plant is more dense and more vigorous that the new variety 'Sahara'. (6) In pollen parent '03-98' (unpatented) the fruit size is bigger than one of the new variety 'Sahara'. Moreover, the new variety 'Sahara' presents an early time of flowering and ripening, whereas in the pollen parent '03-98' (unpatented) time of flowering and ripening are medium.
The average fruit mass (g/fruit) observed for the new variety 'Sahara' on March 18 was 37.9 g, as compared to 39.3 g for 'Sabrosa' and 24.7 g for 'Camarosa' (U.S. Plant Pat. No. 8,708). The luminosity for the fruit of the new variety at 460 nm was 18.6, as compared to 27.3 for 'Sabrosa' and 31.5 for 'Camarosa.'
The accompanying photographs show typical specimens of the new variety, designated 'Sahara' or '06.43H.79' in the photographs.


A support system for mounting a door, such as a pocket door, on an overhead track is disclosed. The system cooperates with carriage that includes a support rod having a discontinuity at a lower end. The system uses bracket that is adapted for mounting along the upper edge of the door, and an engagement bar that is adapted for engaging the discontinuity on the support rod, so that the door is suspended from the track through the carriage by inserting the discontinuity through the bracket and engaging the discontinuity with the engagement bar, trapping the lower end of the support rod through the aperture in the bracket.

(a) Field of the Invention
This invention generally relates to a system for mounting pocket doors. More particularly, but not by way of limitation, to a system that allows quick mounting and removal of a pocket door from its tracks without requiring the removal of trim pieces around the pocket door assembly.
(b) Discussion of Known Art
Pocket doors, that is doors that supported from a horizontal overhead rail that allows concealment of the door into an adjacent wall, have been used for over a century. The space saving advantages of these doors have long been understood, but their popularity has fallen short of expectations. One significant reason for the underutilization of these doors is that the doors are difficult to replace when damaged. The source of these difficulties is that the most of the mechanism for supporting the door is concealed within a wall. Frequently, removing the door for repair or replacement requires destruction of the wall and associated trim, and thus the cost of repairing a damaged door or mechanism often exceeds the value of the whole assemby.
An example of a known device for mounting a moveable door, such as a pocket door, is found in U.S. Pat. No. 6,209,171 to Pelletier et al. The Pelletier et al. device uses a bracket that accepts a locking member or stud that cooperates with a pivoting release lever that engages the locking member, trapping it against the bracket. An important aspect of this type of device is that it may be used to mount pocket doors or other doors that are mounted along an overhead track. A significant disadvantage of the system disclosed by Pelletier et al., as well as other known systems, is that release or removal of the door from the supports requires access to each of the locking members. In the case of a pocket door this access is available only in the situation where the wheels, carriage, or portion of the door that fails is exposed, which if properly trimmed is typically not the case.
A typical failure of mechanisms that support pocket doors is the failure of the bearings or wheels that are used to support the door from the overhead track. Failure of these bearings or wheels can make the door difficult if not impossible to move along the overhead track without damaging the door or other components of the system. Thus, if a component fails while the door is inserted into the "pocket" within the wall, it may be impossible to pull the door back out without damaging other components of the system or the wall around the door.
Therefore, a review of known devices reveals that there remains a need for a simple device that supports a pocket door, and allows removal and installation of the entire door from outside the "pocket" or recess that houses the door, without removing the trim that surrounds the door.
There remains a need for a system that allows the removal and installation of the pocket door without having to remove the surrounding trim or wall structure.
There remains a need for a cost effective easily serviceable system for supporting a movable door along an overhead track.
There remains a need for a pocket door support system that can be easily modified in the field and that allows easy installation and removal of the pocket door from the tracks that support the door.


A method includes, in response to receiving implicit signaling of the radio resource, and in response to a plurality of wireless communication system cell-specific static and semi-static input parameters, determining a plurality of output parameters and allocating radio resources using the determined output parameters. Also disclosed are computer readable storage mediums for storing programs that operate in accordance with the methods, as well as various devices that operate in accordance with the methods and that include the computer readable storage mediums.

Certain abbreviations that may be found in the description and/or in the Figures are herewith defined as follows:
3GPPThird Generation Partnership ProjectCAZACconstant-amplitude zero auto-correlationCDMcode division multiplexingCDMAcode division multiple accessCMcubic metricCPcyclic prefixCQIchannel quality indicatorDFTdiscrete Fourier transformE-UTRANevolved UTRANFDMfrequency division multiplexingFDMAfrequency division multiple accessFFTfast Fourier transformIFFTinverse FFTLBlong blockLTElong term evolutionNode BBase StationeNode BEUTRAN Node B (eNB)OFDMorthogonal frequency domain multiplexPARpeak to average ratioPRBphysical resource blockPUCCHphysical uplink control channelQPSKquadrature phase shift keyingRRCradio resource controlRSreference signalRUresource unitSCsubcarrierSC-FDMAsingle carrier, frequency division multiple accessSFspreading factorSNRsignal to noise ratioTTItransmission time intervalUEuser equipmentULuplinkUTRANuniversal terrestrial radio access networkDFT-S-OFDMdiscrete Fourier transform spread OFDM (SC-FDMA based on frequency domain processing)WCDMAwideband code division multiple access
A proposed communication system known as evolved UTRAN (E-UTRAN, also referred to as UTRAN-LTE) is currently under discussion within the 3GPP. The working assumption is that the DL access technique will be OFDMA, and the UL technique will be SC-FDMA.
Reference can be made to 3GPP TR 36.211, V1.0.0 (2007-03), 3rd Generation Partnership Project; Technical Specification Group Radio Access Network; Physical Channels and Modulation (Release 8), for a description in Section 6 of the UL physical channels.
Reference can also be made to 3GPP TR 25.814, V7.1.0 (2006-09), 3rd Generation Partnership Project; Technical Specification Group Radio Access Network; Physical layer aspects for evolved Universal Terrestrial Radio Access (UTRA) (Release 7), such as generally in section 9.1, for a description of the SC-FDMA UL of E-UTRA.
FIG. 1A reproduces FIG. 12 of 3GPP TS 36.211 and shows the UL slot format for a generic frame structure.
As is described in Section 9.1 of 3GPP TR 25.814, the basic uplink transmission scheme is single-carrier transmission (SC-FDMA) with cyclic prefix to achieve uplink inter-user orthogonality and to enable efficient frequency-domain equalization at the receiver side. Frequency-domain generation of the signal, sometimes known as DFT-spread OFDM (DFT S-OFDM), is assumed.
FIG. 1B shows the generation of pilot samples. An extended or truncated Zadoff-Chu symbol sequence is applied to an IFFT block via a sub-carrier mapping block. The sub-carrier mapping block determines which part of the spectrum is used for transmission by inserting a suitable number of zeros at the upper and/or lower end. A CP is inserted into the output of the IFFT block.
In the PUCCH sub-frame structure for the UL control signaling seven SC-FDMA symbols (also referred to herein as "LBs" for convenience) are currently defined per slot. A sub-frame consists of two slots. Part of the LBs are used for reference signals (pilot long blocks) for coherent demodulation. The remaining LBs are used for control and/or data transmission.
It is pointed out that there are different slot formats that are applicable for the exemplary embodiments of this invention that are described below. Reference in this regard may be made to Table 1 "Resource block parameters" on page 10 of 3GPP TR 36.211, V1.0.0 (2007-03).
The current working assumption is that for the PUCCH the multiplexing within a PRB is performed using CDM and (localized) FDM is used for different resource blocks. In the PUCCH the bandwidth of one control and pilot signal always corresponds to one PRB=12 SCs.
It should be noted that it has yet to be determined whether to support 18 SCs on the PUCCH. However, for the purposes of describing the exemplary embodiments of this invention below the exact number of SCs does not matter (e.g., whether there are 12 or 18 SCs that are supported).
Two types of CDM multiplexing are used both for data and pilot LBs. Multiplexing based on the usage of cyclic shifts provides nearly complete orthogonality between different cyclic shifts, if the length of cyclic shift is larger than the delay spread of radio channel. For example, with an assumption of a 5 microsecond delay spread in the radio channel, up to 12 orthogonal cyclic shifts within one LB can be achieved. Sequence sets for different cells are obtained by changing the sequence index.
Another type of CDM multiplexing may be applied between LBs based on orthogonal covering sequences, e.g., Walsh or DFT spreading. This orthogonal covering may be used separately for those LBs corresponding to the RS and those LBs corresponding to the data signal. The CQI is typically transmitted without orthogonal covering.
Of particular interest to the exemplary embodiments of this invention is control channel signaling and, in particular, the use of the PUCCH.
More specifically, the exemplary embodiments of this invention pertain to ACK/NACK signaling. The ACK/NACK signaling format has been determined and is shown in FIG. 2 (originally presented in 3GPP TSG RAN WG1 Meeting #47bis, Sorrento, Italy, Jan. 15-19, 2007, "CDM based Control Signal multiplexing w/ and w/o additional RS", Nokia, R1-070395.
In this approach the ACK/NACK is transmitted by means of a modulated CAZAC sequence, and the ACK/NACK signals of different UEs can be orthogonally multiplexed within the pilot/data blocks using different cyclic shifts of the same base CAZAC sequence. The length of the CAZAC sequence is equal to 12 symbols. Also, coherent transmission with three reference signal (RS) blocks and four data blocks (ACK/NACK) is applied. In addition, block level spreading with SF=¾ is applied to the RS/data blocks.
Different ACK/NACK UEs are multiplexed by means of CDM. As can be seen in FIG. 3, there are in total 12×3=36 code resources available for reference signals and 12×4=48 resources for data signals. Due to the intra-cell orthogonality issue, only part of the code resources can be used in practice (e.g., one half or one third).
In this case these resources are not allocated explicitly because of the fact that the fixed size resource is always needed in certain situations. Instead, implicit resource allocation is used. It has been determined at RAN1 meeting (#49, Kobe) that for non-persistent scheduling the ACK/NAK resource is linked to the index of the control channel used for (DL) scheduling (see 3GPP TSG RAN WG1 Meeting #49, Orlando Fla.-USA, 25-29 Jun. 2007, "Draft Report of 3GPP TSG RAN WG1 Meeting #49 v0.3.0 (Kobe, Japan, 7-11 May 2007), Section 7.13.2).
The actual need for implicit ACK/NACK resources depends on the number of scheduled UEs in the DL (e.g., bandwidth and scheduling strategy). In a practical sense the ACK/NACK capacity is limited by the inter-cell interference. In general, about 10-12 ACK/NACK UEs/RU/cell can be supported (at least in the DL SIMO case). Reference in this regard may be made to 3GPP TSG RAN WG1 Meeting #47bis, Sorrento, Italy, Jan. 15-19, 2007, "ACK/NACK coverage in the absence of UL data", Nokia, R1-070393.
It has been determined in 3GPP that implicit ACK/NACK resources are configured by means of RRC signaling. However, it can be seen that a number of parameters are needed to characterize one implicit ACK/NACK resource. An implicit ACK/NACK resource here means an ACK/NACK resource that is allocated via implicit signaling, i.e., there is no explicit signaling that tells to the UE which ACK/NACK resource to use for acknowledging a DL transmission. Instead, the ACK/NACK resource is tied to the DL control channel index and is thus signaled implicitly (e.g., by the control channel index). First, static parameters are needed, such as the frequency allocation of the ACK/NACK resource, i.e., the RU for the 1st slot and for the 2nd slot (see FIG. 1). Also needed is the spreading factor of RS blocks (e.g., 3), with two alternatives in the FDD mode (depending on the CP length). Also needed is the spreading factor of data blocks (e.g., 4), with two alternatives in the FDD mode (depending on the CP length). Also needed is the length of the CAZAC sequence (12 frequency bins), with 18 bins being a possible candidate. The static parameters are typically defined in the standard specification.
Second, semi-static parameters are needed, such as the base sequence of the frequency domain CAZAC code (typically separately for the pilot and the data), the base sequence of the block level code (typically separately for the pilot and the data), the cyclic shift allocation for the frequency domain CAZAC code (typically separately for the pilot and the data), the cyclic shift allocation for block level code (typically separately for the pilot and the data), information concerning cyclic shift hopping (typically separately for the pilot and the data), as well as information concerning possible CAZAC sequence hopping (it is for future study whether this feature is supported). The semi-static parameters are typically signaled using higher layer signaling (e.g., RRC signaling) to each UE, or they may be broadcast to the entire cell via a broadcast channel.
It can be noted that most of these parameters are cell-specific, including all of the static parameters, the base sequence indices (both in the frequency and block domains) and the shift/sequence hopping related parameters.
The largest burden is related to the signaling of the cyclic shift resources, which are resource specific. Note that 12 bits are needed to characterize cyclic shifts of one implicit ACK/NACK resource, where six bits are needed to characterize the ACK/NACK RS resource (12*3 available code channels) and an additional six bits are needed to characterize the ACK/NACK data resource (12*4 code channels).
It has been assumed thus far that all of the cyclic shifts of all available implicit ACK/NACK resources are explicitly signaled by means of RRC signaling. This corresponds to a signaling burden with 12, 18 and 36 implicit ACK/NACK resources (there are thus 12, 18 or 36 ACK/NACKs possible to send per TTI) that is equal to 144 bits, 216 bits and 432, respectively. As can be appreciated, this amount of signaling load can be disadvantageous.

Signaling of implicit ACK/NACK resources


The present invention relates to anaerobically curable compositions based on (meth)acrylate-functionalized poly(acrylate) polymers.

1. Field of the Invention
The present invention relates to anaerobically curable compositions based on (meth)acrylate-functionalized poly(acrylate) polymers.
2. Brief Description of Related Technology
Anaerobic adhesive compositions generally are well-known. See e.g., R. D. Rich, "Anaerobic Adhesives" in Handbook of Adhesive Technology, 29, 467-79, A. Pizzi and K. L. Mittal, eds., Marcel Dekker, Inc., New York (1994), and references cited therein. Their uses are legion and new applications continue to be developed.
Conventional anaerobic adhesives ordinarily include a free-radically polymerizable acrylate ester monomer, together with a peroxy initiator and an inhibitor component. Oftentimes, such anaerobic adhesive compositions also contain accelerator components to increase the speed with which the composition cures.
For certain commercial applications, apart from that technology disclosed in U.S. Pat. No. 5,116,558 (Wrobel), conventional anaerobically curable compositions ordinarily do not develop the physical property profile to satisfy performance standards of customers. One such commercial application is a gasketing application.
Certain polyacrylate materials have been designed and developed by Kaneka Corporation for photocurable applications. The route by which these materials are synthesized involves the use of copper (I) salt catalysts, such as copper halides like copper chlorides, copper bromides and copper iodides, which despite extensive purification measures, carry over as impurities at significant concentrations in commercially available polyacrylate materials, such as those referred to by Kaneka under the trade designation XMAP and/or Telechelic polyacrylates. Copper (I) salt impurities such as these lead to poor stability in anaerobically curable compositions. These synthetic routes—commonly referred to as atom transfer radical polymerization or ATRP—are believed to be similar to those described for instance in U.S. Pat. Nos. 5,312,871 (Mardare) and 5,763,548 (Matyjaszewski). Certain polyacrylate materials have been designed and developed by Kaneka Corporation for photocurable applications. The route by which these materials are synthesized involves the use of complexed copper halide catalysts, which despite extensive purification measures, carry over as impurities at significant concentrations in commercially available polyacrylate materials, such as those referred to by Kaneka under the trade designation XMAP and/or Telechelic polyacrylates. Copper compound impurities such as these lead to poor stability in anaerobically curable compositions. These synthetic routes are believed to be similar to those described for instance in U.S. Pat. Nos. 5,312,871 (Mardare) and 5,763,548 (Matyjaszewski). See also H. Tang et al., "Highly Active Catalyst for AGET ATRP Tertiary Amine as Reducing Agent", Polym. Preprints, 47(1), 156 (2006), in which the issues attendent with residual copper catalyst from ATRP syntheses are highlighted.
To date, again, apart from that technology disclosed in the '558 patent, it is not believed that an anaerobically curable composition based on (meth)acrylate-functionalized poly(acrylate)s suitable for gasketing applications has been successfully offered for sale commercially or described in the literature. Cf. THREE BOND product no. 1133EC, which is believed to be a polyester-urethane-methacrylate-based composition. Accordingly, it would be desirable for anaerobically curable compositions to be suitable in terms of physical property profile so that they may be viable candidates to satisfy performance standards of customers for gasketing applications.


An frequency division multiplexing (OFDM) system transceiver for transmitting frequency dividing data in parallel includes antenna elements for receiving known reception and reception data signals. The fast Fourier transforms (FFTs) transform the known reception signals and the reception data signals to obtain known reception sub-carrier signals and reception data sub-carrier signals. The estimator estimates propagation path estimating values of each of the reception data sub-carrier signals with respect to each of the known reception sub-carrier signals. The weight calculator calculates a maximum ratio composition weight to composite the reception data sub-carrier signals. The setting means sets a transmission weight based on the maximum ratio composition weight. The generator generates a transmission data signal by arranging transmission sub-carrier signals on the frequency axis. The multipliers multiply the transmission data signals by the transmission weight and output the multiplied resultant to the respective antenna elements.

Conventionally, an OFDM modulation system is utilized for transmitting large capacity information in parallel. Such a system can transmit one type of data over a long time period, thereby reducing the effects of frequency selectivity phasing.
Further regarding an OFDM system, carrier intervals are shortened on a frequency axis, thereby improving frequency usability. Also, a copy signal of a symbol referred to as a guard interval is attached to the symbol. Therefore, it is possible to prevent symbols from interfering with one another when delayed radio waves are received.
Normally, the OFDM system rebuilds a sub-carrier signal, which includes errors introduced by frequency selectivity phasing, by error correction using other sub-carrier signal information. Accordingly, high precision error correction is required because transmitted data tends to delay spread. In such a case, however, several codes are required for error correction, thereby substantially decreasing signal transmission speed.
To avoid the above-mentioned decrease in signal transmission speed, conventional transceivers utilize an OFDM system with diversity reception. The diversity reception would enable the transceiver to select an appropriate signal from several signals received by several antennas. The transceiver can obtain a highest quality maximum-ratio-combining that maximizes the signal to noise ratio of respective sub-carrier signals.
Specifically, in the transceiver, sub-carrier information in the received signal can be rebuilt using identical sub-carrier signal received at the other antennas even if the received signal level of an antenna decreases. Accordingly, high quality information can be obtained without the need for high precision error correction.
An exemplary transceiver using an OFDM system with maximum ratio combining is described. Referring to FIGS. 7–9C, the transceiver has antenna elements 10–13, serial-parallel (S/P) converters 20–23, Fast Fourier Transformers (FFTs) 30–33, propagation path estimators 40–43, a weight calculator (maximum ratio combining weight calculators) 50, a maximum ratio combining processor 60, a demodulator 70 and a parallel-serial (P/S) converter 80.
Also, an example format of the OFDM signal is described. Referring to FIG. 8, a signal (timing signal) T that is for detecting signal timing and a signal (path signal) CS that is for estimating a propagation path are disposed before respective available symbols D1–DN (N is positive integer) on a frequency axis.
The timing signal T is a predetermined signal periodically transmitted in the time domain. The path signal CS has known sub-carrier signals arranged in a predetermined order in the time domain. The available symbols D1–DN include data symbols (Data (1)-Data (Nsym)) and guard intervals GI that are arranged before each of the data symbols. The data symbols D1–DN have frequency domain sub-carrier signals (data sub-carrier signals). The guard intervals GI are copies of data with respect to predetermined positions of respective data symbols D1–DN in which the guard intervals GI are provided. Incidentally, the available symbols are digital date modulated by a technique such as by BPSK, QRSK, 16 QAM or the like.
In FIG. 7, respective OFDM signals received by the antenna elements 10–13 are amplified and frequency converted by an RF/IF circuit (not shown). The resultant signals are input to a vector demodulator (not shown) and are demodulated to I and Q where I and Q correspond to real and imaginary numbers respectively. The I and Q are then processed by a synchronizing process, an AFC (Auto Frequency Control) process, a guard interval removing process or the like.
Further, respective resultant signals after the guard interval process are serial-parallel converted by the S/P converters 20–23. The parallel signals are then input into respective FFTs 30–33 which in turn generate demodulated OFDM signals.
In the OFDM demodulation, the FFTs 30–33 calculate respective known sub-carrier signals of the path signals CS and respective data sub-carrier signals of the available symbols because the OFDM signals include the path signal CS and available symbols D1–DN in the time domain.
Exemplary data sub-carrier signals calculated by the FFTs 30–33 are shown in FIGS. 9A–9C. In FIG. 9A, the horizontal axis shows frequency, and codes DS1–DS6 are respective data sub-carrier signals calculated by the FFTs 30–33. Phases and amplitudes of the data sub-carrier signals of the OFDM signals are different from each other even if the signals are simultaneously received. This is because the OFDM signals are deformed through the propagation path.
Accordingly, the data sub-carrier signals are compensated by the propagation path estimators 40–43 using, for example, the path signals DS1–DS6. The propagation path estimators 40–43 are provided for respective elements and replicate the known sub-carrier signals of the path signals CS. The propagation path estimators 40–43 complex-divide respective data sub-carrier signals, which are calculated by the FFTs 30–33, using the replicas of the known sub-carrier signals to thereby calculate propagation path estimating values of respective known sub-carrier signals that indicate frequency features of propagation path. In other words, the propagation path estimators 40–43 calculate the respective propagation path estimating values by dividing the received path signals CS by the respective replicas of the known sub-carrier signals.
The weight calculator 50 calculates the maximum ratio combining weight W using the propagation path estimating a value of respective sub-carrier signals. The weight W is shown in mathematical expression (1) below as a matrix of (number of the antenna elements)×(number of the data sub-carrier signals), where the propagation path estimating value is hi(l, k) whose antenna element number is "i", available symbol number is "1" and data sub-carrier signal number is "k".
                    W        =                  (                                                                                h                      1                                                                                (                                              f                        ,                        1                                            )                                                                                                                                                        h                          1                                                                                (                                                      f                            ,                            1                                                    )                                                                                                            2                                                                                h                      1                                                                                (                                              f                        ,                        2                                            )                                                                                                                                                        h                          1                                                                                (                                                      f                            ,                            2                                                    )                                                                                                            2                                                                              ⋯                                                                                h                      1                                                                                (                                              f                        ,                        k                                            )                                                                                                                                                        h                          1                                                                                (                                                      f                            ,                            k                                                    )                                                                                                            2                                                                                h                      2                                                                                (                                              f                        ,                        1                                            )                                                                                                                                                        h                          2                                                                                (                                                      f                            ,                            1                                                    )                                                                                                            2                                                                                h                      2                                                                                (                                              f                        ,                        2                                            )                                                                                                                                                        h                          2                                                                                (                                                      f                            ,                            2                                                    )                                                                                                            2                                                                              ⋯                                                                                h                      2                                                                                (                                              f                        ,                        k                                            )                                                                                                                                                        h                          2                                                                                (                                                      f                            ,                            k                                                    )                                                                                                            2                                                                                ⋮                                            ⋮                                                                                ⋮                                                                                h                      i                                                                                (                                              f                        ,                        1                                            )                                                                                                                                                        h                          i                                                                                (                                                      f                            ,                            1                                                    )                                                                                                            2                                                                                h                      i                                                                                (                                              f                        ,                        2                                            )                                                                                                                                                        h                          i                                                                                (                                                      f                            ,                            2                                                    )                                                                                                            2                                                                              ⋯                                                                                h                      i                                                                                (                                              f                        ,                        k                                            )                                                                                                                                                        h                          i                                                                                (                                                      f                            ,                            k                                                    )                                                                                                            2                                                                                )                                    (        1        )
As shown in mathematical expression (1), respective elements of the weight W correspond to the respective data sub-carrier signals and are divided by an amount of the respective propagation path estimating values (square of the respective propagation path estimating values). The maximum ratio combining processor 60 maximum-ratio-combines the data sub-carrier signals using the weight W.
The resultant signal Z(l,k) generated by maximum-ratio-combining is shown in mathematical expression (2), where the sub-carrier signal is Xi(l,k) whose antenna element number is "i", available symbol number is "1" and data sub-carrier signal number is "k". Also, a total antenna element number is M and * means complex conjugate.
                              z                                (                          ∫                              ,                k                                      )                          =                                            ∑                              i                =                1                            M                                                  hi              *                              (                                  f                  ,                  k                                )                                                          xi                                                  (                                      f                    ,                    k                                    )                                                                                ∑                              i                =                1                            M                                                                                                    hi                                                        (                                          f                      ,                      k                                        )                                                                              2                                                          (        2        )
In the maximum-ratio-combining, the change of the identical data sub-carrier signals of the respective antenna elements 10–13 is compensated for using the propagation path estimating values. Thus, the maximum ratio combining processor 60 outputs combining signals z(l,k) with respect to each data sub-carrier signal.
The demodulator 70 digitally demodulates the combining signals z(l,k) using BPSK, QPSK, 16QAM or the like. Then, the P/S converter 80 converts the parallel demodulated signals z(l,k) to obtain serial demodulated data.
As mentioned above, the maximum ratio combining increases the signal levels of the received signals and decreases corresponding noise levels thereof. Namely, regarding the antennas 10–13, increasing the received signal level directs the main beam for reception toward a high signal energy direction. Also, decreasing the noise signal level decreases the side lobe levels. Accordingly, it is possible to direct the main beam for reception toward the target direction.
The above-mentioned receiver can direct the main beam for reception using the maximum ratio combining weight. That is, the reception beams of the antennas 10–13 are formed using the maximum ratio combining weight.
However, a transceiver that can direct a main beam for transmission in the same direction as that of the transmitter is not suggested.

OFDM transmission system transceiver and method