A power supply circuit is provided for generating a low voltage for powering a control circuit. The power supply circuit is connected to a voltage source having a high voltage relative to the output voltage of the power supply circuit. The power supply circuit includes a power supply capacitor and a charging capacitor connected to the power supply capacitor which charges the power supply capacitor when the power supply circuit is connected to the high voltage source. The power supply also includes an inductor connected to the power supply capacitor which charges the power supply capacitor by resonant action when the power supply circuit is disconnected from the voltage source, so that the power supply capacitor is continuously charged. A pair of diodes are also provided. The first diode is connected between the charging capacitor and the power supply capacitor. The second diode is connected in series with the inductor. The power supply circuit further includes a zener diode connected to the power supply capacitor for regulating the voltage across the power supply capacitor.

In general, the "power factor" of an electrical load refers to the ratio of the active power provided to the load to the apparent power applied to the load. The power factor is closely tied to the phase relationship between the electrical current drawn by the load and the electrical voltage applied to the load from a source of electrical power. If the drawn current is sinusoidal and completely in-phase with an applied sinusoidal voltage, then a unity power factor (i.e., a power factor of 1) is obtained.
High power factors are desirable for various reasons, including energy efficiency. The higher the power factor of a load, the lower the current demanded of the power supply utility for a given output power. Further, electronic loads with rectifier inputs and capacitive filtering frequently draw severely non-sinusoidal currents which results in poor power factor and can lead to distortion of the supply voltage. The higher the power factor, the lower the current that will be drawn from the supply for a given output power and the less the load will tend to distort the voltage waveform provided by the source of electrical power. To minimise supply currents and to avoid significant distortion of the voltage waveforms provided by power utilities, certain countries have promulgated regulations requiring electrical devices above a certain power rating to have a minimum power factor and limiting the harmonic content of the supply current.
In practice, electrical circuits often do not have unity power factors. In certain applications, such as motor control circuits that utilize a converter or inverter operating from a DC bus, the power factor can vary significantly from unity. Such circuits typically use a full wave bridge rectifier in combination with a relatively large DC bus capacitor to convert sinusoidal alternating input voltage into substantially constant direct voltage. In such arrangements, the rectifier output current should ideally follow the rectifier output voltage exactly. This is generally illustrated in FIG. 1 which illustrates an ideal load in which the current drawn by the circuit (indicated by the dashed line) is substantially in-phase with the applied voltage (indicated by the solid line).
Power factor correction (PFC) circuits are often used to improve the power factor of a load by modifying the current drawn by the rectifier such that it approaches that shown in FIG. 1. There are several commercially available PFC circuits. For example, Unitrode offers a family of high power-factor pre-regulator controllers under the model numbers UC1852, UC2852 and UC3852 and Linear Technology offers a power factor controller model number LT1248. These PFC devices are normally fabricated in analog integrated circuit chips and operate as analog circuits. In general, these devices use pulse width modulated switching to improve the power factors of circuits, such as convertors for switched reluctance motors, that utilize DC bus voltages derived from an alternating input.
FIG. 2 illustrates the use of an exemplary analog PFC chip 14. The circuit receives applied sinusoidal alternating voltage at the inputs of a full wave rectifier 4 and produces a full wave rectified sinusoidal voltage. The full wave rectified sinusoidal voltage is applied to one terminal of a filtering inductor 8. Coupled to the other terminal of the inductor 8 is a switching device 10, such as a power MOSFET or an IGBT. The other terminal of the switching device 10 is coupled to the negative rail of the DC bus. A PFC chip 14 provides switching signals to switch the switching device 10 on and off. The PFC chip receives as inputs (a) a measure of the full-wave rectified sinusoidal voltage at the output of the bridge rectifier 4 (V.sub.SIN), (b) a measure of the voltage across the DC bus (V.sub.RET) and (c) a measure or estimate of the current flowing in inductor 8, commonly obtained using a resistive shunt to measure current in the switching device 10. A DC bus capacitor 6 is coupled across the positive and negative rails of the DC bus. A diode 12 is provided to prevent current from flowing back from the load when the switch 10 is closed. In operation, the PFC chip opens and closes switching device 10 so that the load across the full wave rectifier 4 varies from the inductor 8 (when switch 10 is closed) to the inductor and the DC bus capacitor 6 (when switch 10 is opened). By properly switching switch 10 the current in inductor 8 is made to follow a waveform corresponding to the voltage variations at the output of full wave rectifier 4. The power factor of the system is thus improved. The operation of PFC chips is generally understood and is not addressed herein in detail.
The magnitude of the alternating voltage applied to the input of the full wave rectifier 4 is typically fairly high (as much as 240 volts r.m.s.), as is the magnitude of the DC bus voltage that appears across capacitor 6 (as much as 400V d.c.). Most PFC control circuits, however, additionally require a DC supply voltage that is substantially less than the DC voltage that appears across the DC bus typically 10 to 20 volts. Accordingly, some mechanism must be provided for providing a relatively low DC supply voltage 14 to the PFC chip. Moreover, the low DC supply voltage provided to the PFC controller must be referenced to the negative rail of the DC link.
Many approaches have been used to supply the relatively low DC voltage to the PFC control circuit. FIG. 3 illustrates one such approach. In FIG. 3 the low DC supply voltage for the PFC controller is provided by a capacitor 20 that is charged through a high power bleed resistor 16. A zener diode 18 controls the voltage across capacitor 20. One significant disadvantage of DC voltage supplies that employ bleed resistors is that a large voltage is dropped across the bleed resistor, whilst the current flowing through the bleed resistor must be at least equal to the current required by the PFC controller. This current produces considerable heat and represents a source of lost energy. This lost energy introduces inefficiencies into the system. A further disadvantage of DC voltage supplies using bleed resistors is that the bleed resistor must have characteristics capable of handling not only significant power dissipation but also the high voltage output of the full wave rectifier 4. Such resistors are often physically large, relatively expensive, and tend to increase the physical size and cost of systems using them.
Another common DC voltage supply for a PFC control circuit is illustrated in FIG. 4. In this circuit a second inductor 26 is inductively coupled to the filter inductor 8. The second inductor 26 is coupled to a storage capacitor 24 via a diode 30. During operation, PFC controller 14 will switch switching device 10 on and off at a fairly high frequency. The high frequency voltage components appearing across the inductor 8 as a result of this switching induce a current flow and voltage in inductor 26 through transformer action. This current flows through diode 30 and charges capacitor 24 to the desired DC voltage. A bleed resistor 22 is provided to charge capacitor 24 when the circuit is first energized, before the transformer action is sufficient to charge capacitor 24. As before, a zener diode 18 serves to regulate the voltage applied to the control circuit.
While a bleed resistor is still required, it can be of a much higher resistance than that of FIG. 3. The PFC control circuit will generally draw only a very small supply current until the capacitor 24 has charged to sufficient voltage to allow satisfactory operation of the PFC circuit. This allows resistor 22 to be of high resistance and hence the current flowing through it, and the power dissipated by it, will be relatively small. When the voltage across capacitor 24 has reached the required threshold, the PFC controller 14 will come into operation, drawing significant supply current. This current is initially supplied by capacitor 24 but is quickly supplied by the transformer action between inductors 8 and 26.
While the transformer-action DC voltage supply of FIG. 4 reduces the need for a large bleed resistor, it does not eliminate the need for such a resistor. Accordingly, the DC voltage supply of FIG. 4 suffers from the same disadvantages associated with the bleed resistor supply of FIG. 3, although not to the same extent. The additional winding 26 significantly increases the size and cost of the inductor 8 with which it must be combined. Moreover, the transformer action circuit of FIG. 4 is limited in that it is often insufficient to maintain an adequate voltage across capacitor 24. In these circumstances, it is necessary to reduce the value of resistance of the bleed resistor 22 to supplement the current from the inductor 26, bringing back the disadvantages already discussed.
In an effort to overcome the disadvantages associated with the circuits of FIGS. 3 and 4, approaches using more complex diode, capacitor and inductor arrangements have been proposed. FIG. 5 illustrates one such approach.
In FIG. 5 a circuit is provided in which capacitors 34 and 46 are arranged with diodes 32 and 36 to form a charge pump circuit. The charge pump circuit charges capacitor 46 which provides a DC voltage to the PFC chip 14.
As with the circuit of FIG. 4, the circuit of FIG. 5 includes a small bleed resistor 44 that is used to charge the capacitor 46 when the circuit is first energized. Zener diode 40 again serves to regulate the supply voltage to the PFC circuit. Once the PFC chip 14 is operational, the charge on capacitor 46 is maintained through the switching action of the PFC chip as follows. At an initial point in time, switching device 10 will be closed and the voltage at node A will be approximately the value of the negative rail of the DC bus (e.g, 0 volts). At this point, the voltage at node B will also be approximately 0 volts. At a later point in time, the operation of the PFC chip will have necessitated opening of switch 10. This will cause the voltage at node A to rise to approximately the value of the positive rail of the DC bus (e.g., 380 volts). The voltage across capacitor 34 cannot instantaneously change, and a charging current will flow through it as the voltage at node A rises. This charging current will bias diode 36 ON, and current will flow through capacitor 34, through diode 36 into capacitor 46, charging capacitor 46 to the desired low DC voltage level. Eventually, node A will reach the DC bus voltage approximately (e.g., 380 V), and the charging current flowing through capacitor 34 will cease. At this time, node B will be at approximately the same potential as the low voltage supply to the PFC chip and determined by zener diode 40. This might typically be 20 V, so that capacitor 34 is charged to a voltage of e.g., 360 V. In summary, at the time the switching device 10 is opened, capacitor 34 acts as a current source providing charge to the power supply capacitor 46.
At a later point in time, the PFC chip will close switching device 10 and the voltage at node A will again drop to approximately 0 volts. This switching event will also effectively place capacitor 34 in parallel with diode 32. This will discharge capacitor 34 through switching device 10 and diode 32. As the above indicates, in the circuit of FIG. 5 capacitor 34 approximates to a current source where the current is that charging capacitor 46 immediately after switching device 10 is opened, and where the capacitor discharges through switching device 10 immediately after switching device 10 is closed.
The DC voltage supply of FIG. 5 is limited in several respects. Initially it must be noted that all of the charge used to charge the power supply capacitor 46 is provided from capacitor 34. Accordingly, to prevent undesirable variations in the low voltage power supply, capacitor 34 must be sized sufficiently large so that power supply capacitor 46 is always adequately charged. This requirement of a sufficiently large capacitor 34 is disadvantageous in at least two respects. First, the capacitor 34 must be capable of withstanding high voltages. The larger the capacitor 34, the greater the cost and size of the capacitor and, hence, the larger the cost of the system. Second, as discussed above, while the charging current of capacitor 34 is used to charge the power supply capacitor 46 when the switching device is open, the charge across the capacitor 34 is discharged through switching device 10 when the switching device 10 is closed. Accordingly, switching device 10 must be sized to handle not only a portion of the current flowing through the full wave bridge 4, but must also dissipate all the energy stored in capacitor 34 at each instant of switch closure. This generally increases both the cost and the size of switching device 10. Further, the additional power dissipation in switching device 10 due to capacitor 34 results in a loss of power and increases the system inefficiencies. A further disadvantage of the power supply of FIG. 5 is that the power supply capacitor 46 is charged only when the switching device 10 is switched off. A still further disadvantage of the power supply of FIG. 5 is that, while it may be sufficient to drive the PFC controller 14, it typically cannot be used to drive other circuits, e.g., a motor control circuit or a system control circuit. Such a multi-purpose power supply would require significantly more output current than the circuit of FIG. 5. While one could theoretically increase the size of capacitors 34 and 46, such increases would necessitate a corresponding increase in the power handling capability of switching device 10, increase the cost of the system, and decrease the efficiency of the system.
As the foregoing demonstrates, known power supplies for PFC controllers generally result in the use of relatively expensive components or produce significant inefficiencies in the system. Moreover, such power supplies typically can power only PFC controllers and cannot power other circuits. It is an object of the present invention to overcome these and other problems in the prior art.
The present invention is defined in the accompanying independent claim. Some preferred features are recited in the dependent claims.

Power supply circuit for a control circuit


The present invention relates generally to methods of using anti-NGF antibodies in the treatment of various NGF-related disorders, including asthma, arthritis and psoriasis. The methods are effective in treating these disorders in a patient without having a significant adverse effect on the immune system of the patient.

1. Field of the Invention
The present invention relates generally to methods of using anti-NGF antibodies in the treatment of various NGF-related disorders, including asthma, arthritis and psoriasis. The methods are effective in treating these disorders in a patient without having a significant adverse effect on the immune system of the patient.
2. Description of the Related Art
Nerve Growth Factor (NGF)
Nerve growth factor (NGF) was the first neurotrophin to be identified, and its role in the development and survival of both peripheral and central neurons has been well characterized. NGF has been shown to be a critical survival and maintenance factor in the development of peripheral sympathetic and embryonic sensory neurons and of basal forebrain cholinergic neurons (Smeyne et al., Nature 368:246-249 (1994); Crowley et al., Cell 76:1001-1011 (1994)). NGF upregulates expression of neuropeptides in sensory neurons (Lindsay and Harmer, Nature 337:362-364 (1989)) and its activity is mediated through two different membrane-bound receptors. The TrkA tyrosine kinase receptor mediates high affinity binding and the p75 receptor, which is structurally related to other members of the tumor necrosis factor receptor family, mediates low affinity binding (Chao et al., Science 232:518-521 (1986)).
In addition to its effects in the nervous system, NGF has been increasingly implicated in processes outside of the nervous system. For example, NGF has been shown to enhance vascular permeability (Otten et al., Eur. J. Pharmacol. 106:199-201 (1984)), enhance T- and B-cell immune responses (Otten et al., Proc. Natl. Acad. Sci. U.S.A. 86:10059-10063 (1989)), induce lymphocyte differentiation and mast cell proliferation and cause the release of soluble biological signals from mast cells (Matsuda et al., Proc. Natl. Acad. Sci. U.S.A. 85:6508-6512 (1988); Pearce et al., J. Physiol. 372:379-393 (1986); Bischoff et al., Blood 79:2662-2669 (1992); Horigome et al., J. Biol. Chem. 268:14881-14887 (1993)).
NGF is produced by a number of cell types including mast cells (Leon et al., Proc. Natl. Acad. Sci. U.S.A. 91:3739-3743 (1994)), B-lymphocytes (Torcia et al., Cell 85:345-356 (1996), keratinocytes (Di Marco et al., J. Biol. Chem. 268:22838-22846)) and smooth muscle cells (Ueyama et al., J. Hypertens. 11:1061-1065 (1993)). NGF receptors have been found on a variety of cell types outside of the nervous system. For example, TrkA has been found on human monocytes, T- and B-lymphocytes and mast cells.
Consistent with a non-neuronal role for NGF, an association between increased NGF levels and a variety of inflammatory conditions has been observed in human patients as well as in several animal models. These include systemic lupus erythematosus (Bracci-Laudiero et al., Neuroreport 4:563-565 (1993)), multiple sclerosis (Bracci-Laudiero et al., Neurosci. Lett. 147:9-12 (1992)), psoriasis (Raychaudhuri et al., Acta Derm. Venereol. 78:84-86 (1998)), arthritis (Falcini et al., Ann. Rheum. Dis. 55:745-748 (1996)) and asthma (Braun et al., Eur. J. Immunol. 28:3240-3251 (1998)). Chronic inflammatory conditions such as these are a significant public health problem. For instance, it is estimated that arthritis affects 37.9 million people in the United States alone. Current therapies for treating these conditions are severely limited. An understanding of the role NGF plays in these diseases may provide new methods for treating them.
A correlation between stress and psoriasis has been observed. Based on this correlation and the symmetry of the cutaneous lesions that accompany the disease, a relationship with the nervous system has been proposed (Raychaudhuri et al., Acta Derm. Venercol. 78:84-86 (1998)). In particular, neuropeptides have been suggested to play a role in the pathogenesis of psoriasis. Investigators have reported an increased number of terminal cutaneous nerves along with upregulation of one or more of the neuropeptides, such as substance P (SP), vasoactive intestinal polypeptide (VIP) and CGRP. NGF plays a role in regulating innervation in the skin and also is known to upregulate neuropeptides, suggesting that increased NGF levels may be responsible for the upregulation of neuropeptides and the increased cutaneous innervation seen with psoriasis. In fact, increased expression of NGF has been observed in psoriatic keratinocytes (Raychaudhuri et al., Acta Derm. Venercol. 78:84-86 (1998)). It has been suggested that while NGF normally serves as a survival factor for keratinocytes, overexpression of NGF prevents normal cell death, leading to psoriasis (Pincelli at al., J. Derm. Sci. 22:71-79 (2000)).
A number of studies have indicated that neuropeptides such as substance P (SP) and biologically active compounds released from mast cells, such as histamine, also play a role in both naturally occurring arthritis in humans and experimentally induced arthritis in animal models (see e.g. Levine, J., Science 226:547-549 (1984)). NGF has been shown to affect mast cell degranulation (Bruni at al., FEBS Lett. 138:190-193 (1982)) and substance P release (Donnerer et al., Neurosci. 49:693-698 (1992)), implicating it in the pathogenesis of arthritis.
Consistently, an elevated level of NGF in peripheral tissues is associated with both hyperalgesia and inflammation and has been observed in a number of forms of arthritis. The synovium of patients affected by rheumatoid arthritis expresses high levels of NGF while in non-inflamed synovium NGF has been reported to be undetectable (Aloe et al., Arch. Rheum. 35:351-355 (1992)). Similar results were seen in rats with experimentally induced rheumatoid arthritis (Aloe et al., Clin. Exp. Rheumatol. 10:203-204 (1992)). Elevated levels of NGF have been reported in transgenic arthritic mice along with an increase in the number of mast cells. (Aloe et al., Int. J. Tissue Reactions-Exp. Clin. Aspects 15:139-143 (1993)). However, purified NGF injected into the joint synovium of normal rats does not induce knee joint inflammation, suggesting that NGF does not play a causative role in arthritis (Aloe et al., Growth Factors 9:149-155 (1993)).
High NGF levels have been associated with allergic inflammation and it has been suggested that this is related to mast cell degranulation (Bonini et al., Proc. Natl. Acad. Sci. U.S.A. 93:10955-10960 (1996)).
Elevated NGF levels are also observed in both allergic and non-allergic asthma (Bonini et al., supra). Mast cells, eosinophils and T-lymphocytes have all been proposed to play a role in this inflammatory disease and the correlation between NGF serum levels and total IgE antibody titers suggests that NGF contributes to the inflammatory immune response. Allergen induced airway inflammation has been associated with increased local production of NGF in both mice and humans (Braun et al., Int. Arch. Allergy Immunol. 118:163-165 (1999)).
NGF has been shown to regulate the development of increased airway hyperactive response, a hallmark of bronchial asthma (Braun et al., Eur. J. Immunol. 28:3240-3251 (1998)). Indeed, in one study, treatment of allergen-sensitized mice with anti-NGF antibody prevented the development of airway hyperresponsiveness following local allergen challenge (Braun et al., Int. Arch. Allergy Immunol. 118:163-165 (1999)).
Despite the promising results obtained in mice, reported adverse effects of neutralizing anti-NGF antibodies on the immune system have raised serious questions about the feasibility of using anti-NGF antibodies as a therapeutic in the prevention or treatment of asthma or other diseases or disorders in human patients. In particular, Torcia at al., Cell 85:345-356 (1996) identified NGF as an autocrine survival factor for memory B lymphocytes, and demonstrated that in vivo administration of neutralizing anti-NGF antibodies caused a depletion of memory B-cells and abolished secondary antigen-specific immune responses in mice. Garaci et al., Proc. Natl. Acad. Sci. USA 96:14013-14018 (1999) reported that NGF is an autocrine survival factor that rescues human monocytes/macrophages from the cytopathic effect caused by HIV infection. This report, along with the findings of Torcia et al., supra would suggest that anti-NGF antibodies have the potential of compromising the immune system of the subject treated.

Anti-NGF antibodies for the therapeutic treatments


A system and method for controlling a medical device is provided such as a medical imaging device. The preferred embodiment of the present invention receives a verbal command for assigning a function of a medical imaging device from an operator. An additional verbal command from the operator assigns an input device to control the function of the selected medical imaging device. A system control and speech recognition processor then assigns the function specified by the verbal command to the input device selected by the additional verbal command. After the function of the medical device is assigned to the input device, the operator may control the selected function of the medical device with the selected input device. The system and method provides for the unobtrusive and hands-free control of discreet and continuous functions of a medical imaging device.

The present invention generally relates to a system and method for controlling a medical device. More particularly, the present invention relates to a system and method for controlling a medical imaging system using a speech recognition and foot-controlled system.
Medical imaging systems are being used for a growing number of applications in the field of medicine. Medical imaging systems are typically used for diagnosis as well as for monitoring purposes during surgery. One type of medical imaging system commonly used during surgery is an ultrasound imaging system. Typical ultrasound systems operate by transmitting ultrasonic sound waves into a patient's body using a transducer. The transducer is typically a device placed on the patient's body over the area to be imaged that is capable of sending and receiving ultrasonic sound waves. The ultrasonic sound waves sent by the transducer are reflected by the patient's internal bodily structures. The reflected ultrasonic sound waves transmitted into the patient's body are then received by the transducer and processed to display a visual representation of the patient's internal bodily structures to the surgeon.
The received ultrasonic sound waves are typically processed by an ultrasonic imaging system and displayed in real time on a console for viewing by the examining physician. Typically, the console includes a control console as well as a viewing screen on which ultrasonic images are displayed. The control console typically includes a number of control devices. The control devices are typically manual controls such as dials, switches, knobs, or joysticks, for example, that may be used to manipulate the ultrasonic image displayed on the viewing screen. For example, the control devices may be used to control the resolution, magnification, viewing area, or orientation of the ultrasonic image. In typical ultrasound imaging systems, the ultrasonic images may only be adjusted or manipulated solely by the control devices.
Typically, the control console itself may be fairly large and may take up a substantial amount of space in an operating room. However, during surgery, the space around the surgeon may often be very limited. For example, multiple trays containing operating instruments, as well as other medical devices and support personnel may need to be near the surgeon to assist the surgeon during surgery. Therefore, there typically is not free space near the surgeon for the ultrasound imaging system and console during surgery. Consequently, the ultrasound imaging system and console may out of necessity need to be positioned up to several feet away from the surgeon and out of the immediate reach of the surgeon.
Typically when the console is positioned out of reach of the surgeon, the surgeon may be unable to use the control devices to manipulate the ultrasonic images during surgery. Because the surgeon is unable to manipulate the ultrasonic images during surgery, the surgery may become complicated by limiting the adjustibility of the ultrasonic image. If the surgeon must adjust the ultrasonic image, the surgeon may have to continually physically walk over to the console and then adjust the properties of the image using the control devices. Requiring the surgeon to continually walk over to the control devices to adjust the ultrasonic image may be time consuming, inefficient, or impractical if the surgery requires that the surgeon continually monitor the patient. Thus, typical ultrasonic imaging systems operable solely by control devices may be undesirable or inefficient for use by surgeons in situations where the available space around the surgeon is limited.
Even if the ultrasound imaging system and console are able to be positioned within reach of the surgeon, a number of drawbacks still exist in typical systems operable solely by console controls. For example, the surgeon's hands may be in use or too busy during the surgery to manually operate the control devices. Thus, while the surgeon may be able to reach the control devices, in practicality, the surgeon is still unable to use them. Even if the surgeon is able to operate the control devices during surgery, a number of drawbacks exist. For example, during surgery, the surgeon typically wears latex surgical gloves. The surgeon's gloves may often become covered with blood or other materials during surgery. Handling the control devices with soiled gloves may contaminate the control devices with unsterile materials, which may not present optimal sterile conditions. Thus, the control devices may have to be sterilized before, during, and after each surgery since the surgeon typically touches the control devices frequently during operation. Therefore, the control devices typically must to be made of materials that may be handled in a sterile environment.
Additionally, even sterilized control devices may be difficult to operate by a surgeon. After, the surgeon's gloves become covered with blood or other materials during surgery, the gloves may become slippery. Therefore, operation of the control devices by hand may become difficult after the surgeon's gloves become soiled. Thus, adjusting the control devices by hand in a sterile surgical environment may not be the most practical and efficient method of adjusting an ultrasonic image.
One method used to reduce some of the drawbacks associated with the limited amount of free space around a surgeon during surgery is the use of a remote control system. A remote control system is typically a compact hand-held unit including controls such as switches, dials, or joysticks, for example. Remote controls may be operable by either one or two hands. The remote control system may communicate with the medical imaging system by either a wireless transmission system or by a wire-based transmission system. The compact remote control system may be used to operate the functionality typically operated by the control devices and typically does not require that the control devices be within reach of the surgeon. Thus, the compact size of the remote control takes up less free space than the console controls.
However, remote control systems may also suffer from some significant drawbacks. For example, while the space required for the remote control system is typically less than the space required for the entire medical imaging system or console controls, some space is still required. Thus, in situations where there is very little or no free space around the surgeon, even remote control systems may still not be a viable alternative. Furthermore, remote controls systems still may present the same drawbacks that console controls exhibit with regard to sterilization and ease of use as discussed above. Additionally, the controls on the remote control system may actually be smaller than the control devices on the console to save space and thus the remote control system may be more difficult for a surgeon to manipulate, especially with soiled gloves.
One method used to reduce some of the drawbacks associated with medical imaging systems operable by control devices or remote control systems is the use of a foot-controlled console. Foot-controlled consoles are typically comprised of at least one foot-input device such as pedals, switches, or joystick-type devices for example, that may be operated by the surgeon's foot. In the medical imaging system field, such foot-controlled consoles may be used to control the functions typically controlled by the control devices of the medical imaging system.
Typically, the foot-controlled console may be placed on the floor near the surgeon's feet where operation room space is typically available. Thus, the surgeon may still control the functionality of the control console of the medical imaging system, via the foot-controlled console, even though the control console may be positioned out of reach of the surgeon. That is, although the foot-controlled console may not occupy any space in the limited operating space directly around the surgeon's hands and torso, the foot-controlled console allows for hands-free operation of the functionality of the medical imaging system's control devices by the surgeon during surgery. Hands-free operation may reduce the problems associated with manually operating control devices or remote controls as discussed above.
However, the typical foot-controlled console system may still suffer from some significant drawbacks. One drawback that may occur in typical foot-controlled consoles is reduced functionality. That is, the foot-controlled console may lack much of the functionality that the control console has. The reduced functionality of foot-controlled consoles may occur for a number of reasons. One reason a foot-controlled console may lack some of the functionality that the control console has is that there is limited space available on a foot-controlled console. Typical control consoles may include a large number of control devices for a wide variety of features of the medical imaging system. Therefore, in order to accommodate control of each feature of the medical imaging system, a large number of foot-input devices may need to be placed on the foot-controlled console. Because the surgeon's feet are typically larger and less agile than the surgeon's hands, the foot-input devices on the foot-controlled console typically may be larger and spaced further apart than the corresponding control devices on a typical control console. Thus, if the number of control devices on the medical imaging system console is high, the corresponding foot-controlled console may become too large and inefficient to use in practice. Therefore, in order to keep the foot-controlled console compact enough for efficient use, a limited number of foot-input devices, typically less than the number of control devices, may have to be placed on the foot-controlled console limiting the functionality of the foot-controlled console.
In applications outside of the medical imaging field, control and functionality of various systems have been provided by voice-controlled systems. For example, the use of voice-controlled systems has been adopted in fields such as computer science to facilitate hands-free operation of personal computers. Typical voice-controlled systems utilize a microphone and a speech recognition system. Typically, an operator speaks a verbal command into the microphone, and the command is then transmitted to the speech recognition system. The speech recognition system is typically pre-programmed to recognize the command. After recognizing the verbal command, the speech recognition system typically sends a signal to the device being controlled to perform the operator's command. Thus, speech recognition systems allow an operator to control a device in a completely hands-free manner. However, voice-controlled systems may suffer from some significant drawbacks. These drawbacks may make the use of typical voice-controlled systems in the medical imaging field very difficult.
For example, one drawback that may be present in speech recognition systems is the inability to make fine adjustments to continuous controls such as a joystick, trackball, or dial, for example. That is, verbal commands typically are not able to provide small continuous movements of controls, which may often be required of medical imaging systems. Therefore, the use of speech recognition systems may not be optimal for surgical applications because of their inability to perform fine adjustment. Another drawback is that typical speech recognition systems may react slowly to commands. Therefore, in a real-time environment such as during surgery, a significant lag time between a command and execution of the command may complicate the surgery or hinder progress. Furthermore, speech recognition systems are typically sensitive to external noise. That is, speech recognition systems typically need to be operated in a quiet environment to function properly. Noises other than the operator's voice, such as other voices or noise from other devices in surgery for example, may cause the speech recognition system to register false commands.
Thus, a need exists for a medical imaging control system that may be efficiently used in the limited operating area of a surgeon. A need further exists for a medical imaging control system that may be efficiently used in the limited operating area of a surgeon while maintaining all of the functionality available to console controls. Additionally, a need exists for a medical imaging control system that allows for the efficient control of continuous controls such as joysticks, trackballs, or dials, for example, during surgery.

Remote control of a medical device using speech recognition and foot controls

GE Medical Systems Global Technology Company, LLC


The invention relates to a photographic element comprising a paper base, at least one photosensitive silver halide layer, a layer of biaxially oriented polyolefin sheet between said paper base and said silver halide layer, wherein there is located between said biaxially oriented polyolefin sheet and said base paper a layer of adhesive resin comprising an ethylene polymer blended with at least one member selected from the group consisting of metallocene catalyzed polyolefin plastomer, ionomer, ethylene vinyl acetate copolymer, ethylene methyl methacrylate copolymer, ethylene ethyl acrylate copolymer, ethylene methyl acrylate copolymer, ethylene acrylic acid copolymer, ethylene ethyl acrylate maleic anhydride copolymer, or ethylene methacrylic acid copolymer.

It has been proposed in U.S. Pat. No. 5,244,861 to utilize biaxially oriented polypropylene laminated to cellulose grade paper for use as a reflective receiver for thermal dye transfer imaging process. The preferred bonding agent in U.S. Pat. No. 5,244,861, to bond the biaxially oriented polypropylene sheets to paper, is low density polyethylene melt extruded from a slit die. Because of the high processing temperatures required for LDPE, shrinkage of the biaxially oriented sheet is common in the melt extrusion process. Shrinkage can cause undesirable changes in the Poisson ratio of the laminated receiver as well as a reduction in the optical performance of the receiver. It would be desirable to reduce the extrusion temperature of the bonding layer and an maintain acceptable integrity of the laminated support.
In the formation of color photographic paper a waterproof resin is melt extrusion coated on photographic grade cellulose paper. Polyethylene is commonly used and may be any one of a high density polyethylene, a low density polyethylene, a linear low density polyethylene or a mixture of two or more of the above mentioned polyethylenes.
In U.S. Pat. No. 4,339,507 (Kurtz et al.), linear low density ethylene hydrocarbon copolymers for extrusion coating are proposed to improve adhesion. While linear low density polyethylene does improve adhesion when compared to LDPE, LLDPE suffers from a phenomena referred to as draw resonance. Draw resonance is defined as the instability of the melt flow and is manifested as thickness variation both in the cross and machine directions. Draw resonance is a function of machine speed, becoming worse at higher line speeds. As a result, the maximum line speed at which an acceptable photographic paper can be manufactured is restricted by the onset of draw resonance.
During the manufacturing process for photographic papers, while the laminated photographic support is being emulsion coated and slit, the laminated structure is subjected to various forces in manufacturing that will cause delamination of the polypropylene sheet from the paper. The delamination may be a result of bonding layer failure to either the base paper or the polypropylene sheet. Also, when the photographic paper is being processed and finished at photofinishers, the laminated structure is also subjected to various forces in both the wet and dry state. Furthermore, when the photographic paper is kept for years by the final customer, the laminated structure is subjected to forces created by temperature and humidity changes that could cause delamination of the biaxially oriented polyolefin sheets from the cellulose paper base. Delamination of the biaxially oriented sheet from the paper during manufacturing will result in the product being wasted thus increasing the cost of manufacture. Delamination of the biaxially oriented sheet from the paper at either the photo finishing operation or in the final customer format will result in a loss in the appearance of the image and the reduction of the commercial value of the photograph. It would be desirable if a melt extruded bonding adhesive could prevent delamination of biaxially oriented sheets from the base paper during manufacture of a laminated imaging support and in the final customer format.

Adhesives such as metallocene catalyzed ethylene plastomers for bonding biaxially oriented polyolefin sheets to paper


A worker management device comprises time clocks (CL1) provided in a plurality of work places for having worker's entrance and exit times recorded therein and managing means (CL2) for inputting time recording information from the respective time clocks and managing the work place where the worker entered and exited and how long he/she has engaged therein. The time clocks each comprise fingerprint inputting means (CL11) for inputting a fingerprint of each worker, worker specifying means for having personal information and fingerprint information of each worker stored therein in advance, conducting collation of the input fingerprint with the fingerprint inputting means to specify the worker, and time recording means (CL13) for having worker's entrance and exit times recorded therein on the basis of the specification of the worker by the worker specifying means.

1. Field of the Invention
The present invention relates to a worker management device for managing a working situation of employees based on collation of fingerprint.
2. Description of the Related Art
Conventionally, a management of working situations of employees of a work place or an office is conducted by recording entrance and exit times using a time clock.
However, a conventional time clock is not suitable for determining where and how long which employee has been engaged. For example, in the work place, in addition to full-time workers, various workers such as temporary workers, part-time workers, or the like, as well as staggered work shifts exist. Further, in addition to the headquarters, a plurality of work places such as respective branch offices, sales offices, or the like exist. Therefore, it is difficult to accurately determine at which work place each of employees is at present and how long he/she has engaged in that place only by recording of entrance and exit times of each worker using a time clock.


Provided is a method for manufacturing a brass-plated steel wire in which improvement in the quality of the brass-plated steel wire and energy saving in the manufacturing process are balanced and a brass-plated steel wire obtained by the method.

In a manufacturing process of a brass-plated steel wire represented by a steel cord of a reinforcing material for tires, a pre-drawing is performed generally by dry drawing, and then, a perlite steel structure is built by a patenting thermal treatment, followed by brass plating on the surface of a steel wire rod, and the steel wire rod is subjected to a wet drawing which is a final wire drawing process. As a means for brass plating, a thermal diffusion plating method in which zinc plating is performed on a copper plated layer, and then a brass-plated layer is created by thermal diffusion, is generally employed.
Examples of a technique relating to improvement of a method for manufacturing a brass-plated steel wire include Patent Document 1. Patent Document 1 relates to a method for manufacturing a brass plated bead wire in which a steel wire is plated with copper and zinc sequentially, and then a thermal treatment which corrects wire drawing strain and a thermal diffusion of copper and zinc plated on the steel wire are simultaneously allowed to proceed, and thereafter, the steel wire is subjected to a pickling treatment by an inorganic acid. A brass plated bead wire in which the adhesive properties with rubber are improved without compromising the mechanical properties can thus be obtained.

Method for manufacturing brass-plated steel wire and brass-plated steel wire

Abstract: The method is a method for manufacturing a brass-plated steel wire comprising a plating process in which a steel wire rod is brass plated and a final wire drawing process in which the obtained brass-plated steel wire rod is subjected to a final drawing. The method includes a zinc oxide removing process in which the amount of zinc oxide on the surface of the brass-plated steel wire rod is made smaller than 50 mg/m2 before the final wire drawing process.


When switching between a DVD-video mode and a DVD-audio mode in a DVD-A/V player, a current video frame is stored in a current display buffer portion of the memory during the DVD-video mode. The DVD-A/V player is paused in the DVD-video mode and set in the DVD-audio mode. If it is determined that the current display buffer portion of the memory is a reserved display buffer portion of the memory, then the current video frame is copied to a reconstructed display buffer portion of the memory. At least the current display portion of the memory is designated as an ASV buffer and a frame buffer management scheme is changed so as to preserve the ASV buffer.

1. Field of the Invention
This invention relates generally to consumer electronics. More particularly, the invention relates to efficiently storing audio and video data in a memory device.
2. Description of the Related Art
DVD-Audio (DVD-A) is a Digital Versatile Disk (DVD) format that is specifically designed to hold audio data, and particularly, high-quality music. The DVD Forum, consisting of 230 leading companies worldwide, released the final DVD-A specification in March of 1999. The new DVD format is said to provide at least twice the sound quality of audio CD on disks that can contain up to seven times as much information. Various types of DVD-A-compatible DVD players are being manufactured, in addition to the DVD-A players specifically developed for the format.
Almost all of the space on a DVD video disk is devoted to containing video data. As a consequence, the space allotted to audio data, such as a Dolby Digital 5.1 soundtrack, is severely limited. A lossy compression technique—so-called because some of the data is lost—is used to enable audio information to be stored in the available space, both on standard CDs and DVD-Video disks. In addition to using lossless compression methods, DVD-A also provides more complexity of sound by increasing the sampling rate and the frequency range beyond what is possible for the space limitations of CDs and DVD-Video. DVD-Audio is 24-bit, with a sampling rate of 96 kHz; in comparison, DVD-Video soundtrack is 16-bit, with a sampling rate of 48 kHz, and standard audio CD is 16-bit, with a sampling rate of 44.1 kHz.
DVD-Audio allows for a wide variety of audio formats at varying levels of specification. DVD-Audio supports the same multi-channel audio formats used with DVD-Video. Therefore both DVD-Video and DVD-Audio can provide high definition multi-channel audio recorded in the Dolby Digital and DTS audio formats. However, real advantage of the DVD-Audio specification over DVD-Video and CD is in the significantly increased quality of the PCM audio format. PCM or "Pulse-Coded-Modulation" is the audio format standard for CD's and available on many DVD-Video's. DVD-Audio supports a significantly higher quality of PCM audio than is possible on CD or DVD-Video. DVD-Audio PCM can be recorded with a range of frequencies that are more than four times that of a CD while DVD-Audio PCM has a much greater dynamic range that possible on a CD. The greater storage capacity of DVD's allows for much more music to be recorded than possible on CD's.
The following table outlines the technical specifications for PCM on DVD-Audio and standard CD's.
TABLE ISpecificationDVD-AudioCDAudio FormatPCMPCMDisk Capacity4.7 Gb-Single layer650MB8.5 Gb-Dual Layer17 Gb-Double SidedDual LayerChannelsUp to 62(stereo)Frequency Response0–96 khz (max)5–20kHzDynamic Range144 db96dbSampling Rate-244.1, 88.2, 176.4 Khz or44.1kHzchannel48, 96, 192 KHzSampling Rate-44.1, 88.2 Khz orn/amultichannel48, 96 KhzSample Size12, 16, 20, or 24 bits16bits(Quantization)Maximum Data Rate9.6 MBps1.4MBps
In addition to audio, a DVD-Audio disk can contain a limited amount of video, which can be used to display text, such as lyrics or notes; or stills such as a photo album. Such still images, described as Audio Still Video (ASV), are recorded on disc in a separate file. Each still is an MPEG-2 intra-frame and can, optionally, be accompanied by a subpicture for a menu. Still images are pre-loaded into the player's memory before the audio is played or between audio tracks where a mute is acceptable. This allows the images to be presented either as a slide show or browsable by the user. Transitions for still images include cut, fade, dissolve and wipe. Subpictures allow still images to be used as menus or for the display of lyrics etc.
It would be desirable to have the capability of displaying a video frame from a DVD-Video mode during a transition from the DVD-Video mode to the DVD-Audio mode.

Method and apparatus for efficiently allocating memory when switching between DVD audio and DVD video


A read channel component of a magnetic recording system employs equalization of a signal received from the magnetic recording channel, the equalization being modified depending upon the presence or absence of DC shifts in the signal. Equalization corrects for DC shifts, if present, prior to detection and decoding of servo data, such as servo address mark (SAM) and Gray code data. In a first implementation, a DC shift detector detects the presence or absence of DC shifts and modifies equalization in a predetermined manner. In a second implementation, filtering is applied to the signal to enhance equalization in the presence of DC shift, and both filtered and unfiltered signals employed for detection of the servo data.

1. Field of the Invention
The present invention relates to detection of data in a communications system, and, more particularly, to processing of servo data information read from a channel.
2. Description of the Related Art
A read channel integrated circuit (IC) is a component of a modern disk drive, such as a hard disk drive found in many PCs. A read channel component converts and encodes data to enable the (e.g., magnetic) recording head(s) to write data to the disk and then read back the data accurately. The disks in a hard disk drive typically include many tracks containing encoded data, and each track comprises one or more of user (or "read") data sectors as well as "servo" data sectors embedded between the read sectors. The information of the servo sectors aids in positioning the magnetic recording head over a track on the disk so that the information stored in the read sectors may be retrieved accurately.
FIG. 1 shows a conventional magnetic recording system 100 of the prior art. Servo information is encoded by block encoder 101, and block encoder 101 may represent one or more different encoders associated with different fields of the servo information, such as Gray code and servo address mark (SAM) data. The encoded servo information is written to the disk (or other recording medium) as servo sector information.
FIG. 2 shows the format of servo sector information 200. Servo sector information 200 comprises preamble 201 (e.g., a 2T pattern) that allows the system to recover the timing and gain of the written servo data. Preamble 201 may be followed by encoded SAM data 202, which is generally an identical identification address (fixed number of bits) for all servo sectors. SAM data 202 may then be followed by Gray data 203 (i.e., encoded Gray code). Gray data 203 represents track number/cylinder information and may be employed as coarse positioning information for the magnetic head. One or more burst demodulation fields 204 follow Gray data 203. Burst demodulation fields 204 are employed as fine positioning information for the head over the track. Repeatable-run-out (RRO) data field 205 follows burst demodulation fields 204. RRO data in RRO data field 205 provides head-positioning information to correct for RRO, which occurs when the head does not track an ideal path over the disk. RRO information is finer than that provided by the Gray data and coarser than that provided by the burst demodulation fields.
Returning to FIG. 1, the encoded servo information is read back by a magnetic recording head. Together, the process of writing to, storing on, and reading from the disk by the recording head may be modeled as magnetic recording channel 102 with added noise and DC shifts. Data read from the disk is referred to as readback data. The readback data is equalized to a desired target partial response by equalizer 103. Equalizer 103 comprises continuous time filter (CTF) 120 followed by discrete time, finite impulse response (FIR) filter 121. Sampling of the signal from CTF 120 might be accomplished via switch 122. Sampling might be synchronous using the timing information from digital phase locked loop (DPLL) 123 when servo SAM, Gray, and demodulation burst data are read, but might also be asynchronous if DPLL 123 is not used. Sampling of the signal from CTF 120 might be asynchronous when RRO data is read. The output of equalizer 103 is digitized and quantized by analog-to-digital converter (ADC) 104, whose output is shown as Y values.
For either synchronous and asynchronous sampling, the Y values might be applied to data detector 105, which is typically a partial-response maximum-likelihood (PRML) detector employing, for example, a Viterbi algorithm. Detector 105 may also be implemented with a slicer. Constraints imposed by the servo-encoding algorithm of block encoder 101 might be employed in the design of data detector 105 for optimal decoding of the encoded servo information. The output of data detector 105 is applied to SAM detector 107 to detect the SAM data. The output of data detector 105 and the output of SAM detector 107 are applied to Gray code decoder 108 to generate decoded Gray data. The 'Y' values are also applied to burst demodulator 111 to generate fine positioning information for the head over the track.
For asynchronous sampling, such as when DPLL 123 is not used or when reading RRO data, data phase generator 109 and data phase selector 110 might be employed. Data phase generator 109 generates one or more additional sample sequences from the Y values, each additional sample sequence having a different phase relative to the phase of asynchronous samples from ADC 104. The one or more additional sample sequences might be generated either by asynchronous over-sampling or by interpolation of the asynchronous samples from ADC 104. The one or more additional sample sequences and the asynchronous samples from ADC 104 are provided to data phase selector 110. Data phase selector 10 selects of the input sequences for use by data detector 105 based on a determination of which sequence phase is closest to those having ideal timing.
In addition to noise, DC (baseline level) shifts might impair the signal of recording channel 102. Performance of magnetic recording system 100, as measured by SAM detection error rate and Gray code detection error rate, might be degraded considerably when large amplitude DC shifts corrupt the encoded servo information signal. These DC shifts might occur when the read head becomes unstable. FIG. 3A shows a graph of waveforms with DC baseline shift (shown as dashed lines) and without DC baseline shift (shown as solid lines) before sampling, and FIG. 3B shows a graph of waveforms with and without DC baseline shift after equalization and sampling (circles are sample points). Shown in FIGS. 3A and 3B are the input servo signal as well as one phase of the servo signal at the output of ADC 104 before data detection, respectively.
Data detector 105 might detect positive and negative peaks in the servo signal, but DC shifts in the servo signal cause severe signal discontinuities. DC shifts might occur i) randomly within the servo signal, ii) with random duration, and iii) multiple times. Consequently, DC shifts of magnetic recording system 100 are different from a fixed DC offset applied to the entire servo signal or a fixed offset applied to the signal corresponding to individual servo-encoded words. Depending on where the DC shifts occur in the signal, the DC shift might lead to a severe reduction in amplitude of the peaks in the signal, preventing reliable data detection regardless of the type of data detection employed by system 100. For example, in FIGS. 3A and 3B, the amplitude of the negative peak around time 1025 is severely degraded. Less reliable data detection results in an increase in the SAM detection and Gray bit error rates, which inhibits proper operation of the servo system and, in particular, the throughput of the servo system.

Processing servo data having DC level shifts


Methods and systems for controlling entertainment systems are described. In accordance with one embodiment, one or more conditional directives are developed and injected from externally of an entertainment system into the system. The conditional directives direct the entertainment system to conditionally perform some action that is directed to enhancing the user's experience. Typically, the conditional directives come from an external third party. A conditional directive comprises one or more conditions and one or more actions. The conditions define conditions that are to be met before an action is performed. The actions define actions that are to be performed in the event that one or more associated conditions are met.

These days, entertainment systems are becoming increasingly sophisticated. For example, some entertainment systems can utilize sophisticated electronic program guides (or "EPGs") that permit user interactivity in a manner that greatly enhances the user's experience. For example, EPGs can permit a user to actually search for programs that are of interest to them, rather than having to wait until the channel of interest carrying the program scrolls by them. Additionally, entertainment systems often include some type of recording medium on which the user can have programs of interest recorded. For example, systems such as personal video recorders (PVRs) and digital video recorders (DVRs) can enable a user to record, to hard disk, different programs that are of interest to them. In some instances, the user need not know the specific channel and time that a particular program is broadcast in order for the system to record the program. That is, EPGs typically include metadata that is associated with the programs that occur in its listing. This metadata can provide the basis by which interesting and thorough searches can be conducted by the system, for the purpose of identifying and recording programs of interest to the user.
Yet, against the backdrop of these fairly sophisticated and, many times automated systems, problems can exist that result in a user experience that is not all that it could be. For example, it is a fairly simple matter to enable a user to schedule and record a series program such as Friends. The user can, for example, enter the title of the program via their system's user interface, and the system can then automatically record the Friends episodes as they are broadcast. There are instances, however, when there might be television programs that are related in some way to programs that are of interest to the user. Yet, their entertainment system's heuristics and searching mechanism may not necessarily pick up the program. For example, assume that a program called "The Making of Friends" is scheduled to air. This program may contain interviews with the various actors in the series, as well as historical information of interest. Yet, the metadata associated with this program might be sufficiently different from the metadata associated with the series Friends such that the entertainment system does not pick it up. Here then is an instance where a program that is likely to be of interest to the user escapes recording.
The above example is a fairly simple and specific example which illustrates a more general common problem associated with current automated entertainment systems. Specifically, by virtue of the fact that entertainment systems are computerized, they typically do not "think" like humans. Thus, decisions that would be very easy for a human to make (e.g. "let's record The Making of Friends because I really like the series Friends") can be very difficult for an automated system to make.
Accordingly, this invention arose out of concerns associated with providing improved entertainment methods and systems.

Methods and systems for conditionally managing entertainment systems


Described are a network, network device, and method for a method of distributing routing information for a virtual private network (VPN) application through a packet-switched network (PSN) having fully meshed provider edge (PE) routers through Provider Backbone Bridge (PBB) tunnels. Each PE router is configured to support at least one VPN and to run a BGP (Border Gateway Protocol) as an auto-discovery process for finding one or more other PE routers attached to the same VPN. A given VPN is associated with a PBB tunnel. A service instance identifier (I-SID), as defined in IEEE 802.1ah, is assigned to the VPN. Each PE router advertises membership in the VPN by including the I-SID assigned to the VPN in a BGP message issued during the auto-discovery process.

Virtual private networks, or VPNs, are becoming increasingly popular as a cost-effective means for conducting voice and data communications between, for example, corporate data centers, remote offices, mobile employees, customers, suppliers, and business partners. In general, a VPN is a private network configured within a public network, such as a service provider's network or the Internet. The VPN of a given customer appears privately dedicated to that customer, when in actuality the customer's VPN shares the same physical backbone with the VPNs of many other customers.
Increasingly, service providers are using MPLS (Multi-protocol Label Switching) tunneling to implement VPNs across their packet-switched networks. Two major types of IP/MPLS-based VPNs have arisen: (1) layer-3 VPN services, referred to as L3VPNs; and (2) layer-2 VPN services, referred to as L2VPNs. A standard for L3VPNs is described in Internet Engineering Task Force ("IETF") Request for Comments ("RFC") 4364, titled "BGP/MPLS IP Virtual Private Networks (VPNs)", the entirety of which is hereby incorporated by reference herein. Implementations of L2VPNs are described in RFC 4761, titled "Virtual Private LAN Service (VPLS) Using BGP for Auto-Discovery and Signaling", in RFC 4762, titled "Virtual Private LAN Service (VPLS) Using Label Distribution Protocol (LDP) Signaling", and in IETF draft-ietf-12vpn-signaling-08.txt, titled "Provisioning, Autodiscovery, and Signaling in L2VPNs", the entireties of which are hereby incorporated by reference herein.
Implementing a MPLS-based VPN generally requires the distribution of certain routing information between provide edge (PE) routers of the service provider's network. A commonly used inter-domain routing protocol for exchanging such routing information is the Border Gateway Protocol, or BGP. For example, using BGP messages, PE routers exchange VPN routes (layer-3) with customer edge (CE) routers and with other PE routers in the service provider network. For layer-2 VPNs, PE routers use BGP update messages to exchange VPLS (Virtual Private LAN Service) membership and demultiplexer information with other PE routers in the same VPLS.
With the advent of connection-oriented forwarding technologies such as Provider Backbone Transport (PBT) and Provider Backbone Bridge (PBB), native Ethernet is rapidly emerging as a viable packet-switched network technology. Consequently, Ethernet is becoming more widely used, particularly in metro-area networks and wide-area networks. With PBT, service providers are able to establish point-to-point and point-to-multipoint Ethernet tunnels and to specify paths that service traffic will take through their Ethernet networks. With PBB, service providers are able to separate a communications network into customer domains and service provider domains. The separation is achieved by encapsulating the customer packets within a backbone (i.e., service provider) MAC (Media Access Control) header. Network devices in the service provider domain forward packets based on the service provider MAC header while the customer header remains invisible except at the edge of the service provider domain. Having such capabilities, service providers are desirous of supporting layer-2 and layer-3 BGP-VPN applications over their Ethernet networks. Notwithstanding, to support such BGP-VPN applications, control plane mechanisms are needed by which PE routers may control what routing information to accept and distribute.

Border gateway protocol extended community attribute for layer-2 and layer-3 virtual private networks using 802.1ah-based tunnels


A transfer system that supports circuit boards as it conveys them through a reflow oven. The system uses a sag support between edge supports. The sag support does not have to move as the circuit board is transferred along the transfer system. The height of the sag support can be easily adjusted. The sag support's mounts can maintain appropriate tension in the sag support even if the sag support expands during operation. The system may use a mechanism to prevent the friction between the boards and the sag support from interfering with the movement of boards along the transfer system.

1. Field of the Invention
The invention relates generally to an apparatus for supporting printed circuit boards during manufacture and, more particularly, to a thin support for supporting boards as they are transferred through inline pass through ovens.
2. Description of Related Art
Inline pass through ovens are often used to manufacture printed circuit boards. Such ovens have been used, for example, to solder surface mount devices to a board or to cure adhesives securing components to a board. Typically, to solder surface mount devices, solder paste is screen printed onto the circuit board in desired locations. The surface mount components are then appropriately placed in the solder paste. A part of the oven called a transfer system, typically supports the "parallel edges" of the board while it conveys the board through the oven. The inline pass through oven heats the board and solder, soldering the components to the board. "Parallel edges" refers to the edges of a printed circuit board that are basically parallel to direction that the transfer system moves the circuit board. Inline pass through ovens that solder components are often called reflow ovens.
Some conventional transfer systems support the "parallel edges" of a printed circuit board with what we will call "edge supports. " An edge support may comprise support rails, chains and pins extending from the chains. Support rails such as these have been used in ovens and transfer systems manufactured by a company called Research Incorporated, located in Minneapolis, Minn., for example.
Edge supports are typically constructed by fixing support rails to a frame of the pass through oven. Chains typically rotate through these support rails, and pins typically extend from these chains such that a printed circuit board can rest on the pins. The pins typically extend 0.1-0.2 inch beneath the circuit board's parallel edges. As the chains rotate through the support rails, they move the pins such that the chains and pins carry circuit boards through the oven. We will refer to the process of conveying a board along a manufacturing line (e.g. through an oven) as "transferring" the board.
Reflow ovens typically use a specific thermal profile to solder components. For example, a reflow oven may have 4 or more temperature zones through which the board travels. The first 4 or 5 zones, for example, typically bring a board up to a desired temperature in a controlled manner. The final zone, called a spike zone, rapidly brings the board up to reflow temperatures, typically in the range of 183.degree.-210.degree. Celsius, soldering the components to the board. Inline pass through ovens used for purposes other than solder reflow often operate in a similar manner, but may use different thermal profiles.
The transfer systems used with some conventional inline pass through ovens support only the parallel edges of the printed circuit boards. Depending upon the weight of the components on a printed circuit board and/or the thickness of the board, however, the elevated thermal profile used with reflow or other types of inline pass through ovens may cause the circuit board to sag. Sagging can occur because boards, sometimes made of epoxy-glass laminates, may go through a glass transition phase and become weak at certain temperatures. For example, a widely available epoxy-glass laminate that has been used in printed circuit boards, FR4, typically has a transition phase around 150.degree. C. Generally, the higher the temperature above this transition temperature, the weaker a board becomes. Sagging can result in warped boards that may not fit into their final assembly. It can result in open solder joints because the solder cannot bridge the gap between the component and the warped board. Extreme warping can result in boards falling off the transfer system. Accordingly, parallel edge support, alone, may be adequate for some rigid printed circuit boards, but additional support may be required to control sagging of thin printed circuit boards or printed circuit boards with heavy components.
Conventional approaches to preventing sagging have added a dynamic or moving support between the edge supports. One conventional approach has used a steel cable to provide this additional support. This approach attaches the cable to a drive system. The cable's drive system causes the cable to move with the chains and pins of the edge supports as they carry the board through the oven.
Steel cable, however, has an inherent tendency to sag from its own weight. To insure a straight support surface, this approach must apply a high tensile force to the cable. Additionally, because of space constraints, the drive system may force the cable through tight radii. Tight radii in a dynamic system combined with the high tensile force can accelerate cable fatigue and failure. Accordingly, conventional dynamic steel cable systems have failed to provide an optimum solution to the sagging problem.
Another conventional approach to preventing sagging adds a support rail and roller chain between the edge supports. Further, the thermal characteristics of the rail and chain can affect the heating of the board. In fact, conventional systems have employed methods for heating support rails to prevent heating problems caused by the thermal characteristics of such rails. Such heating methods potentially add even further cost and complexity to the system. An additional rail and roller chain between the edge supports can also limit printed circuit board layouts by requiring a wide contact area (an area where no components may exist) on the bottom of the circuit board where the roller chain supports the board. Components in this area might be dislodged or damaged by the rail or roller chain. Accordingly, dynamic rail and roller chain systems have also failed to provide an optimum solution to the sagging problem.
There has been a need for a support system that does not have the complexity of conventional dynamic systems. There has been a need for a support system less sensitive to the fatigue or durability problems of conventional dynamic systems. There has been a need for a system that will have minimal impact on printed circuit board layouts and the thermal characteristics of the heating process. There has been a need for a system that prevents sagging using a sag support that does not have to move during transfer. The present invention meets these needs.

Thin support for PC board transfer system


Apparatus for securing a motor vehicle against theft uses a vehicle control device connected by a signaling link to a first control node of a vehicle module essential to mobilization of the vehicle, the essential module further having a power node connected to a power supply source. The apparatus comprises: an enable/disable control for conditionally disabling signaling along the signaling link, a user interface code input device communicating with the enable/disable control via a secure command link, and an armored encapsulation encapsulating the disable control and the control node and physically configured to render inoperable any of the disable control, the signaling link and the control node upon being tampered with.

The present invention relates to prevention of vehicle theft and, more particularly, to the prevention of vehicle theft using a combination of electronic or mechanical control and physical armored encapsulation.
Vehicle theft is a well-known and widespread crime, causing severe financial damage and affecting the public sense of well-being and morale.
Numerous motor vehicle anti-theft devices are known in the art. These include, inter alia, mechanical locks, electronic alarm systems, electronic engine immobilizers, automatic vehicle location systems and devices for disabling the fuel supply to the engine.
Conventional alarm type theft preventing devices typically comprise a control unit, a loud speaker, a plurality of sensors and a remote controller. The control unit and the loud speaker are installed in the inside of the car, while the remote controller is used for transmitting a control signal to turn the control unit on or off. When the control unit is turned on, it is capable of triggering the loud speaker to give off an alarm signal to frighten the burglar and alert passers-by. Nowadays, it is appreciated that alarm signals themselves are ignored by passers by and thus do not in themselves deter the thief. Thus vehicles are typically further equipped with some kind of engine immobilizer. In one kind of immobilizer the power source of the starter is shut off whilst the legal owner has left the vehicle Such an immobilizer is only partly successful as often the thief is able to start the vehicle before the immobilizer has armed or he is able to disable the alarm by electrically bypassing it Alternatively, the thief can disable or reduce the operation of the alarm loud speaker by inserting a foamy material thereto. Other known systems choke off the fuel supply to the engine.
A common drawback for many of the anti-theft systems is the ability of the thief to access the anti-theft system thereby to disable it. Some systems offer an armored enclosure to serve as defense against violent attacks of the switching device, and against attempts to tamper with the security system.
U.S. Pat. No. 4,288,778 to Zucker teaches a method of using digital electronic anti-theft system integrally formed with a vehicle component.
U.S. Pat. No. 4,209,709 to Betton discloses an electronic ignition system in which electronic circuitry is located in an auxiliary metallic housing which is configured to surround both the solenoid and starter motor of the vehicle. The electrical interconnection between the circuitry and the solenoid are accomplished within the protection of the auxiliary housing. U.S. Pat. No. 4,533,016 to Betton discloses an electronic circuit enclosed within a solenoid housing with means for inhibiting actuation of the solenoid in the absence of entry of a predetermined code from the ignition switch of the vehicle.
French patent FR2764570 to Duval discloses a keypad controlling a starter relay voltage supply, where the starter motor may further be locked. Anti-theft screws with security lugs are used to avoid dismantling of the starter.
U.S. Pat. No. 5,564,376 to Labelle, discloses a housing, which is connectable to the starter motor casing, and serves as enclosure to a control circuit. U.S. Pat. No. 6,227,158 to Labelle discloses a similar arrangement for the starter solenoid; U.S. Pat. No. 6,116,201 and WO0071395 to Labelle teach the use of the starter solenoid housing itself as enclosure for an integrated security chip.
Also of prior art of interest are devices which are used to control the security system. These include, U.S. Pat. No. 4,733,638 which discloses a hand-held transmitter, French Patent No. FR2769562 which discloses the use of an RF signal, U.S. Pat. No. 6,069,411 which discloses the use the Electronic Serial Number of a cellular phone as an ID code, U.S. Pat. No. 5,808,543 which discloses such a system dependent on a radio frequency identification device badge usually used in vehicle key systems, and U.S. Pat. No. 5,704,008 which teaches a voice or password recognition control of the starter solenoid.
Despite the use of the above devices, thieves nevertheless find ways and means of overcoming (e.g., by bypassing) the various protection devices. Mechanical locking devices are broken and by-passed, cut-off valves are circumvented or disabled, electronic engine immobilizers are hot-wired and even sophisticated anti-theft systems are overcome by key-theft or lock picking. A recently observed theft method involves a replacement kit for the vehicle Engine Control Unit (ECU), which is used to bypass the whole security system. A thief equipped with a pre-prepared replacement kit, a key and a key decoder, can fit the unit within minutes and drive away with the vehicle.
It is thus appreciated that the non-accessibility of the anti-theft system to the thief is of utmost importance. Furthermore, it is important that such non-accessible system would be protected against bypassing. Many known physically armored devices are aimed at protecting the vehicle via starter disabling methods. Such devices may be categorized into two groups, built-in control devices, destined to be integrated within the solenoid housing, and add-on armored control devices, destined to be mounted over the solenoid housing. Built-in control devices require re-design of the starter solenoid for new vehicles, or the exchange of a solenoid or solenoid-starter assembly in pre-existing vehicles.
Add-on armored control devices according to prior art require very large protective covers that may interfere with engine compartment serviceability and with proper ventilation of the engine bay. In addition, installation of presently known add-on armored control devices is very complicated, if at all possible, in existing vehicles. Furthermore, prior art built-in or add-on control devices may be unable to withstand the harsh environmental conditions typical of vehicle engine compartments.
Examples of such protective covers are given in U.S. Pat. Nos. 6,351,209 and 5,548,164. These examples suffer from some or all of the disadvantages listed above they further are very complex to manufacture, require considerable installation efforts and require a plurality of models to fit most vehicle types.
There is thus a widely recognized need for, and it would be highly advantageous to have, a system for preventing vehicle theft devoid of the above limitations.

This patent is subject to a terminal disclaimer.

Device system and method for preventing vehicle theft

StarTech Automotive Anti Theft Systems Ltd.


A method for processing a structure. The structure is formed and includes a substrate, a substructure having a sidewall and disposed on the substrate, a first polymer structure disposed on the substrate, and a second polymer structure disposed on the substrate such that the first polymer structure is disposed between the sidewall and the second polymer structure. An aspect ratio of the first polymer structure, the second polymer structure, or both is reduced in a reducing step. One polymer structure (i.e., the first polymer structure or the second polymer structure) is selectively removed from the structure such that a remaining polymer structure (i.e., the second polymer structure or the first polymer structure) remains disposed on the external surface of the substrate after the one polymer structure has been selectively removed, wherein the aspect ratio of the remaining polymer structure was reduced in the reducing step.

Smaller critical dimension (CD) allows denser circuitry to be created and therefore reduces the overall production cost of microelectronic devices. However, there exists a need for a high throughput method to cost effectively pattern features with dimensions or pitches smaller than those which can be fabricated by optical lithography. Self-segregating polymer blends provide a route to generate self-assembled polymer features next to existing topographical features on a wafer. However, such self-assembled polymer features may become distorted or even collapse during their formation or further processing of the resultant polymer pattern. Thus, there is a need to improve the self-assembled polymer features resulting from self-segregated polymer blends in order to retain pattern fidelity.

Method for improving self-assembled polymer features

International Business Machines Corporation


In an intermediate potential generating circuit provided in a DRAM, a reference potential generating circuit includes two combinations of a resistor element constituted by a P channel MOS transistor and a diode constituted by an N channel MOS transistor, and outputs a reference potential Vcc/2+Vthn. A charging circuit constituted by an N channel MOS transistor charges an output node to an intermediate potential Vcc/2 based on the reference potential. A discharging circuit includes one combination of a resistor element constituted by a P channel MOS transistor and a diode constituted by an N channel MOS transistor, and provides a prescribed current flowing from the output node. Reduction of supply voltage Vcc becomes possible by eliminating a diode constituted by a P channel MOS transistor which makes reduction of a threshold voltage Vthp difficult.

1. Field of the Invention
The present invention relates to an intermediate potential generating circuit, and more particularly relates to an intermediate potential generating circuit for producing an intermediate potential which is between a first potential and a second potential lower than the first potential, and outputting it to an output terminal.
2. Description of the Background Art
A dynamic random access memory (hereinafter referred to as DRAM) is conventionally provided with an intermediate potential generating circuit for producing an intermediate potential Vcc/2 which is between a power supply potential Vcc and a ground potential GND. Intermediate potential Vcc/2 generated in the intermediate potential generating circuit is utilized as a precharge potential of a bit line and as a cell plate potential.
FIG. 3 is a circuit diagram showing a structure of a conventional intermediate potential generating circuit. With reference to FIG. 3, the intermediate potential generating circuit is provided with two reference potential generating circuits 31 and 36 and a drive circuit 41.
One reference potential generating circuit 31 includes a resistor element 32, N channel MOS transistors 33, 34 and a resistor element 35 that are serially connected between a line of power supply potential Vcc (hereinafter referred to as a power supply line) 20 and a line of ground potential GND (hereinafter referred to as a ground line) 21. Each of N channel MOS transistors 33 and 34 is diode-connected. Specifically, respective gates of N channel MOS transistors 33 and 34 are connected to their own drains. Resistor elements 32 and 35 have an equal resistance value and N channel MOS transistors 33 and 34 have an equal threshold voltage Vthn. Accordingly, an intermediate node N33 between N channel MOS transistors 33 and 34 is at intermediate potential Vcc/2, and an output node N32 between resistor element 32 and N channel MOS transistor 33 is at a first reference potential Vcc/2+Vthn.
The other reference potential generating circuit 36 includes a resistor element 37, P channel MOS transistors 38, 39 and a resistor element 40 that are serially connected between power supply line 20 and ground line 21. Respective gates of diode-connected P channel MOS transistors 38 and 39 are connected to their own drains. Resistor elements 37 and 40 have an equal resistance value and P channel MOS transistors 38 and 39 have an equal threshold voltage Vthp. Accordingly, an intermediate node N38 between P channel MOS transistors 38 and 39 is at intermediate potential Vcc/2, and an output node N39 between P channel MOS transistor 39 and resistor element 40 is at a second reference potential Vcc/2-Vthp.
Drive circuit 41 includes an N channel MOS transistor 42 and a P channel MOS transistor 43 connected in series between power supply line 20 and ground line 21. The gate of N channel MOS transistor 42 is connected to output node N32 of reference potential generating circuit 31 and the gate of P channel MOS transistor 43 is connected to output node N39 of reference potential generating circuit 36. A node N42 between MOS transistors 42 and 43 is an output node of the intermediate potential generating circuit.
An operation of this intermediate potential generating circuit will next be described. The output potential Vcc/2+Vthn of reference potential generating circuit 31 is supplied to the gate of N channel MOS transistor 42 of drive circuit 41, and the output potential Vcc/2-Vthp of reference potential generating circuit 36 is supplied to the gate of P channel MOS transistor 43 of drive circuit 41.
If a potential Vout of output node N42 is lower than intermediate potential Vcc/2, N channel MOS transistor 42 becomes conductive and output node N42 is charged. At this time, the potential of the gate of N channel MOS transistor 42 is Vcc/2+Vthn, so that output node N42 which is a source of N channel MOS transistor 42 is charged only to intermediate potential Vcc/2.
If potential Vout of output node N42 becomes higher than intermediate potential Vcc/2, P channel MOS transistor 43 becomes conductive and output node N42 is discharged. At this time, the potential of the gate of P channel MOS transistor 43 is Vcc/2-Vthp, so that output node N42 which is a source of P channel MOS transistor 43 is discharged only to intermediate potential Vcc/2. Therefore, potential Vout of output node N42 of the intermediate potential generating circuit is maintained at intermediate potential Vcc/2.
FIG. 4 is a circuit diagram showing a structure of another conventional intermediate potential generating circuit. Referring to FIG. 4, the intermediate potential generating circuit is provided with two reference potential generating circuits 51, 56 and a drive circuit 61.
One reference potential generating circuit 51 includes a P channel MOS transistor 52, an N channel MOS transistor 53, a P channel MOS transistor 54, and an N channel MOS transistor 55 connected in series between power supply line 20 and ground line 21. Each of N channel MOS transistors 53 and 55 is diode-connected. Respective gates of P channel MOS transistors 52 and 54 are connected to respective sources of N channel MOS transistors 53 and 55. Since gates of P channel MOS transistors 52 and 54 are respectively connected to nodes of low potential over N channel MOS transistors 53 and 55, each of P channel MOS transistors 52 and 54 operates as a resistor element. P channel MOS transistors 52 and 54 are identical in size, and N channel MOS transistors 53 and 55 have equal threshold voltage Vthn. Accordingly, an intermediate node N53 between N channel MOS transistor 53 and P channel MOS transistor 54 attains to intermediate potential Vcc/2, and an output node N52 between P channel MOS transistor 52 and N channel MOS transistor 53 attains to first reference potential Vcc/2+Vthn.
The other reference potential generating circuit 56 includes a P channel MOS transistor 57, an N channel MOS transistor 58, a P channel MOS transistor 59 and an N channel MOS transistor 60 connected in series between power supply line 20 and ground line 21. Each of P channel MOS transistors 57 and 59 is diode-connected. Respective gates of N channel MOS transistors 58 and 60 are connected to respective sources of P channel MOS transistors 57 and 59. The gates of N channel MOS transistors 58 and 60 are respectively connected to high potential nodes over P channel MOS transistors 57 and 59, so that each of N channel MOS transistors 58 and 60 operates as a resistor element. N channel MOS transistors 58 and 60 are identical in size, and P channel MOS transistors 57 and 59 have equal threshold voltage Vthp. Accordingly, an intermediate node N58 between N channel MOS transistor 58 and P channel MOS transistor 59 attains to intermediate potential Vcc/2, and an output node N59 between P channel MOS transistor 59 and N channel MOS transistor 60 attains to second reference potential Vcc/2-Vthp.
Drive circuit 61 is provided with an N channel MOS transistor 62 and a P channel MOS transistor 63 connected in series between power supply line 20 and ground line 21. The gate of N channel MOS transistor 62 is connected to output node N52 of reference potential generating circuit 51, and the gate of P channel MOS transistor 63 is connected to output node N59 of reference potential generating circuit 56. A node N62 between MOS transistors 62 and 63 is an output node of the intermediate potential generating circuit.
An operation of the intermediate potential generating circuit is described below. Output potential Vcc/2+Vthn of reference potential generating circuit 51 is supplied to the gate of N channel MOS transistor 62 of drive circuit 61, and output potential Vcc/2-Vthp of reference potential generating circuit 56 is supplied to the gate of P channel MOS transistor 63 of drive circuit 61.
If potential Vout of output node N62 becomes lower than intermediate potential Vcc/2, N channel MOS transistor 62 becomes conductive and output node N62 is charged up to intermediate potential Vcc/2. If potential Vout of output node N62 becomes higher than intermediate potential Vcc/2, P channel MOS transistor 63 becomes conductive and output node N62 is discharged up to intermediate potential Vcc/2. Accordingly, potential Vout of output node N62 of the intermediate potential generating circuit is maintained at intermediate potential Vcc/2.
FIG. 5 is a circuit diagram illustrating a structure of still another conventional intermediate potential generating circuit. With reference to FIG. 5, the intermediate potential generating circuit differs from that in FIG. 4 in that drive circuit 61 is substituted by a drive circuit 70.
Drive circuit 70 includes a P channel MOS transistor 71, an N channel MOS transistor 72, a P channel MOS transistor 73, and an N channel MOS transistor 74 connected in series between power supply line 20 and ground line 21, as well as a P channel MOS transistor 75 and an N channel MOS transistor 76 connected in series between power supply line 20 and ground line 21. The gate of N channel MOS transistor 72 is connected to output node N52 of reference potential generating circuit 51, and the gate of P channel MOS transistor 73 is connected to output node N59 of reference potential generating circuit 56. A node N72 between MOS transistors 72 and 73 is an output node of the intermediate potential generating circuit. Node N72 is connected to drains of MOS transistors 75 and 76.
Gates of P channel MOS transistors 71 and 75 are both connected to a drain of P channel MOS transistor 71, and P channel MOS transistors 71 and 75 thus constitute a current mirror circuit. Gates of N channel MOS transistors 74 and 76 are both connected to a drain of N channel MOS transistor 74, and N channel MOS transistors 74 and 76 thus constitute a current mirror circuit.
An operation of the intermediate potential generating circuit is next described. Output potential Vcc/2+Vthn of reference potential generating circuit 51 is supplied to the gate of N channel MOS transistor 72 in drive circuit 70, and output potential Vcc/2-Vthp of reference potential generating circuit 56 is supplied to the gate of P channel MOS transistor 73 in drive circuit 70.
If potential Vout of output node,N72 becomes lower than intermediate potential Vcc/2, N channel MOS transistor 72 becomes conductive and current is caused to flow from power supply line 20 to output node N72 through MOS transistors 71 and 72. At this time, since P channel MOS transistors 71 and 75 constitute the current mirror circuit, the electric current flowing from power supply line 20 through P channel MOS transistor 75 to output node N72 has a value corresponding to the value of the current flowing through P channel MOS transistor 71. Accordingly, potential Vout of output node N72 attains to intermediate potential Vcc/2 immediately.
On the other hand, if potential Vout of output node N72 becomes higher than intermediate potential Vcc/2, P channel MOS transistor 73 becomes conductive and electric current flows from output node N72 to ground line 21 through MOS transistors 73 and 74. At this time, N channel MOS transistors 74 and 76 constitute the current mirror circuit, so that the value of the electric current flowing from output node N72 through N channel MOS transistor 76 to ground line 21 has a value corresponding to that of the current flowing through N channel MOS transistor 74. Accordingly, potential Vout of output node N72 reaches intermediate potential Vcc/2 immediately. Potential Vout of output node N72 in the intermediate potential generating circuit is thus maintained at intermediate potential Vcc/2.
FIG. 6 is a circuit diagram showing a structure of a further conventional intermediate potential generating circuit. With reference to FIG. 6, the intermediate potential generating circuit differs from that shown in FIG. 3 in that reference potential generating circuit 36 is eliminated and drive circuit 41 is substituted by a drive circuit 81.
Drive circuit 81 is provided with two N channel MOS transistors 82 and 83 connected in series between power supply line 20 and ground line 21. The gate of N channel MOS transistor 82 is connected to output node N32 of reference potential generating circuit 31, and the gate of N channel MOS transistor 83 is connected to power supply line 20. A node N82 between N channel MOS transistors 82 and 83 is an output node of the intermediate potential generating circuit.
Next, an operation of the intermediate potential generating circuit is described. Output potential Vcc/2+Vthn of reference potential generating circuit 31 is supplied to the gate of N channel MOS transistor 82 in drive circuit 81. If potential Vout of output node N82 becomes lower than intermediate potential Vcc/2, N channel MOS transistor 82 becomes conductive and a charging current I82 flows from power supply line 20 to output node N82 through N channel MOS transistor 82. At this time, since the potential of the gate of N channel MOS transistor 82 is Vcc/2+Vthn, output node N82 which is a source of N channel MOS transistor 82 is charged only to intermediate potential Vcc/2.
In the meantime, since a discharging current I83 constantly flows from output node N82 through N channel MOS transistor 83 to ground line 21, potential Vout of output node N82 tends to decrease. Potential Vout of output node N82 is kept at intermediate potential Vcc/2, as discharging current I83 and charging current I82 are balanced.
However, the intermediate potential generating circuits shown in FIGS. 3 to 5 do not operate normally unless requirements of Vcc>2 Vthn+2RI and Vcc>2 Vthp+2RI are met (where R represents a resistance value of the resistor element in the reference potential generating circuit, and I represents a current value flowing in the reference potential generating circuit), since they are provided with reference potential generating circuits 31, 51 including diode-connected N channel MOS transistors 33, 34; 53, 55 as well as reference potential generating circuits 36, 56 including diode-connected P channel MOS transistors 38, 39; 57, 59.
On the other hand, supply voltage Vcc in a DRAM has been reduced in recent years, and threshold voltage Vthn and Vthp of the MOS transistors must be reduced in order to meet the requirements described above.
However, breakdown voltage characteristic is worse in the P channel MOS transistor compared to the N channel MOS transistor in the process technique now available, so that threshold voltage Vthp of the P channel MOS transistor cannot be reduced to the level corresponding to threshold voltage Vthn of the N channel MOS transistor. For this reason, the lowest value of supply voltage Vcc is determined by threshold voltage Vthp of the P channel MOS transistor in the intermediate potential generating circuits shown in FIGS. 3 to 5.
As for the intermediate potential generating circuit in FIG. 6, it is provided with only reference potential generating circuit 31 including diode-connected N channel MOS transistors 33 and 34. Therefore, it is sufficient to meet the requirement Vcc>2 Vthn+2RI only. Therefore, reduction of supply voltage Vcc becomes possible, different from the intermediate potential generating circuits in FIGS. 3 to 5 in which the lowest value of supply voltage Vcc is determined by Vthp higher than Vthn.
However, the balance between charging current I82 and constantly flowing discharging current I83 cannot easily be maintained. In order to strike the balance, a circuit should be designed considering various factors such as supply voltage Vcc, temperature, and device property, and the circuit design becomes difficult.

Intermediate potential generating circuit