In a communication system, an apparatus for receiving a signal transmitted through a channel having time-varying characteristics and for restoring original user information included in the received signal, preferably includes: a first channel characteristic estimator for estimating a signal representing the delay characteristic of the channel; a first channel characteristic predictor for decimating the estimated signal applied thereto, for predicting the characteristic values of the channel, for interpolating the predicted channel characteristic values, and for outputting the interpolated result as a first predictive characteristic signal; and a first receiver for generating a decision value from the first predictive characteristic signal and the received signal, restoring the transmitted signal using the estimated decision value, and outputting a restored transmitted signal, which is fed back to the first channel characteristic estimator.

1. Field of the Invention
The present invention relates to signal reception in a communication system. More particularly, the present invention relates to a signal receiving apparatus and method adapted for use with a time-variant channel.
2. Description of the Related Art
In general, in a communication system including a transmitting side and a receiving side, the receiving side must know a current decision value of a signal received by the receiving side in order to estimate a characteristic of a channel. Herein, the decision value means transmission information restored. At this time, after an elapse of a certain training period of time, the use of the decision value obtains a channel estimate obtained by assuming the characteristic of a channel over which a transmitting signal has been transmitted.
In a case where a conventional signal receiving apparatus estimates the characteristic of a channel varying with time, a reduction in a time delay of the estimated channel characteristic requires a decrease in time spent in estimating the decision value. For this purpose, for example, in Maximum-Likelihood Sequence Estimation (hereinafter, referred to as "MLSE"), a tentative decision value determined with reducing a depth of a traceback is used. Herein, MLSE may also be called Maximum Likelihood Sequence Detection (MLSD), which is described in Proakis, "Digital Communications," Prentice Hall Press, pp. 583–602 (1995). However, since the tentative decision value has a high error rate, there has been a problem in that the characteristic of a channel estimated using the tentative decision value also has an increased error. Consequently, the above-mentioned conventional signal receiving apparatus has a disadvantage in that since there is a significant difference between the estimated characteristic of a channel and the real characteristic of a channel, a decision value used to restore user information cannot be estimated correctly from the estimated channel characteristic with an error.
To overcome this problem, another conventional signal receiving apparatus employs a linear predictor for estimating the characteristic of a time-delayed channel and predicting the characteristic of a current channel from the estimated characteristic of a delayed channel. Herein, the linear predictor can be embodied through a Wiener solution utilizing an autocorrelation of a channel, as is known to those skilled in the art. For the sake of simplicity of the solution, a method of linearly predicting the characteristic of a channel in which only a gradient of the channel is estimated and the channel is approximated with a linear function may be used. However, in the above-mentioned conventional signal receiving apparatus, there also arises a problem in that when the range of the characteristic of a channel to be predicted is long, the above-mentioned predictors cannot correctly predict the characteristic of a channel. Thus, the latter of two conventional signal receiving apparatuses, similar to the former mentioned above, also employs a tentative decision value, so that an error of the tentative decision value contributes to an incorrect restoration of user information.

Signal receiving apparatus and method adapted for use with a time-variant channel


A real-time FROG system provides ultra fast pulse measurement and characterization. The system includes direct, integrated feedback that measures how well the system is retrieving pulses and tracking changes in the pulse train. This feedback is provided in real time and may be in the form of the FROG trace error, the display of the measured and retrieved FROG trace, accuracy of background subtraction or other quality measurement. The system includes preprocessing options that can be used to adjust the dynamic range of the measured signal or to perform different types of filtering. The preprocessing of the FROG trace precedes phase retrieval processing and improves the quality of pulse retrieval.

1. Field of the Invention
The present invention relates to a method and apparatus for the real-time measurement of ultrashort laser pulses based on frequency-resolved optical gating (FROG).
2. Related Art
Ultrashort optical pulses have very short time durations, typically less than a few 10's of picoseconds. As a result, ultrashort optical pulses are spectrally broad. Because the index of refraction of materials is a function of wavelength, different wavelengths of light travel at different speeds in optical materials causing the properties of ultrashort optical pulses to change as they propagate. The shape of the pulse can influence how the pulse itself interacts with materials, further complicating the analysis of ultrashort optical pulse transmission and propagation. Ultrafast laser pulse measurement tries to obtain both the intensity profile of the pulse as well as the "phase of the pulse," which is the actual variation of the frequencies that make up the pulse.
Generally, time resolved measurements of events use shorter events to resolve variations in the characteristics of the events as a function of time. Unfortunately for the analysis of ultrashort optical pulses, shorter events do not exist and modern electronics are insufficiently fast to allow for direct time resolved measurements. One effective measurement technique for ultrashort optical pulses is called "frequency-resolved optical gating" or FROG. FROG is based on the observation that an optically formed spectrogram contains all of the information about the pulse to be measured. A two-dimensional phase retrieval process extracts the pulse from its spectrogram. One disadvantage of frequency-resolved optical gating is the large amount of data required to obtain all of the information needed to measure the pulse. In addition, data is needed both as a function of time and frequency (or wavelength). Aspects of the present invention improve data acquisition for FROG allowing the technique to be more versatile while retaining convenience, accuracy and speed.
Mathematical Representation of an Optical Pulse
The mathematical representation of ultrashort optical pulses is discussed to provide background for the discussion of the FROG technique. The time-dependent variations of an optical pulse are embodied in the pulse's electric field, A(t), which can be written:A(t)=Re[E(t)eiω0t]  (1)where ω0 is the carrier frequency and Re refers to the real part. A(t) can be used in this form for calculations, but it is generally easier to work with a different representation that removes the rapidly varying ω0 part, eiω0t, and uses as a representation a slowly varying envelope together with a phase term that contains only the frequency variations. This representation, which does not include the rapidly varying carrier frequency, isE(t)=[I(t)]1/2eiω(t)  (2)where I(t) and φ(t) are the time-dependent intensity and phase of the pulse. (E(t) is complex.) The frequency variation, Ω(t), is the derivative of φ(t) with respect to time:Ω(t)=−dφ(t)/dt  (3)
The pulse field can be written equally well in the frequency domain by taking the Fourier transform of equation 2:{tilde over (E)}(ω)=[Ĩ(ω)]1/2e−iφ(ω)  (4)where Ĩ(ω) is the spectrum of the pulse and φ(ω) is its phase in the frequency domain. The spectral phase contains time versus frequency information. That is, the derivative of the spectral phase with respect to frequency yields the time arrival of the frequency, or the group delay.
Obtaining the intensity and phase, I(t) and φ(t) (or Ĩ(ω) and φ(ω)) is called full characterization of the pulse. Common phase distortions include linear chirping, where the phase (either in the time domain or frequency domain) is parabolic. When the frequency increases with time, the pulse is said to have positive linear chirp; negative linear chirp is when the high frequencies lead the lower frequencies. Higher order chirps are common, but for these, differentiation between spectral and temporal chirp is required because spectral phase and temporal phase are not interchangeable.
Frequency-Resolved Optical Gating
Frequency-resolved optical gating (FROG) measures the spectrum of a particular temporal component of an optical pulse, shown in FIG. 1, by spectrally resolving the signal pulse in an autocorrelation-type experiment using an instantaneously responding nonlinear medium. As shown in FIG. 1, FROG involves splitting a pulse and then overlapping the two resulting pulses in an instantaneously responding X(3) or X(2) medium. Any medium that provides an instantaneous nonlinear interaction may be used to implement FROG. Perhaps the most intuitive is a medium and configuration that provides polarization gating.
For a typical polarization-gating configuration, induced birefringence due to the electronic Kerr effect is the nonlinear-optical interaction. The "gate" pulse causes the X(3) medium, which is placed between two crossed polarizers, to become slightly birefringent. The polarization of the "gated" probe pulse (which is cleaned up by passing through the first polarizer) is rotated slightly by the induced birefringence allowing some of the "gated" pulse to leak through the second polarizer. This is referred to as the signal. Because most of the signal emanates from the region of temporal overlap between the gate pulse and the probe pulse, the signal pulse contains the frequencies of the "gated" probe pulse within this overlap region. The signal is then spectrally resolved, and the signal intensity is measured as a function of wavelength and delay time τ. The resulting trace of intensity versus delay and frequency is a spectrogram, a time- and frequency-resolved transform that intuitively displays the time-dependent spectral information of a waveform.
The resulting spectrogram can be expressed as:
                                          S            E                                          (                          ω              ,              τ                        )                          =                                                                        ∫                                  -                  ∞                                ∞                                                                            E                                                        (                    t                    )                                                                                g                                                        (                                          t                      -                      τ                                        )                                                                                ⅇ                                                            -                      ⅈ                                                                                ω                                                                                t                                                                                ⅆ                  t                                                                          2                                    (        9        )            where E(t) is the measured pulse's electric field, g(t−τ) is the variable-delay gate pulse, and the subscript E on SE indicates the spectrogram's dependence on E(t). The gate pulse g(t) is usually somewhat shorter in length than the pulse to be measured, but not infinitely short. This is an important point: an infinitely short gate pulse yields only the intensity I(t) and conversely a CW gate yields only the spectrum I(ω). On the other hand, a finite-length gate pulse yields the spectrum of all of the finite pulse segments with duration equal to that of the gate. While the phase information remains lacking in each of these short-time spectra, having spectra of an infinitely large set of pulse segments compensates this deficiency. The spectrogram nearly uniquely determines both the intensity I(t) and phase φ(t) of the pulse, even if the gate pulse is longer than the pulse to be measured. If the gate is too long, sensitivity to noise and other practical problems arise.
In FROG, when using optically induced birefringence as the nonlinear effect, the signal pulse is given by:Esig(t,τ)∝E(t)|E(t−τ)|2  (10)So the measured signal intensity IFROG(ω,τ), after the spectrometer is:
                                          I                          F                                                                                R                                                                                O                                                                                G                                                      (                          ω              ,              τ                        )                          =                                                                        ∫                                  -                  ∞                                ∞                                                          E                                                  (                  t                  )                                                                                                E                                      (                              t                -                τ                            )                                                                                                2                                                                            ⅇ                                                            -                      ⅈ                                                                                ω                                                                                t                                                                                ⅆ                  t                                                                    2                                              (        11        )            The FROG trace is thus a spectrogram of the pulse E(t) although the gate, |E(t−τ)|2, is a function of the pulse itself.
FROG is not limited to the optical Kerr effect. Second harmonic generation (SHG) FROG can be constructed to analyze relatively weak pulses from oscillators and is typically more sensitive than polarization-gating FROG. For SHG FROG, the pulse is combined with a replica of itself in an SHG crystal as illustrated in FIG. 2.
To see that the FROG trace essentially uniquely determines E(t) for an arbitrary pulse, note that E(t) is easily obtained from Esig(t, τ). Equation (11) can then be written in terms of Esig(t, Ω), the Fourier transform of the signal field Esig(t, τ) with respect to delay variable τ. This gives the following, apparently more complex, expression:
                                          I                          F                                                                                R                                                                                O                                                                                G                                                      (                          ω              ,              τ                        )                          =                                                                        ∫                                  -                  ∞                                ∞                                                                                E                    sig                                                                          (                                          t                      ,                      Ω                                        )                                                                                ⅇ                                                                                -                          ⅈ                                                                                ω                                                                                t                                            -                                              ⅈ                                                                                Ω                                                                                τ                                                              )                                                                                ⅆ                  t                                                                  ⅆ                  Ω                                                                          2                                    (        12        )            Equation (12) indicates that the problem of inverting the FROG trace IFROG(ω, τ) to find the desired quantity Esig(t, τ) is that of inverting the squared magnitude of the two-dimensional (2-D) Fourier transform of Esig(t, τ). This problem, which is called the 2-D phase-retrieval problem, is well known in many fields, especially in astronomy, where the squared magnitude of the Fourier transform of a 2-D image is often measured. At first glance, this problem appears unsolvable; after all, much information is lost when the magnitude is taken. The 1-D phase retrieval problem is known to be unsolvable (for example, infinitely many pulse fields give rise to the same spectrum). Intuition fails badly in this case, however. Two- and higher-dimension phase retrieval processes essentially always yield unique results.
FROG Inversion
An iterative 2-D phase retrieval process is used to extract the pulse information from the measured FROG trace as illustrated generally in FIG. 3. This phase retrieval process converges to a pulse that minimizes the difference between the measured and the calculated FROG trace. Application of this phase retrieval process to FROG has been problematic in the past. Some recent applications use a generalized projections algorithm, which converges quickly, along with faster computers, to track pulse changes at rates of 20 Hz or greater, making FROG a real-time pulse measurement technique. Indeed, programs for analyzing FROG traces are commercially available.
The original FROG inversion process, using what is commonly referred to as the vanilla algorithm, is simple and iterates quickly. On the other hand, the process tends to stagnate and give erroneous results, especially for geometries that use a complex gate function such as SHG or self-diffraction. Improved strategies using different algorithms, including brute force minimization, were developed to avoid stagnation, at the expense of both iteration speed and convergence time. Later a numerical technique called generalized projections brought a significant advance in both speed and stability. The generalized projections technique proceeds after an iteration by constructing a projection that minimizes the error (distance) between the FROG electric field, Esig(t, τ), obtained immediately after the application of the intensity constraint, and the FROG electric field calculated from the mathematical form constraint. The projection constructed in this manner is used as the starting point of the next iteration.
The first implementations of the generalized projections technique used a standard minimization procedure to find the electric field for the next iteration (which can still be slow). For the most common FROG geometries, PG and SHG, there are substantial advantages to a different strategy that directly determines the starting point for the next iteration. This strategy, called Principal Components Generalized Projections (PCGP), converts the generalized projections technique into an eigenvector problem. The PCGP technique has achieved pulse characterization rates of 20 Hz.
The goal of phase retrieval is to find the E(t) that satisfies two constraints. The first constraint is the FROG trace itself which is the magnitude squared of the 1D Fourier transform of Esig(t,τ):
                                          I                          F                                                                                R                                                                                O                                                                                G                                                      (                          ω              ,              τ                        )                          =                                                                        ∫                                  -                  ∞                                ∞                                                                                E                    sig                                                                          (                                          t                      ,                      τ                                        )                                                                                ⅇ                                                            -                      ⅈ                                                                                ω                                                                                t                                                                                ⅆ                  t                                                                          2                                    (        13        )            The other constraint is the mathematical form of the signal field, Esig(t,τ), for the nonlinear interaction used. The mathematical forms for a variety of FROG beam geometries are:
                                          E            sig                                          (                          t              ,              τ                        )                          ∝                  {                                                                                E                                                              (                      t                      )                                                                                                                              E                                                                          (                                                      t                            -                            τ                                                    )                                                                                                            2                                                                                P                                                                                G                                                                                F                                                                                R                                                                                O                                                                                G                                                                                E                                                                    (                        t                        )                                                              2                                                                                E                      *                                                                                (                                              t                        -                        τ                                            )                                                                                S                                                                                D                                                                                F                                                                                R                                                                                O                                                                                G                                                                                E                                                              (                      t                      )                                                                                E                                                              (                                              t                        -                        τ                                            )                                                                                S                                                                                H                                                                                G                                                                                F                                                                                R                                                                                O                                                                                G                                                                                E                                                                    (                        t                        )                                                              2                                                                          E                                                              (                                              t                        -                        τ                                            )                                                                                T                                                                                H                                                                                G                                                                                F                                                                                R                                                                                O                                                                                G                                                              }                                    (        14        )            where PG is polarization gate, SD is self-diffraction, SHG is second harmonic generation and THG is third harmonic generation FROG.
All FROG implementations work by iterating between two different data sets: the set of all signal fields that satisfy the data constraint, IFROG(ω, τ), and the set of all signal fields that satisfy equation 14. The difference between FROG implementations is how the iteration between the two sets is completed. In the case of generalized projections, the E(t)'s are chosen such that the distance between the E(t) on the magnitude set and the E(t) on the mathematical form set is minimized. This is accomplished by minimizing the following equation:
                    Z        =                              ∑                          i              ,                              j                =                1                                      N                                                                                                                            E                                          sig                                                                    (                                                  D                                                                                C                                                )                                                                                (                      k                      )                                                                                (                                                                  t                        i                                            ,                                              τ                        j                                                              )                                                  -                                                      E                                          sig                                                                    (                        MF                        )                                                                                (                                              k                        +                        1                                            )                                                                                (                                                                  t                        i                                            ,                                              τ                        j                                                              )                                                                                      2                                              (        15        )            where Esig(DC)(k)(ti,τj) is the signal field generated by the data constraint, and Esig(MF)(k+1)(ti,τj) is the signal field produced from one of the beam geometry equations 14. For the normal generalized projections technique, the minimization proceeds using a standard steepest decent algorithm; the derivative of Z with respect to the signal field is computed to determine the direction of the minimum. The computation of the derivatives is tedious; the derivatives are tabulated in, for example, Trebino, et al., Rev. Sci Instrum., 68, p. 3277 (1997).
An alternative to the minimization of equation 15 is principal components generalized projections (PCGP). PCGP computes the starting point of the next iteration through an eigenvector problem, reducing the computation for the next iteration to simple matrix-vector multiplies. PCGP works for both the PG and SHG beam geometries, is simple to program and is fast, as described in D. J. Kane, IEEE J. Quant. Elec., (1999).
Self Checks in FROG Measurements
Unlike other pulse measurement techniques, FROG can provide a great deal of feedback about both the quality of the measurement (systematic errors) and the quality of the technique's performance. A good check for convergence is the FROG trace error together with a visual comparison between the retrieved FROG trace and the measured FROG trace. The FROG trace error is given by:
                    G        =                                            1                              N                2                                                                                ∑                                  i                  ,                                      j                    =                    1                                                  N                                                                                                                                                            I                                                  F                                                                                R                                                                                O                                                                                G                                                                                (                                                                              ω                            i                                                    ,                                                      τ                            j                                                                          )                                                              -                                          α                                                                                I                                                      F                                                                                R                                                                                O                                                                                G                                                                                (                            k                            )                                                                                (                                                                                ω                              i                                                        ,                                                          τ                              j                                                                                )                                                                                                                2                                                                        (        16        )            where α is a renormalization constant, IFROG is the measured FROG trace and IFROG(k)(ω, τ) is computed from the retrieved electric field. Typically, the FROG trace error of a PG measurement should be less than 2% for a 64×64 pixel trace, while the FROG trace error of a 64×64 pixel SHG FROG trace should be about 1% or less. Acceptable FROG trace errors decrease as FROG trace size increases and increase for smaller FROG traces. These values are only guidelines and variations are expected in specific circumstances. For example, acceptable phase retrievals of large and very complicated FROG traces can produce larger FROG trace errors.
In a good FROG measurement, the spectrum of the retrieved pulse should faithfully reproduce the salient features of the pulse's measured spectrum. SHG FROG even provides an additional check called the frequency marginal. The sum of an SHG FROG trace along the time axis yields the autoconvolution of the pulse's spectrum providing two ways the FROG measurement can be checked. First, the autoconvolution of an independently measured spectrum can be compared to the sum of the FROG trace along the time axis (the frequency marginal) providing an indication of how well the measurement was made. For example, if the doubling crystal was too thick in the pulse measurement, the FROG trace's frequency marginal will be narrower than the autoconvolution of the measured spectrum. Second, comparing the autoconvolution of the retrieval pulse spectrum with the FROG trace marginal can provide a test of convergence in addition to a test of the measurement. Phase matching problems show up as a mismatch between the FROG trace marginal and the autoconvolution of the retrieved spectrum.
Because FROG is a spectrally resolved autocorrelation, summing any FROG trace along the frequency axis yields the autocorrelation of the measured pulse. This autocorrelation can be compared to an independently measured autocorrelation, or a comparison can be made between the frequency sum of the FROG trace and the autocorrelation calculated from the retrieved pulse to determine algorithm convergence and the quality of the measurement.
Making FROG Real-Time
In order to make FROG a real-time pulse measurement technique, the FROG device, the data acquisition, the 2-D phase retrieval process, and the user interface are integrated. One real-time ultrashort laser pulse measurement implementation combined a multishot FROG device with two digital signal processing (DSP) boards in a fast personal computer. This implementation is described in D. J. Kane, IEEE J. Quant. Elec., (1999). One DSP card was devoted to data acquisition and the other DSP card was used to make the calculations. Briefly, as an optical delay line was scanned, the signal from the nonlinear interaction was spectrally resolved. Thus, temporal slices of the FROG trace, or spectrogram were obtained. Appropriate (sequential) ordering and assembly of the temporal slices formed the FROG trace.
The DSP implementation was considered expensive for some applications and so efforts were directed to develop a FROG-based ultrashort laser pulse measurement that did not use DSP cards. One implementation that did not use a DSP, called VideoFROG, used a standard CCD video camera to acquire the FROG trace and a standard computer frame grabber to digitize the video signal. The computer displayed the resulting FROG trace and resized and conditioned the FROG trace for the phase retrieval process. The computer also was the user interface, displaying the retrieved pulse and the raw video image. VideoFROG used the PCGP phase retrieval process, which generally converges well. Consequently, no provision was made to detect stagnation or other faults in the phase retrieval process.
Basic Controllers—Proportional Integral Derivative Control
The simplest and fastest controller is just the basic proportional controller. However, these most basic controllers are prone to calibration errors and offsets. The measured difference will not correspond to the actual difference and the pulse will never achieve the desired pulse shape—the system will have a steady-state error in the set point (Dorf and Bishop, Modern Control Systems, p. 193)
A simple calculation illustrates this effect. The schematic shown in FIG. 4 depicts a typical control loop. Suppose R(s) is the transfer function (Laplace transform) for the input to the pulse shaper, Y(s) is the transfer function for the output, E(s) is the transfer function for the error, Gc(s) is the controller transfer function, G(s) is the transfer function of the servo itself, and H(s) is the transfer function for the position encoder or sensor. The output of the system is given by (Dorf and Bishop, Modern Control Systems, p. 178-180):
                              Y                                (            s            )                          =                                                                              G                  c                                                                  (                  s                  )                                                                          G                                                  (                  s                  )                                                                    1              +                                                                    G                    c                                                                          (                    s                    )                                                                                G                                                        (                    s                    )                                                                                H                                                        (                    s                    )                                                                                R                                      (              s              )                                                          (        17        )            The error of the system, which is the difference between the measured pulse and the desired pulse (R(s)), is given by (Dorf and Bishop, Modern Control Systems, p. 183):
                              E                                (            s            )                          =                              1                          1              +                                                                    G                    c                                                                          (                    s                    )                                                                                G                                                        (                    s                    )                                                                                H                                                        (                    s                    )                                                                                R                                      (              s              )                                                          (        18        )            For purposes of an example, in the case of proportional control only, suppose we set Gc(s)=K, H(s)=1 and G(s)=1 (ideal case). Thus,
                              E                                (            s            )                          =                                            1                              1                +                K                                                                R                                            (                s                )                                              =                                    1                              1                +                K                                                                1              s                                                          (        19        )            when the input is a unit step function (R(s)=1/s). To calculate the steady state error, we use the final-value theorem (Dorf and Bishop, Modern Control Systems, p. 183) which is:
                                          lim                          t              ->              ∞                                                      (                          ⅇ                                            (                t                )                                      )                          =                              lim                          s              ->              0                                                      (                          sE                                            (                s                )                                      )                                              (        20        )            where E(s) is the Laplace transform of e(t). Thus, we find that the steady-state value of the error, e(t) is 1/(1+K).
Therefore, a simple proportional feedback control system is unacceptable for the level of precision and accuracy required by ultrafast laser pulse measurement.
One way to address all of the control issues is to include terms in addition to the proportional feedback term. For example, an integral term will remove steady-state errors. An integral controller has a transfer function of (Dorf and Bishop, Modern Control Systems, p. 703):
                                          G            c                                          (            s            )                          =                              K            p                    +                                    K              I                        s                                              (        21        )            where KI is the integral gain term. Setting G(s) and H(s) to 1 and solving for E(s), we get:
                              E                                (            s            )                          =                              s                                                            K                  p                                                s                            +                              K                I                                                                    1            s                                              (        22        )            Applying the final-value theorem as mentioned above, we find that as t→∞, the offset (error), e(t)→0. Unfortunately, the addition of an integral term does come at a cost—a reduction of the transient response and an increase in the overshoot. A derivative term is usually added to improve transient response, decrease overshoot, and decrease the settling time. Thus, the simplest form of a robust, high precision, high accuracy controller with good speed is the proportional-integral-derivative, or PID, controller (FIG. 5):

Real-time measurement of ultrashort laser pulses


An apparatus and method for serving beer at proper CO.sub.2 content and proper temperature for best taste, comprises a draft tower for distributing beer at the proper gas content and at the proper temperature. A plurality of beer barrels connected in series are in turn connected to the draft tower by a pipeline. A CO.sub.2 gas cylinder is connected to the barrels for supplying gas at an elevated pressure sufficient to move the beer from the barrels along the pipeline to the tower. A heater is provided for heating the barrel to an elevated temperature above the proper temperature for taste, so that the gas content in the barrel does not rise above the proper gas content under the influence of possibly elevated pressure needed to convey the beer. Near or at the draft tower, the beer is recooled to the proper temperature and thus served at a proper temperature and with the proper gas content.

The present invention relates in general to a beer serving system, and more particularly, to a system of serving draft-beer from a barrel through draft towers into jugs, mugs, glasses or the like. Specifically, the invention concerns improvements in the extensibility of beer supplying pipelines in the system, which can be installed in a beer hall, bar, restaurant or the like, while maintaining the taste of the beer.
A system is known, for example, which includes draft towers for dispensing beer, a beer barrel, pipelines connecting the draft towers to the beer barrel, and a gas cylinder connected to the beer barrel for pressurizing the barrel to force the beer through the pipelines and out to the draft towers.
In such a system, the beer at 0.5% by weight CO.sub.2 content in the barrel is cooled down to about 5.degree. C., which is believed to be the best gas content and temperature for taste, and at which to drink the beer. The beer in the barrel is subjected to a pressure which is a little higher than 1.0 atmosphere, namely 1.2-1.5 atmosphere, by means of a carbonic acid gas cylinder, for forcing the beer through the pipelines. If the pressure of the gas cylinder is higher than approximately 1.5 atmospheres and this pressure is applied to the surface of the beer for a long time, too much CO.sub.2 will dissolve into the beer and this will increase its CO.sub.2 content to more than 0.5%. If the beer is warmed to higher than 5.degree. C. under such pressurization, the ideal gas content may be maintained. This cannot be done however, without degrading the beer's taste due to the higher than ideal temperature for the beer.
A pressure of 1.5 atmospheres, however, cannot force the beer very far along the pipelines. The extension of the pipelines between the beer barrel or barrels, and the draft towers, thus, is at most about 2 meters for the known system.
Because of such short supply lines, the prior art systems rely on manpower to carry the beer from the draft towers to the customers. Waitresses and waiters must pass among chairs and tables in bars, beer halls and restaurants having such systems, while carrying the beer in heavy jugs filled at the draft towers. The known systems thus lack efficient beer supplying ability.


Systems, apparatus, and methods are disclosed that include a divided exhaust engine with at least one pair of primary EGR cylinders and a plurality of pairs of non-primary EGR cylinders. The pair of primary EGR cylinders can be connected to an intake with an EGR system that lacks an EGR cooler. In another embodiment, the cylinder pairs include exhaust flow paths that join in the cylinder head to form a common exhaust outlet for each cylinder pair in the cylinder head that is connected directly to the EGR system or to the exhaust system without an exhaust manifold.

Engines operating with one or more cylinders as dedicated exhaust gas recirculation (EGR) cylinders enjoy greatly simplified controls and pressure management, fewer hardware devices, and other benefits. However, these simplifications come at the cost of a loss of control over the system. For example, cylinder deactivation in turbocharged systems that include dedicated EGR can affect the exhaust flow through the turbocharger and the distribution of EGR flow into the intake flow. In addition, twin turbocharger arrangements require complicated plumbing arrangements, additional components, and high temperature capable exhaust manifolds, increasing cost and complexity of these systems. Therefore, further technological developments are desirable in this area.

This patent is subject to a terminal disclaimer.


An engine air-fuel ratio control unit is so constructed that, whenever any abnormal operation is detected in one of O.sub.2 sensors, the control unit makes corrections of a fuel amount injected from an injector on the basis of the output information from the other O.sub.2 sensor performing its normal operation and learned values obtained from both of the O.sub.2 sensors when both performed their normal operations.

The present invention relates to an engine air-fuel ratio control unit which is designed to determine a basic fuel amount for controlling the engine to a prescribed air-fuel ratio by arithmetic operations performed on the basis of the information of the intake air amount of the engine and also to correct the signal of the amount of fuel to be fed into the injector, in such a manner as to operate the engine at a theoretical air-fuel ratio, on the basis of the output information from O.sub.2 sensors installed respectively on the left and right exhaust banks.
In general, the system to which this type of engine air-fuel ratio control unit is applied, is constructed as illustrated in FIG. 3, in which reference number 1 indicates an engine, reference number 2 indicates an air flow sensor, reference number 3 indicates a throttle valve, reference number 8 indicates an air-fuel ratio control unit, and reference number 9 indicates a revolution sensor which detects the revolutions of the engine 1. Moreover, as the exhaust system is divided between the two banks, namely, the left bank and the right bank, O.sub.2 sensors and component parts mentioned in the following are provided each on the left side and the right side. That is, reference number 4 indicates an O.sub.2 sensor (right), which performs the detection of the exhaust gas, and reference number 5 indicates an O.sub.2 sensor (left), which similarly performs the detection of the exhaust gas. Reference number 6 indicates an injector (right), which performs the injection of the fuel, and reference number 7 indicates an injector (left), which similarly performs the injection of the fuel. Reference number 10 indicates a ternary catalytic converter (right), and reference number 11 indicates a ternary catalytic converter (left).
Moreover, FIG. 4 shows a detailed block construction of the air-fuel ratio control unit 8 shown in the construction drawing of the engine control system in FIG. 3. In FIG. 4, reference number 20 indicates a basic fuel amount calculating means for calculating the basic fuel amount on the basis of the detected amount of an intake air, reference numbers 21 and 22 indicate an A/F feedback correction means, which makes corrections of the air-fuel ratio feedback on the basis of the detected output information from the O.sub.2 sensors, and reference number 23 indicates an A/F feedback determining means, which performs control by determining whether the basic fuel amount is to be fed into the injector (right) 6 and the injector (left) 7, respectively, or whether a corrected amount of the fuel as determined by the two systems of the air-fuel feedback correction means 21 and 22 is to be fed into the injectors.
FIG. 5 shows a timing chart illustrating the relationship between the output information from these O.sub.2 sensors and the output time widths of the injectors 6 and 7, which are installed respectively on the left bank and the right bank, in case the individual O.sub.2 sensors are in their normal state. That is to say, FIG. 5(a) shows the waveform of the output from the O.sub.2 sensor (right) 4, and FIG. 5(b) shows the time duration of fuel injection from the injector (right) 6, which corresponds to the above waveform. As shown in these charts, the air-fuel feedback correction means 21 makes a correction in such a manner as to reduce the amount of the fuel fed, when the signal from the O.sub.2 sensor (right) 4 increases and rises above the threshold value voltage V.sub.1, which corresponds to the theoretical air-fuel ratio, and, as the result of this correction, the time T for fuel injection (right) from the injector (right) 6 is shortened. Also, when the output from the O.sub.2 sensor (right) 4 decreases and falls down below the threshold value voltage V.sub.1, the air-fuel feedback correction means 21 makes a correction in such a manner as to increase the amount of the fuel, and, as the result of this correction, the time T for fuel injection (right) from the injector (right) 6 is extended.
In reflection of these results, the waveform of the time T for fuel injection (right) will be such a waveform in amplitude fluctuating upward and downward with respect to the mean value T (right) (central value: the duration of time corresponding to the theoretical air-fuel ratio). Then, the deviations of this amount of feedback correction from the mean value T (right) are constantly renewed and stored in a memory (learning function), and, when the O.sub.2 sensor (right) 4 becomes in any abnormal state, the feedback correction is made on the basis of the corrected value (learned value) thus stored in the memory.
Also, the timing relationship between the waveform of the output from the O.sub.2 sensor (left) illustrated in FIG. 5 (c) and the time for fuel injection (left) from the injector (left) 7 illustrated in FIG. 5(d) shows a transition similar to what is described above.
Generally, the ternary catalytic converters will attain the maximum efficiency in their purification of exhaust gas when the air-fuel ratio A/F is 14.7 (the theoretical air-fuel ratio), and their purifying efficiency will be kept at a favorable level by the O.sub.2 storage effect if control is performed on the correction of the fuel amount by increasing and decreasing it in a prescribed cycle with reference to the line of the value 14.7 of the air-fuel ratio. On the contrary, the purifying efficiency will become extremely low in case the air-fuel ratio deviates from the proximity of the value 14.7 of the air-fuel ratio or in case control is not performed on the correction of the fuel amount by having it fluctuate upward and downward in relation to the line of the value 14.7 of the air-fuel ratio. In case one of the O.sub.2 sensors has a failure, the conventional air-fuel control unit for engine corrects the amount of the fuel for the bank where the failure has occurred by arithmetic operations performed on the basis of the value learned at the time when the failing O.sub.2 sensor was in a normal state, and consequently the corrected value will be a certain fixed value. As the result, the conventional unit presents the problem that it is not capable of correcting the amount of the fuel by moving it upward and downward in a prescribed cycle in relation to the line of the value 14.7 of the air-fuel ratio and consequently that it is incapable of effectively purifying the exhaust gas. Additionally, in case a deviation or the like has occurred in the learned value, the conventional unit fails to make any sufficient correction of the amount of the fuel, so that the ternary catalytic converters cannot be utilized effectively.


A rotating electric machine includes a stator core of a cylindrical shape, a rotor core coaxially rotating inside the stator core, and a plurality of coils each formed by winding magnet wires using a slot formed axially in either the stator core or the rotor core, or both: it is preferable that each of the coils be split into a plurality of coil groups, that starting sections of winding of the magnet wires in one of the coil groups be wound adjacently to ending sections of winding of the magnet wires in another of the coil groups, and that the starting sections and ending sections of winding of the magnet wires in the two coil groups be connected at respective outgoing wire ends to one another, outside the slots.

In recent years, adjustable-speed operation of the motors (rotating electric machines) each using an inverter is prevalent in terms of energy saving. However, when a motor is driven by an inverter that generates a rectangular-wave voltage, if a steep-fronted surge voltage is generated by the inverter and causes a high voltage between the motor coil turns, this could result in insulation deterioration between the coil turns (refer to Non-Patent Reference 1).
Regarding this turn-to-turn insulation deterioration problem, Non-Patent Reference 2 proposes a method of strengthening the turn-to-turn insulation by, for example, increasing the thickness thereof in addition to reducing the number of turns of the lead wire forming a first coil to which the surge voltage from the inverter is applied. However, as described in Non-Patent Reference 2, insulation strengthening has had its limits since the rotating electric machine changes in mechanical output characteristics.
Also, Non-Patent Reference 3 discloses designing a rotating electric machine for reduced capacitance between a turn and a core, against the capacitance between turns, in order to improve the turn-to-turn insulation distribution.
In addition, Patent Reference 1 proposes a method of alleviating a turn-to-turn voltage share in a high-voltage rotating electric machine by connecting a capacitor for adjusting distributed capacitance between turns, from the external section of a winding.
[Patent Reference 1: Japanese Patent Laid-open No. Sho 50(1975)-301 (JP-A-1975-301), What is Claimed is, FIGS. 3, 6]
[Non-Patent Reference 1: IEEJ Technical Report, Vol. 739, pp. 14-20, issued in August 1999] (IEEJ: Institute of Electrical Engineers of Japan)
[Non-Patent Reference 2: "Characterization Tests on Induction Motor with Reduced Number of Terminal Coil Turns", material for technical presentation in the IEEJ Rotating Machines Working Group, RM-00-92 to RM-00-109, p. 19, Hitosugi Kazuyoshi, Shoji Moriyasu]
[Non-Patent Reference 3: "Modern High-Voltage Engineering", pp. 91-93, written by Uchida Masayuki, published by Ohmsha, Ltd.]

Rotating electric machine, winding machine, and rotating electric machine system


An uncatalyzed liquid phase process is provided for the preparation of oligomer free 1233zd via the fluorination of 1230za with a high molar excess of HF. The 1233zd is a known intermediate useful for preparing 245fa.

This invention relates to preparation of an essentially oligomer free (hereinafter simply "oligomer free") 1,1,1-trifluoro-3-chloro-2-propene (1233zd) by the uncatalyzed liquid phase fluorination of 1,1,3,3-tetrachloro-2-propene (1230za), particularly to such processes wherein said 1230za is contacted with a large molar excess of hydrogen fluoride (for convenience hereafter referred to as "HF"). The 1233zd product is a known intermediate for producing 245fa, as taught, for example, in U.S. Pat. No. 5,616,819 and in copending application 08/980,747, filed Dec. 1, 1997.
U.S. Pat. No. 5,616,819 discloses that previous attempts to fluorinate 1230za to 1233zd in an uncatalyzed liquid phase reaction resulted in the formation of a substantial amount of oligomeric products (19 g of oligomeric material in Example 1.i., the result of using a 6:1 molar ratio of the HF and 1230za starting materials). It is thus an object of this invention to provide an uncatalyzed liquid phase fluorination process for successfully converting 1230za to oligomer free 1233zd.

Uncatalyzed liquid phase fluorination of 1230ZA


The continuous passive motion exercise system enables human's joints to recover speedily to original functioning after injuries or surgeries, and thus shortens the period of time needed for joint rehabilitation. The present invention provides a force or torque monitoring device attached onto a continuous passive motion exercise mechanism to measure the driving force of the repeated joint flexing and extending motion, so as to evaluate the change of the viscosity and the stiffness of the injured joints through different rehabilitation periods. This monitoring device also monitor degrees of joint muscle's active contraction and thus slows down or stops the repeated motions of exercise mechanisms to improve safety concerns. In addition, such a monitoring device also includes a data transceiver interface utilized for transmitting and receiving the information regarding a patient's states of using such a exercise system in order to assess the patient and to provide doctors with the basis for evaluating and improving the rehabilitation condition of the injured joints.

1. Field of the Invention
The present invention relates to a patient rehabilitation system, more particularly, a rehabilitation system utilized for rehabilitation and physical therapy of injured limbs and joints. To elaborate further, the present invention is a continuous passive motion exercise system with a driven monitoring device, utilized for reflecting on the state changes of the adhesiveness and the stiffness of patients' injured joints for monitoring their rehabilitation conditions, as well as for controlling the rehabilitation mechanism safely.
2. Description of Related Arts
Limb joints, such as knee joints, hip joints, elbow joints and wrist joints are all crucial mechanisms for humans to conduct various activities, including walking, running, jumping, standing, stepping up and down of stairs, dancing and track-and-field activities that require twists, turns or weight-carrying of joints. In view of the dynamics of a human body, since joints have to sustain most of the weight loading during bodily activities or motions, joints are susceptible to injuries, and abrasions with age, especially for knee and hip joints that carry most of the body weight. After being injured or aged, such joints sometimes have to be cured by replacing with artificial joints, so as to resume original functions.
After surgeries, a long period of rehabilitation therapy soon follows for fully recovery, which might take from a few months to half a year, and thus causes patients great inconvenience. One of the rehabilitation therapy is to maintain continuous exercise of joints to avoid adhesiveness among tendons, and to supply nutrition to joints. In such ways, this procedure enables joints to resume the normal range of motion as soon as possible, and thus shortens the period of time for patients to stay in hospitals or in beds. Therefore, clinically, the continuous passive motion exercise systems are widely utilized in rehabilitation procedures after patients with surgeries on joints.
The continuous passive motion exercise systems currently employed cannot provide information regarding the patients' conditions of rehabilitation such as their progress and the effect of their therapy. Thus, doctors or professional rehabilitation personnel may only rely upon past experience to make decisions to adjust rehabilitation procedures to befit each patient. General clinical rehabilitation procedures usually employ simple and generic processes for all, not being able to vary according to individual states. Furthermore, since continuous passive motion exercise systems engages patients' injured joints and move reciprocatively, to certain joint flexing or extension angles during exercises, patients might feel pain. Such exercise systems, currently available without being able to automatically adjust properly to suit patients' needs, can only be manually stopped by patients themselves. The pain caused by overbending, overextending or speeding can only be inspected and relieved afterwards. Such a drawback might cause secondary injury to patients.
In addition, joints with general joint diseases such as osteoarthritis, after surgeries, are likely to be stiff and cannot act well. Current exercise systems are not able to provide proper indices showing states of stiffness for joints, as well as indices for the evaluation of joint rehabilitation. Therefore, during rehabilitation, precisely and effectively monitoring and recording the degree of stiffness and viscosity for joints not only avoid any ill operations of exercise machines so as to protect patients from further injuries, but also provide doctors and patients with more rehabilitation information, thus, to enable more effective and speedy therapies.

Continuous passive motion exercise system with driven monitoring

Industrial Technology Research Institute


A scanning exposure apparatus for scan-exposing the image of a mask pattern on a mask onto a substrate by synchronously moving the mask and the substrate with respect to an exposing radiation flux is provided. The scanning exposure apparatus includes a movable mask stage for holding the mask, a movable substrate stage for holding the substrate, a first detector outputting a first signal indicating the position of the substrate stage, and a second detector outputting a second signal indicating the position of the mask stage. The scanning exposure apparatus further includes a controller processing the first signal to derive a target position of the mask stage, the controller receiving the second signal to derive a deviation of the mask stage from the target position, the controller further deriving an error anticipation signal in accordance with deviations in mask stage movement measured in advance of scan-exposure, and a mask stage controller for moving the mask stage in accordance with the derived deviation of the mask stage from the target position and the error anticipation signal to maintain the position of the mask stage to the target position.

1. Field of the Invention
The present invention relates to an exposure apparatus for projecting a mask pattern onto a photosensitive substrate in photolithography processes in the manufacture of semiconductor devices, image pickup devices (CCDs), liquid crystal display devices, and thin film magnetic heads, etc. More specifically, the present invention relates to a scanning type exposure apparatus, such as a projection exposure apparatus of a step-and-scan system, which synchronously scans a mask and a photosensitive substrate using a projection optical system.
2. Discussion of the Related Art
In the manufacture of semiconductor devices, for example, a reduction projection type exposure apparatus (stepper) of step-and-repeat type (one-time exposure type) has often been used to transcribe a pattern on a mask (reticle) onto each shot region of a wafer coated with a photoresist. However, in response to the recent demand for transcribing circuit patterns on a larger area with higher accuracy without placing a heavy burden on the projection optical system, a so-called step-and-scan type projection exposure apparatus has been developed. The step-and-scan type projection exposure apparatus successively transcribes the image of the pattern on the reticule onto each shot region on the wafer by synchronously moving (scanning) the reticle and the wafer with respect to the projection optical system in such a way that a portion of the pattern on the reticle is successively projected on a wafer via the projection optical system.
The previously known aligner, which transfers the entire pattern of the reticle onto the entire wafer with a positive image of equal magnification by a single exposure operation using a single body type stage, is a prototype of the scanning type exposure apparatus. In contrast, in the step-and-scan system, since a reduction-type projection optical system is normally used, it is necessary to drive the reticle stage and wafer stage independently at a speed ratio proportional to the reduction ratio of the projection optical system. Also, since movement between shot regions is performed in a stepping mode, the mechanism of the stage system is complex and requires an extremely high level of control.
The following is one example of a method for controlling the stage system in a conventional projection exposure apparatus of step-and-scan type. After aligning the reticle stage with the wafer stage at the beginning of scanning, the wafer stage and the reticle stage are synchronously moved (scanned) at a predetermined respective speeds in predetermined directions. During such scanning operation, the positional deviations in the scanning direction and non-scanning direction (direction perpendicular to the scanning direction) of both stages are monitored using laser interferometers and the positional deviations thus measured are offset by a servo control system. In other words, in the conventional method, these positional deviations are corrected by feeding back the measured positional deviations in a closed feedback loop.
As described above, in the conventional step-and-scan type projection exposure apparatus, the positional deviations in the scanning direction and non-scanning direction of the reticle stage and wafer stage are reduced by feeding back the positional deviations to the servo control system during scanning exposure. However, since there is a limit in improving the response speed of this type of feedback system operating in response to the positional deviation amount, some follow-up errors (positional deviation) remain.
More specifically, even if the scanning directions of the wafer stage and the reticle stage are designed to be parallel to each other, some angular errors may exist between these scanning directions due to mechanical assembly errors, or the like. When such angular errors exist, a positional deviation of the reticle stage relative to the wafer stage is generated in the non-scanning direction during scanning exposure. Using feedback control, a force is generated to correct this positional deviation. Nevertheless, since there is a limit in the response speed of the feedback system, a certain degree of follow-up error will always exist in the non-scanning direction.
One way to reduce the follow-up errors of this type is to increase the gain of the feedback system. However, if the gain of the feedback system is increased, mechanical resonance can be easily excited in the stage system, and accordingly the stage system may vibrate during the scanning exposure.

c/o Nikon Corporation (Intellectual Property Headquarters)Fuji Building, 2-3 Marunouchi 3-chome


An augmentation system for an implantable bone prosthesis, such as a femoral component of a knee joint prosthesis, includes an augmenting block having an aperture disposed therein. A collet having an expandable distal end, extending beyond one surface of the augmenting block, is positioned within the aperture. The proximal end of the collet includes a threaded cavity that receives a set screw which is adapted to communicate with the collet to expand and contract the collet. The distal end of the collet is adapted to fit within a mounting cavity of a joint prosthesis. With the augmenting block placed on the appropriate surface of the joint prosthesis, the distal end of the collet is positioned within the mounting cavity. Tightening of the set screw causes the distal end of the collet to expand, thus frictionally and/or mechanically engaging the interior walls of the mounting cavity to firmly secure the augmenting system to the prosthesis.

This invention relates to devices useful to augment bone deficiencies encountered during joint arthroplasty procedures.
Joint arthroplasty is a well known surgical procedure by which a diseased and/or natural joint is replaced by a prosthetic joint. Joint arthroplasty is commonly performed for knees, hips, elbows, and other joints. In some instances, due to disease or trauma, insufficient healthy boney mass exists at the distal end of a bone to which a joint prosthesis is to be secured. In arthroplasty procedures, it is often necessary to remove a portion of bone (such as a portion of the femur) to ensure sufficient anchoring and a proper fit for a prosthesis. In the example of knee joint arthroplasty, it is necessary to augment the inferior portion of the femoral component of the prosthesis to add additional thickness to the prosthesis to compensate for the lack of sufficient boney tissue.
Augmentation of the boney mass can be accomplished by securing augmenting devices to a component of the joint prosthesis, such as a femoral component. Some earlier devices used to augment knee femoral components must be cemented to the inferior surface of the femoral component. The techniques used to install such devices require accurate measurements and a great deal of precision since the augmenting device is difficult, if not impossible, to remove once it is secured to the prosthesis. These techniques also tend to be rather time consuming.
An improved augmentation device is described in U.S. Pat. No. 4,936,847 (Manginelli). This patent discloses an augmentation system that can be removeably and replaceably secured within a femoral component of a knee joint prosthesis. Such a design is advantageous because it avoids the need to cement augmenting devices to the inferior surface of the femoral component. As a result, augmenting devices of varying thicknesses can be fitted into place on the inferior surface of the femoral component. The augmenting devices can be removed and replaced with devices having different dimensions until the proper augmentation thickness is determined. Such an augmentation system, however, requires a specially constructed femoral component.
Despite existing augmentation systems, there remains a need to provide improved systems that allow standard augmenting components to be easily and securely affixed within a prosthesis component. The cost of joint prostheses that utilize many current augmentation systems can be quite high because the tight tolerances required often result in a high scrap rate. There is thus a need for an augmentation system that allows augmenting devices to be easily and securely fitted to prosthesis components.
It is thus an object of the invention to provide a prosthesis augmentation system that securely and easily fits within a component of a joint prosthesis. A further object is to provide an augmentation system that offers more cost effective manufacturing potential as well as a reduced part rejection rate. Another object is to provide an augmentation system that is well suited for use with femoral components of knee joint prostheses. It is also an object to provide an augmentation system that offers a great deal of versatility in that it can be used with different types of femoral components, including cruciate retaining and cruciate sacrificing femoral components. A further object is to provide an augmentation system that can be used in a prosthesis component without the need for post-casting modification of the prosthesis. Other objects will be apparent to those of ordinary skill in the art upon reading the description that follows.

Augmentation device for joint prosthesis


A surround-vision display system with a very high visual dynamic range is made possible by distributing a limited number of LED's on the inside of a drum and then spinning that drum around a user. The pixel information for each horizontal position in space is sent to each corresponding LED it visits that position. The LED's are arranged in a grid on a panel tile, and the panel tile is tilted slightly, e.g., at 1.1-degrees. The result is each panel tile presents a continuous vertical stripe in the picture frame as all its LED's are swept by in the drum motion. Several panel tiles stacked vertically inside the drum all contribute to the whole height of the picture frame, e.g., several feet. The entire inside circumference of the drum is populated with the LED panel tiles to keep frame refresh rates up to avoid flicker while keeping drum rotation speeds down to reasonable levels. Thus even though the LED's and drum are moving, the image projected appears to be relatively stationary.

Display devices are commonly characterized in terms of their spatial resolution, their dynamic range and their color range. In addition, there are several other important aspects of a display system that contribute to the realism of the presented image, for example artifacts stemming from a limited refresh rate, suppression of reflection of ambient light sources, and problems due to the fact that displays generally produce two-dimensional projections of three-dimensional objects. The visual acuity of the human eye exceeds the capability of conventional displays in all of these measures, and as such poses a tough challenge for display technologies aimed at photo realistic image and movie reproduction.
For example, standard cathode ray tube (CRT) displays fail to reach their 72 dpi ratings. The limited video bandwidth and large electron spot size in CRT's lead to much lower actual resolution. LCD display panels can reach their rated spatial resolutions, but doubling resolution it is problematic because smaller thin film transistors for each pixel limit the light output possible.
The human eye can adapt to a very wide range of lighting conditions. The range of interest for display devices is the lower part of photopic vision range, which provides good color perception and good spatial resolution. Because of the long time constants involved with adaptation, the simultaneous dynamic range of the human eye defines the desirable dynamic range of a display device, which is about 3.5 orders of magnitude.
Conventional displays feature a relatively small dynamic range. For example, the dynamic range of a typical CRT display is limited by many factors, including the limited bandwidth of the video drivers, optical reflections in the front faceplate, large electron spot size, and scattering from the shadow mask, for example. Projection systems suffer from the fact that the required objective lens itself has a limited dynamic range due to imperfect anti-reflection coatings. The net result is that most display systems achieve, at most, about two orders of magnitude less than what would be required to match the human eye.
Practical color display devices pick a set of three primary colors—red, green and blue—and are limited to reproducing the colors within a triangle defined by the color coordinates of these primary colors. The selection of the primary colors of a display device is subject to many technical limitations. For example, it is difficult to generate monochromatic light efficiently. If a white light source is used for the display, narrow band filters would waste a lot of light, hence projection and LCD displays tend to use primary colors quite far from the spectral boundary of human vision. Another factor concerning the primary blue and red colors is the diminishing sensitivity of the eye toward the end of the visible wavelength range.
In at least preferred embodiments, the present invention seeks to provide a surround-vision display system having one or more of improved spatial resolution, dynamic range, and color range than prior art display systems and circuitry suitable for use in such a surround-vision display system.

Hewlett-Packard Development Company, L.P.


A karaoke system is provided with a reproducing section and a remote control device. The reproducing section at least reproduces an accompaniment signal. The remote control device controls the operation of the reproducing section and is provided with an input section, a storage and a sending section. A control signal for controlling the operation of the reproducing section and information showing the order of reproducing an accompaniment signal by the reproducing section are input from the input section. The storage stores reservation information showing the order of reproduction and the personal information of a user both input via the input section. The sending section sends at least one of reservation information and the personal information of a user both stored in said storage to the reproducing section. The reproducing section reproduces an accompaniment signal specified via the input section and the operation of the reproducing section is controlled based upon at least one of personal information and a control signal both sent from the remote control device.

1. Field of the Invention
The present invention relates to a remote control device of a sound-reproducing system, the sound-reproducing system and a karaoke system described later. Particularly, the present invention relates to a remote control device of a sound-reproducing system provided with a storage, the sound-reproducing system and a karaoke system.
2. Description of the Related Art
At present, so-called karaoke for singing to a reproduced piece of music is popular. A karaoke system is provided with a mixing function for mixing and outputting a reproduced piece of music, that is, an accompaniment and a singer's voice input from a microphone. This karaoke system is classified into a system in which a signal of an accompaniment is reproduced using, for example, a so-called optical video disc, a compact disc (CD) or a disc recording medium such as a so-called video CD in which video data is recorded together with audio data, and into a so-called communication karaoke system which is a system for receiving and storing musical instrument digital interface (MIDI) data sent via a telephone line or a telecommunication line such as an integrated services digital network (ISDN) and reproducing MIDI data according to a selected piece of music.
Heretofore, the middle-aged and old mainly use this karaoke system. However, at present, karaoke is popular among wide age-groups owing to the popularization of a so-called karaoke box, that is, a small room provided with a karaoke system, where a group consisting of a few to some persons can enjoy karaoke. The object of going to karaoke is also varied to a pastime like a party.
A variety of methods for meeting a request of users of this karaoke, that is, preparing pieces of music requested for singing are tried based upon the personal information of users obtained by investigating and examining the situation of the use of karaoke.
Incidentally, it takes much time and costs for the above investigation and examination and it is difficult to utilize the result of these investigation and examination in a market timely. As the subject of the investigation is limited, it is difficult to provide service suited to the request of an individual user including a new user.
In the concrete, karaoke box is so popular that a user often must wait until his turn comes while determining pieces of music to sing and tries to sing as many pieces of music as possible when his turn comes. To sing determined pieces of music during waiting, a user must specify pieces of music by operating a operating a karaoke system after the use of the karaoke box is allowed and cannot sing during operating the karaoke system to specify pieces of music.
Generally, reservation for registered pieces of music to be sung beforehand is enabled in a karaoke system. As only a fixed number of pieces of music can be reserved, to reserve pieces of music of a number exceeding a fixed number, reservation of times equivalent to the number is required.

Remote control device, sound-reproducing system and karaoke system


A power semiconductor device (2) has a first main electrode (S) for coupling to a first supply line (3), a second main electrode (D) coupled to a first terminal (4) for connection via a load (L) to a second voltage supply line (5) and an insulated gate electrode (G) coupled to a control terminal (GT) for supplying a gate control signal to enable conduction of the power semiconductor device (2). An open-circuit detection arrangement is integrated with the power semiconductor device (2) for providing an indication that a load (L) coupled to the power semiconductor device (2) is open-circuited. The detection arrangement has a reference current (Ir) providing arrangement (7, R3, R4, R7, Q1, Q2) and a current deriving arrangement (Q3, Q4) for deriving a current (Id) dependent on the voltage at the second main electrode (D). The detection arrangement provides an output signal (OS) to indicate that the load (L) is operating normally when the derived or detected current (Id) is greater than the reference current (Ir). This should allow accurate determination of the status of the load (L) because the use of the reference current (Ir) allows account to be taken of possible leakage current paths around the load (L) so that an indication that the load (L) is operating correctly and is not open-circuit is only given when the derived current (Id) is greater than the reference current (Ir).

This invention relates to a power semiconductor switch, in particular a power semiconductor switch comprising a power semiconductor device having a first main electrode for coupling to a first voltage supply line, a second main electrode coupled to a first terminal for connection via a load to a second voltage supply line and an insulated gate electrode coupled to a control terminal for supplying a gate control signal to enable conduction of the power semiconductor device and an arrangement for determining whether or not the load is present, that is for determining whether or not the load is open-circuited. Such an open-circuit detection arrangement may be used to determine whether, for example, a lamp or bulb operated by the power semiconductor switch has blown. The invention also relates to an automotive electrical system comprising such a power semiconductor switch.
Where a power semiconductor device such as an n-channel enhancement mode power MOSFET is being operated as a low-side switch, that is where the voltage supplied to the second voltage supply line is more positive than the voltage supplied to the first voltage supply line, then the voltage between the first and second main electrodes of the power semiconductor device will depend on the status of the power semiconductor device and of the load.
Thus, when the power semiconductor device is switched off (that is it is not conducting) by removing the gate control signal and the load is connected (that is the load is not open-circuited), then the voltage at the second main electrode of the power semiconductor device will be high because it will be at the potential of the voltage supply, generally battery voltage where the protected switch is being used for an automotive application. If, however, the load becomes open-circuited for some reason, then the voltage between the first and second main electrodes of the non-conducting power semiconductor device will be zero.
In contrast, when the power semiconductor device is conducting, then the voltage between the first and second main electrodes will be very low when the load is operating normally (because the on-resistance of the power semiconductor device will be low compared to the resistance of the load) and will be zero when the load is open-circuited. It is thus very difficult to detect when a load is open-circuited when the power semiconductor device is conducting.
It ought to be relatively easy to check whether a load is operating normally (ie is not open-circuited) when the power semiconductor switch is not conducting, that is when the gate control signal is not applied to the insulated gate electrode of the power semiconductor device. However, even if the load itself is faulty, moisture or dirt in the applications environment may induce a leakage current that may pull the voltage at the second main electrode of the power semiconductor device high , even though the load is open-circuited. Accordingly, the presence of such a leakage current path may make it difficult to detect an open-circuited load even when the power semiconductor device is not conducting.
Previous attempts at detecting an open-circuit condition of a load have involved the user of the power semiconductor switch, for example an automobile manufacturer, coupling an external resistor between the first and second main electrodes of the power semiconductor device to draw off a portion of the current through the power semiconductor device in an attempt to detect whether the load is open-circuited when the power semiconductor device is non-conducting.
The use of such an external resistor means, however, that there is a permanent leakage of current from the power supply that may in time cause a drain on the power supply and could result in a flat battery where the power semiconductor switch is being used for an automotive application. Moreover, there is a danger of the external resistor becoming detached so that the power semiconductor switch is no longer able to indicate whether the load is open-circuited. In addition, such a technique involves additional expense and effort for the user of the power semiconductor switch.

Power semiconductor switch having a load open-circuit detection circuit


The present invention pertains to a synchronizer for transferring data from a first clock domain having a first clock signal at a first clock rate to a second clock domain having a second clock signal at a second clock rate different from the first clock domain. The synchronizer includes a mechanism for transferring data from the first clock domain to the second clock domain. Additionally, the synchronizer includes a mechanism for synchronizing the transfer of data from the first clock domain to the second clock domain by the transferring mechanism. The synchronizing mechanism is self-timing based only on the first clock rate and second clock rate without any additional control signals. The synchronizing mechanism is connected with the transferring mechanism. The present invention pertains to a system. The system includes a first clock domain having a first clock signal at a first rate. The system includes a second clock domain having a second clock signal at a second rate. The second clock domain is different than the first clock domain. Additionally the system is includes of a network on which data vectors travel. The network disposed and operating in the first clock domain. The system includes a memory mechanism for storing data vectors. The memory mechanism is disposed and operating in the second clock domain. The system includes a synchronizer, such as described above. The synchronizer is connected with the transferring mechanism. The present invention pertains to a method for transferring data from a first clock domain having a first clock signal at a first clock rate to a second clock domain having a second clock signal at a second clock rate different from the first clock domain.

In digital circuits, there are frequent needs to interface between asynchronous systems. Asynchronous systems operate at different clocks that have no frequency or phase relationship to one another. The common design method used to pass an asynchronous signal into a system is to sample that signal with a flip-flop running at the system clock. The signal after the flip-flop is synchronous to the system clock. The following are but a few of the many patents in this area. U.S. Pat. Nos. 3,976,949; 4,070,630; 4,873,703; 5,070,443; 5,099,140; 5,256,912; 5,291,529; 5,418,825; 5,487,092, all of which are incorporated by reference herein.
The difficulty arises when one tries to synchronize a set of signals, a data vector. Because of metastability, which can occur when the asynchronous signal transitions close to the system clock and thus violate the set up or hold time of the flip-flop, all the flip-flops that are synchronizing the data vector cannot be guaranteed to resolve, or settle, to the correct value at the same time. Thus, not all the data bits in the vector can be sampled correctly by the system clock and the data vector can have the wrong value.
Conventional technique of "double-clocking" using two flip-flops works only for one bit value. In an n-bit vector, not all the bits will settle out of metastable state at the same time to propagate synchronously to the system clock.
Using Gray coding technique to encode the data vector into Gray codes (only one bit can transition in one time period) requires extra circuitry for encoding and decoding. For large data vector the encoding and decoding can be large and can become a time critical path in the system. Also, this technique only works if the values of the data vector change in sequential order. If the data vector can change value arbitrarily Gray coding cannot insure a single bit transition.
Other esoteric designs rely on delay elements or special cells that are dependent on the silicon process and technology of the synchronization circuitry.
The data vector synchronizer described herein has the following features:
Synchronizes an n-bit data vector from one clock domain to another. PA1 Assumes no relationship between the phase or frequency of the two clocks. PA1 Insures data integrity with minimal lost in time. PA1 Utilizes conventional synchronous digital logic with no delay or asynchronous elements.
Application
Moving data vectors in one clock domain to another. Especially suited for bus interface applications where one clock domain is totally asynchronous to another.
Synchronous FIFO buffering asynchronous data streams. Especially effective when used to synchronize the FIFO pointers across the clock domain to provide word counts or space counts inside the FIFO.

Synchronizer, method and system for transferring data


A system and method for emulating a target system's hardware using highly parallel software emulation. Target system hardware under the control of program instructions is emulated using parallel software activity routines. Each of the program instructions is sequentially provided to a series of the activity routines, where each activity routine is a stage of a software emulation pipeline. Each activity routine emulates a particular function designated by the particular program instruction of the target system. The information is forwarded from each of the stages of the series of activity routines to their successive stages in the series. Each stage forwards its associated information to its respective successive stage when the stage has completed its designated activity, and has recognized that its respective successive stage has completed its respective function and is ready to accept more information.

1. Field of the Invention
This invention relates generally to the emulation of computing systems, and more particularly to a high performance emulation system and method for emulating a computing system through software pipelining and synchronization.
2. Description of the Prior Art
Modern computers are able to achieve much of their high performance due to the parallel operation of the internal hardware. For example, a pipelined computer may be in the process of executing more than one instruction in each hardware cycle. With a deep pipeline, a register file update for one instruction can be performed at the same time as the arithmetic operation for the following instruction, the operand fetch for the third instruction, the address generation for a fourth instruction, the instruction decode for a fifth, and so on. Within each pipeline stage additional parallel operations can take place.
A prior art software emulation technique involves emulating instructions on a host emulation computer by executing a series of native mode instructions which emulate the instructions of the target system. The target system is the system to be emulated, and native mode instructions are instructions which can be executed by the processing environment of the host emulation computer. In such an emulation system, one native mode instruction simulates the operation of a small portion of one stage of a corresponding target system hardware operation. The result is that it may require hundreds of serially executed native mode instructions to emulate a single target system instruction which normally triggers various simultaneous hardware activities.
There is a need, therefore, for a high speed, parallel software emulation system which reduces emulation time. The present invention allows for the use of multiple activities and processors, if desired, to perform emulation in a pipelined fashion. The invention also provides an efficient synchronization mechanism for passing information from one software emulation stage to another software emulation stage. The present invention therefore provides a solution to the aforementioned emulation speed problem and other problems, and offers other advantages over the prior art.
It is a primary object of this invention to provide an improved hardware emulation system.
It is another object of the invention to enhance the speed in which an emulation system can emulate a target system.
It is yet another object of the invention to provide an improved emulation system wherein software emulates hardware functions of the target system.
It is still another object to provide software emulation routines which are executed in parallel.
Still another object of the invention is to provide software emulation routines wherein software activities are executed in a parallel, pipelined fashion,
It is yet another object of the invention to forward information from one software activity to the next software activity in the software emulation pipeline, and to forward the information when a particular stage has completed its activity and when the next stage is ready to accept more information.
Other more detailed objectives will become apparent from a consideration of the Drawings and the Detailed Description of the Preferred Embodiment.