The invention relates to a method for performing positioning in a radio system. At least two different antenna elements of an array antenna of a base station transmit mutually non-interfering identifier signals that are received by a terminal that generates a spatial signature of the received signals. Either the terminal or the network part of the radio system defines the location of a transmitter or receiver by means of the spatial signature of the signals.

1. Field of the Invention
The invention relates to performing positioning in a radio system.
2. Description of the Related Art
In CDMA (Code Division Multiple Access), a narrowband data signal of a user is modulated with a spreading code having a wider band than the data signal to a relatively wide band. Ln a WCDMA (Wideband CDMA) radio system, the bandwidth is yet considerably wider, because current mobile networks want to provide even more versatile services to users.
In radio systems, data transmission is done in duplex mode that can be implemented using the FDD (Frequency-Division Duplex) or TDD (Time-Division Duplex) principle. In an FDD solution, reception and transmission take place on different frequencies. In a TDD solution, reception and transmission take place at different time instants, but the frequency of the received signal and that of the transmitted signal are the same.
In WCDMA radio systems in particular, signal reception and transmission can be done with a diversity antenna that comprises an array of antenna elements. Signals are phased in relation to each other in a desired manner to make sure that the directional pattern formed by the antenna array is in the desired direction and shape. In practice, the phasing is done by multiplying the digital baseband antenna signal of each antenna element by complex coefficients that shape the directional pattern.
One of the most common positioning methods of a terminal is to measure the signal timing between the terminal and at least three base stations to find out the propagation delay of the signals between the terminal and each base station. The distance of the terminal from the base stations can be shown as a circle around each base station, because the direction of the terminal from the base stations is in most cases not known, and the radius of each circle represents the distance of the terminal from the base station. Each of the at least three circles has one common intersection point that is the location estimate of the terminal. Known measuring methods used in terminal positioning include the TOA (Time Of Arrival) measurement and the TDOA (Time Difference Of Arrival) measurement.
The TOA and TDOA positioning methods are, however, hampered by a problem caused by the distance of the subscriber terminal to the base station, i.e. near-far problem. In terminal positioning, this problem is also called a receiving problem. Other base stations do not hear a terminal that is close to one base station and the terminal does not hear the other base stations because of the interfering transmission of the nearby base station. When the signal propagation delay between the terminal and at least three base stations cannot be measured, neither can the position of the terminal be determined in this manner. In addition, methods based purely on delay measurements are often inaccurate, and it is difficult to find out the locations of the desired or first delay components in a multipath channel.
If a receiver comprises a diversity antenna or, more generally, at least a second antenna element, it is also possible to determine the location of a transmitter by means of the direction of reception. Usually, this is done by having several base stations measure the angle of arrival (AoA) of a signal transmitted by a terminal, whereby the location of the terminal can be determined to an intersection of two straight lines. Positioning is described in greater detail in publication: Overview of radiolocation in CDMA cellular systems, J. J. Caffery, Jr and G. L. Stuber, IEEE, 1998, which is incorporated herein as reference.
Determining the direction of reception is, however, a demanding task that requires good array antenna calibration and computationally demanding parameter estimation methods. Determining the direction of reception also requires a great deal of signalling from the transmitter to the base station and thus uses up a lot of capacity in the transmission direction from the terminal to the base station. In addition, the solution only enables positioning by a network element that requires the terminal to transmit something to the network element. There are, however, situations and services, in which the terminal must be capable of exact positioning, but the network element should not be able to position the terminal.


A method (600) for controlling a transmit power of an interface module (230) is provided. The method includes generating a transmit power code at a power code generator (240) (610); receiving the transmit power code at the interface module (620); receiving control signals at the interface module (650); and generating a transmit signal at a transmitter circuit (410) in the interface module based on the control signals and the transmit power code (660). In this method, the power level of the transmit signal is controlled by the transmit power code, and the transmitter circuit and the power code generator are formed on separate circuit elements.

For many electronics devices, it may be desirable to provide an interface with an external device or network. This could be a connection to a local area network (LAN), a wireless LAN (WLAN), a personal area network (PAN), a wireless PAN (WPAN), the internet, a wire connection to a remote device using a coaxial cable or twisted pair cable, etc., or any other desired wireless or wired network. For example, it might be desirable to connect a television set to a WLAN so that its audio can be sent wirelessly to remote speakers; or it might be desirable to connect a digital camera to a WPAN so that its image data can be wirelessly downloaded to a personal computer; or it might even be desirable to connect a video camera to an Ethernet connection to send data to a remote device.
One way to implement this functionality is to provide a separate module for the device that will serve to facilitate the communication between the local device and the desired external device or network. This module will pass signals between the local device and the external device or network, handling all necessary data conversion to and from the different formats used by the local device and by the external device or network.
One parameter that can be of particular importance is the output power of the signal being sent by the module to the external device or network. In many situations, this power must be within a desired range. If the power is too high, then it might either overwhelm the receiver, or it might violate an imposed power level restriction. If the power is too low, the signal might be too faint for the external device or network to properly receive.
However, since different devices have different device parameters, each device may have different internal power requirements for the module it uses, even if the devices themselves desire the same level of power output for the desired transmission media. One device might operate using an antenna with a very small gain for signals sent externally by the module. In this case, the module must output a high power signal to make certain the signal radiated by the antenna will be at the proper power level. Contrarily, if a device employs operates using a high gain antenna for any signals sent externally by the module, the module should output a comparatively lower-powered signal to make certain the signal radiated by the antenna will be at the proper power level.
As a result, a separate module must be designed and tested for each device. And individual modules can't be used with any device save the one they were designed for. This is difficult from a manufacturing standpoint because a manufacturer may have to custom make a large number of different wireless modules for all of the devices that use a module. Each module must be tailored to its intended device to provide the desired level of output power to a particular output source. And using the wrong module could result in an incorrect output power, causing the interface with the external device or network to fail, or perhaps causing the data transmission to violate a set power requirement. As a result it can be difficult for a module maker to achieve certain economies of scale.
It can also be difficult for a device manufacturer to keep a desired level of inventory control for modules if it manufactures multiple devices that could interface with external devices or networks. And it raises the problem of improper modules being used by device manufacturers, resulting in incorrect operation of the devices.
It would therefore be desirable to provide a module design that would enable a single module to be used with multiple different devices, each with arbitrary couplings to the transmission media.

System and method for controlling the transmit power of a wireless module


Timing oscillators as well as related methods and devices are described. A timing oscillator may include a mechanical resonating structure with major elements and minor elements coupled to the major element. The timing oscillator can generate stable signals with low phase noise at very high frequencies which allows a timing oscillator to be used effectively in a number of devices including computers and mobile phones for time and data synchronization purposes. The signal generated by the timing oscillator can be tuned using a driver circuit and a compensation circuit.

The generation of high frequency signals has become extremely important due to the increasing use and application of wireless technology and high frequency devices such as personal digital assistants (PDAs) and mobile phones. Currently, several methods and devices can be used to generate high frequency signals. One example is an oscillator, which produces a signal that resonates or vibrates at a specific frequency. There are several types of oscillators. Oscillators can be mechanical, electrical, or a combination of the two, namely electromechanical, in nature. Electro-mechanical oscillators are commonly used because of their ability to generate a stable signal at a precise frequency. An electromechanical oscillator uses the vibrations of a mechanical element to create an electrical signal. Electro-mechanical oscillator signals are often used in applications involving timers due to the precise and stable nature of the generated signal. Electro-mechanical oscillators used in timing applications are often referred to as timing oscillators.
Timing oscillators can be used in several devices including digital clocks, radios, computers, oscilloscopes, signal generators, and cell phones. Timing oscillators generate a clock signal, for example, as a reference frequency to help synchronize other signals that are received, processed, or transmitted by a device. Often times, multiple processes are run simultaneously on a device and the execution of such processes rely on a clock signal that is generated by the timing oscillator. A designer's or user's ability to effectively manage and synchronize data at high speeds using timing oscillators makes electromechanical oscillators a valuable component of several hardware and software designs and devices.
An example of an electromechanical timing oscillator is a crystal oscillator. When an electric field is applied to a crystal, the crystal becomes distorted. Upon removal of the electric field, the crystal returns to its previous shape and generates an electric field and voltage. This phenomenon is known as piezoelectricity. Depending on the composition of the crystal, the signal produced by the crystal will have a certain resonant frequency. However, using a crystal oscillator for high frequency applications may have several disadvantages. The resonant frequency of the signal generated from a crystal oscillator is dependent on the size and shape of the crystal. Most crystal oscillators are useful for generating signals in the KHz to MHz range whereas most of the latest technology demands signals in the GHz range. Furthermore, the size of a crystal is significantly large occupying more space on a chip compared to other available components.
One solution to overcome the limitation of the generated frequencies of a timing oscillator is to use multipliers. The generated signal can be multiplied using a mixer or a number of other devices known to one of skill in the art to output a new signal at a much higher frequency. For example, a multiplier receiving a signal with a frequency of 50 MHz as an input, can output a final signal of 2 GHz by multiplying the input signal by a factor of 40. However, such an approach is problematic, since doubling a signal's frequency may result in increasing the phase noise, for example, by 6 dB. Hence converting a signal from the MHz range to the upper MHz range or GHz range will result in a significant corruption of the signal quality because of an increase in the phase noise. Conventional timing oscillators may thus not be ideal for generating high frequency signals.

This patent is subject to a terminal disclaimer.


Methods, systems, and computer readable mediums determining a system state of a power system using a convolutional neural network using a convolutional neural network are disclosed. One method includes converting power grid topology data corresponding to a power system into a power system matrix representation input and applying the power system matrix representation input to a plurality of convolutional layers of a deep convolutional neural network (CNN) structure in a sequential manner to generate one or more feature maps. The method further includes applying the one or more feature maps to a fully connected layer (FCL) operation for generating a respective one or more voltage vectors representing a system state of the power system.

Traditionally, power system flow equations are solved by a model-based analytical method, such as the Newton-Raphson method, fast-decoupled method, DC power system flow method, and the like. However, with the increasing penetration of renewable energy and demand side resources into the bulk power grid, conducting accurate system modeling and determining the related solutions becomes more difficult with the uncertainty involved. Namely, the growing number of these energy sources and resources being introduced into power grid systems has presented considerable uncertainty and risk to system-wide operations and stability. Consequently, a significant number of different scenarios that need to be considered for power system flow calculations and traditional power-flow-based contingency screenings may be required. In these instances, traditional model-based power system flow calculation methods may experience significant computational burdens and fail to meet real-time and multi-scenario analysis requirements. For example, in a typical N-k contingency analysis, conventional analytic methods can only function in an off-line (e.g., non-real time) manner due to the requisite heavy computational burden. These analytic methods are likely to fail in instances of resource integration on a large scale or dimension.
Thus, there currently exists a need in the art for determining a system state of a power system using a convolutional neural network.

Methods, systems, and computer readable mediums for determining a system state of a power system using a convolutional neural network

University of Tennessee Research Foundation


An identity state observer for a smooth-rotor permanent-magnet synchronous motor is derived which reconstructs the electrical and mechanical states of the motor from current and voltage measurements. The observer operates in the rotor frame and estimates direct and quadrature stator currents, rotor velocity, and rotor position. Since the rotor position is estimated, the rotor reference frame is approximated using the latest rotor position estimate. The motor dynamics and the transformation into the estimated rotor frame are nonlinear and thus the observer and observer error dynamics are nonlinear. Therefore, small signal stability is analyzed from a linearized model. Simulations including realistic measurement disturbances are used to investigate the global stabilitiy and accuracy of the observer.

This invention relates to a state observer for a permanent magnet or wound rotor synchronous motor.
Electric motors are traditionally used as actuators. However, they may also be used as sensors of the motion they actuate. This simultaneous operation as an actuator and sensor can be beneficial when additional motion sensors are too expensive, large, unreliable, or otherwise undesirable. In particular, the potential cost savings of eliminating a motion sensor can be great.
An electric motor can be successfully used as a sensor of its own motion because this motion effects the voltages and currents in it. Thus, the voltages and currents possess information concerning the motion. What is needed, then, is a means of extracting this information so as to estimate the desired motion. Some form of signal processing is necessary, which places additional demands on the control electronics. However, as digital processors continue to become faster and less expensive, this additional signal processing becomes less of a burden.
In this specification, an identity state observer is used to estimate the electrical and mechanical states of a smooth rotor permanent magnet synchronous motor. The estimated states are the direct and quadrature currents, the rotor velocity, and the rotor position. Thus, the output of the observer is precisely the states desired for typical implementations of field-oriented, or vector, control [5]. (Numbers in brackets refer to the references listed at the end of this specification.) An identity state observer utilizes a model of the dynamics of the machine observed and includes an innovation term multiplied by a gain. By "smooth rotor" is meant a machine having a single harmonic of air gap flux.
The estimation of motor motion using measurements of electrical variables is not a new idea. A wide variety of methods have already been investigated in the literature. At one extreme are waveform detection methods that are commonly associated with permanent-magnet and variable reluctance stepper motors; see references in [6]. These methods attempt to identify specific events such as peaks or zero crossings in the electrical waveforms that are the result of the back-emf of the rotating rotor. They have been successfully demonstrated for commutation needs, and may be implemented using inexpensive electronics, which is advantageous from a manufacturing viewpoint. However, they are not as accurate as possible because they do not utilize all the information present in the waveforms. The fact that the back-emf is always present while the rotor is moving means that back emf information which may reveal the position of the rotor is always present in the waveforms.
One way to extract all the information in the waveforms is to model the dynamics of the motor, drive these dynamics with the same input as is used to drive the real motor, and somehow ensure that errors between the modeled motor and real motor are minimized. If this can be done, then the states of the modeled motor will effectively summarize all the information in the waveforms up to the present time, and the model will accurately reflect the behavior of the real motor. In a state observer, an output is defined as a combination of the states and this output is compared with the equivalent measured output of the real motor. Any error between the two signals is then used to correct the state trajectory of the observer.
State observers have been studied for a wide variety of electric machines. For example, a linear observer based on the linearized model of a generator is studied in [8]. Although the observer is designed for operation at a specific operating point, it is shown that it performs satisfactorily for small perturbations over a wide range of operating conditions.
In [6] and in U.S. patent application Ser. No. 927,532, filed Nov. 5, 1986, an observer is developed for a variable reluctance motor where spatial variations in phase inductances affect the evolution of the flux in the phases as the rotor rotates. An observer is constructed that models the spatially-varying motor dynamics, and a very accurate position estimate is produced under a constant speed assumption. The structure of this observer is nonlinear since the motor dynamics and the feedback gain on the innovation are nonlinear functions of the estimated states.
In [10], an observer for a synchronous generator that has the structure of a linear observer, that is, constant gains on linear innovation, is applied to a nonlinear model. A potential drawback is that the observer in [10] does not exploit the unique qualities of the nonlinearities in the generator, and therefore misses the potential for even faster converging estimates.
A complex observer strcture may be found in [2]. There, optimal control theory is used to develop an observer which is optimal with respect to random system noise and random measurement noise. The system noise may be due either to modeling errors or disturbances. Though not deemed as such in the paper, the resulting observer may be called an invariant embedded observer. It is similar to an extended Kalman filter and, unfortunately, appears to have many of the same drawbacks as the extended Kalman filter, particularly the necessity of carrying along a system of covariance dynamics which increasese the order of the observer from four to fourteen. Although optimization with respect to system noise is clearly desirable, it incurs a heavy price. No consideration is made of the underlying nonlinear system. Rotating machines have strong symmetries that can be exploited to yield simpler structures. The invariant embedded observer would have to be demonstrably superior to other methods before it could be seriously considered for real-time application.
Models for alternating current motors are nonlinear and estimation theory for nonlinear systems is not well developed compared to the wealth of guidance available for linear systems, such as direct-current motors. However, there is considerable symmetry in alternating-current motor models which can be used to simplify an estimation problem. Ignoring this symmetry may lead to estimators that are more complex than necessary. The state observer according to the invention combines the ideas of a linear observer with the d.sub.q transformation that may be applied to smooth-rotor machines. The resulting state observer is one that could not have been developed from a general expression for a nonlinear system.

State observer for the permanent-magnet synchronous motor


A display system includes a base member, a rear support member which is connected to one end of the base member in an upward and downward direction, a display device, a support unit which is disposed on an upper portion of the rear support member, and supports the display device such that the display device is movable in position and changeable in position in at least two axial directions, a sound output device which is disposed on the display device and the rear support member, and a control device which controls operations of the support unit, the sound output device and the display device. A bed, a mattress, a bedding covering or the like is provided on the base member such that a user can watch a screen of the display device in a posture where the user has lain above the base member.

Recently, flat panel display type TVs including liquid crystal display (LCD) TVs and plasma display panel (PDP) TVs occupy most of the TV market. Such flat panel display type TVs have advantages over conventional cathode ray tube (CRT) TVs in that high definition, a large screen size, a light weight and a slim profile can be realized.
Such flat panel display type TVs are configured such that they can be disposed at preset positions in a living room or a bed room using a pedestal or be mounted on a wall using a wall-hanging bracket device, whereby users can watch the TVs.
However, as for such flat panel display type TVs, it is not easy to change positions where the TVs are disposed. In addition, it is not easy for users to comfortably watch TVs with a suitable viewing angle in a posture in which they have lain on the floor, bed or the like, which is problematic.
In addition, at a typical home, a bed is placed inside a room, and a TV is disposed in a living room or on a wall. Considering this, the user who has lain on the bed is spaced too apart from the TV which is disposed on the wall or the like. It is therefore difficult for the user to comfortably watch the TV. It is also difficult to properly transmit sounds from the TV to the user. Therefore, there are problems in that it is difficult to comfortably watch the TV and sound production is not satisfactory.
In addition, since there are no means for providing the user with the comfortable and clean bed state, it is difficult to improve the surrounding environment while staying on the bed for a long time. This causes bad effects to health, which are problematic.
Furthermore, there is a problem in that it is difficult for the user to check or improve his/her health condition while staying on the bed for a long time.


According to the preset mode instructions from the operating part 111, data of a plurality of channels to be included in the search screen are input from the operating part 111. These channel data are transferred to the second channel selection controlling part 109 for storage. When preparing, a search screen, these stored channel data are used for constituting a search screen, which is displayed on the displaying part 112. The channels desired or frequently watched by the user alone are searched and their images are displayed on the multiple-division screen as still pictures.

In addition to conventional ground broadcasting, satellite broadcasting services using broadcasting satellites (BS) and communication satellites (CS) are currently in use. Television sets with a converter to receive such satellite broadcasting are available on the market, but since that the number of such channels is increasing, there is a need for useful broadcasting channel selection.
For the ground broadcasting, on the other hand, the second generation EDTV broadcasting is under study. The aspect ratio of the TV broadcasted images is changing from the current ratio of 4:3 to the wide aspect ratio of 16:9 while maintaining compatibility with the current broadcasting. Television sets for the second generation EDTV broadcasting are designed to maintain the compatibility with the conventional images. Specifically, when displaying an image with the aspect ratio of 4:3 on a display unit for the aspect ratio of 16:9, the image is compressed in the horizontal direction so that the 4:3 image can be displayed on the 16:9 screen.
Some inventions have disclosed using a blank part on the screen generated by compression when a television set for the wide aspect ratio receives the ground and satellite broadcasting, proposing that such blank parts be used for broadcasting channel selection as described above.
One of such proposals is disclosed in Published Unexamined Japan Patent Application No. 62-263781 (No. 263781/1987). This application discloses a television set which displays images of other channels and high definition TV images in the blank part of the screen. By displaying, images of many broadcasting, stations other than the selected one at the same time, this television set provides a useful advantage for channel selection to the user.
A known television set having two tuners which uses one tuner to display images from the selected broadcasting station and the other tuner to repetitively select channels one by one so that images from a plurality of stations can be displayed at the same time as reduced images on the display unit. Thus, the image displayed on the screen can be quickly switched between the images based on both tuners (Refer to Published Unexamined Japan Patent Application No. 6-54269 (No. 54269/1994)).
Another invention proposes a television set having two divisional screens which displays reduced images (search screen) from various stations on one of the screens [Refer to Published Unexamined Japan Patent Application No. 8-223495 (No. 223495/1996) (Japan Patent Application No. 7-30880 (No. 30880/1995))].
With such configuration, the user can check the reduced images from various stations to select a channel and immediately display the selected image on the main screen.
With an increasing numbered channels, however, there is a problem that such television set cannot display all channels to be watched. Therefore, the search screen is designed to allow switching as shown in FIGS. 7(a) and (b). Specifically, after searching Channels 1 to 9 as shown in FIG. 7(a), the search screen is switched to the next screen displaying Channels 10 to 18 for searching as shown in FIG. 7(b).
In the above case, a two-screen displaying television set with the main screen on the left and the sub-screen on the right is provided with a function to display nine reduced images on the sub-screen as the search screen for the broadcasting waves which is received as input signals to the antenna. It has a memory for each of these nine images and the video data channel-searched and selected by the tuner is stored on the memory for later displaying on the screen. Naturally, they are still pictures. It has nine small screens only and, if there are ten or more broadcasting stations, it is necessary to switch to the next set of small screens. The data on the memories are updated by searching again upon completion of the nine small screens, searching so that the image is switched to a new one.
Though this configuration can display all channels on the search screen, it cannot display all channel data on a single screen. It is always necessary to follow a troublesome operation procedure to switch between screens.
Therefore, it is an object of the present invention to provide a television set which can display all channels desired or frequently used by the user on one screen to as the search screen.

Television set having a multiple-division screen


A method for controlling burst assembly, device, and communication equipment thereof are provided. In the present invention, an assembly time and a predetermined time threshold are compared to determine a load change in a cache queue. If the load is increased or decreased suddenly, a length threshold is directly increased or decreased. If the load is to be increased, the length threshold is randomly increased according to a certain probability. If the load is to be decreased, the length threshold is randomly decreased according to a certain probability. If the load is in a normal state, the length threshold is maintained. Therefore, the length threshold is dynamically adjusted with the load change. The method and device provided by the present invention ensure an end-to-end (E2E) delay of the services and optimize the network performance.

With the rapid development of Internet services and the diversity and variability of emerging services (such as web browsing, multimedia services, resource downloading, and network conference services), conventional core switching networks have already lagged behind the requirements of users. Therefore, in order to keep pace with the explosive growth of Internet traffic, the increasing demand on bandwidth, and the diversity and burst characteristic of the services, researchers devote themselves to the study of wavelength division multiplexing (WDM) and even dense wavelength division multiplexing (DWDM) transmission. At present, the DWDM technology is able to raise the transmission bandwidth to about 10 Tbit/s, thus meeting the requirement on the bandwidth of a transport network in a long term. As a network switching node needs to convert transmitted data from an optical domain to an electrical domain and then back to the optical domain, the integrated circuit (IC) technology has become a bottleneck that restricts the node processing capability in an optical network. Though the network bandwidth is large enough, the network transmission rate is still limited. Thereby, an optical switching technology emerges as required, i.e., the whole conversion is completed in the optical domain.
Currently, the optical network adopts optical circuit switching (OCS), which is simple in protocol, mature in technology, and easy to implement. OCS is similar to circuit switching, and is connection-oriented, so as to establish a transmission optical path for end-to-end (E2E) traffic flow. However, due to the changeability and burst characteristic of the Internet services, the establishment and release of the optical path are quite frequent. Thus, on account of the dramatic contrast between the overhead of the establishment and maintenance of the optical path and the duration of the optical path connection (usually very short), the bandwidth utilization is rather low.
Optical packet switching (OPS) is a switching mechanism of fine granularity. Similar to conventional packet switching networks, the OPS also adopts a store-and-forward technology. As OPS allows the statistics and multiplexing of network channel bandwidth resources, it is particularly suitable for burst Internet data services. Therefore, OPS is quite promising in the long run. However, currently some problems difficult to handle still exist. First of all, the optical switch technology capable of effectively supporting the OPS is not mature, and meanwhile the switching speed of an optical switching matrix is far from meeting the requirement. Secondly, it is also difficult to perform packet synchronization at the entry of a switch, and thus the conflict between the packets cannot be effectively decreased. Finally, a random access mechanism in essence has not been proposed yet for the optical domain, and the current optical domain cache may only be realized through fibre delay lines (FDLs). The use of the FDLs is also limited by the signal quality and physical space, and may not provide sufficiently flexible and adequate cache. In view of the above, the OPS still needs to be studied for many years before being utilized.
Optical burst switching (OBS) is proposed as a compromise solution between OCS and OPS. Packets arriving at edge nodes in an OBS network are assembled into burst packets according to destination address and quality of service (QoS) for later transmission and switching. Therefore, the processing granularity is improved. Each burst packet is corresponding to a burst header packet (BHP). The BHP enters a core switching network before burst packets, and reserves resources hop by hop at all intermediate nodes that the burst packets need to pass by. As long as an offset time between the burst packets and the BHP is properly set, it is ensured that before corresponding burst packets arrive at each intermediate node, the processing of the BHP and the configuration of the corresponding switching channels have already been completed, thereby realizing E2E all-optical transparent transmission of the burst packets. Compared with the OCS, the OBS has a moderate switching granularity, and allows the statistics and multiplexing, thereby raising the resource utilization. Compared with the OPS, the OBS is easier to implement, and has a lower requirement for the components. Therefore, though the OBS is immature in terms of standard and protocol, under continuous research and development, it is likely to become a core technology for optical transmission and switching network in the next generation.
Burst assembly is critical to the design of the OBS network, which greatly affects the performance of the OBS network. FIG. 1 is a schematic view showing the assembly of IP packets into burst packets at edge nodes in an OBS network in the conventional art. Referring to FIG. 1, after arriving at the edge nodes in the OBS network, multiple IP packets are classified according to the destination address and priority thereof, and then sent to corresponding assembly queues. When triggering conditions for assembly are satisfied, the IP packets in the corresponding queues are assembled into a burst packet, then scheduled, and sent to an OBS core network for transmission.
The burst assembly method adopted by the edge nodes in the OBS network determines the triggering conditions for assembly, and meanwhile determines the characteristics of the output burst packets, thereby directly affecting the length of the burst packets, packet lost rate of the burst packets in transmission, E2E delay of the IP data packets, and utilization of the network channels. Therefore, a desired burst assembly method may effectively control the E2E delay of the packets, meet the delay requirements of the services, raise the resource utilization in the OBS network, and decrease the packet lost rate.
Currently, three burst assembly methods are disclosed in the conventional art, which are an assembly method based on a fixed time threshold, an assembly method based on a fixed length threshold, and a threshold adaptive assembly method. These methods are respectively introduced as follows.
A burst assembly method based on a fixed time threshold is disclosed in the conventional art. The basic principle of the method is that, a maximum waiting time T is set for each cache queue, and IP data packets arriving within the time T are cached in corresponding assembly queues. A timer starts timing from a first IP packet entering the cache queue, and when reaching the time T, all the data packets in the corresponding queues are assembled into a burst packet. On account of the blocking of control channels, utilization of the channels, and processing speed of core nodes in an OBS network, the length of the burst packet should have a lower limit, and thus the method further restricts a minimum length of the burst packet. If the assembly time reaches the maximum waiting time T while the burst packet length is inadequate, the packet has to be filled to the minimum length and transmitted later. This method is advantageous in that the E2E delay of the IP data packets can be controlled accurately.
Seen from the above, the burst assembly method based on a fixed time threshold has the following disadvantages.
First, when a certain assembly queue is under a low load, a large amount of data will be filled, and thus the utilization of the burst packet is decreased. When the load is high, a great many data packets will arrive during the assembly, and thus the assembly queue requires plenty of cache. Besides, an extremely long burst packet may also lead to the increase of the packet lost rate. The reason is that, when a conflict occurs, the loss of one burst packet means the loss of plenty of IP packets. In addition, with dynamic changes of the load, the length variations of the burst packets are great, which may affect the network performance.
Secondly, when each edge node adopts the same time threshold, each assembly queue periodically generates burst packets. FIG. 2 is a schematic view showing continuous conflicts caused by assembly synchronization at the edge nodes. In FIG. 2, as each assembly queue at the edge node periodically generates burst packets, assembly synchronization occurs between the burst packets, thereby resulting in continuous resource competitions at an egress of a switching structure of the edge node and the core node. At the edge node, packet loss may be avoided through cache, but an additional cache delay may be generated. However, at the core node, due to the lack of optical cache, a large amount of packet loss may be caused continuously, and the channel resources may not be fully utilized.
A burst assembly method based on a fixed length threshold is disclosed in the conventional art. The basic principle of the method is that, a fixed length threshold is set for each assembly queue, and arriving data packets are still sent to corresponding assembly queues for cache. When a total length of the data packets in the cache queues reaches a length threshold, all the data packets in the corresponding queues are assembled into a burst packet. The method is advantageous in that the burst packet length is relatively fixed, which may improve the network performance, optimize the scheduling, and avoid assembly synchronization to some extent. Compared with the burst assembly method based on a fixed time threshold, the method based on a fixed length threshold is advantageous in decreasing the packet lost rate.
However, the burst assembly method based on a fixed length threshold still has the following disadvantages.
First, when a traffic load is low, this method may cause a great assembly delay, and is thus inapplicable to real-time service transmission. An advanced technical solution is also provided in the conventional art, i.e., a time threshold is added to limit the delay, so that the assembly is completed when either of the two thresholds is reached. That is to say, the assembly based on a fixed time threshold is adopted under a low load, and the assembly based on a fixed length threshold is adopted under a high load. This advanced method integrates two thresholds, but still has limited adaptability to the change of the traffic, and the blocking of burst packets caused by assembly synchronization still exists at a low load.
Secondly, when the load is high, a generation period of the burst packets is short. If the length threshold is set too small, plenty of BHPs may flood into the control channel in a short time, thus resulting in the blocking of the control channel. Moreover, due to dynamic changes of the network load, it is difficult to predetermine an optimal length threshold in practice, and thus the delay problem caused by the method is obvious.
In order to enable the assembly threshold to vary with the dynamic changes of the traffic load, a threshold adaptive burst assembly method is further disclosed in the conventional art. The basic principle of the method is to adopt a prolonged length threshold and a fixed time threshold. The fixed time threshold is set to ensure a maximum tolerable delay under a low load, and the prolonged length threshold is set for adjustment according to the current load condition. Moreover, in order to decrease the continuous dramatic changes of the burst packet length, this solution introduces a hysteresis change characteristic, that is, after each assembly is completed, the burst packet length of this assembly is compared to obtain the latest load condition. If it is ensured that the traffic does change through several comparisons, the length threshold is adjusted.
Seen from the above, the threshold adaptive burst assembly method has the following disadvantages.
First, in this method, if the length threshold triggers an assembly, the load will be increased, and thus the length threshold needs to be enlarged. This inappropriate load balancing method may result in an ever increasing length threshold till the time taken to complete the assembly under the current load reaches the fixed time threshold, or the length threshold reaches a predetermined tolerable maximum value, which may cause an unnecessary increase in the assembly delay. Moreover, the judging whether the load changes or not actually depends on the length of the IP packet that enters the queue last to trigger the assembly. The load determination mode adopted by this method cannot be well adapted to the change of the load.
Secondly, though the assembly length threshold is set to change with the load, this method does not clearly state in what relation the two may match up with each other, i.e., the objective of the adaptability is not explicit.
Thirdly, this method introduces the hysteresis change characteristic, so that the assembly threshold is adjusted after it is ensured that the load changes through several comparisons, thereby decreasing the continuous dramatic changes of the length threshold to some extent. However, in this manner, the tracking ability of the length threshold on the burst is also decreased, i.e., the adaptability of this method is lowered.

Method, device and communication equipment for controlling a burst assembly


For use with a protection circuit having an in-rush current control element that controls in-rush current into a power converter, a control circuit, a method of controlling the in-rush current control element and a power supply employing the control circuit and method are provided. In one embodiment, the control circuit includes: (1) a sensor, coupled to the in-rush current control element, that senses an in-rush current through the in-rush current control element and produces an intermediate signal and (2) a controller, coupled to the sensor and the in-rush current control element, that controls the in-rush current control element to limit the in-rush current when the intermediate signal exceeds a first reference current signal and to substantially interrupt the in-rush current when the intermediate signal exceeds a second reference current signal.

Many current power systems employ DC power supplies having DC/DC power converters located on circuit cards (e.g., load cards) which are directly coupled to voltage buses. The voltage buses generally supply voltages of a magnitude of around 40 V to 60 V. The load cards are usually coupled to the voltage bus and to a separate load.
For system reliability, the load cards should be able to connect to and disconnect from the voltage bus without disturbing the operation of other cards and the bus voltage. Furthermore, if a fault condition occurs in a card, other cards should be electrically isolated from the condition until the fault condition is remedied. Typically, an isolation diode and local bus capacitors are mounted on the cards to accomplish the electrical isolation.
Unfortunately, using local capacitors creates an in-rush current that occurs at the moment when the card is coupled to the voltage bus. The narrow-pulsed and high amplitude in-rush current commonly causes current stresses on the associated electrical components and electromagnetic interference in the system as a whole. Therefore, the current that rushes into the local bus capacitors must be introduced in a controlled manner. The in-rush current is commonly regulated using an RC timing circuit in conjunction with a switching component, such as a metal-oxide semiconductor field-effect transistor (MOSFET) . The turn-on timing of the switch is controlled by the timing circuit. When the card is initially connected, the RC circuit begins charging and will slowly allow the switch to reach its turn-on threshold. As the switch turns on, the in-rush current is allowed, via the switch, to begin charging the capacitors at an acceptable rate.
The in-rush current, however, often cannot reach a satisfactory level due to component tolerances. For instance, the threshold voltage for turning on a MOSFET switch is normally about 2 VDC to about 4 VDC. Capacitance tolerances of the capacitor in the RC timing circuit are approximately .+-.20%. Furthermore, the charging rate of the capacitor is high at low capacitor voltages and low at high capacitor voltages. Therefore, at lower turn-on threshold voltages, the switch begins to conduct very quickly; conversely, the switch begins to conducts more slowly (i.e., less current) at high turn-on threshold voltages. When the switch turns on quickly, a dip in the bus voltage could occur. Such fluctuations in the bus voltage are unacceptable in systems that require high reliability.
When the in-rush current is too large, the switch will suffer a short circuit condition. A fuse is normally employed to protect against short circuit situations in higher reliability systems. When a fuse is connected in series with a semiconductor switching device, such as a MOSFET, it is not always able to detect a short circuit or protect the switch from damage.
Accordingly, what is needed in the art is a protection circuit capable of providing overvoltage, current limit and short circuit protection for a power converter that overcomes the deficiencies in the prior art.

Control circuit for an in-rush current control element, and a protection circuit and power supply employing the same


A device for rapidly remotely coupling together two vessels, in particular a first ship or floating support and a second ship, comprises: at least one floating and docking structure fastened to or suitable for being releasably fastened to the side and/or the keel of the hull of a second vessel; and at least two actuators spaced in succession from one another in the longitudinal direction of the first vessel. The actuator cylinder of each the actuator is arranged to be fastened to the side of the hull of the first vessel, using a first fastener and pivot hinge device. The end of the rod of each actuator is arranged to be fastened to or suitable for being fastened to the floating and docking structure via a second fastener and pivot hinge device.

The present invention relates to a device for docking together two vessels at sea and referred to herein as a "coupling device". This docking device serves to keep the two vessels spaced apart laterally from each other at a controlled distance, typically about thirty meters, while conserving their side-by-side longitudinal position, in particular in order to perform transshipment between the two vessels.
The term "vessel" is used herein to designate both a transport ship and also a floating support moored with the sea bed, such as a floating production storage and offloading (FPSO) unit, a floating liquefied natural gas (FLNG) unit for producing, storing, and offloading liquefied natural gas, or a floating storage and regasification unit (FSRU).
This type of device is particularly adapted to enable a first ship or floating support of the type comprising a floating installation for liquefying or regassifying natural gas (FLNG) to be offloaded to a second ship such as a methane tanker or "LNG carrier" via flexible or rigid pipes.
The difficulty encountered is the limit of environments that are acceptable during the offloading, in particular swell, wind, and/or sea current conditions that often make such transshipment operations difficult between two vessels at sea without running the risk of collision between the ships.


Method and apparatus for providing a filtered signal which indicates of a flow of fluid to a fluid pressure communication conduit, the flow being dependent upon the pressure of fluid in a chamber. The pressure in the chamber is controlled by one or more pressure adjusting valves. The invention provides for receiving into a data processing module one or more valve signals indicating open states of the pressure adjusting valve or valves, and also a flowrate signal from a flowrate sensor, which represents the real instantaneous flowrate, and it is this signal which needs to be filtered to remove transients. The invention provides for choosing between a first processing mode and a second processing mode for the flowrate signal. The first processing mode is used when the second processing mode is not in effect, and when neither of the valve signals indicates an open valve. The second processing mode is activated a valve signal indicates an open valve. The second processing mode, when activated, proceeds according to a predetermined procedure and then returns to the first processing mode after no valve signal indicates an open valve. The signal is processed by either the first processing mode or the second processing mode, depending on whether valve signals have been received, and provides the filtered signal as an output to equipment external to the data processing module.

As is generally well known in the railway engineering arts, brakes on a train are generally controlled by a brake line which carries compressed air from the locomotive of the train to braking systems in the individual railroad vehicles of the train. It is also well known that the classical airbrake system as derived from the Westinghouse airbrake maintains compressed air in the brake line when brakes are not required, and signals a need for brake application by dumping air from the brake pipe in the locomotive to decrease the pressure in the brake air line when a brake application is called for. This system has the desirable feature that a failure of the brake air line causes application of brakes throughout the train.
Unfortunately, the time needed for a pressure decrement to propagate down the line of railroad vehicles in a train can be quite large, a minute or more for a mile-long freight train. Hence, brakes on cars remote from the engine do not help in is stopping the train until some time has elapsed following the brake application.
More rapid methods are known for transmitting a signal for brake application down the length of a train, which, for example, include the use of electrical wires, electromagnetic signals, or optical transmission. For railroad braking systems, the classical brake air line may be combined with radio transmission, particularly in a train having locomotives distributed at various locations along the train.
The WABCO EPIC.RTM. brake system combined with a radio communication link from Harris LOCOTROL.RTM. provides a system in which a brake application signaled by the lead locomotive of a train is accompanied by a radio signal sent from the lead locomotive to slave locomotives in portions of the train remote from the lead locomotive. As usual, with railroad airbrake systems, the lead locomotive dumps brakeline air, which sends a pressure decrement down the line of cars, causing a brake application as it proceeds. In addition, the radio signal is immediately received in locomotives remote from the lead locomotive, and these also begin venting brakeline air. Brakeline pressure decrements then begin to travel along the succession of railroad vehicles from each slave locomotive, causing the brakes to be applied as the pressure decrement reaches each vehicle.
Operation of this system requires, in each locomotive which supplies air to the brakeline, a measurement of the flowrate of air from a main air reservoir in the locomotive to the brakeline of the locomotive. The air pressure in the main air reservoir is maintained by a compressor in the locomotive. This flowrate can be used for a number of purposes. One thing it is used for is to determine the leakage flowrate. This is the flowrate of air which leaks out of the brakeline anywhere in the train. An accurate value for this flowrate is also desirable when the train is being prepared for travel, or after a brake application. In both of these cases, the brakeline pressure must be brought up to the operating pressure value. By measuring the flowrate to the brakeline, the system can determine when the bakeline is charged. This occurs when the measured flow through the orifice is approximately equal to the leakage flowrate. A signal indicating the flowrate is also sent to an alarm system, which interprets a sudden increase in flowrate as indicating a failure of the brake air line. This causes brake application, and a signal to the lead locomotive to stop the train.
The flowrate of air is measured by an orifice, such as the air path constriction in the L19 flowblock, which is located between the main air reservoir, and the brakeline.
Another difficulty with the prior art systems is due to the following considerations: One is that during steady state, when the brakes are not being applied, the brakeline continuously leaks air, and demands air from the main reservoir. Hence, the main reservoir loses pressure continuously through the orifice supplying the brakeline, and is resupplied with air by pulses of air originating in the compressor. A brake control valve placed downstream of the orifice, and upstream of the brakepipe, controls the pressure downstream of the valve in the brakepipe. The brake control valve controls airflow by a mechanical linkage to a diaphragm in an equalizing chamber. On one side of the diaphragm, in the equalizing chamber space, pressure is maintained at a constant value by pulses of air from the main chamber, and pulses of air discharged from the reservoir to the atmosphere by means of electric magnet valves. The other side of the diaphragm in the equalizing chamber has the pressure of the brakepipe. Motions of the diaphragm cause changes in the opening provided through the main control valve, to maintain the pressure in the brakepipe at a predetermined value. An unfortunate aspect of this system is that since the pressure in the equalizing space is increased or decreased in pulses, the diaphragm between the equalizing space and the space connected a pressure port on the brakepipe is moved in pulses. This directly causes pulsed variations of flow through the main control valve, and hence through the orifice which is used to measure the flow through the main control valve to the brakepipe. The signal indicating brakepipe flow, therefore, has pulsed variations due to the pulses of air admitted to or exhausted from the equalizing space.
Furthermore, the brakepipe generally has longitudinal waves caused, for example, by changes in acceleration of the train. When these waves encounter the main control valve, they change the pressure of air in the space next to the diaphragm on the side of the diaphragm toward the brakepipe. This causes the diaphragm to move, which causes a pulsed change in the flow through the main brakevalve. This flow is directly read by the pressure transducer or transducers which measure the pressure drop across the orifice, which is immediately upstream of the main brake valve. Hence, a change of flow is indicated. This tends to set off alarms which are intended to signal the situation of a severe leak or break in the brake line.
These pulsed variations have a particularly undesirable effect when these signals are differentiated in time to detect changes in flow to the brakepipe, as is necessary for responding quickly to a failed brakepipe. The differentiated flow signal may cause false indications of a failed brakepipe situation.


A hermetically RF-EMI shielded 3-axis and planarity adjustable combo link assembly for low loss connection between the coaxial RF connector of external instruments (impedance tuners) with 30- or 45-degrees wafer probes allows continuous, micro-positioner controlled independent horizontal and vertical probe movement. Flexible sealing O-rings and RF absorbing sheets and gaskets ensure airtight and RF-EMI shielded operation.

RF and microwave devices and circuits (DUT) are best characterized in chip form "on wafer". This allows eliminating parasitic connection elements, which falsify the test results and must be corrected for by additional elaborate testing. It also allows a much larger number of devices to be tested "in situ" without having to laboriously slice the wafer, mount and wire-bond each individual chip. The "on wafer" testing is currently the preferred testing method, except for very high-power devices, (typically) beyond 20 Watt RF power, because of the limitations of the connecting microscopic contacts (wafer probe tips, FIG. 1). On-wafer testing is also the exclusive testing method in millimeter-wave frequencies above 30 GHz, since, at these frequencies, device packaging is extremely difficult to the point of uselessness.
Several manufacturers (see ref. 1, 2) make wafer probes (FIG. 1, 2A) capable of reliably testing microwave chips. The microwave probes are made, usually, using small coaxial cables with diameters of the order of 1 mm (0.04″), embedded in a solid body; because of the planar structure of the chips to be tested, the coaxial cables of the wafer probes end into a "coplanar" (GROUND-SIGNAL-GROUND) probe tip structure (FIG. 2B), whereby the center conductor of the coaxial cable becomes the signal carrying center conductor of the coplanar transmission line and the ground mantle of the coaxial cable ends up as the ground plane of the coplanar line (FIG. 2B).
Whereas the use of flexible RF (radio frequency) and DC (direct current) cables between the test equipment (impedance tuners) and the device under test (DUT), would allow position control of the wafer probes locally (see ref. 2) and (FIG. 2A), in the case of load pull the use of such cables creates prohibitively high insertion loss between the impedance tuner and the DUT and associated additional reduction of the tuning range or maximum reflection factor amplitude Γ-TUNER (FIG. 3) that the tuner can generate, rendering the DUT impedance un-reachable; Γ-RF-cable<Γ-DUT<Γ-Airline<Γ-TUNER; therefore, ultra-low loss rigid straight or bent airline inter-connections (FIG. 4) from the DUT to the load pull tuners are necessary (see ref. 4). The environmental test conditions for the wafer chips are, typically: controlled temperature, normal (near dust-free) laboratory environment and special RF-EMI (radio frequency, electro-magnetic interference) radiation-free conditions. To create those test conditions a shielded, hermetic enclosure (chamber 605, 74) is needed to protect the wafer 68, 75 on the probe station from the RF-EMI and airborne particle (dust) polluted environment (FIG. 5, 6, 7). Typical commercially available solutions to the encapsulation requirement allowing test cable access are shown conceptually in FIGS. 5 and 6 of ref. 8. In all those commercially available solutions lies some handicap either because the shielding is not leak-free (FIG. 6A, 6B in ref. 8), or because of the back and forth movement procedure of the probe, required to make or break the contact with the wafer (FIG. 5A, 5B in ref. 8) and the resulting scrapping of the probe tips on the fragile microscopic chip connection pads.
The configuration of FIG. 7, 8, 9 in ref. 8, on the other hand, while ensuring hermeticity and simple probe control, it does not include provision for full operation of a load and source pull wafer probe station by allowing controlling a source and load tuner simultaneously, except in a, not further elaborated, simplistic and cumbersome (FIG. 8) embodiment, shown in FIG. 10 of ref. 8, using a simple duplication of the device of FIG. 8 of ref. 8.

Hermetic load and source pull wafer probe station


The invention relates to a novel radiopharmaceutical that inhibits the prostate-specific membrane antigen (iPSMA), containing hydrazinonicotinamide (HYNIC) as a critical chemical group in increasing in the lipophilicity of the molecule for binding to the hydrophobic sites of the PSMA, combined with the conventional use of HYNIC as a chelating agent for radiometal 99mTc, in which the ethylenediaminetetraacetic acid (EDDA) is used to complete the coordination sphere of the radiometal. The novel radiopharmaceutical 99mTc-EDDA/HYNIC-iPSMA detects, with high affinity and sensitivity in vivo, the overexpressed PSMA protein in prostate cancer cells using SPECT molecular imaging techniques in nuclear medicine. The aim of the invention is to provide a novel specific radiopharmaceutical (radiopharmaceutical for molecular targets) for SPECT, with high sensitivity for the detection of tumours with overexpression of PSMA.

Prostate cancer (PCa) is the second commonest cancer in men worldwide [Jemal A. et al. Cancer statistics, 2010. CA Cancer J Clin. 2010, 60: 277-300]. In patients with localized PCa, the five-year survival rate approaches 100%; however, in patients with metastasis, the five-year survival rate is 31% [Wei Q. et al. Global analysis of differentially expressed genes in androgen-independent prostate cancer. Prostate Cancer Prostatic Dis. 2007, 10: 167-174]. Nearly all patients with metastasis respond well initially to antiandrogen treatments. However, the main cause of death in patients with PCa is progression to androgen-independent status.
The enzyme glutamate carboxypeptidase II, also known as prostate-specific membrane antigen (PSMA), is expressed in the epithelial cells of the prostate and is highly overexpressed in 95% of advanced prostate cancers. The expression levels of PSMA are directly correlated with androgen-independence, metastasis and progression of PCa [Santoni M. et al. Targeting prostate-specific membrane antigen for personalized therapies in prostate cancer: morphologic and molecular backgrounds and future promises. J Biol Regul Homeost Agents. 2014, 28: 555-563]. Therefore PSMA is a suitable molecular target for detection by imaging and radiotherapy of metastatic prostate cancer using specific radiopharmaceuticals.
The PSMA gene is made up of 19 exons that comprise ˜60 kb of the genomic DNA. This gene encodes a type II transmembrane protein with a short cytoplasmic fragment (19 amino acids), a hydrophobic transmembrane domain (24 amino acids), and a large extracellular domain (707 amino acids). PSMA contains Zn in the active center of the enzyme, therefore the sequence Glu-NH—CO—NH-Lys(β-naphthyl alanine)=Glu-Urea-Lys(NaI) has been proposed as an effective inhibitor of its activity [Benesová, M. et al. Preclinical evaluation of a tailor-made DOTA-conjugated PSMA inhibitor with optimized linker moiety for imaging and endoradiotherapy of prostate cancer. J Nucl Med, 58, 2015: 914-920]. In the specific chemical interaction, the 3 carboxyl groups of the Glu-Urea-Lys fragment interact electrostatically with the peptide side chains in the active center of PSMA, the oxygen of the urea coordinates with the zinc and the aromatic structure in NaI interacts for coupling with the hydrophobic active site of the enzyme. In recent clinical studies, the application of two different derivatives of iPSMA labeled with Lu-177 showed a significant decrease in the levels of prostate-specific antigen (PSA) in 70-80% of patients with PCa, without showing severe side-effects, significantly increasing the survival of the patients [Ahmadzadehfar H. et al. Early side-effects and first results of radioligand therapy with 177Lu-DKFZ-617 PSMA of castrate-resistant metastatic prostate cancer: a two-centre study. EJNMMI Res. 2015, 5:36 doi: 10.1186/s13550-015-0114-2; Kratochwil C. et al. [177Lu]Lutetium-labelled PSMA ligand-induced remission in a patient with metastatic prostate cancer. Eur J Nucl Med Mol Imaging, 42, 2015: 987-988; Baum, Richard P., et al. Lutetium-177 PSMA radioligand therapy of metastatic castration-resistant prostate cancer: safety and efficacy. J Nucl Med, 2016: DOI: doi:10.2967/jnumed.115.168443; Kratochwil, C. et al. PSMA-targeted radionuclide therapy of metastatic castration-resistant prostate cancer with Lu-177 labeled PSMA-617. J Nucl Med, 2016: doi:10.2967/jnumed.115.171397; Rahbar, K. et al. Response and tolerability of a single dose of 177Lu-PSMA-617 in patients with metastatic castration-resistant prostate cancer: a multicenter retrospective analysis. J Nucl Med, 2016: doi: 10.2967/jnumed.116.173757]. The protein PSMA is multifunctional, since it can act as an internalization receptor, as an enzyme of absorption of nutrients, or as a peptidase that participates in signal transduction in epithelial cells and in cellular migration [Rajasekaran A. et al. Is prostate-specific membrane antigen a multifunctional protein? American Journal of Physiology—Cell Physiology, 2005, 288:C975-C981]. Therefore the PSMA inhibitor radiopharmaceuticals may also be used in another type of neoplasms different from PCa, such as in the case of metastatic breast cancer, osteosarcomas, gliomas and differentiated thyroid cancer, among others [la Fougère, et al. In vivo visualization of prostate-specific membrane antigen in glioblastoma. Eur J Nucl Med and Mol Imaging, 2015, 42: 170-171; Verburg F A, et al. First evidence of PSMA expression in differentiated thyroid cancer using [68Ga] PSMA-HBED-CC PET/CT. Eur J Nucl Med and Mol imaging, 2015, 42: 1622-1623; Zeng C. et al. Prostate-specific membrane antigen: a new potential prognostic marker of osteosarcoma. Medical Oncology, 2012, 29: 2234-2239; Sathekge M. et al. 68Ga-PSMA imaging of metastatic breast cancer. (2015). Eur J Nucl Med and Mol imaging, 2015, 42:1482-1483].
However, before any radiotherapy treatment, the capture of the radiopharmaceutical in the tumors or their metastases must be evaluated by nuclear magnetic imaging in order to confirm whether the treatment will or will not be useful for the patient, as well as for determining the activity required to be administered for imparting the radiation dose for tumor ablation, i.e. personalized medicine is practised. Therefore it is necessary to use PSMA-inhibiting diagnostic radiopharmaceuticals with the aim of obtaining molecular images by positron emission tomography (PET) or by single-photon emission tomography (SPECT). Of these two techniques, PET gives higher spatial resolution and higher sensitivity, therefore the majority of the commercial PSMA-inhibiting diagnostic radiopharmaceuticals have been developed on the basis of 68Ga, which is a radionuclide for PET [Eder M. et al. Novel preclinical and radiopharmaceutical aspects of [68Ga]Ga-PSMA-HBED-CC: a new PET tracer for imaging of prostate cancer. Pharmaceuticals, 2014, 7: 779-796: Eder M. et al. 68Ga-complex lipophilicity and the targeting property of a urea-based PSMA inhibitor for PET imaging. Bioconjugate Chem, 2012, 23: 688-697; Weineisen et al. 68Ga- and 177Lu-labeled PSMA I&T: optimization of a PSMA-targeted theranostic concept and first proof-of-concept human studies." J Nucl Med, 2015, 56: 1169-1176; Afshar-Oromieh, A. et al. Comparison of PET/CT and PET/MRI hybrid systems using a 68Ga-labelled PSMA ligand for the diagnosis of recurrent prostate cancer: initial experience. Eur J Nucl Med and Molecular Imaging 41.5 (2014): 887-897].
However, at national and international level, studies by SPECT represent more than 70% of the total in nuclear medicine owing to its lower cost and greater availability of equipment and radionuclides, since it is not necessary to have a cyclotron within the hospitals or near them. For SPECT images, the radionuclide most used is 99mTc, and there is no publication devoted to a complete clinical study with PSMA inhibitors labeled with 99mTc, for which it has only been investigated up to preclinical studies [Kularatne A., et al. Design, synthesis, and preclinical evaluation of prostate-specific membrane antigen targeted 99mTc-radioimaging agents. Mol Pharmaceutics, 2009, 6: 790-800; Lu, G. et al. Synthesis and SAR of 99mTc/Re-labeled small molecule prostate specific membrane antigen inhibitors with novel polar chelates. Bioorganic & Medicinal Chemistry Letters, 2013, 23: 1557-1563; Hillier S. M. et al. 99mTc-labeled small-molecule inhibitors of prostate-specific membrane antigen for molecular imaging of prostate cancer. J Nucl Med, 2013, 54: 1369-1376]. Only the inclusion of a diagnostic image with 99mTc has been published, as part of a radiotherapeutic study with 177Lu in patients [Kratochwil, C. et al. PSMA-targeted radionuclide therapy of metastatic castration-resistant prostate cancer with Lu-177 labeled PSMA-617. J Nucl Med, 2016: doi:10.2967/jnumed.115.171397].
To be able to equal the high sensitivity of the PET radionuclides, it is necessary to develop molecules that are more competitive from the viewpoint of affinity and feasibility for being labeled with 99mTc, in order to detect tumoral lesions that overexpress PSMA by SPECT techniques and with high sensitivity.

Instituto Nacional de Investigaciones Nucleares


Described is a process for effecting the relaxation of a tensive back, shoulder or neck muscle of a human mammal for a substantial period of time. The process of our invention consists essentially of the step of continuously or periodically administering to said human mammal through inhalation over a stress reactivity-reducing, tensive back, shoulder or neck muscle-relaxing period of time, a stress reactivity-reducing, tensive back, shoulder or neck muscle-relaxing quantity and concentration of one or more fragrance compositions having an aroma profile including one or more of a rose, floral, musky, ambery, sweet and/or powdery aroma notes.

Reactivity to stress is insidious because it does not directly incapacitate a human, but acts as a vector of extreme discomfort in various locations of the body of the human, e.g., back, shoulder and/or neck muscle pain and spasms which are a direct result of excessive back, shoulder and/or neck muscle tension brought about by the reactivity to stress.
The term "stress" hereinafter refers to an event or experience in life of an individual that has specific physiologic and/or subjective consequences that disturb the equilibrium of the individual (see Kopin, I. J., Definitions of Stress and Sympathetic Neuronal Responses Stress: Basic Mechanisms and Clinical Implications, Annals of the NY Academy of Sciences, G. Chrousos, R. McCarty, K. Pacak et al editors, New York, N.Y., New York Academy of Sciences 771: 19–30.). Sources of stress may be an individual's occupation or it may be a life event such as a change of job. The term "reactivity" hereinafter refers to the change generated by stress in the individual's physiologic and/or subjective condition. Within the context of this invention, the term "reactivity" may be ascertained (a) objectively, by measuring, using electrical measuring techniques, the degree of muscular back, shoulder and/or neck tension and (b) subjectively by use of self-report of various symptoms including degree of muscular back and/or neck pain and degree of muscular back and/or neck 'tension' and 'relaxation'. Unlike a drug that is ingested orally or injected subcutaneously, the fragrance compositions utilized in the practice of our invention are inhaled. Hence, for the purpose of practicing this invention, the term "amount administered" hereinafter is intended to mean "amount of stress reactivity-reducing composition calculated to have been inhaled, retained, absorbed and/or provided into the blood-stream".
Methods for reducing physiological and/or subjective reactivity to stress in humans being subjected to stress conditions by administration of stress reactivity-reducing substances such as nutmeg oil, orange oil and valerian oil through inhalation are well known in the prior art, for example, Warren et al, U.S. Pat. No. 4,671,959 issued on Jun. 9, 1987 and France et al. PCT Published Patent Application WO 00/37092 published on Jun. 29, 2000 (corresponding to Provisional Application for U.S. Letters Patent Ser. No. 60/113,347 filed Dec. 21, 1998). The Warren et al patent and the France et al. published patent application does not disclose or suggest the method of inhalation therapy to induce muscle relaxation, and furthermore, no disclosure of back, shoulder and/or neck muscle relaxation is set forth therein.
In addition, Librizzi et al. PCT Published Patent Application WO 01/24807 (corresponding to Provisional Application for U.S. Letters Patent Ser. No. 60/157,371 filed on Oct. 1, 1999) discloses a method of calming a mammal by administering to said mammal a personal care composition which comprises an effective amount of a sensory fragrance wherein the personal care composition is capable of reducing the cortisol level of the mammal by about 0.1% up to about 75% and/or increasing the sigA level of the mammal by from about 10% up to about 150%. Although Librizzi et al. indicates at page 3: "The term 'administering' refers to (i) inhalation of a topically applied personal care composition . . . ", the Librizzi et al. published PCT Patent Application does not, however disclose or suggest the method of inhalation therapy to induce muscle relaxation, and furthermore, no disclosure or suggestion of any methods for the causation of back, shoulder and/or neck muscle relaxation or significant reduction of back, shoulder or neck muscle tension is set forth therein.
The use of "Kava" (Piper methysticum) and Valerian (Valeriana officinalis) to induce muscle relaxation is disclosed by Greadon, J. and Graedon, T. in "The People's Pharmacy Guide to Home and Herbal Remedies", © 1999 by Graedon Enterprises, Inc., at pages 335–337, and 375–377; Published by St.Martin's Press, New York, N.Y. The Graedon et al publication does not disclose or suggest the method of inhalation therapy to induce muscle relaxation, and furthermore, no disclosure or suggestion of back, shoulder and/or neck muscle relaxation is set forth therein.
The use of oil of wintergreen (methyl salicylate), rosemary, aspic, juniper, lavender, nutmeg, pepper, mandarin, neroli, petitgrain, fennel, pine and thyme are disclosed as being useful in reduction of muscular pain by Daniele Ryman in "Aromatherapy: The Complete Guide to Plant and Flower Essences for Health and Beauty",© 1991 by Daniele Ryman at pages 242, 243, 244, 268, 269, 308, 335, 344 and 345; Published, 1991 by Piatkus Books. The Ryman publication does not disclose or suggest the method of inhalation therapy to induce muscle relaxation, and furthermore, no disclosure of back, shoulder and/or neck muscle relaxation is set forth therein.
The use of essential oils including Clary Sage and Jasmine in massage and baths is disclosed to have a relaxing effect on the muscles by Davis, P. in "Aromatherapy An A–Z", © Patricia Davis 1988, at pages 230 and 231; Published 1988 by The C. W. Daniel Company Limited, Saffron Walden, Essex, England. The Davis publication does not disclose or suggest the method of inhalation therapy to induce muscle relaxation, and furthermore, no disclosure or suggestion of back, shoulder and/or neck muscle relaxation is set forth therein.
Reduction of muscle tension "mainly neck and shoulders" by means of application via massage of:                Chamaemelum nobile         Juniperus communis         Lavandula augustifolia         Lavandula X intermedia 'Super'        Origanum marjorana and        Eucalyptus smithii [GULLY GUM]is disclosed by Price, S. and Price, L. in "Aromatherapy for Health Professionals" © Harcourt Publishers Limited 1999, Chapter 12, "Stress", pages 207–215. The Price, et al. publication does not, however, explicitly or implicitly disclose the method of inhalation therapy to induce muscle relaxation.
Accordingly, a need exists for a process for effecting the relaxation to (i) a substantially non stress-reactive state and (ii) a substantially non-tensive state of at least one stress-reactive and tensive back, shoulder or neck muscle of a human mammal for a substantial period of time, using one or more specifically-designed fragrance compositions to be conveniently administered by means of inhalation.

Process for effecting the relaxation of muscles of a human by means of fragrance


A power supply control apparatus includes: a semiconductor switching element switched with PWM control; a PWM signal output unit outputting a PWM signal; a current circuit outputting a current related to a current flowing through the semiconductor switching element; a filter circuit converting the current that is output from the current circuit to a voltage; an overcurrent protection circuit turning off the semiconductor switching element based on a voltage value of the voltage filtered by the filter circuit; a voltage detection unit detecting the voltage value of the voltage at a timing near an end of a pulse of a PWM signal; a temperature estimation unit estimating, a temperature of an electric wire through which a current flows that also flows through the semiconductor switching element; and an electric wire protection unit turning off the semiconductor switching element based on the temperature estimated by the temperature estimation unit.

In a vehicle, a power supply control apparatus is installed that supplies a power with a predetermined voltage value from a battery to a load via a semiconductor switching element in which switching is performed with PWM (Pulse Width Modulation) control. In this kind of power supply control apparatus, a microcomputer (or microcontroller) obtains a voltage value of the battery and calculates a duty ratio of the PWM control based on the obtained voltage value and the predetermined voltage value. Then, the microcomputer generates PWM (Pulse Width Modulation) signals based on the calculated duty ratio, and outputs the generated PWM signals. Based on the PWM signals that are output from the microcomputer, a driving circuit switches the semiconductor switching element on or off.
In this kind of power supply control apparatus, an overcurrent self-protection circuit protects the semiconductor switching element by switching off the semiconductor switching element according to a current value of a current flowing through the semiconductor switching element. Also, the microcomputer obtains the current value of the current flowing through the semiconductor switching element, and estimates, based on the obtained current value and a current flow-through time, a temperature of an electric wire through which a current flows that also flows through the semiconductor switching element. Based on the estimated temperature, the microcomputer turns off the semiconductor switching element to protect the electric wire.
FIG. 1 is a graph showing an example of an overcurrent protection characteristic of a conventional power supply control apparatus and a smoke-developing characteristic of an electric wire. In this graph, the value of the current flowing through the electric wire (flow-through current value) is marked on the vertical axis and time or continuous flow-through time of the flow-through current is marked on the horizontal axis. The waveform F shows a load current including an inrush current, with time being marked on the horizontal axis. Any point, given by flow-through current value and flow-through time, on the upper right of an electric-wire smoke-developing characteristic curve C leads to development of smoke in a material covering the electric wire develops smoke due to overheating. The electric-wire smoke-developing characteristic curve C shows an electric-wire temperature at which smoke development starts in the curve that shows the electric-wire temperature in the case where a current flows for the flow-through time.
In this power supply control apparatus, a curve D indicating a cut-off threshold value pertaining to a temperature calculation is defined for a slightly lower temperature (on the lower current side and shorter time side) of the electric-wire smoke-developing characteristic curve C.
In this case, as shown in the electric-wire smoke-developing characteristic C, the larger the flow-through current value is, the shorter the flow-through time for which the electric wire starts to develop smoke is. Also, time for calculating an electric wire temperature is needed. For this reason, the curve D showing the cut-off threshold value obtained by the temperature calculation is not set in the region in which the flow-through time is, for example, 10 milliseconds or shorter.
JP 4624400B discloses an electric wire protection apparatus in which a sense MOSFET (Metal Oxide Semiconductor Field Effect Transistor) in an IPS (Intelligent Power Switch) outputs a sense current that is proportional to a load current, and a microcomputer obtains a sense voltage value generated by the sense current having been current-voltage converted. The microcomputer periodically obtains the sense voltage value and calculates an electric wire temperature based on the obtained sense voltage value. If a flowing current increases due to a short-circuit of a wire harness and the like, a sense voltage rises, and thus the microcomputer determines that the electric wire is in an overheated state and turns off the semiconductor switching element before the calculated electric wire temperature rises and the electric wire starts to develop smoke.

Power supply control apparatus having a wire protection unit


An HCCI engine is operated by controlling a plurality of engine operating parameters in accordance with a calibration data set representing equilibrium set-points of engine operation characterized by combustion phasing that is relatively least sensitive to cylinder charge temperature deviations.

In homogeneous charge compression ignition (HCCI) engine, combustion is flameless, and spontaneously occurs at the entire cylinder volume. The homogeneously mixed cylinder charge is auto-ignited as the cylinder charge is compressed and its temperature increases. The ignition timing of auto-ignited combustion strongly depends on initial cylinder charge conditions such as temperature, pressure, and composition. Thus, it is important to coordinate the engine inputs, such as fuel mass, injection timing, and valve motion, to ensure robust HCCI combustion. Depending on the valve motion, there are two prevailing operating strategies in an HCCI engine—exhaust recompression strategy and exhaust re-breathing strategy.
In the exhaust recompression strategy, the cylinder charge temperature is controlled by trapping hot exhaust gas from the previous engine cycle by closing the exhaust valve early during the exhaust stroke, while opening the intake valve with late timing symmetrical to the exhaust valve closing timing. In this valve strategy, the cylinder charge composition and temperature depend on how early the exhaust valve closes during the exhaust stroke. For example, if the exhaust valve closes earlier during the exhaust stroke, more hot exhaust gas from the previous engine cycle would be trapped in the cylinder, leaving less cylinder volume for the fresh air mass, thereby, increasing the cylinder temperature, while decreasing the cylinder oxygen level. In the exhaust recompression strategy, the exhaust valve closing timing (thereby, the intake valve opening timing) is typically quantified by valve overlap, which has a negative number. The Negative Valve Overlap (NVO) is defined as the duration in crank angle between exhaust valve closing and intake valve opening. Thus, the initial cylinder charge conditions strongly depend on the intake and exhaust valve timings.
In contrast, in the exhaust re-breathing strategy, the cylinder charge temperature is controlled by re-inducting the hot exhaust gas into the cylinder through re-opening of the exhaust valve during the intake stroke. In this valve strategy, the cylinder charge composition and temperature depend on the lift of the exhaust valve re-opening during the intake stroke. Similarly with the exhaust recompression strategy, if the exhaust valve re-opens higher during the intake stroke, more hot exhaust gas from the previous engine cycle would be re-inducted in the cylinder, leaving less cylinder volume for the fresh air mass which, as a result, would increase the cylinder temperature, while decreasing the cylinder oxygen level. In this case, initial cylinder charge conditions strongly depend on the lift and/or duration of second opening of the exhaust valve.
Either with exhaust recompression strategy or re-breathing strategy, rapid change in inputs to the HCCI engine such as valve timing, EGR valve opening, injection timing, etc., is required to maintain sufficient thermal energy for successful auto-ignited combustions during load/speed transient. Thus, precise and fast actuator control is necessary for a successful transition between set-points, while those set-points are also required to be robust to disturbances that can be inevitably introduced during transient operation. Additionally, HCCI is sensitive to operating factors such as, for example, ambient temperature, engine coolant temperature, altitude, humidity, etc. making robust and stable control even more challenging.

Homogeneous charge compression ignition engine operation


A pulsed CO2 laser is Q-switched by an intracavity acousto-optic (AO) Q-switch including an AO material transparent at a fundamental wavelength of the laser. In one example the AO material is germanium.

High power Q-switched pulsed CO2 lasers have been manufactured in folded waveguide and folded free space Gaussian mode designs. Such a Q-switched CO2 lasers have optical characteristics attractive for a number of specialized, high-value applications in materials processing, such as via hole drilling, and printed circuit board (PCB) polymer ablation. A high-power Q-switched CO2 laser is also useful as an oscillator in a master-oscillator power-amplifier (MOPA) arrangement for use in plasma-EUV (extreme ultraviolet) radiation generation for photolithography. Such a laser combines very high peak instantaneous optical power, for example, about 10 kilowatts (kW) or greater, with modest average power, for example, about 10 watts (W) or greater, in a compact package.
One common method of Q-switching a high power CO2 laser involves use of an intra-cavity electro-optic (E-O) modulator. This provides very fast, for example, tens of nanoseconds (ns), optical switching, resulting in a laser with short optical pulses, for example about 100 ns FWHM (full width at half maximum), and high pulse repetition rate, for example about 100 kilohertz (kHz) or greater. To date, the only material with a suitable combination of high electro-optic coefficient, high bulk resistivity, and low infrared absorption is single-crystal cadmium telluride (CdTe). A pulsed CO2 laser including a CdTe E-O Q-switch is described in U.S. Pat. No. 6,826,204 assigned to the assignee of the present invention.
A complete reliance on CdTe for E-O Q-switching CO2 lasers presents two key issues. The first is high component costs which include the cost of the CdTe crystal itself and the cost of a high-speed, high-voltage modulator driver for driving the CdTe crystal. These component costs result in a cost of the finished pulsed CO2 laser many times higher than that of a continuous wave (CW) CO2 laser of comparable average power.
More significant, however, is that there is only a very limited supply of modulator quality CdTe crystals. For many years there has been only one vendor (Keystone Crystals Corporation, of Butler, Pa.) for such CdTe crystals. Other crystal-growers have attempted to grow CdTe crystals of the required quality but those attempts have been generally unsuccessful. This means in effect that there may be a long-term instability of supply for E-O switch quality CdTe crystals. This is troublesome when considering development of a large-scale commercial product. The high cost associated with problems in growing the CdTe crystals presents a significant barrier to further development of E-O Q-switched CO2 lasers and applying such lasers in material processing systems.
An alternate, relatively inexpensive approach to Q-switching a CO2 laser involves scanning a resonator end-mirror or a resonator fold-mirror such that the mirror sweeps reciprocally from one completely misaligned position to another through an optimally aligned position. This approach is described in U.S. patent application Ser. No. 11/638,645, filed Mar. 13, 2006, assigned to the assignee of the present invention and incorporated herein by reference. A problem with this approach is that the mirrors are preferably scanned at a characteristic resonant frequency to provide a suitable combination of scan-angle and sweep-speed. This limits the range of PRF available in such a laser, and accordingly, such a laser cannot be expected to have as flexible operating parameters as an E-O Q-switched CO2 laser. There is a need for a pulsed Q-switched laser that does not require a CdTe E-O Q-switch, but that can be Q-switched at comparable rates and with the same flexibility as a CdTe E-O Q-switched CO2 laser.