Automatic coin input apparatuses for use with consumer coin counting machines and/or other coin processing machines are disclosed herein. In one embodiment, a coin bowl structure includes a rotatable disk configured to support a plurality of randomly oriented coins deposited thereon. In operation, rotation of the disk in a first direction can automatically drive a first portion of the coins deposited thereon out of the bowl structure through a first coin passage, and rotation of the disk in a second direction opposite to the first direction can automatically drive a second portion of the coins deposited thereon out of the bowl structure through a second coin passage different than the first coin passage.

Various types of coin counting machines are known. Some coin counting machines (e.g., vending machines, gaming devices such as slot machines, and the like) are configured to receive one coin at a time through a slot. These machines are relatively simple and typically designed for relatively low throughput and little, if any, coin cleaning. Such machines, however, are usually ill-suited for counting large quantities of consumer coins received all at once (such as a large quantity of coins poured into a machine from, e.g., a coin jar).
Machines for counting and/or sorting relatively large quantities of consumer coins include those disclosed in, for example, U.S. Pat. Nos. 5,620,079, 7,028,827, 7,520,374, and 7,865,432, each of which is incorporated herein by reference in its entirety. Some of these machines count consumer coins and dispense redeemable cash vouchers, while others may offer other types of products and services either exclusively or in addition to vouchers. Such products and services can include, for example, dispensing and/or topping-up prepaid cards (e.g., gift cards, phone cards, etc.), "e-certificates," and the like, and transfers to online accounts (e.g., Paypal™), mobile wallets, etc. Vouchers can be redeemed for cash and/or merchandise at a point of sale (POS) in a retail establishment, while e-certificates can enable the holder to purchase items online by inputting a code from the e-certificate when making the purchase. Prepaid gift cards can be used to make POS purchases by, for example, swiping the card through a conventional card reader, and prepaid phone cards can be used for making cell phone calls. The term "mobile wallet" can refer generally to an electronic commerce account implemented by a mobile phone or other mobile wireless device. In some embodiments, mobile wallets store "virtual gift cards," virtual loyalty cards, etc.; transfer value; and/or conduct transactions for, e.g., purchasing goods and/or services from suitably enabled merchants. The term "virtual gift card" can refer to an application program operating on the mobile device that performs like a prepaid card, such as a gift card. Virtual gift cards can enable the user to wirelessly purchase items and/or services, pay bills, and/or conduct other transactions with retailers and other merchants via, e.g., a wirelessly enabled point of sale (POS) terminal, the Internet, and/or other computer networks.
Some coin counting and/or sorting machines include a hinged coin input tray that is manually lifted by the user to introduce their coins into the machine for processing. Such an input tray is disclosed in, for example, U.S. Pat. No. 5,620,079. When at rest, the input tray is angled downward and away from a raised hinge line that forms a slight peak. This prevents coins in the tray from flowing into the machine until the user begins rotating the tray upwardly about the peak. As the user continues lifting the input tray, the coins begin to slide out of the tray, over the peak and into the machine for counting and/or sorting. In some instances, the user may be required to use their hands to manually control the flow of coins out of the input tray. For example, if the user lifts the tray too fast, the user may need to place their hands near the peak to prevent coins from leaving the input tray too quickly and jamming the machine. On the other hand, if the user lifts the tray too slowly, the user may need to move some coins out of the tray and over the peak by hand. In either case, user involvement may be necessary to facilitate the coin input process. U.S. Pat. No. 6,602,125, which is incorporated herein by reference in its entirety, disclosed an automatic coin input tray for a self-service coin-counting machine. The input tray employed a spring-loaded rotating disk that would drop if the user poured in more coins than the tray could initially process. This dropping feature can make it difficult to adequately seal gaps between the rotating disk and the surrounding coin bowl.
Speed and accuracy are important considerations in self-service coin counting machines. Consumers are less inclined to use a coin counting machine if they have to wait an appreciable amount of time to have their coins counted. Coin counting machines should also be accurate and relatively easy to operate to encourage use. Accordingly, it would be advantageous to provide coin counting machines with coin input systems that are relatively easy to use, and facilitate accurate and relatively fast counting of large quantities of coins.

Coin input apparatuses and associated methods and systems


A linear sliding guide comprises a longitudinal guide rail with guide flanks that extend in a direction of sliding and a bearing block that lodges therein and has corresponding counterpart flanks. Compensation-for-play spring assemblies are provided between these two components; they are formed by a receptacle and a spring bushing that is inserted therein. The spring bushing acts on the corresponding guide flank of the longitudinal guide rail by way of a gap in the receptacle.

1. Field of the Invention
The invention relates to a linear sliding guide, in particular for inter-adjustable components of office chairs, comprising a longitudinal guide rail with guide flanks which extend in a direction of sliding; a bearing block which lodges in the longitudinal guide rail and has counterpart flanks that are parallel to the guide flanks, with the longitudinal guide rail and the bearing block being inter-adjustable in the direction of sliding; at least one compensation-for-play spring assembly between the longitudinal guide rail and the bearing block, the spring assembly being comprised of a substantially cylindrical receptacle, which lodges in the longitudinal guide rail or in the bearing block, and which is disposed in a marginal area of the longitudinal guide rail or bearing block with its cylinder axis being parallel to the respective flank, and the clear cross section of which is laterally open by a gap towards the respective flank; and an elastically deformable spring bushing which is inserted axially parallel in the receptacle such that it acts via the gap on the opposite flank of the bearing block or longitudinal guide rail.
2. Background Art
As regards the background of the invention, modern office chairs are equipped with multiple possibilities of adjustment of, for example, backrest height, seat depth, armrest height, lengthwise armrest position etc. The linear sliding guides used therefor have a longitudinal guide rail with guide flanks that run in the direction of sliding, with a skid-type bearing block for the to-be-adjusted part being displaceably guided in the guide rail. The bearing block is provided with counterpart flanks that run parallel to the guide flanks of the longitudinal guide rail.
A problem of these linear guides resides in that on the one hand the longitudinal guide rail and the bearing block that runs thereon must have a certain play for smooth sliding to be possible. On the other hand, this play leads to a degree of freedom within the sliding guide, which may result in the sliding part wobbling. This does not only lead to increased wear of the sliding parts, but is felt to be of inferior quality by users of these office chairs. Adjustable parts are supposed to fit properly and stably while a chair is used and to run smoothly with close fit when sliding.
So as to get rid of the above problems, prior art attempts have been made to produce a sort of compensation for play and tolerance by providing nubs in the shape of warts on the flanks of the components involved of the linear sliding guide. However, these nubs will produce point contact with the actuated flank, which may lead to rapid wear in particular upon frequent adjustment.

Linear sliding guide, in particular for inter-adjustable components of office chairs


A plurality of memory cells are connected between two adjacent sub-bit lines. A row decoder 3 selects a word line connected to a memory cell to be read. A selection line selection circuit 2 and a column selection circuit 5 include first and second selection portions that perform selection operations simultaneously and independently. The first selection portion selects a first pair of main bit lines and selection lines in order to select the memory cell to be read. The second selection portion selects a second pair of main bit lines that is different from the first pair of main bit lines and selection lines for selecting a sector different from that for the memory cell to be read in order to select a line to be used for reading a reference voltage.

1. Field of the Invention
The present invention relates to a non-volatile semiconductor memory device. More specifically, the present invention relates to a non-volatile semiconductor memory device having a matrix-shaped memory cell area including a plurality of memory cells arranged in a plurality of rows and a plurality of columns.
2. Description of the Background Art
As one configuration of a non-volatile semiconductor memory device, VGA (virtual ground array) is known. The VGA configuration refers to a configuration in which adjacent memory cells share a diffusion layer. The area of the non-volatile semiconductor memory device can be reduced by adopting the VGA configuration. FIG. 17 shows a non-volatile semiconductor memory device having the VGA configuration. The non-volatile semiconductor memory device shown in FIG. 17 has the VGA configuration and has a hierarchical bit line structure including main bit lines and sub-bit lines.
The non-volatile semiconductor memory device shown in FIG. 17 is divided into three areas A to C. FIG. 17 shows main bit lines MOi, MEi, MOi+1, MEi+1, and sub-bit lines 91Ai to 94Ai, 91Ai+1 to 94Ai+1, 91Bi to 94Bi, 91Bi+1 to 94Bi+1 as the lines in the column direction (vertical direction). FIG. 17 also shows word lines AWLB, BWLB, and selection lines SEL1U to SEL4U, SEL1L to SEL4L as the lines in the row direction (horizontal direction). The word line AWLB is connected to the gates of memory cells AP to AV, and the word line BWLB is connected to the gates of memory cells BP to BV. Selection transistors, which are switches for controlling connection between the main bit lines and the sub-bit lines, are disposed between the main bit lines and the sub-bit lines, if necessary. The main bit lines and the sub-bit lines form a hierarchical structure with the main bit lines at the top and the sub-bit lines at the bottom.
The operation of the non-volatile semiconductor memory device shown in FIG. 17 that is configured as above will be described. As an example, the case where "a read on the source side" with respect to a memory cell BQ is performed will be described. A memory cell storing data "0" is referred to as "being in a program state", and a memory cell storing data "1" is referred to as "being in an erase state". In order to perform a read on the source side with respect to a memory cell, it is necessary to apply voltages of about 4.5 V, about 0 V (substantially ground voltage) and about 1 V to the gate, the source and the drain of the memory cell, respectively.
As shown in FIG. 17, the drain of the memory cell BQ is connected to the sub-bit line 92Bi, and the sub-bit line 92Bi is connected to the main bit line MEi via a selection transistor 95. The source of the memory cell BQ is connected to the sub-bit line 93Bi, and the sub-bit line 93Bi is connected to the main bit line MOi via a selection transistor 96. Therefore, in order to perform a read on the source side with respect to a memory cell BQ, a voltage of about 3 V is applied to the selection lines SEL2L, SEL1U to control the selection transistors 95, 96 to be ON, and voltages of about 1 V, about 0 V and about 4.5 V can be applied to the main bit lines MEi, MOi and the word line BWLB, respectively. Thus, voltages of about 4.5 V, about 0 V (substantially ground voltage) and about 1 V can be applied to the gate, the source and the drain of the memory cell BQ, respectively.
When a predetermined voltage is applied to each line in this manner and the data stored in the memory cell BQ is "0", the threshold value of the gate voltage is high, so that almost no current flows from the main bit line MEi to the main bit line MOi. Consequently, the potential of the main bit line MOi is not substantially increased. On the other hand, when the data stored in the memory cell BQ is "1", the threshold value of the gate voltage is low, so that a current flows from the main bit line MEi to the main bit line MOi. Consequently, the potential of the main bit line MOi is increased by a voltage corresponding to the current that has flown. Therefore, data can be read from the memory cell BQ by detecting the voltage of the main bit line MOi.
Whether the data read from the memory cell BQ is "0" or "1" can be determined in the following manner. A sense amplifier (not shown) amplifies and outputs a difference between a voltage of the main bit line MOi during a read operation and a predetermined voltage that serves as a reference (hereinafter, referred to as "reference voltage"). When the voltage of the main bit line MOi is higher than the reference voltage, it is determined that the read data is "1". On the other hand, when the voltage of the main bit line MOi is lower than the reference voltage, it is determined that the read data is "0".
A reference cell is used to generate the reference voltage as described above. The reference cell is designed to output the reference voltage when it is controlled to be ON. The reference cell is constituted by, for example, a MOS transistor. Such a reference cell is often disposed outside the memory cell area in the non-volatile semiconductor memory device as shown in FIG. 17.
Data read from one memory cell is output to the outside of the memory cell area via the main bit line and the sub-bit line connected to the memory cell. The main bit lines and the sub-bit lines used for reading data are associated with a parasitic capacitance in accordance with the lengths of the bit lines, whether or not there are other bit lines adjacent to the bit lines, or the number of connected memory cells or the like. This parasitic capacitance is a factor that may delay a data read operation. More specifically, when the parasitic capacitance is large, the delay amount during a data read becomes large, and when the parasitic capacitance is small, the delay amount during a data read becomes small.
As described above, the reference cell is often provided in a predetermined position outside the memory cell area. Therefore, unlike the memory cell, it is difficult to connect the reference cell to the main bit lines, which are connected to a plurality of memory cells and the sub-bit lines. Therefore, the line connected to the reference cell is not associated with a parasitic capacitance that is equal to the parasitic capacitance associated with the main bit line connected to the memory cell. Consequently, a phenomenon occurs in which, the parasitic capacitance generated on the side of the memory cell to be read as seen from the sense amplifier does not match the parasitic capacitance generated on the side of the reference cell (this phenomenon is referred to as "capacitance unbalance"). When such capacitance unbalance occurs, the delay amount when the voltage of the main bit line connected to the memory cell is output does not match the delay amount when the reference voltage is output from the reference cell.
Regarding this problem, a method for equalizing the parasitic capacitance associated with the bit line on the memory cell side as seen from the sense amplifier with the parasitic capacitance associated with the line on the reference cell side is known. The method will be described in detail with reference to FIG. 17 below.
As described above, when reading data from the memory cell BQ, the main bit lines MOi, MEi, and the sub-bit lines 92Bi, 93Bi are used. Therefore, when reading data from the memory cell BQ, a parasitic capacitance in accordance with the length of each of the bit lines is associated therewith. Then, the line with which a parasitic capacitance equal to the parasitic capacitance associated with each of the bit lines is associated can be connected to the reference cell. Thus, the parasitic capacitance associated with the bit line connected to the reference cell can be equal to the parasitic capacitance associated with the bit line connected to the memory cell BQ.
More specifically, the same voltage as the voltage applied to the main bit line and the sub-bit line used for a data read is applied to the main bit line and the sub-bit line that are not used for a data read, and these bit lines can be connected to the reference cell. For example, when reading data from the memory cell BQ, voltages of about 0 V and about 1 V are applied to the main bit lines MOi+1, MEi+1, respectively (thus, voltages of about 0 V and about 1 V are also applied to the sub-bit lines 93Bi+1, 92Bi+1, respectively), and the main bit lines MOi+1, MEi+1 can be connected to the reference cell. Thus, the bit line connected to the reference cell is associated with the same parasitic capacitance as that of the bit line connected to the memory cell BQ. Regarding the above-described technique, U.S. Pat. Nos. 5,963,465 and 6,351,415 (hereinafter, referred to as "Reference 1" and "Reference 2") can be referred to.
However, the method for allowing the parasitic capacitance to be associated in the above-described manner has the following problem. Data is read from a memory cell connected to the same word line as the selected memory cell (e.g., memory cell BU in the case where the memory cell BQ is selected), and thus the non-volatile semiconductor memory device may malfunction.
In order to solve this problem, a non-volatile semiconductor memory device shown in FIG. 18 is known. FIG. 18 is a diagram showing the configuration of a non-volatile semiconductor memory device disclosed in U.S. Pat. No. 6,128,226 (hereinafter, referred to as "Reference 3"). The non-volatile semiconductor memory device shown in FIG. 18 will be described.
The non-volatile semiconductor memory device shown in FIG. 18 includes memory cells, bit lines BL, word lines WL, reference bit lines BLR, a Y decoder 1000, a reference unit 1002 and a sense amplifier 1004. The memory cells are arranged in a matrix. The bit lines BL are arranged in the column direction between the memory cells arranged in a matrix. The words lines WL are arranged in the row direction between the memory cells arranged in a matrix. The reference bit lines BLR are lines with which a parasitic capacitance equal to the parasitic capacitance associated with the bit lines BL is associated during a data read operation. The Y decoder 1000 connects the bit line BL to the sense amplifier 1004, the memory cell to be read being connected to the bit line BL. The reference unit 1002 generates a reference voltage Vref that serves as a reference. The sense amplifier 1004 amplifies and output a difference between the voltage Vcell of the bit line BL output from the Y decoder 1000 and the reference voltage Vref.
In the non-volatile semiconductor memory device shown in FIG. 18, a data read is performed in the following manner. First, when reading data from a certain memory cell, data is read out to the Y decoder 1000, using two bit lines BL connected to the diffusion layer of this memory cell and a word line WL connected to the gate of this memory cell. The Y decoder 1000 outputs the voltage Vcell of the bit line BL connected on the drain side to the sense amplifier 1004.
On the other hand, the reference unit 1002 generates the reference voltage Vref, and outputs it to the sense amplifier 1004. In this case, two reference bit lines BLR are selected, and connected to the reference unit 1002. Thus, the parasitic capacitance associated with the bit lines BL connected to the memory cell to be read can be equal to the parasitic capacitance associated with the reference lines BLR connected to the reference unit 1002. That is to say, the problem of capacitance unbalance can be solved. Consequently, the delay amount in reading from the memory cell can be equal to the delay amount in reading from the reference unit 1002.
Regarding the non-volatile semiconductor memory device, U.S. Pat. No. 6,272,043 (hereinafter, referred to as "Reference 4") discloses a method for performing a read on the source side in a non-volatile semiconductor memory device as shown in FIG. 19. In the non-volatile semiconductor memory device shown in FIG. 19, a reference circuit 2045 includes a reference current source 2040 and a current/voltage converting circuit 2050, a memory arrangement 2015 includes a virtual ground array 2000 and a current/voltage converting circuit 2030. The current/voltage converting circuit 2050 is constituted by a resistance element and converts a current generated in the reference current source 2040 to a reference voltage 2055. The current/voltage converting circuit 2030 is constituted by a resistance element and converts a current output from a memory cell in the virtual ground array 2000 to a read voltage 2065. A comparing circuit 2060 is constituted by a differential amplifier and compares the reference voltage 2055 and the read voltage 2065. Thus, the state of the memory cell in the virtual ground array 2000 can be determined.
However, the non-volatile semiconductor memory device shown in FIG. 18 has a problem in that it is difficult to reduce the size of the circuit. More specifically, in the non-volatile semiconductor memory device shown in FIG. 18, extra reference bit lines BLR for generating a read delay in the reference cell are provided. Therefore, it is necessary to provide an area in which the reference bit lines BLR are provided in the non-volatile semiconductor memory device.

Non-volatile semiconductor memory device

Matsushita Electric Industrial Co., Ltd.


A novel catalyst support and a method for preparation thereof, characterized by the presence of selected amounts of a hydrocarbon soluble magnesium-containing compound applied to the support relative to solvent, wherein a surface uniformly and essentially continuously moistened with the solvated magnesium compound in a condition of incipient precipitation to permit the formation of a regular essentially continuous distribution of the magnesium compound onto and conforming to the support material. The drying process under these conditions is effective in maintaining to a high degree the effective surface area and pore volume of the material. The support containing the adsorbed magnesium is then dried in a controlled manner to essential incipient wetness conditions providing a regular spatial distribution of stacked reticular magnesium values at a level of one to five molecular monolayers on the support.. A representative process for polymerizing or copolymerizing olefins providing stereospecific polymers such as propylene is disclosed.

Olefin polymerization using Ziegler-Natta type catalysts has been widely utilized in industry. These catalysts provide polyolefins in high yield possessing the desired physical characteristics of these polymers. However, the use of Ziegler-Natta catalysts do suffer some serious limitations. Thus, new and improved catalysts are continually being sought and developed. An important class of catalysts where improvement is sought is for the polymerization of the very commercially important alphaolefin, propylene.
Commonly in the polymerization of propylene a catalyst containing a magnesium halide support is utilized. However, when propylene is polymerized from such a supported catalyst, the polypropylene molding machine processing this polymer is subjected to extensive corrosion, or other damage, directly caused by the presence of residual halide in the polypropylene product. More importantly, the polypropylene molded article processed in this manner is characterized by aesthetic flaws.
Another serious drawback of art recognized catalysts, in the polymerization of propylene polymers, involves the required use of internal electron donors to insure that the propylene polymer product is highly isotactic. However, unless the amount and type of electron donor compound is carefully selected, not only is the stereoregularity of the resultant polymer deficient but poor catalytic activity often results. This detrimental effect will still occur even if the catalyst is formed with the roper electron donor compound added in the wrong sequence.
The utilization of electron donor compounds often creates additional problems involving the evolution of offensive odors in the final polymeric product. This offensive odor will also be present even if the ideal electron donor compound, in the correct concentration, added at the proper time in the catalyst formation process, is utilized. Thus, polymers formed from these catalysts which include an electron donor compound must oftentimes be deashed or deodorized in order to insure that the final product does not give off any offensive odors.
These difficulties discussed herein has lead workers skilled in the art to produce and develop new catalysts which are aimed to overcome these difficulties.
Drying of catalyst components, especially catalytic supports, is common in the industry. For the supports alone, calcining which implies heating under a nitrogen atmosphere is typically employed to control the presence of adsorbed moisture with these hygroscopic materials, at least in part to avoid hydrolytic reactions with other catalyst reactants. Moisture present in the support is widely recognized as one of several substances which may act as catalyst poisons. For these very reasons, however, such drying is carried out under stringent conditions to essential dryness. See e.g. U.S. Pat. No. 4,593,079.
Short et al. U.S. Pat. No. 4,565,795 discloses an olefin polymerization catalyst which is prepared by the reaction of a chemically treated silica support with a dihydrocarbyl magnesium compound and a halogenated tetravalent titanium compound The solvent for the dihydrocarbyl magnesium compound was removed by evaporation.
U.S. Pat. No. 4,530,913 of Pullukat et al. discloses a method of preparing a solid polymerization catalyst component by reacting a modified support with an organomagnesium compound and a tetravalent titanLum compound with no intermediate drying step. The solvent was subsequently evaporated.
Tachikawa et al. U.S. Pat. No. 4,686,199 describes a catalyst component formed from the reaction of a metal oxide with one or more non-halogenated magnesium alkoxide, a halogen-containing compound, and a titanium compound. Excess inert solvent associated with the magnesium compound is removed in Example 1 under nitrogen by vaporization, and after contact with another magnesium compound the solid was dried in vacuo at 90.degree. C. for 2 hours.
McDaniel et al. U.S. Pat. No. 4,855,271 describes a catalyst component prepared by contacting a support comprising highly porous alumina containing a magnesium compound with an alcohol and subsequently with a halide, alkoxide, or haloalkoxide of titanium and/or vanadium. Once the magnesium compound is impregnated onto the alumina support under the conditions of incipient wetness, drying conditions are selected so as to minimize reduction in the porosity of the support. Typically, this involves heating at a temperature of about 30.degree. C. to about 800.degree. C. More preferably, the drying temperature is from about 100.degree. C. to about 400.degree. C.
By way of background to the invention, U.S. Pat. No. 4,950,631 including inventor Buehler discloses a process for propylene polymerization using a modified silica based catalyst. The catalyst is prepared by first pretreating the silica support either by calcination or chemical treatment with a disilazane compound followed by modifying the silica by reacting it with a hydrocarbon soluble magnesium compound. In this case, the solid product was allowed to settle and the supernatant siphoned off. This modified support is then made chemically active by reacting with a titanium compound having the formula, Ti(OR).sub.n X.sub.m, wherein R is an aryl, alkyl or any mixtures thereof containing C.sub.1 -C.sub.28 atoms, X is a halogen, n=1 to 4 and m=0 or 1 to 3 such that n+m=4. This catalyst precursor is then reacted with another titanium compound having the formula TiX.sub.p '(OR').sub.q where X' is a halogen, R' is an aryl or alkyl group, p=1 to 4; and q=0 or 1 to 3 such that the sum of p and q equals 4. The final catalyst was dried by evaporation to a free flowing solid. Additionally, cocatalyst containing an aluminum compound and/or a silane compound can be used.
The significance of the drying and washing steps in relation to solvents in these catalysts became increasingly apparent to the present inventor(s), such that in copending and commonly assigned U.S. Ser. No. 590,992 of Buehler, Fries and Pullukat filed Oct. 10, 1990 incorporated herein by reference, a "minimum level" of solvent for the magnesium compound is mentioned This application also observes at page 23, lines 28-33 that underdrying or overdrying may be deleterious and expresses preference for drying after reaction with the magnesium compound at 70-80.degree. C. for 11/2 to 21/2 hrs. Then, in copending and commonly assigned U.S. Ser. No. 717,956 of Buehler and Masino filed Jun. 20, 1991 incorporated herein by reference, a drying step is disclosed in Example 19 to remove 67% by weight of the solvent from the magnesium compound.
It has been surprisingly discovered upon further consideration and elucidation of the parametric preparative conditions that drying to a selected solvent level, as it controls the nature of micro-deposition in and upon the surface of relatively high surface area, high pore volume inorganic supports, with applied hydrocarbon soluble magnesium compounds is a key variable in providing desired catalyst characteristics and in turn improved polymer properties in preparation and use. In accordance with the invention, simplified preparation of highly active propylene polymerization catalysts is afforded without internal electron donors, magnesium halide supports, or multiple reactants.

Supported high activity polyolefin catalyst component with regular distribution of magnesium values provided utilizing a controlled drying protocol


An infrared sensor element having an antenna coupled directly into a rectifier. Infrared radiation impinging the antenna induces an alternating current. The rectifier converts the alternating current into a rectified signal. The rectified signal corresponds to a magnitude of the incident infrared radiation impinging the antenna, and can be used for detection and imaging operations. Coupling the antenna directly into the rectifier eliminates the need for a photodetector. The wavelength of sensor elements can be separately tuned. A sensor made up of a plurality of such sensor elements can operate in a time-division multiplexed mode sensing different infrared bandwidths in separate time segments. Two or more sensors each made up of a plurality of sensor elements can be co-located in a single focal plane and operated separately to detect infrared radiation in different bandwidths. A sensor element can also operate in bandwidths other than infrared.

Infrared-based imaging and detection devices have found widespread use and are known to those skilled in the art as "FLIRs" (an acronym derived from "Forward-Looking Infrared"). Since infrared-based imaging devices sense temperature differences between objects in a field of view, they are particularly useful at night and in daytime during periods of reduced visibility. In contrast to the visible spectrum, the infrared spectrum frequently provides high contrast images at night and during periods of reduced visibility. This results from a number of factors, a first factor being that infrared-based imaging devices sense temperature differences between objects in a field of view. At night and during daytime periods of reduced visibility there is frequently a significant thermal contrast between objects of interest (such as, for example, vehicles and persons) and a background. A second factor results from the fact that during periods of reduced visibility atmospheric obscurants (such as, for example, smoke or sand) may attenuate visible light to a far greater degree than infrared radiation.
Infrared-based imaging devices use infrared detectors to detect infrared radiation emanating from objects in a field of view. Conventional infrared detectors are constructed from exotic semiconductor materials such as HgCdTe and InSb. A particular limitation of conventional infrared detectors is their need to be cooled to cryogenic temperatures in order to achieve desired levels of thermal sensitivity. The requirement for active cooling increases the cost, complexity and power consumption of such infrared imaging devices, and distinguishes them from visible-spectrum video cameras that do not require such active cooling. In addition, the need for cooling apparatus introduces complexities into the design of optical elements used in combination with the infrared detectors of infrared-based imaging devices.
Infrared-based imaging devices using detectors that are not cryogenically cooled are known, but lack fast response times for operation in dynamic situations. Uncooled sensors also lack narrow-bandwidth spectral responses for surveillance and identification.
For purposes of surveillance and identification, it is often useful to limit the response of a sensor to a particular bandwidth. If an object that is to be detected is known to have a unique thermal signature because it emanates infrared radiation of a particular wavelength or combination of wavelengths, tuning the response of the sensor to the particular wavelength or combination of wavelengths serves a filtering function by eliminating other objects that have different thermal signatures from consideration.
Progress has been made in providing tunability for infrared detectors through the introduction of antenna-coupled infrared detectors. In antenna-coupled infrared detectors, an antenna element sensitive to infrared radiation is coupled to a conventional infrared photodetector. Such antenna-coupled infrared photodetectors are frequency-tunable and may have an adjustable polarization response. Nonetheless, infrared-based imaging or detection devices that are to be used in dynamic situations require the use of cooled detectors, thereby increasing the cost and complexity of such devices.
Accordingly, those skilled in the art desire improved designs for infrared sensor elements and sensors that eliminate the need for cooling; achieve or exceed the sensitivity of conventional infrared detectors; and preserve the advantages of antenna-coupled detectors.

Antenna-coupled-into-rectifier infrared sensor elements and infrared sensors


An integrated semiconductor circuit has a component formed of a semiconductor substrate with an active pn junction formed between a first semiconductor region of a first conductivity type and a second semiconductor region of a second conductivity type. The semiconductor circuit also has a protective circuit that is associated with the component and serves to dissipate overvoltages and/or electrostatic charges. The protective circuit has a protective pn junction formed in a semiconductor mount with a first semiconductor mount region of the first conductivity type and a second semiconductor mount region of the second conductivity type. The second semiconductor mount region of the second conductivity type is electrically coupled to the first semiconductor region formed in the semiconductor substrate of the first conductivity type.

1. Field of the Invention
The invention relates to an integrated semiconductor circuit having a component formed of a semiconductor substrate with a first semiconductor region of a first conductivity type and a second semiconductor region of a second conductivity type forming an active pn junction therebetween. The integrated semiconductor circuit also has a protective circuit assigned to the component for serving to dissipate overvoltages and/or electrostatic charges.
Integrated semiconductor circuits, in particular fast optoelectronic transmission components such as semiconductor laser circuits or infrared light-emitting diodes are sensitive to electrostatic charges (ESD-Electrostatic Discharge) or voltage spikes in the reverse direction of the pn junction of the transmission component, which may result in damage to, or the destruction of, the component. In order to avoid ESD damage, the known integrated semiconductor circuit has protective circuits formed on the semiconductor substrate of the component and serve to dissipate overvoltages. In particular in the form of protective diodes, which are connected, for example, between a connection point of the active pn junction of the component and earth. The protective circuit is disposed in the immediate vicinity of the component that is to be protected against overvoltages, to be precise on the same semiconductor substrate as the component. The production of the protective circuit on the same substrate can lead to difficulties particularly in the case of optoelectronic transmission components having a semiconductor substrate made of a III-V compound such as, for example, gallium arsenide, indium phosphide and the like. On the one hand, the space available for forming the protective circuit on the semiconductor substrate may be restricted for layout reasons. Furthermore, optoelectronic transmission components, in particular, require, on account of the generally high transmission power, measures for ensuring good dissipation of heat during the operation of the transmission component. In an unfavorable manner, semiconductor materials made of a III-V compound have a much poorer thermal conductivity than silicon, with the result that, in many of such transmission components, a mount which is thermally coupled to the transmission component chip and is made of a material having a low thermal resistance and a large area compared with the chip area of the transmission component is employed in order to ensure good heat dissipation.
From Japanese Patent Abstract No. JP-A-59 188 181 there is disclosed an integrated semiconductor circuit in which a protective diode formed on the semiconductor mount is connected parallel to the active pn-junction. In this manner, the semiconductor mount is to function as a protective circuit for the active component after mounting a reverse voltage on the active pn-junction.


A large scale integrated circuit for a build off self-test and a device to be tested are mounted in a single socket so that their respective electrodes are in direct contact.

1. Field of the Invention
The present invention relates to a build off self-test (BOST) testing method. More particularly, the invention relates to a method for performing tests at high speeds with high precision, and to a socket and a semiconductor device for use with that method.
2. Description of the Background Art
One way to facilitate device testing procedures has been known conventionally as BOST (Build Off Self Test). The BOST involves mounting on a test board an LSI or circuits for expediting examinations of a device under test (called the DUT hereunder, including an LSI or the like to be tested in bare-chip or packaged form). As such, the BOST is designed to carry out tests on high-performance multi-function DUTs by use of an unsophisticated low-performance tester.
In a typical testing setup, a high-frequency oscillator is mounted on the test board so that a clock signal generated thereby is used to perform tests on a DUT at frequencies higher than the tester frequency. In another setup, an operation amplifier is mounted on the test board so as to test analog signals.
FIG. 14 is a schematic view of a setup in which to practice a conventional BOST method. Referring to FIG. 14, an LSI 1008 for use with the BOST (LSI or the like under test in bare-chip or packaged form, called the BOST LSI hereunder) receives a simple control signal from a tester 1001. In turn, the BOST LSI 1008 generates test signals necessary for testing a DUT 1007, checks to see if output signals from the DUT 1007 are correct, and notifies the tester 1001 of the results of the tests on the DUT 1007.
The BOST LSI 1008 and DUT 1007 are mounted on different sockets 1012 attached to a test board 1003 and exchange relevant signals through printed wiring 1010. The BOST LSI 1008 and DUT 1007 use paths indicated by broken lines in the printed wiring 1010 in order to receive from the tester 1001 the power necessary for their operations. The tester 1001 thus causes the BOST LSI 1008 to operate.
Furthermore, the DUT 1007 uses the printed wiring 1010 indicated by solid lines to exchanges signals with the BOST LSI 1008 for testing.
Part of the signals input and output to and from the DUT 1007 may be exchanged directly with the tester 1001. This arrangement is intended to alleviate loads on the BOST LSI 1008.
In the conventional BOST setup outlined above, the BOST LSI 1008 and DUT 1007 are mounted on different sockets 1012 and exchange signals therebetween via the printed wiring 1010. This has led to problems such as a mismatch of impedance due to contactors 1006 of the different sockets 1012, or signal delays over the printed wiring 1010.
The present invention has been conceived to solve the previously mentioned problems, and general a object of the present invention are to provide a novel and useful testing method, a socket for use with the method, and a semiconductor device for use with the method.
The above object of the present invention is achieved by a testing method for use with a BOST setup. In the method, there are mounted on a single socket a BOST semiconductor device and a semiconductor device under test. The semiconductor device under test is tested using the BOST semiconductor device.
The above object of the present invention is also achieved by a socket for use in a BOST setup. The socket includes a socket body with a space to accommodate a first semiconductor device. A first cap with a space to accommodate a second semiconductor device is provided so as to push electrodes of the first semiconductor against contactors. A second cap pushing electrodes of the second semiconductor device against the electrodes of the first semiconductor device is also provided.
The above object of the present invention is further achieved by a semiconductor device for use in a BOST setup described below. The semiconductor device includes a BGA structure having conductive elastic elements attached to electrodes instead of solder balls.
Other objects and further features of the present invention will be apparent from the following detailed description when read in conjunction with the accompanying drawings.

Build off self-test (Bost) testing method


A quadrature amplitude modulator capable of avoiding compensating the linear distortions completely and accurately by a simple configuration. The modulator includes a level signal generation unit for generating level signal for the modulated wave, and a parameter generation unit for determining appropriate values of linear transformation parameters of the linear transformation to be applied to the in-phase and quadrature amplitude input signals according to the level signal. In this modulator, the linear transformation parameters are determined sequentially in such an order that the linear transformation parameters for a DC off-set are determined first, then the linear transformation parameter for an amplitude balance is determined next while compensating distortion due to the DC off-set, and then the linear transformation parameter for an orthogonal phase relationship is determined while compensating distortions due to the DC off-set and the amplitude balance.

1. Field of the Invention
The present invention relates to a quadrature amplitude modulator with distortion compensation to be used in a communication such as a radio communication.
2. Description of the Background Art
In a communication such as a radio communication, a modulator is used for generating narrow band signals such as multi-level PSK (Phase Shift Keying) and QAM (Quadrature Amplitude Modulation).
In particular, in a case of digital modulation and demodulation schemes to be utilized in digital mobile communications, a modulator with higher accuracy is required compared with analog modulation and demodulation schemes because the digital schemes utilize waveform transmission instead of spectral transmission of the analogue schemes. For this reason, such a digital modulation and demodulation scheme uses a quadrature modulator to which an in-phase amplitude and a quadrature amplitude are applied as baseband signal inputs. However, a conventional quadrature modulator has the following problems.
(i) A carrier leak component is superposed onto the modulated wave even when the off-set levels of the baseband signal inputs are balanced, because it is difficult to balance the baseband signal inputs as the effective inputs inside the quadrature modulator.
(ii) An image component is superposed onto the modulated wave even when the amplitudes of the baseband signal inputs are balanced, because it is difficult to balance the baseband signal amplitudes as the effective inputs inside the quadrature modulator.
(iii) The quadrature modulator requires two input carrier signals with 0 phase and .pi./2 phase. It is, however, difficult to manufacture an accurate phase shifter for generating the .pi./2 carrier signal on an IC. When this orthogonal phase relationship between two carrier signals is not accurate, an image component is superposed onto the modulated wave.
(iv) When the quadrature modulator is implemented on an IC, its operational characteristics vary according to the variations of the power source voltage and the surrounding temperature.
In conventional quadrature modulators, these problems are solved by skillful adjustments of manufacturers. As for the variation due to aging, the necessary adjustment is provided by a regular maintenance service.
In order to adjust such a distortion compensation of the quadrature modulator automatically, two simple configurations have been proposed conventionally.
The first proposition is a configuration for compensating the distortion due to the inaccurate orthogonal phase relationships between the baseband signal inputs, disclosed in Japanese Patent Application Laid Open No. 62-13143, which is schematically shown in FIG. 1.
This quadrature modulator of FIG. 1 comprises: an in-phase amplitude input terminal 40 for receiving an in-phase amplitude input signal I(t) in baseband; a quadrature amplitude input terminal 41 for receiving a quadrature amplitude input signal Q(t) in baseband; a modulated wave output terminal 42 for outputting a modulated wave obtained from the baseband signal inputs; an oscillator 46; a .pi./2 phase shifter 45 for shifting the phase of the carrier generated by the oscillator 46; a first mixer 43 for mixing the in-phase amplitude input signal I(t) entered from the in-phase amplitude input terminal 40 with the phase shifted carrier generated by the .pi./2 phase shifter 45; a second mixer 44 for mixing the quadrature amplitude input signal Q(t) entered from the quadrature amplitude input terminal 41 with the carrier generated by the oscillator 46; a combiner 47 for combining the outputs of the first and second mixers 43 and 44; a signal identifier 48 for identifying signal constellation points constituted by the combinations of the in-phase amplitude input signal I(t) and the quadrature amplitude input signal Q(t); and an amplitude calculator 49 for measuring amplitudes of the signal constellation points identified by the signal identifier 48, obtaining a difference or a ratio of the measured amplitudes, and adjusting the .pi./2 phase shifter 45.
The measured amplitudes do not coincide with each other unless the .pi./2 phase shifter 45 is operating accurately, so that the compensation of the distortion due to the inaccurate orthogonal phase relationships between the carriers can be made automatically by adjusting the operation of the .pi./2 phase shifter 45 according to the amplitude difference obtained by the amplitude calculator 49.
However, this quadrature modulator configuration of FIG. 1 is associated with the following drawbacks.
(i) A fine adjustment of the phase shifter required in this configuration is difficult to realize in a practical phase shifter circuit.
(ii) A practical procedure for controlling the phase shifter is not disclosed.
(iii) An error due to the roll off shaping occurs in practice so that the practically achievable accuracy is insufficient.
(iv) The achievable accuracy becomes insufficient under the presence of the other distortions such as those due to a DC off-set and an amplitude imbalance.
The second proposition is a configuration for compensating the distortion due to the DC off-set, disclosed in Japanese Patent Application Laid Open No. 63-62439, which is schematically shown in FIG. 2. In FIG. 2, those elements which are equivalent to the corresponding elements appeared in FIG. 1 above are labelled by the same reference numerals.
In this quadrature modulator configuration of FIG. 2, the in-phase amplitude input signal I(t) applied to the in-phase amplitude input terminal 40 is mixed with a carrier generated by the oscillator 46 at the first mixer 43 and then the DC off-set in the output of the first mixer 43 is calculated by a DC off-set calculation unit 50, in which the output of the first mixer 43 is rectified and integrated for a negative modulation symbol and a positive modulation symbol separately according to a modulation symbol detected by a comparator 51 which compares the entered in-phase amplitude input signal I(t) with a ground level to determine the modulation symbol. The integrated values for the negative and positive modulation symbols are then added together to yield the total DC off-set which is negatively fed back to the in-phase amplitude input signal I(t) inputted into the first mixer 43 by a feed back loop 52 such that the DC off-set can be compensated. The DC off-set for the quadrature amplitude input signal Q(t) is compensated in the similar manner.
However, this quadrature modulator configuration of FIG. 2 is associated with the following drawbacks.
Namely, in this configuration of FIG. 2, the compensation of the distortion due to the DC off-set is carried out for the in-phase amplitude input signal I(t) and the quadrature amplitude input signal Q(t) separately, but this compensation of the distortion due to the DC off-set before the combiner 47 is actually insufficient because the further distortion is caused by the superposition of the local carrier which leaks in an actual high frequency circuit.
In addition, this configuration of FIG. 2 requires a considerably complicated circuit configuration.
In general, the modulated wave obtained by a quadrature modulator is impaired by a linear distortion as well as a nonlinear distortion. The nonlinear distortion is caused by a nonlinear response of diodes used in the mixers in the quadrature modulator, which can be suppressed relatively easily by reducing the input levels appropriately. On the other hand, the compensation of the linear distortion requires selection of well balanced parts or fine adjustment. This situation concerning the linear distortion will now be described in detail by using the quadrature modulator configuration of FIG. 1 described above as an example.
Namely, at the first and second mixers 43 and 44, the in-phase amplitude input signal I(t) and the quadrature amplitude input signal Q(t) are multiplied by a carrier wave r.sub.C (t)=cos (.omega..sub.C t) of an angular frequency .omega..sub.C generated by the oscillator 46 and a phase shifted carrier wave r.sub.S =-sin(.omega..sub.C t) obtained by the .pi./2 phase shifter 45, respectively, and then combined together at the combiner 47. However, a practical quadrature modulator is affected by the carrier leak due to the stray capacitance or the stray inductance as well as by the image generation due to the phase shift error by the .pi./2 phase shifter 45.
Consequently the actual modulator output y(t) is expressed by: EQU y(t)=y.sub.1 (t)+y.sub.2 (t)+y.sub.3 (t)+y.sub.4 (t) (1)
where y.sub.1 (t) is a response for a case the DC off-set .delta..sub.C1 is added to the baseband signal at the first mixer 43, which is given by: EQU y.sub.1 (t)=[I(t)+.delta..sub.C1 ] cos (.omega..sub.C t) (2)
y.sub.2 (t) is a response for a case in which the baseband signal with the DC off-set .delta..sub.S1 added is multiplied with the quadrature carrier of a phase shift error .theta. as the gain of the second mixer 44 is .alpha. times that of the first mixer 43, which is given by: EQU y.sub.2 (t)=[-.alpha.Q(t)+.delta..sub.S1 ] sin (.omega..sub.C t+.theta.)(3)
y.sub.3 (t) is a response for a case in which an amplitude and a phase of the carrier leak for the in-phase component are .delta..sub.C2 and .theta..sub.1, respectively, which is given by: EQU y.sub.3 (t)=.delta..sub.C2 cos (.omega..sub.C t+.theta..sub.1)(4)
and y.sub.4 (t) is a response for a case in which an amplitude and a phase of the carrier leak for the in-phase component are .delta..sub.S2 and .theta..sub.2, respectively, which is given by: EQU y.sub.4 (t)=.delta..sub.S2 sin (.omega..sub.C t+.theta..sub.2)(5)
Thus, the modulator output y(t) can also be expressed as: EQU y(t)=c(t) cos (.omega..sub.C t)-d(t) sin (.omega..sub.C t) (6)
where c(t) is an in-phase modulated signal which is given by: EQU c(t)=I(t)+.delta..sub.C +sin .theta.[-.alpha.Q(t)+.delta..sub.S1 ](7)
and d(t) is a quadrature modulated signal which is given by: EQU d(t)=-.alpha.Q(t) cos .theta.+cos .theta..multidot..delta..sub.S1 +.delta..sub.S ( 8)
in which the off-sets .delta..sub.C and .delta..sub.S are given by: EQU .delta..sub.C =.delta..sub.C2 cos .theta..sub.2 +.delta..sub.S2 sin .theta..sub.S2 sin .theta..sub.2 ( 9) EQU .delta..sub.S =-.delta..sub.C2 sin .theta..sub.1 +.delta..sub.S2 cos .theta..sub.2 ( 10)
However, the actual output of a conventional quadrature modulator such as those described above is not an ideal one in which the in-phase amplitude input signal I(t) and the in-phase modulated signal c(t) are equal and the quadrature amplitude input signal Q(t) and the quadrature modulated signal d(t) are equal.
Namely, as shown in a signal space diagram of FIG. 3, in an ideal case, the Lissajous' figure for the output of the quadrature modulator with the in-phase amplitude input signal I(t)=cos (x) and the quadrature amplitude input signal Q(t)=sin (x) for x=0 to 2.pi. appears as a circle indicated by a solid line in FIG. 3, but the actual Lissajous' figure for the output of the actual quadrature modulator appears as an oblique oval with a center shifted away from an origin which is indicated by a broken line in FIG. 3 because of the linear distortions. In this Lissajous' figure for the actual output, the normal figure shown in FIG. 4A is superposed with the figure involving the DC off-set shown in FIG. 4B, the figure involving the amplitude imbalance shown in FIG. 4C, and the figure involving the inaccurate orthogonal phase relationship shown in FIG. 4D, where the deviation from the origin represents the carrier leak and the deviation from the circular shape represents the image. Thus, in the Lissajous' figure for the actual output shown by the broken line in FIG. 3, the contributions from the distortions shown in FIGS. 4B, 4C, and 4D are mixed up.
In order to compensate such a linear distortion, it is necessary to carry out the fine adjustment for minimizing the DC off-sets .delta..sub.C1 and .delta..sub.S1, the phase shift error .theta. of the quadrature carrier, and the carrier leaks .delta..sub.C2 and .delta..sub.S2, or the selection of the well balanced parts. However, such a fine adjustment or the selection of the well balanced parts requires a highly skilled operation, a time for making the adjustment, and an expensive measurement devices.
In order to solve the above described problems of the conventional quadrature modulator, there has been a proposition to use a linear transformation of the baseband signal inputs for the sake of the compensation of the linear distortion due to the DC off-set, as described in "Direct Conversion Transceiver Design for Compact Low-Cost portable Mobile Radio Terminals" by A. Bateman and D. M. Haines, 1989 Vehicular Technology Conference Proceeding, pp. 57-62, San Francisco, May 1989.
However, in the quadrature modulator described in this reference uses IQ detectors for extracting the in-phase component and the quadrature component of the output of the quadrature modulator in order to derive the linear parameters to be used in the linear transformation, so that it is necessary in this quadrature modulator to provide the IQ detectors having the higher accuracy than the quadrature modulator, but this in turn requires the fine adjustment of the DC off-set, amplitude balance, and orthogonal phase relationship for the IQ detectors themselves.
Furthermore, these conventionally proposed quadrature modulator configurations have a common problem that, since they are designed to deal with a particular type of the linear distortion alone, when the linear distortions due to the DC off-set and the amplitude imbalance are compensated, the linear distortion due to the inaccurate orthogonal phase relationship cannot be compensated, or visa versa.

Quadrature amplitude modulator with distortion compensation

Nippon Telegraph and Telephone Corporation


A cap for retaining a wheel trim with a wheel includes a body portion which slides inside a central opening defined by a hub portion of the wheel trim. The body portion has a first end and a second end. The cap includes at least one engagement portion coupled to the body portion. The at least one engagement portion engages with an internal wall of the hub portion. The cap includes at least one retaining tab coupled to the body portion towards the second end of the body portion. The at least one retaining tab extends radially outwards from the second end of the body portion. The cap further includes a holding portion coupled to the body portion towards the first end. The holding portion extends radially outwards from the body portion such that the holding portion abuts a disc portion of the wheel trim.

Wheel trim systems are widely used on vehicles to create an aesthetic appearance on vehicle wheels, and in some cases provide improvements to the aerodynamics of the vehicles. The wheel trim also protects the wheels from water, road salt and other environmental factors, which may affect corrosion of the wheels.
Numerous methods and systems of releasably securing wheel trims to vehicle wheels are known. One common method relies upon securing elements which are located circumferentially about the periphery of the wheel trim. In such methods, the securing elements are engaged in a press-fit manner to a peripheral surface of rim of the wheel. To remove the wheel trim, an axially outward force is applied to the wheel trim to overcome the axial restraint upon the rim created by the securing elements. However, one of the concerns with such securing systems is the need to optimize the retention force. If retention force is too low, the wheel trim may come off during normal operation of the vehicle. If retention force is too high, it may be difficult to remove wheel trim easily without damaging the wheel trim. It is desirable to provide a safe and secure wheel trim retention system that allows for ease of attachment and removal of the wheel trim. Preferably, the wheel trim should also be removable without a tool.


The invention relates to a means of enabling inspection of engineering components, such as rails used in the railway industry.

Engineering components which are electrical conductors can be inspected using electromagnetic techniques. These include but are not limited to eddy current inspection, magnetic flux leakage, residual magnetism, alternating current flow measurement (acfm), and magnetic induction.
All of these techniques rely on either an electromagnetic or magnetic field to be created in the vicinity of the surface of the component. The sensor then measures either the absolute properties of the field, or far more commonly, variations of those properties as a function of the spatial relationship between sensor and surface position. Typically the sensor will be scanned across the surface of the component either by moving the sensor with respect to the component or vice versa. As this relative movement occurs the interaction between the sensor and the field is monitored. Variations in the material of the component close to its surface will then be revealed as local disturbances of the field.
This type of technique has found wide application in many fields of engineering and there are many instances of proprietary equipment that carry out such inspections routinely. The most important uses are for the detection of surface breaking cracks, and in some cases non-surface breaking but close-to-surface cracks. Other uses are in material sorting to differentiate one type of material from another.
FIG. 1 shows a system, indicated generally by the reference numeral 1, in which the principles described above are applied. The system 1 comprises a component under inspection 2, a sensor 4, and a non-magnetic spacer 6. In use, the spacer 6 and the sensor 4 move relative to the component 2. The spacer is sized to provide the appropriate spacing between the sensor 4 and the component 2. In some exemplary applications, the separation is typically required to be of the order of about 1 or 2 millimeters for the sensor to maintain the appropriate sensitivity.
FIG. 2 shows a system 10 that is a variant of the system 1 described above. The system 10 comprises a component under inspection 12 and a sensor 14 similar to the component 2 and sensor 4 described above. The system 10 also comprises a mechanical arrangement 16 adapted to separate the component 12 and the sensor 14. In use, the mechanical arrangement 16 (and therefore also the sensor 14) moves relative to the component 12. The separation might typically be required to be of the order of about 1 or 2 millimeters for the sensor to maintain the required sensitivity.
Whilst the techniques described above can be very successful, they suffer from a common significant drawback. The response of the sensor is affected by two things: firstly the variation in the properties of the surface in the vicinity of the sensor, as described briefly above, but secondly the separation of the sensor and the surface under inspection. This latter parameter is often referred to as lift-off.
If the surface under inspection is well-controlled and of uniform and regular shape the sensor can be maintained easily at a known distance from the surface, typically less than 1 mm. However, any variations in this separation will cause variations in the nature and magnitude of the response of the sensor so that any variations in the shape or morphology of the surface will cause significant variations in the output of the sensor. These variations commonly dominate those caused by the material variations.
Consequently a great deal of effort has to be applied to establish a mechanical situation that controls this separation and the effects of lift-off. Conversely, where it is impossible to control the mechanical relationship it is often difficult to produce a satisfactory inspection regime. Thus if any component under inspection is subject to unknown amounts of wear which alter its shape then the results of one of these inspections can be unreliable because of the unknown effects of the consequent variation in lift-off.
Many systems make use of mechanical contact between sensor and component to control this separation but these invariably suffer from wear and the potential for damage where there are unexpected variations in the shape of the component.
The present invention seeks to overcome or address one or more of the problems identified above.

Abstract: A rail inspection device comprises a sensor (24) and a compliant spacer (26). The compliant spacer (26) has an inner surface and outer surface. The sensor (24) is urged against the inner surface of the spacer (26) and, when in use, the outer surface of the spacer (26) is in contact with the rail under inspection.

Trent House, RTC Business Park, London Road, Derby


A ventilation control device is disposed in a bathroom. The ventilation control device has a switch board, a controller connected to the switch board, a sensor connected to the controller, and a fan motor connected to the controller. The controller outputs a signal to initiate the fan motor to change a rotating speed.

The present invention relates to a ventilation control device. More particularly, the present invention relates to a ventilation control device for a bathroom.
Referring to FIG. 1, a conventional ventilation control device is a switch 11. The switch 11 initiates an operation of a lamp 12 and an operation of a fan motor 13. A ventilation opening 14 is formed in a bathroom 1. However, the fan motor 13 does not provide a variable speed at all.

Ventilation control device for a bathroom


A cleaning apparatus is provided including a brush assembly adapted to emit water therefrom upon the receipt thereof. Also included is a hose connected to the brush assembly. Next provided is an elastomeric cover connected to the hose and further releasably securable to a water outlet of a bathing stall.

1. Field of the Invention
The present invention relates to scrub brushes and more particularly pertains to a new tile scrubber for conveniently cleaning a shower stall using water from a shower head.
2. Description of the Prior Art
The use of scrub brushes is known in the prior art. More specifically, scrub brushes heretofore devised and utilized are known to consist basically of familiar, expected and obvious structural configurations, notwithstanding the myriad of designs encompassed by the crowded prior art which have been developed for the fulfillment of countless objectives and requirements.
Known prior art scrub brushes include U.S. Pat. No. 4,057,353; U.S. Pat. No. 3,989,391; U.S. Pat. No. 3,271,809; U.S. Pat. No. 2,793,379; U.S. Pat. Des. 331,840; and U.S. Pat. No. 4,004,312.
In these respects, the tile scrubber according to the present invention substantially departs from the conventional concepts and designs of the prior art, and in so doing provides an apparatus primarily developed for the purpose of conveniently cleaning a shower stall using water from a shower head.


An Atmospheric Turbulence Detector utilizes a sensor to detect noise and extracts infrasound having frequencies below a specified infrasound frequency. A threshold is computed from the detection of infrasound in the vicinity of the sensor prior to the arrival of infrasound from the turbulence and an alarm is given when the infrasound from the turbulence are determined with the utilization of two sensors (6-1A, 6-1B), measuring amplitude differences of the infrasound detected at two seperate locations.

1. Field of the Invention
The invention relates generally to the detection of hazardous atmospheric turbulence, and more particularly to the detection of aircraft wake vortices, microbursts, and tornadoes near the ground, and Clear Air Turbulence (CAT) at high altitude.
2. Description of the Prior Art
Hazardous Turbulence Detection: Clear air turbulence, microbursts, aircraft wake vortices, gust fronts, thunderstorms, updrafts, downdrafts, and convective flows all present severe hazards to aircraft. Effects on aircraft encountering any of these hazards range from severe buffeting to the ultimate catastrophe. A system for detecting such hazards is disclosed in U.S. Pat. No. 6,480,142 issued to W. L. Rubin. Hazardous turbulence is detected with the utilization of Doppler shifted frequencies of received radar signals backscattered from sound generated by atmospheric turbulent flows. The radiated radar signals are generally within the UHF or microwave bands. Resulting Doppler frequency bandwidths are used to establish the parameters of the sound emitted by the, hazardous turbulence. Radar reflections are strongest when the acoustic wavelength of the sound is one half the radar wavelength (Bragg effect). The corresponding acoustic frequencies that maximize reflection from UHF and microwave radar fall between 1 kHz and 20 kHz. As will be shown, turbulence generated sound is strongest in the infrasound band below 20 Hz and decreases approximately 6 dB per octave above 50 Hz. Consequently, acoustic reflections are weak for UHF and microwave radar, resulting in short detection ranges.
Wake Vortex Detection: Aircraft generate multiple invisible vortices in their wake as a consequence of lift. The vortices quickly combine to form a pair of primary vortices that are separated by three-quarters of a wingspan. The primary vortices descend after formation and diverge as they approach a height above the ground approximately equal to one-half of their initial separation. Upon reaching this height, the vortices move in opposite directions in calm winds at a speed of 1 to 2 meters per second. A strong crosswind moves both vortices in the same direction at slightly different speeds.
Vortices produced by large and heavy aircraft are hazardous to smaller aircraft since they can induce rolling moments that exceed the smaller aircraft's roll control authority, causing a dangerous loss of altitude, and possible structural failure. Vortex encounters are especially hazardous in the terminal area where light aircraft follow heavy aircraft on the same glide slope and where recovery from an upset may not be possible because of low aircraft altitude.
To minimize wake vortex encounters, the Federal Aviation Administration (FAA) imposes 3 to 6 mile spacing between leader and follower aircraft based on relative aircraft weight categories. Because a strong crosswind moves wake vortices out of the glide path fairly quickly, FAA spacings are overly conservative most of the time. The FAA has projected airline savings of billions of dollars per year if mandated spacings could be safely reduced. Systems have been proposed to adjust aircraft spacing based on vortex duration in the glide slope. Such systems require a real-time vortex sensor.
A Ground Wind Vortex Sensing System (GWVSS) has been developed to determine vortex duration. This system utilizes a row of pole-mounted anemometers perpendicular to the arrival glide slope for detecting the air flow generated by wake vortices as they move away from the glide slope. Each anemometer is mechanically connected to a DC generator, the voltage of which is proportional to the rotational rate of the anemometer. This output voltage is coupled to a graph-like display in which anemometer positions are located along the bottom of the display. A data point above each anemometer position is deflected upward by an amount proportional to the amplitude of the corresponding generator output voltage. A curve is passed through the vertically deflected points. The location of vortices with respect to individual anemometers can be deduced from the shape of the curve. GWVSS, however, is not operationally deployed because poles are not permitted near airport glide slopes for safety reasons.
Another wake vortex detector of the prior art is a horizontally pointing Radar Acoustic Sensing System (RASS). Such a system has detected, tracked, and measured the strength of wake vortices under all weather conditions. RASS radiates overlapping radar and acoustic beams. Because the air's index of refraction is a function of density, the RASS acoustic beam, which consists of a spatial pattern of condensations and rarefactions, generates in situ refractive index variations. Radar waves reflect from these index variations. As previously stated, radar reflections are strongest when the acoustic wavelength is half the radar wavelength. The reflections focus back onto the radar antenna because the radar and acoustic beams have overlapping spherical wavefronts. Radar reflections are Doppler shifted by an amount corresponding to the instantaneous speed of sound. Because a vortex's circular flow speeds up and slows down different parts of a RASS acoustic wave as it passes through the vortex, the radar Doppler spectrum is a mapping of the vortex's line-of-sight velocity distribution. Vortex circulation (vorticity) is deduced from the Doppler spectrum. RASS, however, emits a loud annoying sound, is costly, and expensive to maintain.
Wake vortices radiate audible sounds. These sounds have been recorded 6 and 12 seconds after aircraft passage by a ground-based, vertically pointing 64-element microphone array. A system, having the acronym SOCRATES (Sensor for Optically Characterizing Ring-Eddy Atmospheric Turbulence Emanating Sound), for detecting this sound utilizes a laser-based, opto-acoustic technique to emulate a large array of microphones. SOCRATES has detected sound generated by wake vortices, but not consistently.
Microburst Detection: A microburst is a small-scale, short-lived, intense downdraft produced by cool dense air descending from the base of a convective cloud. When the downdraft reaches the ground, it generates an outburst of damaging wind flow. Microbursts are invisible and can occur anywhere. Nicrobursts typically last less than 10 minutes and are hazardous to aircraft during takeoff and landing.
Winds traveling outward from the touchdown area have an inverted mushroom shape about 5 miles in diameter and 300 to 500 feet high. Outward wind speeds can be as high as 168 mph and vary with height above the ground. There are two types of microbursts—wet and dry. The distinguishing feature between the two is the prevailing environment in which they are produced.
A system, in the prior art, positioned on board an aircraft for detecting microbursts that measures aircraft ground speed and compares it to the air speed of the aircraft is disclosed in U.S. Patent issued to L. M. Green, et al. The difference in speeds is an indication of wind conditions about the aircraft. This system, however, determines the existence of a microburst in the vicinity of the aircraft and does not provide a timely waning for the avoidance of the microburst.
Another method in the prior art, LLWAS (Low Level Windshear Alert System), consists of a centrally located wind sensor and about 12 additional wind sensors located around the airport. A software algorithm detects and measures wind shear based on wind measurements. LLWAS, however, has poor spatial resolution along the arrival glide slope and departure corridor, thereby allowing many microbursts to escape detection.
Still another system in the prior art is the Terminal Doppler Weather Radar (TDWR) a ground based system which detects microburst wind shear in the vicinity of an airport. TDWR, however, detects only radial wind shear relative to its location.
Additionally, its horizontal antenna beams receive strong ground clutter return that reduces wind shear detectability and decreases measurement accuracy. Further, TDWR detects only wet microbursts, it cannot detect dry microbursts.
Tornado Detection: A tornado is a funnel-shaped, swirling turbulent flow having a top that is connected to an overhead thunderstorm cloud and a bottom that is in contact with the ground. Tornados have internal winds that range from 40 to 300 miles per hour and diameters near the funnel bottom that are typically 1000 feet. A tornado travels at an average speed of 25 to 40 miles per hour over an average path length of 4 miles and emits sound over a wide band of frequencies including very low frequency sounds known as infrasound. Tornado damage results from the high internal wind velocities and wind-blown debris. Tornadoes are nearly invisible, marked only by swirling debris at the base of the funnel.
The National Weather Service utilizes radar to identify conditions conducive to the formation of a tornado and issues tornado watches and warnings, which typically average less than 20 minutes, over the broadcast media. This radar detects about 50% of the tornadoes in the Rocly Mountain high plains and about 25% of the tornadoes in the high plains. The relatively low detection rate results from the fact that multiple radar signatures must be simultaneously identified in order to detect atmospheric conditions conducive to tornado formation.
Radar resolution limits the specificity of tornado warnings to several hundred square miles. As a result, National Weather Service warnings mostly serve to heighten people's awareness in the affected area to look for visual and aural signs of an approaching tornado. In these circumstances the chance to reach shelter may have already passed by the time a tornado is sighted.
Other tornado warning devices rely on the detection of electromagnetic, pressure, and acoustic signatures. Electromagnetic signatures are generated by the thunderstorm spawning the tornado. Since all thunderstorms do not generate tornadoes, this method has a high false alarm rate. Devices utilizing low barometric pressure also have high false alarm rates since low pressure may result from an approaching storm having moderate winds. Both methods provide short warning times and no information about tornado distance.
Numerous tornado warning systems have been devised that rely on detecting sound generated by tornadoes. One such system, disclosed in U.S. Pat. No. 5,355,350 issued to H. E. Bass, et al, monitors outdoor noise between 180 and 420 Hz. When the monitored sound intensity exceeds established thresholds, logic is applied to determine whether the level is increasing at a rate indicative of an approaching tornado and therefore whether an alarm should be sounded. This system, because sound in the selected passband is significantly attenuated by the atmosphere, detects tornadoes to a distance of only one-half mile and consequently, provides warning times of only 30 to 60 seconds. This method also does not provide range to the tornado.
Another system for providing tornado warnings is disclosed in U.S. Pat. No. 6,097,296, issued to S. Garza, et al. In this system a narrow pass band acoustic filter is provided which is centered at one Hertz and designed to attenuate signals that are greater than and less than one hertz. In essence the filter has a near zero bandwidth. The wave emanating from this filter is clipped both at the peaks and valleys to obtain a wave that is substantially square. The square wave is coupled to a counter wherein the number of square wave cycles are counted. An alarm sounds when the counter reaches a predetermined number of cycles.
This system is very difficult to implement. A filter that passes only one frequency, if possible, is extremely difficult to design. Further, a filter centered at 1 Hz may not be optimum. The shape of a tornado's infrasound spectrum depends on internal wind speeds and tornado size, giving rise to infrasounds over a bandwidth that may not include one Hertz or may peak at some other frequency. Should the system be operable as described, it would be of limited utility since it does not provide range to the tornado and a concomitant warning time.
CAT Detection: CAT is produced by high altitude atmospheric flows having speeds of 100 to 200 knots. CAT is often hundreds of miles long, tens of miles wide, and thousands of feet high and cannot be seen or avoided by pilots without prior knowledge of its location. NASA has estimated that airlines encounter CAT about 9 times a month, resulting in 24 injuries, principally to flight attendants. Costs due to CAT exceed $100 million a year. These costs include aircraft repair and downtime, flight attendant injuries and time lost, and passenger injuries.
Radar cannot detect CAT because it is composed of clear air which does not contain reflecting aerosols. A system known as LIAR, operating in the intra band, is currently being assessed for its ability to detect CAT. An experimental LIAR system, however, has generated CAT warnings of only a few seconds. As presently configured, LIAR's high cost, size, and weight, in addition to its poor performance to date, make it unattractive for onboard CAT detection.


Examples disclose a method of transmitting audio data to a wireless device and receiving frequency coefficients associated with the audio data from the device. Further, examples disclose analyzing the frequency coefficients to identify a change between the transmitted audio data and the received frequency coefficients. Additionally, examples also disclose adjusting a second audio data based on the identified change.

Audio music users increasingly rely on their wireless devices in networks to listen to music. The audio music files have a broad frequency scope to include the sounds of the music and as such may take a large amount of bandwidth while the wireless networks are limited in bandwidth. This creates a loss of fidelity of the music file, resulting in a poor listening experience for a user.

Identifying a change to adjust audio data

Hewlett-Packard Development Company, L.P.


A friction disk clutch with an electromagnet arrangement which has a coil former and a ferromagnetically conductive coil support, which supports the coil former and at least partially surrounds the latter, and, adjacent to the coil support, there are permanent magnet means, by means of which an axially movable friction section of the friction disk clutch can be connected magnetically to the rotor, and the permanent magnet means are arranged on the rotor in such a way that the permanent magnet means are covered by essentially no magnetically conductive section of the rotor on a side which is transverse to the direction of magnetization thereof. Anchoring means for mounting the permanent magnet means on the rotor include a mounting section formed for this purpose on the rotor, by means of which the permanent magnet means are mounted in the correct position on the rotor.

Friction clutches for transmitting torques between a drivable rotor and an element to be driven by the rotor are known, allowing parts that are to be coupled to one another, such as disk-type clutch components, to be selectively separated or connected, in particular, by nonpositive engagement.
Friction clutches of this kind can be designed as magnetic clutches, which comprise an electromagnet arrangement having a coil former or wire coil, which can be connected to an electric voltage source and energized, and a coil support composed of magnetically conductive material.
These also include arrangements which additionally have permanent magnets, by means of the magnetic action of which, when the electromagnet is not exerting a magnetic effect, a movable clutch element is attracted to a mating section, counter to a spring force or the like, for example, and these components are connected nonpositively for torque transmission. By energizing the electromagnet, it is then possible, by means of the magnetic field produced, to attenuate or cancel out the effect of the permanent magnet to such an extent that the friction clutch disengages, with the movable clutch element leaving the position of nonpositive engagement.
These clutches are used in vehicles, for example.
The disadvantage with the arrangements described is the comparatively high outlay involved in the manufacture of such clutches and the components thereof.


An expansion anchor is provided, which is fixable in a borehole, including an expansion sleeve for anchoring on a wall of the borehole and including an anchor bolt which is guided through the expansion sleeve and which has an expansion area for expanding the expansion sleeve, the expansion sleeve having a first annular groove on its outside, which extends up to the front end face of the expansion sleeve facing the expansion area. According to the invention, it is provided that the expansion sleeve has a second annular groove on its outside, and that an annular connection piece is situated between the first annular groove and the second annual groove.

Generic DE 102004053255 A1 describes an expansion anchor, which includes an expansion body and an expansion sleeve, which is expandable by drawing the expansion body into the expansion sleeve, the expansion sleeve having a reduced wall thickness and a reduced outer dimension on a front edge facing the expansion body.
EP 2029900 A2 discloses an anchor, which has a plurality of annular grooves in the expansion sleeve. EP 0308619 A1 describes an expansion anchor, including an expansion sleeve, which has cutting teeth and an annular groove on its outside. Another expansion anchor having annular grooves in the expansion sleeve is derived from DE 2046341 A1, the annular grooves here being used to accommodate retaining rings.
EP 0365475 A2 describes an anchor, in which the expansion sleeve has an end-face bevel and a first annular groove situated a distance therefrom.


A roll for use in the production of paper having a frame and a roll coating produced by thermal spraying. The coating is sealed by crystals formed out of a sealing-agent solution after the elimination of water therefrom. A coating for a roll for use in the production of paper produced by thermal spraying and having pores. In order to seal the coating, a sealing agent is crystallized in the pores in the coating.

In the past, the only alternative for a center roll in a press section of a paper machine was a rock roll, which was usually made of granite. The popularity of granite was based on its surface properties which enable controlled separation of the paper web from the rock face and, moreover, granite tolerates the abrading effect of a doctor which is used in conjunction with a center roll very well. However, granite has certain drawbacks. Specifically, since it is a natural material, its properties show variation, and internal flaws in granite and its tendency of cracking form a series obstacle for its use in some applications. For this reason, today, as the center roll of the press section of a paper machine, a coated roll is used very frequently, in particular a roll coated with an oxide coating.
In the art of the production of paper, coated rolls are also used in a great variety of other applications, for example as press rolls, suction rolls, and as rolls of calenders and supercalenders.
The coating can be added in a number of different manufacturing processes onto the various rolls employed in the production of paper, and specifically onto a roll frame of the roll. One important process for adding the coating onto a roll is thermal spraying. However, in this type of coating process, the problem occurred that a certain degree of porosity has tended to remain in the roll coatings, i.e., the thermal spraying results in pores in the coating. This tendency occurs in particular in coatings prepared from oxide ceramic by thermal spraying, because oxide materials are poorly meltable. Oxide coatings are, however, highly suitable roll coatings for applications in which requirements are imposed concerning separation of the paper web, resistance to wear, resistance to corrosion, doctoring, and keeping clean. Such applications are, for example, the center roll in a press section of a paper machine and the rolls in calenders.
It has been one of the more prominent problems in thermally sprayed roll faces that the pores in the roll face are readily filled with contaminants existing and forming in connection with the manufacture of paper, for example pitch, or pastes contained in recycled pulps. When the pores in the roll coating are contaminated, the properties of the coating can be changed and, moreover, there is the risk that the contaminants/moisture break the boundary faces between the pores, in which case the coating is also worn by cracking and splitting.
It is one prior art application for sealing the pores in the coatings that so-called sealing agents are made to penetrate into the porous face. One prior art application of such a sealing agent consists of thermosetting plastics which are composed of two or more components and which have a low viscosity when not solidified. In roll faces sealed in this manner, problems have, however, arisen from the fact that the suitability for doctoring is deteriorated because the plastic face tends to become smooth. Moreover, the smoothing may cause problems of runnability, because the properties of separation of paper are also changed.

Roll for use in the production of paper and coating for the same


Provided is a strengthening device to increase the strength of a grout layer. In the strengthening device, clamp members are interconnected and clamped to surround a grout layer on a concrete base, on which an upper structure is installed, so as to increase the strength of the grout layer. Connection members are provided to edges of the clamp members to connect the clamp members. Fastening means is fastened between the connection members in order to enable the clamp members to clamp the grout layer. This strengthening device acts as a mold to facilitate easy construction when grout is placed, and surrounds the grout layer to increase the strength of the grout layer after grout is cured, thereby effectively preventing and minimizing failure of the grout layer caused by loads transferred from an upper structure such as a column member or a bridge bearing.

1. Field of the Invention
The present invention relates to a strengthening device to increase the strength of a grout layer, placed on a concrete base for keeping a structural safety of an upper structure, and more particularly to a strengthening device to increase the strength of a grout layer, which improves a structural safety of an upper structure by fastening the grout layer placed on a concrete base to give a restraining force thereto so that a failure of the grout layer due to loads transferred from the upper structure such as a column member or a bridge bearing installed on the grout layer may be prevented or minimized.
2. Description of the Related Art
As is well known in the art, a grout layer is placed on a concrete base to act for adjusting a vertical position and a horizontal state of an upper structure so that the upper structure such a column member or a bridge bearing may be installed on the concrete base.
In addition, in the case that the upper structure is a column member, the grout layer plays another role of distributing vertical load F (see FIG. 1) and moment load M1 and M2 (see FIG. 1) transferred through a column of the column member.
Thus, such a grout layer should have sufficient strength to resist bearing failure or edge failure.
In the case where the upper structure is a column member for example, the grout layer formed on a concrete base by placing non-shrinkage mortar at a position right below the column of the column member provides a bearing strength sufficiently resisting against a vertical load F (see FIG. 1) and a moment load M1 and M2 (see FIG. 1) transferred down through the column.
However, since an edge of the grout layer has no restraining force applied inward from the outline, the strength of the edge against the moment load is relatively weaker than that against the vertical load.
For example, if a large moment load M1 or M2 (see FIG. 1) is applied to a column member, or an upper structure installed on the grout layer, due to wind or earthquake, a great bearing stress is generated at the edge and corner of a base plate of the column member installed on the grout layer due to rotation of the base plate.
The excessive bearing stress generated along the edge or the corner of the grout layer becomes a fatal factor of crack or failure on the grout layer unless a restraining force is applied inward (or, toward a center) from an outline of the grout layer.
As a result, moment loads repeatedly applied to the upper structure such as a column member may break down the grout layer easily.
In addition, this phenomenon makes it difficult that loads transferred from the upper structure are distributed to the concrete base, resulting in deteriorated structural safety of the grout layer as well as the upper structure such as a column member or a bridge bearing installed thereon.
Accordingly, there has been needed a technique for a strengthening device (or, a reinforcing device) to increase the strength of a grout layer, which may surround and clamp an outline of a grout layer on a concrete base on which an upper structure such as a column member or a bridge bearing is installed so as to improve the safety of the grout layer as well as an upper structure installed thereon, so that the strength of the grout layer, particularly bearing strength at its edge or corner may be enhanced.

Strengthening device to increase strength of grout layer

Research Institute of Industrial Science & Technology