The invention relates to a method and apparatus for determining spatial aor time intervals in the focused image or focused sequence of images of a lens-diaphragm system and/or of spatial, time-related object-parameters (10), spectral and/or local frequency-specific object parameters, in particular to control automatons and to monitor production sequences, focusing of the incident electromagnetic beams being implemented by the lens-diaphragm system (10), position-specific modulation of the direction of propagation, intensity, wavelength, polarization and/or of the time modulation frequency of the electromagnetic beams (11) being implemented by a 3D modulator (12) present behind the lens-diaphragm system (10) in or near the focal plane of the lens-diaphragm system (10), detection of the modulated electromagnetic beams being carried out by a detection system (13) located to the rear of the 3D modulator (12), and calculation of the spatial and/or time intervals (14) of the electromagnetic beams in the 3D modulator (12) being implemented by determining the modulation and/or computation of spatial, time-related, spectral and/or local frequency specific object-parameters

The invention concerns a method for accurately determining spatial and/or time intervals in the focused image of a lens/aperture-diaphragm system, herein called lens-diaphragm system, and/or of spatial and/or time parameters such as speed or depth of an object, in particular to accurately and intelligently control robots.
Cameras operating in a geometric-optics manner are already used in passive methods and they image onto photographic films or on arrays of charge-coupled devices (CCD). Both procedures provide spatial 3-D information in flat, 2-D form in the image plane. While the resolution of optic cameras is restricted only by optical parameters and by film grain size, on the other hand they are limited to storage and are unsuitable for processing in real time. While CCD cameras offer rapid image processing, they are, however, limited in their resolution by the pixel size of the CCD arrays. It is true that holographically imaging and recording devices can both provide and store spatial 3-D information, but only in coherent light. Moreover, a 2-beam procedure, namely an object and a reference beam, are inherently required in picture-taking. Lastly, holographic systems are able to process images in real time only under constrained conditions. Stereo cameras are technically even more complex and demand computations based on two CCD arrays to obtain 3D information.
As regards the technology of optic sensors, and in particular optical-grating spatial-frequency-filtering using the CORREVIT-/CORREFOT measurement technique (CORREVIT is a trademark of CORRSYS GmbH, Wetzlar, Germany), signal processing regarding length, speed and interval measurements of objects moving relative to the sensor (assembly line, industrial conveyor belts and such) are implemented by inserting gratings between geometric-optics imaging means and following photodiodes into the optic beam to act as signal modulators, as described for instance in German Patent Documents A 21 44 487 and AS 2156617. However, the gratings are used therein merely as 1D-/2D- beam splitters, not as diffraction 1D-/2D-/3D- gratings.
Theoretical modeling was carried out at the "Institut fur Optosensorik Dr. Lauinger/Wetzlar" based on a 3D grating and relating to the "inverted retina of the human eye" that were considered relating (N. Lauinger, "A New Interpretation Of The Stiles-Crawford Effects In Human Vision", Journal of Biological Physics, 19: 167-188, 1994; N. Lauinger, "The Relationship Between Brightness, Hue And Saturation When The Inverted Human Retina Is Interpreted As A Cellular Diffractive 3D Chip", SPIE Proceedings, Vol. 2588, Oct. 1995, 208-232) to subjective phenomena known in human vision (aperture effects, the so-called Stiles-Crawford effects I and II, the Bezold bridge phenomenon, for example). Partial aspects were computed by modeling in 4D grating optics (M. Carbon, "Using Diffraction Theory Of Human Vision For Design Of Color Vision Devices", SPIE Proceedings, Vol. 2353, 1994, 550-560; N. Lauinger, "Inverted Retina Of The Human Eye; A Trichromatic 4D Space-Time Optical Correlator", SPIE Proceedings, Vol. 2904, 1996, 344-360). Effects of a diffraction 3D grating were described regarding incoherence-coherence transformation of optic radiation, as known from German Patent Document A 37 40 533.
In spite of vast and comprehensive endeavors especially regarding automation and robotics, there is to date no technically high resolution, rugged and real time determination of 3D motion based on contactless sensor data acquisition. The core significance of optical flux fields for visual determination of motion is known from German Patent Document A 195 03 606, also "Structure From Motion By Linear Transformation", Proc. Workshop "Kognitive Robotiki", B. Krieg-Bruckner, Ch. Herwig. Edts., ZKW-Bericht 3/95, Zentrum fur Kognitionswissenschaften Universitat Bremen, March 1995. The essential performance features of neural networks are parallelism and adaptiveness (H. O. Carmesin, "Theorie Neuronaler Adaption", Koster, Berlin 1994; H. O. Carmesin, "Neurophysics Of Adaption", Physics Essays, 8(1), 38-51, 1995). Even though neural networks are especially well suited for self-adjusting, high resolution image processing, the neural adaptation theory so far has been used neither in determining motion nor in analyzing diffraction reflex sequences in imaging sensors.
The same problems also arise outside the visible spectrum.
The purpose of the invention is to provide a method and apparatus of the initially cited kind allowing accurate determination in practically real time of spatial and/or time intervals in focused image sequences of a lens-diaphragm system and/or of spatial and/or time-related object-parameters such as speed or depth.
The invention is based on the surprising insight that electromagnetic beams focusable in the image plane with a length resolution of about the light wavelength can be modulated with an accuracy of a wavelength in a position-specific manner by a 3D modulator. The electromagnetic beams so modulated subsequently can be detected in a detector of lesser length resolution and the positions in the image plane can be calculated using the position-specific modulation. As a result, the 3D structure of the light field, and hence the 3D structure of objects, can be detected with high accuracy. Illustratively, electromagnetic beams diffracted at a grating assume a direction determined by the crystal orientation corresponding to the condition of reflection and a wavelength corresponding to the Bragg condition.
The advantages offered by the invention are illustratively discussed below for the visible spectrum:
(1) At present the accuracy of determining the intervals, i.e., spacings of a brightness distribution in a focal plane, is bound by the pixel size of an imaging sensor, these pixels being about 20 micrometer (Am). Using a special embodiment of the apparatus of the present invention, that is with a 3D modulator in the form of a 3D diffraction grating, this accuracy will be limited by the wavelength of the light and by the grating spacing. When selecting an optimal grating, such values are about 0.4 .mu.m. Therefore, the length resolution can be increased by a factor of 50 and the surface resolution by a factor of 2,500.
Such an increase in accuracy is important because the human eye comprises in its retina more than 10.sup.8 receptors each with a corresponding length resolution of about 1 .mu.m, because the correspondingly generated cultural environments in the household, at leisure, in traffic and at work are designed for visual systems such as man's, and because future intelligent mobile automatons and robots require comparable visual systems so they may be operated reliably and economically.
(2) The method of the invention allows analyzing by correlation optics and in a simple and economic manner spatial and/or time-related object-parameters such as speeds or depths.
(3) The method of the invention allows universally using algorithms in the form of learning neural networks at appropriately intended sites for particular applications and for corresponding, specific apparatus, without having to develop a new algorithm. Thereby typically occurring development costs are precluded from the beginning.
(4) In the limiting case of large distances from the grating (far field), the theory of diffraction is already very well developed. It is widely used in optics and materials research. On the other hand, the physics of the near field is comparatively complex and is application-specific. Accordingly, the theory might be developed further and to-date no applications are known relating to the determination of motion, while on the other hand the learning, neural networks used herein are applicable on account of flexibility and are especially adequate. Generally, the central significance of flow fields in determining motion has been known since Helmholtz and, moreover, beyond distance of five meters, determining motion from flow fields is advantageous over that from using binoculars, and quite generally visual determination of spatial and/or time-related object-parameters offers the advantage of high resolution, large range, high speed and natural radiation sources when compared to other modes of operation.
(5) A light field in general contains both amplitude and phase information. In most technical visual systems, only amplitude information is used. On the other hand, in the method of the invention and using the 3D diffraction grating, use also may be made of the latent 3D phase information, even when using incoherent light as emitted by natural light sources. Presumably, in natural visual systems, advantageous analysis of phase information in the human eye takes place by means of the 3D diffraction grating of the inverted retina.
(6) In recent decades, explosive development has taken place in neural networks. The essential advantages are parallelism and adaptivity. Thereby, the neural networks are especially well suited to self-adjusting and high resolution image processing in real time even in complex analyses. In a special implementation of the method of the invention, the neural networks may be used at a site where they can in fact perform their best, in the parallel high resolution reconstruction of object-parameters from comprehensive, as modulated as possible, and displaced diffraction patterns.
Further purposes, advantages, features and applications of the invention are elucidated in the following description of illustrative embodiments of the invention and in relation to the attached drawings. All described and/or graphically shown features per se or in arbitrary meaningful combination represent the objective of the invention, even independently of their summary versions in the claims or of relationships between the claims.
FIG. 1 shows a geometrically/optically imaging lens-diaphragm system with a 3D grating in the image plane,
FIG. 2 is a schematic of a possible embodiment of apparatus of the invention,
FIG. 3 shows a concrete embodiment of the apparatus of the invention,
FIG. 4 schematically shows the imaging and resolving configuration of a concrete embodiment of the apparatus of the invention,
FIG. 5a shows a further embodiment of the apparatus of the invention,
FIG. 5b is a detail of FIG. 5a,
FIG. 6 is an illustrative embodiment of a pattern Fourier-Transform,
FIG. 7 shows the Fourier-Transform using a volume holographic optic element,
FIG. 8 shows the Fourier Transform of a bar structure generated by means of the volume holographic element of FIGS. 7 and 8,
FIG. 9 shows the configuration when using a curved 3D diffraction grating of a further embodiment of the apparatus of the invention,
FIG. 10 schematically shows the state when implementing a learning phase of one implementing mode of the method of the invention,
FIG. 11 shows an illustrative embodiment to determine a depth card using the 3D Talbot effect,
FIG. 12 shows the configuration of the 3D diffraction grating with detectors relating to the illustration of FIG. 11,
FIGS. 13, 14 show the intensity distributions at the Bragg grating in front of the focal point,
FIG. 15 shows the superposition of the intensity distributions at the Bragg grating behind the focal point,
FIG. 16 shows the design of the 3D diffraction grating with detector elements in relation to the configuration of FIG. 11, however when using white light and the luminous element being off-axis,
FIGS. 17, 18, show the intensity distribution of the Bragg grating before the focus of FIG. 16,
FIGS. 19, 20 show the intensity distributions for the Bragg grating behind the focus, and
FIG. 21 shows a configuration relating to the 3D Talbot effect.

Process and device for high-definition measurement of intervals in the focused image produced by a lens-aperture diaphragm system

Fraunhofer-Gesellschaft Zur Forderung Der Angewandten Forschung e.V.


A direct conversion tuner for tuning either analog or digital television signals includes a first and second channels, each having first and second mixers and an intervening filter stage, coupled between an RF input and an output combining unit. The first mixers receive respective first local oscillator signals which have the same frequency but a quadrature phase relationship. The frequency of the first local oscillator signals is controlled according to the selected channel so that it is located within the spectrum of the respective RF signal. The second mixers receive respective second local oscillator signals which have the same frequency but a quadrature phase relationship. The frequency of the second local oscillator signal is located above the passband of the filter stages. A digital gain and phase equalization network is included in one of the channels for adjusting the relative gain and phase shift of the two channels and is automatically controlled by a microcomputer in response to signals sampled at respective points within the first and second channels to reduce the relative gain and phase shift. As a result, near perfect cancellation of unwanted components occurs in the output combining unit.

An early type of tuner known as a "tuned radio frequency tuner" (TRF) included several radio frequency (RF) amplifiers which were all tuned to the frequency of the RF signal of desired transmission channel followed directly by a detection section, without an intervening mixer employed in later tuners. Such a tuner could provide relatively distortion free performance due to the absence of a mixer. However, TRF tuners tended to be large in size and subject to stability and gain control problems due to the number RF amplifiers which were needed. Moreover, TRF tuners did not provide a consistent or adequate degree of signal selectivity.
The type of tuner which is primarily used today is known as a "heterodyne" or "superheterodyne" tuner. In its simplest form, known as a "single-conversion" tuner, it comprises a tunable RF amplifier followed by a frequency conversion stage, including a mixer and a local oscillator. The frequency conversion stage produces an intermediate frequency (IF) signal which corresponds to the received RF signal but has a lower frequency. The IF signal is filtered by an IF filter section and the resultant signal is coupled to a detection section. The combination of conversion stage and the following IF filter section provides a significantly better selectivity characteristic than a TRF tuner. The frequency of the local oscillator signal is offset (usually higher) from the frequency of the desired RF signal by the desired frequency of the IF signal. In a television receiver, the local oscillator signal is controlled so that it places the frequency of the IF picture carrier corresponding to the RF picture carrier at a nominal frequency, for example, at 45.75 MHz in the United States and 38.9 MHz in Europe.
A single conversion tuner can be made quite small and relatively inexpensive. However, it produces unwanted intermodulation and cross-modulation products due to third and higher order components of the signal transfer characteristics of the mixer included in the frequency conversion stage. Various unwanted conversion products, known in the tuner fields as "image", "one-half IF" and "IF beats", continue to be a problem. The IF filter is designed to minimize unwanted conversion products and also to provide rejection of responses due to adjacent channels (selectivity). Thus, the selection of the IF frequency is a compromise. As a result the rejection of unwanted conversion products and selectivity of the tuner may not be adequate.
The deficiencies of a single conversion tuner have become especially troublesome due to the increasing number of "contiguous" channels now available in large cable television systems. With the advent of digital television transmission systems, such as for high definition television (HDTV), the problem becomes still more difficult because these systems make full use of the available channel spectrum and only a small guard band of a few hundred kilohertz (KHz) exists between channels. In addition, the overall frequency response of a tuner for tuning digital television signals must be flat to the edges of the channel, but nevertheless, have a very steep "roll-off" (attenuation) at the edges for adequate adjacent channel rejection. This makes the design of an appropriate IF filter more complicated since no Nyquist slopes and sound traps, which tend to ease IF filter design, can be used in digital systems. In addition, it is contemplated that both analog and digital television signals will be transmitted during a transition period. In that case, even more adjacent channel selectivity will be required for good reception of the digital signals because digital television signals will be transmitted with much less power than analog television signals.
The "double-conversion" variation of the superheterodyne tuner was developed to overcome the shortcomings of the single-conversion tuner. In this type of tuner, a first conversion stage is followed by a first IF filter section, a second conversion stage, and a second IF filter stage. The first IF section has a very high frequency range, typically in the order of 620 MHz. The second IF section has a much lower frequency range, typically the same as that of the only IF filter section of a single conversion tuner. The second IF section is followed by a detection section.
The very high frequency of the first IF filter section places the RF signals corresponding to unwanted conversion responses such as the "image" response at frequencies readily rejected by tunable RF stages which precede the first conversion stage. The low frequency second IF provides the required adjacent channel selectivity needed for modern television reception. Unfortunately, an double-conversion tuner system requires additional RF and IF circuitry compared to a single conversion tuner, and much of the additional circuitry must function at relatively high frequencies requiring extensive shielding. As a result a double conversion tuner is relatively large in size and expensive.
Another type of tuner known as a "direct conversion" tuner has improved unwanted conversion product rejection and selectivity properties with respect the TRF and heterodyne types of tuners. A direct conversion tuner operates in accordance with a third tuning method in which the frequency of a local oscillator signal of a first frequency conversion stage is set in the middle of the frequency band of the desired channel. The product of the first conversion stage is at relatively low frequency. There are no image responses because the frequency of the first conversion stage is located with the spectrum of the desired RF signal. In addition, the very low frequency range of the signal produced at the output of the first conversion stage makes it possible to readily provide a filter which can reject adjacent channel signals.
Unfortunately, because the first local oscillator signal is centered in the frequency band of the desired channel, both the upper and lower side band of the desired channel will be converted to the frequency range of the first IF signal so that the lower side band (LSB) is in effect folded over onto the upper side band (USB) in the spectrum of the first IF signal. Since the LSB and USB occupy the same frequency range, the LSB and the USB must again be separated before detection. To accomplish this, a direct conversion tuner is arranged as is shown in FIG. 1.
Basically, the direct conversion tuner contains two channels, each with two conversion stages. The received RF signal is coupled to each of two mixers M1A and M1B via a tuned RF amplifier which provides gain and some selectivity. Desirably, the gain of the RF amplifier is automatically controlled in response to an automatic gain control (AGC) signal (not shown). The local oscillator signal generated by a first local oscillator LO1 is tuned to the center frequency too of the frequency band of the desired channel between the lower sided band (LSB) and the upper side band (USB), as is shown in FIG. 2a. The first local oscillator signal is split by a phase shifting circuit PS1 into quadrature components that are used to drive mixers M1A and M1B. The respective IF output signals of mixers M1A and M1B are filtered by two low pass filters LPF A and LPF B. Low pass filters LPF A and LPF B provide the necessary selectivity to reject the responses from the adjacent channels and higher order products of mixers M1A and M1B.
Each of the output signals of mixers M1A and M1B includes both a lower side band portion and an upper side band portion corresponding to the LSB and USB portions of the received RF signal. However, as earlier indicated, the LSB portion is folded over so that it is superimposed on the USB portion and occupies the same frequency range, as is shown in FIG. 2b. The output signal of low pass filters LPF A and LPF A are coupled to respective ones of a second pair of mixers M2A and M2B. Mixers M2A and M2B are driven by respective ones of a second pair of quadrature local oscillator signals generated by a second local oscillator LO2 and a second phase shifting circuit PS2. Each of the second local oscillator signals has a frequency .omega..sub.N located above the cutoff frequency of the low pass filters LPF A and LPF B filters to fulfill the Nyquist criteria. The output signals of mixers M2A and M2B are added in a summer unit SU to produce an output signal which has a spectrum which includes separated lower and upper side band portions, as shown if FIG. 2c. This output signal is coupled to a demodulator (not shown) which demodulates it, and the demodulated resultant is coupled to further signal processing sections.
The operation of the direct conversion tuner shown in FIG. 1 can be mathematically understood by considering a very simple case in which the received RF signal is assumed to include a sinusoidal upper side band component of sin (.omega..sub.0 +.omega..sub.1) and a sinusoidal lower side band component of sin (.omega..sub.0 -.omega..sub.2), as is indicated in FIG. 3a. It is also assumed that the gains and phase shifts of the two channels are identical. The phases of the signal components produced at various point of the direct conversion tuner are indicated by the vector arrows in FIG. 1. Further, the coefficients of the various mathematically factors corresponding to signal components have been normalized in the following description.
The quadrature first local oscillator signals applied to first mixers M1A and M1B are expressed as sin .omega..sub.0 and cos .omega..sub.0, respectively; and the quadrature second local oscillator signals applied to second mixers M2A and M2B are expressed as sin .omega..sub.N and cos .omega..sub.N, respectively. The following signal is produced at the output of low pass filter LPF A: EQU cos .omega..sub.1 +cos .omega..sub.2
The following signal is produced at the output of low pass filter LPF B: EQU sin .omega..sub.1 -sin .omega..sub.2
The spectra at the outputs of low pass filters LPF A and LPF B are shown in FIG. 3b.
The result of the second mixing operation by mixer M2A produces the following output signal: EQU sin (.omega..sub.N +.omega..sub.1)+sin (.omega..sub.N -.omega..sub.1)+sin (.omega..sub.N +.omega..sub.2)+sin (.omega..sub.N -.omega..sub.2)
The result of the second mixing operation by mixer M2B produces the following output signal: EQU sin (.omega..sub.N +.omega..sub.1)-sin (.omega..sub.N -.omega..sub.1)-sin (.omega..sub.N +.omega..sub.2)+sin (.omega..sub.N -.omega..sub.2)
The addition of the two output signals of mixers M2A and M2B by summer SU results in the following signal: EQU sin (.omega..sub.N +.omega..sub.1)+sin (.omega..sub.N -.omega..sub.2)
The spectrum at the output of summer SU is shown in FIG. 3c.
The operation of the direct conversion tuner depends on the cancellation of unwanted components developed in the two channels (compare the output signals of mixers M2A and M2B indicated above including the terms sin (.omega..sub.N -.omega..sub.1) and sin (.omega..sub.N +.omega..sub.2)). As was stated above, the description of the operation of the direct conversion tuner so far provided assumes that the gains and phase shifts of corresponding elements of the two channels are identical, resulting in perfect cancellation of the unwanted components after the addition of the output signals of the two channels by summer SU. However, in practice, gain and phase characteristics of the two channels are unequal and change with temperature and time. The gain and phase characteristics affect the phase and magnitude of the vectors shown in FIG. 1. As a result, perfect cancellation of the unwanted components no longer occurs causing the generation of unwanted spurious components in the output signal produced by summer SU and the reduction of the quality of the demodulated signal. This is especially the case when the received RF signal is relatively complex, such as a television signal, and does not simply contain a lower and an upper sinusoidal component as assumed in the above description.
The generation of unwanted spurious components when a television signal is tuned by a direct conversion tuner of the type shown in FIG. 1 is illustrated in FIGS. 4a, 4b and 4c. FIG. 4a shows the spectrum of a television signal of a single channel. It includes a picture carrier (PIX), a color subcarrier (SC) and a sound carrier (SOUND). The frequency, .omega..sub.0, of the first local oscillator signal is located approximately midway between the picture carrier and the sound carrier. FIG. 4b shows the spectrum of the signal resulting from the first mixing operation. FIG. 4c shows the spectrum of the output signal of summer SU. For each desired component of the output signal of summer SU to the right of the frequency, w.sub.N, of the second local oscillator signal, an undesired "companion" to the left exists; and for each desired component of output signal of summer SU to the left of frequency .omega..sub.N an undesired "companion" to the right exists. For instance, a "companion" of the picture carrier is present to the right of .omega..sub.N between the color subcarrier and the sound carrier. The presence of the unwanted "companions" causes annoying beat patterns in the demodulated video signal and may also adversely affect the demodulated sound signal. Such unwanted components should desirably be suppressed in the order of 45 to 50 dB for optimum performance of the television receiver. This means that the gain and phase errors should desirably be kept less than 0.05 dB. and 0.5 degrees, respectively, for optimum performance of the television receiver. Such performance standards cannot be obtained and maintained with manual adjustments.


Described are microelectronic devices comprising a dielectric layer formed on a substrate, a feature comprising a gap defined in the dielectric layer, a barrier layer on the dielectric layer, a two metal liner film on the barrier layer and a gap fill metal on the two metal liner. Embodiments provide a method of forming a microelectronic device comprising the two metal liner film on the barrier layer.

Microelectronic devices, such as semiconductors or integrated circuits, can include millions of electronic circuit devices such as transistors, capacitors, etc. To further increase the density of devices found on integrated circuits, even smaller feature sizes are desired. To achieve these smaller feature sizes, the size of conductive lines, vias, and interconnects, gates, etc. must be reduced. Reliable formation of multilevel interconnect structures is also necessary to increase circuit density and quality. Advances in fabrication techniques have enabled use of copper for conductive lines, interconnects, vias, and other structures.
Specifically, for the 5 nm node and below, barrier and liner thickness for copper interconnects becomes even more challenging with respect to device reliability and adhesion of the barrier layer to a dielectric layer. Shrinking geometries also result in higher resistance as well as greater susceptibility to electro-migration (EM) failures in the copper lines. A high-quality bond at the interface between the copper and dielectric barrier layer can reduce or prevent EM failures. A typical thickness of a barrier layer and liner at the 5 nm node is on the order of about 45 Å. Thicker barrier/liner layers result in less space for metal gap fill and tend to increase resistivity. The current approach to improve metal/dielectric to metal adhesion and filling metal mobility during gap fill is to increase film thickness of a single material, and this approach is limited by single material intrinsic properties and thickness.
Therefore, there is a need for material layers and methods for depositing such material layers that improve metal/dielectric to metal adhesion.


A forked shower curtain construction 10 for bathtub receptacles 200. The curtain construction 10 comprises a generally elongated rectangular main shower curtain panel member 20 and an auxiliary shower curtain panel member 30 releasably secured to the lower portion of the main shower curtain panel member 20 by a zip-lock type releasable securing means 25 and 35.

1. Field of the Invention
The present invention relates to the field of shower curtain constructions in general, and in particular to a forked bottom shower curtain construction having a removable auxiliary lower panel member.
2. Description of Related Art
As can be seen by reference to the following U.S. Pat. Nos. 2,776,439; 4,126,172; 5,421,393; and 5,495,628; the prior art is replete with myriad and diverse shower curtain constructions.
While all of the aforementioned prior art constructions are more than adequate for the basic purpose and function for which they have been specifically designed, they are all uniformly deficient with regard to providing a means for simplifying the act of cleaning the lower interior portion of the shower curtain, which seems to act as a magnet for dirt, soap scum, mildew, and the like.
In the past, it was extremely difficult to accomplish this task due to the mass of the conventional shower curtain constructions and the awkwardness in manipulating such a large amount of material.
As a consequence of the foregoing situation, there has existed a longstanding need for a new type of shower curtain construction which not only covers both sides of the rim of the bathtub, but which also includes a relatively small detachable interior panel member which can easily be removed for cleaning. The provision of such a construction is a stated objective of the present invention.


A liquid crystal display device comprises a liquid crystal panel including first and second substrates bonded to each other with a liquid crystal layer positioned therebetween, and the photosensor, formed on the second substrate, for sensing an external light from the surroundings, wherein the photosensor includes a semiconductor layer formed on the second substrate and provided with n+-type ion implantation region, ion non-implantation region and lightly doped region; an insulation film, formed on the second substrate, for covering the semiconductor layer; a passivation film, formed on the second substrate, for covering the insulation film; a first contact hole passing through the insulation film and the passivation film, to expose source and drain regions of the semiconductor layer; source and drain electrodes connected with the source and drain regions of the semiconductor layer through the first contact hole; an ion implanting prevention film formed on the insulation film and overlapped with the ion non-implantation region; and a second contact hole passing through the passivation film and the ion implanting prevention film above the ion non-implantation region, so as to provide the external light to the ion non-implantation region.

1. Field of the Invention
The present invention relates to a liquid crystal display device, and more particularly, to a liquid crystal display device including a photosensor with improved sensing efficiency, and a method of fabricating the same.
2. Discussion of the Related Art
According as various mobile electronic devices, for example, mobile phone, PDA and notebook computer are developed recently, there are increasing demand for flat panel display devices with thin profile and light weight. Examples of flat panel display devices include liquid crystal display devices (LCD), field emission displays (FED), and plasma display panels (PDP). Among the flat panel display devices, the liquid crystal display device has the great attentions owing to the technology suitable for the mass production, the easy driving method and the realization of high resolution.
The liquid crystal display device corresponds to a transmitting-type display device, which controls the light transmittance through a liquid crystal layer by refractive anisotropy, thereby displaying desired images on a screen. For displaying the desired images in the liquid crystal display device, there is a requirement for a backlight unit whose light passes through the liquid crystal layer. Thus, the liquid crystal display device is comprised of the liquid crystal panel and the backlight unit provided at the rear of the liquid crystal panel.
The backlight unit emits the light of constant luminosity to the liquid crystal panel. That is, since the backlight unit emits the light of constant luminosity even in case of the relatively bright environments, it causes the increase of power consumption. Virtually, the backlight unit uses a large percentage of total power, in more detail, about 80% or more of the total power used for driving the liquid crystal display device. In order to fabricate the liquid crystal display device of the low power consumption type, there are various methods to lower the power consumption of backlight unit.
One of the various methods to lower the power consumption of backlight unit is to provide a liquid crystal display device including a photosensor which can sense the luminosity of external light from the surroundings.
As shown in FIG. 1, a liquid crystal display device 100 including a photosensor to sense luminosity of external light from the surroundings includes a liquid crystal panel 150 provided with an upper substrate 110, a lower substrate 120, and a liquid crystal layer 130 between the upper and lower substrates 110 and 120; and a backlight 200 provided at the lower substrate 120 and emitting light to the liquid crystal panel 150. The liquid crystal panel 150 is defined with a display region to display picture images; a non-display region on which picture images are not displayed; and a black matrix region, provided between the display region and the non-display region, for blocking the light.
The upper substrate 110 corresponds to a color filter substrate. At this time, R, G and B color filters 101 are formed in the pixel region of the upper substrate 110, and black matrix films 105 are formed in the black matrix region of the upper substrate 110. Although not shown in detail, the black matrix film 105 is provided in the boundary (not shown) of pixels, thereby preventing the light leakage. The color filter 101 is a resin film including dye or color. In addition, an overcoat layer (not shown) may be formed to planarize the surface of color filters 101. On the overcoat layer, there is a common electrode 103 to apply a voltage to the liquid crystal layer 130.
The lower substrate 120 is provided with a plurality of gate and data lines 125 and 127 crossing each other to define the pixels. Also, a switching device for switching each pixel is provided at each crossing of the gate and data lines 125 and 127. For example, the switching device is formed of a thin film transistor 121 including a gate electrode, a semiconductor layer, and source and drain electrodes. Then, a gate pad 125a is provided at one side of each gate line 125, and a data pad 127a is provided at one side of each data line 127, wherein the gate and data pads 125a and 127a apply signals to the respective gate and data lines 125 and 127. Each pixel is provided with a pixel electrode 123, wherein the pixel electrode 123 of the lower substrate 120 is facing with the common electrode 103 of the upper substrate 110. The common electrode 103 and the pixel electrode 123 are formed of transparent conductive materials which are suitable for transmitting the light to the backlight 200.
Also, a photosensor 140 is formed in the black matrix region of the lower substrate 120, to sense the luminosity of external light and control the brightness of backlight. To expose the photosensor 140 to the external environment, corresponding portion in the black matrix of the upper substrate 110 is partially removed.
As shown in FIG. 2, according as the corresponding portion of the black matrix 105 is removed from the black matrix region of the upper substrate 110, the photosensor 140 of the lower substrate 120 is exposed to the external. At this time, the photosensor 140 is formed simultaneously when forming the thin film transistor 121.
FIG. 3 is a cross section view illustrating a thin film transistor and a photosensor included in a liquid crystal display device according to the related art.
As shown in FIG. 3, a substrate 120 includes a thin film transistor region (I) having a channel of p-type ion implantation region; a thin film transistor region (II) having a channel of n-type ion implantation region; and a photosensor region (III).
Referring to FIG. 3, a p-type semiconductor layer 163, an n-type semiconductor layer 164, and an n-type and p-type semiconductor layer 165 are formed at fixed intervals on the substrate 120 including a buffer layer 162. Then, a gate insulation film 166 is formed on the p-type semiconductor layer 163, n-type semiconductor layer 164 and n-type and p-type semiconductor layer 165. Also, a gate electrode 168 is formed on the gate insulation film 166 above the p-type semiconductor layer 163 and the n-type semiconductor layer 164.
Also, an insulating interlayer 170 including a contact hole to expose the semiconductor layer is formed on the gate electrode 168. Then, source and drain electrodes 172 are formed on the insulating interlayer 170, wherein the source and drain electrodes 172 are respectively connected with the p-type semiconductor layer 163, n-type semiconductor layer 164 and n-type and p-type semiconductor layer 165 through the contact hole to expose the semiconductor layer.
The n-type semiconductor layer 164 is formed such that its region being in contact to the source and drain electrodes 172 is provided with an n+-type ion implantation region 164a, its region being in contact to the gate insulation film 166 is provided with an ion non-implantation region 164b, and its region therebetween is provided with an n−-type LDD layer 164c.
The p-type semiconductor layer 163 is formed without additional LDD layer and is formed such that its region being in contact to the source and drain electrodes 172 is provided with a p-type ion implantation region 163a and its region being in contact to the gate insulation film 166 is provided with an ion non-implantation region 163b.
The n-type and p-type semiconductor layer 165 is formed such that its region being in contact to the source and drain electrodes 172 is provided with p+-type and n+-type ion implantation regions 165a and 165b, and its region being in contact to the gate insulation film 166 is provided with an ion implantation region 165c.
On the ion-implantation process for forming the LDD layer, the n−-type LDD layer 164c of the n-type semiconductor layer 164 is formed by using the gate electrode formed on the gate insulation film as an ion implantation mask, instead of using a mask of photoresist pattern. However, the gate electrode as well as the mask of photoresist pattern is not formed in the photosensor region (III) on the ion implantation process for forming the LDD layer. Thus, n-type ions are doped in the ion implantation region 165c between the p+-type ion implantation region 165a and the n+-type ion implantation region 165b.
In case of the photosensor region (III), if the ion implantation region 165c is formed between the p+-type ion implantation region 165a and the n+-type ion implantation region 165b, it is difficult to check the intensity of current in the photosensor region according to the intensity of external light.
In other words, if the external light becomes strong, it raises the intensity of current flowing through the source and drain electrodes, that is, p+-type and n+-type ion implantation regions 165a and 165b. Meanwhile, if the external light becomes weak, it lowers the intensity of current flowing through the source and drain electrodes. Accordingly, it is possible to check the intensity of current in the photosensor region according to the intensity of external light.
However, the related art photosensor region cannot check the intensity of current according to the intensity of external light since the n−-type ion implantation region 165c formed between the p+-type ion implantation region 165a and the n+-type ion implantation region 165b affects the intensity of current flowing through the p+-type and n+-type ion implantation regions, so that the sensing efficiency of photosensor region deteriorates. That is, as shown in FIG. 4, the related art photosensor region has the non-linear property of drain current according to drain-source voltage Vds, whereby it is difficult to check the difference of current according to the intensity of external light, precisely.

Liquid crystal display device with photosensor and method of fabricating the same


It is attained by being provided with a motor constant calculation unit for calculating electric constants of a motor, and by correcting setting values of electric constants defined on one of the axes of two orthogonal axes, by a functional expression using a state variable defined on the same axis, and by correcting them by a functional expression using a state variable defined on the other axis.

The present invention relates to a control apparatus for an AC motor, in particular, the present invention relates to setting of electric constants of an AC motor.
An AC motor, in particular, a permanent magnet synchronous motor has been extending the application ranges such as consumer electronics, industry, automobiles, by utilization of features of a compact size and high efficiency. In particular, in recent years, products having a motor drive system replaced from a square wave energization type to a sinusoidal wave energization type, and there has increased a controller essentially requiring input of electric constants setting values of a motor, such as resistance, inductance and back EMF (electromotive force) constant, in applications such as rotator position estimation by position sensorless control, or high precision torque control. Therefore, input without correct identification of electric constants of a motor influences on control performance very much. Among others, inductance is strongly influenced by magnetic nonlinearity of a core, and influence of magnetic saturation is expressed very much.
Technology for varying electric constants setting values of an AC motor, in response to electric current, has been shown in JP-A-2001-161099. This technology is a technology for improving torque precision by furnishing the relation between magnetic flux and electric current of a synchronous motor, as a nonlinear function, to the inside of a controller. Further, JP-A-9-327200 has shown a technology for configuring a control apparatus, in consideration of cross-magnetization effect of an armature d-axis and an armature q-axis, and magnetic field system and an armature q-axis (hereafter referred to as conventional technology 2).


A device and a method for purifying and decontaminating air uses an exchanger of the cyclone type and furthermore employs a combination of the action of cyclonic centrifugation and the action of at least one path enhancing field. A tranquilizing chamber, which is separated from the cyclone exchanger by a vent, maintains a volume of liquid into which particles and droplets are received from the peripheral wall. To this end, particulates are partially electrified before entering the enhancing field.

The invention present invention relates to a device for purifying and decontaminating air.
In numerous industries, particularly including the electronics industry, coatings, foods, pharmaceuticals, hospitals, cell cultures, etc., control of the quality of the air in terms of particle content is an absolute necessity. Particulate and microbiological contamination are moreover linked, since it is now known that the micro-organisms present it the environment are almost always attached in the form of biofilms to the particles present in the air, which then serve as a support.
It should also be considered that one particle in 10,000 is associated with biocontaminant presence. It is therefore not surprising that systems for microbiological sanitation of the environment in industries in which the process demands it, are entirely bound to the control of the particulate content of the air.
The parameters which make it possible to monitor this bacteriological content of the air call upon either physics (filtration supplemented with bactericidal action, by UV, ionization, etc.) or chemistry for disinfection (aerosols, bactericides, fumigation, etc.) or usually a combination of the two (i.e.) filtration supplemented with backup from a bactericide).
In most of the solutions currently developed, filtration remains the basic method which makes it possible to limit the number of particles per unit volume and hence, by a proportionate effect, the associated microbiological contamination. These systems have the major drawback of rapidly becoming fouled, and hence of requiring substantial renewal costs if their efficiency is not to see a rapid and irreversible decline.

Cyclone exchanger with tranquilizing tank and method for purifying and decontaminating air


Flexible carrier aggregation is provided for a radio communications system. A capability is determined to communicate over a radio interface using multiple radio frequency component carriers. Each of the multiple component carriers is configurable with one or more control channels in a first mode of operation and with no control channels in a second mode of operation. Configuration information for one of the multiple radio frequency component carriers is signaled to indicate at least one of the component carriers is configured to operate in a selected one of the first mode of operation and the second mode of operation so that a network radio node and a user equipment radio node can communicate using the selected mode of operation.

In a typical cellular radio system, radio or wireless terminals (also known as mobile stations and/or user equipment units (UEs)) communicate via a radio access network (RAN) to one or more core networks. The radio access network (RAN) covers a geographical area which is divided into cell areas, with each cell area being served by a base station, e.g., a radio base station (RBS), which in some networks may also be called, for example, a "NodeB" (UMTS) or "eNodeB" (LTE). A cell is a geographical area where radio coverage is provided by the radio base station equipment at a base station site. Each cell is identified by an identity within the local radio area, which is broadcast in the cell. The base stations communicate over the air interface operating on radio frequencies with the user equipment units (UEs) within range of the base stations.
The Universal Mobile Telecommunications System (UMTS) is a third generation mobile communication system, which evolved from the Global System for Mobile Communications (GSM). UTRAN is essentially a radio access network using wideband code division multiple access for user equipment units (UEs).
In a forum known as the Third Generation Partnership Project (3GPP), telecommunications suppliers propose and agree upon standards for third generation networks and UTRAN specifically, and investigate enhanced data rate and radio capacity. The Third Generation Partnership Project (3GPP) has undertaken to evolve further the UTRAN and GSM based radio access network technologies. The first release for the Evolved Universal Terrestrial Radio Access Network (E-UTRAN) specification has issued, and as with most specifications, the standard will likely evolve. The Evolved Universal Terrestrial Radio Access Network (E-UTRAN) comprises the Long Term Evolution (LTE) and System Architecture Evolution (SAE).
Long Term Evolution (LTE) is a variant of a 3GPP radio access technology where the radio base station nodes are connected to a core network (via Access Gateways (AGWs)) rather than to radio network controller (RNC) nodes. In general, the functions of a radio network controller (RNC) node are distributed between the radio base stations nodes (eNodeB's in LTE) and AGWs. As such, the radio access network (RAN) of an LTE system has what is sometimes termed a "flat" architecture including radio base station nodes without reporting to radio network controller (RNC) nodes.
A key feature of long term evolution advanced (LTE-A) is achieving higher data rates by allowing a user equipment to receive and transmit data on multiple LTE component carriers simultaneously in both uplink and downlink directions, which is referred to as "carrier aggregation."
In legacy LTE release-8, the UE receives and transmits using a single carrier frequency. To limit UE and base station complexity, the radio and performance requirements are specified for a limited number of channel bandwidths: 1.4 MHz, 3 MHz, 5 MHz, 10 MHz, 15 MHz, and 20 MHz. The legacy LTE release-8 UE supports transmission and reception up to 20 MHz of channel bandwidth. In LTE, orthogonal frequency division multiple access (OFDMA) and single carrier—frequency division multiple access (SC-FDMA) are used as access technologies in the downlink and uplink, respectively. Therefore, the physical radio communication resources are expressed in terms of "resource blocks," both in the downlink and the uplink. A resource block is a time-frequency resource comprising of one time slot (0.5 ms) in time and 180 KHz or 12 sub-carriers (1 sub-carrier carrier spacing=15 KHz) in frequency. Each legacy LTE release-8 channel bandwidth contains certain number of resource blocks. For instance, a 20 MHz carrier frequency can theoretically accommodate 110 resource blocks. But in order to meet various practical radio requirements, (e.g., modulation quality, spectrum emission mask, etc.), a guard band is required on each flank of the carrier frequency. For example, in order to fulfill at least one example set of radio requirements, only 100 resource blocks are transmitted in both the uplink and the downlink for a 20 MHz carrier frequency.
Carrier aggregation is described at a general level in the feasibility study for 3GPP LTE release-10, i.e., LTE-Advanced (LTE-A) at 3GPP TR 36.815 V9.1.0 (2010-06) available at the 3GPP website www.3gpp.org. Carrier aggregation means that two or more "component carriers" are aggregated in order to support wider bandwidths. Each carrier frequency is referred to as a component carrier. A goal is to aggregate a different number of component carriers of possibly different bandwidths in the UL and the DL. FIG. 1A illustrates an example of aggregated bandwidth of 90 MHz made up of four 20 MHz component carriers and one 10 MHz component carrier, which are all contiguous. FIG. 1B illustrates an example of aggregated bandwidth of 20 MHz made up of four 5 MHz component carriers, two of which are not contiguous and two of which are contiguous.
Carrier aggregation allows a UE to simultaneously receive and/or transmit data over more than one carrier frequency thereby enabling a significant increase in data reception and/or transmission rates. For instance, 2×20 MHz aggregated carriers theoretically provide a two-fold increase in data rate as compared to the data rate for a single 20 MHz carrier. As shown in the examples in FIGS. 1A and 1B, a component carrier may be contiguous or non-contiguous. Non-contiguous carriers may belong to the same frequency band or to different frequency bands. Aggregated bandwidths of up to 100 MHz using up to 5 component carriers are being considered. A hybrid carrier aggregation scheme with contiguous and non-contiguous component carriers is also contemplated for LTE-advanced.
Implementation of the above contiguous carrier aggregation scenarios requires that a network operator own a very large contiguous frequency band, e.g., 80-100 MHz in the same band. But presently, large contiguous frequency allocations, e.g., on the order of 80-100 MHz, either do not exist or may be difficult to acquire in the future. One way to allow an operator to efficiently use larger chunk of available spectrum is to employ an extension carrier or segment which is generally a smaller carrier compared to other component carriers and can be used by an operator to "fill in" the available spectrum.
To illustrate this point, consider an example in which contiguous 80 MHz spectrum is available. An operator can use 4×20 MHz carriers, each with 100 resource blocks (RBs). This configuration maintains backward compatibility with the legacy LTE-release 8 20 MHz carrier in terms of resource blocks (there are 100 resource blocks per 20 MHz in legacy LTE-release 8). A small portion of spectrum of about 6 MHz (which is less than the 20 MHz legacy spectrum) remains unused as now explained. The 20 MHz carrier has a channel bandwidth of 20 MHz, but a transmission bandwidth of 18 MHz for the 100 RBs because there is 1+1 MHz of guard band inside the channel bandwidth, namely, 1 MHz at each end of the 18 MHz. This means that a densely-packed system for LTE-A of 4×20 MHz uses 4×18 MHz=72 MHz out of the 80 MHz to actually carry resource blocks. In other words, the 4×20 MHz carriers use up all of the available spectrum if the four carriers are configured as: (1+18+1)+(1+18+1)+(1+18+1)+(1+18+1). But if instead the four carriers were configured as: (1+18+18+18+18+1) without the 6 intervening guard bands, that provides room for 6 MHz for additional extension carriers. If a 5 MHz extension carrier is added onto 4×20 MHz carriers, then the available 80 MHz can be more efficiently used. Specifically, if the 25 RBs=4.5 MHz are packed next to the 4×18 MHz=4×100RB=4×100*0.18 MHz, then the result is 1+4×18+4.5+1=78.5 MHz. So even though a guard band of small unused spectrum at the edges of the aggregated carriers must be maintained to meet practical radio requirements, which means 100% spectrum utilization is not practically possible, introduction of an extension carrier can significantly improve the spectral efficiency.
One approach for implementing carrier aggregation in LTE-advanced (LTE-A) is to make the extension carrier backward-compatible to legacy LTE-8 carriers, which means all control channels normally found in the LTE-8 carrier are also found in the extension carrier. An alternate approach that is not backward-compatible to LTE-8 carriers uses an extension carrier with only data resource blocks and no control channels. But this static design choice deprives the operator of flexibility to selectively choose an approach that best suits a current network need or situation. In other words, when configuring new spectrum for LTE-A carrier aggregation, the operator either selects a component carrier design that is an LTE backwards compatible component carrier but which suffers a user data inefficiency due to signaling channel overhead associated with that backwards compatibility, or an LTE-10 extension carrier, which is efficient because there is no signaling overhead, but which is not backwards compatible to earlier LTE releases.
What is needed is flexibility to be able to selectively and dynamically implement either approach including one with one or more extension carriers with control channels and another where extension carriers are configured without control channels. Another need is a signaling mechanism to support such flexible carrier aggregation.

Signaling for flexible carrier aggregation


Method for data acquisition and processing, which uses time-domain averaging to increase the signal to noise ratio (SNR) in single-event measurements. In a preferred embodiment, NMR echo-trains obtained using high-speed NMR logging are provided, the echo trains are time-domain averaged, preferably in real-time, over one or more time intervals to sharpen the spatial resolution of the logging tool and/or to increase the signal to noise ratio (SNR) of the data train.

In oil and gas exploration it is desirable to understand the structure and properties of the geological formation surrounding a borehole, in order to determine if the formation contains hydrocarbon resources (oil and/or gas), to estimate the amount and producibility of hydrocarbon contained in the formation, and to evaluate the best options for completing the well in production. A significant aid in this evaluation is the use of wireline logging and/or logging-while-drilling (LWD) measurements of the formation surrounding the borehole (referred to collectively as "logs" or "log measurements"). Typically, one or more logging tools are lowered into the borehole and the tool readings or measurement logs are recorded as the tools traverse the borehole. These measurement logs are used to infer the desired formation properties.
In recent years nuclear magnetic resonance (NMR) logging has become very important for purposes of formation evaluation and is one of the preferred methods for determining formation parameters. Improvements in the NMR logging tools, as well as advances in data analysis and interpretation allow log analysts to generate detailed reservoir description reports, including clay-bound and capillary-bound related porosity, estimates of the amounts of bound and free fluids, fluid types (i.e., oil, gas and water), permeability and other properties of interest.
NMR tools used in practical applications include, for example, the centralized MRIL® tool made by NUMAR Corporation, a Halliburton company, and the sidewall CMR tool made by Schlumberger. The MRIL® tool is described, for example, in U.S. Pat. No. 4,710,713 to Taicher et al. and in various other publications including: "Spin Echo Magnetic Resonance Logging: Porosity and Free Fluid Index Determination," by Miller, Paltiel, Gillen, Granot and Bouton, SPE 20561, 65th Annual Technical Conference of the SPE, New Orleans, La., Sep. 23–26, 1990; "Improved Log Quality With a Dual-Frequency Pulsed NMR Tool," by Chandler, Drack, Miller and Prammer, SPE 28365, 69th Annual Technical Conference of the SPE, New Orleans, La., Sep. 25–28, 1994. Certain details of the structure and the use of the MRIL® tool, as well as the interpretation of various measurement parameters are also discussed in U.S. Pat. Nos. 4,717,876; 4,717,877; 4,717,878; 5,212,447; 5,280,243; 5,309,098; 5,412,320; 5,517,115, 5,557,200; 5,696,448; 5,936,405; 6,005,389; 6,023,164; 6,051,973; 6,107,796 and 6,111,408. The structure and operation of the Schlumberger CMR tool is described, for example, in U.S. Pat. Nos. 4,939,648; 5,055,787 and 5,055,788 and further in "Novel NMR Apparatus for Investigating an External Sample," by Kleinberg, Sezginer and Griffin, J. Magn. Reson. 97, 466–485, 1992; and "An Improved NMR Tool Design for Faster Logging," D. McKeon et al., SPWLA 40th Annual Logging Symposium, May–June 1999. The content of the above patents is hereby expressly incorporated by reference for all purposes, and all non-patent references are incorporated by reference for background.
NMR tools of the type discussed above generally measure the time for hydrogen nuclei present in the earth formation to realign their spin axes, and consequently their bulk magnetization, either with an externally applied magnetic field, or perpendicularly to the magnetic field, after momentary reorientation due to the application of specific radio frequency (RF) pulses. The externally applied magnetic field is typically provided by a magnet disposed in the tool. The spin axes of the hydrogen nuclei in the earth formation are, in the aggregate, caused to be aligned with the magnetic field induced in the earth formation by the magnet. The NMR tool includes an antenna positioned near the magnet and shaped so that a pulse of radio frequency (RF) power conducted through the antenna induces a magnetic field in the earth formation orthogonal to the field induced by the magnet. The RF pulse has a duration predetermined so that the spin axes of the hydrogen nuclei generally align themselves perpendicular both to the orthogonal magnetic field induced by the RF pulse and to the externally applied magnetic field. After the pulse ends, the nuclear magnetic moment of the hydrogen nuclei gradually relax, i.e., return to their alignment with the externally applied magnetic field; at the same time an antenna, which is typically the same as the one used by the initial pulse, is electrically connected to a receiver, which detects and measures voltages induced in the antenna by precessional rotation of the spin axes of the hydrogen nuclei.
An actual NMR measurement involves a plurality of pulses grouped into pulse sequences, most frequently of the type known in the art as Carr-Purcell-Meiboom-Gill (CMPG) pulsed spin echo sequences. As known in the art, each CPMG sequence consists of a 90-degree (i.e., π/2) pulse followed by a large number of 180-degree (i.e., π) pulses. The 90-degree pulse rotates the proton spins into the transverse plane and the 180-degree pulses generate a sequence of spin echoes by refocusing the transverse magnetization after each spin echo. A wait time interval (TW) of approximately 0.5–10 sec is used first to allow for polarization of the formation by the tool's static magnetic field. Then, a CPMG pulse-echo train is executed, consisting of an excitation pulse and an alternating sequence of refocusing pulses. Following each pair of excitation pulse and a refocusing pulse, an acquisition window is applied next. Complex data points from such a pair of echo trains are co-added on an echo-by-echo basis to remove certain artifacts and to enhance the NMR signal, as known in the art. More pairs may be added to enhance the signal-to-noise ratio. The echo train, consisting of a superposition of exponentially decaying signals is then submitted to a processor, which calculates the underlying decay modes of the received NMR echo signal.
One of the key information sought from the NMR logging data is the T2 decay time distribution, which is related to the pore size distribution of the rock formation. From the T2 decay time distribution one can estimate, for example, the capillary bound water volume. The T2 distribution is obtained by inversion of the echo train data, using well-known methods as described, for example, in U.S. Pat. No. 5,517,115 to the assignee of the present application.
It is also well-known in the art that inversion of NMR echo trains is greatly affected by noise. FIG. 1 illustrates a single-event echo train. In order to reduce the effect of noise and increase the signal-to-noise ratio (SNR), stacking of multiple echo train data is commonly used, by assuming that noise is of random nature. Thus, for example, the average echo train from 10 events is affected with much less noise than the echo train from a single event. (In general, if N is the number of stacked signals, the SNR of the averaged signal improves as approximately sqrt(N)). Various methods exist to perform stacking such as, for example, using two pulse sequences of opposite phase that cancel electronic offsets and 180-degree ringing. Such a pair of pulse sequences is called a phase-alternated pair (PAP).
In actual logging, the tool is moving at a certain speed, and echo train from each event is not collected at the same logging location or depth. When the logging speed is slow, the tool moves only a short distance. Hence one may assume that the earth formation is of nearly constant property over the short distance, and stacking of multiple events from different depths may be acceptable. However, when the logging speed is relatively fast, the tool moves over longer distances and the formation properties may change significantly over the distance. As a result, stacking of echo train data from multiple events, as generally used in the prior art, reduces apparent spatial resolution of measuremenets beyond the intrinsic resolution of the tool.
In particular, in high-signal formations (e.g., high porosity, oil-or water-filled rocks), where the logging speeds can be comparatively fast, the effective vertical resolution of the NMR log becomes dominated by the tool movement during a single PAP. For example, with a recovery time of 10 seconds between echo-trains in a PAP, with PAPs acquired at all possible frequencies, the elapsed time between the first echo-train in the first-frequency PAP and the second echo-train for the last-frequency PAP, is close to 20 seconds. At a logging speed of 900 ft/hr (15 ft/min), the MRIL tool will move approximately 5 feet during this measurement: when combined with the inherent vertical resolution of the antenna (which is approximately 2 feet), the effective vertical resolution becomes roughly 7 feet.
Enhancing the resolution of the logs is a significant problem, because subsurface formations are generally heterogeneous, so that porosity, saturation and lithology vary with position. A common example of heterogeneity is the presence in the formation of geological layers, or beds. Because logging tools have a nonzero volume of investigation, more than one layer may lie within the volume of investigation of a tool. In such cases, the petrophysical evaluation of one layer may be distorted by the presence of another layer falling within the larger volume of investigation of the tool. The above phenomenon leads to a specific problem in the analysis of subsurface formations that include one or more underground layers, especially when the layers are thin compared with the vertical resolution of the measuring tool. Such layers have become subject to significant commercial interest because of their production potential. Any knowledge about the composition and properties of such layered formations that helps better estimate their production potential has thus become increasingly valuable. It is clear therefore that any mechanism that for a given SNR supported by the formation can increase the vertical resolution of the tool without decreasing the logging speed is highly desirable.


A microprocessor has an interface between a reorder buffer and a floating point unit, including a retire signal provided by the reorder buffer and a valid signal provided by the floating point unit. When the reorder buffer detects a floating point instruction which is ready to be retired, the reorder buffer pulses the retire signal. When the floating point unit executes the floating point instruction and produces a corresponding instruction result, the floating point unit pulses the valid signal. Upon assertion of both the retire signal and the valid signal, the floating point instruction is retired by the floating point unit. The reorder buffer retires the floating point instruction upon asserting the retire signal. Either the valid signal or the retire signal may be asserted first (in a temporal sense) for the floating point instruction. The receiving unit for the signal asserted first stores the signal in a shift register until the receiving unit detects the particular floating point instruction. The received signal pulse is then associated with the particular floating point instruction. Additionally, the interface may include a cancel signal provided by the reorder buffer in cases in which a floating point instruction is canceled, as well as an exception signal provided by the floating point unit instead of the valid signal if the particular floating point instruction experiences an exception. Finally, the interface may include a pair of signals for synchronizing the floating point unit and the reorder buffer upon detection of a floating point synchronization instruction.

1. Field of the Invention
This invention relates to the field of microprocessors and, more particularly, to floating point units within microprocessors.
2. Description of the Related Art
Superscalar microprocessors achieve high performance by executing multiple instructions per clock cycle and by choosing the shortest possible clock cycle consistent with the design. As used herein, the term "clock cycle" refers to an interval of time accorded to various stages of an instruction processing pipeline within the microprocessor. Storage devices (e.g. registers and arrays) capture their values according to the clock cycle. For example, a storage device may capture a value according to a rising or falling edge of a clock signal defining the clock cycle. The storage device then stores the value until the subsequent rising or falling edge of the clock signal, respectively. The term "instruction processing pipeline" is used herein to refer to the logic circuits employed to process instructions in a pipelined fashion. Generally speaking, a pipeline comprises a number of stages at which portions of a particular task are performed. Different stages may simultaneously operate upon different items, thereby increasing overall throughput. Although the instruction processing pipeline may be divided into any number of stages at which portions of instruction processing are performed, instruction processing generally comprises fetching the instruction, decoding the instruction, executing the instruction, and storing the execution results in the destination identified by the instruction.
Microprocessors are configured to operate upon various data types in response to various instructions. For example, certain instructions are defined to operate upon an integer data type. The bits representing an integer form the digits of the integer number. The decimal point is assumed to be to the right of the digits (i.e. integers are whole numbers). Another data type often employed in microprocessors is the floating point data type. Floating point numbers are represented by a significand and an exponent. The base for the floating point number is raised to the power of the exponent and multiplied by the significand to arrive at the number represented. While any base may be used, base 2 is common in many microprocessors. The significand comprises a number of bits used to represent the most significant digits of the number. Typically, the significand comprises one bit to the right of the decimal, and the remaining bits to the left of the decimal. The bit to the right of the decimal is not explicitly stored, instead it is implied in the format of the number. Generally, the exponent and the significand of the floating point number are stored. Additional information regarding the floating point numbers and operations performed thereon may be obtained in the Institute of Electrical and Electronic Engineers (IEEE) standard 754. Floating point instructions, therefore, are instructions having at least one operand of the floating point data type.
Floating point numbers can represent numbers within a much larger range than can integer numbers. For example, a 32 bit signed integer can represent the integers between 2.sup.31 -1 and -2.sup.31, when two's complement format is used. A single precision floating point number as defined by IEEE 754 comprises 32 bits (a one bit sign, an 8 bit biased exponent, and a 24 bit significand) and has a range from 2.sup.-126 to 2.sup.127 in both positive and negative numbers. A double precision (64 bit) floating point value has a range from 2.sup.-1022 and 2.sup.1023 in both positive and negative numbers. Finally, an extended precision (80 bit) floating point number has a range from 2.sup.-16382 to 2.sup.16383 in both positive and negative numbers.
The expanded range available using the floating point data type is advantageous for many types of calculations in which large variations in the magnitude of numbers can be expected, as well as in computationally intensive tasks in which intermediate results may vary widely in magnitude from the input values and output values. Still further, greater precision may be available in floating point data types than is available in integer data types.
Many microprocessor architectures define the floating point instructions and operation thereof in such a way that a coprocessor-style design is favored. For example, the x86 microprocessor architecture defines the floating point in this manner. A coprocessor typically executes instructions provided by the processor relatively independent of the processor. An interface between the floating point unit and the integer portion of the microprocessor is therefore needed which preserves the independence of the floating point unit even though the floating point unit is integrated into the microprocessor.

Interface for coupling a floating point unit to a reorder buffer


A communication quality measuring apparatus includes a calculating unit, a specifying unit, and an associating unit. The calculating unit collects information that is used to calculate the communication quality of wireless communication from a base station that performs wireless communication with a mobile station and calculates the communication quality by using the information. The specifying unit collects location information from a measuring device that measures the location information related to the mobile station and specifies a mobile station area that is an area in which the mobile station is located by using the location information. The associating unit associates the communication quality calculated by the calculating unit with each of the mobile station areas specified by the specifying unit.

In recent years, various services have been provided by using wireless communication between mobile stations and base stations. However, if the effect of the environment in a communication area or the effect of noise is large, the communication quality of wireless communication between mobile stations and base stations decreases, and thus it may possibly be difficult to meet a predetermined communication quality.
To cope with the above situation, various technologies have been proposed that measures the communication quality of wireless communication between mobile stations and base stations in order to manage the communication quality for each communication area in which a mobile station is located. For example, a mobile station measures the transmission speed that corresponds to the communication quality of the wireless communication with a base station, and furthermore, the transmission speed is associated with each communication area that is specified by location information obtained by using a global positioning system (GPS) function. Furthermore, there is also a conventional technology in which the Radio Network Controller (RNC) that controls a base station measures the number of mobile stations under the base station and calculates, as the communication quality of a communication area, the communication rate that is obtained by dividing a fixed value of the transmission speed that is previously allocated to a communication area by the number of mobile stations.    Patent Document 1: Japanese Laid-open Patent Publication No. 2003-249887    Patent Document 2: Japanese Laid-open Patent Publication No. 2008-160310
However, with the conventional technology, there is a problem in that it is difficult to accurately measure the communication quality for each area in which a mobile station is located while avoiding an increase in the processing load of the mobile station.
Specifically, with the conventional technology in which a mobile station measures the transmission speed and the transmission speed is associated with each communication area, in addition to the processing load of the wireless communication with the base station, there is also a processing load for measuring the transmission speed; therefore, the processing load of the mobile station may increase.
Furthermore, with the conventional technology in which the RNC calculates the communication rate, an increase in the processing load of a mobile station can be avoided; however, because the communication rate is calculated as the communication quality by using the fixed value of the transmission speed, the accuracy of the calculated communication quality is possibly not maintained.

Communication quality measuring apparatus and communication quality measuring method


This invention relates to a process for papermaking which comprises forming a sheet in the presence of a neutral sizing agent and optionally a filler using as a paper quality improver a cationic polymer which is prepared by copolymerizing an acrylamide compound of the formula: EQU CH.sub.2 .dbd.C(R.sub.1)CONH.sub.2. wherein R.sub.1 is hydrogen atom or methyl group, and an inorganic or organic acid salt of a diallylamine compound of the formula: EQU CH.sub.2 .dbd.C(R.sub.2)CH.sub.2 !.sub.2 N--R.sub.3. wherein R.sub.2 is hydrogen atom or methyl group, R.sub.3 is hydrogen atom or an alkyl having 1 to 6 carbon atoms, in the presence or absence of a vinyl monomer copolymerizable with said acrylamide compound and/or said diallylamine compound salt.. When the above paper quality improver is used, the yield per each additive particularly in case of neutral or alkaline papermaking is excellent and hence the sizing properties are sufficiently given, and there can be obtained a paper having sufficiently improved paper strength.

There has recently been employed a papermaking process such as a so-called neutral or alkaline papermaking process which is carried out at pH 6 to 10, instead of an acidic papermaking process.
The neutral or alkaline papermaking process has advantages such that (1) a smaller energy for beating is used; (2) corrosion of machinery is reduced; (3) the white water system can be closed; (4) selection of fillers is not restricted and inexpensive calcium carbonate can be used; (5) deterioration of paper over time is reduce; (6) printability exhibited by the paper is good, and the like.
On the other hand, the neutral or alkaline papermaking process has drawbacks such that (1) expensive neutral sizing agents such as alkylketene dimers, alkenylsuccinic anhydrides, etc. must be used instead of inexpensive rosin sizing agents, and further, such agents are poor in anchoring onto the pulp; (2) fillers are poor in anchoring onto the pulp; (3) when the amount of calcium carbonate is increased, the strength of the paper is lowered; and the like.
It is known to use Mannich reaction products of an acrylamide polymer or acrylamide-acrylonitrile copolymer as an agent for reinforcing the paper strength in the papermaking under neutral conditions, but these agents are insufficiently anchored onto the pulp, and hence, satisfactory reinforcement of paper strength is not expected. Besides, there are also known polymers prepared by converting a part or whole of the tertiary amines in the above Mannich reaction products into quaternary groups. Although the products can increase the paper strength reinforcing effect in the papermaking step under neutral conditions in comparison with non-quaternary products, the improvement is still insufficient, and the anchoring effect of the neutral sizing agent is poor and hence the desired sizing effect is not achieved. Moreover, there is known a copolymer comprising predominantly (meth)acrylamide and a quaternary product of dimethylaminoethyl(meth)acrylamide, but it is still poor in the paper strength reinforcing effect, and the sizing agent is poor in anchoring effect. There is also known a copolymer comprising predominantly (meth)acrylamide and a quaternary product of dimethylaminopropyl(meth)acrylamide, but it is still poor in the paper strength reinforcing effect like the above polymer, and the sizing agent is poor in anchoring effect and hence a good sizing effect can not be achieved.
Besides, there is known a copolymer comprising predominantly (meth)acrylamide and diallyldimethylammonium chloride, but the sizing agent is poor in anchoring effect and hence the desired sizing effect can not be achieved.
Moreover, it is disclosed in Japanese Patent Second Publication (Kokoku) No. 12521/1972 to use a polymer having a partial quaternary group which is produced by copolymerizing diallylamine and a copolymerizable vinyl monomer e.g. (meth)acrylamide!, followed by modifying with an agent for converting into a quaternary group. However, such a polymer is particularly insufficient in the anchoring of the sizing agent and hence can not provide good sizing effect, while the paper strength reinforcing effect is improved.
Although there are also known one or more cationic polymers (e.g. polyamide/epichlorohydrin resin, etc.) and cationic starch, these are still not satisfactory in providing the desired effects. From these standpoints, it has been desired to develop an improved paper quality which is excellent in the yield of the product per sizing agent and filler and has excellent paper strength reinforcing effects.


A magnetic tunnel junction element (10) includes a configuration in which a reference layer (14) that includes a ferromagnetic material, a barrier layer (15) that includes O, a recording layer (16) that includes a ferromagnetic material including Co or Fe, a first protective layer (17) that includes O, and a second protective layer (18) that includes at least one of Pt, Ru, Co, Fe, CoB, FeB, or CoFeB are layered.

Patent Literature 1 discloses an MTJ element having a structure (double interface structure) in which a recording layer including CoFeB is interposed between a barrier layer (MgO) and a protective layer (MgO) including oxygen. An MTJ element having a double interface structure is also disclosed in Patent Literature 2 (paragraph 0011, FIG. 1). A protective layer in the MTJ element disclosed in Patent Literature 2 includes a conductive oxide layer. Patent Literature 2 further discloses a configuration in which a metal cap layer 13 is formed on an oxide cap layer 31 (protective layer).

Magnetic tunnel junction element and magnetic memory


A connector assembly has a plug connector and a socket connector. The plug connector has a first insulative housing and at least one row of first terminals. The first terminals are mounted on the first insulative housing and each terminal has an inside contacting portion and an interference portion. The socket connector has a second insulative housing and at least one row of second terminals corresponding respectively to the at least one row of the first terminals. The second terminals are selectively engaged and respectively with the first terminals and each second terminal has an inner contacting portion and a shoulder portion. The inner contacting portion selectively contacts the inside contacting portion of one first terminal. The shoulder portion is selectively engaged with the interference portion of one first terminal and prevents the first terminal from disengaging inadvertently from the second terminal.

1. Field of the Invention
The present invention relates to a connector assembly, and more particularly to a connector assembly that has a plug connector and a socket connector may be engaged securely with the plug connector.
2. Description of Related Art
Signal transmission between two printed circuit boards (PCBs) are implemented generally through a connector assembly having a plug connector on one PCB and a socket connector on the other PCB.
With reference to FIG. 9, a conventional connector 100 in accordance with the prior art comprises a plug connector 10 and a socket connector 20. The plug connector 10 has a row of first terminals. The socket connector 20 has a row of second terminals selectively contacting the row of the first terminal.
However, each first terminal contacts a corresponding second terminal merely in one single point without any locking structures. Therefore, the plug connector would be engaged loosely with and easily disengages inadvertently from the socket plug connector after repeated engagements and disengagements of the plug connector and the socket connector.
To overcome the shortcomings, the present invention provides a connector assembly to mitigate or obviate the aforementioned problems.


The invention provides biodegradable implants for treating sinusitis. The biodegradable implants have a size, shape, density, viscosity, and/or mucoadhesiveness that prevents them from being substantially cleared by the mucociliary lining of the sinuses during the intended treatment period. The biodegradable implants include a sustained release therapeutic, e.g., an antibiotic, a steroidal anti-inflammatory agent, or both. The biodegradable implants may take various forms, such as rods, pellets, beads, strips, or microparticles, and may be delivered into a sinus in various pharmaceutically acceptable carriers.

The paranasal sinuses are air-filled cavities within the facial skeleton. Each paranasal sinus is contiguous with a nasal cavity and drains into the nose through a sinus ostium. Although other factors may be involved, the development of sinusitis (inflammation of the mucosal lining of the sinuses) is most often attributed to blockage of one or more of these sinus ostia, followed by mucostasis and microbial overgrowth in the sinus cavity. Ostial blockage may stem from predisposing anatomical factors, or inflammation and edema of the mucous lining in the area of the ostia, arising from such etiologies as viral or bacterial upper respiratory infection or chronic allergic processes.
Traditionally, sinusitis has been medically managed by the oral administration of antibiotics and steroids. However, penetration of these systemically delivered agents into the sinus mucosa is limited due to poor blood flow to the sinuses. Therapeutic agents contained in aqueous solutions, creams, or gels, for topical application in the nose have also been formulated, but usually never travel far enough into the nose to reach the sinuses, are blocked from entering the sinuses due to obstructed ostia, or have such short contact with the sinus mucosa that absorption of the agent is low. For similar reasons, nasally inhaled steroid and anti-infective aerosols that have been developed to treat sinusitis are equally ineffective.
The delivery of ampicillin from a poly(lactic-co-glycolic)acid (PLGA) film to increase residence time of the antibiotic in rabbit sinuses has been investigated for the treatment of sinusitis (Min et al. Mucociliary Activity and Histopathology of Sinus Mucosa in Experimental Maxillary Sinusitis: A Comparison of Systemic Administration of Antibiotic and Antibiotic Delivery by Polylactic Acid Polymer. Laryngoscope 105:835-342 (1995) and Min et al. Application of Polylactic Acid Polymer in the Treatment of Acute Maxillary Sinusitis in Rabbits. Acta Otolaryngol 115:548-552 (1995)). Although clinical signs of sinusitis improved, the procedure for placing the film required that a hole be drilled through the anterior wall of the maxillary sinus.
Consequently, a biodegradable implant for administering a sustained release therapeutic agent to the paranasal sinuses for a prolonged time period without being substantially cleared by the mucociliary lining of the sinuses, and methods for delivering the implant in a minimally invasive fashion may provide significant medical benefit for patients afflicted with sinusitis.

This patent is subject to a terminal disclaimer.

Sinus delivery of sustained release therapeutics


A washing machine provided with a balancing device which may suppress variation in horizontal vibration of a spin basket without performing control operation. The balancing device includes a ring-shaped casing installed to be concentric with the spin basket, and a plurality of the balancing balls and a viscous fluid. A ring-shaped ball accommodation portion, a fluid accommodation portion provided at a lower inner side of the ball accommodation portion in a radial direction of the ball accommodation portion, and a communication channel are formed in the casing. The fluid accommodation portion is provided at a portion of the casing along a circumferential direction of the casing. A bottom surface of the ball accommodation portion is inclined downward such that the balancing balls gather at the opposite position facing the fluid accommodation portion when rotation of the spin basket is stopped.

1. Field
Embodiments of the present disclosure relate to a washing machine having an outer tub, a spin basket rotatably installed in the outer tub, and a balancing device mounted to the spin basket.
2. Description of the Related Art
Conventionally, a washing machine having an outer tub and a spin basket rotatably installed in the outer tub to perform washing, rinsing and drying of laundry such as clothing accommodated in the spin basket has been known.
The conventional washing machine is subjected to vibration when the spin basket having laundry accommodated therein rotates to perform drying operation. This vibration mainly results from uneven distribution of the laundry accommodated in the spin basket. Accordingly, a ball balancer to damp the vibration is mounted to the spin basket.
Specifically, the ball balancer is formed in a ring shape, and is provided with a plurality of balancing balls accommodated in an accommodation chamber and viscous fluid. The balancing balls and viscous fluid are allowed to move in the accommodation chamber in the circumferential direction of the accommodation chamber. The balancing balls are submerged in the viscous fluid and thus movement thereof is limited to an extent. Thereby, self-excited vibration of the balancing balls revolving in the accommodation chamber may be restricted. During rotation of the spin basket, the balancing balls are moved in the accommodation chamber in the circumferential direction by centrifugal force and finally positioned at the side opposite to the maldistributed laundry. As a result, balanced rotation of the spin basket is maintained by the weight of the balancing balls.
In the case that the positioned balancing balls fail to perform balancing during stoppage of rotation of the spin basket before start of the drying operation, the maldistribution of the balancing balls causes variation in magnitude of horizontal vibration (rocking rotation) of the spin basket while the rate of rotation of the spin basket crosses a first resonance section. If the horizontal vibration of the spin basket is large, the outer tub may collide with the outer casing. In addition, when the outer tub collides with the outer casing, control is generally performed to force the drying operation to be terminated. This may waste energy and time.
In the case of the conventional washing machine, a recessed accommodation portion to accommodate the balancing balls is formed on the bottom surface of the accommodation chamber. Thereby, unbalanced positioning of the balancing balls before start of the drying operation of the washing machine is resolved. Accordingly, when the rate of rotation of the spin basket passes the first resonance section, variation of the horizontal vibration of the spin basket due to maldistribution of the balancing balls may be suppressed.
However, the conventional washing machine has a viscous fluid accommodated in the accommodation chamber. Thereby, movement of the balancing balls may be restricted by the viscous resistance of the viscous fluid, and thus it may be difficult to accommodate the balancing balls in the recessed accommodation portion.
In this regard, a control operation may be performed to decelerate or accelerate rotation of the spin basket to move the balancing balls to positions where weight balance is formed between the balancing balls and the viscous fluid.
However, this control may take energy and time.

Washing machine with balancing device having fluid accommodation portion


Various solutions for bandwidth part (BWP) switch operations with respect to user equipment and network apparatus in mobile communications are described. An apparatus may receive a first downlink control information (DCI) to activate a BWP switch. The apparatus may cancel an uplink transmission or a downlink reception during a time duration. The apparatus may perform the BWP switch within the time duration.

Unless otherwise indicated herein, approaches described in this section are not prior art to the claims listed below and are not admitted as prior art by inclusion in this section.
In New Radio (NR), carrier bandwidth part is introduced for resource allocation in view of the wide frequency spectrum and the wide bandwidth. A carrier bandwidth part may be defined as a contiguous set of physical resource blocks. A user equipment (UE) may be configured with multiple BWP (e.g., 4 BWP) for performing uplink transmissions or downlink receptions. Only one BWP may be activated at a given time. Some mechanisms may be used to select a specific BWP as the active one (e.g., BWP selection or BWP switching). For example, the network node may use radio resource control (RRC) signaling or downlink control information (DCI) to configure/indicate a selected BWP.
Different BWP may correspond to different configurations. For example, the configurations may comprise BWP centre frequency, subcarrier spacing (SCS), bandwidth or other numerologies. When a new BWP is activated by a BWP index signalled in the DCI, the UE may need to start preparing for the BWP switch by re-calibrating its radio frequency (RF) chain for a different BWP (e.g., different BWP centre frequency, larger or smaller BWP size, different SCS, etc.). Accordingly, a BWP switch delay may be required for the UE to fulfil the BWP switch procedures.
The UE may know when it should start transmitting or receiving on the new BWP through a slot number indication (e.g., K0 or K2) signalled in the same DCI activating the BWP switch. However, it is unclear whether the UE can still transmit or receive any data or control signals after receiving the DCI triggering the BWP switch. The uplink signal transmission and the downlink signal reception may be affected or interrupted due to the BWP switch procedures.
Accordingly, how to perform the uplink signal transmission and the downlink signal reception when the BWP switch is configured may become important in the newly developed communication systems. Therefore, it is needed to provide proper BWP switch operations to avoid transmission errors or transmission interruptions caused by the BWP switch.

Method and apparatus for bandwidth part switch operations in mobile communications


An improved towing hook structure including a hook body having a surface provided with reinforcing ribs. The exterior of the hook body is wrapped in a layer of tough and wear-resistant rubber or plastic material using hot pressing to reduce possible damage to or scratching of the surface of a car to be towed. A rubber or plastic stop element and a rubber of plastic projection are integrally formed in the process of hot pressing to extend respectively from both ends of an opening of the hook body. The stop element is pushed into the projection prior to being completely cured so that it can elastically press upon and be retained in the opening of the hook body.

(a) Field of the Invention
The present invention relates to an improved towing hook, more particularly to a towing hook structure that is wrapped in a layer of tough and durable rubber or plastic material, and that has a rubber or plastic projection and a rubber or plastic stop element extending integrally from an opening of the towing hook, the stop element being pushed into the projection prior to curing, whereby the stop element can be elastically retained at the opening of the towing hook.
(b) Description of the Prior Art
Conventional towing hooks are usually made of metal. FIG. 1 shows a conventional metal towing hook 10. In use, when the towing hook 10 is fastened to a fastening ring of a car to be towed, it will inevitably scrape or scratch the surface of the car. In addition, as shown in FIG. 1, a stop piece 14 is pivotally connected to the towing hook 10 at its opening 11 by using a rivet 12 and a torsion spring so as to utilize the resetting action of the torsion spring 13 to stop the stop piece 14 at the opening 11 to thereby prevent slippage of the towing hook 10 after it has been attached to the fastening ring of the car. Assembly of the stop piece 14 is inconvenient.