A power supply device includes a first semiconductor switching device for controlling an alternating input current waveform, a smoothing capacitor to which a rectified voltage is applied, and an inverter that converts the rectified voltage into alternating current via a step-up chopper. The step-up chopper includes an inductor and a diode connected between the smoothing capacitor and inverter, and a second semiconductor switching device connected to the inductor and diode. The power supply device further includes an instantaneous voltage drop compensation function whereby the energy of the smoothing capacitor is supplied by an operation of the step-up chopper to the inverter when there is an instantaneous voltage drop in an alternating current power supply voltage. MOSFETs with a breakdown voltage lower than that of the first semiconductor switching device are connected between terminals of the step-up chopper, thus further reducing loss in comparison with when a bypass diode is used.

1. Field of the Invention
The present invention relates to a power supply device having a function of compensating for an instantaneous voltage drop in an alternating current power supply voltage, and specifically, relates to technology for increasing the efficiency and reducing the size of the device.
2. Description of the Background Art
FIG. 3 shows first heretofore known technology of a power supply device having a function of compensating for an instantaneous voltage drop in an alternating current power supply voltage.
In FIG. 3, reference sign 1 is an alternating current power supply, and a rectifier circuit formed of diodes 2 to 5 is connected to both ends of the alternating current power supply. An inductor 6 and a semiconductor switching device 7, such as a MOSFET, are connected in series to either end of a series circuit of the diodes 4 and 5, and a diode 8 and smoothing capacitor 9 are connected in series to either end of the switching device 7.
Direct current input terminals of an inverter INV formed of semiconductor switching devices 10 to 13 are connected to either end of the smoothing capacitor 9, and a primary winding 14a of a transformer 14 is connected to alternating current output terminals of the inverter INV.
A rectifier circuit formed of diodes 15 to 18 is connected to both ends of a secondary winding 14b of the transformer 14. An inductor 19 and a capacitor 20 are connected in series to either end of a series circuit of the diodes 17 and 18, and a load 21 is connected to both ends of the capacitor 20.
Functions required of the above-described power supply device are as follows:
converting an alternating current input voltage to a direct current voltage of a desired size, and keeping output voltage constant regardless of fluctuation in input voltage and load current;
isolating an alternating current input unit and a direct current output unit; and
controlling an alternating input current to a sinusoidal wave with a power factor of practically 1.
Furthermore, when the load 21 is one of which reliability is required, as with an information and communication instrument, the power supply device is required to have a function whereby it is possible to maintain a constant output voltage even in the event of a drop in the voltage of the alternating current power supply 1 for a period of in the region of a few milliseconds to a few cycles, a so-called instantaneous voltage drop (this function will hereafter be referred to as an instantaneous voltage drop compensation function).
A description will be given, while referring to FIGS. 4A to 4E, of operations for realizing these functions.
In FIG. 4A, an input voltage Vin from the alternating current power supply 1 has a sinusoidal waveform, while the output voltage (rectified voltage) of the rectifier circuit formed of the diodes 2 to 5 has the waveform indicated by Vr1 in FIG. 4B. Herein, on the switching device 7 of FIG. 3 being turned on when Vin has, for example, positive polarity, current flows along a path from the alternating current power supply 1 through the diode 2, inductor 6, switching device 7, and diode 5 to the alternating current power supply 1, the rectified voltage Vr1 is applied across the inductor 6, and a current IL increases.
Also, on the switching device 7 being turned off, the current IL flows along a path from the alternating current power supply 1 through the diode 2, inductor 6, diode 8, smoothing capacitor 9, and diode 5 to the alternating current power supply 1. At this time, a voltage equivalent to the difference between a voltage Ed1 of the smoothing capacitor 9 and the input voltage Vin is applied to the inductor 6, but as Ed1 is kept higher than a peak value of Vin by an operation of the circuit, the current IL decreases.
It is possible to control the waveform and amplitude of the current IL by controlling the duty ratio of the turning on and off of the switching device 7. When the waveform of the current IL is the kind of sinusoidal rectified waveform in FIG. 4B (ripple is ignored for the sake of simplification), an input current Iin has a sinusoidal waveform, as shown in FIG. 4A. Also, by controlling the amplitude of IL in accordance with the load power, it is possible to keep the voltage Ed1 of the smoothing capacitor 9 constant, and thus possible to keep the output voltage of the inverter INV constant.
Herein, FIGS. 4C to 4E show operations and voltage waveforms of each portion of the switching devices 10 to 13 when there is and is not an instantaneous voltage drop compensation function.
The inverter INV formed of the switching devices 10 to 13 converts the voltage Ed1 of the smoothing capacitor 9 into a high frequency alternating current voltage. A positive voltage Vt is applied to the primary winding 14a of the transformer 14 on the switching devices 10 and 13 being turned on, while a negative voltage Vt is applied to the primary winding 14a on the switching devices 11 and 12 being turned on, as shown in FIGS. 4C and 4D. By the positive and negative voltages Vt being applied alternately to the primary winding 14a in this way, the high frequency alternating current voltage Vt is input into the transformer 14 (the cycles of Vin and Vt are represented as being of the same extent in FIGS. 4A to 4E for the sake of easier understanding, but in general, Vin is of a commercial frequency of 50 or 60 Hz, while Vt is of a frequency of a few kilohertz or more in order to reduce the size of the transformer).
The transformer 14 isolates and transforms the input high frequency alternating current voltage Vt and, after converting the voltage across the secondary winding 14b of the transformer 14 into a rectified voltage Vr2 of FIG. 4E using the rectifier circuit formed of the diodes 15 to 18, smoothes the voltage using the inductor 19 and capacitor 20, and applies the voltage to the load 21 as an output voltage Vout. The output voltage Vout can be controlled by the duty ratio (hereafter referred to as the inverter duty ratio) of the turning on of the switching devices 10 and 13 or switching devices 11 and 12.
Operations when the power supply device includes an instantaneous voltage drop compensation function and the alternating current power supply 1 is sound are as shown under "Normal time" in FIGS. 4C to 4E.
As opposed to this, under "Time of instantaneous voltage drop", the input power decreases due to the occurrence of an instantaneous voltage drop in the voltage of the alternating current power supply 1, the voltage Ed1 drops, and the amplitude of the voltage Vt also decreases. However, provided that the drop in the voltage Ed1 is within a predetermined range, it is possible to keep the average value of the voltage Vt constant by increasing the duty ratio of the turning on of the switching devices 10 and 13 or switching devices 11 and 12, and to maintain the predetermined rectified voltage Vr2 and, by extension, the output voltage Vout.
However, when an instantaneous voltage drop compensation function is provided, the efficiency of the device decreases. This is for the following reasons.
In order to maintain the constant output voltage Vout even when the voltage Ed1 has dropped to a certain extent, it is necessary that the transformation ratio of the transformer 14 (the value of n in n:1, which is the turn ratio between the primary winding 14a and secondary winding 14b) is smaller than an essentially known optimum value. For example, when the voltage Ed1 is kept constant at 400V and the output voltage Vout is 10V at a normal time, the transformation ratio of the transformer 14 necessary to operate the inverter INV at a maximum duty ratio is 400:10, that is, n=40 (for the sake of simplification, voltage drop in the circuit is ignored here).
Meanwhile, the transformation ratio necessary in order to maintain the output voltage Vout at 10V even when the voltage Ed1 drops as far as 200V is 200:10, that is n=20. When setting the transformation ratio n under this condition, operation is carried out with an inverter duty ratio of approximately 0.5 in order to maintain Vout at 10V at a normal time, when the voltage Ed1 is 400V.
In this case, the amplitude of current flowing on the primary side of the transformer 14 is 1/n that of current flowing through the inductor 19, but when the power supply device has an instantaneous voltage drop compensation function, the transformation ratio n, whose original optimum value is 40, becomes 20, and the value of current flowing on the primary side of the transformer 14 increases due to the providing of the instantaneous voltage drop compensation function. Because of this, loss occurring in the switching devices 10 to 13 and primary winding 14a increases.
The rectified voltage Vr2 is approximately Ed1/n, but as the voltage applied to the diodes 15 to 18 at a normal time increases when the transformation ratio n decreases, it is necessary to use parts with a high breakdown voltage as the diodes 15 to 18. Generally, there is a tendency for loss in a semiconductor part to increase under the same conditions the higher the breakdown voltage, because of which loss occurring in the device increases.
Also, at a normal time and when there is an instantaneous voltage drop compensation function, a period for which the voltage Vr2 is not applied lengthens, because of which the value of the inductance of the inductor 19 necessary in order to smooth the voltage Vr2 increases. In the case in which there is no instantaneous voltage drop compensation function in FIGS. 4D and 4E, the rectified voltage Vr2 is not applied for the very short time in which the polarity of the voltage Vt switches in the period during which the rectified voltage Vr2 drops from a predetermined value to 0V, but at a normal time and when there is an instantaneous voltage drop compensation function, the rectified voltage Vr2 is not applied for a period equivalent to one-half of one cycle, and for this period it is necessary that energy is supplied to the load 21 by the inductor 19.
For these reasons, the inductor 19 increases in size, which results in an increase in the overall size of the device and an increase in loss occurring in the inductor 19.
The circuit shown in FIG. 5 is known as second heretofore known technology whereby the above-described kind of increase in loss from the inverter INV onward is avoided.
In FIG. 5, an inductor 22 and a diode 24 are connected in series to a positive side direct current bus between the smoothing capacitor 9 and inverter INV, and a semiconductor switching device 23 is connected between a connection point of the inductor 22 and diode 24 and a negative side direct current bus. Also, a smoothing capacitor 25 is connected between the cathode of the diode 24 and the negative side direct current bus. A step-up chopper is configured of the inductor 22, diode 24, switching device 23, and smoothing capacitor 25. As the other configurations in FIG. 5 are the same as in FIG. 3, a description thereof will be omitted.
It is possible to control current flowing through the inductor 22 with the step-up chopper, using the same kind of operation as that of the circuit formed of the inductor 6, switching device 7, and diode 8, as a result of which it is possible to obtain an output voltage Ed2 higher than the input voltage Ed1. That is, using an operation of the step-up chopper, it is possible to keep the voltage Ed2 of the capacitor 25 constant (for example, 400V) even when the voltage Ed1 of the capacitor 9 drops (for example, from 400V to 200V) when there is an instantaneous voltage drop in voltage, and a design wherein the transformation ratio is n=40 is possible in the previously described example.
A circuit wherein a drop in alternating current power supply voltage is compensated for using the above-described step-up chopper is shown in JP-A-2-241371 (page 2, bottom right column, line 1 to page 3, top left column, line 4, FIG. 2 and the like).
However, the circuit shown in FIG. 5 has the following drawbacks.
That is, the inductor 22, switching device 23, and diode 24 configuring the step-up chopper generate loss. While inevitable in a case in which the switching device 23 is carrying out a turning on or turning off operation, loss also occurs when the voltage Ed1 is sufficiently high and the switching device 23 is stopped, due to the winding resistance of the inductor 22 and the forward voltage drop of the diode 24. Because of this loss, the amount by which the circuit loss from the inverter INV onward is reduced is cancelled out.
Also, as current is constantly flowing through the inductor 22, an inductor 22 with a large current capacity is needed, despite the time for which the stepping-up operation is carried out being extremely short.
Furthermore, a large capacity smoothing capacitor 25 is needed in order to absorb ripple current generated from the inverter INV. As the one smoothing capacitor 9 has a sufficiently large capacity in order to supply energy when there is an instantaneous voltage drop, the smoothing capacitor 9 can perform the role of absorbing both the ripple current generated by the circuit formed of the inductor 6, switching device 7, and diode 8 and the ripple current of the inverter INV in the circuit of FIG. 3.
However, as the inductor 22 is inserted between the inverter INV and smoothing capacitor 9 in the circuit of FIG. 5, a high frequency ripple current can not pass. Because of this, the separate smoothing capacitor 25 is needed, because of which the size of the device increases.
A circuit wherein energy when there is an instantaneous voltage drop is supplied by another capacitor charged in advance is shown in JP-A-8-185993 (paragraphs [0023] to [0028], FIG. 1 and the like). However, as the circuit disclosed in JP-A-8-185993 (paragraphs [0023] to [0028], FIG. 1 and the like) is also such that a capacitor that absorbs ripple at a normal time and a capacitor that supplies energy when there is an instantaneous voltage drop are separated, it is not possible to avoid an increase in the size of the device.
The circuit shown in FIG. 6 is known as third heretofore known technology whereby the above-described increase in the size of the device is avoided.
In FIG. 6, reference sign 101 is a bypass diode connected between the cathodes of the diodes 8 and 24, while the other configurations are the same as in FIG. 5. At a normal time when the alternating current power supply 1 is sound, the switching device 23 does not operate, and the step-up chopper formed of the inductor 22, switching device 23, diode 24, and capacitor 25 is bypassed by the diode 101. Herein, as the voltages Ed1 and Ed2 are smoothed direct current voltages both at a normal time and when there is a instantaneous voltage drop, the diode 101, unlike the diode 24, does not need to be capable of rectifying a high frequency, so it is possible to use a low speed diode. As the forward voltage of a low speed diode is in the region of one-half compared to that of a high speed diode, and no current flows through the inductor 22 at a normal time, it is possible to reduce loss considerably in comparison with that in the circuit of FIG. 5.
Furthermore, as the time for which the inductor 22 is energized is a few tens of milliseconds or less when there is an instantaneous voltage drop, it is possible to use a short time rated inductor with thin windings as the inductor 22, and thus possible to reduce the size of the inductor 22 considerably in comparison with that in the circuit of FIG. 5. Also, as there is no longer an inductor interposed between the smoothing capacitor 9 and inverter INV owing to the bypassing operation of the diode 101, it is possible for the smoothing capacitor 9 to absorb the ripple current of the inverter INV too at a normal time. Because of this, it is sufficient that the smoothing capacitor 25 withstands only the ripple of the step-up chopper and inverter INV during an instantaneous voltage drop period, and thus possible to use a capacitor with an extremely small capacity.
This method is shown in, for example, JP-A-2010-41910 (paragraphs [0040] to [0046], FIG. 1 and the like).


A vibration-type driving unit includes a vibrator including an electro-mechanical transducer; a moving body provided with the vibrator; a contact member in contact with the vibrator provided on the moving body; a support member configured to support the contact member; and a moving mechanism configured to move the moving body relative to the support member. The vibrator and the contact member are disposed so that the vibrator moves in a first direction relative to the contact member. The moving mechanism is configured to move the moving body relative to the support member in a second direction intersecting the first direction in a plane parallel to a plane in which the vibrator and the contact member are in contact with each other.

1. Field of the Invention
The present invention relaters to a vibration-type driving unit, as well as a two-dimensional driving apparatus, an image-blur correction apparatus, an interchangeable lens, an image capturing apparatus, or an automatic stage including the vibration-type driving unit.
2. Description of the Related Art
Great many vibration-type driving units that excite vibrations in a vibrator to move a moving body that is in pressure-contact therewith have been proposed.
They are positioned as important functional components particularly in optical devices which are required to operate accurately. An example thereof is a two-dimensional driving apparatus including a plurality of linear vibration-type driving units (a type capable of linearly moving a moving body), in which the driving operations thereof are combined, allowing two-dimensional movement of the moving body.
Another known example is an image-blur correction apparatus using this two-dimensional driving apparatus (for example, Japanese Patent Laid-Open No. 2012-44832).
However, the method of combining driving operations of a plurality of vibration-type driving units has the following problems to be solved.
For example, the moving direction of the moving body and the driving direction of some of the plurality of vibration-type driving units sometimes intersect at right angles or at angles close to right angles depending on the moving direction of the moving body.
In this case, not only the vibration-type driving unit whose driving direction intersects the moving direction at right angles or angles close to right angles cannot contribute to driving, but also a frictional force due to the contact between the moving body and the vibration-type driving unit exerts a load on the movement of the moving body to cause an energy loss, resulting in an output loss.
This may cause a decrease in characteristics when used as, for example, a two-dimensional driving apparatus, an image-blur correction apparatus, an interchangeable lens, an image capturing apparatus, or an automatic stage.
To solve these problems, Japanese Patent Laid-Open No. 2012-44832 discloses a configuration having a moving mechanism (a deflecting mechanism) that allows a vibrator to move in a direction (a deflecting direction) intersecting the driving direction.
Therefore, the vibrator and the moving body are configured not to move relative to each other in the deflecting direction.
By forming the deflecting mechanism with a guide mechanism having a remarkably small sliding loss or an elastic spring having low rigidity, a load that acts on the moving body can be made extremely small. This remarkably reduces the output loss.
However, the output-loss reduction unit for the vibration-type driving unit, the two-dimensional driving apparatus, or the image-blur correction apparatus described above mainly has the following problems.
The first problem is that the occupied space is large.
In the related art example described above, a component in contact with the vibrator is moved in the driving direction of the vibration-type driving unit by the vibration of the fixed vibrator, thereby moving the moving body fixed thereto.
To allow a predetermined amount of movement of the moving body, the sum (length) of the dimension (length) of the component in contact with the vibrator and the amount of movement (moving distance) thereof is needed as an occupied space (length), thus increasing the size of the apparatus.
Another problem is that unnecessary vibrations are prone to occur in the component in contact with the vibrator.
Unnecessary vibrations make the state of contact between the vibrator and the component in contact with the vibrator unstable, leading to unstable output.

Vibration-type driving unit, two-dimensional driving apparatus, image-blur correction apparatus, interchangeable lens, image capturing apparatus, and automatic stage


A floating, offshore drilling and/or production platform such as a tension leg platform, semi-submersible, spar or the like is equipped with a ballast tank system that comprises the traditional tank and a shaft that runs typically vertically from top-of-hull level to the top of the tank. This shaft is large enough to allow the ballast pipe, sounding lines, instrumentation piping, etc. to be installed within it. In certain embodiments, the shaft itself functions as a vent line.

1. Field of the Invention
This invention relates to floating offshore platforms. More particularly, it relates to the water ballast and bilge systems used in floating offshore platforms.
2. Description of the Related Art Including Information Disclosed Under 37 CFR 1.97 and 1.98
Various floating structures are used for offshore drilling and production operations. Examples include tension leg platforms (TLP's), semi-submersible floating vessels ("semis"), spars and floating production, storage and offloading vessels (FPSO's). Each of these vessel types may include a water ballast system.
A ship may have a single ballast tank near its center or multiple ballast tanks typically on either side. A large vessel typically will have several ballast tanks including double bottom tanks, wing tanks as well as forepeak and aftpeak tanks. Adding ballast to a vessel lowers its center of gravity, and increases the draft of the vessel. Increased draft may be required for proper propeller immersion.
A ballast tank can be filled or emptied in order to adjust the amount of ballast force. Ships designed for carrying large amounts of cargo must take on ballast water for proper stability when travelling with light loads and discharge water when heavily laden with cargo. Small sailboats designed to be lightweight for being pulled behind automobiles on trailers can be designed with ballast tanks that may be emptied when the boat is removed from the water.
In submarines, ballast tanks are used to allow the vessel to submerge, water being taken in to alter the vessel's buoyancy and allow the submarine to dive. In order for the submarine to surface, water is displaced from the tanks using compressed air, and the vessel becomes positively buoyant again, allowing it to rise to the surface. A submarine may have several types of ballast tank: the main ballast tanks, which are the main tanks used for diving and surfacing, and trimming tanks, which are used to adjust the submarine's attitude (its 'trim') both on the surface and when underwater.
Ballast tanks are also integral to the stability and operation of deepwater offshore oil platforms and similar structures such as floating wind turbines. The ballast facilitates hydrodynamic stability by moving the center-of-mass as low as possible, placing it beneath the air-filled buoyancy tank(s).
Vessels such as TLP's, semis, spars and FPSO's typically require ballast tanks for weight and trim control. These tanks are usually distributed throughout the vessel, and a piping system allows these tanks to be filled and drained (ballasted). The piping system is typically located near the lower parts of each tank.

Ballast system for floating offshore platforms


A USB connector with orientation adjustment is disclosed. The USB connector comprises a top cover, a connector seat with a cable slot, a cable rotating seat, cable and a bottom cover. The top cover is engageable with the bottom cover enclosing the connector seat and the cable rotating seat; the connector seat is a structure for connecting the branched wire of the cable, and the cable slot to the cable; the cable rotating seat is a circular seat plate having a pivot slot at the center of the rotating seat, and the bottom section of the circular seat plate is provided with a plurality of positioning holes and the lateral side of the circular seat plate is provided with through holes; the end of cable passes through the lateral through hole of the pivot slot of the cable rotating seat, and the branched cable is connected to the cable slot of the connector seat; the bottom cover is engaged with the top cover by means of peg and the covers encloses the connector seat and the cable rotating seat, and an engaging body mounted below the bottom cover is engaged with an engaging slot of the connector seat such that the connector seat exposes only a fixed length, and the rear section of the bottom cover is a protruded circular engaging plate which is directly pivotally mounted to the pivot slot of the cable rotating seat and is engaged at the positioning hole of the rotating seat by the positioning points.

(a) Technical Field of the Invention
The present invention relates to USB connector, and in particular, a USB connector with direction adjustment mechanism, allowing a connector cable to turn from 0 degree to 180 degrees direction.
(b) Description of the Prior Art
Computers and digital devices are generally provided with USB interface connection port for external connection of memory. Conventional USB connector is shown in FIG. 1 which is an insertion connector A1 having an exposed USB insertion end A2 which can be inserted to the USB interface connection port. The rear end is a simple cable structure with no direction adjustment mechanism. Thus, the conventional USB connector is inherited with drawbacks as follows:                the connectors of the digital devices and computers together with the cables are entangled with each other as a result of the fixed direction mechanism connecting the cable to the USB connector;        the cable after distorted or twisted shall not be able to restore to the original shape and it may causes the wire within the cable to break; and conventional USB connector is easily dislocated due to constantly pulling and twisting of cable.
As a result of the drawbacks mentioned above, it is an object of the present invention to provide a USB connector with orientation adjustment to overcome or mitigate the shortcomings.


A test circuit for stress testing antifuses before programming. The test circuit provides a voltage to an antifuse detection circuit during antifuse stress testing. In one embodiment, the provided voltage is externally received at a probe pad. In another embodiment, the test circuit controls a voltage generating circuit output voltage from a normal operating voltage to a stress voltage, such as by shifting the ground reference for the voltage generating circuit. The stress voltage can be varied as needed for a particular test setup and/or for different batches of antifuse circuits. Since the stress voltage is independent of the power supply voltage VCC, antifuse stressing can be concurrent with other pre-fuse tests, obviating the need for a dedicated antifuse stress test and reducing test time.

Programmable logic devices, such as antifuses, are commonly used for changing the connectivity of an integrated circuit after fabrication. For example, one application of antifuses is in the repair of integrated circuit memory devices, such as random access memory (RAM) devices. Memory devices are often fabricated with several redundant elements (rows or columns) of memory cells that can be substituted for similar elements that are found to be defective. Antifuses can be selectively programmed to reroute the addresses of failing elements (rows or columns) such that functional redundant elements are addressed instead.
As fabricated in its unblown condition, the antifuse functions as a capacitor and presents a very high resistance, e.g., 10 Megaohms, between its terminals. The antifuse is programmed by shorting its terminals together, providing a relatively low resistance path of approximately between 200 ohms and 10 kiloohms. The programming is carried out by applying, across the terminals of the antifuse, a programming voltage of sufficient amplitude to cause breakdown of the dielectric interposed between its terminals.
Integrated circuit memory devices undergo extensive testing during manufacturing. Antifuses included in such memory devices are also tested prior to programming the antifuses. Antifuses are tested for failure mechanisms including defects in the dielectric between its terminals, defects in the contacts to the antifuse terminals, and other antifuse fault conditions. It is often desirable to provide an antifuse stress test, in which a voltage magnitude greater than an operating voltage, but less than a programming voltage is applied across the terminals of an antifuse to identify weak or defective antifuses, such as may result from dielectric defects, for example. The antifuse stress test is needed prior to actual programming of the antifuses. Since antifuse stress testing requires a different stress voltage than used during testing of the RAM memory cells, a separate testing operation is required to accomplish stress testing of antifuses.
Testing the RAM memory cells involves placing the integrated circuit memory device in a test mode, as distinguished from its normal operating mode, such as by applying a test function voltage, commonly referred to as a "supervoltage", to one of the external pins of the integrated circuit device. In one example, the supervoltage is 4.5 volts higher than a chip power supply voltage VCC. The integrated circuit memory device includes a circuit that detects the supervoltage and generates a test enable signal, which is applied to test circuits of the device for initiating various test procedures as is known in the art.
FIG. 1, which is labeled "Prior Art", illustrates one known stress test circuit that is used in antifuse test procedures for applying an antifuse stress voltage. During normal operation, the test circuit couples a voltage DVC2E produced at node 120 by a voltage generating circuit 102 to an output node 104 which is connected to a terminal 204 of a conventional antifuse detection circuit, shown in FIG. 2, which is labeled "Prior Art". During normal operation, the antifuse detection circuit is used to indicate whether the antifuse 202 is in an "unblown" or "blown" condition. During a test mode of operation, the test circuit decouples the voltage generating circuit 102 from the output node 104 and couples the output of a tristateable driver 106 to the output node 104 for applying to output node 104 a stress voltage that is derived from the power supply voltage VCC.
Voltage generating circuit 102 includes p-channel transistors 108 and diode-connected, n-channel transistors 110. The test circuit includes a gating circuit 112, embodied as a multiplexer, an enabling circuit 114, including a NOR gate 116 and an inverter 118, and the tri-stateable driver 106. The enabling circuit 114 determines the state of the multiplexer 112 based upon the logic state of test signal SUPERVOLT that is applied to an input of NOR gate 116.
During normal memory device operation, test signal SUPERVOLT is at a logic low level, and the output at node 120 of the voltage generating circuit 102 is coupled to the test circuit output at node 104 through the multiplexer 112. When test signal SUPERVOLT is at a logic high level, the multiplexer 112 is disabled, decoupling the voltage generating circuit 102 from the test circuit output at node 104. Tri-stateable driver 106 has inputs connected to receive the complementary state inputs provided at the output 122 of the NOR gate 116 and the output 124 of inverter 118, respectively, for controlling its operation. Tri-stateable driver 106 also receives an antifuse stress enable signal AF.sub.-- STRESS. When stress enable signal AF.sub.-- STRESS and test signal SUPERVOLT are both at a logic low level, tristateable driver 106 is disabled, providing a high impedance level to the output node 104. When the stress enable signal AF.sub.-- STRESS and test signal SUPERVOLT each become a logic high level, the tri-stateable driver 106 applies a voltage corresponding to the power supply voltage VCC to the output node 104. Output node 104 is connected to a node 204 of the detection circuit which, in turn, is connected to a control terminal of a pass gate, e.g. the gate of an n-channel transistor 206 (FIG. 2) of the antifuse detection circuit.
In FIG. 2, one terminal of the antifuse 202 is connected to the power supply voltage VCC through series-connected p-channel transistors 208, 210 and n-channel transistors 206, 212. The gate terminal of n-channel transistor 212 is coupled to the supervoltage Vccp!, which is typically more positive than the positive supply voltage VCC, as described above. The other terminal of the antifuse is connected to ground through series-connected n-channel transistors 214 and 216. Transistor 206 functions as a pass gate that is controlled by the voltage DVC2E. Transistor 206 limits the voltage at node 207 to antifuse 202 to a level corresponding to DVC2E-V.sub.T, which is less than the power supply voltage VCC, where V.sub.T is the threshold voltage drop of the pass gate transistor 206.
Referring again to FIG. 1, during the antifuse testing procedure, a test signal "SUPERVOLT" disables multiplexer 112, decoupling voltage generating circuit 102 from output node 104. When the antifuse stress enable signal AF.sub.-- STRESS is also provided, tri-stateable driver 106 applies a voltage corresponding to the power supply voltage VCC to the output node 104 for application to the pass gate 206 (FIG. 2). Thus, the voltage that is applied to the control input of pass gate 206 during the antifuse stress test is the power supply voltage VCC. However, stress voltage levels that are different from the levels of power supply voltage VCC (e.g., approximately between 2.5 volts and 8 volts, may be desired during the antifuse stress testing. This is difficult to obtain because the power supply voltage level VCC is established by application of a common external power supply voltage. Moreover, the levels of the supply voltage VCC are not optimum for antifuse stress testing. As a result, a dedicated test is required for antifuse stress testing, i.e. separate from other tests of RAM memory cells. This increases test time, and correspondingly increases manufacturing costs.
For the reasons stated above, and for other reasons stated below which will become apparent to those skilled in the art upon reading and understanding the present specification, there is a need in the art for a test circuit and method for stressing antifuses prior to programming the antifuses, which is independent of the level of the chip supply voltage. There is a further need in the art for such a test circuit and method which eliminates the need for a dedicated test time for antifuse stress testing such that antifuses can be stressed while other circuits on the integated circuit die are undergoing normal testing.

Circuit and method for antifuse stress test


An apparatus and method for removing skin and fat from meat parts, a tooth roll assembly attached to a frame and a trimming blade assembly attached to an adjustable frame where the position of the trimming blade assembly is adjusted in relation to the tooth roll assembly based upon the thickness of a layer of fat of the meat part.

This invention is directed toward a trimming device for removing skin and fat from meat parts, and more particularly toward a trimming device that is adjustable based on the fat depth of the meat part.
In the meat processing industry it is necessary and desirable to remove the skin and fat from a meat part, such as a pork leg, before separating the muscles for processing. To date, this has been done manually, using knives. Not only does this method require substantial labor, but it is also time consuming and creates safety hazards creating the potential for lacerations and cumulative trauma injuries. The manual process also has the potential for yield loss because the trimmer has difficulty identifying the lean muscle under the fat layer prior to cutting into the fat layer.
Alternatively, automated systems have been used to remove the skin, but the fat layer is still removed manually, presenting the same problems in the art. With some skinning machines, defatting attachments have been added to remove the layer of fat. These defatting attachments are adjustable to remove different thicknesses of fat layer, but the machine operator must estimate the thickness visually, and then manually adjust the attachment. This process is time consuming and inaccurate such that either additional manual trimming is required, or some lean muscle is lost. Accordingly, a need exists in the art for an improved method and apparatus for removing skin and fat from a meat part.
Therefore, a primary objective of this invention is to provide a method and apparatus for removing skin and fat from a meat part that reduces lost yields and labor costs.
Another objective of this invention is to provide a method and apparatus that is more precise in trimming skin and fat from a meat part.
These and other objectives will be apparent to those skilled in the art based on the following disclosure.

Method and apparatus for removing fat and skin from meat parts


A banana carrier, particularly a banana carrier is provided for transporting bananas on a plurality of trays from the picking fields to the packing area. The banana carrier is particularly adapted to be hung in rolling fashion on a cableway network arranged in the picking fields. The trays are arranged in stacked spaced apart fashion on the banana carrier. Each tray provides quarter panel sloped surfaces, circumscribed by drainage troughs which communicate to a drainage hole located in a central portion of each tray.

This invention relates to a cableway system used in agricultural fields to provide transportation of hand-harvested crops from the fields to the packing area. In particular, this invention relates to a tray arrangement for a banana carrier which hangs from the cableway system for effectively transporting bananas.
Cableway systems arranged throughout picking fields or farms for handling and transporting produce are known. It is also known to transport bananas along a cableway system by hanging the entire banana bunch from the cableway for transport.
It has been found however, that transporting the bananas in this manner by hanging the entire bunch results in a considerable amount of damaged or lost bananas, by the time the bunch reaches the packing area. The banana bunches can swing into stationary objects while being transported, such as supports for the cableway, causing bananas to bruise or separate from the bunch.


A fuel system for an engine is disclosed. The fuel system has a source of pressurized fuel, a plurality of fuel injectors, and a common manifold configured to distribute pressurized fuel from the source to the plurality of fuel injectors. The fuel system also has a first sensor located upstream of the common manifold, and a second sensor associated with the engine. The first sensor is configured to generate a first signal indicative of a fuel temperature. The second sensor is configured to generate a second signal indicative of a speed of the engine. The fuel system further has a controller in communication with the first and second sensors. The controller is configured to estimate a fuel temperature at each of the plurality of fuel injectors based on the first signal, the second signal, and the position of the plurality of fuel injectors along the common manifold.

Internal combustion engines such as diesel engines, gasoline engines, and gaseous fuel-powered engines use injectors to introduce fuel into the combustion chambers of the engine. As the fuel is pressurized, directed through portions of the engine to individual injectors, and returned from the injectors, the fuel absorbs heat from its surroundings and from the work exerted on the fuel. As the fuel is heated, properties of the fuel affecting injection characteristics change. In addition, because fuel heating throughout the engine can vary during operation of the engine, the fuel temperature and, thus, the injection characteristics at one injector may be different from the fuel temperature and injection characteristics at another injector. If these varying temperature and injection characteristics are not accounted for during operation of the engine, the injection of fuel into the engine and subsequent operation of the engine may be unpredictable.
In order to account for these fuel temperature and injection characteristic changes, engine manufacturers have attempted to estimate the fuel temperature at each injector. One such example is disclosed in U.S. Pat. No. 5,865,158 (the '158 patent) issued to Cleveland et al. on Feb. 2, 1999. The '158 patent describes a method and system for controlling the injection of fuel across a plurality of fuel injectors coupled together along a fuel rail in an internal combustion engine. The method includes producing a reference fuel delivery control signal for each fuel injector as a function of a desired fuel mass to be injected. The method further includes adjusting the pulse width of each fuel delivery control signal as a function of the fuel temperature proximate each of the fuel injectors. The temperature of the fuel proximate each injector is ascertained by first measuring the temperature of the fuel near the inlet of the fuel rail. This measured temperature is then offset based on the location of the fuel injector along the rail to determine the temperature of the fuel proximate each injector.
Although the method and system of the '158 patent may estimate the fuel temperature at each injector and control operation of the injectors in response thereto, it may lack accuracy. In particular, the 158 system does not take into account fuel that is directed to other fuel-powered engine accessories or the effect their operation may have on fuel temperature. In addition, the 158 patent does not take into account the current steady-state or transient operation of the engine when determining fuel temperature.
The system and method of the present disclosure solves one or more of the problems set forth above.

Method and system for estimating injector fuel temperature


Disclosed is a device for packaging balls for reaction vessels for an analysis appliance, including: a body including at least one opening and used to house balls, a closing member, that can be moved relative to the body between a closed position in which the closing member closes the opening of the body, and an open position in which the opening of the body is open so as to allow the balls to be dispensed from the body.

Reaction vessels are already known from documents EP 0 325 874 and WO 03/065047 and are used to determine the times of modification of the physical state of a medium, in particular to determine the coagulation time of a blood sample. These vessels are open at their upper ends and have curved bottoms forming raceways of a ball of ferromagnetic material that is that can be moved in the vessel in periodic movement using an external magnetic field. Variations in the amplitude and/or frequency of ball movements are representative of the physical condition of the blood sample.
In WO-A-03/065047, the vessels containing the balls are detachably attached side by side to a flexible carrier film that seals their upper ends and can be wound on a reel to feed an automatic analysis appliance and scroll the vessels in succession through the appliance.

Device for packaging balls for reaction vessels for an analysis appliance


A process for producing a heterologous protein is provided. The process comprises culturing an E. coli strain in a fermentation medium, the E. coli strain harboring a gene which codes for the heterologous protein and is functionally linked to a signal sequence coding for a signal peptide, and releasing the heterologous protein into the fermentation medium, and the heterologous protein being removed from the fermentation medium, wherein the E. coli strain has an attenuated (p)ppGpp-synthetase II activity (PSII activity).

1. Field of the Invention
The present invention relates to a process for the fermentative production of heterologous proteins using an Escherichia coli strain having an attenuated (p)ppGpp-synthetase II activity.
2. Background Art
The market for recombinant protein pharmaceuticals (pharmaceutical proteins/biologics) has shown strong growth in recent years. Particularly important protein pharmaceuticals are eukaryotic proteins, especially mammalian proteins and human proteins. Examples of important pharmaceutical proteins (pharmaceutically active proteins) are cytokines, growth factors, protein kinase, protein hormones and peptide hormones, and antibodies and antibody fragments. Owing to the costs of producing pharmaceutical proteins, which are still very high, there is a continuous search for more efficient and therefore more cost-effective processes and systems for producing the proteins.
Recombinant proteins are usually produced either in mammalian cell cultures or in microbial systems. Microbial systems have the advantage over mammalian cell cultures in that recombinant proteins can be produced in this way more rapidly and with lower costs. Consequently, bacteria are especially suitable for producing recombinant proteins. The gram-negative enterobacterium Escherichia coli (E. coli) is currently the most frequently used organism for producing recombinant proteins, owing to its very well studied genetics and physiology, short generation time and easy manipulation. Production processes for recombinant proteins in E. coli, which involve the correctly folded target protein being secreted with high yield directly into the fermentation medium are particularly useful.
The literature has disclosed a number of E. coli strains and processes using E. coli strains, by which recombinant proteins are secreted into the fermentation medium. Thus, for example, US2008254511 describes various E. coli strains that have a mutation in the Braun lipoprotein (Lpp) gene and export the overproduced heterologous proteins into the medium.
In order to regulate and coordinate the metabolism under conditions, in which the cell is deficient of amino acids or the energy source, E. coli has a mechanism which is referred to as "stringent control". In this, the alarmones guanosine tetraphosphate (ppGpp) and guanosine pentaphosphate (pppGpp), usually combined as (p)ppGpp hereinbelow, play an important part as regulatory signal molecules, inter alia by slowing down the synthesis of stable RNAs (rRNA, tRNA) and enhancing expression of some amino acid biosynthesis genes, due to an increase in the (p)ppGpp level in the cell (Cashel et al., 1996, Escherichia coli and Salmonella Cellular and Molecular Biology, ed. Neidhardt, F. C. ASM Press, Washington, D.C. pp. 1458-1496). E. coli has two enzymes for synthesizing (p)ppGpp from ATP and GDP (ppGpp) and, respectively, ATP and GTP (pppGpp): (p)ppGpp synthetase I (PSI), encoded by the relA gene, is responsible for (p)ppGpp synthesis under stringent control, when the cells are subject to an amino acid deficiency. However, if cell growth is slowed down due to the primary carbon source being exhausted, (p)ppGPP is synthesized predominantly by (p)ppGpp synthetase II (PSII) which is encoded by the spoT gene. In contrast to PSI, the spoT gene product has also still a (p)ppGpp-3′-pyrophosphohydrolase activity in addition to the (p)ppGpp-synthetase activity, and accordingly can also actively control the cellular (p)ppGpp level by breaking down this compound (Gentry and Cashel, 1996, Mol. Microbiol. 19, 1373-84). The active sites of the two different catalytic activities of the PSII enzyme were shown to be non-identical but nevertheless located in close spatial proximity in the N-terminal region of the protein.
Both E. coli relA mutants and E. coli spoT mutants have been disclosed (Cashel et al., 1996, Escherichia coli and Salmonella Cellular and Molecular Biology, ed. Neidhardt, F. C. ASM Press, Washington, D.C. pp. 1458-1496). Strains having a relA1 mutation, with the IS2 insertion element being integrated in the relA gene sequence (relA::IS2), still possess a small residual PSI activity, whereas relA deletion mutants (ΔrelA) no longer have any PSI activity (Metzger et al., 1989, J. Biol. Chem. 264, 21146-52). In contrast to relA wild-type strains which have a functioning stringent control mechanism ("stringent" phenotype), relA mutants with a reduced or missing PSI activity exhibit a "relaxed" phenotype when encountering an amino acid deficiency situation. The latter phenotype manifests itself inter alia in that the cell is unable to accumulate (p)ppGpp above the basal level, and that the synthesis of stable RNA molecules is not stopped but continues undiminished. The basal level means the (p)ppGpp concentration range (10-30 pmol/A450) for exponentially growing cells under non-limiting growth conditions (i.e. before encountering an amino acid deficiency situation), as disclosed in the literature (Cashel et al., 1996, Escherichia coli and Salmonella Cellular and Molecular Biology, ed. Neidhardt, F. C. ASM Press, Washington, D.C. pp. 1458-1496). Gitter et al. (1995, Appl. Microbiol. Biotechnol. 43, 89-92) have disclosed that the E. coli relA mutant CP79, in comparison with the relA wild-type strain CP78, increasingly releases into the culture medium proteins such as β-lactamase or interferonα1, which have been secreted into the periplasm under artificially induced amino acid deficiency conditions.
Depending on the extent of PSII activity impairment, strains with a mutation in the spoT gene have characteristic (p)ppGpp metabolic defects, inter alia: i) an increased basal ppGpp level during normal (balanced) growth, combined with a lower growth rate, ii) a higher induced ppGpp level during the stringent response, and iii) a lower ppGpp turnover rate after the stringent response has ended.
The DNA sequence of the E. coli spoT gene (SEQ ID NO: 1) codes for the spoT protein with the sequence SEQ ID NO: 2. The spoT gene is expressed as part of an operon which comprises the five genes gmk-rpoZ-spoT-spoU-recG. Expression of the operon and therefore also spoT expression are controlled firstly by the P1 promoter located upstream of the gmk gene, and secondly by the P2 promoter which is located in the gmk gene region coding for the C-terminal moiety (Cashel et al., 1996, Escherichia coli and Salmonella Cellular and Molecular Biology, ed. Neidhardt, F. C. ASM Press, Washington, D.C. pp. 1458-1496). In strains with a spoT1 mutation (SEQ ID NO: 3), the spoT protein has an insertion of the two amino acids glutamine and aspartic acid downstream of the aspartic acid residue 84, and a histidine/tyrosine substitution in position 255 (Durfee et al., 2008, J. Bacteriol. 190, 2597-606). In addition, a number of spoT insertion or deletion mutants have also been described (Xiao et al., 1991, J. Biol. Chem. 266, 5980-90).

Process for the fermentative production of heterologous proteins by means of


A lock sensor circuit detects and indicates occurrence of a phase lock condition of an output signal of a phase lock loop (PLL) circuit when the PLL output signal is phase locked to a reference signal. A phase and frequency detector (PFD) has a reference signal input (REF IN) and a feedback signal input (VCO FBK IN) coupled to the output of the PLL circuit. The PFD delivers output UP and DOWN signals according to whether the reference signal leads or lags the feedback signal. A multi-bit up/down counter (FIG. 2 ) has UP and DOWN inputs coupled to the respective UP and DOWN outputs of the PFD and an m bit output (Q0, Q1, . . . Q10). A lock sensor circuit (50) coupled to the m bit up/down counter monitors the nth bit output (QN) of the up/down counter where n<m. A first down counter (F1, F2) counts consecutive output DOWN signals in the absence of an output UP signal. A second up counter (F3, F4) counts consecutive output UP signals in the absence of an output DOWN signal. A lock logic circuit (TLATCH, BLATCH, 64, 66, G7) is coupled through NAND gates (G1, G2) to outputs of the respective first down counter (F1, F2) and second up counter (F3,F4). The lock logic circuit generates an unlock condition indicator signal if consecutive counts of the same signal exceed a specified number and, generates a phase lock condition indicator signal at the LOCK output when no unlock condition is detected.

A conventional phase lock loop incorporates a phase and frequency detector (PFD) with a reference signal input coupled to a reference signal source such as a crystal oscillator providing a stable frequency and phase reference signal. A feedback signal input of the PFD is coupled to the output of the PLL circuit for comparison of the feedback signal with the reference signal. The phase and frequency detector generates output "up" and "down" pulse signals according to whether the reference signal or sync signal leads or lags the feedback signal. Typically, the up and down signals at the output of the PFD are used to drive an analog transistor charge pump and RC loop filter circuit. The duration or pulse width of the "up" and "down" pulse signals is proportional to the skew between the reference signal and feedback signal. The analog charge pump circuit in turn controls the switching speed of current starved inverter stages of a voltage controlled ring oscillator. Varying the switching speed in turn varies the frequency of the feedback signal from the PLL to achieve a "lock" condition between the reference signal and feedback signal.
It would be desirable to sense and detect the occurrence of a phase lock condition at the output of the phase and frequency detector PFD and provide a reliable phase lock condition signal to tell another circuit such as a microprocessor whether to start using the clock signal generated by the PLL clock driver circuit. A disadvantage of prior art lock sensor circuits such as the Motorola 88915 (TM) lock signal output circuit is that the lock signal is ambiguous. The lock indication output signal goes high when the PLL is in a steady state phase and frequency lock condition. However under certain conditions the lock output signal may remain low even though the PLL is in the phase lock condition. Therefore the lock output signal is not suitable to tell other active circuits whether to start using the clock generator signal. The lock output signal can only be used for passive monitoring or evaluation.

Lock sensor circuit and method for phase lock loop circuits


A engine control system for helicopters and the like in which free turbine speed is isochronously controlled by a more robust control system having increased bandwidth due to loop architecture and filter implementation accomplished via closed loop control. Dual notch (feedback compensation) is used with design forward path compensation, including variable lead/position of the lead, variable gain and implementation of integrator/proportional. Superior transient control is achieved by the use of a multi-loop governing design using different error compensations when on "NF" loop control and when not on "NF" loop control, and with "bumpless" inner loop reference by the combination of compensation switching and re-initialization of a common integrator. Parallel select logic with dynamic compensation is used separating the system control loops and the selection process of the controlling functions, as they pertain to the application of control loops designed form small signal stability to large transients. A torque loop limitation as a direct function of the speed loop gains provides a predetermined limit to the governed speed excursion in the event of an in-range failure of the torque signal, or an unannounced fail fix on one engine or system of the twin engine application.

In electronic engine control systems for, for example, twin gas turbine engines used in helicopters, it is desirable to control the free turbine speed isochronously at a fixed reference point, for example, at one hundred (100%) percent. Isochronously refers to there being no error between a reference signal and a feedback signal.
It is also desirable in such a system to have a more robust control system with increased bandwidth. Additionally, it is desirable to provide isochronous control, while expanding the transient bandwidth without associated integrator problems. These desirable attributes are achieved as part of the present invention.
It is also desirable to have superior control system performance in such a system. For example, normal speed governing typically involves a proportional-plus-integral (P+I) approach, when the free turbine speed (NF) loop is commanding fuel flow. When direct free turbine speed (NF) control is overridden by another control loop [e.g., acceleration (accel.), deceleration (decel.)]via the control loop selection logic, the NF error is operated upon by a proportional-plus-differential (P+D) approach. In the invention superior transient control is achieved through the use of a multi-loop speed governing design, as will be explained more fully below.
Additionally, there are a number of problems in the helicopter engine control art currently encountered in present control selection processes (e.g., overshoot/ring or early cut back). For example, a "stiff" rate limit will delay selection of the limit function, allowing an overshoot of the limiter, when used in conjunction with a well damped loop gain function to also reduce the system "ring" or "hard cut back" due to hitting the loop hard. In the invention a rate limit feature, added to the limit loop selection and control process, provides a method for control of the rate of approach to the limits built into the control system. The rate limit technique of the invention allows maximum use of available plant performance.
Thus, three aspects of the present invention meet these desires of the prior art or obviate these prior art problems particularly in the helicopter engine control art, as well as provide innovations applicable to engine control generally, whether single or multiple engine(s) systems and whether in helicopter or other applications, and even in some instances to the control of functions generally.
Additionally, it is noted that controls mounted in twin engine (e.g. gas turbine) helicopter applications generally require control of both output shaft speed and torque matching, presenting significant problems in terms of the failure mode effects of the torque input. A final aspect of the present invention provides a predetermined limit to the governed speed excursion in event of an in-range failure of the torque signal, or an unannounced fail fix of one engine or system.
With respect to this aspect of the invention, it is noted that current torque matching algorithms provide a path for significant speed excursions in the event of a failure, (e.g., incorrect bias, counter problems, "fail fix" on one engine, etc.). To limit this problem, the general, prior art practice has been to severely limit the torque matching gain, and therefore the effectiveness of the loop. The torque matching bias and the speed loop bias both contribute to the inner loop reference.
The torque matching problem arises when the error in the torque match loop cannot be closed (this problem can have many causes), and the difference causes a constant bias to the speed governing loop. In this case the speed loop is required to correct for this torque match error, causing a speed governing error proportional to the relative gains of the speed governing loop and the torque matching loop and opposite in sign. This problem is compounded in isochronous controls, as used in the exemplary electronic engine control design of the invention, because a steady state error can iterate to the extremes of the integrator limits.

Control system for gas turbine helicopter engines and the like


A high performance constant velocity universal joint is based on a body which encloses the constant velocity joint components and provides a smooth spherical outer surface. A one-piece semi-rigid plastic boot in the form of a truncated sphere has a smooth internal spherical surface sized to match the outer surface of the body. The plastic of the boot is sufficiently elastic to allow the boot to snap over the body, yet sufficiently resilient to snap the open end closed after the boot is snapped over the body to provide a substantial seal preventing entry of debris under the seal. A retaining ring is positioned on the boot near the truncated end to resist plastic creep which would result in enlarging the opening and allowing debris to enter the seal.

Constant velocity joints are used in numerous vehicular applications where the rotational velocity oscillation of a conventional cardan joint is unacceptable. For example, in the front suspension of a front wheel drive automobile, there will be two constant velocity joints per axle. They are also used in off-road heavy-duty equipment, in trucks, and in high performance recreation vehicles.
When the application is not overly environmentally adverse, constant velocity joints are excellent. However, in environmentally unfriendly applications they are less desirable because of the problems of keeping dirt and debris out of the joint. A better understanding of that will be appreciated upon review of FIG. 1 which shows a conventional prior art constant velocity joint.
FIG. 1 is, as will be apparent to those skilled in the art, a diagram of a constant velocity joint. An input shaft 21 is attached to a housing 22 which is formed to have a plurality of ball races 23. An inner race member 25 also includes a plurality of ball races, and a plurality of balls 26 connect the two members 25, 22 by means of the balls 26 riding in pairs of the associated races. A cage 28 encircles the balls 26 and keeps them in a constant velocity plane as the joint flexes. Typically the inner race 25 is splined at 30 and receives a shaft 32 splined at 31. By its very nature, the end 40 of the constant velocity joint through which all of the components are assembled is open. A primary seal is provided by boot 35. It is usually a fairly large and flexible member to accommodate the movement of the joint. Thus it is usually made of relatively flexible rubber and is fixed by a ring 36 to the outside of the housing and by another ring 37 to the outside of the shaft 32. A bellows area 38 allows for flexing of the boot 35 as the angle of the output shaft 32 changes with respect to the input shaft 21.
In the normal automotive environment a boot of this type can protect the joint for many thousands of miles of operation. However, as has become apparent to some motorists, once the boot tears, debris and moisture can enter the joint because the boot itself is the primary seal. It is not, like in conventional cardan joints, a simple secondary dust shield, but is the primary seal for keeping foreign material out of the workings of the joint mechanism. Thus, when the boot tears, it is not long thereafter, without attention, that the joint will fail.
Although it is desirable to use constant velocity joints in more environmentally demanding applications, the inability of the seal to withstand tough environmental conditions is a strong negative factor. In off the road applications, for example, rocks and debris thrown up by the tires, or over which the vehicle can skid is readily available to tear the relatively soft rubber boot. In off the road heavy machinery applications, not only is there a substantial danger of tearing the boot during a relatively short period of operation, but almost a guarantee of enough debris to cause the joint to fail shortly after it has torn.
For other recreational applications, such as four wheel drive vehicles, all-terrain vehicles, rock climbers, and the like where the universal joints are flexed to their limits because of the uneven nature of the terrain, the constant velocity joints will also be a benefit. But again, there is also the possibility of likelihood of tearing the rubber boot with almost certain failure of the joint to follow, particularly when running through sand, water, and the like.
There are also suggestions in the art to use closer fitting, less flexible covers over constant velocity joints. However, these are typically characterized as complex, often formed of multiple parts, requiring springs or the like to keep them in operation, and having relatively limited flexing capability, well short of the 40 degree flexing capability demanded in some off road applications.

High performance constant velocity universal joint


A gas/oil separation system for monitoring the performance of a compact, gas/oil separator apparatus which utilizes at least one pair of centrifugal force primary and secondary separators for separating a wellhead fluid containing crude oil, produced water, and hydrocarbon gas from a hydrocarbon production system into its gas and liquid parts employs relatively simple differential pressure instrumentation for producing real-time signals representative of the compact, gas/oil separator apparatus' liquid carryover performance.

Conventional oil and gas separators typically use large vessels that rely on natural or gravity separation to achieve the desired separation performance. The performance of the separator vessel is generally monitored by observing the downstream processes. Various sampling systems can also be used to monitor the separator's performance, and require special hardware to be installed in the outlet gas piping.
An improved apparatus for separating an oil/liquid phase from a gas phase contained in a wellhead fluid from a hydrocarbon production system is disclosed in the U.S. application Ser. No. 08/695,947 filed Aug. 13, 1996 titled COMPACT, HIGH-EFFICIENCY, GAS/LIQUID SEPARATOR METHOD AND APPARATUS, which is a continuation of U.S. application Ser. No. 08/337,359 filed Nov. 10, 1994, now abandoned, both of which are herein incorporated by reference in their entirety. Unless otherwise stated, definitions of terms in Ser. No. 08/695,947 are valid for this disclosure also. Field testing of that compact, gas/oil separation apparatus has demonstrated that it is quite efficient at separating a wellhead fluid mixture containing crude oil, produced water, and hydrocarbon gas from a hydrocarbon production system into its gas and liquid parts. The compact, gas/oil separator preferably utilizes one or more curved-arm, centrifugal force, primary separator(s) and one or more cyclone, centrifugal force, secondary separator(s), always employed in pairs. The compact, gas/oil separator apparatus can be used in multiple pairs (two or more primary and two or more secondary separators) or in a single pair configuration having only a single primary and a single secondary separator.
The trend in the hydrocarbon production industry is to utilize equipment that is more compact than the traditional separation equipment, and the above-identified compact, gas/oil separator apparatus fits these requirements. The compactness has the obvious advantages of smaller size and less weight, that translates into real cost savings when applied to an offshore platform. However, an inherent disadvantage of compact separation equipment due to its smaller size is its reduced capability to handle upsets in the system. Without a means for continuous monitoring of performance, there is the risk of a poor performing separator affecting downstream processes.

Compact gas liquid separation system with real-time performance monitoring


A base plate for magnetic disk drives is provided that includes a hooked protrusion integrated therein between adjacent spindle motor coils. The cross-over wires that span between adjacent coils are secured by the hooked protrusions, thereby optimizing coil height and reducing manufacturing steps. The hook-like protrusions of one embodiment of the present invention are preferably stamped into the base plate.

Micro drives are low profile miniature hard disk drives for portable applications, such as personal digital assistants, cell phones, music players, camera, digital recorders, etc. A micro drive includes the basic components of a conventional hard disk drive, but is much smaller. More particularly it includes rotatable media, i.e., one or more hard disks, a hub on which the one or more disks are mounted, a spindle affixed to a base plate and about which the hub and media rotate, and an electric motor to effectuate rotation of the hub and media. The electric motor typically comprises a rotor and a stator. The rotor may comprise one or more permanent magnets positioned about the hub, and the stator may comprise a plurality of fixed electromagnets disposed in a circular pattern proximate the permanent magnets on the hub or rotor. Controlling current flow through the electromagnets causes the hub and media to rotate.
The stator generally includes a circular member or yoke. Generally, a plurality of teeth extend inwardly from the circular portion of the yoke and are wrapped with wire to form individual electromagnets. However, the teeth can either extend inwardly or outwardly, depending on whether the motor is an interior or exterior stator type, as will be appreciated by one of skill in the art. Typically, an interior stator is employed in a 2.5″ and larger disk drive, wherein smaller form-factor drives employ an exterior stator because there is sufficient radial space for the windings. The yoke and teeth are normally a laminate, constructed from a number of layers of conductive material, although they may be a single core instead of a laminate. Each tooth has a distal and proximate end. The proximate end of each tooth is connected to the circular portion of the yoke and the distal end is opposite the proximate end and adjacent the permanent magnets mounted on the hub. Normally, disk drives use three phase electric motors and, therefore, the number of teeth is a multiple of the number three, e.g. 6, 9, 12 or more. Every third tooth is electrically connected or wired together to form each of the three phases. As outlined herein, every third tooth is wired together to form a single phase (i.e. a 9 slot, 12 pole ABC winding is described), but one skilled in the art will appreciate that other windings are possible depending on the number of poles and the number of slots provided. Thus, three separate wires would be used to individually control each phase of the motor. Coordinating the current flow through the independent wires coiled about the teeth of each phase controls the rotation of the rotor/hub.
Micro drives often utilize in-hub motors, meaning the rotor and stator are positioned inside the hub, or under-hub motors, meaning the stator is positioned underneath the rotor, in order to reduce the height of the motor and thereby reduce the overall height profile of the drive. Another way to reduce height is to alter the windings forming the coils on the stator teeth, which is very applicable with respect to smaller disk drives that must employ an exterior stator motor as described above. Normally, each tooth has multiple layers of windings. The totality of windings on a tooth comprises a coil. Thus, the stator height comprises the height of each tooth, plus two times the coil thickness. The height of each tooth comprises the plurality of conductive laminate layers forming each tooth, also referred to as the core of the electromagnet or a stator core. To alter the stator height, the diameter of the wire may be changed, the number of layers of windings may be changed and/or the height of the core may be changed.
The stator wires that form each coil generally enter and exit the coil at the proximate end of each tooth. This is because, as noted previously, the wires necessarily skip or cross over adjacent teeth in order to be wound in multiple phases. As should be appreciated, routing wires from the distal end of one tooth, either to the distal or proximate end of another tooth, while skipping over two intervening teeth, can cause problems. For example, the cross over wire(s) can interfere with the rotation of the hub, waste valuable space, and pose reliability and noise issues. Thus, in order to avoid interference with the hub, a stator coil will most always have an even number of winding layers in order for the wire to exit the tooth at the proximate end and not cross over any intervening coils. For example, the first layer is formed from the proximate end to the distal end of a tooth and the second layer of the winding is formed from the distal end to the proximate end. The wire will then be routed around the yoke perimeter to the next tooth that is in phase with the previously wound coil. Of course, there may be four or six or more winding layers and not just two. In this way, the wire does not interfere with the rotation of the rotor and, as noted, there will always be an even number of winding layers.
Another drawback of prior art cross-over wire interconnection schemes is that they compel the use of additional hardware. For example, some prior art stators include routing or retainer tabs interconnected to or interleaved within the layers of the stator yoke. The retainer tabs or hooks provide a location around the yoke for engaging and retaining the cross-over wires. This is an acceptable way to retain the cross-over wires in a predetermined location, however, additional complexity is added to the spindle motor which increases costs. Moreover, positioning the retaining tabs on the stator yoke, typically between adjacent teeth, promotes and reinforces the disadvantage that the start and finish of each stator coil still must be located at the proximate end of each tooth near the stator yoke thereby compelling an even number of coil winding layers.
While exiting wires at the proximate end of each stator tooth may simplify the routing of cross-over wires, it can also create other, unrelated limitations. As should be appreciated, to optimally design a disk drive the electric motor should also be optimized. This includes optimizing the characteristics of the stator. In the particular circumstances of a particular motor used in a particular design, it is not always desired simply to maximize the number of windings. For example, it may be desired to increase or decrease the number of laminate layers comprising each stator core, or to utilize an odd number of layers of windings rather than an even number, depending upon the wire diameter or gauge preferably selected for use in the drive. Changing one of these parameters can require changes to one or more of the others to maintain an optimized design. Thus, for small form factor electronic devices, the motor design must accommodate these factors while simultaneously seeking to decrease the height of the motor. Nonetheless, as noted above, in current drives it is more likely that the stator cores will have an even number of winding layers to avoid the problems created by cross-over wiring. As a result, it may not be possible both to optimize the motor and decrease its height. Trade offs may need to be made, such as sacrificing optimum performance to meet height restrictions caused by the number of layers of windings and/or the number of laminate layers, or sacrificing height to meet motor design objectives. The coil may have one less winding layer or one more winding layer than desired. If an additional winding is used, the stator height, and the overall drive height, will increase. If one less layer of windings is used, performance may be negatively affected, as might also be the case with an additional layer, and internal space may be unused and wasted.
Thus, it is a long felt need in the field of micro-drive production to provide a system that allows for an improved cross-over wire retainment strategy thereby providing the ability to decrease the height or thickness of the micro-drive while optimizing the characteristics of the motor for a particular end use application of the disk drive. The following disclosure describes an improved method of routing the cross-over wires between stator teeth that includes the addition of retaining tabs integrated into the base plate of the housing.

Support structure with integral retainer tabs for use in spindle motor wire routing


An arrangement for printing flat workpieces is provided and includes an application roller which rolls on the workpiece surface 6 to be printed and thereby applies a printed image to the workpiece surface 8, the application roller forming a printing cylinder 5 or a transfer roller 1 cooperating with a printing cylinder 5, a counter-roller 2 which forms in cooperation with the application roller a printing gap 3, through which the workpiece 4 passes during the printing process, and a transport device 7 for feeding the workpiece 4 into and out of the printing gap 3 formed by the application roller and the counter-roller 2. A detector device 8 is present in the direction of transport of the workpiece 4, before the printing gap 3, for detecting the position of a workpiece front end 9 or an image beginning mark, and cooperates with the transport device 7, 11 and/or the printing cylinder 5; wherein the transport of the workpiece 4, prior to reaching the printing gap 3, is accelerated or delayed and/or the angular position of the printing cylinder 5 can be changed by accelerating or delaying its rotational motion in order to bring into agreement the beginning of the printed image with the workpiece front end 9 or a determined relative position thereof.

The invention relates to a device for printing flat workpieces.
Such devices generally include an application roller which rolls on the workpiece surface to be printed, and accordingly prints a printed image on the workpiece surface directly or indirectly, according to whether the application roller itself is constructed as a printing cylinder, or cooperates as a transfer roller with such, and applies a printed image to the workpiece surface, and also a counter-roller which, in cooperation with the application roller, forms a printing gap through which the workpiece passes during printing. Furthermore, a transport device is present for feeding the workpiece into and out of the printing gap formed by the application roller and the counter-roller.
In particular, to give derived wood product panels the appearance of real wood, on cost grounds as against real wood veneer or foil coatings, printed images of decorations and wood veining are directly applied to the derived wood product panels.
For the highest quality improved appearance, a single color printing is of course not suitable. Rather, it is desired to apply the veining or decoration to the material panel in multicolor printing. This is not only the case for derived wood products; uses for other materials which can be qualitatively improved by surface printing, such as for example brick or artificial leather, can be upgraded by multicolor printing.
However, specifically for a high-value multicolor printing, it is indispensable that the printed image is positioned on the workpiece within very close tolerances, typically in the region of ±0.1 mm. Only thus can an optical quality comparable with conventional foil coatings be attained.
These requirements are of course more difficult to maintain with the present flat workpieces than in machines for paper or foils, because the flat workpieces here do not run endlessly through the printing machine, and printing according to the pattern of paper sheet printing machines is in all cases not possible, because of the inflexibility of the flat workpieces concerned.

Arrangement for printing flat workpieces


This invention relates to polynucleotide sequences encoding a gene that can confer resistance to the plant pathogen Colletotrichum, which causes anthracnose stalk rot, leaf blight and top dieback in corn and other cereals. It further relates to plants and seeds of plants carrying chimeric genes comprising said polynucleotide sequences, which enhance or confer resistance to the plant pathogen Colletotrichum, and processes of making said plants and seeds. The invention further presents sequences that can be used as molecular markers that in turn can be used to identify the region of interest in corn lines resulting from new crosses and to quickly and efficiently introgress the gene from corn lines carrying said gene into other corn lines that do not carry said gene, in order to make them resistant to Colletotrichum and resistant to stalk rot.

Colletotrichum graminicola (Ces.) (Cg), more commonly known as anthracnose, is the causative agent of anthracnose leaf blight, anthracnose stalk rot (ASR) and top dieback that affects Zea mays (L.), also known as maize or corn. It is the only known common stalk rot that also causes a leaf blight (Bergstrom, et al., (1999), Plant Disease, 83:596-608, White, D. G. (1998), Compendium of Corn Diseases, pp. 1-78). It has been known to occur in the United States since 1855 and has been reported in the Americas, Europe, Africa, Asia, and Australia (McGee, D. C. (1988), Maize Diseases: A Reference Source for Seed Technologists, APS Press, St. Paul, Minn.; White, (1998) supra; White, et al., (1979) Proc. Annu. Corn Sorghum Res Conf (34th), 1-15). In the United States alone, over 37.5 million acres are infested annually with average yield losses of 6.6% nationwide (See FIG. 1). The yield losses are due both to low kernel weight in infected plants and "lodging," that is, the falling over of the plants due to weakness in the stalks caused by the infection (Dodd, J., (1980), Plant Disease, 64:533-537). Lodged plants are more difficult to harvest and are susceptible to other diseases. After infection, typically the upper portion of the stalk dies first while the lower stalk is still green. Externally, infection can be recognized by blotchy black patches on the outer rind of the stalk, while internally the pith tissue is discolored or black in appearance. Inoculation occurs in a number of ways. Roots may grow through stalk debris and become infected. This will become an increasing problem as "no till" methods of agriculture are more widely adopted due to their environmental benefits. The fungus may also infect the stalks through insect damage and other wounds (White (1998) supra). Stalk infection may be preceded by leaf infection causing leaf blight and providing inoculum for stalk infection. There is controversy in the technical literature as to the number of different varieties or races of Cg present in nature. The pathogen is transmitted by wind or contaminated seed lots. Spores remain viable for up to 2 years (McGee (1988) supra; Nicholson, et al., (1980), Phytopathology, 70:255-261; Warren, H. L. (1977), Phytopathology, 67:160-162; Warren, et al., (1975), Phytopathology, 65:620-623).
Farmers may combat infection by corn fungal diseases such as anthracnose through the use of fungicides, but these have environmental side effects, and require monitoring of fields and diagnostic techniques to determine which fungus is causing the infection so that the correct fungicide can be used. Particularly with large field crops such as corn, this is difficult. The use of corn lines that carry genetic or transgenic sources of resistance is more practical if the genes responsible for resistance can be incorporated into elite, high yielding germplasm without reducing yield. Genetic sources of resistance to Cg have been described. There have been several maize lines identified that carry some level of resistance to Cg (White, et al. (1979) supra). These included A556, MP305, H21, SP288, CI88A, and FR16. A reciprocal translocation testcross analysis using A556 indicated that genes controlling resistance to ASR lie on the long arms of chromosomes 1, 4, and 8 as well as both arms of chromosome 6 (Carson, M. L. (1981), Sources of inheritance of resistance to anthracnose stalk rot of corn. Ph.D. Thesis, University of Illinois, Urbana-Champaign). Introgression of resistance derived from such lines is complex. Another inbred, LB31, was reported to carry a single dominant gene controlling resistance to ASR but appears to be unstable, especially in the presence of European corn borer infestation (Badu-Apraku et al., (1987) Phytopathology 77: 957-959). The line MP305 was found to carry two dominant genes for resistance, one with a major effect and one with a minor effect (Carson (1981) supra). MP305 has been made available by the University of Mississippi through the National Plant Germplasm System (GRIN ID NSL 250298) operated by the United States Department of Agriculture. See Compilation of North American Maize Breeding Germplasm, J. T. Gerdes et al., Crop Science Society of America, 1993. Seed of MP305 can be obtained through W. Paul Williams, Supervisory Research Geneticist USDA-ARS, Corn Host Plant Resistance Research Unit, Box 9555, 340 Dorman Hall, Mississippi State, MS 39762.
It has been reported that there are two genes linked on the long arm of chromosome 4 that confer resistance to Cg (Toman, et al., (1993), Phytopathology, 83:981-986; Cowen, N et al. (1991) Maize Genetics Conference Abstracts 33). A significant resistance quantitative trait locus (QTL) on chromosome 4 has also been reported (Jung, et al., (1994), Theoretical and Applied Genetics, 89:413-418). Jung et al. (supra) reported that UMC15 could be used to select for the QTL on chromosome 4 in MP305, and suggested that the QTL is on a 12 cM region of chromosome 4 between UMC15 and UMC66. In fact, as discussed in more detail below, the region between UMC15 and UMC66 as reported on the IBM2 neighbors 4 genetic map is approximately 129 cM, and selection for the QTL in the manner suggested by Jung et al. (1994, supra) would at best select a large chromosomal interval with considerable linkage drag and negative phenotypic effect, and at worst, a double recombination could occur between the two markers resulting in a false positive selection for the Rcg1 locus.
Much work has been done on the mechanisms of disease resistance in plants in general. Some mechanisms of resistance are non-pathogen specific in nature, or so-called "non-host resistance." These may be based on cell wall structure or similar protective mechanisms. However, while plants lack an immune system with circulating antibodies and the other attributes of a mammalian immune system, they do have other mechanisms to specifically protect against pathogens. The most important and best studied of these are the plant disease resistance genes, or "R" genes. One of very many reviews of this resistance mechanism and the R genes can be found in Bekhadir et al., (2004), Current Opinion in Plant Biology 7:391-399. There are 5 recognized classes of R genes: intracellular proteins with a nucleotide-binding site (NBS) and a leucine-rich repeat (LRR); transmembrane proteins with an extracellular LRR domain (TM-LRR); transmembrane and extracellular LRR with a cytoplasmic kinase domain (TM-CK-LRR); membrane signal anchored protein with a coiled-coil cytoplasmic domain (MSAP-CC); and membrane associated kinases with an N-terminal myristylation site (MAK-N) (See, for example: Cohn, et al., (2001), Immunology, 13:55-62; Dangl, et al. (2001), Nature, 411:826-833).
The resistance gene of the embodiments of the present invention encodes a novel R gene related to the NBS-LRR type. While multiple NBS-LRR genes have been described, they differ widely in their response to different pathogens and exact action. To Applicants' knowledge, the novel R gene described in this disclosure is the only one demonstrated to provide resistance to Cg.

This patent is subject to a terminal disclaimer.

Method for identifying maize plants with RCG1 gene conferring resistance to colletotrichum infection


The present specification relates to a carbazole derivative, a coating composition comprising the carbazole derivative, an organic light emitting device using the same, and a method for manufacturing the same.

An organic light emission phenomenon is one of examples converting current to visible light by an internal process of specific organic molecules. A principle of an organic light emission phenomenon is as follows. When an organic material layer is placed between an anode and a cathode and a current is applied between the two electrodes, electrons and holes are injected to the organic material layer from the cathode and the anode, respectively, and the holes and the electrons injected to the organic material layer recombine to form excitons, and light emits when these excitons fall back to the ground state. An organic light emitting device using such a principle is generally formed with a cathode, an anode, and an organic material layer placed therebetween, for example, an organic material layer including a hole injection layer, a hole transfer layer, a light emitting layer and an electron transfer layer.
Materials used in an organic light emitting device are mostly pure organic materials or complex compounds in which organic materials and metals form complexes, and may be divided into hole injection materials, hole transfer materials, light emitting materials, electron transfer materials, electron injection materials and the like. Herein, as the hole injection material or the hole transfer material, organic materials having a p-type property, that is, organic materials readily oxidized and having an electrochemically stable state when oxidized, are generally used. Meanwhile, as the electron injection material or the electron transfer material, organic materials having an n-type property, that is, organic materials readily reduced and having an electrochemically stable state when reduced, are generally used. As the light emitting layer material, materials having both a p-type property and an n-type property, that is, materials having a stable form in both oxidized and reduced states, are preferred, and materials having high light emission efficiency converting, when excitons are formed, the excitons to light are preferred.
In addition to the properties described above, it is preferable that materials used in an organic light emitting device additionally have properties as follows.
First, materials used in an organic light emitting device preferably have excellent thermal stability. This is due to joule heating produced by charge transfer in the organic light emitting device. NPB normally used as a hole transfer layer material currently has a glass transition temperature of 100° C. or lower, and is difficult to be used in organic light emitting devices requiring high current.
Second, in order to obtain a highly efficient organic light emitting device capable of low voltage driving, holes or electrons injected into the organic light emitting device need to be smoothly transferred to a light emitting layer, and at the same time, the injected holes and electrons need to be kept from escaping out of the light emitting layer. For this, materials used in the organic light emitting device need to have a proper band gap and a HOMO or LUMO energy level. PEDOT:PSS currently used as a hole transfer material in an organic light emitting device manufactured using a solution coating method has a lower LUMO energy level compared to a LUMO energy level of organic materials used as a light emitting layer material, and therefore, has a problem in manufacturing an organic light emitting device with high efficiency and long lifespan.
In addition thereto, materials used in an organic light emitting device need to have excellent chemical stability, degree of charge transfer, and interface properties with electrodes or adjacent layers. In other words, materials used in an organic light emitting device need to undergo less material deformation caused by moisture or oxygen. In addition, by having proper hole or electron mobility, the materials need to maximize exciton formation through balancing hole and electron density in a light emitting layer of the organic light emitting device. For device stability, the materials need to improve an interface with electrodes including metals or metal oxides.
Accordingly, development of organic materials satisfying such requirements has been required in the art.

Carbazole derivative and organic light emitting device using same