A retinal prosthesis including an electronic stimulation unit housed inside an eye and including: a plurality of electrodes that contact a portion of a retina of the eye; an electronic control circuit, which is electrically connected to the electrodes and supplies to the electrodes electrical stimulation signals designed to stimulate the portion of retina; and a local antenna connected to the electronic control circuit. The retinal prosthesis further includes an electromagnetic expansion housed inside the eye and formed by a first expansion antenna and a second expansion antenna electrically connected together, the first expansion antenna being magnetically or electromagnetically coupled to an external antenna, the second expansion antenna being magnetically or electromagnetically couple to the local antenna, the electromagnetic expansion moreover receiving an electromagnetic supply signal transmitted by the external antenna and generating a corresponding replica signal.

1. Technical Field
The present disclosure relates to a retinal prosthesis.
2. Discussion of the Related Art
As is known, there are now available numerous retinal prostheses, which are electronic systems with medical purposes, designed for persons having visual impairments.
In general, a retinal prosthesis performs the function of making up, at least in part, for a reduced functionality of the retina, caused by a pathological condition of the retina itself, such as, for example, retinitis pigmentosa.
In greater detail, retinal prostheses are divided into retinal prostheses of an epiretinal type and retinal prostheses of a subretinal type. In use, prostheses of an epiretinal type are arranged, at least in part, on the surface of the retina that is exposed to the light, hence on the surface of the retina facing the crystalline lens. Instead, prostheses of a subretinal type are arranged, at least in part, between the retina and the so-called retinal pigment epithelium, which is the layer of pigment cells that is located on the outside of the retina itself.
This said, irrespective of the type, retinal prostheses each comprise a respective internal unit and a respective external unit. In use, the external unit is set externally with respect to the eye, whereas the internal unit is set inside the eye, and in particular within the vitreous body.
By way of example, FIG. 1a shows a retinal prosthesis 1, the external and internal units of which are designated, respectively, by 2 and 4.
The external unit 2 comprises a transmitter 6 and a first antenna 8, which is electrically connected to the transmitter 6 and is formed, for example, by a coil made of conductive material.
The internal unit 4 comprises a second antenna 10, which is also formed, for example, by a coil of conductive material. Furthermore, the internal unit 4 comprises an integrated electronic device 12 and an electrical connection cable 14, which connects the second antenna 10 to the integrated electronic device 12; for example, the electrical connection cable 14 can be a flexible electrical bus.
The integrated electronic device 12 functions as artificial retina and comprises a plurality of photodetectors 18 (FIG. 1b), an electronic circuitry 19 (FIG. 2), and a plurality of electrodes 20 (FIG. 1b).
As shown in greater detail in FIG. 1b, the integrated electronic device 12 has substantially the shape of a parallelepiped and has a bottom surface 12a and a top surface 12b. The photodetectors 18 face the top surface 12b so as that they can be reached by the light coming from the outside world, whilst the electrodes 20 extend underneath the bottom surface 12a. In turn, the bottom surface 12a is constrained, for example by means of an appropriate adhesive layer (not shown), to the electrical connection cable 14, which, in practice, carries the integrated electronic device 12.
As shown in greater detail in FIG. 2, the electronic circuitry 19 is electrically connected to the photodetectors 18 and to the electrodes 20. Furthermore, the electrical connection cable 14 comprises at least a first conductive element 14a and a second conductive element 14b, and an insulating sheath 14c, which envelops the first and second conductive elements 14a, 14b and carries the integrated electronic device 12. The first and second conductive elements 14a, 14b are electrically connected to the electronic circuitry 19, for example by means of a first via 21a and a second via 21b, respectively. In addition, the first and second conductive elements 14a, 14b are electrically connected, respectively, to a first terminal and a second terminal of the second antenna 10. Furthermore, the first and second conductive elements 14a, 14b, the insulating sheath 14c, and the electrodes 20 are such that the electrodes 20 themselves traverse the electrical connection cable 14 without contacting electrically the first and second conductive elements 14a, 14b, but rather contacting just the insulating sheath 14c. Moreover, the electrodes 20 extend through the adhesive layer set between the bottom surface 12a and the insulating sheath 14c, if this is present.
As previously mentioned, and as shown in FIG. 1a, in use the external unit 2 is set in the proximity of the eye, inside which the internal unit 4 is located. For example, the external unit 2 can be mounted on a pair of glasses, in such a way that the first antenna 8 is arranged within a lens of the pair of glasses, and in particular is arranged along the edge of said lens so as to enable the light to penetrate the eye. The transmitter 6 can be carried by an arm of the glasses.
The internal unit 4 is set inside the eye in such a way that the second antenna 10 is arranged in the proximity of the crystalline lens, possibly surrounding part of the crystalline lens itself.
The integrated electronic device 12 is arranged in the proximity of the retina of the eye, and in particular is arranged in such a way that the electrodes 20 will contact a portion of retina traversed by the optical axis of the crystalline lens, opposite to the pupil and including the so-called macula. Finally, the electrical connection cable 14 is arranged so as to pass along the inner wall of the bulb of the eye, without crossing the optical axis of the crystalline lens.
In greater detail, the second antenna 10 is arranged so as not to obstruct the path of the light rays that traverse the crystalline lens, and hence so as to enable the light that penetrates through the crystalline lens to reach the retina. Consequently, the second antenna 10 is arranged so as to surround the optical axis of the crystalline lens. In practice, in the case where the second antenna 10 is formed by a coil of conductive material, the axis of said coil coincides, to a first approximation, with the optical axis of the crystalline lens, which, amongst other things, intercepts the integrated electronic device 12.
In this way, the light coming from the outside world traverses the crystalline lens without undergoing significant alterations on account of the presence of the second antenna 10, and impinges on the integrated electronic device 12, and in particular on the photodetectors 18, which generate corresponding electrical signals, which in turn are supplied to the electronic circuitry 19. On the basis of the electrical signals supplied by the photodetectors 18, the electronic circuitry 19 generates, on the electrodes 20, corresponding electrical stimulation signals, which stimulate electrically the portion of retina in contact with the electrodes 20. In particular, the electrodes 20 stimulate the so-called internal retina (designated by 22 in FIG. 1b), which is formed, amongst other things, by the ganglion cells, the axons of which form the optical nerve. In this way, the retinal prosthesis 1 makes up, at least in part, for a possible reduced functionality of the so-called photoreceptor cells (designated by 24 in FIG. 1b), which include the cones and rods. In fact, since the ganglion cells are arranged between the photoreceptor cells 24 and the electrodes 20, the electrical stimulation signals do not traverse the photoreceptor cells 24, but rather directly stimulate the optical nerve.
In order to supply the integrated electronic device 12, the transmitter 6 generates a supply signal of an electromagnetic type, which is irradiated by the first antenna 8 and received by the second antenna 10 in such a way that, after prior propagation along the electrical connection cable 14, the supply signal reaches the integrated electronic device 12, supplying thereto the power necessary for its operation.
In greater detail, according to the frequency of the supply signal and to the distance between the first and second antennas 8, 10 a coupling of a magnetic or electromagnetic type is formed in such a way that a transfer of electric power occurs from the first antenna 8 to the second antenna 10. The electric power present on the second antenna 10 is then transferred to the integrated electronic device 12. In greater detail, in the particular case of magnetic coupling, the first and second antennas 8, 10 function as primary and secondary of a transformer.
Retinal prostheses similar to the retinal prosthesis 1, hence of an epiretinal type, are described, by way of example, in "Novel Retinal Prosthesis System with Three Dimensionally Stacked LSI Chip", European Solid-State Device Research Conference, 2006, by Watanabe T. et al., or else in U.S. Pat. No. 6,976,998.
In general, retinal prostheses are known in which, instead of the integrated electronic device 12, a so-called system in package (SiP) is present, or else a stack of integrated circuits, also known as three-dimensional integrated circuit (3D IC). In addition, the electrodes 20 can form an array of electrodes structurally separate from the integrated electronic device 12.
Furthermore, retinal prostheses are known of the type described in US20060282128, where the external unit comprises a system for acquisition and processing of images, which are transmitted to the internal unit by means of coupling between the first antenna and the second antenna. In this case, the integrated electronic device may not comprise any photodetector.
Likewise known are retinal prostheses, and more precisely subretinal prostheses, of the type described in U.S. Pat. No. 7,483,750, where the electrodes are arranged between the internal retina and the external retina.
Once again with reference to the retinal prosthesis 1, this makes possible, after implant, to make up at least in part for a reduced functionality of the photoreceptor cells 24. However, in the case where the integrated electronic device 12 undergoes damage and has to be replaced, it becomes necessary to extract from the eye the entire internal unit 4.


A method and apparatus are disclosed for characterizing multipath-induced distortions in the autocorrelation function of a correlation receiver in order to reduce effects of these multipath-induced distortions on the accuracy of detecting the time of arrival of a received signal. The magnitude of the multipath-induced errors adversely affecting the shape of the autocorrelation function is estimated in real time, for example, through the use of secondary scanning correlators whose time base is independent of a typical receiver's detection-oriented correlators. This error is subtracted from the detection-oriented correlator's timing, thereby yielding a more accurate autocorrelation function.

The Global Positioning System (GPS) as now being implemented utilizes a number of satellites in precise orbits that broadcast navigational information that may be used by anyone with a proper GPS satellite receiver. This so-called navigational information is also useful to surveyors and the like because it can provide accurate position information concerning any point on the globe. Each satellite in the GPS system broadcasts with the same carrier frequency and each broadcast signal includes an individual code that serves to identify the particular satellite. The codes are generally long and are made up of a pattern of 1's and 0's that repeats over long time periods relative to the data rate. A complete study and report on GPS signals has been published by J. J. Spilker, Jr., "GPS Signal Structure and Performance Characteristics", Navigation, 1980. In addition, the basic methods and techniques of GPS are also represented by J. J. Spilker, Jr. in his book "Digital Communications by Satellite", Prentice Hall, Inc. 1977.
Radio receivers for the GPS navigation data bit stream are commonly referred to as correlation receivers and examples of such receivers are described in U.S. Pat. No. 4,754,465 ("465 patent") to Charles Trimble, and assigned to the assignee of the present application. The disclosure of the '465 patent is incorporated herein by reference. Correlation receivers are typically employed because they are designed for situations typically encountered in satellite broadcasting where the strength of the GPS signal is quite weak compared to the noise level. The relative signal level is low at least in part because the receiver must use a wide-angle antenna, which has very low gain, due to the system constraint of having to listen to a number of satellites that might be located anywhere in the sky.
In order to boost the weak signal without also amplifying the noise, it is the practice to use spread spectrum modulation in GPS satellite systems. The spread spectrum technique modulates the satellite transmission by the individual satellite identification code, and this has the effect of spreading the satellite signal over a bandwidth that is determined by the reciprocal of the pulse width. Conversely, the receiver multiplies the signal received on the ground by a replica of the individual satellite code, and this kind of demodulation is generally known as correlation. A typical spread spectrum receiver is described in U.S. Pat. No. 4,965,759 to Uchida et al. Spread spectrum systems in general are more fully described by R. C. Dixon, "Spread Spectrum Systems", J. Wiley & Sons, Inc., 1976.
A particular advantage of using spread spectrum modulation is that it allows the time of arrival of the transmitted signal to be determined by the receiver. This time-of-arrival measurement is the cornerstone of the satellite-based GPS system and is used to calculate the "pseudo-range", which is the first-cut estimate of the distance between the receiver and a GPS satellite.
Determining the time of arrival of a signal requires the recognition of at least a portion of the incoming signal and a comparison of the time of arrival of the incoming signal with the time that it was known to have been transmitted. This measurement is made by aligning the incoming code and the local code using a code tracking loop. Such code tracking loop adjusts the delay time of one code with respect to the other and tries to minimize the time difference between the incoming and local codes. The time-of-arrival measurement is then based on the current local code delay time as measured by the local clock relative to the known time when the incoming signal was transmitted. When this delay time is multiplied by the signal propagation speed, which is assumed to be the speed of light, the path length is derived. As mentioned above, this path length is referred to as the pseudo-range because the local clock is not in perfect synchronism with the transmitter clock. Further, any errors affecting the code tracking loop will also directly affect the time-of-arrival measurement.
As mentioned above, a correlation receiver typically demodulates the spread spectrum signal by multiplying the incoming signal with a locally generated replica of the spread spectrum code. The operation of multiplying the local code and the incoming signal to produce measurable signal power at the receiver requires that the local code and the incoming code be aligned with each other to be within one cycle of the code clocking rate. This one cycle at the clocking rate is also referred to as a "chip". If the two codes are within one chip of each other, some measurable signal power will be observed at the output of the receiver correlator, and the closer the two codes are aligned, the greater is the power that will be observed. The relationship of the delay time between the two codes to the amount of signal power observed at the output of the correlation operation is called the autocorrelation function (AF). It will be appreciated that peak received power will be detected when the two codes are perfectly aligned in time. The autocorrelation function is generally observed as a series of correlation spikes output from a matched filter in the correlator circuit. This type of receiver is commonly known as a "matched filter" receiver. By keeping the local code phase synchronous with the code phase of the received signal, optimum detection of the modulation is accomplished, and from this optimized detection the time-of-arrival of the signal is determined. Through subsequent calculations, the latitude, longitude, and height of the receiver can be determined.
The ideal autocorrelation function between two spread spectrum codes is shown by the spike 10 in FIG. 1. This correlation spike represents the voltage output of a correlating receiver as a function of the relative shift in time between the two correlating codes. Therefore, the maximum voltage at the output of the correlator, as shown at the peak 10a in FIG. 1 will be ideally detected when the two codes are in perfect alignment.
The true autocorrelation function in a real receiver is, however, somewhat different from the ideal and is shown by the curve 12 in FIG. 1. As will be noted, the peak of the curve 12 is not sharp, and the leading and trailing slopes from the peak are not straight. This rounding of the ideal triangular shape is caused by the use of finite-bandwidth filters in the receiver prior to correlation. This rounding has been found to be significant in determining and compensating for multipath effects, described in detail below.
The basic elements of a typical correlation receiver are shown in FIG. 2. The incoming spread spectrum signal is received by an antenna and low noise amplifier 20 and is mixed to baseband in a mixing stage 22 by a locally generated carrier signal from a local oscillator 24. This mixing is performed in mixing stage 22 so that quadrature signals (Q), as well as in-phase signals (I), are available at baseband to facilitate carrier tracking. Both the quadrature and in-phase baseband signals (Q and I) after the mixing are still binary phase-shift keyed (BPSK) modulated by the spreading code and by any lower modulation rate information that might be included in the satellite transmitted signal. For example, in the Global Positioning System the baseband signals contain a coarse/acquisition (C/A) spreading code having a clock rate of 1.023 MHz and a precision (P) spreading code having a clock rate of 10.23 MHz, as well as a lower rate 50 baud data bit stream. The local oscillator 24 output signal also drives a code clock generator 26 having an output signal fed to a local code generator 28 that generates the local code signal fed to the correlators 30. The correlators 30 then do correlation calculations between the local code and the baseband signals and the correlation output signal is fed to a microprocessor 32 for performing the necessary time-of-arrival calculations. The microprocessor 32 also controls the timing of the local oscillator 24 and code clock generator 26, as well as telling the local code generator 28 which individual code to generate.
Determination of the optimum local code delay time requires a feedback technique that either minimizes some error signal based on the difference between the local code and the incoming code or that maximizes the autocorrelation function (AF). Because the signals are so weak, it has generally been the case to elect to use a differencing technique involving the use of "early-late" gates instead of looking for a maximum of the autocorrelation function. This early-late gate approach relies upon the fact that the ideal, uncorrupted autocorrelation function is symmetrical around its peak, the peak being representative of the point in time where the codes are perfectly synchronized. Correlation receivers then have heretofore attempted to locate the peak in time of the autocorrelation function, because this point represents perfect code alignment.
Typically, measurement of the degree of correlation between the incoming code and the local code is performed at three distinct points on the autocorrelation function. These correlation points are:
the punctual (P) point, where voltage output is maximized due to perfect alignment of the two codes; PA1 the early (E) point, which represents voltage output when the two codes are approaching alignment and are about 1/2 chip out of alignment, i.e., where the local code is advanced by 1/2 chip with respect to the incoming code; and PA1 the late (L) point, which represents voltage output when the two codes are receding from alignment, i.e., where the local code is delayed by 1/2 chip with respect to the incoming code. Only one time offset is used to shift all three correlators synchronously. In other words, the time shifts of the correlators are not independent of one another.
The early-late gate method begins by measuring the autocorrelation function and establishing a sample voltage level, and in doing this the receiver samples the autocorrelation function at the beginning of the -1 bit time interval, shown as the i.sup.th sample interval in FIG. 4. The microprocessor easily can keep track of successive samples and so samples that are one chip interval apart are subtracted from one another. The later time sample, the "late" gate, is shown as the k.sup.th interval in FIG. 4 and is subtracted from the early time sample, the "early" gate, resulting in a well-known control function that can be used to drive the local code chip time delay so that the local code is synchronized with the incoming code. This early-late gate error function is shown in FIG. 5 at curve 80 relative to the autocorrelation function shown as curve 82. As the early-late gate sampling function 80 progresses in time, i.e. is shifted in time to the right on the graph of FIG. 5, the difference between the early and late voltages diminishes, and when that difference (the error function) equals zero volts the peak of the autocorrelation function is found and the local code is synchronized with the incoming code. This occurs at 21 time units in the specific example shown in FIG. 5.
Thus, the early-late gate method in effect drives the error voltage to zero, measures the offset in code phase relative to a local clock, and derives an estimate of the time-of-arrival of the incoming signal. The punctual correlator output signal serves as a check on the steering provided by the early-late gate correlators, and in an interference-free environment this signal can confirm the early-late gate derived position of the peak. Nevertheless, the punctual correlator cannot provide any steering information on which way to shift the local code. With no extraneous interfering signals, this early-late gate method works well.
The correlators 30 that are typically employed in a correlating receiver such as that of FIG. 2 are shown in more detail in FIG. 3. In the correlators of FIG. 3, the baseband signals from the mixing stage 22 are fed to respective sets of three mixers corresponding to early, punctual, and late. More specifically the in-phase baseband signals (I) on line 40 are fed to mixers 42, 44, 46 that also receive the local code from the local code generator 28. This local code is fed on line 48 to a distributor unit 50 that might consist of a shift register and that operates to sequentially distribute the local code input on line 48 to the three mixers 42, 44, 46. Similarly, the quadrature baseband signals (Q) on line 52 are fed to three mixers 54, 56, 58 corresponding to early, punctual, and late, and these three mixers 54, 56, 58, respectively also receive the local code from the distributor unit 50.
The extent of coincidence between the received signal (I and Q) and the local code in the three states as determined by the mixers 42, 44, 46 and 54, 56, 58 is accumulated over a number of cycles in six accumulators 60, 62, 64, 66, 68, 70 that are connected respectively to the above-noted mixers. Thus, early, punctual, and late data for both the in-phase and quadrature signals are fed to the microprocessor 32 where the appropriate timing calculations are performed.
The information from the early and late correlators is combined by the microprocessor 32 to generate a delay locked loop tracking signal that is used to close the code tracking loop. The information used to close the code tracking loop is taken from early and late comparisons of the local and incoming codes; therefore, the performance of this code tracking determines the accuracy of the time-of-arrival measurement, which is used to generate the pseudo-range. As described hereinabove, the time-of-arrival measurement is typically performed by comparing in time the local code, which ideally is tracking in perfect alignment with the transmitted code, with the time reference of the receiver.
As indicated below, such prior attempts to locate the peak in time of the autocorrelation function have not yielded entirely satisfactory results. Certain types of interfering signals can distort the autocorrelation function in a way that transfers errors into the tracking loop.
One troublesome kind of interfering signal is known as multipath. Multipath refers to the phenomenon in radio wave propagation wherein a receiver system is able to collect a so-called primary signal, representing the direct path of radio wave propagation between the source and the receiver, and also a plurality of secondary delayed versions of the direct signal, representing reflections of the direct signal from objects adjacent the direct path. This phenomenon is particularly acute in receiver systems with large coverage-area antennas, such as are commonly found in GPS systems. The magnitude of multipath error induced in GPS systems has been reported by J. M. Tranquilla et al., "GPS Multipath Field Observations at Land and Water Sites", Navigation Journal of the Institute of Navigation, Vol. 37, No. 4, 1990-91.
Signal reception at moving vehicles suffers from this phenomenon to an even greater extent. Multipath adversely affects FM reception, cellular mobile telephony, and other voice/data radio systems, whether or not they use spread spectrum digital modulations. In minimizing the adverse affects of multipath, the present invention is not limited to GPS systems and is completely applicable to these other applications, particularly where a spread spectrum technique is employed.
An example of a typical receiver system with multipath signals is shown in FIG. 6, in which a GPS patch antenna 90 receives not only direct path signals 92 from the satellite but also multipath reflected signals 94, 96. The multipath signals 94, 96 represent the signal from the satellite being reflected by a building 98 or some other large object 100, respectively, in the vicinity of the antenna 90.
These secondary signals 94, 96 have been found to have several important characteristics in relation to the primary signal 92. For example, the secondary signals always have a delayed time-of-arrival compared to the primary signal 92, because the secondary signals 94, 96 travel a slightly longer path than the primary signal 92. Furthermore, the respective amplitudes of the secondary signals 94, 96 are nearly always less than that of the primary signal 92, because the reflections are specular and attenuate the signal. In addition, the sense of polarization is reversed by the reflection, and the receiving antenna is not as sensitive to these cross-polarized multipath signals as to the primary signal. For correlation receivers using digital modulation, moreover, multipath code phase signals with delays greater than one chip are completely uncorrelated, and so can be ignored. Finally, the multipath signal distance, that is, the differential path length variation from the direct signal path, varies over the wavelength of both the carrier phase and the code phase. For example, in GPS the carrier phase wavelength is 19 cm (.lambda.=c/1575 MHz, where c=the speed of light), but the code phase wavelength is much longer, because the code frequency is as low as 1.023 MHz. As a result, the multipath carrier signal can either add to the desired direct signal or subtract from it.
These secondary signals have a deleterious effect on the accuracy of the correlation receiver. For example, because multipath signals are replicas of the incoming direct signal and its code, and the principal or primary autocorrelation function is generated by examining the incoming direct signal and the locally generated signal, each multipath signal generates its own secondary autocorrelation function with respect to the locally generated code. The secondary autocorrelation functions will always have lower maximum amplitudes than the correlated primary signal and will always be delayed relative to the primary signal. This phenomenon is illustrated by way of example in FIG. 7. The secondary or multipath signals generate a series of secondary autocorrelation functions 102, 104 and 106 that are smaller than and are delayed relative to the primary autocorrelation function 108. The primary and secondary autocorrelation functions add by superposition, and the resulting net autocorrelation function is shown at 110 in FIG. 7. It should be noted that because the carrier multipath signals can also subtract from the primary signal rather dramatic distortions can be induced in the direct signal autocorrelation function. Also, because of the rounded autocorrelation function curve due to the finite-bandwidth filters, the indicated peak of the composite autocorrelation function may move in time. Therefore, when using the conventional early-late gate method, such distortions translate directly into errors in the steering control or error function for the delay-locked loop and result in erroneous estimates of local code phase and therefore erroneous estimates of pseudo-range.
The adverse effects caused by these multipath signals is readily appreciated by comparing curves 80' and 82' of FIG. 8 with the respective curves 80 and 82 of FIG. 5. Such comparison will reveal that in FIG. 5 the true autocorrelation function peak occurs when the error is zero at 21 time units, whereas in FIG. 8 the peak is falsely indicated to occur by the zero crossing of the error function 80' at 22 time units. This error is introduced because the composite autocorrelation function has experienced a change in the apparent time that a predetermined and fixed amplitude level is intercepted by the early-late gate correlation method.
FIG. 9 illustrates a composite autocorrelation function with additive multipath distortions. In FIG. 9, the uncorrupted autocorrelation function has its arbitrary amplitude, early- and late-gate times identified at E and L, respectively. The delayed multipath signals cause the composite autocorrelation function 112 to have a larger voltage level at the true late-gate time point L (17 time units) as shown at 121. It takes a little time (about one time unit) for the composite autocorrelation function amplitude 112 to fall to the prescribed voltage at level L' normally associated with the true late time point L. As can be seen from FIG. 9, the corrupted autocorrelation function 112 falls to the predetermined level 0.6 at 18 time units. The timing error .DELTA.t is the time difference between L and L' which in this example is one time unit. While the magnitude of this error is small, it is still significant, and can therefore cause an error in the determination of the true time of signal arrival.
The absolute positioning accuracy of a GPS navigation solution is essentially limited by systemic errors in the satellites and other signal degrading influences such as doppler offsets, ionospheric effects and other causes. The GPS specification is such that users heretofore can get position fixes to a three-dimensional accuracy of about 100 meters. The magnitude of the multipath effect has a maximum of around 10 meters for coarse acquisition (C/A) code and 5 meters for precision (P) code. Therefore, in the navigation mode the multipath error is largely masked by other system errors. Differential and survey applications of the GPS remove satellite-based systemic effects by differencing between two co-located receivers, in which two receivers are located located at any arbitrary distance apart in order to get another set of equations relating to satellite uncertainties. In these applications the largest error source affecting the pseudo-range, after receiver and ionospheric effects have been removed, is caused by multipath signals. The objects that are the source of multipath error are dependent on the environment in which the receiver has to operate and are therefore difficult to predict. Previous attempts at substantially reducing the adverse effects of multipath by altering the antenna characteristics of a receiver have not provided a consistent solution and often are very expensive.
The differential survey method achieves a relatively high accuracy from measurements based on the carrier phase of the received signal. Because the wavelengths of the two GPS frequencies are small, 19 and 24 cm, respectively, accuracies of much less than 1 m are possible. The problem with a time-of-arrival measurement based on carrier phase is that the time-of-arrival of each carrier phase cycle is ambiguous, that is, one carrier cycle cannot be distinguished from any other carrier cycle. Previously proposed post-processing techniques typically have required that the two receivers remain stationary for about one hour, so that this carrier cycle ambiguity can be resolved. Thus, multipath-induced errors of 3 to 10 meters are a major stumbling block to decreasing measurement time.
It has also been proposed to use the pseudo-range to resolve the carrier phase ambiguity, thereby allowing the ambiguity to be resolved much faster. The ability of a receiver to utilize pseudo-range to resolve the carrier cycle ambiguity, however, is based on the quality of the pseudo-range measurement. Unfortunately, multipath signals bias the pseudo-range so that the wrong carrier cycle is often chosen when resolving the carrier cycle ambiguity. One possible technique to prevent choosing the wrong carrier cycle is to rely on the fact that the multipath may average out over a period of time. This technique, however, requires the multipath magnitude and direction to change substantially during this period of time. That is, it relies on extensive satellite or user movement.
Other proposed techniques to compensate multipath effects in receivers rely on more commonly understood channel equalization techniques, however, none of these techniques makes use of measurements from the autocorrelation function. Instead, these techniques rely on a less accurate form of channel characterization. For example, U.S. Pat. No. 4,829,543 describes a technique for correcting the multipath effects applicable to digital data demodulation in a Time-Division Multiple-Access (TDMA) data transmission system. TDMA systems operate with regularly repeated short-duration bursts of data and differ in that respect from the Global Positioning System, however, the method of the above-mentioned patent discloses a correlation receiver for this burst start-up sequence. As described in the above-mentioned patent, a known sequence of data bits is transmitted as part of a TDMA burst preamble from which the receiver derives an estimate of the correct time to start demodulation of the data. The system then re-estimates the time to start demodulation in the TDMA system at each time slot of TDMA transmission. While that approach has merit in improving demodulation by obtaining a new and better estimate of the carrier phase timing for each time slot of transmission, it does not address multipath effects on the burst preamble itself.
It can be recognized that removing multipath effects would avoid the delays inherent in the need for averaging periods and would essentially allow signal ambiguities to be resolved instantly. This would result in large time savings for surveyors. In other words, if multipath were removed surveyors would be able to obtain high-quality pseudo-range data in an instant.
Other applications can benefit by eliminating or reducing bit errors in the detection process caused by multipath.


A sun visor assembly is provided that includes a core having a peripheral edge and a foundation folded over the core and fastened to itself to secure the core substantially within the foundation. The core includes a stiffening member and a frame member having an integrally formed pin member to detachably secure the sun visor to a retainer clip. At least a portion of the foundation extends beyond the peripheral edge of the core to define an edge of the sun visor. An upholstery cover is provided over the foundation and forms an exterior surface of the sun visor. In relatively long sun visors, the integrally formed pin member advantageously provides the structural integrity needed to substantially prevent torsional deformation of the sun visor assembly as it is moved from a storage position adjacent a vehicle headliner to a use position adjacent the vehicle windshield.

1. Field of the Invention
The present invention relates generally to sun visors for vehicles and, more particularly, to sun visors having a two piece core.
2. Description of the Related Art
It is well known to provide a sun visor for the windshield of a vehicle, wherein the sun visor is mounted on a roof panel of the vehicle for movement between a storage position adjacent the vehicle headliner, a first use position adjacent the windshield, and a second use position adjacent a side-door window opening.
One known sun visor assembly includes an inner core board of pressed wood attached to a molded plastic extender blade that slides into an out of the sun visor assembly for extending the sun blocking protection of the sun visor. An outer covering of foam-backed cloth, which is adhered to a kraft-paper foundation, is folded over the core board and is edge sewn along a seam circumjacent the periphery of the core board. A separate pin member is secured to the foundation and is used to secure the sun visor in a retainer clip on the vehicle headliner, as is known in the art.
Another known sun visor assembly includes a molded plastic inner core member and an outer covering assembly folded over the plastic core in the manner previously described. The plastic core member is provided with an extender blade for extending the sun blocking protection of the sun visor and an integrally formed pin member to secure the sun visor to the vehicle headliner.
Vehicles having large windshields, such as transport trucks or the like, require relatively long sun visor assemblies. Therefore, a relatively long inner core is required to satisfactorily support the sun visor foundation. However, simply lengthening the inner core of known sun visor assemblies to accommodate the length of the foundation does not provide adequate structural integrity to resist deformation of the sun visor assembly. Unless adequately supported, a sun visor assembly is subject to deformation from the torsional forces imposed by a vehicle occupant moving the sun visor from the storage position to one of the use positions. For example, a relatively long sun visor assembly employing a clip member formed independent of the core member, undesirably deforms proximate the clip member as the sun visor is rotated from the storage position to first use position. In another example, a relatively long sun visor assembly employing a molded plastic core member undesirably deforms through the core""s cross section, due to the flexibility of the plastic. Therefore, a sun visor core is required for use with relatively long sun visor assemblies to substantially prevent the sun visor assembly from deforming during use.
In a preferred embodiment of the present invention, a sun visor assembly is provided that includes an inner core having a stiffening member and a frame member. The frame member includes an integrally formed pin member to detachably secure the sun visor to a retainer clip. A foundation is folded over the core and is fastened to itself to secure the core substantially within the foundation. At least a portion of the foundation extends beyond a peripheral edge of the core to define an edge of the sun visor. The sun visor assembly further includes an upholstery cover covering the foundation that forms an exterior surface of the sun visor.
The stiffening member may be substantially longer than the frame member and/or substantially similar in length to the foundation. This advantageously allows the frame member to be substantially shorter than overall length of the core, providing significant savings in material. The sun visor core of the present invention is advantageously suited for use in a sun visor assembly having a relatively long length, without undesirably compromising the structural integrity of the sun visor assembly.
Various additional aspects and advantages of this invention will become apparent to those skilled in the art from the following detailed description of the preferred embodiment, when read in light of the accompanying drawings.


Example systems and methods are provided for maintaining a system clock during a sleep mode of a mobile communication device. An example system may include a high frequency clock circuit configured to generate a system clock, and a low frequency clock circuit configured to generate a sleep clock, where the sleep clock has a lower frequency than the system clock. The example system may further include a sleep system configured to deactivate the system clock in response to the mobile communication device entering sleep mode. The sleep system may include a sleep counter configured to use the sleep clock to maintain a sleep count for the deactivated system clock during the sleep mode, and a calibration unit configured to activate a calibration clock at periodic intervals during the sleep mode and utilize the calibration clock to calibrate a frequency of the sleep clock.

In a wireless communication system, the battery capacity of a wireless communication device is limited as a result of device size. In order to conserve power, wireless communication devices often utilize discontinuous reception (DRX) techniques that allow the devices to power down certain circuitry when operating in an idle mode, commonly referred to as "sleep mode." During DRX sleep mode, a wireless communication device typically disables a fast system clock and utilizes a lower-frequency clock for sleep counting. When the wireless communication device exits sleep mode, the fast system clock is restored using the sleep count.
The lower-frequency sleep clock in a wireless communication device is typically a 32 kHz clock source, provided by an external crystal oscillator. A crystal oscillator advantageously provides a relatively stable clock source, but may increase the cost and size of the integrated circuit.

Systems and methods for providing a sleep clock on a wireless communications device


An optical isolator includes a first collimator, a second collimator and an isolation assembly arranged between the collimators. The first collimator includes a first capillary retaining an input fiber and a first GRIN lens received and retained in a first glass sleeve. The isolation assembly includes two birefringent crystal wedges with an optical rotator interposed between the birefringent crystal wedges. A magnetic ring receives and retains the birefringent crystal wedges and the optical rotator together. The second collimator includes a second capillary retaining an output fiber and a second GRIN lens which are received and retained in a second glass sleeve. A first stainless steel sleeve receives and retains the first glass sleeve and the isolation assembly together while a second stainless sleeve is fit over the second collimator. Apertures are defined in the first stainless steel sleeve. An end of the second stainless steel sleeve is sized to snugly fit into the end of the first stainless steel sleeve and thus properly aligns the collimators with each other. A portion of the second stainless steel sleeve underlaps the apertures of the first stainless sleeve. Welding is performed through the apertures to permanently secure the two stainless steel sleeves together.

1. Field of the Invention
The present invention relates to an optical isolator and particularly to an optical isolator capable of self-alignment.
2. Description of Related Art
Optical isolators are key elements in optic communication systems in which optical signals are produced by lasers, such as semiconductor lasers. In a projected information highway system, for example, semiconductor lasers are employed in transmitters to produce forward directed optical signals. As is well known, the optical signals produced by such lasers may carry different kinds of information, such as digital, analog, or combined format.
The semiconductor lasers are susceptible to light signal reflections and adverse effects, such as optical wavelength jitter, laser output intensity noise, and uncontrolled optical power modulations are caused.
Uncontrolled optical power modulation leads to a non-linear laser transfer function which represents the relationship between laser drive current and output optical power. This causes an infidelity representing the electrical signal (RF) with a distorted optical signal, which may cause errors in the operation of connected circuitry, such as decision making circuitry to misidentify an intended one state for a zero state. In the case of analog systems, such as multi-channel television (TV) systems, non-linearity in the laser transfer function may cause interference between channels.
Laser output intensity noise is induced with laser operation subject to undesired optical return, which degrades the TV signal to noise ratio and as a result reduces picture quality. An optical wavelength jitter transmitting in an optical fiber further produces TV signal distortion due to signal dispersion, as the optical signals propagate along the path of an optical fiber.
In the case of digital systems, information is carried by bit symbols at a rate of 2.4 billion bits per second (i.e. 2.4 Gb/s) or higher. At such rates, the bit to bit spacing becomes progressively more limited. As optical signal bits progress along the length of an optical fiber, the bits are subject to dispersion, which reduces the signal level of the bits. The reduced signal level results in an increased bit error rate. Signal dispersion causes a spreading of the signal and results in bit overlap. The overlapping of bits in turn causes a high bit error rate and reduces fidelity in the transmission of information.
Accordingly, it is desirable to block the reverse or return transmission of optical signals back to a laser transmitter while providing low attenuation at the forward direction.
Further, it is desirable to reduce undesired levels of reflected optical power in optical systems incorporated in communication systems and in the information super highway.
An optical isolator is used with fiber optic amplifiers in optical systems to prevent oscillation due to reflection and to prevent injection of spontaneous optical emissions to the laser transmitter originally producing the optical signals. Optical interference noise effects such as the spontaneous emissions can occur at a reflection level of one part per million (i.e., below 60 dB) of light. The interference noise will increase transmission noise of the fiber optic transmission system resulting in reducing signal to noise ratio and signal distortion.
An optical isolator consists of a number of elements, which typically include a first GRIN lens, a first birefringent crystal wedge, an optical rotator, a second birefringent crystal wedge and a second GRIN lens The first GRIN lens receives and converges rays emitted from an input optical fiber into parallel rays. The first birefringent crystal wedge split the parallel rays into a first ray polarized along the optical axis and a second ray polarized perpendicularly to the optical axis. The second birefringent crystal wedge recombines the first ray and the second ray The second GRIN lens focuses the recombined rays into an output optical fiber The optical rotator is mounted between the first and the second birefringent crystal wedges for rotating the first ray and the second ray 45?at the same direction. The rays reflected from the output optical fiber will be diverged by the isolator and cannot be focused into the input optical fiber.
The elements of the optical isolator must have precise relative orientation with respect to each other in order to achieve the desired performance. This complicates the assembling process of the optical isolator. Thus a variety of methods have been developed for realizing efficient assemble of the optical isolators.
U.S. Pat. No. 5,446,813 discloses a conventional optical isolator capable to prevent ray reflected from output fiber from focusing into input fiber. As shown in FIG. 1 of the attached drawings, the optical isolator shown in U.S. Pat. No. 5,446,813, designated with reference numeral 10 comprises two standard collimators 20, 40 and an isolation assembly 30. Each collimator 20, 40 comprises a capillary 22 retaining an end of an optic fiber 21, 41, a GRIN lens 23 and a glass sleeve 24, wherein the capillary 22 and the GRIN lens 23 are respectively received in the glass sleeve 24. To hold and protect the collimator 20, the glass sleeve 24 is received in a copper sleeve 25. The isolation assembly 30 includes first and second birefringent crystal wedges 31, 32, an optical rotator 33 mounted between the crystal wedges 31, 32 and a magnetic ring 34 mounted between the first and the second birefringent crystal wedges 31 and 32 The isolation assembly 30 is retained in a copper sleeve 36 for protection.
To assemble, the copper sleeves 25, 36 are all fit into a stainless steel sleeve 55 and a bonding agent, such as epoxy, is applied between the copper sleeves 25, 36 and the stainless steel sleeve 55. A bonding agent is then cured to secure the sleeves 25, 36, 55 together. Curing the bonding agent by heat may cause relative displacement between the copper sleeves 25, 36 and the steel sleeve 55 and consequently poor alignment between the elements of the optical isolator.
Furthermore, the conventional optical isolator has a complicated structure and is thus difficult to manufacture.
It is desired to provide an improved structure of optical isolator for alleviating the above problems.
Accordingly, an object of the present invention is to provide an optical isolator having a simple structure for allowing easy and efficient assembly.
Another object of the present invention is to provide an optical isolator having excellent alignment result.
In accordance with the present invention, an optical isolator is provided, comprising a first collimator, a second collimator and an isolation assembly arranged between the collimators. The first collimator comprises a first capillary retaining an input fiber and a first GRIN lens received and retained in a first glass sleeve. The isolation assembly comprises two birefringent crystal wedges with an optical rotator interposed between the birefringent crystal wedges. A magnetic ring receives and retains the birefringent crystal wedges and the optical rotator together. The collimator comprises a second capillary retaining an output fiber and a second GRIN lens which are received and retained in a second glass sleeve. A first stainless steel sleeve receives and retains the first glass sleeve and the isolation assembly together while a second stainless sleeve is fit over the second collimator. Apertures are defined in the first stainless steel sleeve. An end of the second stainless steel sleeve is sized to snugly fit into the end of the first stainless steel sleeve and thus properly aligns the collimators with each other. A portion of the second stainless steel sleeve underlaps the apertures of the first stainless sleeve. Welding is performed through the apertures to permanently secure the stainless steel sleeves together.


The invention is a board game and method of playing checkers on this specific board game comprising: a planar 10 row by 10 column checker board comprising 100 square spaces to place game pieces upon; a plurality of said 100 square spaces further including vertical riser blocks affixed to the checker board to provide projecting spaces projecting above the plane of the checker board.

The present invention relates to the field of gaming including board games and computerized play of board games including play via networks including the world wide web.
The first references to the game of checkers are found as early as 1600 B.C. in Egyptian paintings and inscriptions at the time of the Pharaohs. In England and Scotland, this game is called xe2x80x98draughtsxe2x80x99 (pronounced as xe2x80x98draftsxe2x80x99). There are many versions played worldwide.
Checkers on an 8xc3x978 board, is the checkers game played mostly in Great Britain (where it is called draughts), USA, Canada, Australia, Ireland, and a few other countries. The most popular version of checkers, however is played on a 10xc3x9710 board in Eastern Europe. In the USA, that game is sometimes called Polish Checkers. Elsewhere, it is called International Draughts. There are other 8xc3x978 varieties played in Spain and Italy and elsewhere.
Anglo-American Checkers or draughts is a game for two players. It is played on an 8xc3x978 checkered board, with a dark square in each player""s lower left corner (see FIG. 1 wherein the xe2x80x9cdarkxe2x80x9d squares are dotted and numbered).
Pieces move only on dark squares (numbered in FIG. 1). Numbers are used to record the moves, for example, if Red moves from square 9 to square 13, then it is recorded as: 9-13.
Each player controls its own army of pieces (men). The player who controls Red pieces moves first. The pieces (also known as xe2x80x98menxe2x80x99) are arranged as shown in FIG. 1.
The goal in the checkers game is either to capture all of the opponent""s pieces or to blockade them. If neither player can accomplish the above, the game is a draw.
Starting with Red, the players take turns moving one of their own pieces. A xe2x80x9cpiecexe2x80x9d means either a xe2x80x9cmanxe2x80x9d an ordinary single checker or a xe2x80x9ckingxe2x80x9d which is what a man becomes if it reaches the last rank.
A man may move one square diagonally only forward, that is, toward the opponent onto an empty square. Thus, for example in FIG. 1, the red pieces can move 12-16, 11-16 or 11-15. Similarly, the white pieces can move 24-20, 24-19 or 23-19.
Checkers rules state the captures or xe2x80x9cjumpsxe2x80x9d are mandatory. If a square diagonally in front of a man is occupied by an opponent""s piece, and if the square beyond that piece in the same direction is empty, the man may xe2x80x9cjumpxe2x80x9d over the opponent""s piece and land on the empty square. The opponent""s piece is captured and removed from the board. Thus, in FIG. 1 red can xe2x80x9cjumpxe2x80x9d 14-21, leaving square (where white man used to stand) 17 empty. Similarly, if it were white turn to move, the white man could xe2x80x9cjumpxe2x80x9d over its red counterpart 17-10, leaving square 14 empty. If in the course of single or multiple jumps the man reaches the last rank, becoming a king, the turn shifts to the opponent. No further xe2x80x98continuationxe2x80x99 jump is possible.
When a single piece reaches the last rank of the board by reason of a move, or as the completion of a xe2x80x9cjumpxe2x80x9d, it becomes a king; and that completes the move, or xe2x80x9cjumpxe2x80x9d.
A king can move in any direction and xe2x80x9cjumpxe2x80x9d in any direction one or more pieces, as the limits of the board permit. The king can only jump diagonally over one adjacent piece at a time, in any of the four diagonal directions. Multiple jumps are possible.
There are two main styles of checkers played in tournaments, Go-As-You-Please (sometimes called Freestyle or Unrestricted) and 3-Move Restriction. In Go-As-You-Please, you can make any opening moves that you want. In 3-Move, the first 3 moves (Red-White-Red) are chosen at random from a list of accepted 3-Move openings. The list contains no openings that are known losses. 3-Move is more popular in serious tournaments and matches, as it decreases the number of draws. After playing a game with one of the 3-Move openings, you play a second game with the same opening, but from the other side of the board, to even out the disadvantage of having to play a weak opening (such as the Octopus or the Skull Cracker).
There are World Championship Matches in both styles. The 3-move World Championship is the more prestigious. There are National Championship Tournaments, District Tournaments, State Tournaments, local tournaments, mail tournaments, mail ladders, International Team Matches (both over-the-board and mail), and other events.
The U.S. National Tournament is currently the strongest and most prestigious tournament in the world. Every 4th year, the winner of that tournament is the official challenger for the World Championship. Midway between these years, the British Championship Tournament determines the official challenger for the World Championship.
There are also other forms of checkers as listed below.
Italian Checkers (Dama):The board is rotated 90 degrees, so a double corner is to the left of each player. A king cannot be captured by an ordinary piece; kings can only be captured by kings. If you have a choice of jumps, you must capture the greatest number of pieces, or (if the number of captured pieces is equal) you must capture a king rather than an ordinary piece.
Spanish Checkers (Dama):The board is rotated 90 degrees, so a double corner is to the left of each player. A king cannot be captured by an ordinary piece; kings can only be captured by kings. If you have a choice of jumps, you must capture the greatest number of pieces, or (if the number of captured pieces is equal) you must capture a king rather than an ordinary piece. A king can move any distance along a diagonal, if not blocked. A king can make long jumps over a piece, any distance beyond the captured piece, if the way is clear of pieces.
International Checkers or Draughts (Polish Checkers): Played on a 10xc3x9710 board, oriented as in our English version. Each player has 20 pieces, which begin in the first four rows. Ordinary pieces move only forward, but may capture backward (in short leaps as in the English version). A king can make long jumps (or a series of such jumps) when capturing. A king can make long jumps over a piece (or a series of such jumps over pieces), any distance in front of, or beyond the captured piece, if the way is clear of pieces. An ordinary piece which jumps onto the back row, must continue jumping off the back row, if possible; and it does not become a king until it lands on the back row at the end of a move (or jump).
Canadian Checkers (Grand jeu de dames): Exactly like International Checkers, but on a 12xc3x9712 board.
Damenspiel/German Checkers/Spanish Pool Checkers: Exactly like International Checkers, but on an 8xc3x978 board. A promoted piece is called a queen (dame).
Russian Checkers (Shashki): Like Damenspiel, except that capturing is not forced. And a piece becomes a queen when it touches the king row, even if it continues to jump off the king row on that move.
Giveaway Checkers (Losing Game): Like our English version, except the object is to give away all of your pieces.
The invention is a xe2x80x9cchuckersxe2x80x9d board game and a new method of playing checkers on this specific board game comprising: a planar 10 row by 10 column checker board comprising 100 square spaces to place game pieces upon; a plurality of said 100 square spaces further including vertical riser blocks affixed to the checker board to provide projecting spaces projecting above the plane of the checker board.
Play is as in Anglo-American Checkers, but there are few different rules. Players may jump their own piece, but not in combination with jumping the opponents piece. However, the player (who can jump their own piece as well as their opponents if allowed) may wait until their next turn to jump their opponents piece which can be referred to as a xe2x80x9cdelayed double jump move.xe2x80x9d The xe2x80x9cdouble jump movexe2x80x9d is frequently referred to Anglo-checkers when a player may jump a piece moving or jumping more than one place at a time.
A game piece on a raised block may be jumped, but the piece is not taken by the opponent. To be kinged, a game piece must make it to the other side of the board, and then flip over the piece to show patterned side respectively. A king may move in any direction, and can jump and take away any game piece, including ones on raised blocks. A player may also jump pieces on raised blocks (even if they are their own). A player may only take away an opponent""s piece located on the raised blocks, if they are kinged. The winner is the player with pieces remaining on the game board.
The invention is at least an improvement over traditional checkers games because play is quickened and more complex strategies emerge than are already known in the art.


The embodiments of the present disclosure provide a touch driving unit and circuit, a display panel and a display device. The touch driving unit comprises a shift control module, a selection module, a buffer module and an amplitude control module; the shift control module includes a start signal terminal, an output terminal and at least one clock signal terminal; the selection module includes a first input terminal, a second input terminal and an output terminal; the buffer module including an input terminal and an output terminal; and the amplitude control module including an input terminal and an output terminal. The embodiments of the present disclosure are used for display manufacturing.

Since the In Cell technique (a technique that integrates a touch element to internal of a display panel) can enable a display panel to be integrated with a touch function, and make a module be thinner and have a higher transmittance, at present, more and more products begin to adopt an In Cell screen. Now, most of the manufacturers use the Hybrid In Cell technique (a hybrid touch technique), wherein in the Hybrid In Cell technique, an RX (a touch receiving electrode) layer is at an outer side of a color film substrate, a TX (a touch emitting electrode) layer is shared with a common electrode layer of a display panel.
The In Cell technique also has its own shortcomings, as to the IC (integrated circuit) alone, a touch driving circuit of the touch emitting electrode (TX) needs to be integrated into a driving IC, that is, it needs to fabricate an independent customized touch driving circuit of the TX on the driving IC. As a result, once an element in the touch driving circuit has a problem, the entire driving IC needs to be re-fabricated, which is not beneficial to cost control and hinders border slimming of the display panel.

Touch driving unit and circuit, display panel and display device


A camshaft driving arrangement for a DOHC engine comprises two pairs of sprockets, each pair being mounted to the adjacent pair of camshafts, and two idler gears rotatably mounted to the cylinder banks, respectively. A timing chain drivingly interconnects the crankshaft and the two idler gears. On the other hand, two camshaft driving chains each interconnect one of the two idler gears and the adjacent two cam sprockets. A front cover is so constructed and arranged as to conceal the two idler gears and the timing chain.

The present invention relates to a camshaft driving arrangement for a double overhead camshaft (DOHC) engine having two camshafts for opening and closing suction and exhaust valves at each of cylinder heads.
Japanese Patent Publication No. 59-74312 (=U.S. Pat. No. 4,553,473) discloses a camshaft driving arrangement of the prior art as shown in FIG. 6.
Referring to FIG. 6, there is shown a so-called V-type DOHC engine in which a cylinder block 1 includes two cylinder banks 1A and 1B angularly disposed to each other. One 1A of the two cylinder banks has a plurality of cylinders displaced along the axis of a crankshaft 2 from a plurality of cylinders of the other bank 1B. There is provided a sprocket 3 at theend of the crankshaft 2 rotatably mounted to the cylinder block 1. Rotatably mounted to each of cylinder banks 1A and 1B are an intake camshaft 6 for opening and closing intake valves 5 and an exhaust camshaft 8 for opening and closing exhaust valves 7. Two idler gears 11 are rotatably mounted to the cylinder heads 4A and 4B, respectively. Each of the idler gears 11 has a large diameter gear 11A and a small diameter gear 11B. Fixdely mounted to the end of each of the camshafts 6 and 8 is a sprocket 9. Rotation of the crankshaft 2 is transmitted to the large gears 11A of the idler gears 11 through a rubber timing belt 10, causing rotation of the small gears 11B. Subsequently, the rotation of each of the small gears 11B is transmitted to the adjacent sprockets 9 through a chain 12. As a result, the intake camshafts 6 and the exhaust camshafts 8 are rotated synchronously, opening and closing the intake and exhaust valves 5 and 7. A tensioner 13 is used for adjusting the tension of the timing belt 10, and an idler gear 14 for stabilizing or ensuring an engagement of the timing belt 10.
With such conventional camshaft driving arrangement, case covers (not shown) for the timing belt 10 and the chains 12 are mounted to an engine body in overlapping the end portions of the case covers each other, and the timing belt 10 is designed to be large in width. This increases the overall length of the engine and the dimension of the case covers, and decreases the assembling efficiency of the timing belt 10 and the chain 12. Additionally, the tensioner 13 and the idler gear 14 are mounted between the two idler gears 11 so that the middle portion defined by the timing belt 10 cannot be used for installing engine's accessories.
An object of the present invention is to provide a camshaft driving arrangement which makes much contribution to reduced overall length and increased space for installing engine's accessories.

Camshaft driving arrangement for double overhead camshaft engine


A golf ball storage and dispensing device for transport in a golfer's bag that is operated to dispense, on demand, a single golf ball into a golfer's hand and consists of a tubular housing, that can be straight or cane shaped, and is for containing a column of golf balls that are supported on a spring biased platform therein to urge that column towards an open ball dispensing end of which tubular housing. A dispenser is provided across the dispensing end of the tubular housing that involves in one embodiment, a pivoting cup and, in another embodiment, a piston that is arranged to slide in a cylindrical sleeve, the piston to pickup and dispense a single golf ball when moved across the open tubular housing end. Both embodiments provide for depressing and locking the platform to load a column of golf balls therein, whereafter the platform is released, the spring biasing urging the column of golf balls against the dispenser.

1. Field of the Invention
This invention relates to golf ball storage and dispensing devices as are maintained in a golfer's bag that operate to dispense golf balls one at a time.
2. Prior Art
Devices for storing and dispensing on demand one at a time round objects, such as golf, tennis balls, and the like, are certainly not new. A number of devices such provide for storing balls within a tube under spring biasing. One such device for dispensing balls one by one out of the top thereof is shown in a patent to Slater, U.S. Pat. No. 3,412,897. The Slater patent, like the present invention, involves a tube with a spring biased platform to urge a column of golf balls therein out of a mouth or top end. The Slater device, however, employs as a dispensing component, flexing leaves, each having an opposing shoulder that extends across the mouth or top end of a cylindrical container. The leaves are to flex outwardly when a golfer grasps a first ball in a column of balls and pulls it upwardly, against the bent shoulders, flexing the leaves apart. Distinct from the Slater patent, the present invention provides a dispensing cup, in one embodiment, that is rotated to pass a single ball out from a tubular housing end and in another embodiment provides a laterally movable head whose passage across the dispenser tube dispenses an individual ball from a tubular housing end into a golfer's hand.
Additional to the Slater patent, a number of other patents have been issued in the area of ball dispensers for both golf and tennis balls. Some examples of such are Patents to Nelson, U.S. Pat. No. 2,966,280, that shows a container that provides spring biasing of a column of tennis balls, but does not involve a dispensing mechanism like those of the present invention; and to McClure, U.S. Pat. No. 1,810,491, that shows a container with balls maintained therein under spring tensioning to urge them out of a mouth end, and includes a dispensing mechanism that consists of opposing rocker arms that pivot to pass a first ball to be pulled therefrom by a golfer operator while blocking passage of a second ball.
Additional to the above, other devices for containing a column of balls are also shown in Patents to Rodriguez, U.S. Pat. No. 4,088,251; and Pose, U.S. Pat. No. 4,253,668, that all involves gravity feed structure, with balls dispensed through a gate that is operated to guide a golf ball onto a practice tee. A salmon egg dispenser is shown in a patent Johnston, U.S. Pat. No. 2,443,861, that includes a gate arrangement that is capable of passing a single salmon egg from a column of salmon eggs, the gate moving across and picking up a single salmon egg and blocking a following egg. The arrangement of Johnston, however, requires an operator to overcome spring biasing of a piston that travels across the tube end, with release of the piston picking up a single egg from the column end. Whereas, the movable dispensing piston and sleeve of one embodiment of the present invention involves passing of an opening in the piston across the tube end, against the spring biasing, for picking up and dispensing a single golf ball.
Additional golf ball dispensing devices that involve gravity feeding of golf balls through a gate to be set on a tee or like driving support, are shown in patents to Wilson, U.S. Pat. No. 3,458,204; Hoffman, U.S. Pat. No. 4,141,558; Eberle, U.S. Pat. No. 4,391,446; Hogeberg, U.S. Pat. No. 2,696,985; and in a United Kingdom Patent No. 374,052 to Breakey. The devices of these patents are for passing a golf ball from a storage container onto a tee and involve lever type structures for selectively passing an individual ball. Further, a device known as a gleaner for gathering nuts, is shown in a U.S. Pat. No. 908,208, that includes a lever arm operated gate, this is for moving a nut into a housing and is structurally and functionally unlike the present invention.


A bird feeder having a feeder unit for retaining seed and the like is provided with a perch of polygonal cross section that is tightly retained in the wall of the feeder unit. A soft pliable surface is provided on the perch that is apparently more appealing to the birds.

Previously known bird feeders have been subject to some disadvantages, particularly with the perches that are used in connection therewith. Feeders, whether they be cylindrical or rectangular, that utilize plastic shells have found it difficult to control the diameter of the hole as it relates to the diameter of the perch. For example, if the fit between the wall of the feeder container and the perch is too loose, the perch will turn when the bird lands on it and this would lead to discouragement of birds feeding at that particular feeding port. Also, if the fit is too loose the perch will eventually work its way out and fall from the feeder. On the other hand, if the fit is too tight it is difficult for anyone to insert the perch into the wall of the feeder.
In the prior art examples of perches that are passed through the walls of the feeding container are seen in the Kilham patent, U.S. Pat. No. 3,568,641, while an example of a perch which is threaded into the container is seen in the Clarke U.S. Pat. No. 4,259,927.

Bird feeder having a non-rotatable perch with protective sleeves


An ultrasonic detector suitable for mounting in a position for surveying an area for a source of airborne ultrasound, which detector comprises a sensor for detecting an ultrasonic sound signal, characterised by a transducer configured to produce a broadband ultrasonic sound signal for reception by said sensor enabling said ultrasonic detector to perform a self-test.

1. Field of the Invention
The present invention relates to an ultrasonic detector for detecting airborne ultrasound and in certain aspects to an ultrasonic detector for detecting high-pressure gas leaks, to methods of surveying an area for a source of airborne ultrasound, to an ultrasonic sensor and to a detector comprising such a sensor.
2. Description of the Related Art
Many industrial operations, such as well drilling, oil production, oil refining and industrial gas production, utilise piping to move a wide variety of high-pressure fluids such as gas and liquids. The pipes move such fluids for operating and controlling industrial processes amongst other things. Frequently the fluids that are piped are potentially explosive and the piping requires careful monitoring for leaks; leaks are required to be identified quickly to enable the appropriate remedial action to be taken.
Other industrial operations involving pressurised gas or those incorporating rotating parts may also produce ultrasound. In the latter case the ultrasound is due to mechanical wear of bearings or stressed components for example. Existing methods to detect such ultrasound use sensors mechanically coupled to machinery to detect vibration through the housing. However, when assemblies have multiple stress points the number of sensors required to detect wear can become prohibitive for cost reasons.
At industrial plants most man-made noise occurs in the acoustic range, whereas a pressurised gas leak produces a sound signal which spans the acoustic and ultrasonic ranges. In particular, the sound signal produced by a gas leak extends into the lower part of the ultrasonic frequency range from about 20 kHz up to about 80 kHz. The amplitude of the sound signal is a function of the size of leak, pressure in the pipe upstream and downstream of leak point, density and temperature of the leaking gas, and the temperature of the environment the gas escapes into. Although the spectra produced by two gas leaks are never identical, the one characteristic shared by the spectra of different gas leaks is that the sound signal is of a broadband type with a gradually decreasing intensity with increasing frequency. This characteristic has made gas leaks susceptible of detection.
Gas leak detectors have been devised which measure the airborne sound pressure waves generated by the turbulent flow when a gas escapes from a high to low pressure. Typically these detectors are mounted at an industrial site in a fixed position (for example on a pole or on a wall about 3 m above ground level) in order to survey a particular part of the plant. Normally several such detectors are used around the plant to ensure all important pipes, etc. are monitored.
The sensing range of such detectors is typically in the ultrasonic frequency range (25 kHz to 80 kHz for example) in order to eliminate most acoustic noise which might otherwise trigger a false alarm. An alarm level is set at some point above peak background noise level in the sensing range measured at the position where the detector is to be installed. For example, current guidelines specify that the alarm level should be approximately 6 dB above the peak background noise level. This is presently done on an installation-by-installation basis by mapping the area of interest prior to installation or by monitoring the detector output after installation. An additional precaution against false alarms is to build a time delay into the alarm circuit; two requirements must then be met to trigger an alarm: the sound pressure level of the detected ultrasonic sound must be both of a certain magnitude and duration. In this way most short-term ultrasonic sound spikes (caused by whistles, metal tags, hammering etc.) are ignored by the detector.
A further requirement is that detectors must have a means of self-test to ensure functionality and achieve Safety Integrity Levels (SIL) appropriate to pressurised gas leaks. This is currently achieved in one of two ways.
An example of the first type of self-test is described in EP 1 522 839. An externally mounted piezoelectric transducer is used to generate at predetermined intervals and for a predetermined time either a single frequency or narrowband test signal within the frequency sensing range of the detector. A microphone receives the test signal and checks whether it falls within a given tolerance range. An alert signal is generated if it does not. The narrow cone of sensitivity in front of the microphone requires that both the microphone and piezoelectric transducer to be mounted on the outside of the enclosure of the detector. Since the piezoelectric transducer and microphone are exposed to the environment, there is a problem of environmentally forced drift (explained in greater detail below), which may trigger a false alarm.
If the drift is caused by a solid build up (ice, dust etc) on the microphone, the amplitude of the test signal as detected will be reduced. This will indicate that the detector has become blocked and may have reduced coverage which is helpful, provided that the build up is actually between the piezoelectric transducer and the microphone. However, if the detector is exposed to extreme environmental conditions such as heat, cold or humidity, the signal strength of the test signal at the microphone may be either increased or decreased outside of the acceptable signal parameters. In particular, the narrowband test signal produced by the piezoelectric transducer is more susceptible to environmental extremes of temperature than the broadband pressurised leak signal to be detected. As a result the test signal emitted by the piezoelectric transducer may fall outside the tolerance levels at point of detection, (causing fault alarms) even though the detector is still functioning well within the parameters of installation. Operator confidence can be lost as a result which is undesirable. If the tolerances for the acceptable signal source are relaxed to address this problem it is possible for the receiver transducer to drift outside of acceptable calibration points.
A further problem with this first type of self-test is that mounting the test transducer on the outside of the enclosure results in a blind spot since it must protrude into the sensor's cone of sensitivity.
A second type of self-test that is more recent utilises the naturally occurring background noise to monitor the detector functionality. In particular, the background noise is used to check whether or not the receiver transducer of the detector is still operational; if the background noise is not detected the detector generates a fault alarm. The problems with this are that the receiver transducer has to have a very low sensing threshold (otherwise the self-test will not work in quiet environments), and there is no alternative way for the operator to check functionality.
In summary, existing ultrasonic gas detectors have reliability problems as far as self-testing is concerned.
Another problem facing ultrasonic gas leak detectors is variation of air temperature and humidity between the source of the gas leak and the detector. These fluctuations cause variation in the amplitude of the ultrasonic sound at the detector; in particular decreasing amplitude with increasing temperature. Therefore it is important to check that such environmental changes have not caused the detector to become 'deaf' to gas leaks at the current alarm threshold. For example, it is possible that a gas leak which would have just triggered an alarm may fail to do so if the temperature has increased at the plant so that the sound pressure level at the detector now falls below the alarm threshold. To overcome this problem it is known either for the detector to monitor external temperature and humidity and adjust the alarm level accordingly, or to perform regular manual calibration of the detector. The latter option is more common as this does not require additional circuitry within the detector. However, this strategy does require that detectors are either returned to the factory or calibrated in situ. Both options are time consuming as the detectors are usually mounted several metres above ground level in order to achieve maximum sensing distances. The return-to-factory method involves replicating on-site conditions such as distance, temperature, humidity and mimicking a gas leak in order to investigate detector sensitivity at various points across the detection frequency range. The on-site method uses a handheld device that comprises a narrowband signal source. The device is placed over the sensor to seal it from the environment and which places the narrowband signal source at a known distance from the sensor. By re-tuning the narrowband source to emit at different centre frequencies the broadband frequency range of interest can be investigated. One problem with both of these methods is that site personnel are required to access the detectors, and on-site calibration can only be carried out during routine maintenance schedule.
It would be beneficial in at least some embodiments of detector to have a calibration system that can recreate factory conditions and check points across the frequency range without the need to access the detector whilst it is in use on site.
Existing forms of fixed ultrasonic gas leak detector utilise a single sensing head due either to the cost of the sensor or the circuitry involved to achieve the necessary certification for use in hazardous areas. Since there is no direct interaction between leaking gas and the detector it is currently not possible to determine which type of gas is leaking. Furthermore due to the small amount of energy and the wavelengths within the sensing range, together with the nature of sound propagation through air, it is extremely difficult to ascertain the direction of the leak source using current detector design.
The amplitude of airborne sound at a given frequency at the sensor is dependant upon inter alia leak size, pressure, gas type, gas temperature and most importantly distance between leak source and sensor. Presently installation of fixed ultrasonic detectors usually provides a large coverage area which may contain multiple leak points from different pressure processes. One of the current strategies in an alarm situation is to shut down all processes within the coverage area. This is clearly costly for the plant operator and may not be necessary if the location of the leak and the associated process could be identified more accurately.
It would be beneficial in at least some embodiments of detector to provide an indication of a gas leak direction with respect to one or more detector.
Installation of gas leak detectors gives rise to further problems. In particular the current method involves mapping the background ultrasonic noise within the detector frequency range in installation area. The alarm level of detector is then manually adjusted to suit the on-site conditions. In particular, the alarm level is often set at approximately 6 dB higher than the background. One problem with this method is that the higher the background noise level, the smaller the detection range of the detector. This is due to the fact that a pressurised gas release will generate a sound pressure level which decreases at approximately 6 dB every time the distance from the source is doubled. For example, if a leak produces 102 dB at a distance of 1 m the sound pressure level will be 96 dB at 2 m, 90 dB at 4 m etc. As alarm level is usually set at 6 dB above background then the alarm level setting for an 84 dB background would be 90 dB. This will give a detection radius of 4 m for the leak example given. If background level is 90 dB current philosophy dictates that the detection radius will be 2 m. If the detection radius is too small other forms of gas detection are usually used instead, such as infra-red point detectors for their ability to identify the gas detected.
A further problem has been identified with this methodology: the background noise environment is dynamic. Therefore maximum safety alarm levels should be reviewed regularly to ensure that the sensing distance of the detector has not deteriorated.
Most ultrasonic background noise is characterised as a spike of narrowband noise (typically 1 to 2 kHz) somewhere within the detector sensing range. Such spikes are often caused by faulty mechanical components in machinery or attenuation from audible alarm devices. Due to the nature of the mechanical noise it is possible for the spikes to change frequency as the wear continues resulting in frequency drift up or down from the initial spike frequency. Such spikes can result in false alarms using existing detector functionality. The current installation method usually deals with such spikes either by reducing the detection radius of the ultrasonic detector or by selecting another detection method.
It would therefore be beneficial to provide a detector that is able to be used in areas containing electronic transducers and/or mechanical equipment and which does not trigger false alarms from ultrasonic signals emitted from the transducers and/or equipment.
To the best of the applicant's knowledge and belief all currently available sensors for detection of airborne ultrasound are of two main types: microphone and piezoelectric; each uses a different method to measure the sound pressure level.
Microphones typically use a diaphragm which moves within an electro-magnetic field or acts as one plate of a capacitor, which in turn, creates a variance in an electrical current. The electrical current can then be processed to determine the frequency and sound pressure level present at the diaphragm.
Due to the nature of construction of microphones the electro-magnetic field or area between the capacitance plates needs to be kept free from moisture and other contaminants to maintain accurate signal tolerances. When used in harsh industrial environments (e.g. on an oil drilling rig) the sensing face must be shielded from the ingress of moisture and contaminants which means they can only be mounted facing downwards which severely restricts the sensing range. A further disadvantage arises due to the use of different materials in the diaphragm and the components of the electro-magnetic field or components in the capacitor assembly and air gap. At extreme temperatures the expansion and contraction rates of said different materials results in signal drift. Microphone construction is also extremely complicated as the tolerances of the assembly need to be carefully controlled and construction must take place in a clean environment so that moisture and contaminants are eliminated during assembly, this leads to very expensive assembly costs.
Piezoelectric sensors have the ability to produce an electrical potential which is proportional to the application of mechanical pressure. Such sensors have been used in detectors, and an example of a detector of airborne ultrasound employing a piezoelectric sensor is sold by Groveley Detection Limited under the product code GDU-01. This detector comprises a piezoelectric sensor on which is mounted a damping block, or matching material as it is often known. The purpose of the damping block is to damp the natural resonance frequency of the piezoelectric so that it has a wider frequency bandwidth of sensitivity; to that end the damping block comprises a material having an acoustic impedance which is close to or that matches that of the piezoelectric.
Detectors of airborne ultrasound are often required to remain operable over a wide range of temperatures so that they can be used in harsh environments, for example on oil rigs. The GDU-01 operates satisfactorily over a temperature range of about −25° C. to +65° C. It is still workable just outside this range, although the piezoelectric crystal begins to suffer a temperature induced drift. Oil rigs are now becoming operational in places where the temperature regularly falls below or above the aforementioned range and there is a demand in the industry for detectors of airborne ultrasound which can operate reliably over a wider range of temperatures, for example between about −55° C. and +85° C.
We have identified the problem that causes the temperature induced drift: the piezoelectric crystal and the damping block each have a different coefficient of thermal expansion. At the aforementioned extremes of temperature, the relative contraction or expansion of each part has an undesired effect on the frequency response of the detector. Due to the relatively small signal (typically mV) produced by the piezoelectric crystal, the damping block must be carefully mounted to the crystal to avoid losses and therefore reduction in output. Current mounting methods include bonding and application of a viscous membrane under clamping pressure. Therefore we have recognised that whilst a piezoelectric crystal is extremely reliable in harsh environments because it is unaffected by moisture and other contaminants, it can suffer from drift at extreme temperatures due to the expansion and contraction differentials of the piezoelectric material and damping block material.
It would be beneficial in at least some embodiments to provide a detector that offers less temperature-induced drift and/or increased linearity across its detection range over a wider range of temperature, for example between about −55° C. and +85° C. The detection range may be between about 25 kHz and 80 kHz, and between about 58 dB to 106 dB.


A second-stage hydrocracking catalyst is provided, comprising: a) a zeolite beta having an OD acidity of 20 to 400 μmol/g and an average domain size from 800 to 1500 nm2; b) a zeolite USY having an ASDI between 0.05 and 0.12; c) a catalyst support; and d) 0.1 to 10 wt % noble metal; wherein the second-stage hydrocracking catalyst provides a hydrogen consumption less than 350 SCFB across a range of synthetic conversions up to 37 wt % when used to hydrocrack hydrocarbonaceous feeds having an initial boiling point greater than 380° F. (193° C.). A second-stage hydrocracking process using the second-stage hydrocracking process is provided. A method to make the second-stage hydrocracking catalyst is also provided.

Improved second-stage hydrocracking catalysts and processes for using them and making them are needed. Earlier second-stage hydrocracking catalysts have not provided the desired improved levels of activity to make high quality middle distillates. Earlier second-stage hydrocracking catalysts have provided good yields of middle distillate and heavy naphtha, but at lower activity that what is optimal.

This patent is subject to a terminal disclaimer.

Noble metal zeolite catalyst for second-stage hydrocracking to make middle distillate


Stabilization of a wireless information collection system for dispenser usage compliance in a facility is provided. This system and method enable network stabilization by providing a network “heartbeat.” The network stabilization operates independently from information collection. Therefore, the system and method are able to improve the reliability of wireless information collection systems where the frequency of data collection is not steady.

The spread of healthcare acquired infections also known as HAI's has been an ever increasing challenge in health care facilities. HAI's include the transmission of bacteria, viruses and other disease-causing micro-organisms from various sources such as a patient or environmental surfaces to another patient or surface via the hands of healthcare workers which results in an infection of a patient that was previously not infected. These problems have been more apparent in recent years with the SARS (severe acute respiratory syndrome) outbreak and the influenza A virus H1N1 pandemic. As well, health care facilities have battled MRSA (methicillin-resistant staphylococcus aureus) and VRSA (vancomycin-resistant staphylococcus aureus) and other drug resistant micro-organisms for many years. Accordingly, there is a need to ensure that health care professionals comply with hand hygiene best practices. Hand hygiene can be accomplished using liquids such as a sanitizing product which does not require water for rinsing off or alternatively it can be accomplished using a soap and water.
As well there are other types of liquids that can be dispensed such as sun screen. The use of the sun screen may need a similar system and method of monitoring, tracking and reporting. For example such a method could be very important in schools in Australia where the incidences of skin cancer are very high.
A need exists for a system and method to reliably monitor, track and report dispenser usage. Such a system should be low-cost, stable and reliable. ZigBee is a low-power, wireless network standard that may offer reliability. However, ZigBee requires network coordinators along with routers to establish the network. This additional level of hardware adds both complexity and cost to a system. Therefore, there is the need for a reliable routing protocol for a wireless network that is tailored to improving the stability for low data flow without the use of network coordinators.

System and method for stabilizing a wireless monitoring network


An optical transmission-reception system includes: a light-emitting element having a first semiconductor multilayer structure with a ring- or disk-like shape and generating a first optical signal and a second optical signal rotating in a direction opposite to the first optical signal; a first optical waveguide optically coupled with the light-emitting element and propagating the first optical signal; a second optical waveguide optically coupled with the light-emitting element and propagating the second optical signal; and a light-receiving element having a second semiconductor multilayer structure with a ring- or disk-like shape, optically coupled with the first and second optical waveguides, and optically receiving the first and second optical signals. The first optical waveguide has a spiral shape at a coupling part with the light-receiving element, and the second optical waveguide has a spiral shape winding in the same direction as the first optical waveguide at a coupling part with the light-receiving element.

Along with recent increase in integration density of LSIs, internal circuit patterns have been scaled-down further. This scaling-down causes reduction of cross sectional area of wiring transmitting signals, thereby increasing the wiring resistance, and moreover causes the space between the adjacent wirings to reduce, thereby increasing the inter-wiring capacitance.
As a result, the wiring delay (or propagation delay) time determined by the wiring resistance and the wiring capacitance increases, which makes it difficult to further increase the operating speed of LSIs. Furthermore, along with core multiplication and the three-dimensional integration of memory in the LSIs, signal propagation in a large amount of capacities between the cores or between the core and the memory has become essential. Thus, the signal propagation speed via electricity is the bottleneck against the higher performance of the LSIs.
As a technique for solving the problem with wiring delay caused along with the increase in density of LSIs, an optical interconnection (optical wiring) technique in which an electric signal is replaced by an optical signal has been attracting attention. The optical interconnection technique is a technique for propagating signals using an optical waveguide instead of using a metal wiring, and can be expected to achieve higher-speed operation because the aforementioned increase in wiring resistance or inter-wiring capacitance due to the scaling-down does not occur.
In a semiconductor laser (LD) used as a light source in the optical interconnection technique, an element conventionally used in optical communication has a width of several micrometers and a length of hundred micrometers, which is extremely huge as compared with a transistor or a wiring pitch of an LSI. This interrupts the replacement of electric wiring with optical interconnection. In view of this, a microring laser including a microring resonator as a compact light source has recently been attracting attention.
For achieving the optical interconnection on an LSI chip, it is necessary to form an optical transmission-reception system by integrating compactly a light-receiving element (reception part) together with a light-emitting element (transmission part) as a light source, and an optical waveguide (propagation part) in addition to a driver circuit and an amplifier circuit on the same chip.
Since a microring laser provides light rays rotating in different directions in a resonator, it is difficult to stably extract and transmit (or propagate) only one light ray as an output. Even though only one light ray can be extracted stably, the loss would increase by the amount of the other light ray.

Optical transmission-reception system and light-receiving unit