An acoustic apparatus for improved bass sound reproduction comprises a resonator, a vibrator, and a vibrator drive means. The resonator has a resonance radiation unit for radiating an acoustic wave by resonance, the vibrator has a diaphragm disposed in the resonator, the vibrator drive means has a motional feedback (MFB) means for detecting the movement of the diaphragm and negatively feeding back a motional signal corresponding to the movement of the diaphragm to the input side of the circuit, and the counteraction of the resonator on the diagram is canceled upon driving of the resonator whereby the vibrator may be invalidated as viewed from the resonator, and the vibrator and the resonator can be independently designed.

1. Field of the Invention
The present invention relates to an acoustic apparatus including or using a resonator as an acoustic radiation member.
2. Prior Art
A speaker system as one type of acoustic apparatus is arranged such that a speaker unit (vibrator) is disposed in a cabinet and is driven by an amplifier (AMP). Of reproduction characteristics of the speaker system, low-frequency reproduction characteristics are mainly determined by the volume of the cabinet.
A dynamic direct radiator speaker (dynamic cone speaker) as a typical direct radiator type speaker has a substantially conical diaphragm. The diaphragm is driven by a voice coil in a magnetic gap attached near the top of the cone. When such a speaker is used in the acoustic apparatus, a direct sound is radiated from the front surface of the diaphragm, and acoustic waves are also radiated from its rear surface. The phase of the acoustic waves from the front and rear surfaces are opposite to each other. Therefore, if a difference in propagation distance of the acoustic waves from the front and rear surfaces to a listener is almost an odd multiple of a half wavelength, sound pressures from these surfaces are in phase with each other, and are superposed.
However, if the difference in propagation distance of the acoustic waves is almost an even multiple of the half wavelength, the sound pressures cancel each other and are attenuated Thus, taking into consideration the fact that sounds having various wavelengths are radiated from the speaker, it is preferable that the sound from the rear surface does not reach the listener or the sound from the rear surface does not adversely influence the direct radiation sound from the front surface.
For this purpose, the direct radiator type speaker employs a baffle. As a baffle for shielding communication of sounds from the front and rear surface of the diaphragm, a plane baffle, back-opening cabinet type baffle, closed baffle, and the like are known, as shown in FIGS. 29A to 29C. Furthermore, as a baffle having a slightly different purpose than the above baffles, a phase inversion type (bass-reflex type) baffle shown in FIGS. 31A and 31B is known. These baffles will be described below.
FIG. 29A shows a sectional view of a plane baffle. In the Figure, a hole having the same size as a vibrator is formed in a single, wide flat plate 1. The vibrator is constituted by a dynamic electroacoustic transducer (dynamic speaker) having a substantially conical diaphragm 2 and a dynamic electoro-mechanical transducer 3, and is mounted in this hole at the diaphragm 2. The dynamic transducer 3 including a voice coil, a magnetic circuit, and the like is attached to the top portion of the cone of the diaphragm 2. According to this plane baffle, since a sound from the rear surface is shielded by the plate 1, if the plate 1 is assumed to have an infinite size, a perfect baffle effect can be obtained However, a plate having an infinite size is not realistic, and in practice, a plate 1 having a finite size is used. If a lowest frequency of sound pressure reproduction characteristics is set to be about 60 Hz, the plate 1 must be a 2.times.2 (m) square, and cannot be put into a practical use.
FIG. 29B is a sectional view of a back-opening cabinet type baffle As shown in the Figure, a hole is formed in a cabinet 4 opened its rear end. A vibrator constituted by a diaphragm 2 and a dynamic transducer 3 is mounted in this hole However, according to the backopening cabinet type baffle, the speaker system must have a large size in order to obtain a necessary baffle effect. An air column in the cabinet 4 constitutes a resonance system, and impairs a transient response.
FIG. 29C is a sectional view of a closed baffle. As shown in the Figure, a hole is formed in the front surface of a closed cabinet 5, and a vibrator constituted by diaphragm 2 and a dynamic transducer 3 is mounted in this hole. With this structure, if the cabinet 5 can be perfectly prevented from being vibrated, the sound from the rear surface of the diaphragm 2 can be perfectly enclosed, thus obtaining a perfect baffle effect. However, air enclosed in cabinet 5 serves as an air spring, and gives an elasticity to the diaphragm 2. As a result, a resonance frequency as a whole undesirably becomes higher than that of the plane baffle.
This principle will be explained below with reference to FIG. 30. FIG. 30 shows a simplified electric equivalent circuit of the system shown in FIG. 29C. In FIG. 30, reference symbol R.sub.v denotes a DC resistance of a voice coil of the vibrator, and m.sub.o, S.sub.o, and S.sub.c have the following relationships:
m.sub.o : equivalent mass of vibration system PA1 S.sub.o : equivalent stiffness of vibration system PA1 S.sub.c : equivalent stiffness of cabinet
Reference symbol A denotes a force coefficient, which is given by A=Bl.sub.v where B is the magnetic flux density in a magnetic gap and l.sub.v is the length of the voice coil conductor A parallel resonance circuit Z.sub.1 by an equivalent motional impedance of the unit vibration system and an equivalent motional impedance A.sup.2 /S.sub.c of the closed cabinet are connected in parallel with each other, and the parallel circuit is connected in parallel with an amplifier (not shown) through the voice coil resistance R.sub.v as a nonmotional impedance.
As can be seen from this electric equivalent circuit, a resonance frequency f.sub.oc of a whole system is increased to be higher than a lowest resonance frequency f.sub.o of the vibrator and is given by: EQU f.sub.oc =f.sub.o (1+S.sub.c /S.sub.o).sup.1/2
An equivalent Q value (Q.sub.oc) of the whole system at the resonance frequency f.sub.oc has the following relationship with respect to a Q value (Q.sub.o) of the vibrator itself at the lowest resonance frequency f.sub.o and is increased as: EQU Q.sub.oc =Q.sub.o (1+S.sub.c /S.sub.o).sup.1/2
Therefore, in order to improve low-frequency reproduction characteristics, the equivalent stiffness S.sub.c of the cabinet must be decreased. For this purpose, a large cabinet must be employed.
A bass-reflex type speaker system has a slightly different purpose from the above-mentioned baffles. FIGS. 31A and 31B are a perspective view and a sectional view of the bass-reflex type speaker system. As shown in FIGS. 31A and 31B, a hole is formed in a cabinet 6, and a vibrator consisting of a diaphragm 2 and a dynamic transducer 3 is mounted in this hole. An opening port 8 having a sound path 7 is arranged below the vibrator. In a conventional bass-reflex type speaker system according to a standard setting, a resonance frequency f.sub.op caused by an air spring in the cabinet 6 and an air mass of the sound path 7 is set to be lower than the lowest resonance frequency f.sub.o of the vibrator (speaker) which is assembled in the bass-reflex type cabinet At a frequency higher than the resonance frequency f.sub.op caused by the air spring and the air mass, the sound pressure from the rear surface of the diaphragm 2 has inverts its phase oppositely in the sound path 7, whereby the direct radiation sound from the front surface of the diaphragm 2 and the sound from the opening port 8 are consequently in phase with each other before the cabinet 6, thus increasing the sound pressure. As a result, according to an optimally designed bass-reflex type speaker system, the frequency characteristics of an output sound pressure can be expanded below the lowest resonance frequency of the vibrator. As indicated by an alternate long and two short dashed curve in FIG. 32, a uniform reproduction range can be extended wider than those of the infinite plane baffle and the closed baffle.
However, when uniform reproduction is realized by the bass-reflex type speaker system, various limitations are posed on the resonance Q value of a unit vibration system and the like, and only when these limitations are met, the characteristics shown in FIG. 32 can be obtained In this manner, in the conventional bass-reflex type speaker system, it is very difficult to obtain an optimal design condition.
On the other hand, an attempt is made to intentionally extremely lower the resonance frequency f.sub.op of the resonator regardless of the standard design idea of the bass-reflex type speaker system and paying attention to only an acoustic radiation power from the opening port.
However, since the bass reproduction power is closely related to the volume of the cabinet, anyhow, a larger cabinet must have been employed as in the closed baffle in order to achieve low-frequency reproduction. This situation will be explained in detail below with reference to FIG. 33.
FIG. 33 shows a simplified electric equivalent circuit of the bass-reflex type speaker system shown in FIGS. 31A and 31B. In FIG. 33, reference symbols A, R.sub.v, m.sub.o, S.sub.o and S.sub.c are the same as those in FIG. 30, and m.sub.p corresponds to an equivalent mass of the sound path (port). A parallel resonance circuit Z.sub.1 by an equivalent motional impedance of the unit vibration system and a series resonance circuit Z.sub.2 by an equivalent motional impedance of a port resonance system are connected in parallel with each other, and this parallel circuit is connected in parallel with a driving amplifier (not shown) through the voice coil resistance R.sub.v as a non-motional impedance.
As can be seen from this electric equivalent circuit, the bass-reflex type speaker system includes two resonance systems according to its major characteristic feature. The impedance characteristics of this speaker system represent a double-humped curve having a total of three resonance points, i.e., two maximum peaks and one minimum peak therebetween. The resonance point of the minimum peak corresponds to the port resonance system (the above-mentioned closed baffle has only one resonance system, and its impedance characteristics exhibit a single-humped curve including only one resonance point). In the bass-reflex type speaker system, the voice coil resistance R.sub.v of the vibrator (unit) serves as both a damping resistance of the parallel resonance circuit Z.sub.1 of the vibrator side and the series resonance circuit Z.sub.2 of the opening port (duct) side. For this reason, the parallel and series resonance circuits Z.sub.1 and Z.sub.2 mutually interfere with each other.
As an example of mutual interference or mutual dependency, if a vibrator having a strong magnetic circuit is used, a resonance Q value of the vibrator is reduced, while the resonance Q value of the opening port is increased. In contrast to this, if a vibrator having a weak magnetic circuit is used, the opposite situation to the above occurs In an essential design of the bass-reflex type speaker system, an optimal point capable of obtaining uniform low-frequency reproduction characteristics must be selected under the conflicting mutual dependency condition.
Assume that the volume of the cabinet is reduced. In this case, the lowest resonance frequency f.sub.o of the unit vibration system exhibits the same tendency as that of the closed baffle, and as a result, is increased The low-frequency reproduction characteristics will finally come to be improved to some extent by the acoustic radiation effect of the opening port. However, if the size of the cabinet is reduced, it cannot be avoided that the low-frequency reproduction power will be decreased as the whole system even in the bass-reflex type speaker system.
In particular, when the resonance frequency f.sub.op of the port resonance system is intentionally decreased from standard setting, as described above, the opening port must be more elongated as the cabinet is smaller in size Therefore, the Q value becomes very small due to an increase in mechanical resistance of air in the port. An extreme decrease in resonance Q value leads to loss of the acoustic radiation power from the opening port. As a result, the function of the opening port as a resonance duct is lost, and the presence of the opening port becomes meaningless. That is, if the size of the cabinet is reduced, bass reproduction is essentially impossible.
As has been schematically described above, in the conventional acoustic apparatus, various countermeasures are taken in order to allow low-frequency reproduction.
The plane baffle, back-opening baffle, and closed baffle shown in FIGS. 29A to 29C are designed such that radiation sounds from the rear surface of the diaphragm do not reach a listener in front of the speaker system as unnecessary sounds. However, in order to improve the bass reproduction characteristics with these baffles, the apparatus (cabinet) will inevitably be made large in size, and even if it is made so to a certain feasible extent, its low-frequency reproduction characteristics will be insufficient.
In the bass-reflex type speaker system shown in FIGS. 31A and 31B, the phase of the backward sound is inverted by the opening port, so that in particular, a bass range of a direct radiation sound from the front surface of the diaphragm is compensated for. For this reason, the resonance system which is originally very hard to deal with is undesirably formed on the two portions, i.e., the diaphragm and the opening port. In order to obtain a satisfactory bass-reflex effect according to the standard setting, the optimal condition of the system must be very critically set while taking the mutual dependency condition of these two resonance systems. Although various attempts have been made in this respect as disclosed in Japanese Patent Publication No. sho 46-12670 and Japanese Utility Model Publication No. sho 54-35068, these attempts could not eliminate difficulty on design.
In order to improve the low-frequency reproduction characteristics, the cabinet undesirably becomes bulky, whether the optimal design of said speaker system has been achieved or not.
In some bass-reflex type speaker systems, the resonance frequency fop of the port resonance system is intentionally decreased from its standard setting. However, if the size of the cabinet is to be reduced, the port resonance system will hardly contribute to acoustic radiation, thus incurring a fatal drawback.
Therefore, when a bass reproduction power of a certain level or higher is to be obtained according to any of the prior arts, the resulting cabinet will inevitably become large in size. As a result, it is difficult to employ an acoustic apparatus having a cabinet of a proper volume and excellent low-frequency reproduction characteristics in a variety of applications such as in halls, rooms, vehicles, and the like.
As is so in the bass-reflex speaker system described above, in an acoustic apparatus, a resonance phenomenon is utilized in a variety of forms. FIGS. 34 to 37 show typical prior art examples in which the resonance phenomenon are utilized.
In a first prior art shown in FIG. 34, a resonator 81 is partitioned into two chambers, i.e., A and B chambers, by a partition wall 82. A dynamic electroacoustic transducer (dynamic speaker) 83 serving as a vibrator is attached to a hole of the partition wall 82. Opening ducts 84a and 84b are respectively provided to the A and B chambers, and resonance acoustic waves are radiated outwards from these ducts, as indicated by arrows in the Figure. The A and B chambers respectively have resonance frequencies f.sub.oa (Hz) and f.sub.ob (Hz) determined by the volumes of cavities (i.e. the volumes of chambers A and B), the dimensions of the opening ducts 84a and 84b, and the like. Therefore, when the speaker 83 is driven by an amplifier (not shown), a resonance phenomenon occurs by the vibration of a diaphragm, and an output energy at that time has maximum values near the above-mentioned resonance frequencies. As a result, the resonance acoustic waves having sound pressure-frequency characteristics illustrated in FIG. 35 can be obtained
In a second prior art shown in FIG. 36, a dynamic electro-acoustic transducer (speaker) 86 serving as a vibrator is attached to a resonance chamber 85' defined by a cabinet 85, and an opening 87 for externally radiating a resonance acoustic wave is formed in the chamber 85'. Another dynamic electro-acoustic transducer (speaker) 88 is separately provided to the cabinet 85, so that an acoustic wave is directly radiated outwards therefrom. In this acoustic apparatus, when the speaker 86 is driven by an amplifier (not shown), a resonance phenomenon occurs in the resonance chamber 85' due to the vibration of a diaphragm of the speaker 86. Therefore, acoustic reproduction illustrated in FIG. 37 is made from the opening 87 to have a peak sound pressure near a resonance frequency f.sub.o inherent in the resonance chamber 85'.
However, according to conventional acoustic apparatuses, the vibrator undesirably causes a decrease in resonance Q value of the resonator serving as an acoustic radiation member. This is because the speaker as the vibrator has an inherent internal impedance Z.sub.v, and the internal impedance becomes to an element which damps the resonance of the resonator. In this manner, as the resonance Q value becomes low, radiation power of the resonance acoustic wave becomes inevitably low, and the presence of the resonator in the acoustic apparatus becomes meaningless.
If the resonance frequency is decreased while rendering the resonator compact, the opening duct must be elongated. Accordingly, the acoustic resistance (mechanical resistance) of the opening duct is inevitably increased, and the resonance Q value is decreased further. For this reason, the acoustic radiation power is further decreased due to the decrease in the resonance Q value, and the acoustic apparatus is not suitable for a practical use.
As a result, neither of the conventional apparatuses shown in FIGS. 34 and 36 have sufficient acoustic radiation power. If a certain level of power is to be maintained, the cabinet becomes extremely large.


An integrated circuit having a regulator circuit capable of tracking reference voltages is provided. The integrated circuit includes shunt regulator circuitry. The shunt regulator circuitry includes a shunt regulator circuit and a voltage tracking circuit. The shunt regulator circuit has an output on which a regulated voltage is provided. The shunt regulator circuit also provides electrical current to the output when the regulated voltage is outside of a voltage range bounded by first and second reference voltages. The voltage tracking circuit may be coupled to the shunt regulator circuit. The voltage tracking circuit may generate the first and second reference voltages. In one instance, the first voltage is greater than the regulated voltage and the second voltage is less than the regulated voltage.

A regulator circuit may generally be used in an integrated circuit to regulate voltages received from an external source. The regulator circuit can be used to ensure that circuitry within the integrated circuit receives supply voltages that are within a predetermined range even when the voltages from an external voltage supply deviate from their nominal levels due to factors such as signal noise, temperature variation, etc.
Typical regulator structures include series voltage regulators and analog shunt regulators. However, other types of regulators such as push-pull shunt regulators have also been increasingly used to regulate voltages in integrated circuits. Unlike linear series voltage regulators and analog shunt regulators, push-pull shunt regulators may overcome the problem of large shunt electrical current getting drawn to ground. Generally, push-pull shunt regulators may reduce voltage headroom and may require fewer area intensive compensation capacitors such as those commonly used in analog shunt regulators.
Conventional push-pull shunt regulators often utilize reference voltages generated using bandgap reference circuits to provide reference voltages that are relatively invariant in the presence of signal noise and power supply voltage fluctuations. When the regulated voltage goes above a reference voltage, the push-pull shunt regulator may source or sink the electrical current so that the electrical current returns to its nominal level. If the average regulated voltage is outside a target range of the reference voltage, the push-pull shunt regulator will continue to sink or source the electrical current, respectively, even in the absence of supply voltage noise. This result in undesirable compensation and it may be necessary to manually program the reference voltage of the bandgap reference circuit to avoid such undesirable compensation.
It is within this context that the embodiments described herein arise.

Regulator circuitry capable of tracking reference voltages


The present invention relates to sulfated oligosaccharides or their derivatives such as the following compound and being useful for diagnosing or treating diseases such as immunity-related diseases and mucopolysaccharide diseases. ##STR1## (M is an alkali metal).

1. Field of the Invention
The present invention relates to sulfated oligosaccharides and their derivatives.
2. Description of Prior Art
Keratan sulfate, which is a kind of glycosaminoglycans (mucopolysaccharides), is contained, in a large amount, in the cornea, discus intervertebralis, cartilago and arteria of mamals.
Keratan sulfate contains, as a repeating unit, a disaccharide comprised of D-galactose and N-acetylglucosamine-6-sulfate, and is treated with, for example, endo-.beta.-galactosidase to produce sulfated oligosaccharides.
It was reported that sulfated oligosaccharides have a site which an antibody can recognize [J. Cell Biology, 101, 53 (1985)]. Further, sulfated oligosaccharides have been confirmed to be largely related to diseases such as polysaccharide disease caused by accumulation of mucopolysaccharide.

Sulfated oligosaccharides and derivatives thereof


According to one embodiment, provided is a nonaqueous electrolyte battery including a positive electrode, a negative electrode, and a nonaqueous electrolyte. The positive electrode includes lithium manganese composite oxide particles having a spinel crystal structure and lithium cobalt composite oxide particles. The negative electrode includes a titanium-containing oxide. The nonaqueous electrolyte contains a propionate ester. The battery satisfies 0.8≤p/n≤1.2 and 1≤w/s≤60. p denotes a capacity per unit area of the positive electrode. n denotes a capacity per unit area of the negative electrode. w denotes a content of the propionate ester in the nonaqueous electrolyte and is in a range of 10% by weight to 60% by weight. s denotes an average particle size of the lithium manganese composite oxide particles.

A lithium-ion secondary battery, in which charge and discharge are performed by movement of lithium ions between a positive electrode and a negative electrode, has advantages of being able to obtain high energy density and high output. Putting such advantages into use, application of lithium ion secondary batteries has been widely under progress from small-scale use such as portable electronic devices to large-scale use such as electric automobiles and electric power supply adjustment systems.
A nonaqueous electrolyte battery using a lithium titanium composite oxide, with a high lithium insertion and extraction potential of about 1.55 V based on a lithium electrode, has also been put to practical use as the negative electrode active material, in place of a carbon material. The lithium titanium composite oxide has excellent cycle performance because of its change in volume accompanying charge and discharge being little. In a negative electrode including the lithium titanium composite oxide, precipitation of lithium metal at the time of lithium insertion and extraction can be suppressed. As a result, a secondary battery provided with this negative electrode is capable of being charged with a large current.
Regarding the positive electrode active material, lithium manganese oxide (LiMn2O4) has advantages of being an abundant resource, being inexpensive, having little environmental burden and having high safety in an over-charged state. Therefore, the lithium manganese oxide is being considered as an alternative material for lithium cobaltate (LiCoO2). On the other hand, with the lithium manganese oxide, deterioration in battery performance due to gas generation, elution of manganese under high-temperature environment, and the like, has been an issue.

Nonaqueous electrolyte battery and battery pack


System and method are provided for regulating a power converter. The system includes a signal processing component configured to receive a first input signal and a second input signal, process information associated with the first input signal and the second input signal, and output a drive signal to a switch based on at least information associated with the first input signal and the second input signal. The first input signal is associated with at least a feedback signal related to an output voltage of the power converter. The second input signal is associated with at least a primary current flowing through a primary winding of the power converter. The signal processing component is further configured to change a peak value of the primary current within a first predetermined range, and change the switching frequency of the power converter within a second predetermined range.

The present invention is directed to integrated circuits. More particularly, the invention provides systems and methods for reducing electromagnetic interference (EMI) using switching frequency jittering in the quasi-resonant mode. Merely by way of example, the invention has been applied to a power conversion system. But it would be recognized that the invention has a much broader range of applicability.
Switching-mode power conversion systems often have switching frequencies in the range of tens of kHz and hundreds of kHz. Parasitic capacitance and parasitic inductance usually exist in the switching-mode power conversion systems. Electromagnetic interference (EMI) generated by a switching-mode power conversion system often needs to be reduced in order to avoid interfering significantly with other power components. To reduce the size of a switching-mode power conversion system, the switching frequency of the switching-mode power conversion system usually needs to be increased. But increasing the switching frequency often exacerbates the problems related to EMI and switching loss.
A fixed-frequency controller with frequency jittering is often used in a power conversion system to reduce the EMI. FIG. 1 is a simplified diagram showing a conventional power conversion system including a fixed-frequency pulse-width-modulation (PWM) controller. The power conversion system 100 includes a fixed-frequency PWM controller 102, a primary winding 118, a secondary winding 120, a power switch 122, a current sensing resistor 124, a feedback-and-isolation component 126, a rectifying diode 128 and a capacitor 130. The fixed-frequency PWM controller 102 includes an oscillator 104, a PWM comparator 106, a flip-flop 108, a gate driver 110, a diode 112, and two resistors 114 and 116. The fixed-frequency PWM controller 102 further includes three terminals 132, 134 and 136. For example, the power switch 122 is a field-effect transistor (FET), a bipolar junction transistor (BJT), or an insulated-gate bipolar transistor (IGBT).
The switching frequency of the power conversion system 100 is often determined by the oscillator 104. To reduce the EMI, the oscillator 104 is often used to perform switching frequency jittering in different ways, e.g., ramping up and down the switching frequency of the power conversion system 100, or using a pseudo-random generator to change the switching frequency.
FIG. 2 is a simplified conventional timing diagram for the power conversion system 100 showing frequency jittering by ramping up and down the switching frequency by the oscillator 104. The waveform 202 represents the switching frequency of the power conversion system 100 as a function of time. The waveform 204 represents turned-on and turned-off conditions of the switch 122 as a function of time. For example, if the waveform 204 is at a logic high level, the switch 122 is closed (e.g., on), and if the waveform 204 is at a logic low level, the switch 122 is open (e.g., off). Two time periods T1 and T2 are shown in FIG. 2. The time period T1 starts at time t0 and ends at time t1, and the time period T2 starts at time t1 and ends at time t2. For example, t0≤t1≤t2.
During the time period T1, the switching frequency of the power conversion system 100 ramps up from a minimum frequency 206 (at t0) to a maximum frequency 208 (at t1) as shown by the waveform 202. The frequency of the switch 122 being turned on and off increases as shown by the waveform 204. During the time period T2, the switching frequency of the power conversion system 100 ramps down from the maximum frequency 208 (at t1) to the minimum frequency 206 (at t2) as shown by the waveform 202. The frequency of the switch 122 being turned on and off decreases as shown by the waveform 204.
The magnitude of the frequency jittering is usually controlled within a certain range to avoid audio noise caused by excessive frequency jittering. For example, if the fixed-frequency system 100 has a switching frequency of about 60 kHz, the range of frequency jittering can be ±4%. That is, the switching frequency range of the system 100 is 60 kHz±2.4 kHz. The frequency range for energy distribution of an nth-order harmonic is ±2.4 nkHz. The total harmonics energy of the fixed frequency system 100 remains the same, while the harmonic energy amplitude at each harmonic frequency reduces. Thus, the conduction EMI of the system 100 is often improved.
In operation, the system 100 usually has a parasitic inductance Lk in the transformer including the primary winding 118 and the secondary winding 120. The power switch 122 often has a parasitic capacitance Cp between the terminals 138 and 140. The parasitic inductance Lk and the parasitic capacitance Cp often not only reduce the system efficiency, but increase the EMI. For example, when the power switch 122 is open (e.g., off), the parasitic inductance Lk and the parasitic capacitance Cp often cause resonation, and a high peak voltage is generated at the terminal 138. Then after the primary inductance Lp of the transformer is demagnetized, the primary inductance Lp and the parasitic capacitance Cp often cause resonation, and the voltage of the terminal 138 oscillates with decreasing amplitude. But the amplitude of the voltage of the terminal 138 is often at a high level. When the power switch 122 is closed (e.g., on) during a next period, the parasitic capacitance Cp often discharges through the power switch 122, and generates a high peak current including many harmonics, which usually exacerbates the problems related to switching loss and EMI.
Quasi-resonant (QR) techniques can be implemented to take advantage of the parasitic inductance and the parasitic capacitance existing in a power conversion system to improve system efficiency. FIG. 3 is a simplified diagram showing a conventional fly-back power conversion system including a QR controller. The power conversion system 300 includes a QR controller 302, a primary winding 304, a secondary winding 306, a power switch 308, a rectifying diode 310, a capacitor 312, an output load 316, and an auxiliary winding 330. For example, the power switch 308 is a field-effect transistor (FET), a bipolar junction transistor (BJT) or an insulated-gate bipolar transistor (IGBT).
As shown in FIG. 3, the fly-back conversion system 300 has a parasitic inductance 318 (e.g., Lk) in the transformer including the primary winding 304 and the secondary winding 306, and a primary inductance 334 (e.g., Lp) of the transformer. The power switch 308 has a parasitic capacitance 314 (e.g., Cp) between the terminals 320 and 322.
In operation, if the power switch 308 is closed (e.g., on), a current 324 (e.g., Ip) flows through the power switch 308. For example, the voltage between the terminals 320 and 324 of the power switch 308 is determined based on the following equation:Vds=Vin+N×Vout  (Equation 1)wherein Vds represents the voltage between the terminals 320 and 324 of the power switch 308, Vin represents an input voltage 326 on the primary side of the system 300, and Vout represents an output voltage 328 on the secondary side of the system 300. Additionally, N represents a turns ratio between the primary winding 304 and the secondary winding 306.
FIG. 4 is a simplified conventional timing diagram for the power conversion system 300. The waveform 402 represents the voltage between the terminals 320 and 322 of the power switch 308 (e.g., Vds) as a function of time. The waveform 404 represents turned-on and turned-off conditions of the switch 308 as a function of time. The waveform 406 represents the current 324 that flows through the power switch 308 as a function of time. For example, if the waveform 404 is at a logic high level, the switch 308 is closed (e.g., on), and if the waveform 404 is at a logic low level, the switch 308 is open (e.g., off).
Five time periods Ton, Tr, Tdemag, Toff, and Ts are shown in FIG. 4. The time period Ton starts at time t0 and ends at time t1, and the time period Tdemag starts at time t1 and ends at time t2. The time period Tr starts at the time t2 and ends at time t3, and the time period Toff starts at the time t1 and ends at the time t3. The time period Ts starts at t0 and ends at t3. For example, t0≤t1≤t2≤t3. The time periods Tdemag and Tr are within the time period Toff. The time period Ts is the switching period of the system 300, and includes the time period Ton and the time period Toff.
During the time period Ton, the power switch 308 is closed (e.g., on) as shown by the waveform 404. The voltage between the terminals 320 and 322 (e.g., Vds) keeps at a low value 412 (e.g., approximately zero as shown by the waveform 402). The current that flows through the power switch 308 increases over time from a low value 410 (e.g., approximately zero at t0) to a peak value 408 (e.g., Ipk at t1) as shown by the waveform 406.
At the beginning of the time period Tdemag (e.g., at t1), the power switch 308 becomes open (e.g., being turned off) as shown by the waveform 404. The voltage between the terminals 320 and 322 (e.g., Vds) increases from the low value 412 (e.g., approximately zero at t1) as shown by the waveform 402. The current that flows through the power switch 308 drops from the peak value 408 to a low value 414 (e.g., approximately zero) as shown by the waveform 406.
During the time period Tdemag, the power switch 308 remains open (e.g., off) as shown by the waveform 404. The current that flows through the power switch 308 remains at the low value 414 (e.g., approximately zero) as shown by the waveform 406. For example, the voltage between the terminals 320 and 322 (e.g., Vds) can be determined according to Equation 1. The parasitic capacitance 314 (e.g., CP) and the parasitic inductance 318 (e.g., Lk) cause high frequency resonation. The primary inductance of the transformer 334 (e.g., Lp) is demagnetized.
At the end of the time period Tdemag (e.g., at t2), the demagnetization of the primary inductance 334 (e.g., Lp) is completed. The QR controller 302 detects a falling edge of a voltage 332 (e.g., VDEM) at the auxiliary winding 330 that indicates the completion of the demagnetization.
During the time period Tr, the power switch 308 remains open (e.g., off) as shown by the waveform 404. The current that flows through the power switch 308 remains at the low value 414 (e.g., approximately zero) as shown by the waveform 406. The primary inductance 334 (e.g., Lp) and the parasitic capacitance 314 (e.g., Cp) cause resonation. The voltage between the terminals 320 and 322 (e.g., Vds) decreases from a high value 418 (e.g., at t2) to a valley value 416 (e.g., at t3) as shown by the waveform 402. When the switch 308 is closed (e.g., on) at the beginning of a next switching period, the voltage between the terminals 320 and 322 (e.g., Vds) is at the valley value 416. The current peaks caused by the parasitic capacitance 314 (e.g., Cp) through the power switch 308 is often reduced, and thus the switching loss and the EMI of the system 300 is often improved.
As shown in FIG. 4, the time period Tr during which the parasitic capacitance 314 (e.g., Cp) and the primary inductance 334 (e.g., Lp) resonate is less than the switching period Ts of the system 300. The system 300 operates approximately in a boundary-conduction mode (BCM). The output power of the system 300 can be determined based on the following equation.
                              P          out                =                  η          ×                                    I                              p                                                                                k                                      2                    ×                      [                                          N                ×                                  (                                                            V                      out                                        +                                          V                      F                                                        )                                ×                                  V                                      i                                                                                n                                                                                _                                                                                D                                                                                C                                                                                N                  ×                                      (                                                                  V                        out                                            +                                              V                        F                                                              )                                                  +                                  V                                      i                                                                                n                                                                                _                                                                                D                                                                                C                                                                        ]                                              (                  Equation                                                              2                )            where Pout represents the output power of the system 300, η represents the power transfer efficiency of the system 300, and Ipk represents the peak current on the primary side of the system 300. Additionally, N represents a turns ratio between the primary winding 304 and the secondary winding 306, Vout represents the output voltage on the secondary side of the system 300, and VF represents the voltage drop on the rectifying diode 310. Furthermore, Vin_DC represents the voltage level of the input voltage 326 (e.g., a power-factor-correction output voltage, or a line input rectified voltage), and D represents a turn-on duty cycle of the power switch 308.
The switching frequency of the system 300 can be determined based on the following equation.
                              f          s                =                                                            η                ×                                  V                                      i                                                                                n                                                                                _                                                                                D                                                                                C                                    2                                                            2                ×                                  L                  P                                ×                                  P                  out                                                      ×                          D              2                                =                                    η                              2                ×                                  L                  P                                ×                                  P                  out                                                      ×                                          [                                                      N                    ×                                          (                                                                        V                          out                                                +                                                  V                          F                                                                    )                                                                            1                    +                                                                  N                        ×                                                  (                                                                                V                              out                                                        +                                                          V                              F                                                                                )                                                                                V                                                  i                                                                                n                                                                                _                                                                                D                                                                                C                                                                                ]                            2                                                          (                  Equation                                                              3                )            where fs represents the switching frequency of the system 300, Lp represents the inductance of the primary winding 304, and Pout represents the output power of the system 300. Additionally, η represents the power transfer efficiency of the system 300, Ipk represents the peak current on the primary side of the system 300, and N represents a turns ratio between the primary winding 304 and the secondary winding 306. Furthermore, Vout represents the output voltage on the secondary side of the system 300, VF represents the voltage drop on the rectifying diode 310, Vin_Dc represents the voltage level of the input voltage 326 (e.g., a power-factor-correction output voltage, or a line input rectified voltage), and D represents the turn-on duty cycle of the power switch 308.
According to Equations 1 and 2, the switching frequency of the system 300 often remains approximately constant, if the output load 316 and the input voltage 326 do not change. The frequency range for energy distribution of the conduction EMI is usually narrow. The conduction EMI cannot be reduced significantly in the conventional fly-back power conversion system 300 using the QR techniques.
FIG. 5 is a simplified diagram showing a conventional power conversion system including a QR controller. The power conversion system 500 includes a QR controller 502, a primary winding 504, a secondary winding 506, a power switch 508, a rectifying diode 510, a capacitor 512, an error-amplification-and-isolation component 516, a current sensing resistor 518, an auxiliary winding 520, and two resistors 522 and 524. The QR controller 502 includes a demagnetization detection component 526, a flip-flop 528, a gate driver 530, a PWM comparator 532, a diode 534, and two resistors 536 and 538. The QR controller 502 further includes four terminals 540, 542, 544 and 546. For example, the power switch 508 is a field-effect transistor (FET), a bipolar junction transistor (BJT) or an insulated-gate bipolar transistor (IGBT).
As shown in FIG. 5, the error-amplification-and-isolation component 516 receives the output voltage 548 on the secondary side of the system 500, and outputs a feedback signal 550 to the terminal 540 (e.g., terminal FB). The feedback signal 550 is received by at least the diode 534. In response, the resistors 536 and 538 generate a signal 552 to a non-inverting input terminal of the PWM comparator 532.
A primary current 554 that flows through the primary winding 504 is sensed by the current sensing resistor 518, which in response outputs the sensed signal 556 to the terminal 546 (e.g., terminal CS). The sensed signal 556 is received at an inverting input terminal of the PWM comparator 532. The PWM comparator 532 generates a comparison signal 558 based on the signal 552 and the sensed signal 556.
The resistors 522 and 524 receive an auxiliary current 560 that flows through the auxiliary winding 520, and in response outputs a demagnetization signal 562 to the terminal 542 (e.g., terminal dem). The demagnetization detection component 526 receives the demagnetization signal 562, and outputs a detection signal 564. The flip-flop 528 receives the comparison signal 558 at one input terminal, and the detection signal 564 at another input terminal. In response, the flip-flop 528 outputs a signal 566 to the gate driver 530. The gate driver 530 outputs a gate drive signal 568 through the terminal 544 (e.g., terminal gate) to drive the power switch 508.
The feedback signal 550 generated by the error-amplification-and-isolation component 516 is often used to control the peak values of the current 554 in order to regulate the output voltage 548. To make the system 500 operate stably, the error-amplification-and-isolation component 516 is usually compensated, and the bandwidth is often limited to below one-tenth or one-fifteenth of the switching frequency. Similar to what is discussed in FIGS. 3 and 4, the conduction EMI cannot be reduced significantly in the conventional power conversion system 500 using the conventional QR controller 502.
Hence it is highly desirable to improve techniques for reducing the EMI.

This patent is subject to a terminal disclaimer.

Systems and methods for reducing electromagnetic interference using switching frequency jittering

On-Bright Electronics (Shanghai) Co., Ltd.


The present invention provides antigen delivery compositions and methods of using same to prevent or to treat cancers and other infectious diseases.

To prevent and treat many of the deadly diseases and cancers, effective vaccines are needed. The new generation vaccine has many advantages over traditional vaccine. However, the potency of the new generation vaccine needs to be enhanced. Unfortunately, after 80 some years of development, Alum is still the only single adjuvant approved by the Food and Drug Administration (FDA) in the United States for use in humans. Therefore, novel vaccine delivery systems and/or adjuvants are desperately needed and desired.

Antigen delivery compositions and methods of use


A perforate projection screeen for movies, planetariums and the like having uniform reflectivity throughout its visible surface.. The screen is made from a plurality of perforate panels and is mounted, when in use, with a dark non-reflective chamber behind the screen. Any overlapped areas of the panels and any frame members or other objects, in close proximity to the back of the panels are covered with a black velour type material having vertically extending fibers to absorb the light rays entering the holes in those areas of the panels and thereby prevent light reflection from such holes which would cause undesirable light strips or areas to be visible in the overlapped portions of the screen or in the area of the screen where frame members or other objects are in close proximity to the back of the panel.

The use of perforate projection screens is well known in the prior art as shown and described in U.S. Pat. Nos. 2,366,761 issued to R. O. Walker, U.S. Pat. No. 3,992,841 issued to R. Ward, Jr. and U.S. Pat. No. 4,750,807 issued to G. Chamayou dit Felix.
Perforate screens are particularly useful as spherical or geodesic dome shaped screens used in planetariums to permit the flow of heating and cooling air through the screen to the inside of the domed screen from the space between the screen and the exterior dome of the planetarium thereby assuring the proper ventilation and temperature is maintained inside the domed screen and the viewing area of the planetarium.
One of the problems encountered when using a perforate screen is that when lap joints are used to connect adjacent perforate panels of the screen, the holes in overlapping edges of the adjacent panels are not always in alignment with each other and the surface of the overlapped panel sometimes shows through the holes in the overlapping panel edge thereby creating a strip of greater reflectivity than the rest of the surfaces of the panels where the holes are not blocked by the surface of an overlapped panel, or by a supporting frame member or some other object in close proximity to the back of the panel.
The prior art devices have attempted to eliminate these areas of greater reflectivity by applying black paint or black tape to the surface of any overlapped panel edges or frame members which would show through the holes in the overlapping perforate panels.
It has been found however, that even when black paint or tape is used to cover the overlapped surfaces there is still sufficient reflectivity from the black surfaces that the overlapped surfaces showing through the holes still create strips or areas of higher reflectivity than the rest of the screen. These strips or areas show up as light strips or other shapes on the screen and thereby break the desired continuity of the screen and distract from the images being projected on the screen.


A computer system comprising a first and second computer entity having program-accessible information; wherein the second computer entity is arranged to modify the second computer entity's program-accessible information in accordance with a received change to the first computer entity's program-accessible information if the second computer entity determines that the received change to the first computer entities program-accessible information complies with a first predetermined criteria and irrefutably establishes that the received change is associated with the first computer entity.

The present invention relates to a method and system for updating an information state.
Organisations are increasingly using the Internet to offer their services and to utilise the services of others.
However, even though the Internet provides tremendous potential for allowing parties to interact it can prove difficult, when using the Internet, to provide assurances that parties to an agreement are acting in a trusted manner.
In particular, the provision of services require that all associated parties mutually agree certain conditions, where a party needs confidence that once a condition has been agreed another party can not repudiate what has already been agreed.
One solution to this problem has been the use of trusted third parties, where each party to an agreement interacts with a trusted third party, thereby allowing the trusted third party to maintain a master copy of an information state that defines the terms agreed between the parties and, as such, prevent the repudiation by any party of any of the agreed terms.
However, due to the nature of the relationship with a trusted third party such relationships can take a long time to establish.
It is desirable to improve this situation.

System for updating an information state received from another computer based upon validation rules that apply to the information state

Hewlett-Packard Development Company, L.P.


An infant warmer apparatus includes a base, a crib and a frame. The frame includes a support plate assembly comprising a plate member defining a pivot aperture and a notch. The heater assembly includes a canopy, and a mounting shaft. The mounting shaft defines a first terminal end secured to the canopy and a second terminal end passing through the pivot aperture of the plate member. The mounting shaft includes a biasing pin configured to settle within the notch of the plate member when the heater assembly is in a neutral position relative to the frame. The mounting shaft is rotatable within the pivot aperture such that the heater assembly can be pivoted in either a clockwise or counter-clockwise direction from the neutral position. A controller is configured to turn off the heater assembly when the heater assembly is pivoted out of the neutral position.

The subject matter disclosed herein relates to an infant warming apparatus with a pivotable heater assembly.
Warming devices may be implemented to warm an infant and to supply the necessary heat to maintain the infant at a predetermined temperature. Infant warmers commonly have an overhead radiant heater that is located above the infant and which thus radiates energy in the infrared spectrum to impinge upon the infant to maintain the predetermined temperature.
One problem with conventional infant warmers is that the heater assembly limits access to the infant. As an example, diagnostic and/or imaging equipment that is introduced from a vertical position above the infant can interfere with the heater assembly. It is therefore generally necessary to remove the infant from a conventional warmer device before implementing such diagnostic and/or imaging equipment. A more specific problem attributable conventional warmers is that they require a user to handle the infant during many diagnostic and/or imaging procedures, which can increase the risk of infection.


A color cathode ray tube comprises a vacuum envelope including a substantially rectangular faceplate having a first axis and a second axis perpendicular to the first axis and a plurality of necks opposing to the faceplate. A plurality of electron gun assemblies are received in the necks respectively. A phosphor screen is formed on the inner surface of the faceplate and includes a plurality of continuous segment regions scanned with electron beams emitted from corresponding ones of the assemblies. The mask member is received in the vacuum envelope and has a plurality of effective regions through which the electron beam pass. A plurality of effective regions are formed into a plurality of groups.

1. Field of the Invention
The present invention relates to a color cathode ray tube and, more particularly, to a color cathode ray tube of a multineck structure.
2. Description of the Related Art
Color cathode ray tubes in which a fluorescent screen in a vacuum envelope is scanned by electron beams and an image is displayed on the screen, are conventionally used as high definition broadcasting devices or computer terminal high resolution graphic display devices. For these applications, increased resolution is desirable. High resolution in a color cathode ray tube can be achieved by minimizing an electron beam spot on its screen. Conventional tubes have been improved by elongating the electron gun assembly and enlarging its diameter. These improvements have not been completely satisfactory because as the diameter of the tube is increased, the distance between the electron gun assembly and the screen also increases, resulting in undesirably large magnification of the electron lens. In other words, in order to achieve high resolution, it is important to shorten the distance between the electron gun assembly and the screen. For this purpose, the tube may be designed as a wide angle deflection tube. However, in such a tube, the magnification at the center region of the screen differs from that at the peripheral region thereof.
In order to solve the above problem, Japanese Patent Disclosure (Kokai) No. 48-90428 describes a multitube structure display device having a plurality of small or medium cathode ray tubes arranged in the horizontal or vertical direction to display an image on a large screen with high resolution.
A conventional display device of the multi-tube structure can be effectively used outdoors to display an image on a very large screen divided into blocks.
However, such a display device is not suitable for a medium-size screen, i.e., about 40 inches, since the joints of the divided blocks of the screen stand out and result in a poor image. In particular, when this display device is used as a computer-aided design graphic terminal, the presence of joints is a significant shortcoming.
In addition, U.S. Pat. No. 3,071,706, Japanese Utility Model Publication (Kokoku) No. 39-25641, and Japanese Patent Publication (Kokoku) No. 42-4928 and No. 50-17167 describe a multi-tube structure in which a plurality of independent tubes are integrated into a screen. However, in such an arrangement, when a screen is divided into a plurality of separately scanned segment regions, rasters in the adjacent segment regions overlap each other at their boundaries or have a blank therebetween and result in a poor image.
In order to solve this problem, Japanese Patent Disclosure (Koki) No. 61-256551 (U.S. Pat. No. 4,714,856) describes a color cathode ray tube having a multineck structure. As shown in FIG. 1, the color cathode ray tube comprises a vacuum envelope having a single faceplate 1, on which a phosphor screen 2 is formed, and a plurality of necks 3a, . . . 3d. A plurality of electron gun assemblies 4a, . . . 4d are respectively received in necks 3a, . . . 3d. Phosphor screen 2 is defined by a plurality of continuous segment regions each of which is scanned with electron beams 5R, 5G, and 5B from a corresponding electron gun assembly. A shadow mask 6 is received in the panel and faces the screen 2. The shadow mask 6 has a plurality of effective row and column regions 7 corresponding to the segment regions and a non-effective region 8 for surrounding and partitioning the respective segment regions. A plurality of deflection units 9a, . . . 9d, generating deflection magnetic fields, are respectively mounted in the vicinity of the electron gun assemblies 4a, . . . 4b to deflect the electron beams 5R, 5G, and 5B.
Generally, in a color cathode ray tube with a shadow mask, only 30% or less of the electron beams originally emitted by the electron guns pass through the apertures formed in the shadow mask. The remaining 70% or more of the beams impinge on the shadow mask, which may become heated. As a result, the shadow mask thermally expands and deforms. When it deforms, landing error occurs and color purity is degraded. In conventional color cathode ray tubes with one phosphor screen and one electron gun assembly, deformation of the shadow mask is symmetric about the center of the phosphor screen corresponding to the center of the shadow mask. In order to correct landing error, a means for adjusting the distance between mask and screen in response to the temperature of the shadow mask and the mask frame has been used.
In a color cathode ray tube having a multineck structure, as mentioned above, deformation is symmetric about the center of the shadow mask, similar to conventional tubes. As a result, the direction of deformation of each effective region relative to the corresponding segment region is different in accordance with its position in the shadow mask. Therefore, compensation of the landing error as used in a conventional tube is not effective in this type of tube.

Color cathode ray tube having a plurality of masks


A method for enhancing read performance in an ARMR system includes: obtaining a first reader offset profile corresponding to a first reader of a multi-reader array head in the ARMR system; obtaining at least a second reader offset profile corresponding to at least a second reader of the multi-reader array head in the ARMR system; combining the first and second reader offset profiles to generate a combined reader offset profile; and controlling a location of the multi-reader array head in the ARMR system relative to at least one target track associated with a magnetic storage medium to be read as a function of a peak amplitude of the combined reader offset profile.

The magnetic disk drive recording industry continues to pursue advances in technology that will sustain enhancements in recording density in a cost-effective manner. Two approaches currently under investigation are bit patterned media recording (BPMR) and heat-assisted magnetic recording (HAMR). An objective of these approaches is to overcome challenges posed by the super-paramagnetic limit that imposes a trade-off among three fundamentally competing recording parameters: media signal-to-noise ratio (SNR), writability and thermal stability. BPMR and HAMR, however, require modifications to the media and heads which significantly increase costs. Another technology, two-dimensional magnetic recording (TDMR), which uses conventional media and a new multiple-head configuration, relies on powerful signal processing in an attempt to achieve a theoretical one bit-per-grain recording density.
As a practical near-term milestone, array-reader based magnetic recording (ARMR) has been proposed to increase areal density with an array-reader and associated signal processing.

Skew-tolerant reader configurations using cross-track profiles for array-reader based magnetic recording


There is provided a position detection system including an imaging unit to capture an image of a projection plane of an electromagnetic wave, an electromagnetic wave emission unit to emit the electromagnetic wave to the projection plane, a control unit to control emission of the electromagnetic wave by the electromagnetic wave emission unit, and a position detection unit including a projected image detection section to detect a projected image of an object existing between the electromagnetic wave emission unit and the projection plane based on a difference between an image of the projection plane captured during emission of the electromagnetic wave by the electromagnetic wave emission unit and an image of the projection plane captured during no emission of the electromagnetic wave, and a position detection section to detect a position of the object based on a position of the projected image of the object.

1. Field of the Invention
The present invention relates to a position detection system, a position detection method, a program, an object determination system and an object determination method.
2. Description of the Related Art
Studies of new interfaces for inputting information to information processing units such as a PC (Personal Computer) and a cellular phone are widely conducted today. For example, an interface is proposed that captures the image of an operator by an imaging unit, extracts the hand, finger or the like of the operator from the captured image and detects the motion (gesture) of the extracted hand or finger as an input operation.
In order to appropriately detect the input operation of an operator in such an interface, it is important to efficiently and accurately extract the hand, finger or the like of the operator from a background image. Thus, studies on the extraction of the hand or finger of an operator are widely conducted. For example, Japanese Patent No. 3997566 discloses a method that stores an image captured during non-operation by an operator as a reference image and then calculates a difference between the reference image and an image captured during operation by the operator to thereby detect the hand or finger of the operator.

Position detection system, position detection method, program, object determination system and object determination method


Embodiments of a filter for electromagnetic radiation are disclosed herein. The filter includes a first glass-ceramic substrate having a first refractive index, a second glass-ceramic substrate having the first refractive index, and a first region disposed between the first glass-ceramic substrate and the second glass-ceramic substrate. The first region has a second refractive index that is less than the first refractive index. Further, the second glass-ceramic substrate is arranged substantially parallel to and spatially disposed from the first glass-ceramic substrate. The filter transmits at least 70% of electromagnetic radiation within a band of frequencies and reflects at least 80% of electromagnetic radiation outside the band of frequencies. The band of frequencies is located within the frequency range of 20 GHz to 100 GHz.

The disclosure relates generally to filters for electromagnetic radiation and, in particular, to a filter including a plurality of glass-ceramic substrates configured to transmit a desired 5G signal. Deployment of the 5G network has required the installation of many new antennas to send and receive 5G signals. Such antennas relay data throughout the network in a highly directional manner. Efficient sending and receiving of these 5G signals allows for the 5G network to be built out in an economical manner.


A seal for a turbine engine is provided that includes a seal body disposed at a base of a turbine engine blade and a wing portion extending axially from the seal body. The wing portion has a first portion substantially parallel with a centerline of the engine and an angled upturn portion. The first portion extends between the seal body and the angled upturn portion. An angle between the engine centerline and the angled upturn portion is between about 0 degrees and about 90 degrees.

The present invention relates generally to rotary machines and, more particularly, to angel wing seals.
Rotor assemblies used with turbine engines include a row of circumferentially-spaced rotor blades. Each rotor blade, sometimes referred to as a "bucket" includes an airfoil that includes a pressure side and a suction side that are connected together along leading and trailing edges. Each bucket extends radially outward from a bucket platform. Each bucket typically includes a dovetail that extends radially inward from a shank extending between the platform and the dovetail. The dovetail is used to couple the rotor blade to a rotor disk or spool.
Wheel space cavities, defined between the rotating parts, such as the buckets, and the stationary parts of gas turbines, may be purged with cooling air to maintain the temperature of the wheel space and rotor within a desired temperature range, and to prevent hot gas path ingestion into the cavities. Seals are provided to seal the wheel space cavity. At least some known rotor blades include "angel wing seals" that extend generally axially away from the blades to form a seal by overlapping with nozzle seal lands extending from fixed components in the gas turbine. Typically, angel wing seals are cast integrally with the blade and are generally substantially planar, in cross-section, or include a preformed 90° bend at the tip that enables a portion of the angel wing to extend substantially perpendicular to the turbine engine centerline. Thus, if the angel wing seals need to be adjusted, for example to change a gap size with respect to an adjacent engine component, the gap size may be changed with respect to one direction only (i.e., axial or radial) by grinding down the angel wing seal.

Seals for rotary devices and methods of producing the same


A microfluidic method and device for testing and analyzing chemotaxis by providing a stable, static fluid gradient. The device includes a sink reservoir for receiving biological cellular material and a source reservoir for receiving a chemoattractant. The biological cellular material migrates through a low fluid volume microfluidic gradient channel located between the source and sink reservoirs. The fluid in the gradient channel is static and stable due to a high fluid volume closed circuit bypass microfluidic channel also in fluid communication with the source and sink reservoirs, whereby the bypass channel relieves any pressure differential imparted across the gradient channel.

To study chemotaxis, several forms of gradient forming devices have been developed that vary in terms of ease-of-use and achievable throughput. The Dunn chamber consists of a central well surrounded by an annular well, whereby both structures are etched in a glass slide. The two wells are separated by an annular region that is shallower than the wells and lower than the top of the glass slide. The annular well is filled with a soluble factor solution, and the central well is filled with an appropriate buffer solution or cell culture medium.
Cells are seeded on a glass coverslip. The coverslip is laid on the slide containing the wells such that the cells are located within the area of the center well. The cells closing the shallow annular region separating the wells will see a gradient of the soluble factor. The cell movement in response to that gradient is recorded via video microscopy and analyzed using suitable software.
Microfluidic devices have been employed to create gradients and pattern cells. A gradient may be created in a microfluidic device by employing a network of successive branching and diffusive mixing units that form a gradient across the width of a microfluidic downstream channel. The stability of that gradient is reliant upon a constant flow through the device. Once constant flow ceases, the gradient dissipates and concentration becomes uniform across the channel. However, such microfluidic devices have important disadvantages. The flow biases the migration of cells in the direction of flow, which affects the results. The flow also removes soluble factors secreted by the cells, which abolishes important cell-to-cell signaling. Constant flow also utilizes an excessive fluid source that wastes expensive reagents.
Other microfluidic devices produce gradients without flow. Static gradients avoid some disadvantages of flowing microfluidic devices, however, they have other disadvantages. Static microfluidic devices employ microfluidic channels having a small volume. Small disturbances can easily distort or eliminate a gradient that exists in a small-volume microfluidic channel. Several important sources of disturbance can be identified. A fluid level difference between access ports on the ends of a microfluidic channel may produce flow due to gravity. Surface tension significantly influences the length scale of microfluidic channels. Any differences in radius of curvature of the air-liquid interface of access ports produce a pressure differential between the ports that tends to produce flow. Evaporation is proportional to several parameters, such as the surface area of the air-liquid interface, the exponent of negative one over the ambient temperature, ambient humidity, and air convection. Evaporation causes an imbalance in fluid level or radius of curvature, which causes fluid flow and an unstable gradient. Evaporation is proportional to the vapor pressure of the fluid. The vapor pressure is proportional to exp(−H/RT), where H is the enthalpy of vaporization.
Therefore, since there is a long felt need in the industry for a microfluidic gradient device, alternative methods and microfluidic devices described herein below make a desirable contribution.

Microfluidic device having stable static gradient for analyzing chemotaxis


There is disclosed certain heteroaroyl 10-deazaaminopterin and 5, 10 and 8, 10 di deazaminopterin compounds and their use for treatment of rheumatoid arthritis and related diseases and preparative process.. Also disclosed are 10 alkenyl-(and alkynyl) 10-deazaminopterins also disclosed for treatment of rheumatoid arthritis and for leukemia and ascites tumors and preparative process.

1. Field of the Invention
The current invention concerns novel antiinflammatory and antineoplastic 10-deazaaminopterin compounds. In particular, the invention concerns heteroaroyl-10-deazaaminopterins and 10-alkenyl or 10-alkynyl-10-deazaaminopterins having pronounced antiinflammatory activity, antileukemic and antitumorigenic activity, as well as a method for treatment of inflammatory diseases, leukemia and tumors. Pharmaceutical compositions containing these heteroaroyl-10-deazaaminopterin compounds are also disclosed. The invention further concerns a process for preparation of these compounds.

Heteroaroyl 10-deazaamino-pterine compounds and use for rheumatoid arthritis