A device for implementing control or regulating functions in a motor vehicle having at least one control unit, at least one peripheral element, and at least one control module. The peripheral element is connected to the control unit and receives and/or sends signals, and the control or regulating functions are implemented by a triggering and/or analysis of the signals of the peripheral element. The device also includes at least one control module connected between the control unit and the peripheral element, the control module being composed of an electronic unit and a flexible extension, and the peripheral element being contacted to the flexible extension.

The present invention relates to a device for implementing control or regulating functions and a method of controlling or regulating in combination with a motor vehicle.
In regulating or control functions in a vehicle, various functions are typically implemented by different devices, each having control units adapted individually to the given functions. Examples include devices controlling the running gear functions for anti-lock brake systems, traction control, electronic stability programs, or other systems which influence wheel speed and in particular increase driving stability and/or safety, as well as drive control or transmission control systems. Different control devices, such as a brake control device, engine control device, or transmission control device, are used, depending on the respective application, often connected by an internal communication system within the vehicle, in particular a bus system such as CAN or TTP/C. In addition, there are also functions in a vehicle necessitating the use of multiple comparable control devices, i.e., having a similar hardware design. This is true, for example, of lock systems or the control of actuator motors such as those used for window lift motors or in adjusting outside rearview mirrors, where the functions and thus also the comparable control devices must be implemented multiple times, e.g., according to the number of vehicle doors. In general, for reasons of cost and complexity, an attempt is made to use standard control devices, i.e., universally usable control devices which may be used for different functions. This means that although the functionality varies, e.g., in the case of a lock system, taking into account the passenger door to the driver""s door and the front doors to the rear doors, an attempt is made to use one and the same standard control device for each individual door for reasons of simplicity or at least to minimize the variety of models of control devices and/or control units as much as possible, because otherwise up to three different control units would have to be used with such a lock system.
German Patent No. DE 37 38 915 describes a universal control device for regulating systems. A universal control device that determines the system design and stores it on the basis of an external command only after installation is used. However, such a universal control device must also have a variety of variant-determining components, i.e., specially designed circuit parts and/or components and individualized interfaces via which corresponding sensors or actuators may interact with the control device at the same time in order to ensure a large number of control and/or regulating functions, in particular different control and/or regulating functions.
Integration of such variant-determining circuit parts and interfaces for the desired control functions into one control device has proven to be relatively complicated and cost intensive. The need for such a universal control device to fulfill a variety of different control or regulating functions has a negative effect on the complexity of the control unit due to the variant-determining circuit parts and interfaces which must be present concurrently. With regard to the space required, such a universal control unit will also be larger than a control device variant tailored specifically to the respective regulation or control function.
Conventionally, several electronic control devices having the same hardware design are used in a vehicle. The specific function of these control devices, which communicate via a bus, is impressed upon them from the outside, depending on their specific application. Here again, variant-determining circuit parts and interfaces corresponding to the universal control device must be present in each control device for implementation of the possible or desired preselectable regulation or control functions.
Conventional solutions do not yield optimum results. Thus, in accordance with an example embodiment of the present invention, a wide variety of different control and regulating functions may be implemented, while optimizing cost and complexity aspects.
According to an example embodiment of the present invention, a device for implementing control and/or regulating functions in a motor vehicle is provided, having at least one control unit and at least one peripheral element, the peripheral element being connected to the control unit and receiving and/or sending signals, and the control or regulating functions being implemented by triggering and/or analyzing the signals of the peripheral elements. The device contains at least one control module which may be connected between the control unit and the peripheral element and may be composed of an electronic unit and a flexible extension, and the peripheral element may be contacted to the flexible extension. In this way, the specific variant-determining control or regulation electronic systems may be used expediently, based on the on-site controlled system, so that the peripheral elements, i.e., in particular actuators and sensors, may be positioned very flexibly and appropriately with respect to the controlled system.
A film extension, for example, a film conductor, may be used as a flexible extension. Printed conductors or conductor structures for power and/or signal transmission to and/or from the peripheral elements may be encased in a nonconducting flexible material, in particular a moldable material such as plastic. It is advantageous that in addition to the flexible positioning of the control modules and/or the peripheral elements, problems such as unfavorable ambient influences for the conductor structures and/or the signal and/or power lines may also be largely ruled out. In addition, in view of problems involving electromagnetic compatibility, the conductor structures and/or the flexible extension may advantageously also be shielded with respect to such interference, and/or the transmission of their own interfering radiation by the control module, i.e., the flexible extension and/or components placed thereon, in particular over the conductor structures may be suppressed or prevented.
Use of a uniform, standardized interface between the flexible extension, in particular the conductor film and the electronic unit may allow different electronic units having flexible extensions of any desired design may be combined to form control modules.
Electronic components may be installed on the flexible extension and controlled there or analyzed to advantage. In this way, it is possible to produce an optimum contact or an optimum working relationship, e.g., between sensors and the elements to be sensed. Components for controlling actuators, in particular power components, may also be mounted on the flexible extension to advantage in such a way that they migrate physically to the actuators and thus interfering radiation and/or power loss due to a high-power signal may be prevented or diminished through radiation of the conductor structure.
The electronic unit may be designed to be scalable, i.e., composed of predefinable uniform components or combinations of components and circuit parts, so that the variants of the electronic units and thus the control modules may be limited according to a modular principle. A wide variety of different functionalities for control or regulation of operating sequences or processes in a vehicle may be implemented expediently through combined use of several mutually supplementary control modules or electronic units so that nevertheless any missing functionality may be achieved by installing appropriate components on the flexible extension.
Finally, the device and the method of control or regulation of operating sequences of a motor vehicle is an advantageous embodiment of the present invention, where the control or regulation may be implemented by at least one control unit, and the control unit contains programs and/or data for controlling and/or regulating certain first operating sequences, at least one peripheral element being connected to the control unit and receiving data in the form of signals therefrom or sending them thereto. The respective peripheral element may be advantageously used to control or regulate second operating sequences, and before reception and/or after transmission, the signals are processed in a control module, which is connected between the control unit and the peripheral element. The processing of the signals by the control module may be used to supplement the electronic components and/or functions required for the control or regulation of the second processes or operating sequences, when these electronic components and/or functions are missing in the control unit and/or peripheral element for controlling or regulating the second processes or they may not be used for that purpose, because they are available to the first processes, for example, the control or regulation of the first processes being implemented in addition to and independently of the control or regulation of the second processes. Thus, a control unit which is actually used for first processes and is already present for that purpose may also be simply used to advantage additionally if it still has capacity for the programs and/or data of the second processes, which are not being executed in the control module having a simple design. Thus, other possibilities for establishing contact may also be used expediently in addition to or instead of the interfacing of peripheral elements via flexible extensions.
Simple product care may be possible according to the present invention, because faulty or outdated control modules, i.e., control units, may be replaced easily and in an uncomplicated manner by control modules that are adequate for the function or that belong to a new generation.

Device for implementing control or regulating functions and a method of controlling or regulating in a vehicle


A driving device includes a first member, an annular second member arranged outside the first member, an annular first plate for connecting the first and second members to each other, and an annular second plate for connecting the first and second members to each other. The first and second members are moved relative to each other by supplying or exhausting a fluid into or from a space surrounded by the first and second members and the first and second plates. Also, at least one of (i) an inner diameter a1 of a portion of the second member at which the second member is connected to the first plate is different from an inner diameter a2 of a portion of the second member at which the second member is connected to the second plate and (ii) an outer diameter b1 of a portion of the first member at which the first member is connected to the first plate is different from an outer diameter b2 of a portion of the first member at which the first member is connected to the second plate.

The present invention relates to a driving apparatus suitable for precise position adjustment of members and an exposure apparatus including this driving device and, more particularly, to a driving device for adjusting the optical characteristics, such as magnification, aberration, and distortion, of a projection optical system which can be mounted on an exposure apparatus and the like, and an exposure apparatus including this driving device. The present invention is applicable to an optical apparatus serving as an additional optical system in an exposure apparatus, such as a semiconductor exposure apparatus, for accurately forming an image when projecting and exposing the image of an original plate (e.g., a mask and reticle) onto a target (e.g., a wafer), and an exposure apparatus including this optical apparatus.
A semiconductor exposure apparatus transfers different patterns formed on a plurality of original plates (reticles) onto a silicon wafer (substrate) in an overlaying manner. To form a high-integration circuit, an increase in not only resolution performance but also overlay accuracy is indispensable.
Factors causing overlay errors in the semiconductor exposure apparatus are classified into an alignment error, image distortion, and magnification error. The alignment error is reduced by adjusting the positions of the original plate (reticle) and the substrate (wafer) relative to each other. The magnification error can be adjusted by moving at least one optical element of the projection optical system in the direction of the optical axis. When moving the optical element in the direction of the optical axis, components other than the moving direction, particularly parallel eccentricity and a tilt error, of the optical element should be kept small.
Conventionally, as a projecting magnification adjusting device for a semiconductor exposure apparatus, an optical element moving device having a mechanism using parallel leaf springs is proposed (Japanese Patent Laid-Open No. 9-106944).
FIGS. 17A and 17B are plan and sectional views, respectively, of a conventional optical element moving device.
As shown in FIGS. 17A and 17B, the conventional optical element moving device has a movable table 1 for holding an adjustment lens 7 which adjusts the magnification, aberration, and the like of an optical system, and a cell 8 which supports the adjustment lens 7, and a stationary table 2 which forms part of the stationary portion of the projection optical system. A driving element 4 is comprised of bellows 9 and the like. One end of the driving element 4 is fixed to the stationary table 2, and the other end thereof is fixed to a clamp top plate 5 connected to the movable table 1. The movable and stationary tables 1 and 2 are connected to each other through a spring mechanism having a plurality of sets of opposing leaf springs 3 and a plurality of leaf spring retainers 6 for supporting the leaf springs 3. The driving element 4 is arranged in a gap between the plurality of sets of leaf springs 3 arranged on the movable table 1 to be equidistant from its center and symmetric with respect to an optical axis 20.
FIG. 18 shows the driving element 4 in detail.
The driving element 4 is constituted by the bellows 9 and two flanges 10a and 10b. The flange 10a is connected to the clamp top plate 5 at the gap between a plurality of sets of leaf springs 11. The other flange 10b is arranged in the stationary table 2 to transmit an air pressure applied to the interior of the bellows 9 to the movable table 1.
When a heavy, large-diameter movable portion such as a lens is driven by a mechanism utilizing leaf springs, since the eigenfrequency is low, the movable portion is largely vibrated by a disturbance or the like. To improve the transfer performance of the exposure apparatus, the mechanism desirably has a high eigenfrequency.
It is an object of the present invention to provide a high-precision driving device.
A driving device according to the first aspect of the present invention comprises a first member, an annular second member arranged outside the first member, an annular first plate for connecting the first and second members to each other, an annular second plate for connecting the first and second members to each other, and a fluid controller for supplying or exhausting a fluid into or from a space surrounded by the first and second members and the first and second plates, thereby moving the first and second members relative to each other. As a result, at least one of the first and second members can be driven at high precision.
A volume in the space can change as the first and second members move relative to each other.
The first and second members can move relative to each other in a direction substantially parallel to an axial direction of the annular second member. The first and second members can move relative to each other in a direction substantially parallel to a direction of gravity.
The first and second plates preferably have different shapes.
Either one of a1xe2x89xa0a2 and b1xe2x89xa0b2 is preferably satisfied where a1 is an inner diameter of a portion of the second member at which the second member is connected to the first plate, a2 is an inner diameter of a portion of the second member at which the second member is connected to the second plate, b1 is an outer diameter of a portion of the first member at which the first member is connected to the first plate, and b2 is an outer diameter of a portion of the first member at which the first member is connected to the second plate. In this case, (a1xe2x88x92b1) and (a2xe2x88x92b2) are further preferably substantially equal to each other.
A gas such as an inert gas is preferably used as the fluid.
The fluid controller preferably has a pressure control valve for controlling a pressure in the space.
The driving device preferably has a sensor for detecting a relative distance or displacement between the first and second members. The fluid controller preferably controls the pressure control valve, which controls the pressure in the space, on the basis of an output signal from the sensor. A target for the sensor preferably has an annular shape. A fixing member for fixing either one of the first and second plates to either one of the first and second members preferably also serves as the target for the sensor. The fixing member may have a portion formed by welding a metal to a nonconductive material. The sensor may be arranged in the space. The sensor may be, e.g., an electrostatic capacitance sensor.
The driving device preferably has a damper between the first and second members. The damper may have a piston, cylinder, and viscous substance. The damper may be arranged in the space.
The first member may have a step structure on its outer side surface. The second member may have a step structure on its inner side surface.
The first and second members may move relative to each other within a range having as one boundary a position where rigidity of the first and second plates and a mass of at least one of the first and second members balance with each other. Alternatively, the first and second members may move relative to each other within a range including a position where rigidity of the first and second plates and a mass of at least one of the first and second members balance with each other.
A driving device according to the second aspect of the present invention for driving a first member and an annular second member, which is arranged outside the first member, relative to each other, comprises an annular electrode formed on at least one of the first and second members, and a detector for detecting a change in electrostatic capacitance concerning the electrode upon movement of the electrode, thereby detecting positions of the first and second members relative to each other. As a result, since the electrostatic capacitance concerning the electrode can be increased, the change in electrostatic capacitance upon movement of the electrode can be increased. Accordingly, the position detection precision can be improved.
A driving device according to the third aspect of the present invention comprises a first member, a second member arranged outside the first member, a first plate for connecting the first and second members to each other, a second plate for connecting the first and second members to each other, and a fluid controller for supplying or exhausting a fluid into or from a space surrounded by the first and second members and the first and second plates, thereby moving the first and second members relative to each other.
A driving device according to the fourth aspect of the present invention comprises a first member for supporting an optical element, an annular second member arranged outside the first member, an annular first plate for connecting the first and second members to each other, an annular second plate for connecting the first and second members to each other, and a fluid controller for supplying or exhausting a fluid into or from a space surrounded by the first and second members and the first and second plates, thereby moving the first member. As a result, the optical element can be driven at high precision.
A driving device according to the fifth aspect of the present invention comprises a first member for supporting an optical element, a second member arranged outside the first member, a first plate for connecting the first and second members to each other, a second plate for connecting the first and second members to each other, and a fluid controller for supplying or exhausting a fluid into or from a space surrounded by the first and second members and the first and second plates, thereby moving the first member. As a result, the optical element can be driven at high precision.
An exposure apparatus according to the sixth aspect of the present invention comprises a projection system and a driving device for driving at least one optical element of the projection optical system. The driving device has a first member for supporting the optical element, an annular second member arranged outside the first member, an annular first plate for connecting the first and second members to each other, an annular second plate for connecting the first and second members to each other, and a fluid controller for supplying or exhausting a fluid into or from a space surrounded by the first and second members and the first and second plates, thereby moving the first member. As a result, a high-precision exposure apparatus can be provided.
An exposure apparatus according to the seventh aspect of the present invention comprises a projection optical system and a driving device for driving at least one optical element of the projection optical system. The driving device has a first member for supporting the optical element, a second member arranged outside the first member, a first plate for connecting the first and second members to each other, a second plate for connecting the first and second members to each other, and a fluid controller for supplying or exhausting a fluid into or from a space surrounded by the first and second members and the first and second plates, thereby moving the first member. As a result, a high-precision exposure apparatus can be provided.
A device manufacturing method according to the eighth aspect of the present invention comprises the steps of controlling a driving device to arrange at least one optical element of a projection optical system of an exposure apparatus at an appropriate position, and transferring a pattern onto a device material by utilizing the exposure apparatus, the driving device having a first member for supporting the optical element, an annular second member arranged outside the first member, an annular first plate for connecting the first and second members to each other, an annular second plate for connecting the first and second members to each other, and a fluid controller for supplying or exhausting a fluid into or from a space surrounded by the first and second members and the first and second plates, thereby moving the first member. As a result, a precise device can be manufactured.
A device manufacturing method according to the ninth aspect of the present invention comprises the steps of controlling a driving device to arrange at least one optical element of a projection optical system of an exposure apparatus at an appropriate position, and transferring a pattern onto a device material by using the exposure apparatus, the driving device having a first member for supporting the optical element, a second member arranged outside the first member, a first plate for connecting the first and second members to each other, a second plate for connecting the first and second members to each other, and a fluid controller for supplying or exhausting a fluid into or from a space surrounded by the first and second members and the first and second plates, thereby moving the first member. As a result, a precise device can be manufactured.
Other features and advantages of the present invention will be apparent from the following description taken in conjunction with the accompanying drawings, in which like reference characters designate the same or similar parts throughout the figures thereof.


Techniques for optimizing application specific integrated circuit (ASIC) and other IC pin assignment corresponding to a high density interconnect (HDI) printed circuit board (PCB) layout are provided. Applying the techniques described herein, pin assignments may be systematically and strategically planned, for example, in an effort to reduce the PCB layer count and associated cost, increase signal integrity and speed, reduce the surface area used by an ASIC and its support circuitry, reduce plane perforations, and reduce via crosstalk when compared to conventional designs with an ASIC mounted on a multilayered PCB.

1. Field of the Invention
Embodiments of the present invention generally relate to integrated circuits (ICs) and multilayer printed circuit boards (PCBs).
2. Description of the Related Art
As processing applications become increasingly more complex, the number of input/output (I/O) pins, configuration pins, and other types of pins of ICs required for power and to interface with other components continues to grow. It is not uncommon for some application-specific ICs (ASICs) to possess pin counts that exceed 2000. Advancements in technology continually push the envelope to design smaller and smaller surface mount ASICs. Corresponding reductions in package size present a challenge to package designers faced with accommodating the increased pin count.
One approach to meeting the packaging requirements has been to encapsulate ASICs in a surface-mountable ball grid array (BGA). Using a grid of solder balls as its connectors, a BGA enables the package to be only slightly larger than the integrated circuit housed within. All of these trends towards increased functionality in an obligatory diminished package have led to very high density ASICs. However, properly connecting a high density BGA ASIC to a PCB to route signals and maintain signal integrity creates numerous problems for both the ASIC designers and the printed circuit board designers.
Accordingly, techniques for efficiently routing signals from high pin count ICs on multilayer PCBs are needed.
Overview
Embodiments of the present invention provide techniques for optimizing application specific integrated circuit (ASIC) and other integrated circuit (IC) pin assignment corresponding to a high density interconnect (HDI) printed circuit board (PCB) layout. Such techniques may be applied, for example, in an effort to increase signal integrity and speed, reduce the surface area used by an ASIC and/or its support circuitry, reduce the PCB layer count, reduce plane perforations, and/or reduce via-routed signal crosstalk when compared to conventional layout techniques.
One embodiment of the present invention provides a method. The method generally includes dividing an array of pins of an IC into a plurality of rings, creating a mapping of each of the rings to a net category based on at least one criterion, and assigning each of a plurality of nets of the IC to a pin of one of the plurality of rings based on the mapping.
Another embodiment of the present invention provides a method. The method generally includes classifying nets of an ASIC according to categories based on at least one criterion; dividing an array of pins of the ASIC into rings and quadrants; assigning power nets to pins aggregated at or near a center of the array to form a power core; assigning ground nets to the ASIC pins to form L-shaped patterns; determining a mapping based on the categorized nets and the number of rings; and assigning the remaining nets to the remaining ASIC pins based on the mapping and one of the quadrants corresponding to the general direction in which each of the nets will be routed.
Yet another embodiment of the present invention provides an IC. The IC generally includes a plurality of pins, wherein a first portion of the pins are assigned to ground and arranged in L-shaped patterns.
Yet another embodiment of the present invention provides an IC. The IC generally includes a plurality of pins divided into a plurality of rings. The pins are assigned to signal nets grouped into the plurality of rings according to at least one of signal speed or an amount of associated circuitry.


A ferroelectric memory of the present invention comprises: a plurality of normal cells, each of which includes a first ferroelectric capacitor for holding data and a first transistor connected to a first electrode of the first ferroelectric capacitor; a first bit line connected to the first transistor; a first bit line precharge circuit which is a switch circuit provided between the first bit line and a ground; and a word line connected to a gate of the first transistor. The word line is deactivated to disconnect the first ferroelectric capacitor from the first bit line before the first bit line precharge circuit is driven to discharge a potential of the first bit line.

The present invention relates to a ferroelectric memory wherein a ferroelectric capacitor is used for a memory cell and to a method for reading data from the ferroelectric memory.
In recent years, with the progress of process miniaturization and increase in capacity, a larger number of ferroelectric memories have employed a memory cell formed by one transistor and one ferroelectric capacitor, which has a reduced memory cell size, in substitution for an existing memory cell formed by two transistors and two ferroelectric capacitors. A ferroelectric memory employing the memory cell formed by one transistor and one ferroelectric capacitor requires a reference cell for generating a reference potential (reference level) in addition to a normal cell for holding data, i.e., memory cell. Thus, preventing deterioration of data which can be caused due to a decrease in the amount of polarized charges held in a ferroelectric capacitor of a memory cell, or the like, and shortening the bit line discharge time of the memory cell have been more important for improving the reliability and operation speed of the ferroelectric memory.
Hereinafter, a conventional ferroelectric memory is described with reference to the drawings (see, for example, Y. Chung et al., "A 3.3-V 4-Mb Nonvolatile Ferroelectric RAM with a Selectively-Driven Double-Pulsed Plate Read/Write-Back Scheme", Symposium on VLSI Circuits Digest, June 1999, pp. 97-98).
FIG. 5 shows a circuit structure of a conventional ferroelectric memory. As shown in FIG. 5, the conventional ferroelectric memory includes a first normal cell 101 and a second normal cell 102 as memory cells for storing data. The first normal cell 101 is formed by a first transistor T1 and a first ferroelectric capacitor C1. The gate of the first transistor T1 is connected to a word line WL, and the drain of the first transistor T1 is connected to a first bit line BL1. The first electrode of the first ferroelectric capacitor C1 is connected to the source of the first transistor T1, and the second electrode of the first ferroelectric capacitor C1 is connected to a cell plate line CP. The second normal cell 102 is formed by a third transistor T3 and a third ferroelectric capacitor C3. The gate of the third transistor T3 is connected to the word line WL, and the drain of the third transistor T3 is connected to a third bit line BL3. The first electrode of the third ferroelectric capacitor C3 is connected to the source of the third transistor T3, and the second electrode of the third ferroelectric capacitor C3 is connected to the cell plate line CP.
The conventional ferroelectric memory of FIG. 5 includes a first reference cell 103 and a second reference cell 104 as memory cells for storing data used for generation of a reference potential. The first reference cell 103 is formed by a second transistor T2 and a second ferroelectric capacitor C2. The gate of the second transistor T2 is connected to a reference word line RWL, and the drain of the second transistor T2 is connected to a second bit line BL2. The first electrode of the second ferroelectric capacitor C2 is connected to the source of the second transistor T2, and the second electrode of the second ferroelectric capacitor C2 is connected to a reference cell plate line RCP. The second reference cell 104 is formed by a fourth transistor T4 and a fourth ferroelectric capacitor C4. The gate of the fourth transistor T4 is connected to the reference word line RWL, and the drain of the fourth transistor T4 is connected to a fourth bit line BL4. The first electrode of the fourth ferroelectric capacitor C4 is connected to the source of the fourth transistor T4, and the second electrode of the fourth ferroelectric capacitor C4 is connected to the reference cell plate line RCP.
The conventional ferroelectric memory further includes a reset circuit 105 which is connected to the first reference cell 103 and the second reference cell 104 for writing predetermined data in the reference cells 103 and 104. The reset circuit 105 is formed by a fifth transistor T5 and a sixth transistor T6. The drain of the fifth transistor T5 is connected to the first electrode of the second ferroelectric capacitor C2 at a node ST2. The drain of the sixth transistor T6 is connected to the first electrode of the fourth ferroelectric capacitor C4 at a node ST4.
The gates of the fifth transistor T5 and sixth transistor T6 of the reset circuit 105 are connected to a reset signal line RST. The source of the fifth transistor T5 is connected to an output line RFD0 of a reset driver circuit 106. The source of the sixth transistor T6 is grounded.
The reset driver circuit 106 includes a seventh transistor T7, an eighth transistor T8 and a ninth transistor T9. The drains of the seventh transistor T7 and the eighth transistor T8 are connected to the output line RFD0. The sources of the seventh transistor T7 and the eighth transistor T8 are grounded. The drain of the ninth transistor T9 is connected to the output line RFD0. The source of the ninth transistor T9 is connected to a power supply line. The gate of the seventh transistor T7 is connected to a first reset driver control line XRDIN2. The gate of the eighth transistor T8 is connected to a third reset driver control line RDIN4. The gate of the ninth transistor T9 is connected to a second reset driver control line XRDIN3.
The first bit line BL1 and the second bit line BL2 are grounded through a first bit line precharge circuit 107, which is formed by a tenth transistor T10, and a second bit line precharge circuit 108, which is formed by an eleventh transistor T11, respectively. The first bit line BL1 and the second bit line BL2 are connected to a first sense amplifier 115 as a bit line pair.
As well, the third bit line BL3 and the fourth bit line BL4 are grounded through a third bit line precharge circuit 109, which is formed by a twelfth transistor T12, and a fourth bit line precharge circuit 110, which is formed by a thirteenth transistor T13, respectively. The third bit line BL3 and the fourth bit line BL4 are connected to a second sense amplifier 116 as a bit line pair. The second bit line BL2 and the fourth bit line BL4 are connected to each other through a level equalizer circuit 111 which is formed by the fourteenth transistor T14.
The conventional ferroelectric memory of FIG. 5 further includes a control circuit 120 for controlling the above-described circuits. The control circuit 120 drives the word line WL, the cell plate line CP, the reference word line RWL and the reference cell plate line RCP to control the operations of the first normal cell 101, the second normal cell 102, the first reference cell 103 and the second reference cell 104.
The control circuit 120 controls the reset circuit 105 by driving the reset signal line RST. The control circuit 120 controls the reset driver circuit 106 by driving the first reset driver control line XRDIN2, the second reset driver control line XRDIN3 and the third reset driver control line RDIN4. The control circuit 120 controls the first bit line precharge circuit 107, the second bit line precharge circuit 108, the third bit line precharge circuit 109 and the fourth bit line precharge circuit 110 by driving a bit line precharge control line BP. The control circuit 120 controls the level equalizer circuit 111 by driving a level equalizer control line REQ. The control circuit 120 controls the first sense amplifier 115 and the second sense amplifier 116 by driving a sense amplifier activation line SAE.
The first normal cell 101, the second normal cell 102, the first reference cell 103, the second reference cell 104, the reset circuit 105, the reset driver circuit 106, the first bit line precharge circuit 107, the second bit line precharge circuit 108, the third bit line precharge circuit 109, the fourth bit line precharge circuit 110 and the level equalizer circuit 111 constitute a single memory unit. A plurality of such memory units are connected to the control circuit 120 such that the control circuit 120 can simultaneously control the memory units.
The circuit operation of the conventional ferroelectric memory is described with reference to the drawings. Herein, it is assumed that data "1" is stored in the first normal cell 101, data "0" is stored in the second normal cell 102, data "1" is stored in the first reference cell 103, and data "0" is stored in the second reference cell 104. Data "1" means that the ferroelectric capacitor has a residual dielectric polarization such that the first electrode is positive. Data "0" means that the ferroelectric capacitor has a residual dielectric polarization such that the second electrode is positive.
FIG. 6 illustrates an operation timing of a read operation of the ferroelectric memory shown in FIG. 5. As shown in FIG. 6, in the initial state, the bit line precharge control line BP has "H" level voltage (supply voltage) to be in an active state. Accordingly, the first bit line precharge circuit 107, the second bit line precharge circuit 108, the third bit line precharge circuit 109 and the fourth bit line precharge circuit 110 are driven. Therefore, the potentials of the first bit line BL1, the second bit line BL2, the third bit line BL3 and the fourth bit line BL4 are the ground voltage. At this point in time, the word line WL, the cell plate line CP, the reference word line RWL, the reference cell plate line RCP, the level equalizer control line REQ, the reset signal line RST, the third reset driver control line RDIN4, and the sense amplifier activation line SAE have "L" level voltage (ground voltage) to be in an inactive state. The second reset driver control line XRDIN3 has "H" level voltage to be in an inactive state. The first reset driver control line XRDIN2 has "H" level voltage to be in an active state.
In the first place, at time t00, the bit line precharge control line BP is pulled down to "L" level, whereby the first bit line BL1, the second bit line BL2, the third bit line BL3 and the fourth bit line BL4 are set afloat. At time t01, the level equalizer control line REQ is pulled up to "H" level. At time t02, the word line WL and the reference word line RWL are pulled up to "H" level. At time t03, the cell plate line CP and the reference cell plate line RCP are pulled up to "H" level.
As a result, the first transistor T1, the second transistor T2, the third transistor T3 and the fourth transistor T4 are turned on so that the first ferroelectric capacitor C1, the second ferroelectric capacitor C2, the third ferroelectric capacitor C3 and the fourth ferroelectric capacitor C4 are connected to the first bit line BL1, the second bit line BL2, the third bit line BL3 and the fourth bit line BL4, respectively. In the meantime, the second electrodes of the first ferroelectric capacitor C1, the second ferroelectric capacitor C2, the third ferroelectric capacitor C3 and the fourth ferroelectric capacitor C4 result in "H" level, so that data "1" stored in the first normal cell 101 and data "0" stored in the second normal cell 102 are output to the first bit line BL1 and the third bit line BL3, respectively, and data "1" stored in the first reference cell 103 and data "0" stored in the second reference cell 104 are output to the second bit line BL2 and the fourth bit line BL4, respectively.
At this point in time, the level equalizer control line REQ is at "H" level. Thus, the level equalizer circuit 111 is driven so that the second bit line BL2 and the fourth bit line BL4 are connected. Therefore, the potentials of the second bit line BL2 and the fourth bit line BL4 are equal so that a potential at the midpoint between the potential corresponding to data "1" and the potential corresponding to data "0" is output as the reference potential (reference level) to the second bit line BL2 and the fourth bit line BL4.
Then, at time t05, the reference cell plate line RCP and the cell plate line CP are pulled down to "L" level. At time t06, the reference word line RWL is pulled down to "L" level, whereby the second ferroelectric capacitor C2 and the fourth ferroelectric capacitor C4 are disconnected from the second bit line BL2 and the fourth bit line BL4, respectively. Thereafter, at time t07, the level equalizer control line REQ is pulled down to "L" level so that the level equalizer circuit 111 is turned off.
Then, at time t09, the sense amplifier activation signal SAE is pulled up to "H" level, whereby the difference between the potentials read into the first bit line BL1 and the second bit line BL2 is amplified by the first sense amplifier 115, and the difference between the potentials read into the third bit line BL3 and the fourth bit line BL4 is amplified by the second sense amplifier 116. As a result, the first bit line BL1 is at the supply voltage, the second bit line BL2 is at the ground voltage, the third bit line BL3 is at the ground voltage, and the fourth bit line BL4 is at the supply voltage.
At this point in time, the word line WL is at "H" level. Thus, the first electrode of the first ferroelectric capacitor C1 is at "H" level, and the first electrode of the third ferroelectric capacitor C3 is at "L" level.
In the meantime, at time t09, the first reset driver control line XRDIN2 is pulled down to "L" level, and the second reset driver control line XRDIN3 is pulled down to "L" level. As a result, the reset driver output line RFD0 rises to "H" level. At the same time, the reset signal line RST is pulled up to "H" level. Thus, the first electrode of the second ferroelectric capacitor C2 (node ST2) results in "H" level, and the first electrode of the fourth ferroelectric capacitor C4 (node ST4) results in "L" level.
Then, at time t10, the cell plate line CP and the reference cell plate line RCP are pulled up to "H" level, whereby data "0" is rewritten in the third ferroelectric capacitor C3 of the second normal cell 102 and the fourth ferroelectric capacitor C4 of the second reference cell 104.
Thereafter, at time t12, the reference cell plate line RCP and the cell plate line CP are pulled down to "L" level, whereby data "1" is rewritten in the first ferroelectric capacitor C1 of the first normal cell 101 and the second ferroelectric capacitor C2 of the first reference cell 103.
At time t14, the second reset driver control line XRDIN3 is pulled up to "H" level so that the ninth transistor T9 is turned off, and the third reset driver control line RDIN4 is pulled up to "H" level so that the eighth transistor T8 is turned on, whereby discharge of the node ST2 is started. In the meantime, the sense amplifier activation signal SAE is pulled down to "L" level. Then, at time t15, the first reset driver control line XRDIN2 is pulled up to "H" level so that the seventh transistor T7 is turned on, and the third reset driver control line RDIN4 is pulled down to "L" level so that the eighth transistor T8 is turned off, whereby discharge of the node ST2 is continued. Further, the bit line precharge control line BP is pulled up to "H" level, whereby discharge of the first bit line BL1, the second bit line BL2, the third bit line BL3 and the fourth bit line BL4 is started.
Then, at time t16, after the end of the discharge of the node ST2 and the first bit line BL1, the reset signal line RST and the word line WL are pulled down to "L" level so that the ferroelectric memory is restored to the initial state, whereby one cycle of the read operation is terminated.
In such a conventional ferroelectric memory, if the bit line precharge control line BP is pulled up to "H" level at time t15, the first bit line precharge circuit 107 is driven, and the first bit line BL1 is grounded. Thus, the first bit line BL1 transitions from "H" level to "L" level. At this point in time, the word line WL is at "H" level, and therefore, the first transistor T1 is on. The potential at the first electrode of the first ferroelectric capacitor C1 also abruptly transitions from "H" level to "L" level. This abrupt potential transition influences the cell plate line CP due to coupling of the first ferroelectric capacitor C1. Thus, as indicated by broken line circle A, undershoot noise is caused in the cell plate line CP which is at "L" level.
If the undershoot noise is caused in the cell plate line CP, the amount of polarized charges of a ferroelectric capacitor which holds data "0" is decreased by a mechanism described below, so that the reliability of the ferroelectric memory deteriorates.
FIG. 7 shows a hysteresis curve for a memory cell of the conventional ferroelectric memory. In FIG. 7, the horizontal axis represents the voltage applied to an electrode of a ferroelectric capacitor, ant the vertical axis represents the amount of polarized charges of the ferroelectric capacitor.
At time t15, the first electrode and the second electrode of the third ferroelectric capacitor C3 are both at the ground potential, and therefore, the polarized charge amount of the third ferroelectric capacitor C3 is at point G. However, if the voltage of the cell plate line CP connected to the second electrode is decreased from the ground potential to voltage —Vp due to the undershoot noise, voltage —Vp is applied between the first electrode and the second electrode of the third ferroelectric capacitor C3, and therefore, the polarized charge amount of the third ferroelectric capacitor C3 temporarily moves to point X. Since the polarized charge amount changes to draw a hysteresis curve, the polarized charge amount moves to point Y without returning to point G even when the cell plate line CP is restored to the ground potential, and the polarized charge amount decreases from 2P to 2Q. As a result, the potential difference between data "1" and data "0" becomes small in a data read operation, so that the potential difference between data "1" and data "0" is smaller than a potential which can be read by the sense amplifier. Therefore, there is a possibility that stored data is corrupted. Thus, the noise caused in the cell plate line deteriorates the nonvolatility of the ferroelectric memory and therefore greatly deteriorates the reliability of the ferroelectric memory.
In order to prevent the above-described deterioration of the reliability of the ferroelectric memory, the speed of discharging the bit lines is decreased as much as possible by using, for example, a slow transistor as the eighth transistor T8 such that the potential at the first electrode of the ferroelectric capacitor does not abruptly change as shown in FIG. 8.
However, if the speed of discharging the bit line is decreased, the cycle time of the ferroelectric memory becomes longer, and the operation speed of the memory decreases.

Ferroelectric memory and method for reading data from the ferroelectric memory

Matsushita Electric Industrial Co., Ltd.


The present invention relates to a polypeptide including an Fc variant produced by substituting a portion of the amino acid sequence of the Fc domain of a human antibody with a different amino acid sequence. The present invention also relates to an antibody including the polypeptide. The Fc variant can find application in a wide range of antibodies and Fc-fusion constructs. In one aspect, the antibody or Fc fusion construct of the present invention is a therapeutic, diagnostic or laboratory reagent, preferably a therapeutic reagent. The Fc variant is suitable for use in the treatment of cancer because its in vivo half-life can be maximized by optimization of the portion of the amino acid sequence. The antibody or Fc fusion construct of the present invention is used to kill target cells that bear a target antigen, for example cancer cells. Alternatively, the antibody or Fc fusion construct of the present invention is used to block, antagonize or agonize a target antigen. For example, the antibody or Fc fusion construct of the present invention may be used to antagonize a cytokine or a cytokine receptor.

With recent advances in biotechnology such as genetic recombination and cell culture, a great deal of research has been conducted on the structure and function of proteins throughout the world. Biotechnology promotes a better understanding of vital phenomena and plays a decisive role in elucidating the mechanism of pathogenesis of various diseases to pave the way for effective diagnosis and treatment of diseases, greatly contributing to an improvement in the quality of life. Particularly, since hybridoma technology for monoclonal antibody production by fusing B cells with myeloma cells was developed in 1975 (Kohler and Milstein, Nature, 256:495-497, 1975), extensive research and development has been conducted on immunotherapy using therapeutic antibodies in clinical applications, including cancer, autoimmune disease, inflammation, cardiovascular disease, and infection.
Therapeutic antibodies have much higher specificity for targets, are less biotoxic, and cause fewer side effects than existing small-molecule drugs. Another advantage of therapeutic antibodies is their long blood half-life (about 3 weeks). Due to these advantages, therapeutic antibodies are considered the most effective approaches for cancer treatment. Indeed, large pharmaceutical companies and research institutes have concentrated their R&D capabilities on therapeutic antibodies that specifically bind to and effectively remove cancer cells, including carcinogenic factors. Roche, Amgen, Johnson & Johnson, Abbott, and BMS are major pharmaceutical companies that are currently developing therapeutic antibody drugs. Particularly, Roche, who developed three innovative therapeutic antibodies, i.e. Herceptin, Avastin, and Rituxan, for applications in anticancer therapy, reached approximately 19.5 billion US dollars in sales for the therapeutic antibodies in 2012 to gain huge profits in the global market and is currently leading the global market for antibody drugs. Johnson & Johnson, who developed Remicade, is rapidly growing in the global market for antibodies due to the increased sales volume of Remicade. Other pharmaceutical companies such as Abbott and BMS are known to possess many therapeutic antibodies in the final stage of development. As a consequence, biomedicines, including therapeutic antibodies that are specific for target diseases and cause few side effects, are rapidly replacing small-molecule medicines that have led the global pharmaceutical market.
The Fc region of an antibody recruits immune leukocytes or serum complement molecules, which in turn triggers the clearance of defective cells such as tumor cells or infected cells. The Fc interface between Cγ2 and Cγ3 domains mediates interactions with the neonatal Fc receptor (FcRn) and its binding recycles endocytosed antibody from the endosome back to the bloodstream (Raghavan et al., 1996, Annu Rev Cell Dev Biol 12: 181-220; Ghetie et al., 2000, Annu Rev Immunol 18: 739-766). This process, coupled with preclusion of kidney filtration due to the large size of the full-length IgG antibody molecule, results in favorable antibody serum half-life in the range of 1-3 weeks. Further, binding of Fc to FcRn plays a key role in antibody transport. Accordingly, the Fc region is crucial for the prolonged serum persistence of circulating antibodies through an intracellular trafficking and recycle mechanism.
The administration of an antibody or an Fc-fusion protein as a therapeutic agent requires a predetermined frequency of injection taking into consideration the half-life of the therapeutic agent. A longer in vivo half-life allows for less frequent injection or lower dosing. Thus, in many clinical studies that are currently underway, many efforts have concentrated on the development of next-generation anticancer therapeutic antibodies and anticancer therapeutic proteins by the introduction of mutations into the Fc domain to increase the half-life of antibodies or the introduction of variants into the Fc domain to achieve a maximal ADCC effect (Modified from Cancer Immunol Res. 2015/Thomson Reuters).
However, despite the research groups' efforts aimed at developing some proteins and antibodies with increased binding affinity for FcRn and extended in vivo half-life by introducing some mutations into the Fc domain, a significant increase in in vivo half-life is not yet achieved. Under these circumstances, there is an urgent need to develop optimally mutated antibodies.
The description of the Background Art is merely provided for better understanding the background of the invention and should not be taken as corresponding to the prior art already known to those skilled in the art.

Antibody Fc variants for increased blood half-life


An industrial vehicle includes: an exhaust gas treatment device arranged in an engine room so as to be located above an engine, the exhaust gas treatment device including a pretreatment unit accommodating a diesel oxidation catalyst and extending in the forward/rearward direction, a selective catalytic reduction unit located adjacent to the pretreatment unit and extending in the forward/rearward direction, a connecting pipe including a mix portion, a first bent portion, and a second bent portion, the mix portion being located between the pretreatment unit and the selective catalytic reduction unit and extending in the forward/rearward direction, the first bent portion connecting the mix portion and an outlet of the pretreatment unit, the outlet being located close to a dividing wall, the second bent portion connecting the mix portion and an inlet of the selective catalytic reduction unit, the inlet being located far from the dividing wall, and a urea water injection device attached to the first bent portion of the connecting pipe so as to be located on an extended line of the mix portion and configured to inject urea water into an exhaust gas; a urea water tank arranged outside the engine room and storing the urea water; and a urea water pipe configured to guide the urea water from the urea water tank to the urea water injection device and penetrating the dividing wall.

In some of industrial vehicles, for example, a machine room is arranged behind a driver's cab and includes an engine room and a ventilation room which are separated by a dividing wall in a forward/rearward direction (a front direction of the driver's cab and its opposite direction). An engine is arranged in the engine room, and a radiator for the engine is arranged in the ventilation room.
In recent years, it has been proposed that to reduce NOx in an exhaust gas, such industrial vehicle includes an exhaust gas treatment device that executes a selective catalytic reduction (SCR) using urea water as a reducing agent. For example, as shown in FIGS. 5A and 5B, PTL 1 discloses an industrial vehicle 100 in which an exhaust gas treatment device 140 is arranged in an engine room 110 together with an engine 120.
Specifically, according to the industrial vehicle 100 disclosed in PTL 1, the exhaust gas treatment device 140 includes: a pretreatment unit 141 having a diesel particulate filter (DPF); and an SCR unit 142 that causes the urea water to react with the exhaust gas.
The pretreatment unit 141 is cylindrical, is arranged between the engine 120 and a dividing wall 130, and extends in a vehicle width direction. The SCR unit 142 is also cylindrical, is arranged above the pretreatment unit 141 and the engine 120, and extends in the vehicle width direction.
The pretreatment unit 141 includes an outlet 141a at a right side in the vehicle width direction, and the SCR unit 142 includes an inlet 142a at a left side in the vehicle width direction. The outlet 141a and the inlet 142a are connected to each other by a connecting pipe 143 extending in the vehicle width direction. Further, a urea water injection device 144 is attached to an end portion of the connecting pipe 143, the end portion being located at the right side in the vehicle width direction. The urea water injection device 144 injects the urea water into the exhaust gas in a direction along a center line of a linear portion of the connecting pipe 143, the linear portion extending in the vehicle width direction.
A radiator 170 for the engine is arranged in a ventilation room 150 that is separated from the engine room 110 by the dividing wall 130. Further, a urea water tank 160 storing the urea water is arranged under the ventilation room 150. Then, the urea water is guided from the urea water tank 160 to the urea water injection device 144 through a urea water pipe 161 penetrating the dividing wall 130. A urea water pump 162 is provided on a portion of the urea water pipe 161.


The present invention provides a system for measuring biomarker analytes indicative of various diseases comprising an array of sensors sensitive to volatile organic compounds. Particularly, the system is composed of a random network of single-walled carbon nanotubes (SWCNTs) coated with non-polar small organic molecules in conjunction with learning and pattern recognition algorithms. Methods of discriminating between breath samples of healthy individuals and of lung cancer patients are disclosed.

Breath testing has long been recognized as a reliable technique for diagnosing certain medical conditions through the detection of specific volatile metabolites in exhaled breath (Buszewski et al., Biomed. Chromatogr., 2007, 21, 553-566). The analysis of breath offers several advantages being a non-invasive technique that possess the potential for direct and real-time monitoring (Cao et al., Crit. Rev. Analy. Chem., 2007, 37, 3-13).
Although exhaled breath is mainly composed of N2, O2, CO2, water vapors and other atmospheric constituents (e.g., argon and the like), many volatile compounds which are produced by metabolic processes within the body are found therein. Such metabolic mixtures are relatively less complicated than those obtained from serum or urine samples, and include nitric oxide, nitrogen dioxide, sulfur-conta ining compounds, hydrogen peroxide, carbon monoxide, hydrogen, ammonia, ketones, aldehydes, esters, and alkanes.
Mixtures containing unique compositions of metabolites are indicative of various medical conditions including tissue inflammation (e.g., asthma), immune responses (to cancer cells or bacteria), metabolic disorders (e.g., diabetes), digestive processes, liver and kidney dysfunction, cardiac disorders, gum disease, as well as other physiological conditions. Moreover, mixtures of volatile compounds are characteristic of a certain disease and often display different patterns at different stages of the disease.
Detection of volatile compounds as biomarkers for the diagnosis of medical conditions can be performed using olfactometry systems. These systems, also known as electronic nose devices perform odor detection through the use of an array of cross-reactive sensors in conjunction with pattern recognition algorithms. In contrast to the hitherto "lock-and-key" based devices, wherein each sensor produces an electronic response from a single analyte, in the electronic nose device each sensor is widely responsive to a variety of odorants. In this architecture, each analyte produces a distinct signature from the array of broadly cross-reactive sensors. This configuration allows to considerably widen the variety of compounds to which a given matrix is sensitive, to increase the degree of component identification and, in specific cases, to perform an analysis of individual components in complex multi-component mixtures. Pattern recognition algorithms can then be applied to the entire set of signals, obtained simultaneously from all the sensors in the array, in order to glean information on the identity, properties and concentration of the vapor exposed to the sensor array. Various algorithms and computer controlled systems for olfactometry known in the art are disclosed, for example, in U.S. Pat. Nos. 6,411,905, 6,606,566, 6,609,068, 6,620,109, 6,767,732, 6,820,012, and 6,839,636.
US Patent Application No. 2002/0117659 discloses electrical devices comprised of selectively functionalized nanowires or nanotubes for the detection of analytes.
U.S. Pat. No. 7,153,272 provides methods of collecting and detecting compounds in a human breath sample, comprising a breath collector and a breath analyzer with a plurality of sensors. The methods are directed towards detecting breath compounds for monitoring various medical conditions.
U.S. Pat. No. 7,076,371 discloses methods for characterizing a particular condition or disease from a set of volatile markers which are found, for instance, in the exhaled breath of a person. These markers are detected using a volatile substance detector, such as an artificial olfactory system that includes an artificial neural network/fuzzy filter system equipped with an algorithm for general screening or accurate diagnosis and monitoring, according to need.
U.S. Pat. No. 7,052,854 discloses systems and methods for ex vivo diagnostic analysis of samples of bodily fluids, including exhaled breath. The diagnosis is performed using nanostructure-based assemblies in combination with sensor technology to provide identification of a target analyte/biomarker in a sample.
US Patent Application No. 2007/0048180 discloses a breath analyzer comprising nano-electronic sensors for detecting analytes in human breath. The analyzer contains an integrated multivalent monitor system equipped with a microprocessor capable of analyzing measurements and storing measurement history. The system allows for two or more analytes to be measured in breath samples, and is directed to monitor pulmonary conditions such as asthma.
The analysis of organic vapors in exhaled breath was performed using an array of four polymer-coated surface acoustic wave (SAW) sensors and a thermally desorbable adsorbent pre-concentrator for rapid breath analysis (Groves et al., Analy. Chimi. Acta., 1998, 371, 131-143). The adsorbent used in the pre-concentrator was found critical for achieving adequate sensitivity and further compensate for the high background of water vapors.
Recent studies of gas-chromatography linked to mass-spectrometery (GC-MS) showed that several breath analytes appear to be elevated in instances of lung cancer. Additionally, these analytes are usually detected in distinctive mixture compositions indicative of lung cancer.
Lung cancer was detected by analyzing the breath of patients using a GC column coupled with arrays of polymer-coated surface acoustic wave (Yu et al., Proc. IEEE Sens., 2003, 2, 1333-1337). An artificial neural network has been used, instead of an alveolar gradient, in order to separate the samples into healthy and diseased populations. Using a breath pre-concentrator prior to the GC column, a group of volatile organic compounds characteristic of lung cancer was identified.
In another study, an array of coated quartz crystal microbalance (QCM) sensors was used to detect lung cancer in patients immediately prior to surgical procedure for removing the tumor (Natale et al., Biosens. Bioelectron., 2003, 18(10), 1209-1218). The measurements used the QCM array with no pre-concentration of the breath. The analytes detected by the QCM showed a unique signature.
Recently, sensor arrays of carbon nanotubes coated with different non-polymeric organic layers were shown to be highly potent for the diagnosis of lung cancer via breath samples (Peng et al., Nano Lett., 2008, 8, 3631-3635 and Peng et al., Nano Lett., 2009, 9(4), 1362-1368, published after the priority date of the present application), the contents of which are incorporated by reference herein in their entirety as if fully set forth herein.
In yet another study, carbon nanotubes functionalized with self-assembled sponge like structures of discotic hexa-peri-hexabenzocoronene (HBC) derivatives increased the sensitivity as well as selectivity of the carbon nanotubes to specific VOCs indicative of cancerous breath (Zilberman et al., Langmuir, 2009, 25(9), 5411-5416, published after the priority date of the present application), the contents of which is incorporated by reference herein in its entirety as if fully set forth herein.
There still remains an unmet need for apparatuses and methods for detecting minute quantities of analytes released from human breath with high sensitivity. Furthermore, the advantages of these apparatuses and methods would be particularly striking for measuring mixtures of volatile organic compounds indicative of various types of cancers.

Carbon nanotube structures in sensor apparatuses for analyzing biomarkers in breath samples

Technion Research and Development Foundation Ltd.


Techniques for processing audio data are described. An example of a computing device includes a microphone array to generate audio data and a camera. The computing device is to receive video data from the camera and identify a beam forming target based, at least in part, on the video data. The computing device also includes a beam former to process the audio data to aim a beam of the microphone array at the beam forming target.

When recording audio, background noise may tend to interfere with the suitable recording of the intended audio source. To reduce background noise, beam forming may be used to amplify the audio signal corresponding with the intended audio source while diminishing background noise.
The same numbers are used throughout the disclosure and the figures to reference like components and features. Numbers in the 100 series refer to features originally found in FIG. 1; numbers in the 200 series refer to features originally found in FIG. 2; and so on.


The invention relates to apparatus comprising a container, such as a silo or hopper for storing and distributing flowable material, to receptacle such as an I.B.C. via an outlet, there being a device adapted for removing and replacing a closure of the receptacle, particularly on passage of material into the receptacle.. In a preferred arrangement, outlet of the silo is normally obturated by a conical (or frusto-conical) closure device which is raised and lowered by pneumatic devices in the form of an air bellows mounted in the silo on a frame. The bellows is within a cone, and there is also a device for oscillating the cone, such as a vibrator, in order to obviate bridging or rat-holing of material during discharge. The bellows is housed within a hygienic cover for cleanliness. The cone is connected with the bellows via a connector which may be in the form of a telescopic tube. The tube passes through the apex of the cone and is secured thereat as by a nut and bolt arrangement. The tube passes down interiorly of the cone and terminates in a suction cup such as a rubber cup, the cup itself being generally conical and its periphery being generally in a plane which also contains the periphery of the cone. There is a vacuum line connecting the interior of the tube, and hence the interior of the cup, with a vacuum source.

The invention relates to apparatus for handling flowable material, and to a system incorporating same.
Containers such as bulk storage vessels, silos, hoppers and the like are often used for handling and storing flowable material such as powder or like particulate material, though the material could be a liquid, slurry or similar.
Such material is often required to be transferred from the container through an outlet thereof to a receptacle such as an intermediate bulk container (I.B.C.), for further handling. Where the material can contaminate the surrounding environment as well as the container and receptacles themselves, complicated, and thus expensive, equipment has to be utilized in removing a closure or lid of a charging aperture of the receptacle. Even where contamination is not of such import, it is still a time-consuming process to ensure correct opening (and replacing) of the lid

Apparatus for transferring flowable material to a receptacle and system incorporating the same


A cross-differential amplifier is provided. The cross-differential amplifier includes an inductor connected to a direct current power source at a first terminal. A first and second switch, such as transistors, are connected to the inductor at a second terminal. A first and second amplifier are connected at their supply terminals to the first and second switch. The first and second switches are operated to commutate the inductor between the amplifiers so as to provide an amplified signal while limiting the ripple voltage on the inductor and thus limiting the maximum voltage imposed across the amplifiers and switches.

High efficiency saturated amplifiers and switching power amplifiers are known in the art. One drawback with the use of such amplifiers is the high peak voltages relative to the dc supply that the active devices must withstand in these modes of operation. In order to improve the gain, switching speed, and on-resistance of transistors, the breakdown voltage of the device is usually reduced. This tradeoff is exhibited by all modern semiconductor device technologies, including but not limited to field-effect transistors (FET), bipolar junction transistors (BJT), heterojunction bipolar transistors (HBT), high electron mobility transistors (HEMT), metal-semiconductor field-effect transistors (MESFET), metal-oxide semiconductor field effect transistors (MOSFET), and junction field-effect transistors (JFET). The effect is also independent of the semiconductor material system from which the devices are constructed, including but not limited to gallium arsenide (GaAs), indium phosphide (InP), silicon-germanium (SiGe), and silicon (Si) processes such as silicon bipolar (Si BJT), complementary metal oxide field effect transistor (CMOS) processes, and silicon-on-insulator (SOI) technologies.
In high efficiency switching amplifiers, such a reduction in breakdown voltage can be problematic. Unlike many applications in which the maximum voltage seen by any device is typically limited to the dc voltage of the power source, high efficiency switching amplifiers such as class E, class F, class inverse-F, current-mode class D and class E/F can require that the peak voltage seen by the devices be several times the dc supply. Class F, for instance, can require a peak voltage at least twice the supply voltage, whereas class E can require the device to withstand over 3.5 times the supply voltage without breaking down.
This high peak voltage relative to the dc power supply voltage applied results from the use of an inductor to connect the active device to the dc supply voltage. FIG. 1 is a diagram of a generalized circuit topology typically used in saturated and switching amplifiers such as class E, class F, and class E/F. The active device is connected to the dc supply through the inductor. Since the dc (or average) voltage drop across any inductor at steady state can be zero, the voltage waveform can have an average voltage equal to the supply voltage. This corresponds to a limitation on the waveform that the average area above the supply voltage and the area below it must be the same. This can be seen in FIG. 2, depicting typical waveforms for a representative switching amplifier, with equal areas above and below the supply voltage shaded.
As can be seen in FIG. 2, the active device spends a significant portion of its time in a low voltage state. This is so that the active device can conduct the bulk of its current during this time, thereby reducing the power dissipation in the device, resulting in high efficiency. Unfortunately, this results in a very large area below the supply voltage, necessitating an equally large area above it. Thus the voltage during the times when the switch is not low can be significantly greater than the supply voltage, usually by a factor of two to four.
In a typical CMOS process, for instance, the device breakdown can be less than 6 V whereas the supply voltage is in many cases 3.3 V or higher. With a 3.3 V supply, the class E amplifier can produce waveforms with peak voltage greater than 11V, almost twice that which a CMOS device with 6 V breakdown can tolerate. Thus in this application, the supply voltage can be changed, a more expensive high-voltage process can be used, or a less efficient type of power amplifier with a lower peak voltage can be employed. If the supply voltage cannot be changed, such as if it is coming from a battery or if other circuits on the same supply cannot change their supply voltage, the high peak to supply ratio of the traditional switching amplifiers thus forces a sacrifice in either cost or performance.

This patent is subject to a terminal disclaimer.


Parallel and series tuned resonator circuit topologies are disclosed that permit implementation of narrow band-pass filters having high loaded Q and optimal coupling (for low insertion loss) at frequencies in approximately the 50 MHz to 1 GHz. The topologies employ an additional series capacitor for the parallel-tuned topology, and an additional series capacitor for the series-tuned topology, the capacitors having values counterintuitive to those of skill in the art to produce 6.sup.th order transfer functions out of what were once lower order resonators. Multistrip transmission lines are exploited in the parallel-tuned circuit to reduce the inductance of the resonators to achieve a very large skew in the ratios of inductance to capacitance, to increase Q while maintaining optimal coupling between the resonators of the tuned resonator circuits. Air coils are used in the series-tuned topology to achieve the large inductor values desirable for this topology.

1. Field of the Invention
This invention relates to magnetically coupled resonators, and more particularly to their use as narrow band filters having a high-loaded Q (quality factor) for increased selectivity, low insertion loss and improved out-of-band rejection for advantageous application in high frequency (HF), very high frequency (VHF) and ultra high frequency (UHF) bands, but which are simple and inexpensive to manufacture with a high degree of repeatable accuracy.
2. Background of the Related Art
The processing of broadband multi-carrier signals presents a particularly rigorous and stringent context for signal processing circuitry such as filters. The base-band television signal for example, which has a bandwidth on the order of about 5-6 MHz, is typically mixed with (to modulate) an RF (radio frequency) carrier signal in the range of 50 to 1000 MHz or greater, to achieve frequency division multiplexing (FDM). Applications which require the processing of broadband signals such as the broadcast and reception of television signals can present situations which require filters to pass only a small fraction of the total bandwidth (those frequencies fall within the pass band), while rejecting the rest of the frequencies over the total bandwidth (those falling within the stop-band). This is accomplished using a narrow band-pass filter.
Noise and image signals, as well as various undesired spurious signals, can be injected or generated at various points in processing, and thus band-pass filters are often called upon to reject (i.e. attenuate) out-of-band signals to significantly low levels, depending upon the sensitivity of the application. For example, even signals attenuated up to 60 dB can still be seen in received video transmissions. Thus, it is often critically important that any signals present other than the base-band signal modulated on the desired carrier be sufficiently attenuated. This often requires band-pass filters to be very selective (i.e. ideally passing only that fraction of the total bandwidth that contains the base-band signal of interest), with little or no loss of energy in the pass-band (i.e. low insertion loss), while meeting and maintaining the requisite measure of attenuation for all other frequencies in the stop-band. Moreover, because the fraction of the total bandwidth occupied by base band signals in broadband applications are so small relatively speaking (on the order of 1-2%), such filters must produce the requisite frequency response with a high degree of accuracy and must maintain that response over time (i.e. the response should not drift). Further, they must be relatively immune to RF noise from external sources, as well as from coupling between their own components. Finally, it is always desirable that the filters be inexpensive, and easy to manufacture with a high degree of repeatable accuracy.
There are several known techniques for implementing band-pass filters. As previously discussed, the Q value of a filter indicates its selectivity; a filter's selectivity is defined by how quickly the filter's response transitions from the pass band to the stop band. The higher the Q of a filter, the steeper the roll-off from pass band frequencies to stop band frequencies. Because the input and output loading of a filter affects its Q, a more useful and practical measure is its "in-circuit" or loaded Q (i.e. Q.sub.L). The Q.sub.L of a filter is roughly equal to the reciprocal of the fractional bandwidth of its frequency response, which is typically measured between the points on the response curve that are 3 dB below the peak of the response (i.e. the half-power points of the response). Thus, the Q.sub.L of a filter passing a 1-% fractional bandwidth is roughly 100. Narrow pass band filters for broadband signal processing applications often require a high value of Q.sub.L while exhibiting low insertion loss (i.e. the amplitude of signals in the pass band should not be significantly attenuated), and attenuation off signals in the stop-band should meet the requirements of the applications.
One known technique for implementing band-pass filters involves the use of lumped LC components to produce classical filters based on the technique of low-pass to band-pass transformation. Several variations of topologies can be synthesized for producing desired band-pass filter responses. The shortcomings of such filters are numerous for purposes of processing broadband signals in the VHF and UHF bands, the most serious of which is that the lumped components (particularly the coil inductors) are highly susceptible to parasitic effects at frequencies much above 100 MHz. Moreover, several stages of circuit components must be cascaded together to achieve the complexity of transfer function requisite for a high value of Q.sub.L. Thus, such filters take up valuable space and make their cost of manufacture relatively high.
Another known technique for implementing filters employs helical resonators. Filters employing helical resonators are magnetically and/or capacitively coupled and are capable of producing a response with the high Q.sub.L and low insertion loss requisite for many broadband signal-processing applications. They are not, however, suitable for frequencies much below 150 MHz, because very large inductor values would be required for the resonators below that frequency. Such inductors are impractical or impossible to construct. Moreover, even at higher frequencies they are rather large mechanical structures (they require shielding both for proper operation and to reduce susceptibility to RF noise), which makes them relatively expensive to manufacture (even in high volumes). They also are highly susceptible to environmental shock and drift, and they typically require an adjustment in value during the manufacturing process to make sure that they resonate accurately at the proper frequency.
Yet another known technique for building band-pass filters employs magnetically and/or capacitively coupled dielectric resonators, implemented either as cylindrical coaxial transmission lines, or as printed strip transmission lines sandwiched in between two ground plane shields. These resonators are short-circuited transmission lines, and as such are exploited for their ability to resonate at a particular frequency as a function of their length relative to the wavelength of the transmitted input signal (the length of the line is typically .lambda./4 for the wavelength .lambda. of the resonant frequency). Such resonators are capable of producing high Q.sub.L values to achieve responses having the fractional bandwidth characteristic requisite for many broadband signal-processing applications (i.e. 1-2%). Because the trace length increases as the desired resonant frequency decreases, however, such resonators are not suitable for anything other than UHF (i.e. between about 400 MHz and several GHz). They become cost prohibitive for HF and VHF applications because transmission line lengths increase to a prohibitive size. Moreover, their length must also be trimmed at production to ensure that they resonate at the requisite frequency.
Another well-known circuit topology for producing a band-pass filter response is that of the magnetically coupled double-tuned resonant circuit. Band-pass filters so implemented are the least expensive to manufactured relative to the other various prior art techniques discussed herein (they can be manufactured for a few cents each). Implementations of such filters heretofore known have been unable to achieve the large Q.sub.L values necessary to produce responses having small fractional bandwidths and low insertion loss requisite of many applications such as broadband signal processing (they have typically achieved no better than about 15% fractional bandwidth or greater). The reasons for their shortcomings in such applications will be apparent to those of skill in the art in view of the following discussion.
The generic topology of a series double-tuned circuit 10 is illustrated in FIG. 1a, and that of a parallel double-tuned circuit 100 is illustrated in FIG. 1b. The series tuned circuit has an input resonator circuit 12 that is magnetically coupled to an output resonator circuit 14. Likewise, the parallel tuned circuit 100 has an input resonator circuit 120 magnetically coupled to an output resonator circuit 140. The input resonators 12, 120 are coupled to an input source modeled by sources V.sub.S 18, 180 and associated source impedances R.sub.S 16 and 160 respectively. The output resonators 14, 140 are coupled to an output load impedance modeled by resistor R.sub.L 15, 150 respectively.
The input and output resonators 12, 14 of the series tuned circuit 10 are formed as a series connection between lumped series capacitors C.sub.S1 and C.sub.S2 13 respectively, and inductors L.sub.1 17 and L.sub.2 19 respectively. The two series tuned resonators 12, 14 and the two parallel tuned resonators 120, 140 are magnetically coupled as a function of the physical proximity between their inductors, whereby a mutual inductance M 21 is created. M=kL.sub.1 +L .multidot.L.sub.2 +L , where k is the coupling coefficient which has a value commensurate with the geometry and physical proximity (and thus reflects the percentage of total coupling) between the two resonators. The closer in proximity the two inductors 17, 19 or 170, 190 are, the greater the value of k and therefore the greater the mutual inductance between the resonators; likewise, the further they are apart, the lower the degree of coupling and thus the lower the value of k.
The parallel double-tuned circuit 100 is the theoretical dual of the series double-tuned circuit 10, and thus operates quite similarly. The resonators 120, 140 of the parallel tuned circuit 100 are formed as a parallel connection between lumped capacitors C.sub.P1 110 and C.sub.P2 130, and inductors L.sub.1 170 and L.sub.2 190 respectively. The parallel tuned resonators 120, 140 are also magnetically coupled as a function of the physical proximity between their inductors, whereby a mutual inductance M 210 is created. The mutual inductance of the parallel tuned circuit is given by the same equation, M=kL.sub.1 +L .multidot.L.sub.2 +L , with its value of k dictated by the same considerations.
FIG. 2 illustrates three typical responses of a double-tuned resonant circuit (either series or parallel), for different values of the coupling coefficient k. Response 22 is obtained when the two resonators are critically coupled, which is the point at which the tuned resonator exhibits minimum insertion loss with average selectivity at the resonant frequency. Response 24 illustrates the response of the double-tuned resonators 10 and 100, when their input, output resonators are under-coupled. This occurs for values of k approaching zero. As the resonators are moved further apart, the value of its QL increases (the fractional bandwidth decreases) but the insertion loss also increases, which is not desirable. Response 26 occurs when the two inductors are so close together they become over-coupled (i.e. k approaches a value of 1). Response 26 is characterized by two maxima on either side of the resonant frequency, and the circuit exhibits its lowest Q.sub.L value (and thus its largest fractional bandwidth). From these responses, it can be seen that there is a tradeoff, for this type of filter, between the maximum attainable Q.sub.L value and insertion loss as a function of the coupling.
The Q.sub.L for a series tuned circuit is roughly determined as the reactance X of the tuned circuit network at the resonant frequency, divided by the load or source impedance coupled to it. Thus, Q.sub.L for the output resonator ##EQU1##
Thus, for a given resonant frequency w.sub.o, one could increase the Q.sub.L by increasing the value of L.sub.2. (Of course, to increase the overall Q.sub.L for the double-tuned resonator, one would do the same for the input resonator 12 by increasing the value of L.sub.1 as well). The problem with this approach is that there are practical limitations on the size of the inductors L.sub.1, L.sub.2 that can be manufactured and implemented at a reasonable cost. Moreover, as the values of L.sub.1, L.sub.2 are increased, the parasitic shunt capacitance associated with a lumped value inductor (typically a coil) degrades the frequency response of the filter at frequencies above 200 MHz. Finally, because the resonant frequency is determined by the equation ##EQU2##
(for the output resonator 14), the value of C.sub.S2 must be reduced commensurately to maintain the value of w.sub.o. There are also practical limitations on how small C.sub.S2 can be built accurately.
FIG. 3 illustrates the series double-tuned circuit 10 of FIG. 1 with values for k, C.sub.S1 11 and C.sub.S2 13, and L.sub.1 17 and L.sub.2 19, designed to push the value of Q.sub.L for the circuit at a resonant frequency of 400 MHz. FIGS. 4a and 4b show the simulated response for the circuit 30 having the indicated component values as shown in FIG. 3. The pairs of values across the bottom of FIGS. 4a and 4b indicate the frequency (in MHz) and attenuation (in dB) values for the points 1-4 as indicated on the response curve. The response as shown in the scale provided in FIG. 4a illustrates the unacceptable performance of the filter at high frequencies for television signal processing applications. The smaller scale provided by FIG. 4b shows the 3 dB fractional bandwidth to be about 16% (and thus the approximate value of Q.sub.L is 6.25). As previously discussed, this is unacceptable for many broadband signal processing applications.
The Q.sub.L for a parallel tuned circuit is roughly determined as the admittance of the network at the resonant frequency, multiplied by the load or source impedance coupled to it. Thus, Q.sub.L for the parallel tuned output resonator 140 is.congruent.w.sub.o.multidot.C.sub.P2.multidot.R.sub.L. Thus, it can be seen that to increase Q.sub.L for the parallel tuned output resonator, one could increase the value of C.sub.P2 and R.sub.L. R.sub.L can't be increased much above 100 ohms, as the signal would tend to be shunted to ground through parasitic shunt elements. Increasing C.sub.P2 requires that L.sub.2 be made very small. To manufacture lumped inductors on the order of 5 nH with acceptable accuracy is very difficult, as such inductors are very sensitive to geometric variation, especially longitudinally. Further more, obtaining and maintaining proper coupling between such small coils on a repeatable basis is nearly impossible, primarily due to high sensitivity of the coupling coefficient to dimensional variations of a small gap between the two coupled coils (because the coils must be dimensionally small for small inductance values, the gap between them must also be small to achieve the desired coupling). The requisite small gap magnifies sensitivity to small dimensional variations. Such component and dimensional variations cannot be tolerated when fractional bandwidths on the order of 1% are required.
FIG. 5 illustrates the parallel double-tuned circuit 100 of FIG. 1 with values for k, C.sub.P1 110 and C.sub.P2 130, and L.sub.1 170/L.sub.2 190, with an L to C ratio designed to push the value of Q.sub.L for the circuit at a resonant frequency of 400 MHz. FIGS. 6a and 6b show the simulated response for the circuit 50 having the indicated component values as shown in FIG. 3. The pairs of values across the bottom of FIGS. 6a and 6b indicate the frequency (in MHz) and attenuation (in dB) values for the points 1-4 as indicated on the response curve. The response as shown in the scale provided in FIG. 6a illustrates the unacceptable performance of the filter in the stop-band, even though it operates more symmetrically at high frequencies relative to the series tuned circuit 30 of FIG. 3. Even though the coils values used in this example of the prior art are being pushed to the limit, the bandwidth of this filter is still not narrow enough for many applications. The smaller scale provided by FIG. 6b shows the 3 dB fractional bandwidth to be about 15.5% (and thus the approximate value of Q.sub.L is 6.45. As previously discussed, this is unacceptable for many broadband signal processing applications that require fractional bandwidths of 1 to 2% (i.e. Q.sub.L values in the 50 to 100 range).
Thus, those of skill in the art will recognize the need for band-pass filter circuits that provide characteristics required for many broadband signal processing applications over bandwidths spanning about 50 to 1000 MHz. Those characteristics are namely high Q.sub.L values to provide high selectivity and therefore small fractional bandwidths, high attenuation in the stop-band, low insertion loss in the pass-band but which can be manufactured as cheaply and repeatably as the tuned resonator circuits of the prior art.

Magnetically coupled resonators for achieving low cost narrow band pass filters having high selectivity, low insertion loss and improved out-of-band rejection


In an internal combustion engine, inside of an engine exhaust passage, a hydrocarbon feed valve (15) and an exhaust purification catalyst (13) are arranged. The concentration of hydrocarbons which flows into the exhaust purification catalyst (13) is made to vibrate by within a predetermined range of amplitude of a 200 ppm or more and within a predetermined range of period of 5 second or less, whereby the NOx which is contained in exhaust gas is reduced at the exhaust purification catalyst (13). At this time, the nitrogen-containing intermediate which is produced in the NOx reduction process is exhausted from the exhaust purification catalyst (13). An intermediate purification catalyst (14) for removal of the exhausted nitrogen-containing intermediate is arranged downstream of the exhaust purification catalyst (13).

Known in the art is an internal combustion engine which arranges, in an engine exhaust passage, an NOx storage catalyst which stores NOx which is contained in exhaust gas when the air-fuel ratio of the inflowing exhaust gas is lean and which releases the stored NOx when the air-fuel ratio of the inflowing exhaust gas becomes rich, which arranges, in the engine exhaust passage upstream of the NOx storage catalyst, an oxidation catalyst which has an adsorption function, and which feeds hydrocarbons into the engine exhaust passage upstream of the oxidation catalyst to make the air-fuel ratio of the exhaust gas flowing into the NOx storage catalyst rich when releasing NOx from the NOx storage catalyst (for example, see Patent Literature 1).
In this internal combustion engine, the hydrocarbons which are fed when releasing NOx from the NOx storage catalyst are made gaseous hydrocarbons at the oxidation catalyst, and the gaseous hydrocarbons are fed to the NOx storage catalyst. As a result, the NOx which is released from the NOx storage catalyst is reduced well.

Exhaust purification system of internal combustion engine


A method of controlling inkjet printing including firing at least one fluid drop from an inkjet printhead at one or more firing frequencies; measuring a parameter of said fired at least one fluid drop to determine a present firing performance of the inkjet printhead at each of the one or more firing frequencies; determining if the present firing performance at any of the one or more firing frequencies is different from a predetermined firing performance; and changing, in accordance with a difference between the present firing performance and the predetermined firing performance, and for at least one of the one or more firing frequencies, a fluid quantity parameter for subsequent fluid drops to be fired by the printhead.

Inkjet printing mechanisms fire drops of ink onto a print medium to generate an image. Such mechanisms may be used in a wide variety of applications, including computer printers, plotters, copiers, and facsimile machines. An inkjet printing apparatus may include a printhead having a plurality of independently addressable firing units. Each firing unit may include a fluid chamber connected to a fluid source and to a fluid outlet nozzle. A transducer within the fluid chamber provides the energy for firing fluid drops from the nozzles. In thermal inkjet printers, the transducers are thin-film resistors that generate sufficient heat during application of a voltage pulse to vaporize a quantity of fluid. This vaporization is sufficient to fire a fluid drop.
It is known to control drop quantity for inkjet printing of colored inks, for example by comparing the color of a calibration pattern printed on a print medium with a desired color output. This comparison can be useful to compensate for deterioration of a printhead over time, for example due to kogation. However, such techniques cannot be used for a colorless fluid, such as a pretreatment fluid for improving the fixing of a colored ink to the print medium, to reduce image quality defects.

Hewlett-Packard Development Company, L.P.


A high power solid state semiconductor fiber optic ring laser which includes an optical amplifier diode having its waveguide region oriented at an angle between Brewster's angle and 0.degree. with respect to the normal to the diode cleavage planes together with antireflection coatings disposed on the diode cleavage planes. The optical amplifier diode being optically coupled with the ends of an optical fiber.

The present invention generally relates to lasers, and more particularly is concerned with fiber optic semiconductor ring lasers.
In recent years, the aviation industry has been involved in a seemingly never ending quest to enhance the performance of modern avionics. One particular area in which avionics engineers have expended much effort and achieved many successes, is in the substitution of completely electro-optical systems for the conventional mechanical systems of the past. A prime example of these efforts is in the area of inertial navigation systems.
Mechanical gyroscopes have been, for years, a central component to any inertial navigation system. Recently, the helium neon (He-Ne) ring laser gyroscope has replaced many of the mechanical gyros. However, these He-Ne ring laser gyros have disadvantages in cost and size, among others. With the recent advent of the global positioning system, which uses satellite communications to assist inertial navigation systems, the fiber optic ring laser gyroscopes are becoming more attractive.
Numerous ring lasers which incorporate optical fibers have been used in the past for several applications. One example of a ring laser which uses a fiber optic resonant ring is described in U.S. Pat. No. 4,405,236, entitled "Semiconductor Ring Laser Apparatus", which was issued on Sept. 20, 1983 to Mitsuhashi et al., which patent is incorporated herein by this reference.
While the Mitsuhashi apparatus is capable of producing a ring laser, it does have several disadvantages. A major problem with this approach is that it has a reduction in internal diode reflections, but only at the cost of the overall performance of the laser. The Mitsuhashi design utilizes a diode with its waveguide region being oriented at Brewster's Angle with respect to the normal to the diode cleavage plane. Also at this angle with respect to the cleavage plane is each end of the fiber optic ring. The use of Brewster's Angle was intended to reduce the reflections from within the diode and thereby allowing for the fiber ring to act as the resonant cavity. However, when Brewster's Angle is used, the overall power of the laser is often less than optimal.
Consequently, a need exists for the improvement in fiber optic ring lasers which reduce internal diode reflections, while still providing a high degree of power out of the ring laser.

Solid state fiber optic semiconductor ring laser apparatus