There is disclosed a whispering gallery oscillator for producing a low noise microwave output signal. The whispering gallery oscillator comprises a dielectric resonator 1 comprising paramagnetic ions 34. A pump signal 5 is coupled by a coupler 7 to the paramagnetic ions to excite 19 the paramagnetic ions 34. The paramagnetic ions 34 decay 17 and thus excite a whispering gallery mode 15 of the dielectric resonator 1. The whispering gallery mode 15 is coupled by a coupler 10 to give an output signal 9. The whispering gallery mode 15 has a high Q (Q≈107) resonance. The high Q resonance results in a low-noise output signal 9. The paramagnetic ions 34 act as a MASER that is distributed about the dielectric resonator 1. In some embodiments, the whispering gallery oscillator may be locked to a frequency standard.

Oscillators are vital elements of clocks and have many important applications in telecommunications, remote sensing (e.g. radar), and signal processing. An oscillator is a device that supplies a measurable output. This output can take, for example, the form of the deflection of a mechanical dial or the voltage across a pair of electrical terminals. An oscillator regulates the value of its output as a function of time such that it is periodic (with respect to time). The inverse of the output's temporal period is known as the oscillator's frequency.
Performance, Requirements, and Fundamental Anatomy of an Oscillator
Performance:
An oscillator's fitness of purpose, with respect to any particular application, is determined by the extent to which its frequency remains constant as time progresses. Oscillators that exhibit the most stable output frequencies are the ones, and the only ones, that can enable the most exacting applications, and they are thus valuable.
A mathematical phase can be associated with an oscillator's output, where an increase of 2π in the value of this phase is associated with each new period of the oscillator's output. The phase can be notional compared to that of a corresponding ideal oscillator, whose phase accumulates perfectly linearly with respect to time. The difference in the phases of these two oscillators as a function of time, particularly when transformed into the so-called Fourier domain, provides an alternative way of viewing and characterising the real oscillator's frequency stability.
Several established measures of performance, well known to experts in the art of oscillator characterization, are used to quantify an oscillator's frequency stability. These measures have been reviewed by Rutman and Walls ['Characterization of Frequency Stability In Precision Frequency Sources', J. Rutman and F. L. Walls, Proceedings of the IEEE, vol. 79, pp. 952-960 (1991)] and by Stein ['Frequency and Time—Their Measurement and Characterization', S. R. Stein, pp. 191-416, in 'Precision Frequency Control', edited by E. A. Gerber and A. Ballato, Academic Press, New York (1985)], as well as in many other technical articles and monographs.
Two Measures of performance that have gained particularly wide industrial acceptance are:    (i) The oscillator's 'fractional frequency stability' defined mathematically as its square-root Allan variance, also known as its Allan deviation; this measure is a function of the temporal sampling interval.and    (ii) The oscillator's 'phase-noise spectral density', defined mathematically as the one-sided spectral density of the oscillator's phase fluctuations (with respect its ideal equivalent); this measure is a function of the offset frequency with respect to oscillator's average, or so-called 'carrier' frequency.
Note that each of these two measures of performance is a curve, i.e. a function whose value varies with the value of the function's argument, as opposed to being a single numerical value. With either of these measures, for a given value of its argument (i.e. for a given value of the temporal sampling interval in the case of the fractional frequency stability, or for a given value of the offset frequency, in case the phase-noise spectral density), the lower the measure the greater the oscillator's frequency stability. An oscillator of high performance with regard to it frequency stability is one that exhibits a low fractional frequency stability over a range of temporal sampling intervals or one that exhibits a low phase-noise spectral density, over a range of frequency offsets.
Requirements:
Beyond its frequency-stability performance, an oscillator has various other properties that can affect its utility in particular applications. For a start, an oscillator has a certain mass and occupies a certain amount of space. An oscillator requires the supply of various resources to it and also the keeping of it in an environment whose qualities should allow the oscillator to attain the level of frequency stability that is required of it. All oscillators require both an adequate supply of power and an adequate means of discharging waste heat. Certain oscillators can only operate over limited ranges of temperature. Many other environmental parameters can affect the oscillator's operation, and fluctuations in them may affect the oscillator's frequency stability. These parameters include: the magnitude and direction of either ambient or deliberately applied electric or magnetic fields; the humidity; the intensity of ionising radiation to which the oscillator is exposed; mechanical acceleration (caused by the movement/vibration of the platform to which the oscillator is attached) and acoustic noise.
Fundamental Anatomy:
Every oscillator contains, in conjunction with peripheral supporting equipment, the following three core functional elements:    (A) a means of defining a particular (absolute) frequency; this 'frequency reference' is typically associated with a 'resonance', as exhibited by some or other form of 'resonator';    (B) a means for ensuring that the oscillator oscillates at the reference frequency; this process of regulation or 'locking' is typically achieved through some of other form of 'interference' or 'feedback'.    (C) a means of sustaining the oscillator's oscillation (to prevent it from decaying through unavoidable dissipative processes); this energy-supplying function is typically accomplished by some or other form of 'amplifier'.
The frequency reference associated with any individual oscillator need not be perfectly reproducible but it should, by definition, be sufficiently constant with respect to time. This in turn requires that those physical properties of the resonator whose resonance defines the frequency reference should remain sufficiently constant with time. The present invention proposes a new type of oscillator with operating frequencies in or around the broadly interpreted 'microwave' region, i.e. between a few hundred MHz and several hundred GHz. Here, to provide some technological context for the invention, and to appreciate its significance, some relevant existing microwave oscillators and associated components are briefly reviewed.
Passive Electromagnetic Resonators and Whispering-Gallery Modes
Many successful designs of microwave oscillator incorporate a passive electromagnetic (e.m.) resonator to supply the frequency-reference function [(A) above], where the resonator supports one or several resonant, electromagnetic modes exhibiting high quality factors. Here, the resonator functions as an essentially passive, linear (either 1- or 2-port) device, as can be modelled with scattering parameters (e.g. S21). Because of their regular mention within the descriptions that follow, the quality factors of e.m. modes shall often be denoted simply by the (italicised) letter 'Q'. The frequency and Q of an e.m. mode is determined by the dimensions, shape and electromagnetic properties (viz. electric permittivity, magnetic susceptibility, surface conductance . . . ) of the resonator's constituent materials. The oscillator's frequency is regulated, by one of several available means, to lie at or close to the centre frequency of a particular high-Q mode. One says that the oscillator 'runs on', or is 'locked to' the mode. Besides the resonator itself, the oscillator requires a means of amplification to sustain its oscillation and also a means for locking its operating frequency to the mode on which it is intended to run. The most stable microwave oscillators based on high-Q resonators were recently reviewed by A. G. Mann. ['Ultrastable Cryogenic Microwave Oscillators', A. G Mann, pp. 37-66, in 'Frequency Measurement and Control, Advanced Techniques and Future Trends', Edited by A. N. Luiten, Topics in Applied Physics Vol. 79, Springer-Verlag 2000).] Many (though not all) of them are locked to electromagnetic mode of so-called 'whispering-gallery' type.
Without any further associated qualifications, the word 'dielectric', is used below to denote materials that have a complex electric permittivity, at the frequency of the electromagnetic wave to which the material is exposed, whose real part is greater than that of free space and whose complex part is several orders of magnitude smaller than its real part; in other words, a dielectric is, by default, a 'good' one, in the sense of it having a low loss tangent. Mica, silica, rutile, polystyrene and p.t.f.e. are examples of such dielectrics at microwave frequencies. A sinusoidal electromagnetic wave is characterized by its associated wavelength, λ, which is in general a function of the wave's frequency, polarization and direction of propagation. Through unbounded regions of a (good) dielectric, such a wave can propagate over distances equal to many wavelengths before suffering significant attenuation.
The external surface of a dielectric body that is surrounded by free space defines an extended two-dimensional electromagnetic interface. If the body's form is suitably convex, this interface can support a distinct class of electromagnetic waves known as 'whispering gallery' (henceforth 'WG') waves or modes, as were first analysed by Lord Rayleigh in the case of analogous acoustical systems ['The problem of the whispering gallery', J. W. S. Rayleigh, Philosophical Magazine, Vol. 20, pp. 1001-4 (1910)]. The WG waves flow tangential to the interface and are predominantly confined to the dielectric side of it, in a layer of thickness √{square root over (λR)} that lies immediately below the surface; here, λ is the wavelength of the equivalent freely-propagating wave in an infinite region of the same dielectric and R is the surface's appropriate radius of curvature. The mechanism of confinement can be considered as the radio- and/or microwave-frequency equivalent of what in ray optics is known as 'total internal reflection'. Within the ray-optics description, the inverse sine of the WG wave's angle of incidence at the interface is always greater than the dielectric's refractive index (equal to the square root of the ratio of the dielectric's relative electric permittivity to that of free space). Since no corresponding transmitted (refracted) wave propagating in free space can satisfy Snell's law, the WG wave is totally reflected back into the dielectric. Outside of the dielectric structure, the wave is evanescent as opposed to propagating, where its amplitude (and also its field-energy density) decays exponentially with distance away from the interface. All these features are most pronounced when the wavelength λ is small compared to the convex dielectric body's characteristic radius of curvature R. If the dielectric body is compact or at least 'closed', in the sense that it confines WG waves to run in closed loops, then multiple interference will select a discrete spectrum of frequencies to each of which a particular whispering-gallery standing wave or 'mode' is associated.
Finite dielectric bodies whose geometric shapes have rotational symmetry can be made relatively easily with commonly available tools and established methods of fabrication. Most work on electromagnetic WG-modes to date has concerned dielectric bodies with cylindrical symmetry in the form of spheres, or cylinders, or more general 'solids of rotation' such as rings or toroids. Such dielectric bodies will support whispering-gallery modes that exhibit (discrete) rotational symmetry. The rotational symmetry significantly aids, moreover, in the mathematical analysis and electromagnetic modelling of these WG modes as they can then be quite accurately represented in terms of special functions (such as Bessel functions and their 'modified' variants) or at least relatively compact series-expansions thereof. However, as pointed out below, rotational symmetry, though often convenient, is not strictly necessary for supporting or analysing WG-modes.
If the electric permittivity tensor of the dielectric material out of which such a body is made is anisotropic, as is the case with many crystalline dielectrics whose losses are desirably low, yet the tensor exhibits a rotational symmetry about a particular axis, as is also often the case with the same, this axis can be oriented parallel to the body's geometric axis of rotational symmetry such that the electric permittivity of the space within and about the dielectric body exhibits continuous rotational symmetry. The requisite alignment of the crystal can be accomplished by either viewing it optically through crossed polarizers or using x-ray diffraction. Monocrystalline sapphire is an example of such a material; its c-axis is oriented parallel to the cylindrical axis of the dielectric body that is comprises it. The dielectric loss of sapphire at microwave frequencies, especially at liquid-helium temperatures, is, moreover, extremely low.
Dielectric resonators incorporating rings, cylinders or pucks of high-purity monocrystalline sapphire have been used in electromagnetic resonators at cryogenic temperatures to support WG modes at microwave frequencies exhibiting unloaded Qs in excess of 109. Microwave oscillators built around such resonators have excellent frequency stability. [See 'Improved cryogenic sapphire oscillator with exceptionally high frequency stability', S. Chang et al, Electronics. Letters 36, 480-481 (2000); 'Cryo-cooled sapphire oscillator with ultra-high stability', G. J. Dick et al, Proceedings of 1998 IEEE International Frequency Control Symposium, pp. 528-533 (1998); 'A cryogenic open-cavity sapphire reference oscillator with low spurious mode density', P.-Y. Bourgeois, IEEE Transanctions on Ultrasonics. Ferroelectrics and Frequency Control, vol. 51, pp. 1232-1239 (2004).] Here, the sapphire resonator is maintained at a temperature lying conveniently above 4.2 Kelvin, where a frequency-versus-temperature turnover point is located. The turnover point can be precisely sat upon through a temperature-control servo loop that employs a resistive heater, with no need to pump on the helium of bath. It should be noted here, and as Bourgeois et al explicitly have demonstrated, a reflective electromagnetic shield or cavity placed around the dielectric ring is not necessary for attaining attractively high Qs.
The whispering-gallery modes can be classified through their associated electromagnetic field configurations. Assuming rotational symmetry, the modes can be broadly divided into two classes: quasi-transverse-magnetic (WGH) modes, whose magnetic field lines lie approximately orthogonal to the ring's cylindrical axis, and quasi-transverse-electric (WGE) modes whose electric field lines lie approximately normal to the mode's cylindrical axis. The azimuthal mode order, 'n', of a WG mode equals the number of full waves made by the mode's field pattern around the perimeter of its supporting ring. The WG mode's axial and radial mode orders, 'a' and 'r', respectively, equal the number of nodes in the mode's field pattern along these two respective directions. A 'fundamental' WG mode has no nodes in either of these two directions. In general, a WG mode can be identified using the notation WGHn,r,a for quasi-transverse-magnetic (WGH) modes and WGEn,r,a for quasi-transverse-electric modes. Without external perturbations (from coupling probes, for example) each WG is doubly degenerate; this degeneracy can be associated with two travelling waves circulating in opposite directions around the ring; alternatively, the doublet can be regarded as comprising two otherwise identical standing-wave WG modes whose azimuthal phase differs by 90°.
In general, the existence and form of a given whispering-gallery mode is robust to isolated defects, such as the odd inclusion or void within the dielectric body, or the odd surface scratch or chipped edge on the body's surface, provided the dimensions of these defects are small compared with the mode's associated wavelength λ (within the dielectric). Though the form of the whispering-gallery mode may remain qualitatively the same, its quality factor Q can be significantly degraded by even small defects through their scattering or absorption of energy from the WG mode.
The vast majority of the electromagnetic WG modes that have been studied to date have been planar in form. Planarity is not strictly essential to WG modes and indeed, non-planar whispering-gallery modes can be supported by finite cylinders and 'bottle-shaped' solids of rotation; as have recently been analysed by Sumetsky. ['Whispering-gallery-bottle microcavities: the three-dimensional etalon', M. Sumetsky, Optics Letters, Vol. 29, pp. 8-10 (2004)].
Furthermore, rotational symmetry of the dielectric body itself is not necessary for supporting 'generalized' whispering-gallery electromagnetic modes, which exhibit the same desirable features concerning confinement, low evanescent leakage and (thus) high Q, as their rotationally symmetric relatives. These generalized WG mode can, moreover, offer significant advantages and design flexibility with regard to controlling how they couple electromagnetically to surrounding structures (such as coupling probes and 'launchers') located at a particular azimuthal positions. S. Ancey et al have, for example, analysed whispering-gallery modes in dielectric bodies of elliptical shape ['Whispering-gallery modes and resonances of an elliptic cavity', S Ancey, A Folacci and P Gabrielli, Journal of Physics A (Mathematical and General), Vol. 34, pp 1341-1359 (2001)].
To support a generalized WG mode, the external surface of the dielectric body should in general contain a closed, convex 'band', where the band's curvature in its 'long' or generalized-azimuthal direction should be small enough at all positions around the band to ensure total internal reflection and (thus) sufficiently suppress evanescent leakage and radiation losses. Furthermore, the variation in the azimuthal curvature around the band needs to be sufficiently limited to avoid anomalously large leakage and losses due to a phenomenon known are 'chaos assisted tunnelling', as for example has been discussed by Nöckel and Stone ['Ray and wave chaos in asymetric resonator optical cavities', J. U. Nöckel, and A. D. Stone, Nature. 385, pp. 45-47 (1997)]. Provided this and related wave-chaotic phenomena (such as so-called 'dynamical localization') are taken in to account, non-rotationally symmetric dielectric bodies that support generalized whispering-gallery modes with both high Qs and advantageous coupling features can be rationally designed and constructed.
Oscillator Loops and Locking Configurations
In connection with (B) stated above, the various means through which an oscillator can be compelled to run at the frequency that is defined by its frequency reference (as embodied by a resonator) are reviewed here.
The simplest resonator-based oscillators comprise a passive resonator [i.e. (A) above] and an amplifier [i.e. (C) above] that are connected directly together by cables in a loop to form what is known as a free-running loop oscillator. The phase-noise performance of such oscillators has been by considered quantitatively by Leeson ['A Simple Model of Feedback Oscillator Noise Spectrum'. D. B. Leeson, Proceedings of the IEEE, vol. 54, pp. 329-330 (1966)] and more recently by Everard in somewhat more detail ['Fundamentals of RF Circuit Design with Low Noise Oscillators', by J. Everard, John Wiley & Sons Ltd. (2001)]. Through such considerations, it is known how a loop oscillator's phase noise depends on its operating parameters. With regard to attaining low phase noise:    (i) the mode supported by the oscillator's resonator should have an unloaded Q that is as high as possible;    (ii) the mode's centre frequency should be as constant as possible;    (iii) the oscillator's feedback power, which flows through the resonator, should be as large as possible;    (iv) the thermal noise generated by the amplifier, as characterized by its finite noise temperature, as well as that generated by other parts of the oscillator's loop, should be as low as possible; thermal noise can be reduced by reducing the temperature;    (v) the flicker noise in the oscillator's sustaining amplifier should be as low as possible;    (vi) the electromagnetic properties of the loop's interconnecting cables, i.e. their phase length and loss, should be as stable as possible—lest fluctuations in the cables come to pull the loop oscillator's frequency.
The optimisation of a practical free-running loop oscillator involves judicious trade-offs between the above parameters. More complicated designs of microwave oscillator, incorporating additional components within various servo loops, attempt to suppress or circumvent the free-running loop oscillator's sensitivities to fluctuations in certain of them.
So-called Pound-stabilized loop (PSL) oscillators [see 'A High Stability Microwave Oscillator Based on Sapphire Loaded Superconducting Cavity', by A. J. Giles et al, Proceedings of the 43rd Annual Symposium on Frequency Control, pp. 89-93 (1989)] have been built around cryogenic sapphire whispering-gallery-mode resonators maintained at cryogenic (often liquid-helium) temperatures within a refrigerator (i.e. a so-called 'cryostat'). Within the bandwidth of such an oscillator's Pound stabilizer, the frequency instabilities that would otherwise be introduced by either the loop amplifier's phase noise and/or by fluctuations in the loop's interconnecting cables, or both, are compensated.
In contrast to the present invention proposed below, it can be remarked that a PSL oscillator is a spatially extended system: at least two microwave lines, each typically greater than one meter in length, are required to connect the cryogenic resonator to the room-temperature section of the oscillator's loop. In reality, each line comprises a series of semi-rigid microwave cables that are connect by feedthroughs between the cryostat's different chambers and sections. To achieve frequency stabilities at the 1×10−14 level, several auxiliary cables and sensors, supporting the control of the resonator's temperature, the so-called Pound servo, and well as loop-power regulation [see, for example, 'Latest results of the U.W.A. cryogenic sapphire oscillator', A. N. Luiten et al, Proceedings of IEEE International 49th Frequency Control Symposium, pp. 433-437 (1995)] are all required to be wired into the cryostat. Pound stabilization does suppress, within a finite bandwidth, the phase (thus frequency) shifts associated with mechanical vibrations and/or temperature fluctuations along the microwave lines. Residual amplitude modulation, causing offsets in the Pound servo's d.c.˜error signal remains a problem, however.
Despite their configurational complexity, PSL oscillators have demonstrated their worth in several demanding applications. They have been used as 'flywheels' for cold-atom frequency standards [see 'A high stability atomic fountain clock using a cryogenic sapphire interrogation oscillator', A. G. Mann et al, Proceedings of 1998 IEEE International Frequency Control Symposium, Pasadena, Calif., USA, pp. 13-17 (1998)], as reference oscillators for (close-in) phase-noise measurements [see 'Microwave Frequency Discriminator with a Cryogenic Sapphire Resonator for Ultra-Low Phase Noise', G. J. Dick & D. G. Santiago, Proc. of 6th European Frequency and Time Forum, Noodwijk, The Nederlands, 17-19 March 1992, pp. 35-39 (1992)], or in fundamental-physics experiments testing, for example, Lorentz invariance [see 'Tests of Lorentz Invariance using a Microwave Resonator', P. Wolf et al, Physical Review Letters, vol. 90, pp. 060402 (2003)].
Solid-State Masers
In the designs of oscillators considered above, the means through which the oscillator's oscillation is sustained [(B) above] takes the form of a conventional microwave amplifier, typically based on semiconductor technology, and typically operating at room temperature. A wholly different means of amplification is Microwave Amplification by Stimulated Emission of Radiation, which is based on atomic (also often described as 'quantum-electronic') principles. The term 'maser' (as both a noun and adjective) shall henceforth be generally used to refer to it; the gerundive 'mas(er)ing' shall also on occasions be used to refer to those physical entities that participate in the 'maser action'—to distinguish them from other's that don't.
The maser phenomemon can be realized in both solid-state systems, where the masering atoms (or ions) reside in condensed matter, and also in more rarified systems, where the masering atoms, in the form of propagating beams or clouds, reside in what is otherwise a vacuum. With regard to the former, solid-state masers have been reviewed pedagogically by Siegman ['Microwave Solid-state Masers', A. E. Siegman, McGraw-Hill (1964)]. Here, the masing medium is a dielectric solid, most often a crystal, containing a distribution (a so-called 'solid dilution') of paramagnetic ions. The frequencies that are associated with transitions between the electronic-spin states, or 'levels', of these paramagnetic ions typically lie in the microwave region (i.e. GHz). With a few exceptions, maser action in solids requires operation at liquid-helium temperatures. The most comprehensively studied and applied solid-state maser system to date has been (artificial) 'pink ruby', that is, crystalline sapphire doped with chromium Cr3+ ions, at a substitutive concentration on the order of 1 part per thousand. This system has been used to realize low-noise, sufficiently wide-band amplifiers for boosting the power of weak microwave signals in telecommunication and astronomy. [See, for example, 'Solid State Masers', N. Bloembergen, pp. 396-429 (Chapter IX) in 'Progress in Low Temperature Physics', edited by C. J. Gorter, vol. 111, North-Holland (1961).]
The strength of a paramagnetic transition between two levels in so-called 'free-spin units' is equal to the ratio, expressed as a fraction, between the said strength and that of the equivalent 'strongly allowed' transition between the upper (m=−½) and lower (m=+½) spin levels of a free electron (S=½), where these two free-spin levels have been separated by an applied static magnetic field. Maser action requires that both its so-called 'pump' and 'signal' transitions are sufficiently strong, i.e. a sufficiently large fraction of a free-spin unit. Except in a few specific systems, which shall be subsequently mentioned, this generally requires that the masering paramagnetic ions be exposed to a d.c. magnetic 'bias' field that splits and mixes, quantum mechanically, the 'zero-field' quantum states that the ions would otherwise have. In addition, the application of a magnetic field of a judiciously chosen strength and orientation (relative to the crystal's axes), allows the maser's signal and pump transitions to be 'engineered' with regard to their strengths, polarizations, and frequencies [see Siegman, already referenced above.]
Dick et al ['Development of the Superconducting Cavity Maser', G. J. Dick and D. M. Strayer, Proceedings of 38th Annual Frequency Control Symposium, pp. 435-446 (1984); 'Ultra-Stable Performance of the Superconducting Cavity Maser', G. J. Dick, and R. T. Wang, IEEE Transactions on Instrumentation and Measurement, vol. 40, pp.174-177 (1991)] developed a 'Superconducting Cavity Maser Oscillator' (SCMO), operating at a temperature near 1.6 K, that exhibited extremely good frequency stability (Allan deviation of 4–5×10−15 for sampling intervals between 1 and 1000 s) and exceptionally low (flicker-) phase noise (−80 dBc/f3, where the frequency f is in Hz). Dick et al's oscillator incorporated a solid-state ruby maser amplifier. This amplifier was maintained at a cryogenic temperature and its operation required an applied d.c. magnetic 'bias' field. With regard to the present proposed invention, it should be, noted that the maser amplifier within the SCMO was physically separated from the oscillator's passive frequency-defining resonator cavity. This physical separation was in fact an essential feature of the SCMO: the resonator's high Q, and hence the oscillator's frequency stability, would have been severely degraded, due to the trapping of magnetic flux in the walls of the oscillator's superconducting cavity (made of lead), had the cavity been exposed to the same magnetic field necessary to bias the ruby crystal. In other words, the amplifier and resonator of Dick et al's SCMO could not operate in the same region of space.
In contrast to a typical embodiment of the Pound-stabilized loop oscillator, however, the SCMO's all-cryogenic loop was relatively more compact (centimeters as opposed to meters) and benefited from the uniformity and stability of temperature provided by its wholly cryogenic environment. By dint of the ruby maser's combined low thermal and low flicker noise, the phase noise exhibited by the SCMO was superior to that which could have been achieved by swapping the maser for a conventional microwave (e.g. 'GaAsFET' or 'HEMT') amplifier, even if operated at a cryogenic temperature. The SCMO offered several other advantages that are shared with the present invention proposed below, which shall be stated in due course.
For completeness, and as shall be relevant to the description of the present invention's first experimental embodiment, it is noted here that there exist a few solid-state systems that enable maser action without the application of a d.c. magnetic bias field. Sapphire that is substitutively doped with Fe3+ ions, or sumarium sulphate doped with Gd3+ ions, provide two examples (there are several others). Such so-called 'zero-field masers'. have been reviewed by Bogle and Symmons ['Zero-Field Masers', G. S. Bogle and H. F. Symmons, Australian Journal of Physics, 12, pp. 1-20 (1959)].
Atomic Maser Oscillators
If the linewidth of a maser's so-called 'signal' transition is sufficiently narrow, the maser action itself can provide the means of realizing an oscillator's frequency reference [(A) above], where the oscillator's output frequency is predominantly determined by the frequency of the said signal transition—acting as the reference—as opposed to the centre frequency of the maser's associated electromagnetic mode, whose linewidth is broader than that of the signal transition, with which the signal transition interacts. This possibility was in fact immediately appreciated upon the very first experimental demonstration of maser action ['Molecular Microwave Oscillator and New Hyperfine Structure in the Microwave Spectrum of NH3', J. P Gordon, H. J. Zeiger and C. Townes, Physical Review, 95, pp. 282-284 (1955)]. In such 'atomic' maser oscillators, frequency stability demands that the linewidth of the maser transition be as narrow as possible, i.e. that the transition's line Q be as high as possible. To reduce 'pulling' of the atomic maser oscillator's frequency, the Q of the electromagnetic mode with which the signal transition interacts should, on the other hand, be as low as is compliant with the conditions required for sustained, above-threshold active maser oscillation, at a reasonable output power. Here, the desired relative linewidths of the signal transition and the electromagnetic mode lie in stark contrast to Dick's SCMO, where the operating electromagnetic mode had a Q (˜109) that was approximately seven orders of magnitude higher than the line Q of the ruby's atomic (i.e. paramagnetic) signal transition (˜102). This fundamental 'reversal of roles' between the atomic maser transition and the electromagnetic resonance was indeed explicitly noted by Dick and Strayer, when comparing the workings of their SCMO to that of the (atomic) hydrogen maser. [See 'The Superconducting Cavity Stabilized Ruby Maser Oscillator', Dick, G. J. and D. M. Strayer, Proceedings of the Fifteenth Annual Precise Time and Time Interval (PTTI) Applications and Planning Meeting, pp. 723-739, (1983).]
When describing maser oscillators, it is often necessary to distinguish between and quantify the various processes through which energy is lost from or gained by an electromagnetic mode. In this regard, is useful to define a mode's 'non-magnetic' Q as the Q that it would have were its interaction with the maser oscillator's paramagnetic ions (somehow) turned off.
As early as 1960, Bloembergen considered the realization of atomic maser oscillators of high frequency stability in solid-state systems (as opposed to those based on atomic/molecular beams). [See 'The Zero-Field Solid State Maser as a Possible Time Standard', in Quantum Electronics. A Symposium, Columbia University Press, New York, pp. 160-166 (1960).] He speculated, perhaps rather optimistically, that certain solid-state maser materials could exhibit paramagnetic transitions with linewidths almost as narrow as those exhibited by atoms in a vacuum. He also pointed out several advantages of those special solid-state systems (viz. the 'zero-field masers' previously mentioned) where maser action is possible without the application of a d.c. magnetic bias field.
Despite Bloembergen's imaginative early conjectures, by far the most successful atomic maser oscillator to date has been the (active) hydrogen maser oscillator, whose masering atoms compose a rarified (state-selected) gas, held within what is otherwise an evacuated bulb. Vanier provides a review ['The Active Hydrogen Maser: State of the Art and Forecast', J. Vanier, Metrologia, vol. 18, pp. 173-186 (1982)]. A conventional hydrogen maser operates at near-room temperature (the temperature of the hydrogen atoms emitted from the system's rf-dissociator is higher though the emitted atoms become quickly thermalized to the temperature of the walls of the maser's storage bulb.) The short-term stability (<100 s) of the hydrogen maser is limited by the low power of its output, which is typical no greater than −90 dBm or 1 pW. Its longer term frequency stability is limited by, amongst other processes, fluctuations in the dimensions of the maser's electromagnetic cavity.
Cryogenic or 'cold' hydrogen masers (CHMs), whose storage bulbs are coated with super-fluid helium, necessarily operate below the so-called lambda-point temperature for liquid helium at 2.17 K. They derive several benefits from their cryogenic operation: (i) lower thermal 'Schawlow-Townes' noise in the maser oscillator, (ii) lower thermal (Johnson) noise imparted by the (now potentially cryogenic) receiving amplifier, and (iii) a reduction in cavity pulling due to the lower thermal expansion, as well as lower change in the dielectric permittivity, of the materials that compose the maser's electromagnetic cavity. These significant benefits come, however, at the cost of other phenomena that significantly affect stability, most notably (the temperature and pressure dependence of) hyperfine spin-exchange, which cannot be compensated through (conventional) so-called 'spin-exchange tuning'.
Fundamental Limits
The fundamental limits on a maser oscillator's short-term frequency stability are well understood by experts in the art of their design and construction. [See, for example, 'Analysis of Fundamental and Systematic Effects Limiting the Hydrogen Maser Frequency Stability', by E. Mattison and R. F. C Vessot, Proceedings of Twenty-first Annual Precise Time and Time Interval (PTTI) Applications and Planning Meeting, Redondo Beach, Calif., USA, pp. 433-444 (1989)].
Thermal noise that is generated within and amplified by the maser oscillator causes fluctuations in the frequency of the maser oscillator's signal output. These frequency fluctuations can be described by their corresponding Allan deviation, as introduced above, which takes the form:
                                          σ            M                                          (            τ            )                          =                              1                          Q              L                                                                                k                                                                                T                  M                                                            2                                P                                                                                τ                                                                        (        1        )            where σM(τ) is the Allan deviation of the fractional frequency fluctuations of the maser oscillator's signal output as a function of the temporal sampling interval, τ; QL is the loaded, non-magnetic Q of the signal electromagnetic mode with which the maser oscillator's signal transition interacts, k is Boltzmann's constant, TM is the absolute temperature of the maser oscillator, and P is the overall rate at which energy is removed or dissipated from the maser signal mode due to both loading (i.e. the coupling out of signal power) and internal electromagnetic losses.
If the power of the maser oscillator's output signal is low, then the noise that is imparted onto it by its so-called 'receiving amplifier', which is generally necessary for boosting the power of the maser oscillator's output signal to a usable level, can adversely affect the frequency stability of the boosted signal. The Allan deviation that describes these additional frequency fluctuations takes the form:
                                          σ            M                                          (            τ            )                          =                              1                          2                            π                                                                                f                M                                          τ                                                                                3                                k                                                                                T                  R                                                B                                            2                                                  P                  0                                                                                (        2        )            where σR (τ) is the Allan deviation of the additional fractional frequency fluctuations at the output of the receiving amplifier as a function of the sampling interval τ, and fM is the frequency of the maser oscillator's signal transition, TR is the so-called 'noise temperature' of the receiving amplifier, B is the effective noise bandwidth of the same, and P0 is the power of the signal delivered to the input of the receiving amplifier from the maser oscillator's output. P0 is some finite fraction of the total power dissipated in the maser's signal electromagnetic mode, P [as introduced in connection with equation (1) above]; the fraction P0/P is typically 0.25—though could be higher or lower depending on the application for which the maser oscillator's operating parameters are optimised. The receiving amplifier may also impart significant frequency flicker (phase) noise onto the boosted signal.
In general, the above two formulae indicate that it is advantageous with regard to frequency stability for the temperature, be it TM or TR (or both), to be low, and for the maser oscillator's operating power, be it P or P0 (or both), to be high. In these two regards, conventional hydrogen masers are limited by both their high operating temperature and, in particular, by their low output power.


Disclosed herein are compositions for use in an energy providing devices and methods of preparing such devices. Also included herein is energy providing devices that comprise a charged compound modified substrate or zwitterion-modified substrate or energy providing devices that comprise an electrolyte that comprises a perhalogenatedphenyl azide charged or zwitterionic compound.

Renewable energy resources such as wind, solar, and hydroelectricity are alternative energy sources to traditional fossil energy. In some instances, the shortcomings of renewable energy include their variability and intermittent power generation. As such, energy storage devices such as batteries, flywheels, and supercapacitors have been used to ensure delivery of continuous and stable energy supplies.

Energy providing devices and applications thereof

The Regents of the University of California


A transmission apparatus in a Multiple Input Multiple Output (MIMO)-based wireless communication system. The transmission apparatus includes at least two antenna groups each having at least two antennas, wherein the antenna groups are spaced apart by a first distance and transmit antennas in each antenna group are spaced apart by a second distance which is shorter than the first distance. A channel coding and modulation unit channel-codes and modulates a desired transmission data stream. A precoding unit precodes the channel-coded and modulated signal separately for each antenna group and each antenna in the same antenna group. Thereafter, a transmission processing unit transmission-processes the output signal from the precoding unit.

1. Field of the Invention
The present invention relates generally to an apparatus and method for transmitting data using multiple antennas in a wireless communication system, and in particular, to an apparatus and method for precoding transmission data in transmitting data using Multiple Input Multiple Output (MIMO).
2. Description of the Related Art
Wireless communication systems have been developed to offer users mobility that allows users to travel freely. A typical example of such wireless communication systems may include a mobile communication system. The mobile communication system converts voice signals into electrical signals, processes them and carries them on a Radio Frequency (RF) thereby providing voice services to users.
With the rapid progress of the communication technology, the mobile communication system is evolving into a high-speed, high-quality wireless packet data communication system providing data services and multimedia services beyond the earlier voice-oriented services. Recently, various mobile communication standards, such as High Speed Downlink Packet Access (HSDPA) and High Speed Uplink Packet Access (HSUPA), both defined by 3rd Generation Partnership Project (3GPP), High Rate Packet Data (HRPD) defined by 3rd Generation Partnership Project-2 (3GPP2), and 802.16 defined by Institute of Electrical and Electronics Engineers (IEEE), have been developed to support the high-speed, high-quality wireless packet data transmission services.
The present 3rd generation wireless packet data communication systems, such as HSDPA, HSUPA and HRPD, use such technologies as an Adaptive Modulation and Coding (AMC) method and a channel-sensitive scheduling method to improve transmission efficiency. With application of the AMC method, a transmitter can adjust the amount of transmission data according to the channel state. That is, in a poor channel state, the transmitter reduces the amount of transmission data to match a reception error probability to a desired level, and in a good channel state, the transmitter increases the amount of transmission data to efficiently transmit a large amount of information while matching the reception error probability to the desired level. With application of the channel-sensitive scheduling-based resource management method, the transmitter, since it selectively services a user with a good channel state among several users, increases system capacity when compared to a method of allocating a channel to a single user and servicing the user using the allocated channel. Such capacity increase is called multi-user diversity gain. In sum, the AMC method and the channel-sensitive scheduling method are methods for applying proper modulation and coding techniques at the most efficient time determined by partial channel state information fed back from a receiver.
To implement the AMC method and the channel-sensitive scheduling method, the receiver should feed back its channel state information to the transmitter. The channel state information fed back by the receiver is called a Channel Quality Indicator (CQI).
Recently, intensive research is being conducted to replace Code Division Multiple Access (CDMA), the multiple access scheme used in the 2nd generation and 3rd generation mobile communication systems, with Orthogonal Frequency Division Multiple Access (OFDMA) for the next generation mobile communication systems. 3GPP and 3GPP2 have started standardizing work on an evolved system using OFDMA. It is known that OFDMA will increase system capacity. In OFDMA, one of several causes of the capacity increase is its capability of performing scheduling in the frequency domain (Frequency Domain Scheduling). Having obtained capacity gain in the communication system using the time-varying channel characteristic through the channel-sensitive scheduling method, OFDMA is expected to obtain higher capacity gain with application of the frequency-varying channel characteristic. However, in order to support frequency-domain scheduling, the transmitter should have channel state information for each frequency. That is, there is a need for CQI feedback for each frequency, causing an increase in a load of CQI feedback.
In the next generation system, active research is being conducted on a MIMO technology based on multiple transmit/receive antennas. The term "MIMO" as used herein refers to a technology for simultaneously transmitting multiple data streams over the same resources using multiple transmit/receive antennas. In the good channel state, transmitting multiple low-modulation order data streams can increase the throughput at the same error probability, compared to increasing the modulation order. In the MIMO terminology, the dimension over which an individual data stream is transmitted is called a layer. As for the layer, a method of separately applying AMC according to the channel state is efficient in increasing system capacity. For example, Per Antenna Rate Control (PARC) is a technology for transmitting a different data stream via every transmit antenna, and its layer is a transmit antenna. Multiple transmit antennas experience different channels, and the PARC technique applies AMC such that a larger amount of data can be transmitted via a transmit antenna having a good channel state and a lesser amount of data can be transmitted via a transmit antenna with a poor channel state.
As another example, in the Per Common Basis Rate Control (PCBRC), its layer is a fixed transmission beam. Therefore, the PCBRC technique transmits a larger amount of data with a transmission beam having a good channel state, and transmits a lesser amount of data with a transmission beam having a poor channel state.
When MIMO is realized using multiple antennas, a precoding method is used to adaptively form a transmission beam according to the channel state. The term "precoding" as used herein refers to an operation in which a transmitter pre-distorts a transmission signal before transmitting the signal via a transmit antenna. If precoding is realized by linear combination, a precoding process can be expressed as shown in Equation (1),x=Es  (1)where "s" is a K×1 vector and denotes a desired transmission signal, and "x" is an M×1 vector and denotes an actual transmission signal. Further, "K" denotes the number of symbols simultaneously MIMO-transmitted over the same resources, and "M" denotes the number of transmit antennas. In addition, "E" is an M×K matrix and denotes precoding. That is, Equation (1) expresses an operation in which a MIMO transmitter with M transmit antennas applies a precoding scheme E when simultaneously transmitting K signal streams.
A precoding matrix E is adaptively determined according to a transmission MIMO channel. However, when a transmitter cannot acquire information on the transmission MIMO channel, it performs precoding according to the feedback information reported by a receiver. Accordingly, a precoding codebook including a finite number of precoding matrixes E is preset between the transmitter and the receiver. The receiver selects a precoding matrix E most preferred in the current channel state from among the precoding codebook, and feeds it back to the transmitter. Therefore, the transmitter performs MIMO transmission by applying this precoding matrix E.
The transmission signal of Equation (1), received over a MIMO channel H, is expressed as shown in Equation (2),y=Hx+z=HEs+z  (2)where "y" and "z" are both an N×1 vector and denote signals and noise signals received at N receive antennas, respectively, and 'H' is an N×M matrix and denotes a MIMO channel. The received signals undergo a reception combining process so that a Signal-to-Interference and Noise Ratio (SINR) of a transmission signal stream for each layer is improved. A signal r that has undergone the reception combining process can be expressed as shown in Equation (3).r=Wy=WHx+Wz=WHEs+Wz  (3)In Equation (3), "W" is an N×N matrix and denotes a reception combining process, and "r" is an N×1 signal vector. Reception techniques such as interference cancellation and Maximum Likelihood (ML) reception can be further performed to normally receive the transmission signal stream of each layer.
The conventional precoding techniques can be classified into two types. First, there is a Discrete Fourier Transform (DFT)-based precoding matrix as a precoding technique designed so as to use a spatial correlation taking into account a linear array antenna whose elements are arrayed at regular intervals. Equation (4) represents a DFT-based precoding matrix E.
                    E        =                  LF          =                                    diag              (                                                ⅇ                                      j                                          ϕ                      1                                                                      ,                                  ⅇ                                      j                                          ϕ                      2                                                                      ,                …                                                                                ,                                  ⅇ                                      j                                          ϕ                      M                                                                                )                        ·                                                            1                                      M                                                  [                                  exp                  (                                                            ±                      j                                                                                2                                                                          π                                                                                (                                                          m                              -                              1                                                        )                                                                                (                                                      n                            -                            1                                                    )                                                                    M                                                        )                                ]                                                              m                  =                  1                                ,                                                                                …                                                                                M                                                                              n                  =                  1                                ,                                                                                …                                                                                ,                M                                                                        (        4        )            In Equation (4), "F" denotes a DFT matrix or an Inverse Discrete Fourier Transform (IDFT) matrix, and "L" denotes a diagonal matrix for adjusting only the phase. A DFT-based precoding matrix, which uses the characteristic that the DFT matrix F forms spatial beams, shows excellent performance in the channel having a high spatial correlation.
Next, there is a Grassmannian Line Packing (GLP) precoding matrix designed in the antenna structure where there is no spatial correlation. The GLP precoding matrix, which improves precoding gain by extending a distance between precoding matrixes, is designed on the assumption that there is no spatial correlation. Although the DFT-based precoding matrix can also be applied to the antenna structure where there is no spatial correlation, it is known that the GLP precoding method has a better performance. Meanwhile, if there is no spatial correlation, applying PARC without precoding can also increase channel capacity. In this case, it is possible to express the precoding matrix as a unit matrix.
The precoding schemes are classified into a Single CodeWord (SCW) scheme and a Multi-CodeWord (MCW) scheme according to the number of coded packets from which multiple signal streams transmitted by the MIMO technique are generated. In the SCW scheme, one codeword is transmitted through a plurality of layers regardless of the number of layers. The MCW scheme transmits one codeword over each of multiple layers. The MCW scheme is advantageous in that a receiver can obtain additional gain through interference cancellation, because the receiver can determine success/failure in decoding of each codeword through Cyclic Redundancy Check (CRC) applied to every codeword. However, the MCW scheme, as it increases the number of transmission codewords, may consume additional resources which linearly increase to apply CRC, and may increase in the complexity of the receiver. A Dual CodeWord (DCW) scheme has been designed as a proposed trade-off for obtaining the rate improvement effect of the MCW scheme while compensating its disadvantages. In the DCW scheme, a maximum of two codewords are transmitted through multiple layers regardless of the number of layers.
FIG. 1 is a diagram illustrating an example of a general SCW scheme-based MIMO (SCW MIMO) transceiver structure. With reference to FIG. 1, a description will be made of a general SCW MIMO transceiver structure.
A desired transmission data stream is input to a channel coding and modulation unit 101 where it is converted into one coded packet signal stream after undergoing a channel coding and modulation process. The signal converted into a packet signal stream is input to a demultiplexer 103 for MIMO transmission. The demultiplexer 103 demultiplexes the signal stream into K signal streams. The K signal streams demultiplexed in this way are linearly converted into M signal streams to be transmitted via associated transmit antennas by means of a precoder 105. The precoding process performed in the precoder 105 can be considered that K signal streams are processed such that they are transmitted through different transmission beams. The precoded M signal streams are transmitted via associated transmit antennas 109a through 109m by means of associated transmission processors 107a through 107m. The transmission processors 107a through 107m each includes not only the process of making CDMA or OFDMA signals but also the filtering and RF processing process performed in each antenna. The transmitted signals are received at N receive antennas 111a through 111n, and the signals received at the receive antennas 111a through 111n are restored to baseband signals by means of associated reception processors 113a through 113n. The reception-processed signals are output to a reception combiner 115 where they undergo signal combining separately for each transmit antenna. The signals combined separately for each transmit antenna are input to a multiplexer 117 where they are restored to the desired original transmission signal stream. Then the restored signal undergoes demodulation and channel decoding in a demodulation and channel decoding unit 119, restoring the desired original transmission data stream.
SCW MIMO is characterized in that since the transmitter generates multiple transmission signal streams by applying one channel coding and modulation unit 101, it only needs to receive one CQI feedback. However, the number of MIMO transmission signal streams, i.e., the number K of transmission MIMO layers, should be adjusted according to the channel state. The number K of transmission MIMO layers will be referred to herein as 'Rank'. Therefore, a feedback of SCW MIMO is composed of one CQI representing a channel state of transmission MIMO layers and the number Rank of transmission MIMO layers.
FIG. 2 is a diagram illustrating an example of a general MCW scheme MIMO (MCW MIMO) transceiver structure. With reference to FIG. 2, a description will now be made of an example of a general MCW MIMO transceiver structure.
In MCW MIMO, unlike in SCW MIMO, different coded packet signal streams are transmitted through individual MIMO layers. Therefore, a desired transmission data stream is input to a demultiplexer 201 where it is demultiplexed into as many signal streams as Rank. The demultiplexed signal streams are converted into signal streams associated to corresponding MIMO layers by means of different channel coding and modulation units 203a through 203k. In the following transmission process, the signal streams are transmitted using the same structure as that of the SCW MIMO transmitter. That is, the signals to be transmitted via M transmit antennas are generated by means of the precoding process and the transmission process for individual transmit antennas. It should be noted that the same or similar elements are denoted by the same reference numerals throughout FIGS. 1 and 2. However, in FIG. 2, channel coding and modulation processes 203a through 203k can be implemented with one channel coding and modulation unit. In this case, FIG. 2 is similar to FIG. 1 in terms of the channel coding and modulation process.
The MCW MIMO reception process is also similar to the process performed in the SCW MIMO receiver in several steps just after reception. That is, the MCW MIMO receiver and the SCW MIMO receiver use the same reception process of converting signals received via individual antennas into baseband signals, and the reception combining process of combining the received signals into transmission signals for individual layers. Therefore, the same elements are denoted by the same reference numerals. The restored signals may include mutual interference. The MCW MIMO receiver, since the transmission signals have undergone different coding and modulation for individual layers, can cancel a first restored signal of a particular layer to prevent the signal from causing interference to another layer. With application of such an interference canceller 205, it is possible to improve channel capacity of MIMO layers. As a result, it is possible to transmit more data through MCW MIMO transmission.
Although the receiver structure shown in FIG. 2 uses the interference canceller 205 by way of example, it can use a reception method of another type. Since the interference canceller 205 is used herein, a detailed description will be given below of the reception process based on the interference canceller 205 given in FIG. 2 by way of example.
When a signal of one layer is successfully restored by means of a demodulation and channel decoding unit 203, the interference canceller 205 cancels interference using the restored signal. The interference-canceled signal stream 207 is delivered back to the demodulation and channel decoding unit 203, and the restoration and interference cancellation is repeated until signals of all layers are successfully restored or there is no more signal of layers that will undergo restoration. Finally, the restored multiple signal streams for individual layers are restored to one desired transmission data stream by means of a multiplexer 209.
The MCW MIMO transmitter is characterized in that it should receive a CQI feedback for each layer since it generates multiple transmission signal streams by applying different channel coding and modulation units 203a through 203k for individual layers. Meanwhile, Rank can be implicitly expressed by setting a predefined CQI value indicating 'no-transmission' among CQI values, rather than separately feeding back CQIs. Therefore, in the MCW MIMO receiver, a feedback is composed of multiple CQIs representing channel states of individual transmission MIMO layers.
As described above, the SCW and MCW MIMO techniques have their advantages and disadvantages. Therefore, there is a need for a more effective MIMO technique.

Precoding apparatus and method in a wireless communication system using multiple input multiple output


The present disclosure relates to an IC chip holder capable of alleviating an impact if the IC chip holder is dropped or collides with another object. In the IC chip holder, an upper housing body is provided, on a side of electric contacts of an IC chip, with an elastic deformation portion capable of elastically deforming toward the inside of the upper housing body. Thus, since the elastic deformation portion elastically deforms toward the inside of the upper housing body, if the IC chip holder is dropped or collides with another object, the elastic deformation portion bends and the impact is alleviated. Thus, a possibility of the IC chip being damaged is reduced.

The present disclosure relates to an IC chip holder in which an IC chip is mounted.
A memory cartridge is known that stores, inside a housing body, a circuit board including a storage element such as a ROM or the like. For example, in a memory cartridge, an engagement protrusion is formed on an inner surface of a side wall of a housing body. A tab, which can be elastically deformed and which engages with the engagement protrusion, is provided on a circuit board, and the circuit board is fixed inside the housing body. Further, a connector is known that causes contact portions of a circuit board stored inside a housing body and a jack portion to be fitted with each other. For example, in a connector, a tab of a fixing arm on a side surface of a housing body storing a circuit board engages from the inside of a hole of the jack portion, and the circuit board and the jack portion are held in a stable contact state.


Techniques for improving parametric chip yield of manufactured chips are provided. In one aspect, a method for optimizing parametric chip yield is provided. The method includes the following steps. Parametric chip yield is computed based on performance and power consumption of a plurality of manufactured chips subject to a given voltage binning scheme. It is then determined whether the parametric chip yield computed is optimal. If the parametric chip yield is not optimal, the voltage binning scheme is altered and the compute and determine steps are repeated. Otherwise the binning scheme is left unaltered.

Process variation during chip manufacture is recognized as a major source of parametric yield loss. In the future, due to scaling down of complementary metal-oxide semiconductor (CMOS) transistor sizes, the situation is expected to get worse. Process variation causes high variability in gate delays and leakage current, which leads to high variability of chip operational frequency and power consumption. Due to chip-to-chip and within chip (intra-chip) variability, only some manufactured chips satisfy both performance and power requirements. The rest are either too slow or consume too much power. It is these non-compliant chips that represent parametric yield loss.
A number of approaches have been proposed to address this problem. A conservative design approach sets chip timing and power to a stricter target than required. To illustrate this point, ideally there would be no variability and chip timing and power requirements would just come from the specifications on the chip. If, however, due to variability the power varies, e.g., by 25 percent (%), and performance varies, e.g., by 20%, then in order to make all chips operational the chips have to be designed to be 25% faster than is required and consume 20% less power than is required. The main disadvantages of this approach are larger chip area and higher design costs.
Another approach, speed binning, reduces parametric yield loss by accepting low performance chips and selling them at discount price. See, for example, A. Datta et al., "Profit aware circuit design under process variations considering speed binning," IEEE Trans. on VLSI vol. 16, no. 7, pgs. 806-815 (July 2008). However, application-specific integrated circuit (ASIC) chips often have strict requirements for frequency and power. Chips not satisfying those frequency and power requirements have no value at all. Additionally, selling chips at a discount price reduces profits.
Yet another approach, body biasing, (see, for example, C. Neau et al., "Optimal body bias selection for leakage improvement and process compensation over different technology generations," ISLPED, pgs. 116-121 (August 2003) and T. Chen et al., "Comparison of adaptive body bias (ABB) and adaptive supply voltage (ASV) for improving delay and leakage under the presence of process variation," IEEE Trans. on VLSI, vol. 11, no. 5, pgs. 888-899 (October 2003) (hereinafter "T. Chen")) either reduces transistor leakage or improves gate delays. This approach requires connecting transistor bodies to a biasing voltage source. The wiring needed to do this, however, negatively affects chip routability, as well as increases design time and cost and is impractical for silicon-on-insulator (SOI) technology.
Therefore, improved techniques for reducing parametric yield loss would be desirable.

Yield computation and optimization for selective voltage binning

International Business Machines Corporation


A data processing apparatus in which a trade-off between over-learning prevention and calculation load prevention is eliminated when creating a model formula is provided. The data processing apparatus includes: a recording unit that records electronic data; and a computing unit that performs computing using the electronic data, in which the computing unit includes a feature amount selection unit used for computing, and the feature amount selection unit performs feature amount selection including: a first step (S101) of ranking feature amounts and rearranging the feature amounts from top; a second step (S103) of creating a plurality of data groups using only a part of the feature amounts according to the order; a third step (S104) of calculating a value that is an index for evaluating prediction performance of a regression or classification problem using each of the data groups using only a part of the feature amounts; a fourth step (S105) of deleting feature amounts based on the calculated prediction performance index; and a fifth step (S106) of updating the order of the feature amounts, which are feature amounts other than the deleted feature amount, using the prediction performance index, in which the second step to the fifth steps are iterated (S102) until an optimal value of the prediction performance index calculated in the third step is no longer updated.

In recent manufacturing industry, efforts to effectively utilize data obtained from a manufacturing apparatus and to improve productivity are attracting attention. For example, in a semiconductor manufacturing apparatus, prediction of a processing result using a method such as a multiple regression analysis as disclosed in PTL 1 is widely used for process control and the like. On the other hand, in the modern manufacturing industry where a manufacturing process is becoming more sophisticated and complicated, the amount of data obtained in an entire production plant is enormous, and selection of significant data and non-significant data becomes an indispensable problem in operating the manufacturing apparatus utilizing the data. For example, in the semiconductor manufacturing apparatus, it is not uncommon that one manufacturing apparatus is provided with more than several tens of sensors for acquiring the data. When the time required for the manufacturing process is long, more data is accumulated, and the amount of the data to be handled is enormous.
In normal data processing, a time-series signal obtained from measurement performed by a sensor is not used as it is, and a feature amount that well represents a feature of the signal or a value referred to as a feature is often used. An average value, a standard deviation value, or the like of the time-series signal is one of the feature amounts. By extracting the feature amount from the time-series signal, the amount of the data can be compressed to a certain extent, but at a stage of extracting the feature amount, it is desirable to extract as many feature amounts as possible. However, in many cases, many extracted feature amounts are meaningless in the analysis, and such meaningless feature amounts not only increase a calculation time, but also may adversely affect an analysis result as noise in the analysis. Furthermore, in creation of a model formula for predicting a processing result or the like, as the feature amount to be used increases, a possibility of falling into a state of lacking generalization performance referred to as over-learning when creating the model formula increases.
Akaike information criteria (AIC) is well known as an evaluation index of validity of the model formula for preventing the over-learning, and selection of significant variables using the AIC is disclosed in, for example, PTL 2. In addition, many algorithms have been proposed for selecting the feature amount that is likely to be analytically significant from the extracted feature amounts.
For example, NPL 1 discloses a feature amount selection method using iteration calculation referred to as sequential forward selection. This method is one of the feature amount selection methods and referred to as a wrapper method. NPL 2 discloses a method of creating a ranking of significance of feature amounts referred to as Fisher criterion and selecting the feature amount according to the ranking. This method is one of the feature amount selection methods and referred to as a filter method. NPL 3 discloses a method of creating a ranking of significance of feature amounts in a regression method referred to as random forest and selecting the feature amount according to the ranking. This method is one of the feature amount selection methods and referred to as an embedded method.

Data processing apparatus, data processing method and semiconductor manufacturing apparatus


A stamped and formed terminal for use in establishing an insulation displacemnt termination to wires extending in either of two orthogonal directions is disclosed. The stamped and formed terminal has a box-like conductor terminating portion in which wires can be inserted into slots in opposed plates on two of the four sides. The other plates can then be deformed about weakened sections to overlap a conductor inserted into plates on the adjacent sides to provide strain relief to prevent removal of the wire from the slots. Interconnection to other terminals can then be made to a termination portion which will have the same orientation relative to the insulative body regardless of the direction of the conductors.

1. Field of the Invention
This invention relates to insulation displacement terminals made from a flat stamping in which electrical contact with a wire or other conductor is established by the edges of a slot into which an insulated conductor is inserted.
2. Description of the Prior Art
Insulation displacement electrical terminals have been employed successfully in a number of applications. An insulation displacement terminal, normally consists of a stamped and formed member in which a slot is defined by opposed edges. When an insulated wire is inserted into such a slot, the opposed edges penetrate the insulation to establish intimate electrical contact with the underlying core of the insulated conductor. Since the slot edges penetrate the insulation, an insulation displacement terminal can be employed without first having to strip the insulation surrounding the conductive core.
One of the more widely and sucessfully employed insulation displacement terminals employs a terminal in which a slot is stamped in a plate-like member with a conductor being inserted into the slot with the conductor axis locally perpendicular to the plane of the plate-like terminal. The stresses induced in the terminal are not only sufficient to penetrate the insulation surrounding the conductive core of the wire but the slot edges can score or scape metal on the conductor to establish a clean, intimate contact, substantially free from contaminants such as oxide located on the exterior of the wire.
Although conventional insulation displacement contacts have proved suitable for many applications, the fact that the wire is not trapped within the insulation displacement slot formed in the terminal has resulted in some concerns, since the wire is not positively held in place. A number of different techniques have been employed to provide strain relief to the wire so that it cannot move upwardly out of the wire slot. A number of insulation displacement connectors have employed separate plastic strain relief caps to trap the conductors in the slots. U.S. Pat. No. 4,159,158 discloses a connector in which a wire is received in an insulation displacement slot. The insulative connector housing includes projections which prevent upward movement of the wire once it has been inserted into the wire receiving slot. U.S. Pat. No. 4,097,106 discloses an insulation displacement connector in which a portion of the insulative housing is deformed to provide conductor strain relief.
Other insulation displacement terminals have employed upper and lower opposed slots to firmly trap the conductor in place. U.S. Pat. No. 4,324,450 shows the essential features of such a termination. U.S. Pat. No. 4,019,801 discloses a similar configuration in which the conductors are first inserted through holes in the terminal and the terminal is then deformed to force plate-like portions having wire engaging slots down over the wire. Note that in each of these configurations the wire is firmly retained by the terminal itself.
Another technique for providing strain relief to conductors received in an insulation displacement slot is to employ a separate strain relief member which is a part of the terminal itself. U.S. Pat. Nos. 4,456,317; 4,277,124; 4,113,341 and 4,421,375 disclose insulation displacement terminals in which a separate strain relief barrel is crimped around the conductor after termination in an insulation displacement slot.
U.S. Pat. No. 3,950,065 discloses another terminal in which the strain relief is an integral portion of the stamped and formed terminal. This insulation displacement terminal has integral retaining fingers extending from interconnecting straps extending between parallel plates with aligned wire receiving slots. The fingers are angled inwardly and keep a fully inserted wire securely retained within the slot.
U.S. Pat. No. 4,039,239 discloses an insulation displacement terminal in which a wire positioned within an insulation displacement slot is held in place after the clip, in which the slot is located, is inserted over a free standing pin to which the wire is to be interconnected.
Each of these insulation displacement terminals employ a configuration in which the wire is inserted into a slot in a terminal initially held in a fixed position. For example, the terminal will normally be retained within an insulative housing having one or more terminals positioned with outwardly facing wire receiving slots. Insulation displacement slots can also be employed in connectors in which the wire is first secured to the housing and the terminal subsequently inserted. For example, U.S. Pat. No. 3,979,615 discloses an insulation displacement terminal which is inserted into a cavity in an insulative body to engage a wire secured in slots on opposite sides of a cavity. The wire spans the cavity and when the slots engage the conductor, a sound termination is established. Since the wire is trapped between the insulative housing and the terminal, a configuration of this type normally does not require additional wire strain relief. Similar terminals are disclosed in U.S. Pat. Nos. 4,118,103; 4,130,331 and 4,132,460. U.S. Pat. No. 4,557,544 shows a similar terminal having a wire crimp barrel on the opposite end of the terminal and U.S. Pat. No. 3,977,754 discloses a four sided terminal having insulation displacement slots in opposite sides at one end and a poke in contact on the opposite end.
Although each of these terminals is suitable for use in certain applications, there remains a need for simple, cost effective means to provide strain relief to prevent wires from moving upward out of insulation displacement slots. Preferably this strain relief should be provided by the terminal itself since the use of the insulative housing to supply the strain relief could result in relative movement between the conductor and the terminal because of the relative movement between the housing and the terminal itself.

Bidirectional insulation displacement electrical contact terminal


To provide a vapor deposition source of which material usage efficiency is higher as compared with the related art. A vapor deposition source (10) includes a vapor deposition particles ejecting unit (30) configured to include multistage of nozzle units layered apart from each other in a vertical direction, each of the nozzle units including at least one vapor deposition nozzle (32, 52), and at least one space part (43) provided between the multistage of vapor deposition nozzles, and a vacuum exhaust unit (14) connected with the at least one space part (43).

An electro luminescence (hereinafter, abbreviated as "EL") display device including EL elements using EL of organic materials or inorganic materials, which is an all solid-state type, has self-luminosity, and is excellent in low voltage driving and high responsiveness, has been being developed as a candidate of a next generation display technology.
The EL element is generally formed through film formation by a vacuum vapor deposition technique in which vapor deposition particles (component formed into film) are vapor-deposited on a target film forming substrate via a vapor deposition mask (also referred to as a shadow mask) under a reduced pressure (under a high vacuum), the vapor deposition mask having an opening of a prescribed pattern formed thereon. At this time, as a large substrate film formation technology using a large substrate as a target film forming substrate, a scan vapor deposition technique is promising which does not require a vapor deposition mask or vapor deposition source having a size equivalent to a large target film forming substrate. The scan vapor deposition technique performs scan film formation in which the vapor deposition source smaller than the target film forming substrate, or the vapor deposition mask and vapor deposition source smaller than the target film forming substrate are used to perform film formation while scanning the target film forming substrate.
In the film formation through the vacuum vapor deposition technique, a vapor deposition source including a heating portion and an emitting port is located within a vacuum chamber which is capable of keeping an inside thereof in a reduced pressure state such that the vapor deposition material is heated under a high vacuum to evaporate or sublimate the vapor deposition material. The vapor deposition material heated by the heating portion to be evaporated or sublimated is externally emitted to outside from an emitting port as the vapor deposition particles and deposited on the target film forming substrate.
However, the vapor deposition material heated by the heating portion to be evaporated or sublimated is scattered on an inner wall of the vapor deposition source (i.e., inner wall of a holder housing the heating portion), or the vapor deposition particles repeatedly collide with each other, and after that, the vapor deposition material is emitted from the emitting port.
Such scattering of the vapor deposition particles causes the vapor deposition particles emitted from the emitting port to be emitted in various directions.
In the vacuum vapor deposition technique, the vapor deposition particles emitted toward the target film forming substrate contribute to the film formation, but other vapor deposition particles do not contribute to the film formation. Therefore, in the vacuum vapor deposition technique, the vapor deposition particles other than the vapor deposition film deposited on the target film forming substrate are all loss of the materials. For this reason, the lower a directivity of the vapor deposition particles, the lower a material usage efficiency.
In recent years, a method has been proposed in which a directivity of the vapor deposition particles is heightened by limiting scattering directions of the vapor deposition particles, to thereby lead the vapor deposition particles to a vapor deposition region (e.g., PTL 1 or the like).
PTL 1 discloses that a flow of the vapor deposition particles (vapor deposition flow) is controlled by use of regulation plates regulating coming directions of the vapor deposition particles to improve usage efficiency of the vapor deposition material and improve film formation quality such that uniform vapor deposition is performed.
In a vapor deposition device disclosed in PTL 1, a target film forming substrate to be subjected to vapor deposition and a vapor deposition source are located within a vacuum chamber, and the vapor deposition particles discharged from the vapor deposition source are made to deposit on the target film forming substrate to form a vapor deposition film, not illustrated, on the target film forming substrate.
The vapor deposition source disclosed in PTL 1 includes three layered frame structures. A heating coil is wound around these frame structures.
The bottom frame structure is a heating portion (vapor deposition particles generation portion) which accommodates and heats the vapor deposition material to generate the vapor deposition particles. Each of the other two frame structures is a vapor deposition flow regulating layer (vapor deposition flow controller) which regulates directions of the vapor deposition particles from the bottom frame structure as the heating portion toward the target film forming substrate.
In the above two frame structures used as the vapor deposition flow regulating layer, there are formed multiple nozzle-like passing zones (vapor deposition nozzle, emitting port) sectioned by the regulation plates which are provided along the directions from the bottom frame structure used as the heating portion toward the target film forming substrate.
With this configuration, the scattering directions of the vapor deposition particles discharged from the heating portion via the respective passing zones are regulated to a direction along lateral surfaces of the respective regulation plates in the passing zones.

Vapor deposition source and vapor deposition device for producing vapor deposition film with high material usage efficiency


A metal target, at least the surface region of which has been oxidized, is prepared in a chamber. Then, a sputtering process is performed on the metal target with an inert gas ambient created in the chamber, thereby depositing a first metal oxide film as a lower part of a gate insulating film over a semiconductor substrate. Next, a reactive sputtering process is performed on the metal target with a mixed gas ambient, containing the inert gas and an oxygen gas, created in the chamber, thereby depositing a second metal oxide film as a middle or upper part of the gate insulating film over the first metal oxide film.

The present invention relates to a semiconductor device including a gate insulating film made of a material with a high dielectric constant (which will be herein referred to as a xe2x80x9chigh-dielectric-constant materialxe2x80x9d) and also relates to a method for fabricating the device.
Recently, there has been a growing demand for high-speed-operating semiconductor devices. To meet this demand, the gate insulating film of MOSFETs has been further thinned for the purpose of increasing the drivability thereof.
However, if the gate insulating film is a thin film of SiO2 (with a relative dielectric constant xcex5 of 3.9), the gate leakage current increases noticeably because a tunneling current flows therethrough.
Thus, to prevent the gate leakage current from increasing while enhancing the drivability of MOSFETs, various methods for increasing the actual thickness of the gate insulating film and obtaining a desired gate capacitance have been researched. For example, according to one of those methods, the gate insulating film is made of a high-dielectric-constant material (high-xcexa material) such as HfO2 (hafnium dioxide with a relative dielectric constant xcex5 of about 30) or ZrO2 (zirconium dioxide with a relative dielectric constant xcex5 of about 25).
To deposit a gate insulating film of a high-dielectric-constant material, a reactive sputtering process is performed using a target of Hf or Zr, for example, in a mixed gas ambient containing Ar (argon) and O2 gases, for example. In this manner, a gate insulating film of a high-dielectric-constant material such as HfO2 or ZrO2 can be deposited over a semiconductor substrate.
However, if the gate insulating film of the high-dielectric-constant material is deposited over a silicon substrate by the reactive sputtering method, for example, the surface of the silicon substrate is oxidized by a plasma created from the O2 gas during the reactive sputtering process. Thus, an unwanted silicon dioxide film is formed between the silicon substrate and gate insulating film. It should be noted that the unwanted film will be herein referred to as a xe2x80x9csilicon dioxide filmxe2x80x9d but can actually be any other silicon oxide film with a non-stoichimetric composition. Consequently, the gate insulating film becomes a stack of the silicon dioxide film with a relatively low dielectric constant and the high-dielectric-constant film. As a result, the gate insulating film has its effective dielectric constant decreased as a whole.
That is to say, the known method for fabricating a semi-conductor device cannot obtain the desired gate capacitance. Thus, it is difficult to enhance the drivability of MOSFETs.
FIG. 7 is a cross-sectional view showing the known method for fabricating a semiconductor device.
As shown in FIG. 7, a target 80 of Zr is placed in a chamber (not shown) and a silicon substrate 90 is loaded thereto. Then, a reactive sputtering process is performed using the target 80 with a mixed gas ambient containing Ar and O2 gases created in the chamber. During this process, the surface of the target 80 is oxidized, thereby forming a Zr oxide layer 81 thereon. At the same time, the surface of the silicon substrate 90 is also oxidized to be covered with a silicon dioxide film 91. Further, as a result of the reactive sputtering process, a Zr oxide film 92 is formed over the silicon substrate 90 with the silicon dioxide film 91 interposed therebetween. Accordingly, the resultant gate insulating film becomes a stack of the silicon dioxide film 91 and Zr oxide film 92. As a result, the gate capacitance decreases compared to a gate insulating film that has the same thickness but consists essentially of a Zr oxide film alone.
It is therefore an object of the present invention to enhance the drivability of MOSFETs by getting a gate insulating film, consisting essentially of a high-dielectric-constant material alone, formed by a sputtering process without allowing any silicon dioxide film to exist on the surface of a semiconductor substrate.
An inventive method for fabricating a semiconductor device includes the steps of: a) preparing a metal target in a chamber, at least a surface region of the target having been oxidized; b) performing a sputtering process using the metal target with an inert gas ambient created in the chamber, thereby depositing a first metal oxide film as a lower part of a gate insulating film over a semiconductor substrate; and c) performing a reactive sputtering process on the metal target with a mixed gas ambient, containing the inert gas and an oxygen gas, created in the chamber, thereby depositing a second metal oxide film as a middle or upper part of the gate insulating film over the first metal oxide film.
According to the inventive method, in the step of depositing a first metal oxide film over a semiconductor substrate, i.e., the initial stage of a process for forming a gate insulating film, no reactive sputtering process is performed but a sputtering process is performed using a metal target, at least the surface region of which has been oxidized, in an ambient containing no oxygen gas. Thus, the first metal oxide film can be deposited over the semiconductor substrate without allowing any silicon dioxide film to exist on the surface of the semiconductor substrate. Also, in the step of depositing a second metal oxide film over the first metal oxide film, i.e., after the initial stage of the process for forming the gate insulating film is over, a reactive sputtering process is performed in an ambient containing an oxygen gas with the surface of the semiconductor substrate covered with the first metal oxide film. Thus, the second metal oxide film can be deposited over the first metal oxide film without allowing any silicon dioxide film to exist on the surface of the semiconductor substrate. Accordingly, the gate insulating film can be essentially made up of the first and second metal oxide films alone. In other words, a gate insulating film consisting essentially of a high-dielectric-constant material alone can be formed. As a result, the resultant MOSFET can have its gate capacitance increased and its drivability enhanced. In addition, a gate leakage current can be minimized because the gate insulating film can be thick enough with a desired gate capacitance maintained.
In one embodiment of the present invention, the step a) may include the step of performing a provisional reactive sputtering process on the metal target to be oxidized with a mixed gas ambient, containing the inert and oxygen gases, created in the chamber, thereby oxidizing the surface region of the metal target before the semiconductor substrate is loaded into the chamber.
Then, the metal target with the oxidized surface region can be prepared easily.
In this particular embodiment, the provisional reactive sputtering process is preferably performed on another semi-conductor substrate that has been loaded into the chamber before the step a) is started.
Then, no insulating metal oxide is deposited on a wafer stage (which will be used as a gas-discharge electrode during the subsequent sputtering process steps) in the chamber when the surface region of the metal target is oxidized. As a result, it is possible to avoid the inability to apply a voltage to the semiconductor substrate in the subsequent process steps.
In another embodiment, the step c) may include the step of introducing the oxygen gas into the chamber with the inert gas, used in the step b), left in the chamber and with a gas-discharge continued from the step b) to carry out the reactive sputtering process.
Then, the steps b) and c) of depositing the first and second metal oxide films can be performed continuously. As a result, the throughput of the process improves.
In an alternative embodiment, the inventive method may further include, between the steps b) and c), the step of introducing the oxygen gas into the chamber with the inert gas, used in the step b), left in the chamber and with a gas-discharge for the sputtering process suspended.
Then, the mixture ratio of the inert and oxygen gases can be fixed before the step c) of depositing the second metal oxide film is started. As a result, the oxygen concentration of the second metal oxide film is controllable more easily.
In another alternative embodiment, the inventive method may further include, between the steps b) and c), the step of exhausting the inert gas, used in step b), from the chamber and then newly introducing the inert gas along with the oxygen gas into the chamber with a gas-discharge for the sputtering process suspended.
Then, the mixture ratio of the inert and oxygen gases should be fixed before the step c) of depositing the second metal oxide film is started. As a result, the oxygen concentration of the second metal oxide film is controllable much more easily.
In yet another embodiment, the step c) may include the step of supplying the oxygen gas at a controlled flow rate into the chamber to deposit the second metal oxide film with a different oxygen concentration from that of the first metal oxide film.
Then, the structure of the gate insulating film can be optimized with the reliability and the dielectric constant of the gate insulating film both taken into account. As a result, a highly reliable, high-performance MOSFET is realized.
An inventive semiconductor device includes a gate insulating film that includes: a first metal oxide film deposited on a semiconductor substrate; and a second metal oxide film deposited on the first metal oxide film. In this device, the first and second metal oxide films are made of the same type of metal oxide and have mutually different oxygen concentrations.
In the inventive device, the structure of the gate insulating film has been optimized with the reliability and the dielectric constant of the gate insulating film both taken into account. Thus, the device is implementable as a highly reliable, high-performance MOSFET.

Semiconductor device including a gate insulating film made of high-dielectric-constant material

Matsushita Electric Industrial Co., Ltd.


Carrier mobility in a heterojunction field effect device is increased by reducing or eliminating alloy scattering. The active channel region of the field effect device uses alternating layers of pure silicon and germanium which form a short period superlattice with the thickness of each layer in the superlattice being no greater than the critical thickness for maintaining a strained heterojunction. The gate contact of the field effect device can comprise quantum Si/Ge wires which provide quantum confinement in the growth plane, thereby allowing the field effect device to further improve the mobility by restricting phonon scattering. The structure can be used to improve device speed performance.

This invention relates generally to field effect semiconductor devices having strained heterojunction structures, and more particularly the invention relates to strained heterojunction field effect devices which have improved carrier mobility.
Electrical and optical silicon devices are known which use silicon-germanium alloy superlattice heterostructures and heterojunctions. A major advantage of the Si.sub.1-x Ge.sub.x /Si devices lies in the fabrication compatibility with semiconductor integrated circuit technology. With the advent of molecular beam epitaxy (MBE) and low temperature chemical vapor deposition (CVD) techniques, high quality Si.sub.1-x Ge.sub.x /Si heterostructures or superlattices can be easily obtained. Recently, heterojunction bipolar transistors using the Si.sub.1-x Ge.sub.x /Si strained layers have demonstrated potential applications in high speed circuits. See, for example, Plummer and Taft, U.S. Pat. No. 4,825,269 and Iyer, et al. "Heterojunction Bipolar Transistors Using Si/Ge Alloys" IEEE Transactions on Electron Devices, Vol 36 No. 10, October 1989, pp. 2033-2064. Cutoff frequency of above 75 GHZ has been reported. Furthermore, strained Si.sub.1-x Ge.sub.x /Si quantum well P-channel MOSFETS have been demonstrated using the enhancement and transconductance (g.sub.m) due to the high mobility of the strained silicon-germanium alloy layer. See Nayak et al. "Enhancement Mode Quantum Well Ge.sub.x Si.sub.1-x PMOS" , IEEE Electron Device Letters, Vol. 12 No. 4, April 1991, pp. 154-156. The transconductance (g.sub.m) is shown to be about twice that of the conventional p-doped silicon MOSFET with the same dimension.
A major advantage in using strained Si.sub.1-x Ge.sub.x /Si MOSFETS is improved carrier mobility in the channel by proper control of strain for increasing germanium concentration. However, the mobility enhancement of strained Si/Ge layers for both electrons and holes has a limiting factor. Mobility in semiconductors is in part governed by scatterings due to phonon, impurity or alloy. The dominant scattering mechanism for silicon-germanium alloys lies in the scattering due to the random mixture of silicon and germanium atoms.
The present invention is directed to increasing carrier mobility by reducing scattering in germanium-silicon superlattice structures.

Field effect devices having short period superlattice structures using Si and Ge

Board of Regents of the University of California