First electrode layer includes a plurality of first electrode lines (W1, W2) extending parallel to each other. State-variable layer lying on the first electrode layer includes a plurality of state-variable portions (60-11, 60-12, 60-21, 60-22) which exhibits a diode characteristic and a variable-resistance characteristic. Second electrode layer lying on the state-variable layer includes a plurality of second electrode lines (B1, B2) extending parallel to each other. The plurality of first electrode lines and the plurality of second electrode lines are crossing each other when seen in a layer-stacking direction with the state-variable layer interposed therebetween. State-variable portion (60-11) is provided at an intersection of the first electrode line (W1) and the second electrode line (B1) between the first electrode line and the second electrode line.

In recent years, along with the advancement of the digital techniques in electronic devices, demands for nonvolatile memory devices have been increasing for storage of data, such as a picture, and the like. Further, demands for increasing the capacity of a memory device, reducing the write power, shortening the read and write times, and prolonging the device's life have been escalating. To meet such demands, U.S. Pat. No. 6,204,139 (Patent Document 1) discloses a technique for forming a nonvolatile memory device using a perovskite material whose resistance value varies according to an applied electric pulse (e.g., Pr(1-X)CaXMnO3 (PCMO), LaSrMnO3 (LSMO), GdBaCoXOY (GBCO), etc.). According to the technique disclosed in this publication, a predetermined electric pulse is applied to these materials (hereinafter, generically referred to as "variable-resistance material(s)") to increase or decrease the resistance value of the materials. The resistance value which has varied as a result of application of the pulse is used for memorization of different values. Based on this scheme, the materials are used for memory devices.
U.S. Pat. No. 6,673,691 (Patent Document 2) discloses a method for changing the resistance value of a variable-resistance material by changing the pulse width of an electric pulse. U.S. Pat. No. 6,673,691 further discloses an example of a 1D1R (1 diode/1 resistor) memory cell array wherein a memory cell is formed using these variable-resistance materials, and a diode is used as a memory cell selection device. An advantage of this structure is a smaller memory cell size as compared with a structure which includes a transistor as a memory cell selection device.
FIG. 21 shows a memory device (1D1R nonvolatile memory device) 900 formed using a conventional variable-resistance material disclosed in U.S. Pat. No. 6,673,691. In this conventional example, the memory device 900 includes a substrate 901, a P/N junction diode (N-type Si region 902, P-type Si regions 903-1 and 903-2) formed on the substrate 901, a lower electrode 904-1 formed on the P-type Si region 903-1 of the diode, a lower electrode 904-2 formed on the P-type Si region 903-2 of the diode, a contact plug 905 formed on the N-type Si region 902 of the diode, a variable-resistance material layer 906 formed over the lower electrodes 904-1 and 904-2, and upper electrodes 907-1 and 907-2 formed on the variable-resistance material layer 906. In this conventional example, the lower electrodes 904-1 and 904-2 and the upper electrodes 907-1 and 907-2 are formed of Pt, and the variable-resistance material layer 906 is formed of P0.7Ca0.3MnO3.
In the memory device 900 shown in FIG. 21, when a predetermined pulse is applied between the upper electrode 907-1 and the lower electrode 904-1, the resistance value of a portion of the variable-resistance material layer 906 between the upper electrode 907-1 and the lower electrode 904-1 (variable region 906α) is varied. When a predetermined pulse is applied between the upper electrode 907-2 and the lower electrode 904-2, the resistance value of a portion of the variable-resistance material layer 906 between the upper electrode 907-2 and the lower electrode 904-2 (variable region 906β) is varied. That is, in this memory device, each of the variable region 906α and the variable region 906β is used as a single memory cell.
In the memory device 900 shown in FIG. 21, the P/N junction diode formed on the substrate 901 is used as a diode for selection of memory cells. Thus, an electric current flows from the upper electrode 907-1 (907-2) to the lower electrode 904-1 (904-2) (forward direction) but does not flow from the lower electrode 904-1 (904-2) to the upper electrode 907-1 (907-2) (reverse direction) or between the upper electrode 907-1 and the upper electrode 907-2.
FIG. 22 shows an equivalent circuit of the memory device 900 of FIG. 21. In FIG. 22, a word line W1 corresponds to the upper electrode 907-1, a word line W2 corresponds to the upper electrode 907-2, and a bit line B1 corresponds to the contact plug 905. A memory cell MC911 corresponds to the variable region 906α, a diode D911 corresponds to the diode (N-type Si region 902, P-type Si region 903-1), a memory cell MC912 corresponds to the variable region 906β, and a diode D912 corresponds to the diode (N-type Si region 902, P-type Si region 903-2).
<Operation>
Next, an operation of the memory device 900 of FIG. 21 is described with reference to FIG. 22. Herein, a process with the memory cell MC911 is described.
[Set (Memorization) or Reset]
In a memorization process, the word line W2 and the bit line B1 are pulled down to the ground, and a predetermined electric pulse is applied to the word line W1. As a result, the resistance value of the memory cell MC911 changes to a low resistance state (reset) or high resistance state (set). In an example disclosed in U.S. Pat. No. 6,673,691, when a pulse voltage having a voltage value of +4 V and a pulse width of 100 nsec is applied, the resistance value of the memory cell MC911 changes from the high resistance state to the low resistance state. When a pulse voltage having a voltage value of +2.5 V and a pulse width of 10 μsec is applied, the resistance value of the memory cell MC911 changes from the low resistance state to the high resistance state.
[Reproduction]
In a reproduction process, the word line W2 and the bit line B1 are pulled down to the ground, and a predetermined reproduction voltage (e.g., a voltage having a voltage value of +0.5 V) is applied to the word line W1. As a result, the electric current flowing through the memory cell MC911 is released to the bit line B1. On the other hand, no electric current flows through the memory cell MC912. Since the diode D912 (the N-type Si region 902 and P-type Si region 903-2 of FIG. 21) is provided to the memory cell MC912, no electric current flows from the word line W1 to the word line W2. Thus, only the resistance value of the memory cell MC911 can be detected.
According to the schemes as described above, the conventional memory device (1D1R nonvolatile memory device) 900 performs recording or reproduction in each memory cell.
U.S. Pat. No. 6,531,371 (Patent Document 3) realizes a larger-capacity memory device by constructing a cross point type memory device using a variable-resistance material. Specifically, as shown in FIG. 23, the intersections (cross points) of word lines W1 and W2 and bit lines B1 and B2 are respectively provided with memory cells 90-11, 90-12, 90-21, and 90-22, whereby the cross point type memory device is realized. The memory cells 90-11 through 90-22 are formed of a variable-resistance material.
However, in the memory device of FIG. 23, if a memory cell adjacent to a target memory cell from which information is to be read (e.g., variable-resistance material portions 90-11, 90-12, 90-22 adjacent to variable-resistance material portion 90-21) has a low resistance value, a current flows not only through the target memory cell but also through the adjacent memory cell as shown in FIG. 23, so that there is a possibility that the state of resistance in the target memory cell cannot be determined. In view of such, as disclosed in Japanese PCT National Phase Laid-Open Patent Publication No. 2002-530850 (Patent Document 4), a cross point type memory device has been proposed wherein a state-variable portion functioning as a memory cell and a diode called a steering portion are connected in series.
[Patent Document 1] U.S. Pat. No. 6,204,139
[Patent Document 2] U.S. Pat. No. 6,673,691
[Patent Document 3] U.S. Pat. No. 6,531,371
[Patent Document 4] Japanese PCT National Phase Laid-Open Patent Publication No. 2002-530850
[Non-Patent Document 1] 2002 IEDM, Article No. 7.5, December 2002

Memory device and semiconductor integrated circuit


In a clutch manufacturing method, inner peripheral portions of drum members, of two clutches are connected to a sleeve member, and a piston, a seal plate and a return spring are provided between the drum members, an insulator to apply electric isolation between the piston and the drum member or between the seal plate and another drum member is attached to either the piston or the seal plate. Then, the drum member, to which the electric isolation with the insulator is not applied, is connected to the sleeve member. Then, the piston, return sprig and seal plate are assembled to the sleeve member to which the drum member has been connected. Finally, the drum member, to which the electric isolation with the insulator has been applied, is connected to the sleeve member with electric welding.

The present invention relates to a manufacturing method of a clutch for an automatic transmission which comprises two clutches to conduct connection/disconnection between of an input shaft and rotational elements of a transmission mechanism, and the automatic transmission equipped with such two clutches.
Conventionally, an automatic transmission, in which two clutches are provided on an input side of a transmission mechanism including plural planetary gears and a power transmission path of an input rotation to the transmission mechanism is selectively changed by these clutches to provide multistage shifts of the transmission, is known as an automatic transmission for automotive vehicles. Herein, respective input members (drum members) of the two clutches are connected to the input shaft. For example, according to the automatic transmission which is shown in FIG. 11 of Japanese Patent Laid-Open Publication No. 2007-78191, an inner peripheral portion of a drum member of one of clutches is welded to a flange portion of an input shaft, and to this drum member is coupled an inner peripheral portion of a drum member of the other of the clutches via a spline connection, thereby connecting the two clutches to the input shaft.
There are other types of automatic transmission in which the connection of the drum members of the two clutch members and the input shaft is conducted by connecting the inner peripheral portions of the drum members of the clutches to a sleeve member which is coupled to the input shaft via the spline connection by means of welding or spline connection.
Herein, in case the drum member of at least one of the clutches is coupled to the sleeve member via the spline connection, the spline connection may require a specified length in its axial direction. Accordingly, the sleeve member may be relatively long, so the entire length of the automatic transmission in the axial direction may become improperly long.
The connection with the electric welding between the drum members of the two clutches and the sleeve member may be considered as one of countermeasures for this problem. Herein, various elements of the clutch, such as a piston, a return spring, a spring receiver (a seal plate to form a centrifugal balance oil-pressure chamber), are provided between the drum members of the two clutches, so the electric-welding connection of the two drum members needs to be conducted via these various elements. Accordingly, the drum member of one of the clutches is welded to a specified position of the sleeve member, then the drum member of the other clutch is made contact the specified position of the sleeve member in a state in which the piston, spring and so on are disposed properly between the drum members. Here, in this state, an electric current is applied to the drum member of the other clutch and the sleeve member. Thus, the electric current flows through not only the contact portion between the drum member of one of the clutches and the sleeve member but the piston, spring, drum member of the other clutch, and so on. Accordingly, there is a possibility that welding would occur at a portion between the spring and the piston as well as the contact portion between the drum member of one of the clutches and the sleeve member. Further, there is a possibility that the piston, spring and the like have a welding deformation.

Manufacturing method of clutch for automatic transmission and automatic transmission


The present invention discloses a method of implementing traversal of multimedia protocols through Network Address Translation device, wherein the Network Address Translation (NAT) device establishes a mapping relationship between address of a terminal in a private network and address of a terminal in a public network, and enables the terminal in the private network to access the public network with a common address, so as to implement the interaction of media streams between the terminal in the private network and the terminal in the public network, in which the common address is a address of the private network for accessing the public network.

All multimedia communication protocols involve transmission of media streams (audio/video streams); and it is more suitable to use UDP protocol for transmission of media streams in IP networks. Therefore, presently, most of the multimedia communication protocols employ UDP to carry media streams. However, as UDP is a non-connection-oriented transport layer protocol, Network Address Translation (NAT) devices are unable to determine which private network the incoming UDP packets should be forwarded to; as a result, UDP packets transmitted from a public network to a private network will be rejected by the NAT devices, resulting in the final fail of multimedia communication.
The modes for address translation of NAT device can be classified into: static NAT, dynamic NAT, and NAPT (Network Address Port Translation). At present, solutions for traversal of multimedia communication protocol through NAT in either of the first two NAT modes described above are available: static NAT solution and dynamic NAT solution.
The static NAT solution is implemented as follows: The private network IP addresses of terminals requesting for multimedia communication are mapped to public network IP addresses at NAT, that is, NAT reserves public network IP addresses for the multimedia terminals in private networks in advance.
Based on the static NAT solution, a dynamic NAT solution is introduced: when there is multimedia communication (detected at a known port, for example, the calling known port for H.323 protocol is 1720), NAT assigns a certain public network IP address to the multimedia terminal in the private network dynamically, till the multimedia communication is terminated. Though this solution has been improved compared with the static NAT solution, it is still impossible to save IP address resources in public networks. This is because once a certain public network IP address is assigned to a certain private network terminal, other terminals cannot use this public network IP address before the multimedia communication of this terminal is terminated.
However, for NAPT mode, there is still no solution for traversal of multimedia communication protocol through NAT. NAT devices are designed to save public network IP address resources. But it is impossible for a public network IP address to be shared by multiple multimedia communication terminals in a private network in the existing solutions. In many cases involving private network terminals (or other H.323 nodes), waste of public network IP addresses is inevitable, causing the major function of NAT devices—saving public network IP addresses—to be meaningless in multimedia applications.

This patent is subject to a terminal disclaimer.

Method of implementing traversal of multimedia protocols through network address translation device


Branched polycarbonate prepolymers are produced by adding phosgene, one or more dihydric phenols, one or more polyhydric phenols, a halogenated organic solvent, and an aqueous caustic solution together with mixing in motionless mixers to form fine dispersions of partially phosgenated phenols, allowing for interfacial reactions to occur in residence time sections and repeating the steps after the addition of caustic to form high molecular weight prepolymers. These prepolymers are then polymerized with amines to form branched high molecular weight polycarbonates.

This invention relates to an interfacial process for the production of high molecular weight polycarbonates wherein high molecular weight prepolymers are first formed using one or more motionless mixers and are then coupled to form the polycarbonates.
It is known from U.S. Pat. No. 3,674,740 that polycarbonates can be made from carbonate forming monomers in a tubular reactor using a series of mixing tubes and residence time tubes. However, this known process is designed to produce emulsions which are difficult to process for the recovery of the polycarbonate resins. Other related processes for the production of polycarbonates are known from U.S. Pat. Nos. 3,974,126 and 3,530,094, but none of these patents teaches the making of high molecular weight prepolymers and the polymerization thereof.
A continuous process for making linear polycarbonate is disclosed in U.S. Pat. No. 4,847,352. However, this patent does not disclose use of that process for making branched polycarbonate.

Continuous motionless mixer reactor process for the production of branched polycarbonate


According to the disclosure the amount of wind for an air cleaner is controlled regardless of the initial value of the air pollution level sensed by a sensor in an air cleaner. This allows the air cleaner to efficiently perform the air cleaning regardless of the area in which the air cleaner is installed. The method for cleaning the air requires that an initial value of the air pollution is sensed. Thereafter, an air pollution grade from a plurality of air pollution grades having a plurality of blow control sections set according to each of the air pollution grades. A present value of the air pollution is then sensed. An increment of air pollution is determined from the initial value of the air pollution and the value of the air pollution sensed at a later time. Based on this increment that has been determined, the amount of air from the air cleaner is controlled depending on that increment.

1. Field of the Invention
The present invention relates in general to an air cleaner having an air pollution level sensor, and more particularly to a method for controlling the amount of wind for air-cleaning regardless of an initial air pollution level sensed by the air pollution level sensor in the air cleaner.
2. Description of the Prior Art
Referring to FIG. 1, there is shown a schematic block diagram of a conventional air cleaner. As shown in this drawing, the conventional air cleaner comprises a sensor 1 having a heating coil 2, electrodes 3 and a variable resistor 4, for sensing an air pollution level, a microcomputer 5 for outputting a wind amount control signal in response to an output signal from the sensor 1 corresponding to the sensed air pollution level, and a blower 8 for generating a desired amount of wind for the air-cleaning in accordance with the wind amount control signal from the microcomputer 5. The variable resistor 4 functions to adjust sensitivity of the sensor 1.
The operation of the conventional air cleaner with the above-mentioned construction will hereinafter be described.
Upon powering on the air cleaner, the sensor 1 is supplied with a DC voltage Vcc (about +5 V). Then in the sensor 1, the heating coil 2 is heated and a voltage based on the heating of the heating coil 2 appears between the electrodes 3. The variable resistor 4 cooperates with the electrodes 3 to divide the +5 V DC voltage Vcc and applies the divided voltage to the microcomputer 5.
For example, provided that the air pollution level is increased, the voltage between the electrodes 3 drops abruptly and the voltage across the variable resistor 4 rises to the contrary. The increased voltage across the variable resistor 4 is applied to an input terminal I1 of the microcomputer 5. In response to the increased voltage across the variable resistor 4 inputted therein, the microcomputer 5 recognizes that the air pollution level is increased and outputs the wind amount control signal in accordance with the increased air pollution level. As a result, the blower 6 generates a desired amount of wind for the air-cleaning based on the wind amount control signal from the microcomputer 5.
Thereafter, when the air is cleaned and, the air pollution level becomes low, the voltage between the electrodes 3 is increased and the voltage across the variable resistor 4 is reduced to the contrary. The reduced voltage across the variable resistor 4 is applied to the input terminal I1 of the microcomputer 5. In response to the reduced voltage across the variable resistor 4 inputted therein, the microcomputer 5 recognizes that the air pollution level is reduced and outputs the wind amount control signal via output terminal O1 in accordance with the reduced air pollution level. As a result, the blower 6 decreases the amount of wind based on the wind amount control signal from the microcomputer 5.
Referring to FIG. 2, there is shown a flowchart illustrating a system program stored in the microcomputer 5 in the conventional air cleaner of FIG. 1. Upon powering on the air cleaner, it is checked whether a reference value of the air pollution level has been defined. If the air cleaner has not been powered on, the system program returns to an initial state. Herein, the reference value means an initial value of the air pollution level sensed by the sensor I after powering-on of the air cleaner.
If the reference value of the air pollution level has been defined, the defined reference value of the air pollution level is compared with a present air pollution level being provided by the sensor I with the lapse of the time. Obtained in accordance with the compared result is an increment of the air pollution level.
If the reference value of the air pollution level has not been defined, the system program defines an initial value of the air pollution level provided by the sensor 1 as the reference value and then returns to the initial state.
When the increment (referred to hereinafter as .increment.PPM) of the air pollution level is smaller than 3, the blower 6 is deactivated. If the .increment.PPM is greater than or equal to 3 and smaller than 6, the blower 6 is activated to generate one-stage amount of wind. If the .increment.PPM is greater than or equal to 6 and smaller than 9, the blower 6 is activated to generate two-stage amount of wind larger than the one-stage amount of wind. If the .increment.PPM is greater than or equal to 9 and smaller than 12, the blower 6 is activated to generate three-stage amount of wind larger than the two-stage amount of wind. Also, if the .increment.PPM is greater than or equal to 12 and smaller than 15, the blower 6 is activated to generate four-stage amount of wind larger than the three-stage amount of wind. Finally, when the .increment.PPM is greater than or equal to 15, the blower 6 is activated to generate five-stage amount of wind as the maximum amount. In this connection, the amount of wind generated by the blower 6 can be defined in the following order: EQU 1-stage<2-stage<3-stage<4-stage<5-stage (1)
As mentioned above, in the conventional air cleaner, the initial value of the air pollution level sensed by the sensor 1 is defined as the reference value and the increment of the air pollution level is obtained by subtracting the reference value from a present air pollution level being provided by the sensor 1 with the lapse of the time. Then in accordance with the increment of the air pollution level , the blower 6 is activated to generate one of the 1-stage to 5-stage amount of wind. Namely, the more the increment of the air pollution level is, the generated amount of wind is increased.
However, the conventional air cleaner is desirable in the place of relatively low air pollution level such as houses and offices, but has a disadvantage in the place of relatively high air pollution level such as underground shopping arcades, factories and public places crowded with many people. In other words, the relatively high air pollution level has been present in the factory from powering-on of the air cleaner. For this reason, a great amount of wind must be generated just from powering-on of the air cleaner. However, according to the conventional air cleaner, the initial value of the air pollution level sensed by the sensor 1 is defined as the reference value and much amount of wind is not generated until the increment of the air pollution level is considerably large. Moreover, when the increment of the air pollution level is considerably large, the amount of wind will, in fact, be required more for the air-cleaning in the factory than that of wind generated from the air cleaner at that time. That is, in the case where the conventional air cleaner is employed in the place of relatively high air pollution level, it cannot perform an efficient air cleaning because too high value is defined as the reference value.


A flexible and very thin touch screen panel is implemented by forming sensing patterns as touch sensors on a first surface of a flexible thin glass substrate and by forming a supporting film on a second surface of the glass surface. A method of fabricating a touch screen panel for securing the strength of a unit cell touch screen panel includes forming sensing patterns as touch sensors in every unit cell touch screen panel on a mother glass substrate, etching the glass substrate in the thickness direction, forming a supporting film under the glass substrate, and by cutting the glass substrate and the supporting film cell by cell using dual cutting.

1. Field
Aspects of embodiments according to the present invention relate to a touch screen panel, and particularly, to a flexible touch screen panel and a fabrication method thereof.
2. Description of the Related Art
A touch screen panel is an input device enabling a person to select an instruction displayed on a screen of an image display device with his/her finger or a tool and to input a user command.
To this end, the touch screen panel is provided on a front face of the image display device and converts a contact position on the screen, which is directly contacted by a finger or a tool, into an electrical signal. By doing so, an instruction selected at the contact position is received as an input signal.
The touch screen panel has an increasing range of applications because the touch screen panel can be a substitute of a separated input device such as a keyboard or a mouse that is coupled to an image display device.
There are known touch screen panels such as a resistive touch screen panel, a light sensitive touch screen panel, and a capacitive touch screen panel. The capacitive touch screen panel converts a contact position into an electric signal by sensing change of electrostatic capacity generated by a conductive sensing pattern in association with another sensing pattern or a ground electrode when a finger or an object contacts the touch screen.
Such touch screen panel is attached to an outer surface of a flat panel display device such as a liquid crystal display (LCD) or an organic light emitting display (OLED) device. Therefore, the touch screen panel has high transparency and thin thickness.
In addition, a flexible flat display device has been developed recently and the touch screen panel attached to the flexible flat display device should also be flexible.
However, since processes such as a thin film membrane and a pattern forming process are used in order to form a sensing pattern for implementing a touch sensor, the capacitive touch screen panel has high heat resistance and chemical resistance.
Therefore, in a known capacitance touch screen panel, a sensing pattern is formed on a glass substrate suitable for the process. However, in this case, since the glass substrate should have a sufficient thickness to allow the glass substrate to be transferred, the glass substrate does not have a thin thickness to provide for flexibility.

Touch screen panel and fabrication method thereof


The present invention provides a method, an apparatus, and a system for reducing digital subscriber line interference. The method includes identifying at least one legacy digital subscriber line in a digital subscriber line DSL legacy line set, obtaining a target power spectral density mask value of the legacy digital subscriber line, and reducing power spectral density of a transmit signal on the legacy digital subscriber line according to the target power spectral density mask value.

A digital subscriber line (DSL) technology is a high-speed transmission technology for performing data transmission by using a telephone twisted pair, that is, an unshielded twisted pair (UTP), and includes asymmetric digital subscriber line (ADSL), very-high-bit-rate digital subscriber line (VDSL), integrated services digital network (ISDN) digital subscriber line (IDSL), and the like. These digital subscriber lines are collectively referred to as xDSL.
Except baseband transmission DSLs such as IDSL and symmetric high-bit-rate digital subscriber line (SHDSL), a passband transmission xDSL enables, by using a frequency division multiplexing technology, an xDSL service to coexist on the same twisted pair as a plain old telephone service (POTS), where the xDSL service occupies a high frequency band, the POTS occupies a baseband part below 4 kilohertz (KHz), and a POTS signal and an xDSL signal are split by using a splitter. The passband transmission xDSL uses discrete multi-tone (DMT) modulation. A system providing access to multiple xDSLs is referred to as a digital subscriber line access multiplexer (DSLAM). According to the electromagnetic induction principle, mutual interference is caused among multiple channels of signals received by the DSLAM, which is referred to as crosstalk and includes near end crosstalk (NEXT) and far end crosstalk (FEXT). NEXT and FEXT energy increases as a frequency band increases. On one hand, because frequency division multiplexing is used for xDSL upstream and downstream channels, NEXT does not cause serious harm on system performance. On the other hand, because a frequency band used by the xDSL becomes wider, FEXT affects transmission performance of a line more seriously. According to Shannon's equation C=B·log2 (1+S/N) (where C is a channel capacity, B is a signal bandwidth, S is signal energy, and N is noise energy), because crosstalk in an xDSL reflects a part of noises, serious FEXT significantly reduces a channel rate. When an xDSL service is provisioned in a bundle of cables upon the requests of multiple subscribers, FEXT may cause a low rate, unstable performance, and even a failure in service provisioning for some lines, which eventually results in a low line activation ratio of the DSLAM.
Currently, a technology known as a vectored digital subscriber line (Vectored-DSL) is proposed in the industry. The Vectored-DSL mainly uses a possibility of joint sending and receiving on a DSLAM end to cancel FEXT interference by using a signal processing method, thereby eventually eliminating FEXT interference in each channel of signals. As shown in FIG. 1a and FIG. 1b which respectively provide a schematic diagram of synchronous sending and synchronous receiving on a DSLAM end, an operating principle thereof is described as follows:
In FIG. 1a and FIG. 1b, a shared channel H on a kth tone of a frequency domain may be expressed in a matrix form:
                    H        =                              [                                                                                h                    11                                                                                h                    12                                                                    …                                                                      h                                          1                                            M                                                                                h                    21                                                                                h                    22                                                                    …                                                                      h                                          2                                            M                                                                                h                      ij                                                                                ⋮                                                  ⋮                                                  ⋱                                                  ⋮                                                                                h                                          M                                                                                1                                                                                h                                          M                                                                                2                                                                                …                                                                      h                    MM                                                                        ]                                M            ×            M                                              (                  equation                                                              1                )
In equation 1, hij is a transmission equation from a line pair j to a line pair i. Actually, i equals j and equals the number of channels having a crosstalk relationship with each other in a shared channel. On an assumption that the number of channels having a crosstalk relationship with each other in a shared channel is M, H is an M×M channel transmission matrix. Moreover, it is respectively assumed that x is an M×1 channel input vector, y is an M×1 channel output vector, and n is an M×1 noise vector. Eventually, the channel transmission equation is expressed in the following form:y=Hx+n  (equation 1)
In an upstream direction, joint receiving processing on a signal is performed on a central office (CO) end. That is, a crosstalk canceller (expressed by using a matrix W) is added to a vectored digital subscriber line access multiplexer (Vectored DSLAM; refer to FIG. 1c for functional modules thereof) on the CO end, and therefore a signal received by the CO end is:{tilde over (y)}=Wy=WHx+Wn  (equation 2)
When a matrix WH is a diagonal matrix, the crosstalk is eliminated.
In a downstream direction, joint sending processing on a signal is performed on the CO end. That is, a crosstalk pre-coder (expressed by using a matrix P) is added to the Vectored DSLAM on the CO end, and therefore a signal sent by the CO end is:{tilde over (x)}=Px  (equation 3)
A signal received by a receiving end (subscriber end) is:{tilde over (y)}=H{tilde over (x)}+n=HPx+n  (equation 4)
When a matrix HP is a diagonal matrix, the crosstalk is also eliminated.
According to the above analysis, a Vectored-DSL performs joint processing in the upstream and downstream directions to achieve an effect of eliminating far end crosstalk, where a key point thereof is to estimate a downstream precoding matrix P and an upstream cancellation matrix W.
Generally, a Vectored-DSL system is implemented by using the following method. Firstly, synchronization is performed using a synchronization symbol (Sync Symbol. Then, joint modulation is performed on a pilot sequence on Sync Symbols of all lines. Finally, a receiving side feeds back an error to a vectoring control entity (VCE). Hence, a downstream precoding matrix P and an upstream cancellation matrix W may be estimated in the VCE, thereby canceling FEXT by using the vectoring technology described above.
A VDSL2 technology is prior to the Vectored-DSL technology and has been widely applied; therefore, compatibility with an existing VDSL2 legacy (Legacy) customer premises equipment (CPE), that is, a VDSL2 Legacy CPE, on an existing network must be considered when upgrading a VDSL2 to a Vectored-DSL. However, the VDSL2 Legacy CPE does not support pilot sequence sending and receiving or error feedback on a synchronization symbol.
In order to support pilot sequence sending and receiving and error feedback on a synchronization symbol (Sync Symbol) to estimate a downstream precoding matrix P and an upstream cancellation matrix W, a method provided by the prior art is to upgrade all VDSL2 Legacy CPEs on an existing VDSL2 network to or replace them with VDSL2 Vectored CPEs.

Method, apparatus, and system for reducing digital subscriber line interference


A digital broadcasting receiver, a driving method, and a self-diagnosis method thereof, are provided. The digital broadcasting receiver includes: a digital demodulator for demodulating a digital broadcasting signal; a lock detector for detecting lock signals from the digital demodulator; and a lock processor for controlling the digital demodulator according to the detected lock signal.

1. Field of the Invention
The present invention relates to a digital broadcasting receiver, and more particularly, to a digital broadcasting receiver, a driving method and a self-diagnosis method thereof that can diagnose and settle its problems by itself.
2. Discussion of the Related Art
In general, a digital broadcasting receiver can be roughly divided into three receiving methods.
First, there is a ground wave broadcasting receiving method, which includes a VSB (Vestigial Side Band) method and an OFDM (Orthogonal Frequency Division Multiplexing) method. Second, a cable broadcasting receiving method, which includes QAM (Quadrature Amplitude Modulation) method. Third, a satellite broadcasting receiving method, which includes QPSK (Quadrature Phase Shift Keying) method
These three receiving methods are different each other in their analog reception block, signal synchronization method, channel equalizing method, matched filter, and channel demodulation method.
However, recently, a multi-medium digital broadcasting receiver capable of incorporating the above-mentioned receiving methods is under development.
FIG. 1 is a block diagram illustrating a construction of a general multi-medium digital broadcasting receiver. As shown in FIG. 1, the general multi-medium digital broadcasting receiver includes: an analog reception block 101 for receiving a radio frequency (RF) signal and converting the signal into an intermediate frequency (IF) signal; a digital demodulation block 102 for demodulating a converted digital signal using an A/D (Analog/Digital) converter 101g.
The analog reception block 101 receives a radio frequency signal of 50-860 MHz, converts the signal into an IF signal, converts the converted signal into a digital signal using an A/D converter 101g, and delivers the digital signal to the digital demodulation block 102.
The analog reception block 101 includes: a tuner 101b for converting a RF signal of 50-860 MHz received through an antenna 101a into a first IF signal of about 44 MHz; a surface acoustic wave (SAW) filter 101c for filtering a signal outputted from the tuner 101b; a first oscillator 101d for generating an oscillating frequency so as to generate a second IF signal; a mixer 101e for converting the filtered signal into a second IF signal by down-converting the signal filtered by the SAW filter 101c using an oscillating frequency generated at the first oscillator 101d; a second oscillator 101f for generating a sampling frequency; and the A/D converter 101g for converting a signal converted by the mixer 101e into a digital signal according to a sampling frequency generated at the second oscillator 101f.
At this point, the analog reception block 101 among elements of the multi-medium digital broadcasting receiver has a standardized construction as shown in FIG. 1 regardless of demodulation methods.
On the contrary, the digital demodulation block 102 may have a variety of forms depending on demodulation methods such as VSB, OFDM, QAM, QPSK.
Generally, the digital demodulation block 102 may include a gain recovery, a phase splitter, a complex multiplier, an interpolator, a match filter, a symbol clock recovery, a channel equalizer, a carrier recovery, and an FEC (Forward Error Correction).
However, since the multi-medium digital broadcasting receiver is operated in response to an individual lock of each element, there is a problem in that stability of a system is deteriorated if a false lock is generated between the elements.
For example, if a gain recovery operates after a system is reset, other elements should be a reset mode.
However, with a gain of a received signal not tracked up to an appropriate level, if a channel equalizer of a blind state is operated, there is high possibility that the channel equalizer operates in its state of local minima. Such characteristics are fatal to system stability.
Further, since a training sequence does not exist in the QAM method, there is high possibility that a symbol clock recovery and a carrier recovery diverge.
Further, the symbol clock recovery operates in response to an automatic gain control (AGC) lock signal from a lock detector of the gain recovery.
However, the elements at the end should maintain a reset state constantly, but if the channel equalizer operates, there is high possibility that the channel equalizer operates in the state of local minima, which is fatal to system stability.

Digital broadcasting receiver, driving method thereof, and self-diagnosis method thereof


An electronic device includes a memory device including a power switch configured to provide one of a first voltage and a second voltage to an internal circuit in response to a control command. A power management device is configured to generate the first voltage, the second voltage, and the control command and to provide the first voltage, the second voltage, and the control command to the memory device. The power switch provides the second voltage to the internal circuit while a level of the first voltage is changed and provides the first voltage to the internal circuit after a level change of the first voltage is completed.

Embodiments of the inventive concepts described herein relate to dynamic power management, and more particularly, to power management of a memory device by dynamically changing a supply voltage.
Electronic mobile devices typically include semiconductor integrated circuits that are increasingly required to have high integration, high performance, and low power consumption. Low power consumption of a semiconductor integrated circuit may be implemented in various ways, typically by lowering a drive voltage of the semiconductor integrated circuit.
Another method for lowering power consumption of the electronic device is through the use of dynamic voltage frequency scaling (DVFS) technology. The DVFS is a technology to reduce power consumption by changing an operating speed and a drive voltage of a semiconductor integrated circuit according to a workload of an integrated circuit. The semiconductor integrated circuit can change drive voltages of a logic circuit and an interface circuit by the DVFS operation.
When a drive voltage is changed by the DVFS operation, an operating characteristic of the logic circuit and the interface circuit can change. Additionally, the drive voltage may remain unstable for at least a micro unit (e.g. microsecond) when the drive voltage is changed.

Power management of a memory device by dynamically changing supply voltage


In a tablet computer stand for supporting a computer shaped as a tablet on a support surface, at least first and second footprints joined at an angle are provided. At a front end of the first footprint, a receiving surface is provided receiving an end of the tablet to support the tablet on the stand. The footprints are dimensioned and positioned relative to the tablet end receiving surface such that the stand with the tablet thereon is stable when either the first or the second footprint is resting on said support surface.

Tablet computers, or simply tablets, enjoy more and more popularity as sales figures of recent years show, and will probably play an even bigger role in the near future. Examples of such prior art tablets are the "iPad" and "iPad mini" by Apple, "Kindle" by Amazon, "Slate" by Hewlett-Packard, "Iconia" by Acer, "XPS 10" by Dell, "Ideatab" and "Thinkpad" by Lenovo, "Galaxy Tab" by Samsung, and "EeeSlate" by Asus. Minimal slim design, simplicity, and light weight are principal aspects that make the tablets extremely mobile and therefore attractive. Nevertheless, the handling and support for the actual use of these devices is not solved and even offered by their design. It is necessary to use an additional device for handling and support.
Support devices in the prior art are principally the following design types:                One-piece stands that offer one position with a fixed viewing angle, mostly for horizontal and vertical formats. Some of the stands offer additional angles by changing the position on the same stand. One thing they all have in common is having only one single footprint.        Multipartite stands with adaption mechanisms offer a variety of viewing angles and mostly function with retaining elements on the back.

Multiple footprint stand for a tablet computer


A rotor blade assembly for a wind turbine is presented. The rotor blade assembly includes a rotor blade having a surface, where the surface of the rotor blade includes an inclined groove. The rotor blade assembly further includes at least one add-on element mounted on the surface of the rotor blade via a bonding interface downstream of the inclined groove such that particulate matter in an airflow upstream of the at least one add-on element is deflected away from the bonding interface between the surface of the rotor blade and the at least one add-on element. The wind turbine having the rotor blade assembly is also presented.

The present disclosure relates in general to rotor blades for wind turbines, and more particularly to a rotor blade having an inclined groove and one or more add-on elements.
Wind power is considered one of the cleanest, most environmentally friendly energy sources presently available, and wind turbines have gained increased attention in this regard. A modern wind turbine typically includes a tower, a generator, a gearbox, a nacelle, and one or more rotor blades. The rotor blades capture kinetic energy of wind using known foil principles. The rotor blades transmit the kinetic energy in the form of rotational energy so as to turn a shaft coupling the rotor blades to a gearbox, or if a gearbox is not used, directly to the generator. The generator then converts the mechanical energy to electrical energy that may be deployed to a utility grid.
In many cases, accessory components are attached to the rotor blades of wind turbines to perform various functions during operation of the wind turbine. For example, it is known to change the aerodynamic characteristics of wind turbine rotor blades by adding protrusions or other structures (often referred to as "vortex generators") to the surface of the blade in order to increase the energy conversion efficiency during normal operation of the wind turbine by increasing the lift force of the blades while decreasing the drag force. Vortex generators serve to increase the attached-flow region and to reduce the detached-flow region by moving the point of flow separation nearer to the trailing edge of the blade or to delay it from occurring altogether. In particular, vortex generators create local regions of longitudinally rotating, turbulent airflow over the surface of the blade as a means to delay flow separation and thus optimize aerodynamic airflow around the blade contour.
Laminar boundary layer instability noise occurs, however, when flow instabilities are scattered by a uniform discontinuity (e.g. the edge of a vortex generator panel or other blade add-on elements) on the rotor blade surface. These scattered acoustic waves travel upstream, where they interact with and amplify the initial amplitude of the flow instabilities. The result is a feedback loop that produces multiple acoustic tones, regularly-spaced in frequency, which create undesirable noise for the wind turbine. Moreover, particulate matter contained in an airflow may also damage or lift-off add-on elements disposed on the rotor blade surface. Further, in some conditions, the add-on elements are also prone to erosion, especially, at a bonding interface between the add-on elements and the rotor blade surface, due water droplets contained in the airflow and impinging on the bonding interface.

Rotor blade assembly and a wind turbine having the rotor blade assembly


A pre-trained source encoder generates a source encoder representation for each image of a labeled set of source images from a source domain. A target encoder generates a target encoder representation for each image of an unlabeled set of target images from a target domain. A generative adversarial network outputs a first prediction indicating whether each of the source encoder representations and each of the target encoder representations originate from the source domain or the target domain. The generative adversarial network outputs a second prediction of the latent code for each of the source encoder representations and each of the target encoder representations. The target encoder and the generative adversarial network are trained by repeatedly updating parameters of the target encoder and the generative adversarial network until one or more predetermined stopping conditions occur.

The development of deep neural networks including deep convolutional neural networks (CNN) has boosted a large number of computer vision applications. Significant advances that deep neural networks have achieved in various computer vision applications are to a large extent due to the availability of large-scaled labeled data. However, the manual annotation of large amounts of training data for various applications remains expensive. Further, neural networks trained on one set of images do not generalize well to novel sets of images and tasks. This is due to domain shift, the differences between the data distributions of the two domains from which the sets of images are sampled. For example, a neural network trained on synthetic images of objects will not generalize well to photographs of the objects in the real world that include different backgrounds as well as light and shadow. Accordingly, the aim of domain adaptation is to generalize the learned knowledge from a labeled source domain to a new unlabeled domain, called the target domain.
Unsupervised domain adaptation (UDA) aims at compensating the data shift between different domains by learning domain-invariant feature representations using labeled data of the source domain and unlabeled data of the target domain. Deep networks contribute to UDA by disentangling explanatory factors of variations underlying domains to learn more transferable features. Features are transferable (or domain invariant) if they are designed in such a way they can be used in both source and target domains without an accuracy drop.
More specifically, labeled source images Xs having labels Ys are drawn from a source domain distribution ps(x,y) and unlabeled target images Xt are drawn from a target domain distribution pt(x). Due to the domain shift, the data distributions of the source domain and the target domain are different, i.e. ps(x)≠pt(x). Moreover, the label space, Ys, formed by source labels Ys is often different from the target label space, Yt.
Some approaches rely on the comparison of marginal distributions between the source and target domains (i.e., where p(x,y)), the marginal distribution variables x, given as the probability distribution p(x)=Σyp(x,y), and until recently, they implicitly assumed a good class alignment between the source and target domains, such that reducing distribution divergence makes the transfer between domains easier. In the class alignment case, the two spaces are identical, Yt=Ys, and the source and target class distributions are similar, pt(y)≈ps(y). Under such an assumption, different families of domain adaptation methods have been developed, by matching the statistic moments, by optimal transportation between the domains, or by relying on domain adversarial networks. However, such an alignment assumption is invalidated in real-world scenarios where source classes are often under-represented or simply absent in the target domain. This shift between the source domain and the target domain is referred to as a class imbalance problem.
Domain adaptation may be performed in various ways. One example includes trying to match statistic moments on domain distributions, i.e. maximum mean discrepancy (MMD) and its joint and conditional variants. As another example, one may align representations in source and target domains using optimal transportation or by associating source and target data in an embedding space. As yet another example, adversarial learning may be used. Borrowing the idea of Generative Adversarial Networks (GAN), these examples encourage samples from different domains to be non-discriminative with respect to domain labels.
Class imbalance occurs when each class does not make up an equal portion of the dataset. Class imbalance in domain adaptation may be addressed based on projecting source and target data into a subspace and selecting a set of source subspaces that are most relevant to the target domain.
In the moment based family of domain adaptation methods, a weighted MMD model may be used to introduce an auxiliary weight for each class in the source domain when the class weights in the target domain are not the same as those in the source domain. The class weights may be either known or estimated from iterative soft labeling of the target instances using the EM method.
In the adversarial learning family, partial domain adaptation relaxes the fully shared label space assumption to that the source label space subsumes the target label space. A Selective Adversarial Network (SAN) may be used in order to distinguish between the relevant and irrelevant (outlier) source classes.
Another strategy to identify the importance score of source samples involves two domain classifiers, where the activations of first domain classifier are used as an indicator of the importance of each source sample to the target domain. By using these activations as weights, the weighted source samples and target samples are fed into the second domain classifier for optimizing the feature extractor.
Selection bias may be addressed using semi-supervised learning and deep learning with noisy labels. Selection bias refers to the difference between the training and the test data. Specifically, selection bias causes the distribution of collected data used in training data to deviate from the overall distribution. Predictive models may be sensitive to selection bias, and this may be corrected by adding pseudo-labeled data to the training sample, while instance weighting may be used to make training samples similar to the distribution observed in the test data.
The iterative self-training procedure, where the problem is formulated as latent variable loss minimization, can be addressed by alternatively generating pseudo labels on target data and re-training the model with these labels. However, jointly learning the model and optimizing pseudo-labels on unlabeled data is naturally difficult as it is impossible to guarantee the correctness of the generated pseudo-labels. Semi-supervised learning can correct modest selection bias, but if the domain gap is too wide, initial predictions in the target domain may be poor, and there is a risk to increase bias during the training rather than decrease it.
Class imbalance (CI) may be measured as the Kullback-Leibler (KL) divergence of target class distribution pt(x) from the source class distribution ps(y). A KL divergence of 0 indicates similar behavior of two different distributions, while a KL divergence of 1 indicates that the two distributions behave in such a different manner that the expectation given the first distribution approaches zero.
FIG. 1 illustrates the impact of class imbalance on the performance of three domain adaptation methods: optimal transport, adversarial discriminative domain adaptation (ADDA), and correlation alignment (COREL). Specifically, FIG. 1 plots the classification accuracy as the function of CI values for the six domain adaptation tasks of the Office31 dataset (shown in FIG. 7). Large circles indicate the accuracy domain class alignment, where CI values lie between 0.03 and 0.08. Comparing the UDA methods for small CI values provides little information regarding their resistance to severe class imbalance. Meanwhile, when CI values approach 1, the accuracy drop is 30% to 50%.
Under a severe class imbalance, it may be difficult to reduce the domain shift by comparing source and target distributions directly. In other words, reducing distribution shift will not benefit the target task, since the marginal distributions between domains should not be the same.
In this case, transferring from the source domain to the target domain may be accomplished by re-weighting the source domain samples whose classes are likely to appear in the target domain. Since the target domain is unlabeled, it may be challenging to uncover which classes are presented and which source domain samples are important for transferring. Success of any re-weighting schema may rely on capacity to correctly estimate the target labels. While this may be relatively straightforward in the alignment case, this may be challenging in the case of severe class imbalance.
Two main approaches to cope with the class imbalance in domain adaptation are discussed above. The first approach is based on the source classifier. Methods count on the source classifier to correctly predict the soft target labels. However, if the gap between source and target is too large, iterative optimization with soft labels often suffer from estimation error, inadequate initialization and convergence to local minima.
The second approach is to count on the discriminator and its activations as an indicator for re-weighting the source instances. However, under class imbalance, it is challenging to distinguish between a poor domain discriminator and a low class probability in the target domain.
Accordingly, there is a need in the art to address the problem of severe class imbalance in unsupervised domain adaptation, when the class spaces in source and target domains diverge considerably.
It is therefore desirable to provide a method of improved image classification that overcomes the above disadvantages. Specifically, it is desirable to provide a method of training neural networks trained on labeled images from one domain to be able to accurately classify unlabeled images from another domain.

Latent code for unsupervised domain adaptation


A technique for the simultaneous measurement, in real time, of the changes in mode size, pulse length, and power of a wavetrain of laser pulses delivered by mode-locked or other pulsed laser systems contemplates directing respective portions of the beam along each of three beam paths. The beam portions along the first and second beam paths are subjected to respective, and different, degrees of focusing, and then passed through respective non-linear elements which generate radiation at the second harmonic. The first harmonic components are removed, and the second harmonic components are passed to respective first and second detectors which generate signals representative of the total second harmonic energy along each of the first and second beam paths. The beam portion along the third beam path is passed to a detector which generates a signal representative of the total energy in the beam pulse. The three signals are combined to give values proportional to peak power, pulse length, and mode size.

Picosecond light pulses are employed in a number of applications to measure the temporal response of electronic and optical materials and devices. Such pulses are conventionally produced by phase-locking the longitudinal modes of a free-running laser oscillator. Such mode-locked lasers produce a train of short pulses with a width inversely proportional to the homogeneously broadened gain linewidth of the lasing medium and a pulse-to-pulse separation equal to the round-trip time of light through the laser or to an integral sub-multiple of this time. Such mode-locked lasers are capable of producing pulses with widths as short as 0.1-10 picoseconds at pulse-to-pulse separations of the order of 1-10 nanoseconds. The length of the pulse train produced by such a laser is determined by the lasing time of the laser oscillator, which will typically vary from about a microsecond to many seconds.
Because lasing conditions may vary with time, the energy, width, and spatial distributions of the picosecond pulses produced by such lasers may evolve in time. To effectively use these short pulses for the applications referenced above, it is necessary to identify the extent of these variations over the duration of the pulse train produced by the laser.
The characterization of ultra-short (picosecond and femtosecond) laser pulses presents a number of problems. Since conventional detectors cannot be used due to their limited bandwidth, various techniques, most of them based on the measurement of the second or higher order intensity autocorrelation functions, have been proposed and tested. These techniques can be used to reconstruct, through numerical convolution, the temporal profile of the pulses. However, they typically yield only the average of the pulse shape over the wavetrain, and do not provide any information on the possible transient evolution of either the pulse shape or mode size.

Real time analyzer for pulsed laser systems


A power supply regulation circuit is provided which can achieve a preferable constant current drooping characteristic without the need for a constant current control circuit on the secondary side. During a constant current operation, a switching element (1) is turned on so as to keep constant the on duty of secondary current passing through a secondary winding (110B) of a transformer (110), so that a constant current drooping characteristic is achieved. Further, at this moment, a current limit dictating the maximum value of drain current (ID) passing through the switching element (1) is changed between first and second levels during the on time of the switching element such that the drain current (ID) has a constant peak value.

Conventionally, switching power supplies have been widely available as a power supply regulation circuit for a charger, each of which achieves a constant current drooping characteristic by means of a constant current control circuit provided on the secondary side. The switching power supply comprises, on the secondary side, a resistor for detecting an output current, a constant current control circuit for controlling current passing through the resistor to a constant level, and a photocoupler for transmitting the signal of the constant current control circuit to the primary side. When the output current is equal to or higher than a fixed value, the constant current control circuit is operated. According to the switching power supply having such a constant current drooping characteristic, batteries or the like can be charged with a constant current by using the constant current drooping characteristic.
Japanese Patent Laid-Open No. 2003-189612 proposes a switching power supply for achieving a constant current drooping characteristic without the provision of a constant current control circuit. The conventional switching power supply will be described below.
In the conventional switching power supply, a current limit dictating the maximum value of element current passing through a switching element is reduced as the output voltage reduces during a constant current operation, so that the constant current drooping characteristic is achieved.
FIG. 16 is a block diagram showing the configuration of the conventional switching power supply.
In FIG. 16, a control circuit 200 comprises, as input/output parts, a PIN terminal serving as a start-up power supply input part, a VCC terminal serving as an auxiliary power supply voltage input part, a CS terminal serving as a current detection/input part, an OUT terminal serving as a switching element driving/output part, and a GND terminal serving as the GND output part of the control circuit. Further, the control circuit 200 comprises an oscillator circuit. The control circuit 200 drives the gate of a switching element (MOSFET) 210 connected to the OUT terminal and controls the switching operation of the switching element 210.
The switching element 210 controls the on/off of current passing through the switching element 210 from a primary winding 220A of a transformer 220. Element current (drain current) ID passing through the switching element 210 forms a triangular wave which inclines in proportion to input voltage VIN due to the inductance of the primary winding 220A.
A rectifying/smoothing circuit comprising a diode 231 and a capacitor 232 converts alternating-current power, which is generated on a secondary winding 220B of the transformer 220 in response to the switching operation of the switching element 210, to direct-current power and supplies the power to a load 233.
A rectifying/smoothing circuit comprising a diode 241 and a capacitor 242 is used as the auxiliary power supply part and the output voltage detection part of the control circuit 200. The rectifying/smoothing circuit converts alternating-current power, which is generated on an auxiliary winging 220C of the transformer 220 in response to the switching operation of the switching element 210, to direct-current power and supplies the power to the VCC terminal through a resistor 243 and a capacitor 244. An alternating-current voltage generated on the auxiliary winging 220C is proportionate to an alternating-current voltage generated on the secondary winding 220B.
The control circuit 200 detects the drain current ID passing through the switching element 210, based on the voltage of the CS terminal fed with voltage generated on a resistor 250. The control circuit 200 has an overcurrent protection function for preventing overcurrent in the switching element 210. The overcurrent protection function is provided for automatically turning off the switching element 210 when a peak value Ip of the drain current ID increases to a current limit ILIMIT.
When the voltage of the VCC terminal (VCC voltage) is higher than a fixed voltage, the control circuit 200 reduces the on duty of the switching element 210 as the VCC voltage increases, so that a constant voltage characteristic is achieved.
Conversely when the VCC voltage is lower than the fixed voltage, the control circuit 200 reduces the current limit ILIMIT as current passing through the VCC terminal decreases, so that a constant current drooping characteristic is achieved. In this case, in order to keep an output current IO constant, the current limit ILIMIT is changed as a function of current passing through the VCC terminal.
Referring to the accompanying drawings, the following will describe the operations of the switching power supply thus configured. FIG. 17 is a timing chart showing the operations of the parts of the conventional switching power supply.
When the VCC voltage is higher than a fixed voltage VCC_A, the switching power supply reduces an on duty Don of the switching element 210 as the VCC voltage increases, so that a constant voltage characteristic is achieved.
When a load increases, the peak value Ip of the drain current ID passing through the switching element 210 is increased to the current limit ILIMIT, output power PO is maximized, and then the output voltage VO is reduced, the VCC voltage decreases in response to the reduction.
Thereafter, when the VCC voltage falls below the fixed voltage VCC_A, the switching power supply reduces the current limit ILIMIT according to a reduction of current passing through the VCC terminal, so that a constant current drooping characteristic is achieved. In this case, in order to keep the output current IO constant, the switching power supply changes the current limit ILIMIT as a function of current passing through the VCC terminal.
As described above, in the conventional switching power supply which achieves a constant current drooping characteristic without the need for a constant current control circuit, the maximum output power is determined by the current limit ILIMIT and an overload is detected.
In an actual circuit, however, the peak value Ip of the drain current ID passing through the switching element is higher than the current limit ILIMIT. This is because the switching element is not turned off immediately after the detection of the drain current ID having increased to the current limit ILIMIT. To be specific, a certain delay time called an overcurrent detection delay time Td is present between the detection of the drain current ID having increased to the current limit ILIMIT and the time when the switching element is actually turned off.
FIG. 18A shows the waveforms of the drain current ID when the current limit ILIMIT is constant relative to a given input voltage VIN. As described above, the drain current ID forms a triangular wave which inclines in proportion to the input voltage VIN. Meanwhile, the overcurrent detection delay time Td is constant. Thus when the current limit ILIMIT is constant relative to a given input voltage VIN, an increase in the current value of the drain current ID inevitably changes with the input voltage VIN in the overcurrent detection delay time Td. In other words, even when the current limit ILIMIT is constant, the peak value Ip of the drain current ID changes with the input voltage VIN. To be specific, as shown in FIG. 18A, a high input voltage VIN increases the inclination of the drain current ID and the peak value Ip of the drain current ID. A low input voltage VIN reduces the inclination of the drain current ID and the peak value Ip.
Therefore, in the case of a high input voltage VIN, the maximum value of the drain current ID is high as compared with a low input voltage VIN, and the maximum output power PO increases. This means that overload detection moves to the heavy load side. Thus, as shown in FIG. 19A, in the case of a high input voltage VIN, the output current IO in a constant current operation is high as compared with a low input voltage VIN. This relationship will be described below using the equation of the output power PO.
The output power PO in a discontinuous mode is expressed as below:PO=A×L×Ip2×foscwhere 'fosc' represents the oscillation frequency of the switching element, 'A' represents a constant, 'L' represents the inductance of the primary winding of the transformer, and Ip represents the peak value of the actual drain current ID.
As is evident from this equation, as long as the oscillation frequency fosc and the peak value Ip of the drain current ID passing through the switching element are constant, the output power PO remains constant even when the input voltage VIN changes. On the other hand, even when the oscillation frequency fosc remains constant, the output power PO changes with a change of the peak value Ip of the drain current ID. Therefore in the case where the current limit ILIMIT is constant relative to a given input voltage as described above, the maximum value of the drain current ID determined by the current limit ILIMIT changes with the input voltage VIN and thus the maximum output power PO also changes. Hence, as shown in FIG. 19A, in the case of a high input voltage VIN, overload detection moves to the heavy load side and the output current IO in a constant current operation is high as compared with a low input voltage VIN.
As described above, when the current limit ILIMIT is constant, the maximum value of the drain current ID changes with the input voltage VIN and the maximum output power PO also changes. Thus, the output current IO changes with the input voltage VIN in a constant current operation.
In order to solve this problem of the conventional switching power supply, the following technique is used: as shown in FIG. 20, in the conventional switching power supply, the current limit ILIMIT is linearly increased for a certain time after the switching element is turned on, and then the current limit ILIMIT is reduced to a value obtained when the switching element is turned on.
When the inclination of the increase of the sawtooth current limit ILIMIT is set at a proper value, as shown in FIG. 18B, a high input voltage VIN has a low current limit ILIMIT relative to the drain current ID, and a low input voltage VIN has a high current limit ILIMIT relative to the drain current ID. Therefore a change in the peak value Ip of the drain current ID according to the input voltage VIN decreases and thus a change in the output current IO also decreases in a constant current operation.
Strictly speaking, when the overcurrent detection delay time Td is constant, the current limit ILIMIT which keeps the peak value Ip of the drain current ID constant relative to a given input voltage VIN does not linearly change with time. In other words, it is necessary to change the current limit ILIMIT as expressed in the equation below:ILIMIT(t)=Ip×(t−Td)/twhere 't' represents a time from when the switching element is turned on.
A current limit ILIMIT1 shown in FIG. 21 represents a current limit ILIMIT which sets, when the overcurrent detection delay time Td is 150 ns, the peak value Ip of the drain current ID at 1 A regardless of an input voltage. As shown in FIG. 21, the current limit ILIMIT1 is a function which simply increases and forms a convex shape with a primary time derivative acting as a positive derivative and a secondary time derivative acting as a negative derivative. The current limit ILIMIT1 does not linearly change with time.
On the other hand, a current limit ILIMIT2 is a current limit ILIMIT which linearly changes with time. The current limit ILIMIT2 is set such that the peak value Ip of the drain current ID has an error of about ±3% when the overcurrent detection delay time Td is 150 ns and an on time Ton of the switching element changes over a range of 1.0 to 4.5 μs. Further, a peak value Ip2 is an actual peak value Ip when the current limit ILIMIT2 serves as the current limit ILIMIT.
In this case, the on time Ton is a period during which the switching element is turned on. Strictly speaking, the on time Ton is a time during which the switching element is turned on, the drain current ID increases to the current limit ILIMIT, and then the switching element is turned off after a delay of the overcurrent detection delay time Td.
As shown in FIG. 21, even the current limit ILIMIT2 linearly changing relative to the on time Ton of the switching element can reduce the dependence of the peak value Ip of the drain current ID on the input voltage within a predetermined range of the on time Ton. The range of the on time Ton (1.0 to 4.5 μs) in this example is a 4.5-times range which can sufficiently respond to, for example, the input voltage range of world wide input (AC 85 to 282 V). Thus in the conventional switching power supply, the inclination of the increase of the current limit ILIMIT is set at a proper value, for example, like the current limit ILIMIT2 shown in FIG. 21.
However, in the case where the on time Ton is short relative to the overcurrent detection delay time Td, that is, "Td/Ton" is large, it is difficult to reduce the input voltage dependence of the peak value Ip of the drain current ID passing through the switching element. Thus the long overcurrent detection delay time Td and the short on time Ton are disadvantageous to a reduction of the input voltage dependence of the peak value Ip of the drain current ID. The input voltage dependence is reduced by the current limit ILIMIT linearly changing with time. This is proved from a fact that the peak value Ip2 of FIG. 21 is considerably deviated from 1 A at the on time Ton of 1 μs or less.
In the conventional switching power supply, as shown in FIG. 17, the current limit ILIMIT is reduced to supply lower energy to the secondary side, so that a constant current drooping characteristic is achieved. Therefore, in the conventional switching power supply, the peak value Ip of the drain current ID passing through the switching element is reduced and the on time Ton is shortened.
As described above, as the on time Ton becomes shorter, the peak value Ip of the drain current considerably changes. Thus in the conventional switching power supply, it is quite difficult to make the output current IO constant relative to a change of the input voltage VIN and a load.
FIG. 18C shows the waveforms of the drain current ID when the current limit ILIMIT decreases and the peak value Ip of the drain current ID passing through the switching element decreases.
As shown in FIG. 18C, in the case where the peak value Ip of the drain current ID decreases, the on time Ton of the switching element is shortened, and the inclination of the current limit ILIMIT is the same as the current limit ILIMIT shown in FIG. 18B, the peak value Ip of the drain current ID changes with the input voltage VIN. For this reason, in the conventional switching power supply which reduces the sawtooth current limit ILIMIT during a constant current operation and achieves a constant current drooping characteristic, it is expected that the constant current drooping characteristic is shaped like FIG. 19B. In other words, it is expected that the output current IO changes with the input voltage VIN when the output voltage VO decreases.

Power supply regulator circuit and semiconductor device

Matsushita Electric Industrial Co., Ltd.


Disclosed is a measurement method for a glycated hemoglobin ratio. According to a measurement method for a glycated hemoglobin ratio of the present invention, reagents are easy and convenient to use because of their sequential leakage during the rotation of a cassette, are all discharged by the rotation with no remaining reagent, and are not mixed with each other. Therefore, measurement results are accurate, with fewer errors in the quantities of used reagents and sample blood.

In the fields of medical diagnosis or drug-based therapy, the concentration measurement of analytes of anesthetics or harmful chemicals has recently been useful in medical or environmental fields. Above all, the concentration measurement of biological samples used in the fields of medical diagnosis and therapy is increasingly drawing interest along with an increase in human desire to be free from various diseases. Particularly, with regard to diabetes, the glycated hemoglobin test capable of measuring blood sugar allows a relatively long-term average value of the blood sugar to be detected by one-time measurement, and thus has an increasing interest.
Hemoglobin A1c (HbA1c) is also called glycated hemoglobin, and is present in human red blood cells as a part of hemoglobin. When a concentration of blood sugar (glucose) in the blood rises, a glucose moiety in the blood binds to hemoglobin. This hemoglobin conjugated with glucose is referred to as glycated hemoglobin. Blood sugar levels can be determined by this glycated hemoglobin test, which has an advantage in that it can be conducted by collecting blood regardless of the meal time.
Meanwhile, U.S. Pat. No. 6,300,142 discloses an apparatus for reacting a test sample with a first reactant in a first inlet port and sequentially reacting the reacted test sample with a second reactant in a second inlet port to measure the analyte present in the test sample. In this case, the measurement of the analyte has to be conducted periodically and sequentially. Further, a user has to intervene in the measuring process in such a manner that he or she injects the test sample sequentially to react the test sample with other materials. Furthermore, since beads conjugated with the glycated hemoglobin have to be filtered once, the measuring process is complicated and takes a long time. That is, since the conventional measuring process requires the user's direct intervention in various processing steps, the user may feel inconvenient. Also, the user's direct intervention makes the measuring process more complicated, thereby further increasing the measuring time.
Therefore, there is a demand for research on the method for measuring glycated hemoglobin ratio, which is easy to measure and can provide accurate measurement results.

Measurement method for glycated hemoglobin ratio


A method and apparatus for data recovery in a system involving a first data store acting as a standard device and a physical moving mirror data store that operates as moving mirror with a first mode to be synchronized and in a second, isolated mode. In response to a command to establish a third or protected restore operating mode, the data to be transferred in response to that command is identified. A restoration procedure copies data from the second data store to the first store to recover any data that may have been corrupted in the second data store. An update procedure acts on the restored data concurrently with the restoration procedure.

1. Field of the Invention
This invention generally relates to the maintenance of data redundancy in a data storage facility and more particularly to recovering data in such a facility.
2. Description of Related Art
Many data processing applications now operate on digital data processing systems that incorporate large scale memory facilities. These large scale memory facilities take many forms. One such form is a disk array storage device (DASD) in which data is stored at a local large scale disk memory with many physical disk drives. In other applications it may be desired to distribute the functions of equivalent devices over a network at different network nodes in a local-area or wide-area network.
When data integrity is critical, such large scale disk memory facilities include some type of data redundancy. Memory mirrors provide on-site redundancy to protect against equipment failure. U.S. Pat. No. 5,206,939 to Yanai et al. and assigned to the same assignee as this invention discloses a remote system that normally acts as a mirror of a local system. The geographical separation between the local and remote systems provides an extremely high degree of reliability, particularly as protection against a natural disaster.
More recently U.S. Pat. No. 6,101,497 granted Aug. 8, 2000 discloses a tool that is particularly well adapted for facilitating redundancy. This device, called a BCV device, establishes a special physical disk structure that acts as a moving physical mirror to attach to and to synchronize with a standard device with one or more fixed physical mirrors without interfering with normal operations between an application program being run by a host using data on the standard device. Once the data in this BCV moving mirror is synchronized with the data in the standard device, it can be split or isolated from the standard device and thereafter be available for backup or for use by another application.
As known, data on a standard device can, for a variety of reasons, become corrupt. As described in the foregoing U.S. Pat. No. 6,101,497, a restore or an incremental restore command initiates a restoration procedure that copies data from the redundant BCV physical mirror to the standard device, assuming that data on the BCV physical mirror has not been changed. For example, a BCV physical mirror used as a source for a tape backup will maintain the data without change. In this situation, the data on the BCV physical mirror represents the data that existed when the BCV physical mirror was isolated from the standard device. So it does not represent the data that actually exists on the standard device if an event occurs that requires restoration; rather it represents the data that existed at some earlier point in time.
However, many databases and other applications maintain a log file, or like file that records each transfer and the sequence of each transfer. Entries from such a file from a point in time after the prior isolation can be used to recover or update the data fully on the standard volume. One such update procedure is known as xe2x80x9cunrolling the redo logsxe2x80x9d. During this procedure, the BCV device may be attached to the standard device. As each log entry is processed, a write operation transfers the data associated with that log entry to both the standard device and the BCV physical mirror. Consequently, after the first write operation occurs, the data on the BCV physical mirror is no longer identical to the data that had been on the BCV physical mirror at the time of a prior split.
Oftentimes errors can occur while performing an update procedure like unrolling the redo logs thereby corrupting the data on the standard device and, due to the mirroring function, the data on the BCV physical mirror. When this occurs it may be required to recover data from a tape backup made even earlier in time and again try to reconstruct the current data by unrolling the redo logs for a longer time interval. Data recovery involving tape backups can be time consuming and tedious. What is needed is a process by which the data used on a mirror for restoration, like data on a BCV physical mirror, is not altered until an update procedure, such as unrolling the redo logs, has been completed without error.
With prior data recovery operations, data from a BCV physical mirror is restored to a standard device before an update procedure, such as unrolling the redo logs, is initiated. Conducting the restoration and update procedures in series can result in long downtime for a production volume. What is needed is a data recovery operation that enables the restoration and update procedures to occur concurrently.
Therefore it is an object of this invention to improve a facility for recovering data from a redundancy device.
Another object of this invention is to provide a facility for improving data recovery involving specially configured physical mirrors that can interact with multiple applications.
Still another object of this invention is to provide a data recovery operation that assures the integrity of data used in a restoration and stored on a device providing the restoration data.
Yet another object of this invention is provide a data recovery procedure during which restoration and update procedures operate concurrently.
This invention facilitates the recovery of data that is characterized by a restoring procedure for copying data to a first data store for a primary copy of the data from a second data store. The second data store operates in a first mode as a mirror to the first data store and in a second mode isolated from the first data store. An update procedure updates data on the first data store. The data recovery operation further includes preventing data transfers to locations in the second data store during the data recovery procedure. Restoration of the data includes identifying locations in the first data store and copying data to the identified locations in the first data store from corresponding identified locations in the second data store.


Provided are a data storage device and a method of tracking data stored in the data storage device. The device includes a data storage medium on which data can be recorded and erased, a plurality of probes for scanning the data storage medium to detect data, a scanner for driving the data storage medium or the probes, and a controller for applying a control signal to the scanner. The method includes (a) oscillating the probe or the data storage medium; (b) detecting an off-track error of the probe on the data storage medium by the oscillation; and (c) adjusting a scanning position of the probe on the data storage medium to compensate for the off-track error.

1. Field of the Invention
The present invention relates to a data storage device and a method of tracking data stored in the data storage device, and more particularly, to a method of finely tracking a position of data recorded in a data storage medium using scanning probe microscopy (SPM) technology.
2. Description of the Related Art
Generally, a high-density data storage device using SPM technology includes a data storage medium, a probe, an x-y stage, a controller, and a signal processor. Data can be stored in or erased from the data storage medium. The probe includes a tip for recording or reproducing data and a cantilever on which the tip is mounted. The x-y stage moves the data storage medium, and the controller issues commands to control the data storage device. To record or reproduce high-density data, a method of tracking data with high precision is required.
Three different types of conventional methods of tracking a position of data have been employed to record data on or reproduce data from such a high-density data storage device using SPM technology.
First, the position of data can be tracked using a mechanical or electric signal of a data storage medium by distinguishing a signal of recorded data from a signal of reproduced data. This method is proposed in U.S. Pat. Nos. 5,132,934, 5,396,483, 5,856,967, and 6,370,107. The foregoing patents disclose a method of forming metal patterns or mechanical grooves on a data storage medium, but it is difficult to form sufficiently fine patterns or grooves on the data storage medium. Also, since the mechanical grooves themselves serve as a mechanism for recording and reproducing data, the practical applications of this method are limited.
Second, there is a method of modulating a central position of data recorded in a data storage medium by oscillating a probe or the data storage medium, as disclosed in U.S. Pat. No. 5,404,349. This method is based on a principle of modulation and demodulation by oscillating the data storage medium in a high frequency range. However, it is difficult to oscillate the data storage medium in the high frequency range. In addition, a circuit for detecting the oscillation of the data storage medium becomes complicated and detection is delayed owing to the principle of modulation and demodulation.
Third, a method of using a central position of data stored in a data storage medium and a relative displacement of a probe for recording or detecting the data is disclosed in U.S. Pat. Nos. 5,202,879 and 6,195,313, for example. Unfortunately, this method necessarily requires very precise recording of data patterns in the data storage medium. In particular, the method proposed in U.S. Pat. No. 5,202,879 cannot serially track data. Accordingly, the data can only be tracked at a relatively low speed. Thus, a data tracking process becomes very sensitive to the external environment.

Data storage device and method of tracking data stored in the same


Techniques pertaining to scalable video codec are disclosed. According to one aspect of the present invention, a video image is analyzed and a region of interest (ROI) and a region of non-interest (non-ROI) are identified. By comparing the non-ROI image with that of a previous image, a background ignored identifier is created indicating whether the non-ROI can be ignored during encoding and decoding processes. Based on the status of the background ignored identifier, the encoder encodes the images into a basic layer (BL) and an enhanced layer (EL), and transmits the coded bit streams along with the identifier to a decoder. The decoder reconstructs the image based on the identifier and the BL and the EL bit streams.

1. Field of the Invention
The present invention relates to image processing fields, more particularly to video codec methods and systems.
2. Description of Related Art
With continuous advancements in electronics and information technologies and use of internet, remote video surveillance techniques have been developed and used in various applications, e.g., traffic management, security monitoring, or forest fire monitoring. However, users in different types of networks may have different requirements for the video images because different types of networks have different bandwidth performances. Hence, scalable video coding techniques are developed to meet the above requirements.
The scalable video coding (SVC) technique is provided to encode image frames of a video sequence in multiple layers. The first layer, also called base layer (BL), contains a minimum required image data, and the remaining layers, also called enhancement layers (EL), contain the refinements to the data carried by the base layer. This technique makes scalability possible as a receiver can choose to receive only the layers desired and ignore the rest depending on its required resolution and the bit transfer rate capacity of the network.
The spatial scalability is one type of scalable video coding techniques capable of encoding an image frame of a video sequence into two layers at the same frame rate but in different spatial resolutions. The original image of the video sequence is down-sampled to a low spatial resolution base layer image, and the original image is to be coded as the enhancement layer depending on the base layer. Subsequently, the bit stream of the coded base layer is decoded independently, and the bit stream of the coded enhancement layer is decoded based on the reconstructed base layer image. Referring to FIG. 1, there is a block diagram showing a spatial SVC coder in the prior art. An original whole image of size 4xCIF (Common Intermediate Format) is down-sampled by a whole-image down-sampling module 102 to generate a BL whole image of size CIF. A whole-image BL encoder module 106 encodes the BL whole image and generates a BL bit stream and BL data. A whole-image EL encoder module 104 encodes the original whole image and generates an EL bit stream.
In some video monitoring applications, it is only necessary to pay attention to special regions (e.g., doors and windows etc.) in an image and ignore the background regions. In other words, the content of the regions of interest (ROI) may be retained as detailed as possible, while the details of the region of non-interest (non-ROI) may be ignored.
In the current video SVC codec technique, the coding and decoding are performed on the whole video image. But the resolution of the BL image is relatively low, resulting in the loss of details of the image. If the whole image is encoded as the enhancement layer to improve the resolution, the network bandwidth is required to increase to accommodate the higher bit transfer rate. Applications of current video codec technique is, therefore, limited either by the resolution of the image or the bandwidth of the network.
Thus, improvements on the current video codec techniques are desired to overcome the above-mentioned disadvantages and others.


A three-dimensional printing system includes a print engine and a controller. The controller in turn includes a processor coupled to a non-transient memory. The non-transient memory stores software instructions that when executed by the processor cause the controller to operate the print engine to fabricate a body. The body includes a three-dimensional arrangement of 3D articles that are pairwise coupled together. An individual pair of the arrangement of 3D articles includes a first article and a second article. A curved body couples a first surface of the first article to the second article. The curved body intersects the first surface to define a closed intersection curve that bounds a break surface. Along the closed intersection curve a surface of the curved body subtends an acute angle with the first surface.

Additive manufacturing is in wide use for prototyping and manufacture of high value articles. One challenge facing a wider adoption of additive manufacture is cost and quality compared to alternatives. A key advantage of additive manufacturing is a rapid turn-around of design changes and customization. There is an ever present desire to bring the advantages of additive manufacturing to higher volume manufacturing applications.

High volume manufacturing method for precision articles based on three-dimensional printing