text:
A mobile radio telephone having a simplified architecture including a reduced number of integrated circuits and a reduced number of RF connections. A dual band, dual mode mobile phone can be constructed using either a single crystal reference oscillator or two crystals such that alternative symbol rates, channel spacings, or transmit/receive duplex spacings can be achieved.
1. A dual-mode cellular radiotelephone apparatus for receiving and transmitting coded and modulated information symbols at a first symbol rate alternately at a second symbol rate comprising:
a dual reference frequency oscillator responsive to a select signal to oscillate at a first accurate frequency or a second accurate frequency according to the select signal level and to provide an output signal of the selected frequency;
a digital signal processing using said selected reference oscillator frequency signal to generate said coded and modulated information symbols for transmission at said first rate or said second rate in dependence on the selected reference frequency;
a second local oscillator frequency synthesizer producing an accurate second local oscillator frequency signal phase-locked to said selected reference frequency signal;
a first local oscillator frequency synthesizer producing an accurate first local oscillator frequency signal;
a transmit oscillator producing a signal at a desired frequency of transmission in dependence on a control signal;
a transmit oscillator frequency control comprising transmit mixer mixing said transmit frequency signal with said first local oscillator signal to produce a transmit offset frequency signal and transmit phase lock loop for comparing said transmit offset frequency signal with said second local oscillator frequency signal to generate said control signal.
2. The apparatus of
3. The apparatus of
4. The apparatus of
5. The apparatus of
6. A dual-mode digital cellular radiotelephone comprising:
a reference oscillator oscillating at an accurately controlled frequency of 19.5 MHz;
a transmitter generating a signal having a transmitted bit rate derived from said reference oscillator frequency by division by an integer factor of 72, and alternatively generating a signal having a symbol rate derived from said reference oscillator frequency by dividing by a non-integer factor.
7. The radiotelephone of
This application is a divisional, of application Ser. No. 08/974,227, filed Nov. 19, 1997 now U.S. Pat. No. 6,308,048.
The invention relates to mobile communications using a selected pair of transmit and receive frequencies and relates to the derivation of transmit and receive frequency steps and digital clock rates from a common crystal reference oscillator.
It is well known in the art of mobile radiotelephones to employ a receiver for receiving a receive frequency signal while a transmitter simultaneously transmits a transmit frequency signal in the other direction, the transmit frequency being separated from the receive frequency by a constant offset known as the duplex spacing.
Although the duplex spacing is nominally a constant, it can be a different constant depending on the frequency band in which the mobile phone is operating. Complications can then arise in constructing mobile phones that operate in more than one frequency band.
U.S. patent application Ser. No. 08/795,930 entitled "Transmit Signal Generation with the Aid of Receiver" (Dolman) describes the use of the second local oscillator of the receiver as a reference frequency against which a transmit frequency is controlled relative to a receive frequency to achieve either a first or a second duplex spacing. The Dolman application is hereby incorporated by reference.
It is also known in the prior art, when packaging two synthesizer PLL circuits into a common integrated circuit, to synchronize or otherwise relate the reference dividers of the two PLLs so that their phase comparators do not mutually interfere. The Philips UM1005 and 8026 dual synthesizer integrated circuits available on the open market use this technique. These circuits include the use of fractional-N dividers and programmable loop bandwidth, such as described in U.S. Pat. Nos. 5,095,288 and 5,180,993 which are hereby incorporated by reference. Novel ways to employ such synthesizers in dual mode satellite/cellular telephones in order to achieve different tuning step sizes in different frequency bands are described in U.S. Pat. Nos. 5,535,432 and 5,610,559 which are also hereby incorporated by reference.
Continuous advances in electronics allow for smaller mobile phones complying with a variety of national and international protocols. The international mobile phone standard known as GSM in Europe and as PCS 1900 in the USA operates with a transmit/receive duplex spacing of 45 MHz in the European 900 MHz band; 95 MHz in the European 1800 MHz band, and 80 MHz in the U.S. 1900 MHz PCS band. The channel spacing is 200 KHz (13 MHz/65) and the transmitted symbol rate is 13 MHz/48. All timing in this standard is related to a 13 MHz clock, as is well known. The U.S. IS 136 system known as DAMPS operates with a 45 MHz duplex spacing in the US 800 MHz cellular band, and with an 80.4 MHz duplex spacing in the U.S. 1900 MHz PCS band, with a tuning step size of 30 KHz and a transmitted symbol rate of 24.3 Kilosymbols/sec. In IS 136, as is well known, the tuning step sizes and symbol rates and internal timing are all derivable from a 19.44 MHz clock. Yet another U.S. standard known as IS95 uses Code Division Multiple Access at a transmitted chip rate of 1228.8 MHz, with a duplex spacing of 45 MHz combined with tuning steps of 30 KHz in the 800 MHz band, alternatively 50 KHz steps combined with 80 MHz duplex spacing in the 1900 MHz band. In IS95, the chip rate and frequency step sizes are not easily derivable from the same crystal oscillator. It may be easily understood that combining two or more of the abovementioned protocols in the same handheld unit is hindered by the variety of tuning step sizes, duplex spacings and symbol rates that must be synthesized. Consequently, there exists a need for an improved radio architecture to facilitate such combination.
A mobile phone receiver according to the invention comprises a first superheterodyne downconversion means using a first local oscillator frequency which can be tuned in frequency steps by a programmable digital frequency synthesizer phase lock loop (PLL). The first downconversion means converts received signals to a first intermediate frequency (IF) for filtering. A second downconversion means using a second local oscillator converts first IF signals to a second IF or to the complex baseband for further filtering and processing. The second local oscillator is generated using a second digital frequency synthesizer PLL which locks the second oscillator to a crystal reference oscillator. The crystal reference oscillator provides a buffered clock output signal from which digital logic derives transmit symbol rates and receiver processing sampling rates.
According to a first aspect of the invention, the second local oscillator provides a buffered output signal at the second local oscillator frequency. The buffered output signal is used as the reference frequency for the first local oscillator's synthesizer PLL, thus eliminating the need to distribute the crystal oscillator signal to the first oscillator's PLL circuit. According to a second aspect of the invention, the first oscillator PLL comprises a phase comparator to compare the divided down first local oscillator signal with the divided down reference frequency signal from the second local oscillator, the divided down frequencies being equal to the desired receiver frequency tuning steps or a multiple thereof. It should be appreciated that this frequency would not have been available by dividing down the crystal frequency in an integral ratio without practicing this aspect of the invention.
According to a third aspect of the invention, a third digital frequency synthesizer PLL controls the transmitter frequency to be equal to the first local oscillator frequency plus or minus a transmit offset frequency. The transmit frequency can for example be heterodyned with the first local oscillator frequency to produce a transmit offset frequency signal; the transmit offset frequency signal is then divided down in a digital divider and compared with a phase reference frequency which is also derived by dividing the second local oscillator frequency by an integer factor.
Since according to the third aspect of the invention, the transmit offset synthesizer PLL and the first local oscillator PLL both utilize the second local oscillator as a common frequency reference, they can furthermore be packaged in a common integrated circuit and can share at least part of the reference divider which divides the second local oscillator frequency to produce a first and second phase comparator reference frequency signal for the two PLLs respectively. The two PLLs' respective phase comparators are arranged to respond to opposite polarities of a signal at a lowest common multiple frequency of their respective first and second phase comparator reference signals in order to minimize mutual interference between the two PLLs.
The invention will be more fully understood upon reading the following description with reference to the accompanying drawings, in which:
FIG. 1 illustrates prior art reference frequency distribution;
FIG. 2 illustrates another prior art scheme;
FIG. 3 illustrates the improved system described in the incorporated Dolman reference;
FIG. 4 illustrates a new inventive reference distribution scheme;
FIG. 5 illustrates more detail of the inventive frequency synthesis scheme;
FIG. 6 illustrates an inventive dual band scheme using frequency doublers
FIG. 7 illustrates an inventive dual band scheme using frequency halvers;
FIG. 8 illustrates an inventive scheme using one frequency halver and one frequency doubler;
FIG. 9 illustrates an inventive scheme in which the frequency halver and the frequency doubler are switched in position;
FIG. 10 illustrates an inventive scheme in which a frequency doubler can be powered down for AMPS reception;
FIG. 11 is a modification of FIG. 10, using frequency halvers;
FIG. 12 illustrates an alternative reference frequency distribution for dual-mode radios
FIG. 13 illustrates divider ratios for the dual-mode radio of FIG. 12;
FIG. 14 illustrates a dual-mode radio using I.F. Homodyne for a PCS1900 mode;
FIG. 15 illustrates a dual-mode radio using a single crystal;
FIG. 16 illustrates divider ratios for the dual-mode radio of FIG. 15;
FIG. 17 illustrates a dual-mode radio using two reference crystals;
FIG. 18 illustrates divider ratios for the dual-mode radio of FIG. 17;
FIG. 19 illustrates divider ratios to eliminate the second crystal of FIG. 17; and
FIG. 20 illustrates a skip-counter for generating 194.4 KHz from 19.5 MHz.
Referring now to FIG. 1, a prior art cellular phone comprises an antenna (10) connected to a receiver and a transmitter by means of a transmit/receive duplexor (11). When simultaneous transmit and receive (frequency duplex) is used, as in the analog FM AMPS standard or the IS95 CDMA standard, the duplexor (11) is a duplexing filter. Alternatively, for a TDMA system such as GSM/PCS1900 or D-AMPS/IS136 that employs time-duplex, the duplexor can be a T/R switch. For dual band phones employing frequency duplex in one band and time duplex in another band, duplexor (11) can be a dual-band duplexor having both a switch and a duplexing filter. When frequency duplex is used in both bands, duplexor (11) could comprise duplexing filters for both bands, and when time duplex is used in both bands, a single T/R switch might serve for both bands.
The duplexor allows the transmitter to be connected to the antenna without affecting receiver sensitivity. The receiver comprises a low noise amplifier and downconvertor known as "the front end", (12). The front end can be fabricated in a single integrated circuit comprising a low-noise amplifier, a downconverting and possibly image-rejecting mixer and a first local oscillator, for each of two or more different frequency bands (such as the 800 MHz and 1900 MHz bands).
The first local oscillator mixes with the desired receive frequency signal to produce a first intermediate frequency signal. Filtering can take place with a fixed frequency bandpass filter, IF filter (15). The desired receive frequency is selected by tuning the local oscillator to a frequency equal to the sum or the difference of the desired receive frequency and the first IF, by means of first local oscillator synthesizer phase lock loop (14). The 1st LO PLL tunes the 1st LO to a programmable integer multiple of a basic tuning step size, which is derived from a crystal reference oscillator (21) by dividing the crystal frequency by another integer to get the step size. For small step sizes, the synthesizer may alternatively derive a greater step size from crystal oscillator (21) by dividing by a smaller integer, and then interpolate between these larger steps to get the desired smaller steps using the technique of fractional-N synthesis described in the above-incorporated references. First LO PLL circuit (14) compares the 1st LO frequency to the crystal reference signal and generates an error signal. The error signal is filtered and integrated in Loop Filter (24) to produce a control signal to control the oscillator frequency until the frequency is exactly as intended.
The receiver amplifies the filtered 1st IF signal and then customarily performs a second frequency downconversion using a second heterodyne mixer and second local oscillator. The IF amplifiers, second local oscillator and second mixer are all contained in a conventional second integrated circuit (16). After downconverting a second time to a second or final Intermediate frequency, further amplification may take place at the final IF and a detector circuit can be employed to produce a Radio Signal Strength Indication (RSSI) related to the strength of the received signal. The second IF amplifier may be hardlimiting, and then outputs a hardlimited final IF signal to digital signal processing (20) where phase information is extracted and digitized using the second IF signal simultaneously with digitizing the RSSI signal, as described in U.S. Pat. No. 5,048,059 entitled "Log-polar signal processing", which is hereby incorporated by reference herein. The second local oscillator part of IF amplifier circuit (16) is also controlled to the desired frequency by means of a synthesizer PLL circuit (17) and loop filter (23). The 2nd LO frequency is compared with the crystal oscillator (21) and an error signal produced as before. Thus both synthesizer circuits (14) and (17) use the crystal as the frequency reference or standard of accuracy for controlling both the first and second LO. The digital signal processing logic (20) can also require an accurate frequency standard for producing receiver sampling and processing rates and transmit symbol rates, and so is also fed with an output from crystal oscillator (21).
The transmitter comprises transmit frequency generation circuit (19) for producing a signal offset from the receive frequency by the fixed duplex spacing. The transmit frequency is thus offset from the 1st LO frequency by the duplex spacing combined with the first intermediate frequency, which nevertheless is still a constant offset. The constant transmit offset is either equal to the first IF minus the duplex spacing or the first IF plus the duplex spacing, depending on whether the 1st LO is lower or higher than the receive and transmit frequencies.
The transmit frequency signal is then modulated with information from digital signal processor (20) using modulator (18), which is for example a quadrature modulator having I and Q input signals. The modulated signal is then amplified to a transmit power level using power amplifier (13), which may be a dual-band power amplifier in a dual-band phone.
Transmit offset PLL forms the difference between the transmit and 1st LO frequencies and tests to see if it equals the desired offset, by comparing the offset with the crystal reference. The TX offset PLL thus also has need of a crystal reference frequency signal from oscillator (21), making four places to which the oscillator signals must be distributed.
The four separate outputs from oscillator (21) must be sufficiently isolated from one another by buffer amplifiers and conditioned to drive copper tracks on the printed circuit motherboard. This consumes battery power and presents a radiated interference hazard. Often, to save battery power during standby, outputs which are momentarily not required, such as that feeding the transmit offset PLL, may be turned off by control signals from a control processor (part of digital signal processing 20), which is a further complication. It is thus desirable to reduce the distribution of the crystal reference signal by means of printed circuit board tracks to multiple destinations.
A first step in this direction is already taken in prior art products sold by Ericsson in the USA. By combining the first and second LO PLLs in a single chip, such as the Philips UM1005 or 8026 part, a single input for the crystal reference may be used, since both use the crystal as the reference. Further, by combining crystal oscillator (21) into a transmit signal generator chip together with TX offset PLL and modulator (18), no external output connection between oscillator (21) and offset PLL (19) is needed.
Thus, referring to FIG. 2, the number of crystal reference signal outputs required is reduced to two, one feeding dual synthesizer circuit (14+17) and the other feeding digital signal processing (20).
In U.S. patent application Ser. No. 08/795,930 to Dolman, incorporated above, it is explained that all PLLs desirably operate by dividing the crystal reference frequency by the smallest possible integer for comparison with the frequency of the oscillator they are controlling, which is also divided by the smallest possible integer. Expressing this another way, it is desired to have the largest possible common factors between the controlled oscillator frequencies and the reference frequency. Dolman discloses that this is facilitated when the transmit offset frequency is generated using the 2nd LO as the reference frequency rather than the crystal (22). Dolman's inventive arrangement is shown in FIG. 3.
The second LO provides a first output signal to its controlling PLL (17) and a second output to TX offset PLL (19). Since the crystal oscillator is now not used for any purpose in transmit circuits (18,19) oscillator (21) is shown once more as a separate circuit (21), having two buffered outputs. The total number of radio frequency signal outputs has however increased, as has the number of separate integrated circuit chips. The radio frequency signals distributed on the printed circuit board are as follows:
1) 1st LO signal from frontend (12) to PLL (14);
2) 1st LO signal from frontend (12) to TX offset PLL (19);
3) Crystal reference frequency from oscillator (21) to PLL (14+17);
4) Crystal reference frequency from oscillator (21) to processing (20);
5) 2nd LO from IF chip (16) to controlling PLL (17); and
6) 2nd LO from IF chip (16) to TX offset PLL (19).
It is an object of the current invention to reduce the number of RF distribution tracks from the six enumerated above.
FIG. 4 shows one implementation of the invention. The two places to which the 1st LO signal is routed, that is the first LO PLL (14) and the TX offset PLL (19) are colocated with modulator circuit (18) in first (transmit) integrated circuit. Thus there is only a single 1st LO output connection from front end chip (12) to transmit chip (14,18,19).
However, when two synthesizer PLLs are colocated on the same chip, they should produce output pulses at different times. This is difficult or impossible to arrange if the two phase comparators have independent reference frequency sources, such as when the second LO is used for the TX offset PLL reference and the crystal is used as the first LO reference. Therefore, according to this invention the second LO is also used as the reference for the first LO synthesizer PLL. Moreover, as will be shown below, there are numerical advantages in using the second LO as the reference source for the first LO, particularly when it is desired to construct dual-band/dual-mode radios. Thus, a single reference input from 2nd LO part of IF chip (16) is provided for both PLLs (14) and (19).
Crystal oscillator circuit (21) is now combined with 2nd LO PLL and IF circuit (16) so that the reference signal from oscillator (21) to PLL (17) is an internal connection only. Likewise, the 2nd LO signal to its controlling PLL (17) is an internal signal only. The only remaining external signal is from reference oscillator (21) to digital processing (20).
The radio frequency oscillator signal distribution has now been reduced to the following signals:
1) 1st LO signal from frontend (12) to PLLs (14 & 19);
2) Crystal reference frequency from oscillator (21) to processing (20); and
3) 2nd LO from IF chip (16) to TX offset PLLs (14 & 19).
It would have been equally conceivable to place crystal oscillator (21) into digital processing chip (20), however, the oscillator (21) is more logically associated with other analog/RF circuits that use the same integrated circuit fabrication processes, and is therefore envisioned to be preferably integrated with IF chip (16,17,21). It is possible that in some applications a Very High Frequency (VHF) crystal, such as an overtone crystal, could be used directly to control the frequency of the second local oscillator without the use of a digital frequency synthesizer PLL circuit; however, VHF overtone crystals are more difficult to adjust to a desired oscillation frequency than fundamental-mode crystals, so a fundamental-mode crystal reference oscillator with a digital PLL is preferred.
FIG. 5 gives more details of the reference frequency distribution and frequency synthesis arrangements according to the inventive block diagram of FIG. 4.
The basic source of an accurate frequency reference in the apparatus is quartz crystal resonator (22) of FIGS. 1-4, connected to oscillator circuit (21). Even a quartz crystal cannot provide the necessary accuracy required for cellular phones operating in the 2 GHz region of the radio spectrum, and therefore means contained within digital processing (20) determine the receiver frequency error relative to signals received from land-based network stations or satellite relays, which error is ascribed to crystal (22), and an adjusting signal is sent to frequency adjustment components (such as a varactor diode for example) connected to crystal (22) so as to annul the error.
In FIG. 5, the oscillator circuit (21) is incorporated in IF chip (30) along with second local oscillator (33) and its control PLL comprising reference divider (35), first variable divider (32), phase comparator (31) and loop filter (34). The crystal oscillator signal is divided in frequency by counter/divider (35) which divides by a first integer M1 to produce a phase comparison frequency Fref/M1, where Fref is the crystal frequency. The second local oscillator signal is divided in frequency by an integer N1 in first variable divider (32), to produce a second phase comparison signal, which is compared with the phase comparison frequency signal from divide-by-M1 circuit (35) to produce a phase and frequency error signal from first phase comparator (31).. The phase error signal is filtered sand integrated using loop filter (34) to produce a frequency control signal to 2nd local oscillator (33) free of comparison frequency ripple. The higher the comparison frequency from divider (35), the easier it is for, loop filter (34) to eliminate this unwanted ripple while otherwise maintaining a fast speed of response to correct unwanted fluctuations of the 2nd LO's frequency due to noise or vibration, for example. Therefore an objective of the invention is to obtain a high comparison frequency, that is a low reference divide ratio M1. The second local oscillator frequency is thus accurately controlled to equal Fref.N1/M1.
According to Dolman's prior invention and the current invention, a buffered second local oscillator signal is output from 2nd LO (33) to be used as a reference for other frequency generation, in particular the transmit offset frequency (according to Dolman's prior application referenced above) and now also the first local oscillator frequency according to this invention. Since both the TX offset and 1st LO synthesizer PLL circuits are to be colocated in transmit signal generation chip (40) in order to reduce distribution of the second LO signal to a single cross-board connection, it is desirable that the phase comparators of the respective PLLs should pulse at different times separated as far as possible within some longest common period. This ensures that when one charge-pump phase comparator receives a current pulse from the supply, the other charge pump is in its tristate, that is, a high impedance state or open circuit output, with no current flowing to its respective loop filter. This minimizes the risk of interference from one charge pump to the other. The design and operation of charge pump phase detectors is described more fully in above-incorporated U.S. Pat. No. 5,095,288.
In order to provide the preferred out-of-phase relationship between the charge pumps (43,49), internal frequency plans are sought in which the phase comparison frequency for the TX offset loop is an integer multiple M3 of the phase comparison frequency for the first LO. Since the second LO frequency is also, according to Dolman's incorporated application, an integer M2 times the TX offset reference, the first LO comparison frequency must now be related to the second LO frequency divided by M2.M3.
Thus the second LO frequency signal is input from IF chip (30) to TX chip (40) and divided by integer M2 in a second reference divider (41) to obtain the phase reference to TX offset phase comparator (43) according to Dolman as
This frequency is then further divided in third reference divider (42) by integer M3 in order to obtain the phase comparison frequency for first LO phase comparator (49). Moreover, divider M3 and phase comparator (43) are arranged to respond to opposite edges of the output of divider M2, for example one responding to a rising edge (low voltage or '0' state transitioning to high voltage or '1' state) while the other responds to a falling edge (1 to 0 transition). This ensures that they respond one half cycle apart in time of their lowest common multiple frequency at the output of divider (41).
The phase comparison rate for charge pump phase comparator (49) is thus
This frequency is compared with the 1st LO frequency from first LO (51) divided down by a factor N3 in third variable divider (48), to produce a frequency and phase error signal from comparator (49) which is filtered in loop filter (52) to obtain a feedback control signal to control oscillator (51) to the desired 1st LO frequency
Preferably, N3 is not an integer factor but comprises a whole part and a fractional part, the components (48,49 and 52) of the first LO PLL forming a fractional-N synthesizer according to above-incorporated U.S. Pat. No. 5,180,993. Optionally, both M3 and N3 can be varied in a pattern generated by a fractional-(N,M) controller according to U.S. Pat. No. 6,236,275 entitled "Digital Frequency Synthesis by Sequential Fraction Approximations" by Dent, which is hereby incorporated by reference. Both the fractional-N and the fractional (N,M) techniques have the desirable effect of allowing the 1st LO phase comparison frequency to be higher than the desired tuning step size, thus making it easier for the loop filter (52) to filter out unwanted comparison frequency ripple while otherwise maintaining a fast control loop response to correct errors.
The transmitter frequency signal, when transmission is required, is generated by transmit frequency oscillator (45). A transmit frequency signal from oscillator (45) is mixed in TX mixer (46) with a first LO signal from first LO (51). The first LO signal preferably comes from receive chip (12) via a single cross board connection to minimize RF tracks. A single cross board connection for any of the interchip signals mentioned can nevertheless be a balanced connection comprising two tracks driven in antiphase, as balanced connections to and from RF chips at high frequencies reduce unwanted stray coupling and radiation effects.
TX mixer (46) mixes the transmit frequency and the 1st LO frequency to produce a difference frequency signal at the TX offset frequency Ftxoff. The difference frequency signal output from mixer (46) may be low-pass filtered to ensure the original, higher input frequencies are removed, and then drives second variable divider (47) which divides by a factor N2. The output signal at frequency Ftxoff/N2 is then compared in second phase comparator (43) with the phase reference from divider (41) to produce a frequency and phase error signal. The error signal from comparator (43) is filtered and integrated in loop filter (44) to produce a control signal which controls TX oscillator (45) until the desired TX-offset frequency is accurately achieved. Thus the TX offset frequency is given by
It is possible for the TX offset PLL comprising components (41,43,44,45,46,47) also to be a fractional-N synthesizer, however fractional-N synthesizers are more complex than integer synthesizers and thus it is desired to avoid having more than one in the apparatus. Thus, factor N2 is preferably an integer.
Obtaining the highest possible phase comparison frequencies for phase comparators (31,43,49) is rarely a problem in single-band radios having a single duplex spacing between transmit and receive frequency channels. It is more difficult first in the context of two-band radios that must operate with more than one duplex spacing. Therefore two-band radio designs according to the invention will now be described with the aid of FIGS. 6,7,8 and 9.
A two-band radio according to FIG. 6 would comprise a transmit frequency oscillator (45) for generating transmit frequencies in the lower of the two possible transmit frequency bands, A frequency doubler (45 a) is then used to double the frequency when operation in the higher of the two frequency bands is desired, the lower and higher bands being approximately one octave apart. An output from the oscillator (45) directly is used to drive the modulator when lower-band operation is required while an output from the doubler (45 a) is used when operation at the higher band is desired. However, as indicated in FIG. 6, the lower frequency direct from oscillator (45) enters TX mixer (46).
Likewise, first local oscillator (51) operates on a frequency adapted to the lower of the two-possible receive frequency bands to convert received signals to the desired first Intermediate Frequency; the signal from first LO (51) is doubled in frequency using doubler (51 a) when operation in the higher of the two receive frequency bands is desired, the LO frequency for the higher band being approximately an octave higher than that for the lower band. This approximation can be manipulated to be a close approximation by suitable choice of the first Intermediate frequency and by suitable choice of either high-side or low-side mixing in front-end chip (12).
For example, for low-band receive operation, we have
F101(lo) is the low-band first LO frequency,
Frx(lo) is the low-band receive channel frequency and
Fifl is the chosen first intermediate frequency.
F101(hi) is the high-band first LO frequency,
Frx(hi) is the high-band receive channel frequency and
Fifl is the same chosen first intermediate frequency as for low-band.
Thus, for F101(hi) to be twice F101(lo), we have
The latter two equations give negative results which is impossible. A possible alternative is to make the first Lo range at high band triple the first LO range at low band, giving
Examples of preferred internal frequency plans for a radio according to FIGS. 4 and 5, operating according to the IS54 "D-AMPS" single-band standard, will now be described. A search of frequency plans giving the highest possible phase comparison frequencies at phase comparators (31,43 and 49) yielded, among others, the results:
The above results provide phase comparison frequencies of
Fxtal/M1=19.44/9=2.16 MHz for 2nd LO phase comparator (31);
Flo2/M2=101.52/9=11.28 MHz for TX offset phase comparator (43), and
Flo2/(M2.M3)=11.28/47=240 KHz for 1st LO phase comparator (49). The first LO tuning steps are reduced from the above 240 KHz to 30 KHz by employing a fractional-N divider for N3 giving steps of ⅛th, i.e. the fractional-N modulus is 8.
The above solution provides a high TX offset phase comparison frequency of 11.28 MHz. Other criteria might be to obtain the highest 2nd LO phase comparison frequency. An alternative result for which the 2nd LO is just a harmonic of the crystal is for example:
The above values result in a 2nd LO phase comparison frequency of 19.44 MHz at phase comparator (33), and divider (35) is not necessary because M1=1. The transmit offset and first LO phase comparators (43,49) both operate at 480 KHz, and divider (42) may be omitted as M3=1. The first LO tuning steps are reduced from 480 KHz to 30 KHz by employing a modulus 16 fractional-N divider (48) allowing N3 to be varied in steps of {fraction (1/16)}th.
Attention is now turned to dual-band radios with internal frequency reference distribution according to FIG. 6. The above two exemplary solutions were illustrated because they are also compatible with a dual-band radio operating according to the dual-band D-AMPS standard IS136. Solutions for dual-band radios are given in the tables below for cases where the first local oscillator is on the high side for 800 MHz hand operation and on the low side for 1900 MHz band operation, and the second IF is fixed at 120 KHz.
Table 1 illustrates solutions in which the 2nd LO is a harmonic of the crystal, that is the second LO has the highest possible phase comparison frequency, M1 being equal to unity.
When using the figures in table 1 above to determine the phase detector comparison frequencies for TX offset phase comparator (43), it must be taken into account that the arrangement in FIG. 6 controls the TX oscillator (S1) frequency BEFORE doubling to 1900 MHz in doubler (51 a).
Therefore, phase comparator (43) must operate at half the frequency provided by divider (41) and the indicated value of M2.
Thus, either phase comparator (43) must contain a further divide by two circuit to halve the frequency from divider (41) when operating in the 1900 MHz band, or else the value of M2 for 1900 MHz must be doubled.
In the latter case, the value of M3 for 1900 MHz operation must be halved (which is impossible as M3 is always odd at 1900 MHz), or else the fractional modulus for 1900 MHz operation must be halved. The latter is preferable and so the preferred fractional-modulus at 1900 MHZ is 4 or 12 combined with an M2 value double that shown in table 1 for 1900 MHz operation. The TX offset phase comparison frequency at phase comparator (43) is thus 1080 KHz for 1900 MHz operation and not 2160 KHz as would be obtained by dividing the second LO frequency in table 1 by the indicated values of M2.
Furthermore, note that in FIG. 6 it is always the doubled frequency from doubler (S1 a) that is fed to fractional-N first-LO synthesizer loop beginning with variable divider (48) for N3. Since the frequency used for the receiver mixer during 800 MHz band operation is half of the synthesized frequency, the synthesizer need only provide 60 KHz steps in order to tune the receiver in 30 KHz steps. Thus the fractional-N modulus shown in table 1 for 800 MHz operation may be halved.
It may be desirable to operate in both bands using the same fractional-N modulus, and this may always be accomplished by use of a modulus which is the lowest common multiple of the 800 MHz and 1900 MHz moduli, accepting that the frequency steps in one or both bands may then be finer than needed, it being acceptable to exceed the required frequency resolution.
The above issues are one motivation for considering frequency halving circuits (45 b,S1 b) in FIG. 7 as opposed to frequency doubling circuits (45 a, 51 a) of FIG. 6. Another motivation is that phase noise is doubled by frequency doubling circuits but halved by frequency halving circuits. Thus there is an expectation of lower unwanted phase noise and ripple when using frequency halving circuits. Yet another motivation is that a frequency doubling circuit requires a filter to remove unwanted leakage of the fundamental, as well as other unwanted higher harmonics; the output of a frequency divide-by-2 circuit is however relatively free of other unwanted spectral components.
Referring now to FIG. 7, it is seen that it is the non-divided output of oscillator (S1) which is fed to the TX offset synthesizer loop beginning with mixer (46). Therefore the phase comparator for 800 MHz band operation must operate at double the frequency implied by table 1, i.e. the value of M2 must be halved, alternatively the value of N2 must be double that shown in table 1 for 800 MHz operation. The former is not possible for cases where M2 is odd, but is possible when the first IF is 155.64 MHz and M2=162. Thus, when table 1 is applied to FIG. 7, the values of N2 for 800 MHz should be doubled except in the case of first IF=155.64 MHz, in which case a better option is to halve M2 to 81; then it is necessary to double the value of M3 (to two) for 800 MHz operation in order to maintain the same 1st LO phase comparison frequency at phase comparator (49), alternatively to increase the fractional-N modulus from 32 to 64. On the other hand, since the frequency of oscillator (51) is halved before use in the receiver for 800 MHz operation, it is sufficient that oscillator (51) be tuned in 60 KHz steps, allowing the fractional-N modulus to be halved again back to 32.
The above considerations with respect to FIG. 7 also apply to the arrangement of FIGS. 8 and 9, in which the TX frequency signal and the first LO are always controlled at the higher frequency and halved for 800 MHz use.
In choosing between the implementations of FIGS. 6,7,8 and 9, another motivation is power consumption. In FIG. 6, the �2 circuit 51 a must be powered up during 800 MHz band receive operation, which has most impact on standby time before the battery must be replenished. Still, the �2 circuit 45 a need only be powered up for 1900 MHz transmission, thus saving power in 800 MHz band transmission. In FIG. 7, divider 51 b need only be powered up for 800 MHz reception, and may be powered down for 1900 MHz reception. Divide by 2 circuit 45 b likewise need only be powered up for 800 MHz transmission and is not needed for 1900 MHz transmission.
In FIG. 8, �2 circuit 45 a must be powered up for transmission in either frequency band, but this is of little consequence as the power amplifier (13) dominates transmit power consumption. Divide by 2 circuit 51 b may be powered down during 1900 MHz receive. In FIGS. 6 and 9, �2 circuit 51 a must always be powered up for reception in either frequency band. Therefore this is not as desirable for standby battery life at 1900 MHz as FIGS. 7 or 8.
1900 MHz D-AMPS operation uses TDMA, which affords much longer standby times due to a low receive duty factor. 800 MHz operation however includes the analog FM AMPS mode, in which receive standby duty factor is longer. The 800 MHz AMPS operation is therefore the limiting factor for battery life and we are therefore led to consider FIG. 10, in which the first LO is always controlled at the lower frequency, allowing doubler 51A to be powered down during 800 MHz reception.
Referring to FIG. 10, the first LO is always controlled at the lower frequency, i.e. before doubling. This allows doubler 51 a to be powered down in 800 MHz operation. A disadvantage, however, is that the oscillator 51 must be tuned in 15 KHz steps in order to provide 30 KHz steps at 1900 MHz, requiring the fractional-N modulus to be doubled, which is undesirable. Taking into account that a divide-by-2 circuit in present semiconductor technology consumes very little power, and likely less than a frequency doubler circuit, together with the other advantages outlined above for frequency halving rather than frequency doubling, FIG. 7 is likely to be the best practical implementation.
The above Table 1 listed solutions for which the second LO was a harmonic of the crystal, giving the lowest value of unity for divider (35). Table 2 lists solutions in which 2nd LO phase comparator (31) operates at 6.48 MHz, which is the crystal frequency divided by 3 (M1=3).
There are also many solutions with the 2nd local oscillator a multiple of 2.16 MHz (crystal/9, i.e. M1=9), or 720 KHz (crystal/27 or M1=27), and at least one solution with M1=6. Table 3 below only lists other solutions that have particularly interesting characteristics such as high comparison frequencies for TX offset comparator (43) in either 800 MHz or 1900 MHz operation.
The above solutions are remarkable for their relatively low values of (M2,N2) in one or other frequency band, giving very high TX offset phase comparison frequencies in those cases.
As described above, it is of interest that the first local oscillator in the higher frequency range should tune over a range approximately equal to twice the range of frequencies needed for operation in the lower band.
The receive frequency range for the 800 MHz cellular band is 869.04 to 893.97 MHz, while the receive frequency range of the 1900 MHz PCS bands is 1930.08 to 1990.08 MHz. Substituting into equations (1),(2),(3) and (4) above gives desirable first intermediate frequencies of 192 MHz, 64 MHz, 338.52 MHz and 169.26 MHz respectively. The 64 MHZ IF is too low to provide sufficient image rejection when operating over the 60 MHz wide 1900 MHz receive band. The 338.52 MHz IF is difficult to choose because of the unavailability of SAW or crystal filters with a 30 KHz bandwidth at that frequency. The solution of equation (1) or equation (4) is therefore preferred.
All the above solutions in tables 1-3 were for 1st LO high at 800 MHz and low at 1900 MHz, i.e. for the solution of equation (2). These can be employed providing the range of the local oscillator (51) is band-switched between 800 MHz and 1900 MHz operation. It can be undesirable to attempt to cover in one band the entire tuning range that would be required for operation at both 800 and 1900 MHz.
A search for solutions for the case of equation (1) yielded the following result with the closest first IF to 192 MHz:
TX offset comparison frequency AT 800 MHz=2640 KHz
(actually 5280 KHz for the arrangement of FIG. 7, with N2=89,M2=36)
TX offset comparison frequency at 1900 MHz=2160 KHz
2nd LO comparison frequency=2160 KHz
The first LO phase comparison frequency is for example 240 KHz if a fractional-N modulus of 8 is selected for both bands.
Alternatively, a fractional-N modulus of 24 may be selected to give 720 KHz phase comparison frequency at 1900 MHz, but the phase comparison frequency will still be 240 KHz at 800 MHz. The tuning step size at 800 MHz will be 10 KHz with the same modulus of 24, or even 5 KHz for the arrangement of FIG. 7. This is finer than the 30 KHz needed, but is acceptable. 240 KHz is an adequate comparison frequency for 800 MHz operation, and the higher comparison rate of 720 KHz is desirable for 1900 MHz operation where the oscillator phase noise is double that at 800 MHz.
A solution in accordance with equation (4) assumes division of the high-band first local oscillator frequency by 3 for 800 MHz operation. In other words, the divider (51 b) of FIG. 7 must be changed from a divide-by-2 to a divide-by-3 circuit. It is also necessary then to change divider 45 b to a divide by 3 circuit, in order for the transmit frequency steps at 800 MHz to be correct. This solution is not investigated further here, as it is not preferred for a dual-band IS136 cellular phone, and in any case is an obvious extension of the disclosed methods.
The invention can be used for dual-band/dual-mode radio telephones in which compatibility with AMPS and IS54 (DAMPS) is desired in the 800 MHz band together with compatibility with the PCS1900 (GSM-based) standard.
The problem to be solved is that a radio is normally designed for D-AMPS operation based upon the use of a 19.44 MHz crystal as the most convenient common multiple of the 24.3 KS/S transmission symbol rate, the 30 KHz channel spacing and the 8 KS/S voice digitization. On the other hand, a radio is normally designed for GSM, DCS1800 or PCS1900 operation based upon a 13 MHz crystal, which is the lowest common multiple of the transmission bitrate of 270.833 KB/S (13 MHz/48), the channel spacing of 200 KHz (13 MHz/65) and the 8 KS/S voice digitization rate. This makes it difficult to merely integrate a radio of one design with a radio of the other design, due to the increase in parts count. Therefore it is desired to find internal frequency plans that allow components to be designed that can operate from either crystal frequency, and as another objective it is desired to find reference frequency distribution schemes that will allow operation with the same crystal reference frequency in any of an AMPS mode at 800 MHz, a D-AMPS mode at 800 or 1900 MHz or a PCS1900 mode at 1900 MHz.
FIG. 12 illustrates a solution using both a 13 MHz and a 19.44 MHz crystal connected to reference oscillator (21), only one of which however is activated at a time via a "select crystal" control signal from digital logic (20).
A single intermediate frequency amplifier chip comprise a dual-crystal reference oscillator (21), second LO and its control PLL (17) and a dual-bandwidth second IF amplifier and second mixer (16). The reference oscillator operates in one mode at 13 MHz and the second LO is then controlled to 12�13 MHz. Alternatively, in a second mode, the reference oscillator operates at 19.44 MHz and the second LO is controlled to, for example, 155.52 MHz, which is sufficiently close to 156 MHz that the same oscillator can be used, while also being a multiple of 19.44 MHz ( eight times 19.44 MHz).
The IF amplifier chip receives a downconverted signal from front end chip (12) which is filtered either using Wideband IF filter (15 WB) or Narrowband IF filter (15 NB). The filter center frequency in the wideband mode is 150 MHz, which mixes with the 2nd LO of 156 MHz in that mode to produce a second IF of 6 MHz, which is fed to digital signal processing (20) along with the RSSI signal. The narrowband first IF filter operates at a center frequency of 120 KHz higher than the second LO of 155.52, that is at 155.64 MHz, giving a second IF in the narrowband mode of 120 KHz, which is then fed to the signal processing chip (20). The second IF signal at either 120 KHz in the narrowband mode or 6 MHz in the wideband mode is preferably further filtered in IF amplifier (16) using second IF filters (not shown). In one implementation, the 120 KHz second IF filters are integrated active bandpass filters having a passbandwidth of approximately 30 KHz, and are fabricated as part of IF amplifier chip (16,17,21). The 6 MHz 2nd IF filtering is performed by external ceramic filters (not shown) of approximately 170 KHz bandwidth, as used for TV sound IF stages.
When operating in the narrowband AMPS mode at 800 MHz, the duplex spacing is 45 MHz and so the transmitter frequency is 45+155.64 MHz below the first LO. The TX offset would therefore be 200.64 MHz. However, as shown in FIG. 13, the TX mixer (46) mixes transmit and receiver oscillators (45,51) at double the 800 MHz frequency, and so produces an offset of 401.28 MHz. This has a highest common factor of 1920 KHz with the second LO of 155.52 MHz, so divider (47) divides the TX offset from TX mixer (46) by a first integer N2 to obtain a first 1920 KHz signal, and divider (41) divides the second LO from IF chip (30) by and integer M2=81 to produce a second 1920 KHz signal. The two 1920 KHz signals are compared in transmit phase comparator (43) to produce an error signal. The error signal is filtered and integrated in loop filter (44) to produce a control signal for TX oscillator (45) to maintain it at the desired frequency, which, when halved in divider (45 b), is the desired 800 MHz transmit frequency.
This frequency plan at 800 MHz may also be used for the D-AMPS mode in the 800 MHz band. For operating in the D-AMPS mode at 1900 MHz, the duplex spacing is 80.04 MHz, so that the transmit offset is 80.04+155.64 MHz=235.68 MHz. This is not simply related to second LO frequency of 155.52 MHz; however, since only time-duplex modes are used at 1900 MHz such that transmission and reception occur in different timeslots and not simultaneously, the first local oscillator may be sidestepped by the relatively small amount of 240 KHz between transmit and receive so that a TX offset of 235.44 MHz may be used instead of 235.68 MHz.
The slightly modified TX offset of 235.44 MHz shares a common factor of 2160 KHz with the second LO of 155.52 MHZ. Thus in the 1900 MHz D-AMPS mode, divider (47) divides by an integer N2 reprogrammed to divide 235.44 MHz to 2160 KHz, while divider (41) is reprogrammed to divide by an M2 of 72 to obtain 2160 KHz, the phase comparator (43) now comparing signals at 2160 KHz instead of 1920 KHz.
Finally, to obtain the PCS1900 mode, where the duplex offset is 80 MHz, the transmit offset is 80+150 MHz, as the first IF is 150 MHz in that mode. The 230 MHz TX offset shares a common divisor of 2 MHz with the second LO that is now 156 MHz. This mode is also time duplex, and the first LO could be sidestepped to modify the TX offset from 230 MHz to for example 234 MHz, which has the much larger common factor of 78 MHz with the 2nd LO of 156 MHz. Nevertheless, it may be advantageous to keep a phase comparison frequency of 2 MHz, which makes all phase comparison frequencies (1920,2160 and 2000 KHz) sufficiently close to facilitate the use of a common loop filter (44) and phase comparator (43). Otherwise, if desired to take advantage of a larger common factor such as 78 MHz, a different loop filter and even phase comparator may become necessary to provide the desired closed loop characteristics of stability and lock-in time. Thus the arrangement of FIG. 13 has deliberately aimed to maintain roughly the same TX offset loop operational characteristics of loop-bandwidth and lock-in time in all bands and modes.
The dual-mode, dual-band transmitter-receiver of FIGS. 12 and 13 assumes a double superheterodyne receiver is used in all modes. In the narrowband AMPS and D-AMPS modes, the second intermediate frequency is 120 KHz and the second IF filters are integrated, on-chip, active filters; in the wideband PCS1900 modes, which can include all GSM voice and data modes, satellite communication modes and GPRS packet data modes, the second IF is 6 MHz, and the second IF filters are more difficult to integrate at that frequency. An alternative receiver architecture for the wideband mode is shown in FIG. 14, in which the second IF in the wideband mode is zero frequency, otherwise known as an "IF Homodyne", as opposed to an RF Homodyne, which converts directly from the frequency received at the antenna to zero frequency in one conversion step. The receiver of FIG. 14 converts from the frequency received at the antenna to zero frequency in two steps, the first step converting to a first Intermediate Frequency of 156 MHz and the second step converting from 156 MHz to zero frequency by mixing with the 156 MHz local oscillator. Since the first IF in FIG. 14 is now 156 MHz as opposed to the 150 MHz of FIGS. 12 and 13, the TX offset for 1900 MHz is now 156+80=236 MHz, which still shares a common factor of 2 MHz with the 156 MHz local oscillator. Thus the only change to FIG. 13 is that the value of N2 for PCS1900 operation would change from 230/2=115 to 236/2=118. If desired, a higher common factor of 4 MHz could be used by changing N2 to 236/4=59 and M2 from 78 to 39, and M3.from 2 to 4 (or alternatively changing the fractional-N modulus of N3 to accept a higher reference frequency for third phase comparator (49).
The implementations of FIGS. 12, 13 and 14 use two different reference crystals, although only one is active at any time. Nevertheless this adds the complication that both crystals must be independently temperature compensated, as every crystal has different individual temperature compensation needs. Temperature compensation is carried out by a "self-learning" technique, whereby the receiver locks to a base station signal and then uses the base station signal frequency as a basis for correcting crystal error. The prevailing temperature is measured using a thermistor, and the correction applied to the crystal is stored in a table against the prevailing temperature in microprocessor memory in digital signal processor (20).
To simplify the temperature compensation as well as eliminating the cost and board area associated with a second crystal, it is therefore of interest to consider the solutions of FIGS. 15 and 16 using a single crystal. The solution of FIG. 15 is to choose a compromise crystal frequency of 19.5 MHz. This is 1.5 times the 13 MHz from which PCS1900 bitrates are derived, and the bitrate is still derivable as 19.5 MHz/72 as opposed to 13 MHz/48. 19.5 MHz is also close to the 19.44 MHz needed for D-AMPS modes, from which the symbol rate of 24.3 KS/S is derived by dividing by 800. When 19.5 MHz is used, the error is 0.3%, which would cause a timing drift in the transmitted symbol stream of exactly half a symbol period during transmission of a TDMA burst of 6.667 mS or 162 symbols duration. In principle, such an error is no greater than must in any case be anticipated by the receiver due to multipath propagation causing transmission path delay variations of up to one symbol. Nevertheless it is desirable to correct the transmitted signal so that its errors do not compound the imperfections introduced by the propagation path. To a first approximation, the symbol rate error can be reduced by dividing the crystal frequency by 802 to obtain the symbol rate with a residual error of 0.0585%, giving a timing drift of less than one tenth of a symbol over a 162-symbol burst duration. A further refinement can be made by means of a skip-counter, which divides sometimes by 802 and sometimes by 803 in order to create a more accurate approximation to the 24.3 KS/S symbol rate. However, in one implementation, the 24.3 KS/S modulation is generated digitally at the rate of 8 samples per bit. Several samples per bit are used to represent the curved waveform of a symbol stream that has been filtered using a root-raised-cosine filter frequency response. Thus it is really desired to create an accurate approximation to 8 times the symbol rate, or 194.4 kilosamples per second, by dividing the crystal frequency sometimes by 100 and sometimes by 101. The number of times N1 division by 100 occurs and the number of times N2 that division by 101 occurs will now be derived.
The D-AMPS frame repetition period of 20 mS represents 390,000 cycles of a 19.5 MHz clock as opposed to 388,800 cycles of a 19.44 MHz clock.
A timing generator is thus programmed to divide by 390,000 when a 19.5 MHz clock is used as opposed to 388,800 when a 19.44 MHz clock is used, in order to create the 20 mS repetition period. The D-AMPS TDMA frame is divided into 3 timeslots, and one timeslot is thus 130,000 cycles of a 19.5 MHz clock in duration as opposed to 129,600 cycles of a 19.44 MHz clock. The first equation for N1 and N2 is therefore that
In addition, the total number of ⅛th symbol sample periods to be created is 8�162=1296 as before, so the second equation for N1 and N2 is
Solving these equations gives N2=400, N1=896.
Thus a skip counter is programmed to divide by 100 a total of 896 times interspersed with dividing by 101 a total of 400 times, creating a total of 1296 ⅛th symbol periods with no timing error greater than of the order of half a clock period of the 19.5 MHz clock, or 25 nanoseconds. FIG. 20 shows a skip counter design accomplishing the above. A divider (100) is configured to divide either by 100 or by 101 according to a control input from accumulator (101), so that successive output pulses from the divider (100) will be spaced by either 100 cycles of the 19.5 MHz clock or by 101 cycles. The accumulator (101) is configured as a modulo-81 accumulator, which means that if after adding an increment, the value in the accumulator is equal to or greater than 81, then 81 is subtracted from the accumulator value and an overflow or carry pulse is generated. The carry pulse output from accumulator (101) is used to cause divider (100) to divide by 101.
If no carry is generated by accumulator (101) upon being caused to increment by the last divider (100) output pulse, then divider (100) counts 100 cycles of the 19.5 MHZ clock input to produce the next output sample rate pulse. Else, if upon the last divider output pulse causing the accumulator to increment and overflow, then the accumulator carry output fed back to divider (100) causes the divider to count 101 cycles of the 19.5 MHz clock input before producing the next divider output sample rate pulse.
By setting the accumulator increment equal to 25, the accumulator produces a carry pulse {fraction (25/81)}ths of the time, which is equal to {fraction (400/1296)}ths of the time, this being the proportion of divide-by-101's calculated above needed to produce the exact number 1296 of 8�symbol rate pulses in a D-AMPS timeslot.
FIG. 16 shows the internal frequency plan using a 19.5 MHz crystal. The first IF in D-AMPS mode is changed to 154.32 MHz to give high phase comparator frequencies of 1320 KHz and 1080 KHz for transmit phase comparator (43) in 800 and 1900 MHz operation respectively, while also giving a high comparison frequency of 780 KHz at 2nd LO phase comparator (31).
Yet another implementation of the invention is shown in FIG. 17, this time using a 13 MHz crystal to derive all radio oscillator frequencies, and a 19.44 MHz crystal connected to digital chip (20) only to derive bit and digital sampling rates for the AMPS and D-AMPS modes. The frequency plan for this case is shown in FIG. 18, in which substantially the only difference from FIG. 16 is that the second LO phase comparator now operates at 520 KHz.
In both FIG. 16 and FIG. 18, the main receiver synthesizer (the first LO) operates as a fractional-N synthesizer with a modulus of 5 (optionally 10 or 20) in PCS1900 mode, and 12 in AMPS and D-AMPS modes.
It is possible to eliminate the 19.44 MHz crystal used only for generating digital clocks by using the arrangement of FIG. 19, in which the digital chip (20) creates its own 19.44 MHz clock by means of an internal PLL when needed. To facilitate this, dividers (41) and (42) are split into two dividers (41 a, 41 b) and (42 a, 42 b). Divider 41 a divides the 2ND LO frequency of 154.44 MHz by 117 in D-AMPS mode at 800 MHz to obtain 1320 KHz at which the transmit phase comparator (43) operates. Selector switch 41 c is operated to select the output of divider 41 a in this mode. During-this mode, divider 41 b operates simultaneously, dividing by 11 to provide a 14.040 MHZ output to the digital chip (20). This frequency shares a common factor of 1080 KHz with the 19.44 MHz generated on the digital chip (20) by means of a local PLL when needed. Divider (42 a) operates at this time to divide the operation frequency of phase comparator (43) by a further factor of 2 to obtain 660 KHz, which is used along with a modulus-11 fractional-N divider (43) to provide 60 KHz steps of oscillator (51), which provides 30 KHz steps after division by 2 for 800 MHz AMPS or D-AMPS operation. For 1900 MHz D-AMPS operation switch 41 c selects instead the output of divider (42 b) which is 14.04 MHz divided by 13, that is 1080 KHz. This is the desired frequency to provide the duplex offset of 80.04 MHz at 1900 MHz compared to 45 MHz at 800 MHz. The selected 1080 KHz for phase comparator (43) is then further divided by 3 in divider (42 a) reprogrammed to divide by 3, giving a 360 KHz phase comparison frequency for phase comparator (49), which, together with the use now of a fractional-N modulus of 12 for divider (48), gives 30 KHz tuning steps for oscillator (51) during D-AMPS operation at 1900 MHz. For PCS1900 operation, divider (41 b) is programmed to divide by 12, dividing the second LO of now 156 MHz to provide a 13 MHz clock output to the digital chip (20). The 13 MHz is divided by 13 in divider (42 b) to 1 MHz, which is the operating frequency of phase detectors (43,48). Divider (42 a) is thus programmed to make M3=1. Using a fractional-N modulus of 5 for divider (48) provides the desired 200 KHz steps in this mode.
It has thus been shown above that the invention permits the construction of dual-band, dual-mode transceivers using either a single crystal reference or two crystals in a variety of ways to derive alternative symbol rates of 270.833 KB/S and 24.3 KS/S, alternative channel spacings of 30 KHz or 200 KHz, and transmit-receive duplex spacings of 45 MHz, 80.04 MHz or 80.00 MHz.
Moreover, this flexibility is achieved with an improved architecture as compared with the prior art, allowing the radio hardware to be reduced to essentially three integrated circuit chips having a reduced number of RF interconnections there between, thus minimizing risks of internal interference and reducing power consumption.
The invention may be useful in contexts other than cellular radio, whenever it is inconvenient to derive all clock frequencies and radio frequencies that have to be synthesized directly from a common reference crystal oscillator. The invention may be adapted by a person skilled in the art using the above teachings while remaining within the spirit and scope of the invention as described by the following claims.
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Aided by online resources, concerns about tuition and a broader national trend, Jewish home schooling is increasing in popularity.
Squeezed into a child's seat, Nora Lavender, mother of two young sons, sits across from her oldest, 7-year-old Shalom Yehuda, at a small wooden table in a corner of an office in Bay Ridge, Brooklyn, leading him through spelling lessons. In the background are the sounds of children at play, which mother and son ignore.
Shalom Yehuda, his kipa featuring a soccer ball design, aces this afternoon's spelling lesson. His reward is a Harry Potter sticker, which he adds to the growing collection in his workbook.
The two are in the waiting room at Sensory Freeway Therapy Services, a therapy center that treats children with a wide variety of special needs and where Lavender's youngest son, 3-year-old Dovid Efraim, comes several days a week for treatment for mild cerebral palsy.
The setting is also a school for Shalom Yehuda.
The Lavenders are home schoolers, which for them means education that takes place at home, in therapy centers, museums, parks or other places where they can carry some books. While Dovid Efraim gets therapy, Shalom Yehuda learns; Sensory Freeway has become as familiar to the family as their living room.
Paralleling a steady rise in the number of American families who have decided in recent decades to take their children's education into their own hands — a movement especially in vogue among Evangelical Christians who distrust the secular bent of public education — home schooling has also become an option for a growing number of Jewish families. While Jews of various denominational backgrounds have started to home school, Orthodox Jews are thought to represent a majority of Jewish home schoolers. The evidence is heavily anecdotal however; while there are about 1.5 million home schoolers in the United States, according to the most recent statistics compiled by the U.S. Department of Education, no one has figures breaking out the Jewish numbers.
Home schoolers themselves, along with outside educational experts, speculate that at least several hundred American Jewish families, probably as many as several thousand, will be home schooling their children when the school year begins in September. That's a steep increase from the estimated few hundred a decade ago.
Home schooling "is growing" in Jewish circles, says Jonathan Woocher, chief ideas officer of the Jewish Educational Service of North America. "People are looking for more customized ways of dealing with their Jewishness."
Ilana Masri, a home schooling mother in Brooklyn says she loves "the freedom that home schooling provides us. Our children are not bound by the confines of traditional school in terms of what they learn and how fast they have to learn it."
Susan Lapin, a longtime home-schooling parent who now lives in Baltimore, says, "It is only in recent years that the number of Jewish home schooling families has grown significantly."
"There are reasons why Jews lagged behind in this American trend, including loyalty and well-deserved respect and affection for the system of religious private schools which has burgeoned in America since the end of World War II," she writes on her blog, susanlapin.typepad.com.
Yehudis Eagle, conference coordinator of the Torah Home Education Group, which sponsors an annual Jewish home schooling conference, says she has witnessed an increase in the last few years of Jewish home schoolers. With approximately 200 participants, this year's conference in Baltimore was more than double the size of last year's gathering, Eagle says.
"We are not alone anymore," she says. Home schooling, while still practiced by a distinct minority of Jewish families, is achieving a critical mass and a recognition as a valid, no longer fringe, educational alternative.
Why do Jewish parents decide to do home schooling?
It's not just for financial reasons, like out-of-reach day school tuitions, Eagle says. She estimates that a third of Jewish home schooling is done for "grass-roots, philosophical reasons," a third for economic reasons and the other third for a variety of reasons.
Primarily, she says, "many of us don't want to delegate [responsibility for children's education] to a teacher."
"Sometimes," says author and home-schooling mother Evelyn Krieger, who lives in Sharon, Mass., "when you are in the system, such as working in a school, you see practices, administration, curriculum and social issues that bring you dismay. Much time is wasted in the typical school. Often, I would say to myself, 'I could do better for my own kids.'"
A decade-old study of home schooling in the general community conducted by the National Center for Education Statistics found that the desire for a "better education," "religious" motivation and a "poor learning environment" at established schools were the reasons most often given for home schooling.
"I wish there were fewer kids doing home schooling," for the sake both of the children and of established Jewish schools, but sometimes it's "in the child's best interests," says Rabbi Shmuel Klein, a spokesman for Torah Umesorah, the national association of mostly Orthodox Jewish day schools. "It's a good thing that [a choice] exists. Bless the good Lord that it exists as an opportunity."
The growth of the Jewish home schooling movement poses no threat to day schools, most of which are flourishing as the Orthodox population has grown in the last few decades, Rabbi Klein says. "The system's thriving. Our schools are growing."
For Jewish home schoolers, the setup varies. Some parents arrange joint lessons for several children in Hebrew or other subjects; others follow a traditional pattern of classes, homework and grades. Lessons can be parent-led or can involve children developing their own, independent learning skills; sometimes families participate in home school-geared programs offered by institutions like museums and galleries and the Hudson Valley's Eden Village Camp, which last year established a Farm & Forest Homeschool Program whose holidays-oriented fall session starts Sept. 6.
Some children enter home schooling at nursery school age and stay there through high school; others start out at home, then transfer to an established school. Sometimes it's vice versa. Some participate in what is called "unschooling," which is more free form, following a student's interests or abilities.
Jewish home schoolers are not a monolithic group, says Rebecca Masinter, Baltimore mother of five home-schooled children.
"Being a home school parent," she says, "is only being a parent." She grew up in a home-schooled family, and decided, with her husband, to carry on the tradition.
At the recent Jewish home schooling conference in Baltimore, a vendors' exhibit featured books, curriculum guides and such high-tech educational aids as the KeyboardTownPals.com "instruction coloring storybook" and the LivingLessons.com program for teaching "Torah subjects." Sessions included such topics as money management, home schooling the bar/bat mitzvah-age student, "Advancing the Relationship Between Homeschoolers and the Community" and "Bringing Technology Into Our Homes."
The latter topic is a nod to online classes and other tech tools that have made the growth of Jewish home schooling possible. Home schooling parents have created a virtual community of like-minded parents.
"The Internet has changed everything," Evelyn Krieger says. "The [home schooling] pioneers in the 1960s and '70s simply did not have the resources and connections we have now."
"Social media has made it much easier to connect and exchange ideas with Jewish home school families all over the world," Michelle Kleinman, a home schooling mother in Brooklyn, tells The Jewish Week in an e-mail interview.
Do home schooling families "burn out" from the 24/7 parent-child/teacher-student time together? No, they report. The home schooling serves as a bond, they say. But, Eagle says, "everyone needs a break and some personal space sometimes — home schooling moms too."
Since the emergence of the home schooling phenomenon, home schoolers have faced the criticism that their stay-at-home children don't get the same quality of education as that provided in a standard classroom, and that, by spending much of their time among their parents and siblings, they don't develop the socialization skills necessary for dealing with other people in other venues.
The statistics of home schooled students being accepted into universities and flourishing there academically refutes the first charge, home school parents say.
As for social skills — "This is the most frequently asked question of home schoolers and the most dreaded," Evelyn Krieger says. "We call it the 'S word.'"
Her children, like most home-schooled kids, get out of the house on a frequent basis, she says. "Most cities and towns now have home school social groups and co-ops that meet for field trips, park days, open gym, as well as academic classes. Museums, YMCAs, libraries are now opening their doors to home school programs during the school day."
For Shalom Yehuda Lavender, New York City is his classroom, and his place to meet other kids. There's a joint Hebrew class with other home-schooled children, a home-school soccer league set up by parents, and clubs geared to students' other academic interests.
He says he enjoys being a home-schooled student. "You get to do cool stuff." Like the Bronx Zoo, Museum of Natural History and the Liberty Science Center (part of his individualized science curriculum). His favorite part is the integrated lessons that his mother offers, teaching, for instance, history through literature; he's an avid reader, often of books above his grade level.
Today, he turns from spelling to penmanship. At his mother's instruction, he traces letter after letter in English in a workbook.
His mother — his teacher — is satisfied with his writing.
He gets another sticker.
Home Schooling Resources Online
Among the home schooling resources available on the Internet:
♦ Room613.net — classes, curriculum resources and networking opportunities.
♦ Homeschooling.gomilpitas.com/religion/jewish.htm — clearinghouse for educational resources.
♦ ChinuchAtHome.info — curriculum guides and essays from an Orthodox perspective.
♦ Chinuch.org — printable worksheets, project ideas and lesson plans.
♦ Chesska.com/homeschool/chevra_faq.html — answers to "frequently asked questions" about Jewish home schooling.
♦ Home-school.LoveToKnow.com/Jewish_Homeschooling — a hands-on guide to specific concerns of Jewish home schooling parents.
♦ Successful-Homeschooling.com/jewish-homeschooling.html — the experience of one home schooling family.
♦ JewishHomeSchool.blogspot.com — the musings of "a Busy Jewish mom married to a Fabulous Jewish dad raising 5 amazing blessings."
♦ JewishHomeSchooling.blogspot.com — a similar personal collection of insights and resources.
♦ Chayas.com/homeschoolindex.htm — a guide to home schooling by the now-defunct League of Observant Jewish Homeschoolers. Though LOJH no longer operates, its duties have been taken over by other organizations that serve the home schooling community, and its guide still appears on the chayas.com website. "The number of requests we receive about this topic is overwhelming," the website states.
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We are new to homeschooling - my son just started homeschooling and he is high school
I would like to know if there are any Jewish high school homeschoolimg groups anywhere in south Florida .
Thank you for this terrific and balanced article on Jewish homeschooling. As one of the parents you quoted, I'd like to add that there are wonderful things happening out there in Jewish day schools, as well as not so wonderful practices. I have one child who homeschooled all the way and one who graduated from day school and another who homeschooled for a few years. Jewish parents should know that homeschooling can be a viable option either short or long term.
A group of Orthodox Jewish parents in South Florida have been successful at using a hybrid, low cost administrative model called 'cooperative schooling'. The results have been stellar, small class-size education with a dual curriculum. jewishcooperativeschool.org
Hi Everyone! Jewish Farm School has established the Farm and Forest Homeschool Program on the Farm at Eden Village. Our groundbreaking homeschool program starts Thursday, September 6th- Join us! http://www.jewishfarmschool.org/homeschool
I think this is cool. I hope the children do get some socialization activities, like, hmmm, basketball or something where they need to interact with others. Also, there have been studies that show children not put in daycare have worse allergies. This last bit may be off topic, but it might not be. Some kind of regular group interaction, like basketball, would be nice.
And I don't even play basketball. Well, I did once, several months ago. Nothing compares as an exercise in group dynamics/cooperation/interaction/working together etc, and etc. And it's fun and exciting! And good exercise!
You're right, but I can assure you that most homeschoolers get a lot more exercise than schooled kids who spend a lot of time at desks.
Another resource that is secular and open to any and all who feel they belong there: Gifted Homeschoolers Forum http://giftedhomeschoolers.org
GHF works to support, educate & advocate for families regarding their choice of educational path for their gifted and twice-exceptional (2e) children around the world. We're not specifically for Jewish homeschooling, but we have members from a multitude of countries and backgrounds.
In my personal situation, my (Jewish) family was not allowed to join what was, at the time, the only local (rural) homeschool group because I could not sign their Christian statement of faith. As Executive Director of GHF, I believe it's very important to cross those kinds of lines. Everyone is welcome.


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Somehow I imagined an inaugural ball would be … less gross.
Presidents are supposed to rise above that for inaugural addresses.
This was the president, after all, who memorably called in his second inaugural for " ending tyranny in our world."
1680s, from French inaugural (17c.), from inaugurer "to inaugurate" (see inauguration). The noun meaning "an inaugural address" is recorded from 1832, American English.


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Music, entertainment, current events, literature, cinema and opera will all be integral parts of a new summer adult lifelong learning program at KleinLife: Northeat Philadelphia, 10100 Jamison Avenue...
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Prison Terms Reduced
Posted: 08/31/2006 - The alleged leader and two others involved in the Operation Sandshaker drug ring in Pensacola had their prison sentences reduced.
Posted: 08/31/2006 - The alleged leader and two others involved in the Operation Sandshaker drug ring in Pensacola had their prison sentences reduced.
Updated: 08/30/2006 - Tyndall Air Force Base held a big sendoff for some of the troops being deployed across seas. Some troops will be leaving Thursday, some at a later date, but the mission at hand was to make sure all the families know what services can be the base offers.
Updated: 08/30/2006 - For the first time in awhile, we're seeing a little relief at the gas pump. Prices have dropped, even since last week, and could be going down even more, but why and how long will these prices remain?
Updated: 08/31/2006 - The intense fighting in Iraq continues to claim lives every day. Saturday several U.S. servicemen died, including the son of an area man.
Updated: 08/30/2006 - There's an old saying, "You can't fight City Hall." Well, the folks in Bay County would disagree based on what's happened at Panama City City Hall in the past 24 hours.
Updated: 08/30/2006 - Floridians checking for damage from Tropical Storm Ernesto won't find much.
Updated: 08/30/2006 - Sixteen-year-old Tavish Greene remains in the hospital after police say someone shot him in the leg while in his cousin's car.
Updated: 08/30/2006 - There are some complaints surfacing about the way local health officials are treating an instance of a contagious disease. It involves 2-dogs that died from rabies.
Updated: 08/30/2006 - The president of a local veterans group says he's disappointed that his fellow veterans may be behind a recent wave of dirty politicking.
Updated: 08/29/2006 - State emergency officials are urging Floridians not to do anything foolish as Tropical Storm Ernesto moves ashore in South Florida.
Posted: 08/29/2006 - Two public figures are facing problems after brushes with the law.
Posted: 08/29/2006 - It's been five years since Valparaiso city leaders promised to turn a swampy area into a park where a nine-year-old died of a mosquito-borne virus.
Posted: 08/28/2006 - Tropical Storm Ernesto Moving Over Eastern Cuba...Forecast to Impact the Keys and the Southern Peninsula as a Strong Tropical Storm or Category 1 Hurricane Late Tuesday Night into Wednesday.
Posted: 08/28/2006 - A 27-year-old Fort Walton Beach man suspected of beating, choking and pistol-whipping his girlfriend has been arrested by the Okaloosa County Sheriff's Office.
Updated: 08/28/2006 - A 16-year-old boy is facing charges for allegedly attacking a school bus driver last week.
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Updated: 08/28/2006 - With a little more than a week before voters pick a District 6 State House Republican nominee, things are turning ugly.
Posted: 08/28/2006 - A volunteer coach with the Carrabelle High School football team could face charges for allegedly attacking a player.
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Posted: 08/25/2006 - The Walton County Sheriff's Office is currently investigating the death of 23-month-old Carolyn Ann Martin at 214 Cottage Hill Road in Mossy Head, Florida.


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The world's most passionate online gaming community can now connect, compete and improve their Call of Duty® multiplayer experience through Activision's innovative new online service, Call of Duty Elite.
Developed from the ground up by Activision's new Beachhead studio, Call of Duty Elite offers cutting-edge features that significantly enhance the franchises' multiplayer experience and deliver a new level of social engagement that will unite more than 30 million players worldwide. Additionally, the service will offer exclusive,…
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Day baseball, natural grass, broadcast television, print newspaper ...
Day baseball, natural grass, broadcast television, print newspaper ...
-LTlabnerIf you can't build a winning team with that core a fire-sale isn't the solution. Selling the franchise, moving them to Nashville and converting GABP into a used car lot is.
I love to read. If people didn't like to read, internet news sites would have flourished the way they have. I just want to read something that's current and intelligent. Recycled, out-of-date, formulaic "news" just doesn't get the job done. Especially if I'm paying for it and have to search through dozens of pieces of unwieldy paper to find the particular thing I'm interested in.
I think the newspaper industry is going to have to change fundamentally due to the fact that it no longer controls the distribution of timely written information. It's an era of specialization. If I want comics, I'll get them at a place specializing in comics, perhaps directly from the source. If I want political news, I'll find a site specializing in up to date politics. If I want an in-depth investigative report, opinions, restaurant reviews, etc, there are places who do that better and who are, now, equally if not more accessible -- and often free. And the things people want to read about are also being changed. People are no longer as interested in reading about what the newspaper editor feels is the top story of the day. Chances are, they've already heard about and if they wanted to learn more, could have done so already. They no longer have a captive audience.
And I'm not stuck to my computer. I can print what I need/want to take with me with increasing ease. Heck, I can even get it on my Blackjack if I'm so inclined. It's simply a business model built on a fee-based product that is no longer unique it in it's ability to serve its purpose.
Last edited by RedsManRick; 03-12-2008 at 03:12 PM.
Games are won on run differential -- scoring more than your opponent. Runs are runs, scored or prevented they all count the same. Worry about scoring more and allowing fewer, not which positions contribute to which side of the equation or how "consistent" you are at your current level of performance.
Nearly every newspaper has a Web site, so it's not a matter of whether to do online news, but to do it well. As for the paper, the best thing to do is to put out a paper for the people who want news on paper, for however long that lasts. Some people, as mentioned above, still like to sit down with a paper and a cup of coffee and read the news that way. The paper should step up and meet their needs.
But I'm working on my concept of Pizza 2.0: Give away the pizza and sell advertising on the box.
Increasingly, you will see "free" papers, given away at Subway stations and other high-traffic places. Europe is full of them, to the point that some cities were considering taxing them (the papers) due to the refuse they generate.
"A person is smart. People are dumb, panicky, dangerous animals and you know it."
-LTlabnerIf you can't build a winning team with that core a fire-sale isn't the solution. Selling the franchise, moving them to Nashville and converting GABP into a used car lot is.
Just of out curiosity, if you were Castellini and you had bought The Cincinnati Post instead of the Reds, what would you do with it?
Google Reader and RSS Feeds have pretty much eliminated any need whatsoever for me to buy printed copy any more.
Just about anything I want to read has an RSS feed (except Redszone) and I have 200+ feeds in my Google Reader.
Slowly news organizations are figuring out a way to make money off their web content - it's just taken them way too long to do so. I have a feeling newspapers will be like Radio is decades after the explosion of television. You'll still be able to pick one up, but their function and importance in our daily lives has been reduced significantly.
Still believe that no one comes close to covering local, community news like newspapers. Our free paper for my community of 30k people is excellent and I never miss picking it up at Walgreens every Friday when it comes out. I'm on the extreme suburban fringe of a major metro area and our two major papers might combine for 3 stories a week about our town. That's why I wouldn't bother dropping a dime to pick either up anymore.
Do you think our Fantasy Post newspaper should try to be bigger than the Enquirer, or should it be a lot of little Florence Recorders?
I know it's a fantasy but the first things you're going to need are a printing press and a way to get the papers to the subscribers. The Post was using the Enquirer's facilities for that and that ship has sailed. If you're going to compete against the Enquirer you also have to take into account that you're not just going up against the Enquirer but Gannett as well.
The Rally Onion wants 150 fans before Opening Day.
While I can appreicate the newspaper afficanados, the change in the delivery system is a wonderfull stepforward. It's not without it's pitfalls (as has been noted), but moving to a more decentralized, pseudo real-time system with controbutions from a wider number of sources is far more valuable for delivering news than the old model.
Newspapers are great if you want to read yesterdays news, stock figures, sports transactions, etc. But if you want more current info, in some cases real-time, an electronic delivery system simply makes far more sense. Especially with advances into newsreaders, RSS, internet capabile cellphones, etc.
The electronic media makes even more sense if you travel frequently, as I do. Paying for a paper that will be 2 to 4 days old when I get back home to read it is a complete waste of money and paper. While I could only get the Sunday paper now the time factor is even more of an issue.
Although my poor Grandfather simply can't compute that my computer screen can deliver the news just as his paper does to him. Several times now I've commented, "I read in the news that blah blah blah........" and I can see in his eyes he has no idea what I am talking about since he knows I don't get a paper.
Last edited by Ltlabner; 03-13-2008 at 06:55 PM.
a super volcano of ridonkulous suckitude.
I simply don't have access to a "cares about RBI" place in my psyche. There is a "mildly curious about OBI%" alcove just before the acid filled lake guarded by robot snipers with lasers which leads to the "cares about RBI" antechamber though. - Nate
My local newspaper recently stopped providing delivery and newsstand service to a few cities over 100 miles distant because the cost of getting papers there had become greater than the revenue it was generating. The few thousand subscribers who were cut off were hopping mad at being told the only way they could now receive their San Antonio paper is to have it emailed as a PDF file. As fuel prices increase, you have to figure this will start happening in other places around the country.
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