text:
|Numéro de publication
|Type de publication
|Numéro de demande
|Date de publication
|26 avr. 1988
|Date de dépôt
|30 mars 1981
|Date de priorité
|30 mars 1981
|État de paiement des frais
|Numéro de publication
|David C. House
|Cessionnaire d'origine
|Data General Corporation
|Exporter la citation
|BiBTeX, EndNote, RefMan
|Citations de brevets (8), Référencé par (10), Classifications (6), Événements juridiques (4)
|Liens externes: USPTO, Cession USPTO, Espacenet
1. Field of the Invention
The invention relates generally to circuits for synchronization of received data and more particularly to a circuit for recovering a data clock pulse from an NRZI coded data stream.
2. Description of the Prior Art
Synchronous transmission of data, i.e. data which is synchronous with a timing signal, is commonly coded in NRZI (non-return to zero inverted) form. In this form, a zero, or low, data state is indicated by a transition in the data stream from low to high or high to low. A one, or high, data state is indicated by the absence of a transition in the data stream. In cases of data transmission using a modem where the data clock is derived or "recovered" from the received data, NRZI coding is typically used to insure that an adequate number of data stream transitions occur for accurate generation of the timing signal.
In addition, "protocols" or generalized techniques of formating and structuring the data to be transmitted have been developed. Two common protocols are Synchronous Data Link Control and the High-level Data Link Control. These techniques vary in some respects, but have many features in common. One of these features is termed "zero insertion", which ensures that, except for specifically designated control characters, no more than five consecutive high signals will be received before a low signal will be seen. In other words, a data stream transition will appear after no more than six bit times.
Synchronous modems typically provide both a transmit clock for use in performing the NRZI encoding and a receive clock for use in decoding the received data. Some modems, however, provide only a transmit clock and lower cost asynchronous modems generally provide neither. Timing for the received data stream in the absence of a receive clock is generally accomplished through use of an analog phase-locked loop. This technique typically compares the transitions in the received data with a source frequency. If they are not coincident, an error voltage is generated which raises or lowers the source frequency, within limits, until the clock is coincident with the data stream transistions.
The phase-locked loop technique has certain disadvantages. For example, individual source frequencies must be generated by discrete components having varying tolerances. Costly components must generally be used to obtain a precise source frequency. In addition, if multiple data frequencies must be supported, separate circuits must typically be used to determine each frequency.
The present invention relates to a novel circuit for recovery of a data clock from an NRZI coded data stream which is free of the above mentioned problems.
The present invention relates to apparatus and method for recovering a data clock signal from a received NRZI coded data stream. A circuit for implementing the invention includes a source clock signal, apparatus for detecting a data stream state transition, apparatus for generating a data clock signal, and apparatus for resetting the data clock generation apparatus when a data stream transition is detected.
It is a feature of the present invention that the source clock has a frequency which is an integer multiple of the data rate.
It is a further feature of the invention that the apparatus for generating the data clock signal is a counter clocked by the source clock signal.
It is an advantage of the invention that it can be constructed using small- and medium-scale integration TTL circuits.
It is another advantage of the invention that multiple data rates can be accommodated.
It is yet another advantage of the invention that multiple synchronization resolution requirements can be accommodated.
Other features and advantages of the present invention will be understood by those of ordinary skill in the art after referring to the detailed description of a preferred embodiment and drawings herein.
FIG. 1 is a block diagram showing an overview of the present invention.
FIG. 2 is a schematic diagram of an NRZI decoding and data clock recovery circuit included within FIG. 1.
FIG. 3 is an NRZI data decoding and data clock recovery timing diagram.
Referring to FIG. 1, a block diagram of the present invention is shown. Line Multiplexer 100 could be utilized in various ways, for example, it might occupy an input/output slot in a host computer (not shown) or might be resident in a communications chassis which is, in turn, connected to a host computer. Line Multiplexer 100 includes, among other features, Encode Logic 101 and Decode/Clock Recovery Logic 102. Encode Logic 101 receives UXDATA (uncoded data to be transmitted) from the host computer or a processor in Line Multiplexer 100 and XDCLOCK (transmit clock signal) from Modem 103. If Line Multiplexer 100 can produce coded XDATA with its own internal clock, XDCLOCK from Modem 103 can be eliminated, thereby allowing use of a lower cost asynchronous modem. Encode Logic 101 supplies XDATA (NRZI coded data) to Modem 103.
Decode/Clock Recovery Logic 102 receives RDATA (NRZI coded received data) from Modem 103 and supplies URDATA (uncoded received data) and RDCLOCK (recovered data clock signal) to the Line Multiplexer 100 processor or the host computer.
Modem 103 either modulates the XDATA bit stream to generate CDATA (communication line compatible data) for transmission or receives a CDATA bit stream and demodulates it to generate the RDATA bit stream.
Referring now to FIG. 2, a schematic diagram of a preferred embodiment of Decode/Clock Recovery Logic 102 is shown.
SCLOCK is a source clock signal selected to have a frequency which is an integer multiple of the data frequency. In the preferred embodiment discussed below, the source frequency has been chosen to be 16 times the data frequency. That is, 16 SCLOCK pulses will occur for each data bit time. SCLOCK is supplied as the clocking input to flip flops 203 and 205 and to counter 207. Pull up voltage UPS, maintained high, is provided to the S and R inputs of flip flops 203 and 205, the K input of flip flop 209, and the P, T, PE, 0, 1, 2, and 3 inputs of Counter 207.
RDATA is inverted by invertor 204 and supplied to the D input of flip flop 203. The Q output of flip flop 203 is connected as the D input of flip flop 204 and as one of the inputs to X-OR gate 206. The Q output of flip flop 205 is also connected as an input of X-OR gate 206. The output of gate 206 is supplied to the reset input of counter 207 and, after inversion by inverter 210, to the clocking input of flip flop 209.
The 0 and 3 outputs of counter 207 are provided as inputs to NAND gate 208, the output of which is connected to the S input of flip flop 209. The 3 output of counter 207 is the source of the RDCLOCK signal. The R input of flip flop 209 is connected to the RA signal from the receiver hardware indicating receiver active status. The J input of flip flop 209 is held low. The Q output of flip flop 209 is the source of URDATA.
SCLOCK, as mentioned above, has been selected in this embodiment to be 16 times the data rate (i.e. N=16). The selection of the SCLOCK frequency depends on such variables as data rate and bit distortion. An SCLOCK to RDCLOCK ratio of 32 to 1 (N=32) might be required at higher data rates, while a ratio of 8 to 1 (N=8) or less could be acceptable at slower data rates. While the ratio of frequencies need not be a power of two, such ratios are normally convenient to work with and lend themselves to making several SCLOCK rates available to the user by repetitive division of a master clock signal source. For whatever SCLOCK-to-data rate multiple is used in a particular application, the outputs of the one or more SCLOCK pulse counters must be connected such that the RDCLOCK pulse occurs N/2 SCLOCK pulses after data transition (sustantially equal to 1/2 of a bit time) and every N SCLOCK pulses thereafter until the one or more counters are reset by another data transition. This RDCLOCK signal generation timing ensures data sampling will occur substantially at the center of the data bit. Because of the possibility of bit distortion, sampling at or very near the center of a data bit may be critical to avoid erroneous data readings.
Referring to FIG. 3, a sample timing diagram for operation of Decode/Clock Recovery Logic 102 is shown. For illustration, FIG. 3 shows an NRZI coded 0010 RDATA bit stream.
Prior to time t1, RDATA is high. Therefore 203Q is high, 205Q is low, and the output of X-OR gate 206 is high. A low bit is being transmitted, therefore a data transition will occur. At the first SCLOCK rising edge (time t1) after the change in RDATA state, 203Q is driven low. Since 205 Q will not go high until the following SCLOCK pulse, both inputs to gate 206 are now low. This causes the output of gate 206 to go low, which resets up counter 207.
At the next SCLOCK pulse after time t1, 205Q is driven high. This changes the output of gate 206 to high and, because of inverter 210, presents a falling edge clocking pulse to flip flop 209. Clocking of flip flop 209 forces URDATA low.
At time t2 (i.e. the eighth SCLOCK pulse after time t1) the 3 output of counter 207 goes high. This output is the source of data clock signal RDCLOCK. As discussed above, it can be seen that counter 207 generates RDCLOCK at substantially one-half of a data bit time after the received NRZI data transition has been detected. The RDCLOCK rising edge will be used as the clock pulse for sampling of the URDATA bit stream. To ensure it has been properly read, URDATA is held for one SCLOCK pulse time after generation of RDCLOCK. At the next SCLOCK pulse (i.e. the ninth pulse after time t1), both inputs to NAND gate 208 will be high. This presents a low signal to the S input of flip flop 209. If 209Q (i.e. URDATA) is not already high, the low signal at 209 S will force it high in preparation for the next data bit.
Another low bit is now being transmitted. Since the previous low bit was indicated by a transition from high to low, the next low bit will be indicated by a transition from low to high. At the next SCLOCK pulse after the data state transition (time t3), 203Q is driven high. Both inputs to X-OR gate 206 are now high, therefore the output of 206 goes low. This again resets counter 207. It can be seen that counter 207 is being reset at each data stream transition (ie. low data bit), therefore, RDCLOCK is being frequent1y resynchronized with the incoming data.
At the first SCLOCK pulse after time t3, 205Q is driven low, causing the output of gate 206 to go high and presenting a falling edge clocking pulse to flip flop 209. This forces 209Q to a low state.
At time t4, the 3 output of Counter 207 (i.e. RDCLOCK) again goes high. At the next SCLOCK pulse after t4 the inputs to gate 208 are again satisfied and 209Q is driven high. Since the following bit received is high, no transition is detected in the data stream at time t5. RDCLOCK goes low at time t5, but does so because it has filled up and restarted, not because it has been reset by a data transition. The greater the number of consecutive high data bits received, the more the RDCLOCK derived from Counter 207 will begin to diverge from actual received data timing. However, as stated earlier, typical protocols require that no more than five consecutive high data bits be transmitted. RDCLOCK is, therefore, ensured of being resynchronized with the data after no more than six data bit times of "free running". In the example shown in FIG. 3, Counter 207 is reset at time t3 and is not reset again until time t7. This is because a high bit was received between the two low bits.
The advantages and benefits of the invention can now be clearly understood. The invention provides a digitally recovered data clock signal for use in reading the uncoded received data stream. This technique is significant1y less expensive than a comparable phase lock loop circuit, since the more expensive discrete components and phase lock loop chips are not required. In addition, SCLOCK can be jumperable to several different frequencies to allow the circuit to accommodate different data rates.
The invention may be embodied in yet other specific forms without departing from the spirit or essential characteristics thereof. For example, the synchronization resolution can be increased by increasing the source frequency to 32 times the data bit rate and ganging two up counters.
The present embodiments are therefore to be considered in all respects as illustrative and not restrictive. The scope of the invention is indicated by the appended claims rather than by the foregoing description, and all changes which come within the meaning and range of equivalency of the claims are therefore intended to be embraced therein.
|Brevet cité
|Date de dépôt
|Date de publication
|14 sept. 1965
|6 déc. 1966
|Bell Telephone Labor Inc
|Digital frequency control circuit
|31 juil. 1967
|27 oct. 1970
|Digital phase lock loop
|15 mai 1970
|6 juin 1972
|Hughes Aircraft Co
|Receiver timing and synchronization system
|16 nov. 1970
|10 juil. 1973
|North American Rockwell
|Coarse initial timing recovery circuit
|6 nov. 1972
|25 juin 1974
|Trt Telecom Radio Electr
|System for synchronous data transmission through a digital transmission channel
|17 oct. 1977
|17 juin 1980
|Sperry Corporation
|System and method for clocking data between a remote unit and a local unit
|13 déc. 1978
|7 oct. 1980
|Nippon Telegraph And Telephone Public Corporation
|Clock pulse regenerator
|1 févr. 1980
|5 janv. 1982
|Telecommunications Radioelectriques Et Telephoniques T.R.T.
|Circuit for rapidly resynchronizing a clock
|Brevet citant
|Date de dépôt
|Date de publication
|13 janv. 1989
|8 janv. 1991
|Plessey Company Plc
|Circuit arrangement for use in the time division multiplexed signalling system
|14 juin 1989
|18 juin 1991
|Telettra-Telefonia Elettronica E Radio S.P.A.
|System and circuits for the processing and transmission of a clock signal, in particular of the video synchronization frequency
|20 déc. 1990
|7 juil. 1992
|Unisys Corporation
|Non-return to zero synchronizer
|5 mars 1991
|15 déc. 1992
|National Semiconductor Corporation
|Single channel serial data receiver
|1 avr. 1991
|16 mars 1993
|Nec America, Inc.
|Clock recovery and decoder circuit for a CMI-encoded signal
|27 mars 1991
|7 sept. 1993
|Kabushiki Kaisha Komatsu Seisakusho
|Encoding method and code processing circuitry
|22 déc. 1992
|9 juil. 1996
|3Com Corporation
|Port circuit for a token ring concentrator having phase lock loop timing recovery with additional circuitry to verify appropriate signals
|11 mars 1996
|9 févr. 1999
|Adtran, Inc.
|Mechanism for automatically adjusting the phase of a transmission strobe clock signal to correct for misalignment of transmission clock and data signals
|17 févr. 1999
|26 déc. 2000
|Silicon Storage Technology, Inc.
|Circuit for turning on and off a clock without a glitch
|27 juil. 2001
|3 aoűt 2004
|The Pulsar Network, Inc.
|System for extracting a clock signal and a digital data signal from a modulated carrier signal in a receiver
|Classification aux États-Unis
|375/360, 375/371, 327/166
|Classification internationale
|Classification coopérative
|Classification européenne
|30 mars 1981
Owner name: DATA GENERAL CORPORATION, WESTBORO, MA, A CORP. OF
Free format text: ASSIGNMENT OF ASSIGNORS INTEREST.;ASSIGNOR:HOUSE DAVID C.;REEL/FRAME:003875/0776
Effective date: 19810327
Owner name: DATA GENERAL CORPORATION, WESTBORO, MA, A CORP. OF
Free format text: ASSIGNMENT OF ASSIGNORS INTEREST;ASSIGNOR:HOUSE DAVID C.;REEL/FRAME:003875/0776
Effective date: 19810327
|17 sept. 1991
|Fee payment
Year of fee payment: 4
|8 sept. 1995
|Fee payment
Year of fee payment: 8
|20 sept. 1999
|Fee payment
Year of fee payment: 12


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Most actors would shy away from the responsibility of portraying a character with Asperger syndrome, an autism-spectrum disorder marked by difficult social interactions and highly repetitive behavior. But Max Burkholder, who portrayed the Asperger-afflicted teen Max Braverman on NBC's "Parenthood" (2010-15), took pride in his collaboration with medical experts and researchers, and at the positive reception given by critics and audiences alike. Though comedy ruled in his debut film "Daddy Day Care" (2003), Burkholder quickly established himself as a thoughtful and talented actor. A string of minor TV appearances in the mid-2000s led the California-born actor to voice-over roles on a number of series, including "Family Guy" (FOX, 1999-2003, 2005- ) and "The Land Before Time" (Cartoon Network, 2007-08). After switching gears with supporting roles in dramas such as "Brothers & Sisters" (ABC, 2006-2011) and HBO's "In Treatment" (2008-2010), Burkholder took on the highly intelligent Max in "Parenthood." Put together with his expansion into horror in "The Purge" (2013), Burkholder proved capable of handling just about anything.

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text:
|Publication number
|Publication type
|Application number
|Publication date
|Mar 4, 1997
|Filing date
|Jun 23, 1993
|Priority date
|Jun 23, 1993
|Fee status
|Publication number
|Bryan L. Burns, J. Thomas Cordaro
|Original Assignee
|The United States Of America As Represented By The United States Department Of Energy
|Export Citation
|BiBTeX, EndNote, RefMan
|Patent Citations (22), Non-Patent Citations (12), Referenced by (98), Classifications (10), Legal Events (5)
|External Links: USPTO, USPTO Assignment, Espacenet
The invention herein relates to the field of imaging synthetic aperture radars. More particularly, the invention relates to a linear-FM, synthetic aperture radar (SAR) which implements overlapped-subaperture real-time processing to correct for motion through resolution cells.
As with many applications of imaging devices, the finer the resolution the more useful the imaging devices become; however, the spatial resolution achievable by an antenna, without any coherent processing, is inversely proportional to the antenna's aperture. Thus, for fine resolution, a large antenna is required. To overcome the inability to place a large antenna on an imaging system in motion such as an aircraft or satellite, SAR samples a large aperture by positioning a small antenna at well defined locations along the large aperture. These samples are then coherently combined in the radar's signal processor to generate, synthetically, the resolution appropriate for the aperture traveled by the small antenna. In order to coherently combine the samples, the relative position of the antenna must be known to within a fraction of a wavelength. If the synthetic aperture to be sampled by the radar is a straight line, components of motion away from the line degrade the coherent combination of the samples of the synthetic aperture. The angular relationship between the line-of-sight vector to any point in the scene to be imaged and the undesired motion vector appears as phase modulation of the radar echo when viewed at the radar receiver. Spatially varying phase errors, also dependent on the motion of the radar, occur because the effect is not constant for all locations in the scene.
To achieve finer resolution image processing, the signal processing burden must increase at least as rapidly as the inverse of the resolution squared, assuming that the image area is constant and the X and Y dimensional resolution change at the same rate. Thus, it appears that the requirements for both a real-time, imaging SAR, and fine resolution are in conflict. For this reason, radar systems which efficiently utilize computing devices are particularly important to advancing the state-of-the art in real-time SAR.
Synthetic aperture radars are known in the art. For example, much background is described in "Developments in Radar Imaging", IEEE Transactions On Aerospace and Electronic Systems, Volume AES-20, Number 4, July 1984.
U.S. Pat. No. 4,771,287 to Mims describes a method of correcting for focus errors and higher order errors that occur in radar processing. The system includes first and second stages for processing radar signals over a plurality of time samples. The first stage derives the signals at each time sample and forms a set of complex signals of point-like scatterers to form a time history. Focusing errors are corrected by generating a set of correcting signals. The method, however, compensates for phase errors common to range and azimuth locations within the imaged area, but cannot compensate for phase errors that vary as a function of location.
U.S. Pat. No. 4,758,838 to Maeda et al. describes a method of reconstructing original images from synthetic aperture radar image data. In order to speed up the reconstruction of the original image the process for generating a point image pattern and the Fast Fourier Transformation process are carried out once for every certain number of lines. Positional displacement is corrected by multiplying a phase rotation factor by the product of the Fourier image pattern and one line of the image data. The method uses range compression before azimuth compression. However, the method does not use two stages of azimuth compression which implements a subaperture approach.
U.S. Pat. No. 4,034,370 to Mims describes a method using two stage azimuth processing with range migration compensation being processed after range compression. The method does not combine initial stage azimuth filtering with linear-FM transmitter pulse processing to implement a range-migration correction with a phase rotation.
The prior art processing schemes require inefficient interpolation to compensate for range migration. Moreover, prior art radar architectures suffer from unduly large computational burden and memory requirements of the signal processor by not correcting the received radar phase for undesirable, motion-induced, errors using phase rotations in the local oscillator and sampling rate changes in the A/D converter.
Prior art radar architectures calculate in general purpose computers the time-varying coefficient vectors which are needed in the signal processor and do not synthesize these vectors with digital hardware. Note that range and coarse-resolution azimuth positions are known before the final processing stage, so that some spatially dependent phase errors that limit other techniques such as polar format processing are eliminated.
It is thus an object of the present invention to account for spatially varying effects of motion induced phase errors to achieve high image quality over large scene areas. This object is achieved in the present invention by, inter alia, correcting the radar phase for undesirable, motion-induced, errors using phase rotations in the transmitter, receiver, and signal processor and sampling rate changes in the A/D converter.
It is another object of the present invention to provide a radar and method for imaging that can be efficiently realized in digital hardware.
It is yet another object of the present invention to provide an apparatus and method for radar imaging which implements the method with signal processing operations.
It is yet another object of the present invention to minimize the computational burden and memory requirements of signal processing equipment.
It is yet another object of the present invention to organize processing requirements so that slow time varying coefficient vectors may be produced in general purpose computers and rapidly time varying coefficient vectors may be synthesized in hardware synthesizers.
It is thus an object of the invention to improve image quality at fine resolution of synthetic aperture radar which is degraded because the relative range of near and far images of a scene changes with motion of the radar. The radar herein is designed to change the radar center frequency, fo, and either the signal sampling rate or the chirp rate based on actual radar motion using a digitally-controlled waveform synthesizer. A further advantage is that this improved image quality can be accomplished in a real-time system without adding to the burden of the radar's signal processor.
It is a further object of the invention to obtain fine resolution with minimal memory requirements. Because of the filtering operation at coarse resolution processing, only A<M coarse-resolution azimuth bins need to be saved, thus achieving the reduction in real-time memory. Moreover, some of the signal processing operations are realized by varying the radar center frequency and either the chirp rate or the A/D sample rate. This process reduces memory requirements.
It is yet another object of the invention to achieve a synthetic aperture radar compensating for motion through range cells. This object is achieved when the spectrum of the signal in each subaperture is shifted by an amount that is proportional to the azimuth bin number of that column.
It is yet another object of the invention to allow for real-time processing concurrent with data collection. Certain functions, namely the coarse-resolution azimuth and the range processing, operate on one subaperture at a time wherein the data are reduced.
It is yet another object of the invention to remove the geometric distortion caused by range-doppler interaction. The feature of the invention which removes the geometric distortion is the use of the chirp-z transform as a last step in the processing. Moreover, a further advantage of this feature is that the invention produces an image with constant azimuth-sample spacing.
These and other objects are achieved with an imaging radar including a linear-FM type doppler radar for producing a received signal; a coarse-azimuth process for producing a plurality of subaperture arrays from the received signal, each subaperture array including a plurality of coarse-azimuth bins; a range process for producing a plurality of range bin vectors, each range bin vector having a plurality of range bins and being produced from a corresponding coarse-azimuth bin of the plurality of coarse-azimuth bins of each of the plurality of subaperture arrays according to the coarse-azimuth bin; and a fine-azimuth process for producing a radar image including a plurality of image lines, each image line being produced from a corresponding range bin of the plurality of range bins of each range bin vector of the plurality of range bin vectors.
These and other objects are achieved with a method of radar imaging including the steps of producing a received signal from a doppler radar having a linear-FM waveform; producing a plurality of subaperture arrays from the received signal, each subaperture array including a plurality of coarse-azimuth bins; producing a plurality of range bin vectors, each range bin vector having a plurality of range bins and being produced from a corresponding coarse-azimuth bin of the plurality of coarse-azimuth bins of each of the plurality of subaperture arrays according to the coarse-azimuth bin; and producing a radar image including a plurality of image lines, each image line being produced from a corresponding range bin of the plurality of range bins of each range bin vector of the plurality of range bin vectors.
The invention will be described in detail in the following description of the preferred embodiments with reference to the following figures.
FIG. 1 is a drawing illustrating the geometry of the radar and a point scatterer.
FIG. 2 is a functional block diagram of the synthetic aperture radar of the invention.
FIG. 3 is a signal processing flow chart of the invention.
FIG. 4 is a subaperture formation timing diagram.
FIG. 5 is an image graphic of the imaging area slant range versus doppler frequency for Vy =Vz =0.
The imaging radar is a linear-FM, synthetic-aperture radar (SAR). Because of the challenging system requirements to achieve fine resolution, real-time image processing, and an ability to synthesize a large aperture, the invention embodies new SAR signal analysis processes using fast fourier transform (FFT) and vector-multiplication operations. These operations may be implemented with special-purpose hardware.
With reference to FIG. 1 which describes the geometry of the SAR described herein and illustrates the first of an XYZ coordinate system 1 which is fixed with the XY plane tangent to the earth's surface and the X axis along a nominal motion path 3. The central reference point 10 is in the XY plane at a range rc 12 from the antenna phase center 4. A second coordinate system 2, defined by the orthonormal vectors ua, ur, and uz, is centered at the central reference point 10, with uz parallel to the Z axis. The coordinates of the point scatterer 8, =(sa, sr, sz), are expressed in the second coordinate system 2. The range from point scatterer 8 to the antenna phase center 4 is rs 14. For broadside imaging ua is parallel to the X axis. The variables rs and rc are time functions, and is a constant.
Other geometric variables in FIG. 1 describe motion compensation. During motion, the center of the antenna beam on the ground moves along a straight line 22 in the XY plane parallel to the X axis wherein the variable x 24 is the distance along the X axis from the central reference point 10 to the beam center. The velocity of the beam center along this line is Vx. At the center of an aperture, x=0. For the situation in FIG. 1, x is negative. Finally, the line of sight from the antenna phase center 4 to the beam center on the ground makes an angle φ 20 with the XY plane and has length rb 16. The variable rb 16 and φ 20 are functions of Vy and Vz, the Y and Z components of the antenna velocity. If the radar traveled exactly along the nominal motion path 3, then φ 20 and rb 16 would be constant, and rb 16 would equal the nominal range rm 18, but the radar travels along the actual motion path 6. If rsc is defined as the differential slant range, rsc =rs -rc then the range to the point can be written as:
r.sub.s ≃r.sub.sc +r.sub.b +x.sup.2 /(2r.sub.b) (1)
where the last two terms are from the power series for rc.
Given the geometry of the path of the motion of the radar and the scene to be imaged, the basic approach of the SAR invention is summarized in the functional block diagram of FIG. 2.
The purpose of motion compensation is to keep the radar returns from various scatterers in fixed range-doppler resolution cell during a coherent processing interval. So to start, motion compensation can be divided into two parts. First, the radar-return signals are adjusted so the range and doppler frequency corresponding to the central reference point (10 in FIG. 1) in the scene are constant, see block 30 in FIG. 2. For signals from scatterers sufficiently near the central reference point (10 in FIG. 1), this adjustment holds the range and doppler-frequency variations to a fraction of a resolution cell, and no other compensation is needed.
But for imaging a large scene at fine resolutions, the signals from scatterers farther away from the central reference point (10 in FIG. 1) have range and doppler-frequency variations of several resolution cells during a coherent processing interval. Thus, the second aspect of motion compensation of the invention to achieve fine-resolution image of a large scene is the need to correct for motion through resolution cells. The imaging radar described herein employs an overlapped subaperture processing approach to compensate for motion through resolution cells in a novel way. Motion through range cells is largely a function of range rate or doppler frequency. An initial coarse-resolution azimuth-processing stage 32 sorts dechirped baseband signals into doppler-frequency bins corresponding to a coarse-azimuth position. Next, the signal in each bin is corrected for range variations 34. Because the radar transmits a linear-FM pulse, the correction is a frequency shift in the dechirped baseband signal that is proportional to the doppler bin number implemented by a simple vector multiplication compatible with signal processing equipment. The range-stabilized dechirped baseband signals are Fourier transformed to obtain fine resolution in range.
The remaining motion is through doppler cells and is a function of both the range and azimuth coordinates of a scatterer within a scene to be imaged. The azimuth dependence is weak, so the process corrects for motion through doppler cells using fine-resolution range and coarse-resolution azimuth information as another vector multiplication. Some spatially dependent phase errors are eliminated because the range and coarse-resolution azimuth positions are known before the final processing stage.
The doppler-stabilized signals are then Fourier transformed to obtain fine resolution in azimuth, see block 36 of FIG. 2. The bandwidth and frequency-sample spacing in this transform are varied as functions of range to hold the azimuth width and sample spacing constant in the output image.
The first stages of the SAR image processing illustrated in FIG. 2 begins with the transmitted signal from transmitter 40. The transmitter 40 of the invention generates a linear-FM waveform with a transmitter starting frequency ft, a center frequency fo, starting phase φt, and chirp rate γt. Digitally-controlled waveform synthesizer 44 permits the transmitter starting frequency ft, starting phase φt, and chirp rate γt to be changed on a pulse-to-pulse basis based on actual radar motion. These radar pulses are transmitted at a pulse repetition frequency PRF. The achievable range resolution of a linear-FM waveform is inversely proportional to the total transmitted bandwidth; for example, BW≈150 MHz/ρr where ρr is the desired range resolution in meters. If the total transmitted bandwidth is τt �γt then γt =150 MHz/(τt �ρr). The transmitted waveform is pulse modulated with pulse width τt by an RF switch 42 and amplified by the transmitter 40. The amplified transmitter signal is coupled to antenna 48 through a circulator 46.
The energy radiated by the antenna 48 is then propagated to the scene in the field of view of the antenna 48 and scattered by the objects in the field of view. The intensity of the back scattered energy is proportional to the radar cross section of the objects the wave strikes. The backscattered energy propagates back to the antenna 48 where it is received and coupled to receiver 50 through the circulator 46.
At the receiver 50, each pulse is dechirped, mixed to baseband, and sampled to produce a complex dechirped baseband signal. More specifically, the receiver 50 amplifies the signal, and mixes it in mixer 52 with receiver local oscillator 54 which has a starting frequency f1, starting phase φ1, and chirp rate γl. The receiver local oscillator 54 is pulse modulated with pulse width τ1. The receiver local-oscillator pulse width is greater than the transmitter pulse width by the difference in the time it takes an electromagnetic wave to propagate from the radar to the near side of the scene and back to the antenna and the time of propagation from the antenna to the far side of the scene and back to the antenna. The local oscillator chirp rate γl is set equal to the transmitter chirp rate and both shall be herein represented as γ. The local oscillator starting frequency and starting phase of the receiver local oscillator 54 are adjusted on a pulse-to-pulse basis depending upon actual motion of the radar. When coarse-resolution changes in the radar frequency are needed, the local oscillator frequency f1 of oscillator 54 is changed directly; however, second oscillator 56 with very fine-resolution control also generates a frequency translation, fm, determined by the motion of the radar.
The mixing operation in the mixer 52 is a quadrature mixing operation. In-phase, I, and quadrature, Q, components of the receiver signal are produced by mixing the receiver signal with local oscillator signals separated ninety degrees in phase. A pair of analog-to-digital (A/D) converters 58 quantize the data for use by a digital signal processor. The sampling frequency fv and starting time of the dechirped baseband samples are modified on a pulse-to-pulse basis based on measured radar motion. Samples of the dechirped baseband pulses are collected for a length of time τv, where typically, τt ≦τv ≦τ1.
A novel circuit 60 using digital waveform synthesizer 62, bandpass filter 64, one-bit quantizer 66, and switch 67 permits the sampling location and the A/D sampling frequency to be controlled with extremely fine resolution without using a very high frequency oscillator. The clock edges from this circuit can be far more precisely controlled than other methods within the state-of-the art. This circuit 60 generates a burst of clock edges to cause the A/D to sample the dechirped baseband signal.
Inputs to the circuit are the chirp rate of the A/D strobe signal γv, starting frequency fv determined by the desired initial sampling frequency, starting phase φv determined by the location of the A/D samples, the number of samples of the dechirped baseband signal to be taken by the A/D, F, and a delay determined by the delay from the start of the local oscillator waveform from 54 until the switch 67 is closed.
Digital waveform synthesizer 62 is triggered at the same time as the local oscillator waveform synthesizer 54. Bandpass filter 64 is a combination of high-pass filter and a low-pass filter which performs the function of interpolating between the discrete voltage amplitudes from the digital waveform synthesizer so that zero crossings occur with a period equal to the inverse of the frequency synthesized by the digital waveform synthesizer. If the interpolation filter is not used, the zero crossings occur at an integer multiple of the period at which the digital waveform synthesizer 62 is clocked rather than with a period which is inversely proportional to the frequency being synthesized by the digital waveform synthesizer. The one-bit quantizer 66 is a zero-crossing detector which converts the interpolated waveform from the bandpass filter 64 into a square wave. The switch 67 is closed just prior to the rising edge corresponding to the desired time of the first sample of the dechirped baseband signal and held closed until F samples have been taken. The switch 67 is not closed until the quantizer output 66 is in an "off" state so that an extraneous clock is not generated at switch closing.
This circuit 60 provides control of three fundamental parameters of the A/D clock signal: (1) position of the rising A/D strobes, (2) period of the A/D strobes, and (3) rate of change in frequency (chirp rate) of the A/D strobes.
The location of the first A/D strobe may be controlled to a small fraction of the period of the A/D strobes by adjusting the starting phase φv and the delay until the switch 67 is closed. For current digital waveform synthesizer technology the starting phase is settable to within 1 part in 4096 for the period of the signal being synthesized. If 100 MHz A/D strobes are being generated this means that the starting location has a relative accuracy of 2.4 ps using the starting phase. To achieve similar performance using typical digital counter circuit technology would require a counter circuit to function at an input clock frequency of more than 400 GHz. This requirement exceeds today's technology by approximately 2 to 3 orders of magnitude. The precision with which the A/D strobe location may be set is a unique feature of this A/D strobe generator circuit topology.
Additionally the period of the A/D strobes may be precisely controlled by setting the frequency fv. Current digital waveform synthesizer technology provides from 24 to 32 bits of precision in the value of fv at clock frequencies of greater than 400 MHz. If a 10 ns A/D strobe period (100 MHz frequency) is desired, 24 bits of precision is assumed, and the input clock frequency for the digital waveform synthesizer is 400 MHz, the A/D strobe frequency may be set to a precision of approximately 23 Hz. The two adjacent values of the A/D strobe periods (1/100 MHz and 1/100.000023 MHz) would therefore be separated by approximately 2.4 fs which is a smaller period step size by several orders of magnitude than other methods for generating A/D strobe signals for sampling deramped baseband signals. If the 400 MHz clock frequency is used directly in a divide-by-N circuit, the next shorter period than 10 ns is 7.5 ns which is a step size of 2.5 ns, 6 orders of magnitude larger than possible with the strobe generator circuit 60.
Returning now to the discussion of SAR, the functional block diagram of the invention will be read in conjunction with the data flow diagram of FIG. 3. The AC-coupled analog to digital A/D converters 58 in principal sample the received signal from mixer 52 shown to produce received vector 202 shown in FIG. 3. Conceptually, it is useful to think that coarse-resolution azimuth processing 32 begins with bandpass filtering 68. The bandpass filtering 68 is implemented first as a high-pass filter 60 and a presummer 62 wherein the low-pass filters in embodied in dedicated special purpose hardware that executes operations as the radar samples are collected. Gallium arsenide integrated circuits have been developed for each of these functions.
The high-pass filter 70 of the bandpass filter 68 removes the non-zero DC-bias resulting from the drift of the A/D input. The time constant of the high-pass filter 70 is many times longer than a synthetic aperture of data to avoid removing any of the radar data with the high-pass filter 70.
The presummer 72 reduces the data rate for the signal processor by a factor of Np where Np is the number of dechirped baseband pulses integrated into a sample. This is also described as a point-by-point accumulation of the data in Np data vectors, each of length F where F is the number of A/D samples in each pulse and is a function of the time duration of the dechirped baseband pulse and the sample rate. This feature is extremely advantageous in real-time signal processing. This presummer 72 is actually many filters operating in parallel, one for each dechirped baseband sample output from the A/D converter 58 and passed through the high pass filter 70. While the folding frequency of this filter, in the azimuth dimension, is located at approximately the 4 dB point on the filter rejection, the azimuth beamwidth of the radar antenna and the output sampling rate of this filter can be chosen to limit the amplitude of aliased energy. After Np vectors have been accumulated, the resulting vector 204 is sent to the remainder of the radar processor.
To reduce the amount of the dechirped baseband and doppler bandwidth required, the radar removes the range and doppler-frequency components contained within the first rb term in Equation (1). This is accomplished by changing, as functions of rb, the local oscillator phase and local oscillator starting frequency f1.
Let xv (FI,n) be the FIth time sample of the nth dechirped baseband pulse at the presummer output, i.e., the FIth element of the received vector 204 in FIG. 3, where FI=0, 1, . . . , F-1, and n=0, 1, . . . , Na -1, where Na is determined by, inter alia, the azimuth resolution. Then ##EQU1## where AR is an amplitude factor, c is the speed of light, and Ts is the A/D sample period, γ is the chirp rate, and fo is the transmitter center frequency. F is the total number of samples in each dechirped baseband pulse and x (24 in FIG. 1) is the distance along the X axis from central reference point (10 in FIG. 1) to the beam center. The constant rm is set by the time position of the local oscillator pulse, and is taken here to be the nominal range (18 in FIG. 1). Equation (2) shows that varying fo and either γ or Ts as functions of the out-of-plane, in other words, the actual motion applies a scale factor to rsc and removes some motion-induced errors. In one approach, Ts and fo can be varied as functions of cosφ. In view of the radar equation, the variable AR depends on both rs and the position of in the antenna pattern.
Given the xv (FI,n) values, the task of signal-processing is to estimate the position and relative reflectivity of the point scatterers from the expression on the first line of Equation (2). The other two expressions are extraneous terms that are removed at suitable stages. The expression on the second line contains the part of central reference point motion that is quadratic in x. For a coarse-resolution SAR, the dependence of this term on FI is negligible, and it reduces to the usual azimuth chirp. There are two reasons why the x2 /(2rb) factor is not included in the function that controls the receiver dechirp pulse which would eliminate the whole central reference point-motion term at one time. First, it is sometimes useful to record the dechirped baseband signal continually and process it off-line with assorted central reference points and resolutions. But calculation of x at the time a return pulse arrives supposes that the central reference point and the azimuth resolution are determined before data collection begins. And the invention herein accommodates off-line processing. Second, and more importantly, the radar may have multiple range channels, each with the same azimuth resolution. This will require different central reference point locations and aperture lengths in the separate channels. The expression on the third line of Equation (2) is negligible in the case requiring coarse-resolution in azimuth. If the radar varies the time position of the dechirp signal as a function of rc, then the rm is replaced with rc and the term can be omitted if |s| is not too large. Neither of these conditions are applicable.
Motion through resolution cells can be significant over the integration time required to obtain fine resolution in azimuth. The invention handles this problem by dividing the dechirped baseband data collected for the whole synthetic aperture into smaller, overlapping subapertures in which the motion through resolution cells is insignificant within a subaperture.
FIG. 4 shows how the subapertures are defined. Each vertical line 400 represents a single dechirped baseband pulse, i.e., it represents xv (FI,n) for all FI and a fixed n. Recall that for xv (FI,n), FIth represents a particular time sample of the nth dechirped baseband pulse, where FI=0, 1, . . . , F-1, and n=0, 1, . . . , Na -1. The whole F�Na phase-history array is divided into S subapertures 402. A subaperture array 402 is an F�M complex matrix indexed by the integer SI, where 0≦SI≦S-1. Within a subaperture 402, the dechirped baseband pulses are indexed by another integer m, where 0≦m≦M-1. The delay from the beginning of one subaperture to the next is Δ 404. Δ 404 is chosen to constrain the azimuth sidelobes far away from the main lobe below system requirements. From FIG. 4, the number of azimuth samples is related to M and Δ by
N.sub.a =(S-1)Δ+M. (3)
Because the data are initially processed by subapertures, it is convenient to define a new notation for the dechirped baseband samples. Let
x.sub.v (FI,m,SI)=x.sub.v (FI,n) (4)
for n=SIΔ+m. Also, whenever an index is replaced by a �, the notation denotes the array of data elements corresponding to all values of the missing index. For example, xv (�,m,SI) is the F�1 vector of dechirped baseband samples for the mth pulse in the SIth subaperture. Similarly, xv (�,�,SI) is the F�M matrix of dechirped baseband samples for the SIth subaperture.
The differential slant range, cf with Equation (1), between the central reference point and evaluated at the transmission time of the nth pulse, can be written as
r.sub.sc =r.sub.o -(λ/2)[n-(N.sub.a /2)]f.sub.D +η.sub.n, (5)
where ro is the slant range and fD the normalized doppler frequency, both evaluated at the center of the synthetic aperture, and ηn is an expression that accounts for higher-order variations in rsc. Equations for the quantities ro and fD are given in the sections on the range process and coarse-azimuth process, respectively.
The function ηn varies slowly with n, so for n=m+SIΔ, it can be approximated by ηSIΔ, its value at the center of subaperture. By using (5), and making several other approximations, the expression for the dechirped baseband samples reduces to ##EQU2##
This equation describes the basic form of the processes of the invention. Each term in Equation (6) is operated on by a specific process of the invention. The complex exponential on the first line is the dechirped baseband signal that would result if the radar were stationary. In this case, the range ro could be resolved through a simple FFT operation. The expression on the second line is the doppler signal resulting from a linear change in rsc. The product on line three is just the central reference point-motion term from line 2 of Equation (2) and is further discussed with reference to Equations 8 and 14. Here it is factored into separate azimuth and range components.
For some combinations of resolution and scene size, only the first three lines of Equation (6) are required to describe the dechirped baseband signal. In such a case, processing reduces to three simple steps. First, the dechirped baseband data are multiplied by the conjugate of the expression on line three. This holds the slant range and doppler frequency of the signal from the central reference point at the origin in a range-doppler plane. Second, the data are windowed and transformed with respect to FI to resolve ro. And third, the data are windowed and transformed with respect to n (=m+SIΔ) to resolve fD. The order of the last two steps is unimportant. For fine resolution and large scene size, the terms on lines four and five of Equation (6) are not negligible. The expression on line four describes the motion through range cells. Only the motion that is linear in SI, known as range walk, is included. Because the expression does not depend on m, range walk during a subaperture is neglected. The expression on line five describes the motion through doppler cells. The function ξSI is a combination of ηSIΔ from Equation (5) and the expression from the last line of Equation (2). It depends on the coordinates of as well as the exact trajectory of the antenna phase center.
At the next stage of coarse-resolution azimuth processing, the subapertures are processed to obtain coarse resolution in doppler frequency and hence, in azimuth position. The columns of a data block are sorted into doppler frequency bins. In view of Equation (6), this corresponds to sorting by the amount of range walk. The dechirped baseband pulses out of the bandpass filter 68 read into the first corner-turning memory CTM0 76 in FIGS. 2 and 3. Each dechirped baseband pulse, e.g., each received vector 204 (FIG. 3) corresponds to an F�1 column vector xv (•,m,SI). In CTM0 76, sequences of M column vectors 208 are arranged into F�M matrices called subaperture arrays 206. The coarse-resolution azimuth process operates on one subaperture at a time. During the length of a synthetic aperture, a total of S subapertures are formed, as shown with respect to FIG. 4. The subapertures are overlapped as shown in FIG. 4.
After an F�M subaperture 206 is formed, the data are read out of CTM0 as a 1�M row vectors 208. For the SIth subaperture, the input to this process consists of a sequence of 1�M vectors 208, xv (FI,•,SI) for 0≦FI≦F-1, read from CTM0 76. The first operation on the vectors 208 read from CTM0 is multiplication by w(•) 82 and Fca (•,SI). The second vector, Fca (•,SI), is called the coarse-resolution azimuth focus vector and is a synthesized waveform 80 having a starting frequency fc, starting phase φc, and chirp rate γc. Let w(m) be the window function, 0≦m≦M-1, and let w(•) be the corresponding 1�M vector. The 1�M vector whose mth element is w(m) xv (FI,m,SI) is represented as the product w(•).*xv (FI,•,SI) wherein the symbol .* denotes element-by-element multiplication of two vectors. Also each row vector 208 is multiplied at 86 by a coarse-resolution azimuth focus vector that stabilizes the doppler frequency of a reference point at the center of the scene on the ground. The coarse-resolution azimuth focus vector is produced in multiplier 84 and the synthesized waveform 80 is produced in waveform synthesizer 78. This operation does not stabilize points that are far away from the reference point. Such points are stabilized at the fine-resolution step.
The radar may use a variable pulse repetition frequency PRF to transmit pulses at equal-distance increments dx along the X axis, so the position x in (2) is
x={(m-M/2)+[SI-(S-1)/2]Δ}d.sub.x, (7)
and the focus vector has components
F.sub.ca (m,SI)=exp[j(2π/λ)(x.sup.2 /r.sub.b)], (8)
where rb is evaluated at the center time of the SIth subaperture. Since Fca (•,SI) does not depend on FI, the same vector can be used for all the rows in a subaperture. Also, because of the equal x increments, the focus vector can be synthesized with a hardware chirp generator as the data are read from CTM0. Both these observations aid the implementation process in a real-time processor. In situations, however, when the process described herein is used off-line, consideration must be given to the facts that the factored central reference point-motion expression is an approximation. In certain situations, however, it is desirable to vary the PRF as a function of the squint angle to the central reference point. This option is further discussed later with respect to other embodiments of the invention.
Next, the vector 208
w(•).*F.sub.ca (•,SI).*x.sub.v (FI,•,SI) (9)
0≦F≦F-1 is transformed using a length-M Fast Fourier Transform (FFT) 90 to yield the vector. The components of the resulting 1�M vectors are indexed by frequency bin having width 1/M in normalized frequency units. The resulting vector 210 is ordered by coarse azimuth position into coarse-azimuth bins. Because the azimuth sample rate is chosen to oversample the doppler spectrum from the scene of interest, only A<M coarse-resolution azimuth bins are saved. This azimuth filtering operation reduces memory requirements of the hardware of the system. To determine A, it is necessary to estimate the maximum doppler bandwidth. For a scatterer at =(sa,sr,0), the normalized doppler frequency is
f.sub.D =[2d.sub.x /(λr.sub.b)]{s.sub.a [1-s.sub.r cos φ/r.sub.b)]}, (10)
where rb, φ, and the velocity components are evaluated at the center time of the aperture. For real-time operation, the maximum doppler frequency is estimated from Equation (10) based on the scene size. The bandwidth of the normalized doppler spectrum is no greater than A/M, where A is the smallest integer satisfying
A≧2M max|f.sub.D |. (11)
For some combinations of parameters, it is necessary to shift the spectrum to make it fit in A bins. This is accomplished by including a frequency shift when the input vectors are multiplied by w(•). Now AI is the frequency-bin index that replaces m after the FFT, and let W(f) be the discrete Fourier transform (DFT) of w(m). An output-vector from the coarse-resolution stage, i.e., the output row vector 208a, denoted by xc (AI,•,SI), has components ##EQU3## where 0≦AI≦AI-1 and a constant-phase term is omitted. The azimuth sample index in SI instead of N indicates that the signal in each coarse resolution azimuth bin is downsampled by a factor of Δ.
One of the approximations implicit in Equation (12) is that motion through coarse-resolution doppler cells is neglected. A more detailed analysis treats the fD in the argument of W [ ] as a function of SI. The analysis has shown that motion through half a coarse cell during an aperture has no important effect on the point-scatterer response function. Current radar and scene-size parameters restrict the maximum motion to this range, so the approximation is justified.
The columns of the matrix xc (•,•,SI) contain linear combinations of the dechirped baseband pulses collected during the SIth subaperture. Because of the window factor, W[(AI/M)-fD ], a dechirped baseband-signal component with doppler frequency fD will have maximum amplitude in the column with AI/M closest to FD. In this sense, the dechirped baseband returns are sorted by doppler frequency into the columns indexed by AI.
Because the bin width in frequency is 1/M, it follows from Equation (10) that the coarse-resolution sample spacing in azimuth, for sr =sz =0 is
δ.sub.ca =λr.sub.b /(2Md.sub.x). (13)
The amplitude factor AR in Equation (6) contains roll-off effects from the antenna gain and presummer. So at this point in the process, it is convenient to apply corrections at 92 for the azimuth antenna-pattern and the presummer frequency response. To accomplish this, each FFT-output vector is multiplied by a correction vector at 92 to produce a new amplitude factor AC, that is independent of azimuth position.
Then the result, the output row vector 210 is written into the second corner-turning memory CTM1 94 which forms an F�A matrix 212. For range processing, F�1 column vectors 214 are read from CTM1 94. The signal in a column is the superposition of returns from points in one coarse-resolution azimuth bin and all ranges in the scene. In a linear-FM radar system, the frequency of the signal component from each point scatterer is proportional to its range. Thus to compensate for changes in range, the spectrum of the signal in each column is shifted.
There are two components of the required frequency shift. The first frequency shift at 98 stabilizes the central reference point in range and is applied equally to each vector in a subaperture. The expressions on line two of Equation (12) describe how the dechirped baseband frequency of the signal from a point scatterer varies during an aperture. The first expression, Fr *(FI,SI), represents the term of the central reference point range that is quadratic in x. Its effect is removed by multiplying each input vector by Fr (•,SI), where
F.sub.r (FI,SI)=exp{j2πx.sup.2 γT.sub.s [FI-(F/2)]/(cr.sub.b)}. (14)
In Equation (14), x and rb are evaluated at the center of the SIth subaperture. After multiplication by Fr (•,SI), the dechirped baseband frequency from a scatterer at the central reference point is zero. If only the first frequency shift is performed, the signals from points far away from the reference point will still move through range-resolution cells.
A second frequency-shift at 96 is necessary to correct for the remaining motion through range, range walk. The second expression on line two of Equation (12) describes the dechirped baseband-frequency variation that corresponds to range walk which is a function of FD. For any column of xc (•,•,SI), the total range walk over an aperture is (S-1)Δλ|fD |/2, which can be a number of range cells. By design, the doppler frequency fD is in one of the frequency bins 0≦AI≦AI-1. It is possible to remove most of the range walk as follows. Define a motion through range or range-walk correction vector Frw (•,AI,SI) by
F.sub.rw (FI,AI,SI)=exp{-j2π[γT.sub.s λΔAI/(cM)][SI-(S-1)/2][FI-(F/2)]} (15)
where FI=0, 1, . . . , F-1, and SI=0, 1, . . . , S-1, and AI is as above. Then for each AI, multiply xc (•,AI,SI) by Frw (•,AI,SI). In column AI, the residual range walk over an aperture is
rw(AI)=(S-1)Δλ|f.sub.D -(AI/M)|/2. (16)
Let AIo be the bin where fD lies, that is, |fD -AI/M| is a minimum at AIo. The effect of the range-walk correction is to decrease the range walk for the signal in bin AIo and to increase it for the signals in the other bins. If fD is in the center of bin AIo, then fD =AIo /M, and the residual range walk is zero. In the worst case, fD is at the edge of the bin, and the residual range walk is (S-1)Δλ/(4M). To avoid amplitude loss and spreading in the point-scatterer response function, rw(AIo) should be held to a maximum of about one range-resolution cell. Let ρr be the range resolution cell size. By relating M to the coarse azimuth resolution ρca, and SΔ to the fine azimuth resolution ρa, this constraint is:
ρ.sub.r ≧(λ/4)(ρ.sub.ca /ρ.sub.a). (17)
Because ρr and ρa are system parameters, the inequality is best thought of as a constraint on ρca, which is the product of δca given by Equation (13) and the coarse-resolution window bandwidth.
Thus, the spectrum of the signal in each column is shifted by an amount that is proportional to the azimuth bin number AI of that column, i.e., an azimuth position of a center of an azimuth bin vector. This simple vector multiplication which performs range frequency translation is a key operation in the invention and is unique to this radar. The frequency shifts are added together to arrive at the value of fr.
A vector is then synthesized at 96 with frequency range translation fr and range translation starting phase φr. The range translation chirp rate, γr, of this synthesizer is set to zero.
After multiplication by the two-motion compensation vectors, the resulting columns are multiplied by a range window wr (FI), 0 FI=0, 1, . . . , F-1 at 98. The resultant data vector 214 is then multiplied at 100 by a F�1 column vector from CTM1 and Fourier transformed at 102 to resolve ro.
The input to the range FFT is
w.sub.r (•).*F.sub.r (•,SI).*F.sub.rw (•,AI,FI).*x.sub.c (•,AI,SI). (18)
If necessary, this vector can be zero filled to Ffft elements, where Ffft is a power of two. Then the sample spacing in range for the FFT-output vector is
γ.sub.r =c/(2γT.sub.s F.sub.fft). (19)
Assume that R output samples are saved; that these are indexed by RI where 0≦RI≦R-1; and that range bin RI corresponds to a slant range r=-(RI-R/2)δr. At this point in the process, the data vectors are available as functions of slant range. The slant range ro that the radar measures is approximately
r.sub.o ≃s.sub.r cos φ-s.sub.z sin φ+[(s.sub.r sin φ+s.sub.z cos φ).sup.2 +s.sub.a.sup.2 ]/(2r.sub.b) (20)
where φ and rb are evaluated at the center of the aperture. The last expression in Equation (20) accounts for wavefront curvature which is a significant factor for fine resolution and large scene size.
Let xr (•,AI,SI) denote the R�1 output vector from the range-processing stage. An element of this vector is
x.sub.r (RI,AI,SI)=A.sub.o R(r-r.sub.o) •W[(AI/M)-f.sub.D ]exp(j2πf.sub.D SIΔ) exp(-j2πξ.sub.SI), (21)
where R(r) is the DFT of wr (FI), and Ao is the amplitude factor corrected for range-dependent variations. Because of the factor R(r-ro), the columns of the vector xr (•,•,SI) have fine resolution in range. Because of W[(AI/M)-fD ], the rows have coarse resolution in azimuth. FIG. 8 is a diagram of the range-processing operations. The output R�1 column vectors have fine-resolution in range. Next, they are corrected at 104 for the elevation antenna gain pattern, range law path loss, and filter imperfections. The output vectors xr (•,AI,SI) are written into CTM2 106.
The coarse-resolution azimuth process 32 and range process 34 are repeated as data are collected for each of the S subapertures. The operations up to this point can be performed in parallel with data collection, another advantage for real-time processing. Other subaperture techniques compensate for motion through range cells by interpolating and resampling signals that have already been range compressed.
The output matrices from range processing are arranged into an RAS array 218. This three-dimensional array is the input to the azimuth fine-resolution processing 36. For each of the RAS-vectors in the three-dimensional array, the fine-resolution azimuth process 36 performs four operations: windowing at 112, fine-resolution focus 118, Discrete Fourier Transform (DFT) 120, and correction for the coarse-resolution window roll off 122. The initial windowing operation at 112 is just a standard multiplication by a window function selected to meet sidelobe requirements. The fine-resolution focus operation 118 is another key element of the invention. It is the operation that corrects for motion through azimuth-resolution cells. The fine-resolution focus vector is the product of three separate vectors.
Each S-vector, i.e., each subaperture direction vector 220 from the input array 218 is multiplied at 110 by a first component that is a function of both coarse-azimuth and range bin numbers which corrects for spatially varying phase errors. The first component of the fine-resolution focus vector changes most rapidly and is synthesized directly in azimuth correction waveform 108, using parameters azimuth starting frequency fa, azimuth phase φa, and azimuth chirp γa to save time.
The focus vector is also a function of dynamic motion information which is supplied from an external inertial measurement unit. This second component 116 of the fine-resolution focus vector is a slowly-varying part component dependant upon inertial measurement unit data and may be prepared in a computer.
The third component of the fine-resolution focus azimuth vector is necessary because the inertial measurement unit cannot measure the radar motion precisely enough to satisfy image quality requirements for high-resolution SAR. Thus, the invention incorporates a data-driven estimate of the unsensed phase error generated by an autofocus process 114. The autofocus process 114 requires input from the above-described fine-resolution azimuth processes; the invention has been designed to incorporate the autofocus process at this time to minimize the amount of processing required. A sharper image is obtained when the data is corrected for spatially variant phase errors prior to autofocus 114. Typically, autofocus algorithms presume that the spatially-variant phase errors are invariant which is not true in many actual cases. The process may be the phase-gradient autofocus process described in U.S. Pat. No. 4,924,229 herein incorporated by reference. Other known techniques for estimating the unsensed phase error may also be used, such as in U.S. Pat. No. 4,771,287 to Mims. Once the unsensed-phase errors are estimated by an autofocus process, they are removed by multiplying all of the radar data by an autofocus phase correction vector. The autofocus phase correction vector is multiplied by the fine-azimuth window vector, which is in turn multiplied by the product of the first and second components of the fine resolution focus vector.
After multiplication of each xr (RI,AI,•) vector by the window and fine-resolution focus vectors at 118, the resulting S-vector is Fourier transformed at 120. For each fixed range, a DFT is performed over a doppler frequency band of the vector from each coarse-azimuth bin which provides fine resolution in doppler frequency. The doppler frequency of a point scatterer is a function of both range and azimuth. In order to keep the pixel spacing in azimuth constant, it is necessary to vary the doppler frequency sampling as a function of range when the DFTs are performed. Therefore, the number of output samples, P, required from the DFT operation is approximately the ratio of the coarse-resolution sample spacing to the desired fine-resolution sample spacing. To implement the DFT, the chirp-z transform 120 is used. The fact that the process is able to remove the geometric distortion caused by range-doppler interaction is another of the key characteristics of the invention. SAR imaging maps the point =(sa,sr,0) in the XY plane into (ro,fD) in the range-doppler plane. Under this mapping, a rectangle in the XY plane is transformed into the shape shown in FIG. 65. Because of wavefront curvature, lines of constant sr map into curves 600, 602, 604 described by Equation (20). Also, the doppler bandwidth corresponding to a rectangular radar image is a function of slant range. This range-doppler interaction is indicated by the presence of sr in Equation (10) and accounts for the apparent greater bandwidth at near slant range shown in FIG. 5. It also explains why curves of constant sa ≠0 i.e., curves 606 and 608 are not vertical. Thus, by modifying the chirp-z transform parameters as a function of slant range, the data is mapped from slant range ro and doppler frequency fD grid onto a grid of ro and azimuth position. In this way the invention compensates for range-doppler interaction and produces an image with constant azimuth-pixel spacing. This is another advantage of the invention.
Another vector operation at 122 corrects for amplitude and phase effects of the coarse-resolution azimuth processing, see 32 of FIG. 2. The result is a part of one range line corresponding to one coarse-resolution bin; and the compilation of all sequential range lines yield the complete radar image 226 at 126.
One of the joys of the invention is the inherent flexibility. Just about any parameter of any process step of the invention can be varied. As an example, the variable-PRF scheme used by the radar is important for real-time implementation, because it allows the motion-compensation vectors Fca, Fr, Frw, and F1 to be produced in hardware. Note, however, that the invention does not require variable PRF. If the radar PRF is constant, it is necessary to determine the X-axis position by integrating Vx. Then the resulting x(t) is substituted into the equations for the motion-compensation vectors. When used this way, the invention is more sensitive to X-axis acceleration ax. For the constant-PRF case, nonzero ax produces a quadratic phase error that is proportional to sa. Small errors of this kind can be removed in the fine-resolution focus process. But if ax is too large, the amount of (coarse) azimuth resolution available before the last DFT is not adequate to remove all the phase error. So the variable-PRF embodiment of the invention is preferable.
The description of the invention given above specifies varying the radar center frequency and A/D sample rate, but it is possible to hold these two parameters constant. In this case, the processor can still form a high-quality image, but there are two disadvantages. First, the doppler-frequency band corresponding to a particular ground patch is increased, so more coarse bins must be retained. This increases the processor memory requirement. Second, some range-dependent phase errors remain in the dechirped baseband signal as indicated in Equation (6) that must be removed before fine-resolution processing. This can be accomplished by augmenting the fine-resolution focus vector at the expense of increased processing time.
Another modification relates to the second line of Equation (2) which stabilizes the dechirped baseband frequency associated with a central reference point. This stabilization can occur in two ways: as described by range processing 34 in FIG. 2; or by appropriate changes in the starting frequency and phase of the local oscillator. If the local oscillator is operated this way, then the vectors defined in Equations (8) and (14) are unnecessary. This modification simply involves some changes in programming of the specific processing steps and does not alter the structure of the radar.
FIG. 1 shows the radar operating in a broadside mode, i.e., the vector ua is parallel to the nominal motion path. The situation where ua is not parallel to the nominal motion path is termed squint mode. The modifications of the previous paragraph and the variable PRF scheme are extended to include a dependence on Vy. To operate the radar in the squint mode, the radar structure need not change. Also the radar center frequency and either the chirp rate or the A/D sample rate are modified as functions of the depression angle to the central reference point.
Although the invention has been described with respect to its preferred embodiment, for example a coarse-azimuth process using an FFT, it will be appreciated by persons skilled in the art that an FFT is only one mechanization of a filter bank. Therefore, alternative filter bank mechanizations are also disclosed. For example, special-purpose digital hardware can realize a bank of infinite-impulse-response filters.
Having described preferred embodiments of a novel imaging radar and method for compensating a radar (which are intended to be illustrative and not limiting), it is noted that modifications and variations can be made by persons skilled in the art in light of the above teachings. It is therefore to be understood that changes may be made in the particular embodiments of the invention disclosed which are within the scope and spirit of the invention as defined by the appended claims.
|Cited Patent
|Filing date
|Publication date
|Sep 5, 1973
|Nov 4, 1975
|Raytheon Co
|Time compression system adding noise to allow one bit quantization
|Mar 21, 1975
|Mar 8, 1977
|Cubic Industrial Corporation
|Single frequency radio ranging system
|Apr 6, 1976
|Jul 5, 1977
|Westinghouse Electric Corporation
|Second order motion compensator for high resolution radar
|Feb 7, 1975
|Aug 26, 1980
|Hughes Aircraft Company
|Synthetic array autofocus system
|Dec 15, 1978
|Sep 29, 1981
|Real-time multiple-look synthetic aperture radar processor for spacecraft applications
|Jun 15, 1971
|Jul 13, 1982
|Hughes Aircraft Company
|Synthetic array processor
|Oct 9, 1980
|Feb 1, 1983
|The United States Of America As Represented By The Administrator Of The National Aeronautics And Space Administration
|Servomechanism for doppler shift compensation in optical correlator for synthetic aperture radar
|May 12, 1981
|Jul 24, 1984
|Philippe Martin
|Radar signal processing process and circuit
|Oct 26, 1981
|Sep 11, 1984
|The United States Of America As Represented By The Administrator Of The National Aeronautics And Space Administration
|Pipelined digital SAR azimuth correlator using hybrid FFT/transversal filter
|Mar 18, 1982
|Apr 2, 1985
|The United States Of America As Represented By The Administrator Of The National Aeronautics And Space Administration
|Method and apparatus for ΔK synthetic aperture radar measurement of ocean current
|Jun 30, 1983
|Feb 16, 1988
|X-Cyte, Inc.
|System for interrogating a passive transponder carrying phase-encoded information
|Aug 19, 1985
|Jul 19, 1988
|Hitachi, Ltd.
|Method of reconstructing images from synthetic aperture radar's data
|Jun 23, 1983
|Sep 13, 1988
|Westinghouse Electric Corp.
|Method of correcting for errors in radar imaging
|Jan 12, 1987
|Apr 25, 1989
|Grumman Aerospace Corporation
|Simultaneous triple aperture radar
|Dec 30, 1987
|Jul 11, 1989
|Westinghouse Electric Corp.
|Coordinate system transformation apparatus for a high resolution radar
|Feb 1, 1988
|Jul 25, 1989
|The United States Of America As Represented By The Secretary Of The Navy
|Frequency agile synthetic aperture radar
|Dec 17, 1987
|Jan 9, 1990
|Caterpillar Inc.
|Velocity reference system
|Oct 31, 1990
|Jul 7, 1992
|General Electric Company
|Modulated range tone system
|Jul 15, 1991
|Jan 12, 1993
|Raney R K
|Synthetic aperture radar processor to handle large squint with high phase and geometric accuracy
|Jul 1, 1991
|Oct 5, 1993
|Duane Roth
|Method of correcting rotational motion error in SAR and ISAR imagery
|Sep 24, 1992
|Aug 30, 1994
|Hughes Aircraft Company
|Doppler tracking method for object imaging from radar returns
|Dec 31, 1992
|Jul 4, 1995
|Raytheon Company
|Synthetic aperture radar guidance system and method of operating same
|B. Arambepola, "New Architecture For a Real-Time SAR Processor", Proceedings of the IGARSS '88 Symposium, pp. 691-694, Sep. 1988.
|B. Arambepola, New Architecture For a Real Time SAR Processor , Proceedings of the IGARSS 88 Symposium, pp. 691 694, Sep. 1988.
|D. Ausherman et al., "Developments in Radar Imaging", IEEE Transactions On Aerospace and Electronic Systems, vol. AES-20, No. 4, Jul. 1984.
|D. Ausherman et al., Developments in Radar Imaging , IEEE Transactions On Aerospace and Electronic Systems, vol. AES 20, No. 4, Jul. 1984.
|J. Mims et al., "Synthetic Aperture Imaging With Maneuvers", IEEE Transactions on Aerospace and Electronic Systems, vol. AES-8, No. 4, pp. 192-200, Jul. 1972.
|J. Mims et al., Synthetic Aperture Imaging With Maneuvers , IEEE Transactions on Aerospace and Electronic Systems, vol. AES 8, No. 4, pp. 192 200, Jul. 1972.
|K. Liu et al., "Spacecraft On-Board SAR Image Generation for Eos-Type Missions", IEEE Transactions on Geoscience and Remote Sensing, vol. 27, No. 2, pp. 184-192, Mar. 1989.
|K. Liu et al., Spacecraft On Board SAR Image Generation for Eos Type Missions , IEEE Transactions on Geoscience and Remote Sensing, vol. 27, No. 2, pp. 184 192, Mar. 1989.
|K. Tomiyasu, "Tutorial Review of Synthetic-Aperture Radar (SAR) with Applications to Imaging of the Ocean Surface", Proceedings of the IEEE, vol. 66, No. 5, pp. 563-584, May 1978.
|K. Tomiyasu, Tutorial Review of Synthetic Aperture Radar (SAR) with Applications to Imaging of the Ocean Surface , Proceedings of the IEEE, vol. 66, No. 5, pp. 563 584, May 1978.
|W. Brown et al., "Range-Doppler Imaging with Motion Through Resolution Cells", IEEE Transactions on Aerospace and Electronic Systems, vol. AES-5, No. 1, pp. 98-102, Jan. 1969.
|W. Brown et al., Range Doppler Imaging with Motion Through Resolution Cells , IEEE Transactions on Aerospace and Electronic Systems, vol. AES 5, No. 1, pp. 98 102, Jan. 1969.
|Citing Patent
|Filing date
|Publication date
|Jun 24, 1996
|Jan 13, 1998
|Northrop Grumman Corporation
|Multi-mode radar system having real-time ultra high resolution synthetic aperture radar (SAR) capability
|Dec 23, 1996
|Sep 8, 1998
|Nec Corporation
|Synthetic aperture radar and target image production method
|Apr 30, 1998
|Jun 8, 1999
|Method for the processing of the reception signal of a synthetic aperture radar with frequency ramps
|Feb 6, 1998
|Jul 20, 1999
|Ems Technologies Canada, Ltd.
|Synthetic aperture radar
|Sep 20, 1996
|Oct 19, 1999
|Hellsten; Hans
|SAR radar system
|May 28, 1998
|Mar 14, 2000
|Multispec Corporation
|Method for automatic focusing of radar or sonar imaging systems using high-order measurements
|Mar 19, 1999
|Dec 19, 2000
|National Space Development Agency Of Japan
|Radio wave receiving apparatus having an improved antenna resolution
|Jul 29, 1999
|Dec 26, 2000
|Nec Corporation
|Image synthesizing method using a plurality of reflection radar waves and aircraft image radar apparatus using the method
|Feb 22, 2000
|Jan 30, 2001
|Veridian Erim International, Inc.
|Reference-based autofocusing method for IFSAR and other applications
|Sep 20, 1999
|Mar 20, 2001
|National Space Development Agency
|Radio wave receiving apparatus having an improved antenna resolution
|Sep 2, 1999
|Jan 29, 2002
|Telefonaktiebolaget Lm Ericsson (Publ)
|Method and arrangement for duty cycle enlargement
|Jun 13, 2001
|Dec 10, 2002
|Raytheon Company
|System and method for processing squint mapped synthetic aperture radar data
|Oct 15, 2001
|Jun 10, 2003
|Sandia Corporation
|Transponder data processing methods and systems
|May 30, 2003
|Mar 22, 2005
|The Boeing Company
|System and method for increasing resistance to target vibrations in synthetic aperture radar signals
|Dec 22, 2005
|Jan 23, 2007
|Sensors & Software Inc.
|Data acquisition for a ground penetrating radar system
|Dec 10, 2005
|Jan 30, 2007
|Landray Technology, Inc.
|Linear FM radar
|Oct 15, 2003
|Aug 21, 2007
|Sandia Corporation
|Quadrature mixture LO suppression via DSW DAC noise dither
|Jan 17, 2006
|Dec 11, 2007
|Raytheon Company
|Non-statistical method for compressing and decompressing complex SAR data
|Oct 5, 2005
|Jun 24, 2008
|Sandia Corporation
|Correction of motion measurement errors beyond the range resolution of a synthetic aperture radar
|Jan 29, 2007
|Jun 24, 2008
|Land Ray Technology, Inc.
|Linear FM radar
|Jun 5, 2006
|Jul 8, 2008
|Sandia Corporation
|SAR image formation with azimuth interpolation after azimuth transform
|Mar 10, 2005
|Aug 12, 2008
|Itt Manufacturing Enterprises Inc.
|Synthetic aperture radar (SAR) data compression
|Feb 13, 2007
|Mar 3, 2009
|Sandia Corporation
|Synthetic aperture design for increased SAR image rate
|Mar 29, 2006
|Sep 8, 2009
|Saab Ab
|Method and system for mapping a target scene using scanning radar
|Jul 30, 2008
|Jan 12, 2010
|The United States Of America As Represented By The Secretary Of The Navy
|Synthetic aperture radar motion estimation method
|Aug 15, 2007
|Feb 16, 2010
|General Dynamics Advanced Information Systems, Inc.
|Methods for two-dimensional autofocus in high resolution radar systems
|Nov 12, 2003
|Mar 23, 2010
|Fuji Xerox Co., Ltd.
|Image processing apparatus, image processing method, image processing program, printed matter inspection apparatus, printed matter inspection method and printed matter inspection program
|Sep 28, 2007
|May 11, 2010
|Raytheon Company
|Non-statistical method for compressing and decompressing complex SAR data
|May 14, 2008
|Jul 20, 2010
|Sandia Corporation
|Decreasing range resolution of a SAR image to permit correction of motion measurement errors beyond the SAR range resolution
|May 14, 2008
|Aug 17, 2010
|Sandia Corporation
|Comparing range data across the slow-time dimension to correct motion measurement errors beyond the range resolution of a synthetic aperture radar
|Nov 17, 2008
|Jan 18, 2011
|Adc Dsl Systems, Inc.
|System for multi-frame alignment
|Oct 29, 2007
|Feb 1, 2011
|Adc Telecommunications, Inc.
|Systems and method for orthogonal frequency divisional multiplexing
|Oct 29, 2007
|Feb 1, 2011
|Adc Telecommunications, Inc.
|Systems and method for orthogonal frequency divisional multiplexing
|Feb 26, 2007
|Mar 22, 2011
|Adc Telecommunications, Inc.
|Timing and symbol alignment in multipoint-to-point communication using orthogonal frequency division multiplexing
|Sep 17, 2008
|Apr 5, 2011
|The United States Of America As Represented By The Secretary Of The Army
|Systems for measuring backscattered light using rotating mirror
|Oct 16, 2008
|Apr 19, 2011
|The United States Of America As Represented By The Secretary Of The Army
|Systems and methods for producing radar images
|Oct 30, 2007
|May 3, 2011
|Adc Telecommunications, Inc.
|Ranging and round trip delay timing adjustment in a multi-point to point bidirectional communication system
|Oct 29, 2007
|Jun 7, 2011
|Adc Telecommunications, Inc.
|Systems and method for orthogonal frequency divisional multiplexing
|Feb 26, 2007
|Jul 19, 2011
|Geile Michael J
|Synchronized multipoint-to-point communication using orthogonal frequency division
|Oct 29, 2007
|Aug 9, 2011
|Htc Corporation
|Systems and method for orthogonal frequency divisional multiplexing
|Aug 27, 2009
|Nov 22, 2011
|Honeywell International Inc.
|Method for cross-range enhancement of real-beam radar imagery
|Oct 29, 2007
|Jan 3, 2012
|Htc Corporation
|Systems and method for orthogonal frequency divisional multiplexing
|Oct 29, 2007
|May 8, 2012
|Htc Corporation
|Systems and method for orthogonal frequency divisional multiplexing
|Oct 29, 2007
|Jun 12, 2012
|Htc Corporation
|Systems and method for orthogonal frequency divisional multiplexing
|Aug 26, 2009
|Jul 3, 2012
|Htc Corporation
|Method for multiple use subchannels
|Sep 28, 2009
|Jul 3, 2012
|Htc Corporation
|System for multiple use subchannels
|Jun 10, 2011
|Nov 20, 2012
|Htc Corporation
|Synchronized multipoint-to-point communication using orthogonal frequency division
|Oct 29, 2007
|Jan 8, 2013
|Htc Corporation
|Systems and method for orthogonal frequency divisional multiplexing
|Jun 10, 2011
|Mar 26, 2013
|Htc Corporation
|Systems and methods for orthogonal frequency division multiplexing
|Apr 11, 2012
|Oct 1, 2013
|Htc Corporation
|Systems and methods for orthogonal frequency divisional multiplexing
|Jun 2, 2009
|Oct 29, 2013
|Astrium Gmbh
|Method for optimizing the operation of an active lateral-view sensor when the height above the surface to be detected is variable
|Dec 5, 2011
|Nov 5, 2013
|Htc Corporation
|Systems and method for orthogonal frequency division multiplexing
|Dec 6, 2012
|Jan 28, 2014
|Htc Corporation
|Systems and method for orthogonal frequency divisional multiplexing
|Dec 15, 2011
|Nov 4, 2014
|Industrial Technology Research Institute
|Image stabilization method and image stabilization device
|Oct 7, 2008
|Aug 18, 2015
|The Boeing Company
|Correction of spatially variant phase error for synthetic aperture radar
|May 23, 2012
|Mar 29, 2016
|Garmin International, Inc.
|Electronically steered weather radar
|Jul 19, 2011
|Jun 21, 2016
|The Wilfred J. and Louisette G. Lagassey Irrevocable Trust
|Modular intelligent transportation system
|Jul 12, 2012
|Sep 13, 2016
|Mitsubishi Electric Corporation
|Radar system and data processing device
|Nov 12, 2003
|Jul 29, 2004
|Fuji Xerox Co., Ltd.
|Image processing apparatus, image processing method, image processing program, printed matter inspection apparatus, printed matter inspection method and printed matter inspection program
|May 30, 2003
|Dec 2, 2004
|The Boeing Company
|System and method for increasing resistance to target vibrations in synthetic aperture radar signals
|Dec 22, 2005
|Dec 14, 2006
|Sensors & Software Inc.
|Data acquisition for a ground penetrating radar system
|Feb 9, 2007
|May 17, 2007
|Adc Telecommunications, Inc.
|Contention-based access to tones in orthogonal frequency division multiplexing (ofdm) waveform
|Jan 29, 2007
|Jun 14, 2007
|Beckner Frederick L
|Linear FM radar
|Jan 17, 2006
|Jul 19, 2007
|Raytheon Company
|Non-statistical method for compressing and decompressing complex SAR data
|Aug 15, 2007
|Feb 21, 2008
|General Dynamics Advanced Information Systems, Inc.
|Methods for two-dimensional autofocus in high resolution radar systems
|Oct 29, 2007
|Feb 28, 2008
|Adc Telecommunications, Inc.
|Systems and method for orthogonal frequency divisional multiplexing
|Oct 29, 2007
|Mar 13, 2008
|Adc Telecommunications, Inc.
|Systems and method for orthogonal frequency divisional multiplexing
|Oct 29, 2007
|Oct 2, 2008
|Adc Telecommunications, Inc.
|Systems and method for orthogonal frequency divisional multiplexing
|Dec 16, 2005
|Dec 11, 2008
|Thomas Brosche
|Radar System For Monitoring Targets in Different Distance Ranges
|Nov 17, 2008
|Mar 19, 2009
|Adc Telecommunications, Inc.
|System for multi-frame alignment
|Mar 29, 2006
|Sep 17, 2009
|Saab Ab
|Method and system for mapping a target scene using scanning radar
|Jun 2, 2009
|Dec 3, 2009
|Astrium Gmbh
|Method for Optimizing the Operation of an Active Lateral-View Sensor When the Height Above the Surface to be Detected is Variable
|Sep 9, 2008
|Mar 11, 2010
|Thomas Patrick Yunck
|Cellular Interferometer for Continuous Earth Remote Observation (CICERO)
|Sep 17, 2008
|Mar 18, 2010
|Videen Gorden W
|Systems for measuring backscattered light using finite speed of light
|Oct 7, 2008
|Apr 8, 2010
|Cho Kwang M
|Correction of spatially variant phase error for synthetic aperture radar
|Oct 16, 2008
|Apr 22, 2010
|Huebschman Benjamin D
|Systems and methods for producing radar images
|Aug 27, 2009
|Mar 3, 2011
|Honeywell International Inc.
|method for cross-range enhancement of real-beam radar imagery
|Jul 12, 2012
|Feb 26, 2015
|Mitsubishi Electric Corporation
|Radar system and data processing device
|Jan 7, 2011
|Sep 18, 2012
|Htc Corporation
|System for multiple use subchannels
|Jan 14, 2011
|Aug 27, 2013
|Htc Corporation
|Systems for synchronous multipoint-to-point orthogonal frequency division multiplexing communication
|Apr 20, 2006
|Oct 13, 2010
|Synthetic aperture radar impulse phase shift method
|Aug 2, 2012
|Feb 12, 2014
|Imaging method of synthetic aperture radar in large squint angle mode
|Jan 26, 2014
|Jul 30, 2014
|Multi-sub-band reception synthetic method for SAR echo signals
|Jan 26, 2014
|Feb 1, 2017
|Oct 16, 2014
|Dec 24, 2014
|Dechirp receiving method of radar echo signal
|Oct 16, 2014
|Nov 16, 2016
|May 5, 1998
|Nov 11, 1998
|Method for processing the received signal of a SAR radar with frequency ramps
|Mar 18, 1999
|Sep 29, 1999
|National Space Development Agency Of Japan
|Radio wave receiving apparatus having an improved antenna resolution
|Mar 18, 1999
|Feb 2, 2000
|National Space Development Agency Of Japan
|Radio wave receiving apparatus having an improved antenna resolution
|Sep 16, 1999
|Mar 29, 2000
|National Space Development Agency Of Japan
|Radio wave receiving apparatus having an improved antenna resolution
|Sep 16, 1999
|Mar 29, 2000
|National Space Development Agency Of Japan
|Radio wave receiving apparatus having an improved antenna resolution
|Aug 1, 2013
|Oct 16, 2013
|Institute of Electronics, Chinese Academy of Sciences
|Imaging method for synthetic aperture radar in high squint mode
|Feb 23, 2000
|Sep 8, 2000
|Veridian Erim International, Inc.
|Reference-based autofocusing method for ifsar and other applications
|Feb 23, 2000
|Mar 1, 2001
|Veridian Erim International In
|Reference-based autofocusing method for ifsar and other applications
|Dec 7, 2006
|Jul 5, 2007
|Landray Technology, Inc.
|Linear fm radar
|Dec 7, 2006
|Nov 27, 2008
|Landray Technology Inc
|Linear fm radar
|Jan 8, 2007
|Jul 17, 2008
|Rafael - Armament Development Authority Ltd.
|Method and system for synthetic aperture radar mapping and navigation using digital pulse compression
|Jan 8, 2007
|Apr 16, 2009
|Rafael Armament Dev Authority
|Method and system for synthetic aperture radar mapping and navigation using digital pulse compression
|U.S. Classification
|International Classification
|Cooperative Classification
|European Classification
|May 27, 1994
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About This Activity
The Dragon's fury starts in Messina, an altitude of 1410m (4625 ft) travels up to Dochula at 3150m (10,330 ft) then it is all downhill till the Thimphu city square finish.
Distance: 60Km. Join us on September 3rd, 2016.
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Get the latest inside info on the development of Irrational's Freedom Force vs. the Third Reich with a new designer diary on IGN:
For this article I'm going to mention a few of the technical graphics enhancements we've added and how these improve the look of the game.
Some of the new features include among others; projected character shadows, improved collision objects for physics, vertex colors on terrains, dynamic lighting and improved building shadows. Other enhancements were simply methods we have used to improve the run-time performance of frame rate and memory usage. Many of these features have been added only in the last couple of months and have produced some significant improvements.


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