CROSS-REFERENCE TO RELATED APPLICATION 
   This application claims priority to and the benefit of Korean Patent Application No. 10-2006-0020351 filed in the Korean Intellectual Property Office on Mar. 3, 2006, the entire contents of which are incorporated herein by reference. 
   BACKGROUND OF THE INVENTION 
   (a) Field of the Invention 
   The present invention relates to a switching mode power supply (SMPS) for supplying power by switching of a switch. 
   (b) Description of the Related Art 
   An SMPS is a device for converting a DC voltage into a square wave voltage by using a semiconductor device (such as a power MOSFET) as a switch, and then acquiring a DC voltage controlled through a filter. An SMPS is more efficient and durable compared to a conventional linear power supply since it controls power by using a semiconductor device as a switching regulator. In addition, an SMPS is a stabilization power device, having small size and less weight as advantages. An SMPS may be applied to industrial fields, including communication devices, computers, OA devices, and home appliances. In general, an SMPS controls the current (the drain current of a MOSFET) flowing to the main switch of the primary coil and maintains the output voltage of the secondary coil through a turn on/off operation of the main switch of the primary coil according to the current output to the load from the SMPS. In this instance, the SMPS feeds the output voltage back to a capacitor of the primary coil, and controls the duty cycle of the main switch of the primary coil through the feedback voltage charged in the capacitor of the primary coil to thereby maintain the output voltage. That is, the SMPS reduces the feedback voltage charged in the capacitor when the current output to the load is reduced to increase the output voltage, and the SMPS increases the feedback voltage charged in the capacitor when the current output to the load is increased to reduce the output voltage. 
     FIG. 1  is a diagram illustrating a variation of the sense voltage (Vsense) representing the drain current (Ids) flowing through the main switch of the primary coil when an overload condition occurs in a conventional SMPS. The sense voltage (Vsense) is generated when the drain current (Ids) of the main switch of the primary coil flows through a resistor (Rsense). As shown in  FIG. 1 , the current flowing through the main switch of the primary coil (as represented by sense voltage (Vsense)) increases when the main switch of the primary coil is turned on because of periodic signals from an oscillator. The main switch of the primary coil is turned off when the sense voltage (Vsense) of the main switch of the primary coil reaches the control voltage (Vc). 
   The current flows to the load through a diode of the secondary coil when the main switch of the primary coil is turned off. In this instance, the control voltage (Vc) is used to control the level of the sense voltage (Vsense) of the main switch of the primary coil, and it functions to maintain the output voltage (Vo). That is, when the current (Io) output to the load is increased to reduce the output voltage (Vo), the feedback voltage (V FB ) charged in the capacitor (C FB ) of the primary coil is increased, the control voltage (Vc) is gradually increased within the threshold voltage range of Vc sat  voltage level in correspondence to the feedback voltage (V FB ) to increase the current (Ids) flowing through the main switch. 
   Therefore, as shown in  FIG. 1 , the level of the control voltage (Vc) is maintained when the current (Io) applied to the load is constant from the time 0 to T 1  on the time axis. In this instance, when the load is increased at the time T 1 , the current (Io) increases and the output voltage (Vo) of the secondary coil is reduced, thereby causing an overload condition. The control voltage (Vc) level is increased, and the sense voltage (Vsense) is gradually increased until the time T 2 . In this instance, the slope of the drain current (Ids) of the main switch of the primary coil and the slope of the current (Id) flowing through the diode of the secondary coil are in proportion to the voltage as expressed in Equation 1. 
   
     
       
         
           
             
               
                 
                   
                     ⅆ 
                     i 
                   
                   
                     ⅆ 
                     t 
                   
                 
                 = 
                 
                   V 
                   L 
                 
               
             
             
               
                 ( 
                 
                   Equation 
                   ⁢ 
                   
                       
                   
                   ⁢ 
                   1 
                 
                 ) 
               
             
           
         
       
     
   
   Since the input voltage of the primary coil is generally constant, the rising slope of the drain current (Ids) of the main switch of the primary coil is maintained by Equation 1 when the main switch of the primary coil is turned on. When the current (Io) applied to the load is increased, the output voltage (Vo) of the secondary coil is reduced, and the slope of the drain current (Ids) of the main switch of the primary coil is gradually decreased when the main switch of the primary coil is turned off. 
   Therefore, the main switch of the primary coil is repeatedly turned on/off because of the signal periodically output by the oscillator. As shown in  FIG. 1 , the rising slope of the sense voltage (Vsense) becomes greater than the falling slope, and the sense voltage (Vsense) is increased corresponding to the control voltage (Vc) after the time T 1  each time the main switch of the primary coil is turned on. However, as further shown in  FIG. 1 , the control voltage (Vc) has the Vc sat  voltage as a threshold voltage. Therefore, after the time T 2 , the main switch of the primary coil is turned off when the voltage (Vsense) is increased to the Vc sat  voltage in the case in which an overload condition is generated and the feedback voltage (V FB ) is increased. 
   The main switch of the primary coil is shut down when the feedback voltage (V FB ) is increased to the protection voltage (Vp). The protection voltage (Vp) is a reference voltage for protecting the circuit when an overload is generated. As shown in  FIG. 1 , when the feedback voltage (V FB ) is increased to the protection voltage (Vp) at time T 3 , the main switch of the primary coil is shut down. In this instance, the period from the time T 2  (in which the sense voltage (Vsense) is increased to the Vc sat  voltage) to the time T 3  (in which the main switch of the primary coil is shut down) may be considered a delay time (T delay ), which is expressed in Equation 2. 
   
     
       
         
           
             
               
                 
                   T 
                   delay 
                 
                 = 
                 
                   
                     
                       C 
                       FB 
                     
                     × 
                     
                       ( 
                       
                         
                           V 
                           P 
                         
                         - 
                         
                           V 
                           c 
                           sat 
                         
                       
                       ) 
                     
                   
                   
                     I 
                     delay 
                   
                 
               
             
             
               
                 ( 
                 
                   Equation 
                   ⁢ 
                   
                       
                   
                   ⁢ 
                   2 
                 
                 ) 
               
             
           
         
       
     
   
   Here, I delay  is the current flowing to the capacitor (C FB ) during the delay time (T delay ). Accordingly, when an overload condition is generated in the conventional SMPS, the main switch of the primary coil is shut down when the delay time (T delay ) is passed, and hence, the circuit is protected by terminating the operation of the main switch. 
     FIG. 2  is a diagram illustrating a variation of the sense voltage (Vsense) for representing the current (Ids) flowing to the main switch of the primary coil when an output short is generated for a conventional SMPS. As shown in  FIG. 2 , at the time T 1 , the load is increased, which substantially increases the current (Io) applied to the load. The output voltage (Vo) of the secondary coil becomes 0V to thus generate the output short condition. The falling slope of the sense voltage (Vsense) becomes 0 according to Equation 1 when the main switch of the primary coil is turned off. Therefore, the sense voltage (Vsense) for the drain current (Ids) of the main switch of the primary coil is continuously increased each time the main switch of the primary coil is turned on. 
     FIG. 3  is a diagram illustrating an extended drawing of the sense voltage (Vsense) when the main switch of the primary coil is turned on for a conventional SMPS. Although not illustrated in  FIG. 1  and  FIG. 2 , in the sense voltage (Vsense) waveform, a leading edge overshoot is generated to instantly generate a peak voltage (PC) when the main switch of the primary coil is turned on. Therefore, during the time Tmin in which the peak voltage (PC) is generated, a leading edge blanking (LEB) process is performed so that an excess voltage is not sensed when the sense voltage (Vsense) exceeds the control voltage (Vc) and the main switch of the primary coil is not turned off. Hence, as shown in  FIG. 2 , when the sense voltage (Vsense) is greater than the Vc sat  voltage, the main switch of the primary coil is turned on during the time Tmin for performing the LEB process and the sense voltage (Vsense) is further increased. Therefore, as to the output short condition, the sense voltage (Vsense) is not decreased when the main switch of the primary coil is turned off, differing from the overload condition. 
   As a result, when the sense voltage (Vsense) becomes greater than the Vc sat  voltage, the sense voltage (Vsense) is consecutively increased during the time Tmin when the main switch is turned on. However, in the prior art, in a manner similar to the overload condition, the main switch of the primary coil is shut down after the output short is generated and the delay time (T delay ) is passed, but the circuit elements may be damaged because a large overload is applied during the delay time (T delay ). 
   The above information disclosed in this Background section is only for enhancement of understanding of the invention and is not an admission that such information is in the prior art that is already known in this country to a person of ordinary skill in the art. 
   SUMMARY OF THE INVENTION 
   In one aspect of the present invention, a switching mode power supply includes a power supply circuit including a main switch coupled to a primary coil of a transformer, and supplying power to a secondary coil of the transformer according to an operation of the main switch. A feedback circuit generates a feedback voltage corresponding to an output voltage at the secondary coil of the transformer. A switching controller controls the turn on/off operation of the main switch, the switching controller comparing a first voltage corresponding to a current flowing through the main switch and a first reference voltage corresponding to the feedback voltage. The switching controller turns off the main switch when the feedback voltage is greater than a second reference voltage and the duty ratio of the main switch is less than a reference value 
   In another aspect of the present invention, a switching mode power supply includes a transformer having a primary coil and a secondary coil. A main switch coupled to the primary coil of the transformer is turned on in synchronization with clock signals alternately having a high level and a low level. A feedback circuit generates a feedback voltage corresponding to an output voltage applied to the secondary coil of the transformer. A first logic operator controls the turn off operation of the main switch, the first logic operator having an output terminal coupled to a gate of the main switch. A second logic operator outputs a first signal to the first logic operator, the second logic operator having a first input terminal for receiving the clock signals. A first comparator outputs a second signal, the first comparator having a non-inverting terminal and an inverting terminal for receiving the feedback voltage and the first reference voltage. A third logic operator has input terminals for receiving the second signal, a third signal generated by delaying the clock signal by a first period, and the first signal, and an output terminal coupled to the first logic operator. 


   
     BRIEF DESCRIPTION OF THE DRAWINGS 
       FIG. 1  is a diagram illustrating a variation of a sense voltage for a drain current flowing through the main switch of the primary coil when an overload is generated for a conventional SMPS. 
       FIG. 2  is a diagram illustrating a variation of a sense voltage for a drain current flowing through the main switch of the primary coil when an output short phenomenon is generated for a conventional SMPS. 
       FIG. 3  is a diagram illustrating an extended drawing of the sense voltage when the main switch of the primary coil is turned on for a conventional SMPS. 
       FIG. 4  is a schematic circuit diagram of an exemplary SMPS according to an embodiment of the present invention. 
       FIG. 5  is a diagram illustrating a method for sensing an output short condition, according to an embodiment of the present invention. 
   


   DETAILED DESCRIPTION OF THE EMBODIMENTS 
   An exemplary embodiment of the present invention will hereinafter be described in detail with reference to the accompanying drawings. In the following detailed description, only certain exemplary embodiments of the present invention have been shown and described, simply by way of illustration. As those skilled in the art would realize, the described embodiments may be modified in various different ways, all without departing from the spirit or scope of the present invention. Accordingly, the drawings and description are to be regarded as illustrative in nature and not restrictive. The same parts over the specification have the same reference numerals. An SMPS according to an embodiment of the present invention will now be described in detail with reference to drawings. 
     FIG. 4  is a schematic diagram of an exemplary SMPS, according to an embodiment of the present invention. As shown in  FIG. 4 , the SMPS includes a power supply circuit  100 , an output circuit  200 , a feedback circuit  300 , a switching controller  400 , and a protection circuit  500 . The power supply circuit  100  includes a full wave bridge rectifier (BD)  102  for rectifying AC input signals, a capacitor (Cin)  104  for smoothing the rectified voltage, a primary coil L 1   106  of a transformer coupled to an input voltage (Vin), a switching transistor (M)  108  for functioning as a main switch coupled to the primary coil L 1  of the transformer, and a sense resistor (Rsense)  110  coupled between a source of the switching transistor  108  and a ground and sensing the current flowing through the switching transistor  108 . In this instance, the main switch of the exemplary SMPS can be implemented as a MOS field effect transistor (MOSFET). 
   The power supply circuit  100  of the exemplary SMPS receives an input voltage (Vin) and supplies power to the secondary coil  202  of the transformer according to the duty cycle of the switching transistor  108  to output a predetermined output voltage (Vo) to the output circuit  200 . In this instance, the output voltage (Vo) is fed back, and the SMPS uses the feedback value to control the duty cycle of the switching transistor  108  of the supply  100 , thereby regulating the output voltage (Vo). 
   The output circuit  200  of the exemplary SMPS may include a diode D 1   204  having an anode coupled to the secondary coil L 2   202  of the transformer, a capacitor C 1   206  coupled between a cathode of the diode D 1   204  and a ground, a resistor R 1   208  coupled to the cathode of the diode D 1   204 , a photodiode (PD)  210  coupled between the resistor R 1   208  and the ground, and a Zener diode (ZD)  212  coupled between the photodiode (PD)  210  and the ground. The output circuit  200  outputs an output current (Io) to the load, and provides information corresponding to an output voltage (Vo) to the feedback circuit  300  to regulate the output voltage (Vo). The photodiode (PD)  210  implements a photocoupler together with a photo transistor (PT)  302  of the feedback circuit  300 , and provides information corresponding to the output voltage (Vo) to the feedback circuit  300 . 
   The feedback circuit  300  of the exemplary SMPS includes the photo transistor (PT)  302  implementing a photocoupler together with the photodiode (PD)  210  of the output circuit  200 , and a capacitor (C FB )  304  coupled in parallel to the photo transistor (PT)  302 . The photo transistor (PT)  302  works to control the current corresponding to the output voltage (Vo) of the output circuit  200 , and the photo transistor (PT)  302  can be equivalently given as a dependent current source. That is, the photo transistor (PT)  302  of the feedback circuit  300  controls the flow of the current corresponding to the output voltage (Vo) of the output circuit  200  so that a relatively large amount of current may flow to the photo transistor (PT)  302  to further reduce the feedback voltage (V FB ) charged in the capacitor (C FB )  304  when the output voltage (Vo) is high, and a relatively small amount of current may flow to the photo transistor (PT)  302  to increase the feedback voltage (V FB ) charged in the capacitor (C FB )  304  when the output voltage (Vo) is low. 
   Accordingly, the information corresponding to the output voltage (Vo) is sensed by the feedback circuit  300  and is then input to the switching controller  400 , and is used to control the duty cycle of the switching transistor (M)  108 . 
   As shown, the switching controller  400  of the exemplary SMPS includes a first current source (I delay )  402 , a second current source (I FB )  404 , a first comparator  410 , a second comparator  420 , an latch  430  (e.g., a set-reset (SR) flip-flop), a NOR gate  440 , a gate driver  450 , an LEB  460 , and an OR gate  470 . The capacitor (C FB )  304  is coupled to the first current source (I delay )  402  for generating a current and the second current source (I FB )  404  for generating a relatively greater current compared to that of the first current source (I delay )  402 . The diode D 2   406  has a cathode coupled to the first current source (I delay )  402  and an anode coupled to the second current source (I FB )  404  to prevent the current of the first current source (I delay )  402  from flowing to the second current source (I FB )  404 . The diode D 3   408  has an anode coupled to the second current source (I FB )  404  and a cathode coupled to an inverting terminal (−) of the second comparator  420 , and a resistor R 2   412  is coupled between a contact point of the diode D 3   408  and the second comparator  420  and the ground. 
   When the switching transistor (M)  108  is turned on, the capacitor (C FB )  304  is charged by the first current source (I delay )  402  and the second current source (I FB )  404 . The control voltage (Vc) corresponding to the value generated by multiplying the current value of the second current source (I FB )  404  and the value of the resistor R 2   412  is applied to the inverting terminal (−) of the second comparator  420 . In this instance, when a large output current (Io) is applied to the load, the second current source (I FB )  404  outputs more current so as to compensate for the application of large current, and the control voltage (Vc) is increased within the range of the threshold voltage level, Vc sat . 
   Therefore, when the voltage (V FB ) charged in the capacitor (C FB )  304  is greater than the Vc sat  voltage level, the current of the second current source (I FB )  404  cannot be applied to the capacitor (C FB )  304  because of the diode D 2   406 , and the current of the first current source (I delay )  402  flows to the capacitor (C FB )  304 . 
   The first comparator  410  has a non-inverting terminal (+) and an inverting terminal (−). The non-inverting terminal (+) is coupled to the capacitor (C FB )  304  for charging the feedback voltage (V FB ), and the inverting terminal (−) receives the protection voltage (Vp). When the feedback voltage (V FB ) has a voltage level greater than the voltage (Vp) according to the comparison result of the first comparator  410 , a high level signal is output, and the switching transistor (M)  108  is shut down. The second comparator  420  has a non-inverting terminal (+) and an inverting terminal (−). The inverting terminal (−) receives a control voltage (Vc) and the non-inverting terminal (+) receives a sense voltage (Vsense) corresponding to the drain voltage of the switching transistor (M)  108  of the power supply circuit  100 . 
   In this embodiment, the control voltage (Vc) for controlling the sense voltage (Vsense) is increased when the current of the second current source (I FB )  404  is increased. That is, as described above, when the current (Io) output to the load is increased to reduce the output voltage (Vo), the control voltage (Vc) level is increased to increase the current flowing through the switching transistor (M)  108 . Therefore, when the sense voltage (Vsense) has a voltage level greater than that of the control voltage (Vc) according to the comparison result of the second comparator  420 , a high level signal is output and the switching transistor (M)  108  is turned off. In this instance, it is possible to control the sense voltage (Vsense) level by adding an offset power source (Voff) between the non-inverting terminal (+) of the second comparator  420  and the sense resistor (Rsense). A set terminal (S) of the SR flipflop  430  is coupled to an oscillator (OSC), the reset terminal (R) of the SR flipflop  430  is coupled to the second comparator  420 , and an inverted output end (/Q) is coupled to input terminals of a NOR gate  440  and an AND gate  520 . The SR flipflop  430  outputs an inverting output signal (/Q) to the NOR gate  440  and the AND gate  520  according to the logic of signals input to the set terminal (S) and the reset terminal (R). Table 1 shows a logical operation result according to the inputs of the SR flipflop. 
   
     
       
             
             
             
             
             
           
             
             
             
             
             
           
             
             
             
             
             
           
         
             
                 
               TABLE 1 
             
           
           
             
                 
                 
             
             
                 
               INPUTS 
               CURRENT STATES 
               NEXT STATES 
                 
             
           
        
         
             
               S 
               R 
               Q(t) 
               Q(t + 1) 
               /Q(t + 1) 
             
             
                 
             
             
               0 
               0 
               0 
               0 
               1 
             
             
               0 
               0 
               1 
               1 
               0 
             
             
               0 
               1 
               0 
               0 
               1 
             
             
               0 
               1 
               1 
               0 
               1 
             
             
               1 
               0 
               0 
               1 
               0 
             
             
               1 
               0 
               1 
               1 
               0 
             
           
        
         
             
               1 
               1 
               0 
               X 
                 
             
             
               1 
               1 
               1 
               X 
             
             
                 
             
           
        
       
     
   
   When the current state is given as the Q(T) in Table 1, the next state is determined depending on the signal input to the reset terminal (R) and the set terminal (S), and the signal level corresponding to a logic value of the /Q(T+1) is output from the output end (/Q). 
   The above-noted process is not allowed when high level signals are input to both the reset terminal (R) and the set terminal (S). The oscillator (OSC)  414  generates clock signals (CLK) having a predetermined period and outputs them to the NOR gate  440 , the AND gate  520 , and the SR flipflop  430 . The NOR gate  440  receives the clock signals (CLK) from the oscillator (OSC)  414 , the inverting output signals (/Q) from the SR flipflop  430 , and the signals from the OR gate  470  through the first, second, and third input terminals, and transmits signals generated by a NOR logic operation to the gate driver  450  of the switching transistor (M)  108  to thus control turning on/off the switching transistor (M). The NOR operation outputs a high level signal when the three input signals have a low level, and outputs a low level signal when one of the three input signals is a high level signal. The gate driver  450  outputs a gate voltage (Vg) for turning on/off the switching transistor (M)  108  according to the signal level output by the NOR gate  440 . That is, the gate driver  450  turns on the switching MOS transistor (M) when receiving a high level signal from the NOR gate  440 , and it turns off the switching MOS transistor (M) when receiving a low level signal. 
   An LEB circuit  460  is coupled between a gate of the switching transistor (M)  108  and the second comparator  420 . In this instance, the LEB circuit  460  provides an enable signal to the second comparator  420 . The LEB circuit  460  prevents the switching transistor (M)  108  from being turned off even though the drain current (Ids) is abruptly increased to be greater than the control voltage (Vc) when the switching transistor (M)  108  is turned on at the falling edge time of the clock signal (CLK). The LEB circuit  460  can be implemented with an RC filter. The OR gate  470  receives signals from the first comparator  410  and the AND gate  520  through the first and second input terminals, and outputs a signal to the NOR gate  440 . 
   The protection circuit  500  of the exemplary SMPS includes a third comparator  510  and an AND gate  520 . The third comparator  510  includes a non-inverting terminal (+) and an inverting terminal (−). A reference voltage (V TH ), which may have a value less than the Vc sat  voltage, is coupled to the inverting terminal (−), and the voltage (V FB ) charged in the capacitor (C FB ) is coupled to the inverting terminal (+). 
   Therefore, the feedback voltage (V FB ) is compared with the reference voltage (V TH ) to sense the voltage rising of the feedback voltage (V FB ) before the feedback voltage (V FB ) rises to the Vc sat  voltage. The AND gate  520  receives the signal A output by the inverting output terminal (/Q) of the SR flipflop  430 , the signal B generated by time delaying (Td) the clock signal (CLK) of the oscillator (OSC)  414 , and the output signal C of the third comparator  510  through the first, second, and third input terminals, and outputs a signal to the OR gate  470 . The AND gate  520  outputs a high level signal when the three input signals are high level signals. The high level signal in this case means that the output short (OS) condition is sensed or detected. 
   Referring to  FIG. 5 , an exemplary method for sensing the output short condition, according to an embodiment of the present invention, will now be described. In one embodiment, this method may be performed by the protection circuit  500 .  FIG. 5  is a graph of a signal input to the AND gate  520  of the exemplary protection circuit  500  when an output short is generated. The oscillator (OSC)  414  generates clock signals (CLK) having a predetermined period, and the switching transistor (M)  108  is turned on when the signal is switched from the high level to the low level. The AND gate  520  receives the clock signals (CLK), time-delayed (Td) by the predetermined period of D TH , through the second input terminal of the AND gate  520 . The AND gate  520  receives the signals (A) output by the inverting output terminal (/Q) of the flip-flop  430  through the first input terminal of the AND gate  520 . That is, the signal A has the opposite phase of the clock signals (CLK) while the switching transistor (M)  108  is turned off. The switching transistor (M)  108  is turned off when the signal A is switched from the low level to the high level. 
   Therefore, when the sense voltage (Vsense) is increased to the control voltage (Vc) while the switching transistor (M)  108  is turned on, the second comparator  420  senses the increase and outputs a high level signal to the reset terminal (R) of the SR flipflop  430 . In this instance, since the oscillator (OSC)  414  outputs a low level signal to the set terminal (S), the SR flipflop  430  outputs a high level signal through the inverting output terminal (/Q). 
   Therefore, when the sense voltage (Vsense) is increased to the control voltage (Vc), the signal A output through the inverting output terminal (/Q) of the SR flipflop  430  is switched from the low level signal to the high level signal, and the switching transistor (M)  108  is turned off. In this instance, since the oscillator (OSC)  414  turns on the switching transistor (M)  108  with a predetermined period, the duty ratio of the switching transistor (M) is substantially reduced. That is, the switching transistor (M)  108  is turned on during a time interval D from the time when the clock signal (CLK) is switched from the high level to the low level to the time when the signal (A) is switched from the low level to the high level. Therefore, it is determined that the duty ratio of the switching transistor (M)  108  is substantially reduced when the time interval D in which the switching transistor (M) is turned on is less than the period (D TH ) of time delay (Td). 
   The AND gate  520  receives the output signal C of the third comparator  510  through the third input terminal of the AND gate  520 . Therefore, the third comparator  510  outputs a high level signal when the feedback voltage (V FB ) has a level greater than that of the reference voltage (V TH ) because of the increase of current (Io) applied to the load. In this instance, since the AND gate  520  outputs the high level when the input signals are high level, the duty ratio is substantially reduced. When the feedback voltage (V FB ) has a level greater than that of the reference voltage (V TH ), the output signal (OS) of the AND gate  520  has a high level as shown in  FIG. 5 . Since the OR gate  470  outputs a high level signal when at least one of its input signals is high level, the OR gate  470  outputs a high level signal to the NOR gate  440  when receiving a high level signal from the AND gate  520 . The NOR gate  440  turns off the switching transistor (M)  108  when at least one high level signal is input. As a result, when the signals input to the AND gate  520  become high level signals, the AND gate  520  outputs a high level signal (OS) to shut down the switching transistor (M)  108 . 
   Therefore, according to an embodiment of the present invention, when the output short condition occurs in the SMPS, the output short is sensed before the delay time (T delay ) has elapsed, and the main switch of the primary coil  106  is then shut down. That is, as shown in  FIG. 5 , the switching transistor (M)  108  is shut down at the time T 4 , before the time T 3 . Although the present invention and its advantages have been described in detail, it should be understood that various changes, substitutions, and alterations can be made therein without departing from the spirit and scope of the invention as defined by the appended claims. That is, the discussion included in this application is intended to serve as a basic description. It should be understood that the specific discussion may not explicitly describe all embodiments possible; many alternatives are implicit. It also may not fully explain the generic nature of the invention and may not explicitly show how each feature or element can actually be representative of a broader function or of a great variety of alternative or equivalent elements. Again, these are implicitly included in this disclosure. Where the invention is described in device-oriented terminology, each element of the device implicitly performs a function. Neither the description nor the terminology is intended to limit the scope of the claims.
A switching mode power supply includes a power supply circuit, a feedback circuit, and a switching controller. The power supply circuit includes a main switch coupled to a primary coil of a transformer, and supplies power to a secondary coil of the transformer according to an operation of the main switch. The feedback circuit generates a feedback voltage corresponding to an output voltage provided to the secondary coil of the transformer. The switching controller controls the main switch to turn off according to a sense voltage corresponding to the current flowed by the main switch. In this instance, the switching controller quickly senses the output short phenomenon by using the duty ratio of the main switch and the feedback voltage and shuts down the main switch to protect the circuit.