This application claims priority from Provisional application Ser. No. 60/211,463, filed Jan. 13, 2000. 
    
    
     BACKGROUND 
     1. Field of the Invention 
     The invention relates generally to the field of electronic circuits, and more particularly, to enhancing slew rates in electronic circuits such as amplifier circuits. 
     2. Background Information 
     The present invention relates generally to high speed equipment such as communication systems, displays, and the like, ranging from cell phones to computer displays, in which the large signal behavior of the amplifier is an important consideration. The present invention involves a variety of designs which may provide techniques to achieve good large signal behavior in a variety of circuit arrangements. 
     Analog circuit elements in a system need to apply a specified signal swing to the next element in the circuit. The next element often has a required signal swing for maximum dynamic range, such as the full scale level of an analog to digital converter. As systems evolve for use at higher bandwidths, analog circuit elements, particularly elements with feedback, reach their large signal limits. Large signal performance is usually measured as slew rate, defined as the maximum rate of voltage change possible for a circuit at a given node. 
     The classic slew rate limit in an amplifier often occurs when a stage in the amplifier (usually an input stage) has a fixed maximum current, and this stage must charge capacitance. This is illustrated in FIG. 1, which shows an amplifier circuit  10  that includes an input stage  20  and an output stage  30 . Input stage  20  supplies a current with a fixed maximum to output stage  30 , and this current can be applied to a capacitor  104  that resides in output stage  30 . 
     As shown in FIG. 1, input stage  20  includes a differential pair of transistors  101  and  102  that receive a differential input signal  111  at their respective base terminals  110  and  112 . Differential input signal  111  controls how much current flows through transistors  101  and  102 . A current generator  103  is coupled to transistors  101  and  102  at their respective emitter terminals  114  and  116 . Current generator  103  provides amplifier circuit  10  with current, and this current typically has a fixed maximum. Therefore, the maximum current that transistors  101  and  102  can apply to output stage  30 , and ultimately to capacitor  104 , is the current generated by current generator  103 . This can then determine the maximum charging rate of capacitor  104  by the current-voltage relationship for a capacitor:              ∂   V       ∂   t       =     I   C                                         
     where          ∂   V       ∂   t                            
     is the time rate of change of voltage (slew rate), I is the current  103  and C is the capacitance  104 . 
     To complete the circuit of input stage  20 , transistors  101  and  102  include a pair of collector terminals  118  and  120  that are electrically coupled to a pair of resistors  105  and  106 . Resistors  105  and  106  are in turn coupled to a path to ground  122 . 
     Output stage  30  includes a transistor  107  with an emitter terminal  124  that is electrically coupled to collector terminal  120  of transistor  102 . Transistor  107  is operated by a reference voltage  129  that is applied to a base terminal  126  of transistor  107 . Transistor  107  also has a collector terminal  128  that is coupled to capacitor  104  via output line  132 , and capacitor  104  is in turn coupled to a path to ground  130 . Output line  132  is typically connected to additional circuitry that is unrelated to amplifier circuit  10  for the purposes of this description, and is therefore not shown. 
     While there are a variety of techniques in use to improve slew rate, no single design achieves its goals without limiting performance in other areas. For instance, a common technique well known in the art is input stage degeneration. This technique does provide larger slew rates, but it increases noise and degrades open loop gain. Another common technique uses an input stage which is not current limited, one example of which is set forth in U.S. Pat. No. 5,049,653 issued to Smith, et al., which is hereby incorporated by reference. This technique is commonly used in current feedback amplifiers, and has been implemented in voltage feedback amplifiers as well. The drawback of a non-current limited input stage is that more voltage is required to bias the stage, making it not usable in low supply voltage or battery applications. 
     In addition, it is not feasible to use a non-current limited input stage in single supply applications which require the input common mode range to include one or both supplies. Thus, even though solutions to the problems mentioned in this disclosure have existed, none are believed to have provided the proper balance of competing concerns in most applications and certainly none have met the various criteria which may now be met by the present invention, especially in the low voltage, single supply or battery operated area or the like. Accordingly, there is a need for an improved amplifier system and method that does not limit the slew rate of a circuit. 
     SUMMARY 
     The disadvantages and problems associated with current limited stage amplifiers and other similar circuits have been improved using the present invention. 
     In accordance with an embodiment of the invention, an amplifier circuit comprises an input stage capable of receiving and amplifying an input signal, a gain stage capable of further amplifying the input signal, wherein the gain stage is electrically coupled to the input stage, and an output stage capable of charging a capacitance of the amplifier circuit and outputting the amplified input signal, wherein the output stage is electrically coupled to the gain stage. 
     According to another embodiment of the invention, the input stage of the amplifier circuit comprises a differential pair of transistors, each transistor comprising a base, an emitter, and a collector, a current generator electrically coupled to the emitters of the differential pair of transistors, a pair of input lines electrically coupled to the bases of the differential pair of transistors, the input lines configured to carry an input signal, a pair of resistors electrically coupled to the collectors of the differential pair of transistors, and a path to ground electrically coupled to the pair of resistors. 
     According to yet another embodiment of the invention, the gain stage of the amplifier circuit comprises a pair of gain transistors, each gain transistor comprising a base, an emitter, and a collector, wherein the bases of the gain transistors are electrically coupled to the input stage, wherein the collectors of the gain transistors are electrically coupled to a path to ground, and wherein the emitters of the gain transistors are electrically coupled to the output stage. 
     And according to yet another embodiment of the invention, the output stage of the amplifier circuit comprises a pair of output transistors, each output transistor comprising a base, an emitter, and a collector, wherein the emitters are electrically coupled to the gain stage, a reference voltage line electrically coupled to the bases of the output transistors, the reference voltage line configured to carry a reference voltage signal, and a pair of output lines electrically coupled to the collectors of the output transistors. 
     An important technical advantage of the present invention includes applying the amplified input signals produced in the input stage of an amplifier circuit to the base terminals of transistors in a gain stage of the amplifier circuit. This configuration breaks the typically direct relationship between the input stage and the output stage of an amplifier circuit. So here, when the low current, input stage signals are disassociated from the output stage of the amplifier circuit, the output stage is free to operate at higher currents. The result is an increase in the slew rate of the amplifier circuit. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     For a more complete understanding of the present invention and for further features and advantages, reference is now made to the following description taken in conjunction with the accompanying drawings, in which: 
     FIG. 1 is a schematic diagram of a circuit that shows the source of slew rate limitation; 
     FIG. 2 is a block diagram illustrating an embodiment of the invention; 
     FIG. 3 is a schematic diagram of a known circuit with slew rate limitation; and 
     FIG. 4 is a schematic diagram of a circuit constructed according to an embodiment of the invention. 
    
    
     DETAILED DESCRIPTION 
     The embodiments of the present invention and their advantages are best understood by referring to FIGS. 2 through 4 of the drawings. Like numerals are used for like and corresponding parts of the various drawings. 
     The invention includes systems and methods for providing electronic circuits that have a high slew rate relative to known systems. In addition to increasing the slew rate, the invention has many other practical considerations, such as not increasing noise while maintaining the gain of the circuit. Also, the invention allows the input common mode range to include one or both supplies. The invention produces this slew rate enhancement through the use of gain stages that can be incorporated in a variety of electronic circuits. For instance, the invention can be incorporated into telecommunications-type devices, including cellular or PCS telephones, battery powered or low power devices, and most electronic circuits that utilize amplifier circuits. 
     FIG. 2 is a block diagram illustrating a high level view of an embodiment of the invention. This embodiment begins with an input signal  200 , which can be a differential input signal as shown in FIG.  1 . Input signal  200  is applied to a current limited stage  201 , and this stage is similar to input stage  20  of FIG.  1 . Input signal  200  is amplified by current limited stage  201 , and the resulting amplified signal is then applied to a high current stage  202 . In this embodiment, and unlike the circuit shown in FIG. 1, the amplified signal is applied to high current stage  202  in such a manner that high current stage  202  is not constrained by the limited current of stage  201 . High current stage  202  thus has a much higher maximum current than current limited stage  201 . High current stage  202  then passes a higher current signal on to a capacitor  203 , and capacitor  203  can be connected to a path to ground  204  (as shown in FIG. 2) or to additional circuit elements. The use of a higher current signal can increase the slew rate of the overall circuit. 
     FIG. 3 is a schematic diagram of an input circuit  300  commonly used in single supply amplifiers. Input circuit  300  includes an input stage  312  and an output stage  314 . Input stage  312  receives and amplifies an input signal, and then passes this amplified signal on to output stage  314 . Output stage  314  can utilize the current supplied by input stage  312  via the amplified input signal to charge a pair of capacitors  306  and  307 , and can pass the current on to other circuit elements outside input circuit  300 . 
     Input stage  312  has a pair of transistors  301  and  302  that form a simple differential pair. Transistor  301  includes an emitter (or source) terminal  316 , a base (or gate) terminal  318 , and a collector (or drain) terminal  320 . Transistor  302  includes an emitter terminal  322 , a base terminal  324 , and a collector terminal  326 . Transistors  301  and  302 , as well any or all of the other transistors disclosed herein, can be bipolar junction transistors such as p-n-p or n-p-n bipolar junction transistors. Alternatively, the transistors disclosed herein can be other types of transistors or devices, such as field effect transistors. Transistors may be made of single devices as disclosed herein, or may use multiple devices such as added emitter followers, Darlington connectors, or other composite transistor connectors. 
     A differential pair is a two-transistor amplifier in which a differential input signal  328  is applied to base terminals  318  and  324  of the two transistors  301  and  302  over a pair of input lines  327  and  329 . The output is then taken differentially from collector terminals  320  and  326 . In the differential pair, emitter terminals  316  and  322  are connected together to a constant current generator  303 . Current generator  303  supplies current to input stage  312 , and typically has a fixed maximum limit. 
     Collectors  320  and  326  of transistors  301  and  302  are loaded with a pair of resistors  304  and  305 . Resistors  304  and  305  provide a buffer between the differential pair and a path to ground  330 , without which the current in input stage  312  would flow directly and fiercely to ground and the circuit would not function. Path to ground  330  can include additional circuitry that is not shown here. 
     Output stage  314  includes a pair of transistors  308  and  309  that are electrically coupled to input stage  312 . More specifically, transistors  308  and  309  have emitter terminals  332  and  334  that are electrically coupled to collectors  320  and  326  of transistors  301  and  302 , and that are electrically coupled to resistors  304  and  305 . Transistors  308  and  309  also include base terminals  336  and  338  that are coupled to a reference voltage input line  344 , and collector terminals  340  and  342  that are coupled to output lines  310  and  311 . Transistors  308  and  309  can drive a pair of capacitors  306  and  307  (which may be a combination of parasitic and compensation capacitance) via output lines  310  and  311 , and can drive additional circuitry which is not shown. 
     Capacitors  306  and  307  may be connected to a path to ground  346  (as shown), or connected to other circuit elements that are not relevant to input circuit  300  for the purposes of this description. Reference voltage input line  344  carries a reference voltage  345 , which is typically set such that transistors  308  and  309  conduct the same amount of current as transistors  301  and  302  when input circuit  300  is in a balanced, quiescent state (i.e. when differential input signal  328  is around zero). 
     The transconductuance gain of input circuit  300  in FIG. 3 for a small differential input voltage signal  328  applied between base terminals  318  and  324  of transistors  301  and  302  is approximately the transconductance of a differential pair, given as follows:          l   out     =     gm   ·       V   in     2                              
     where I out  represents the change (from the balanced state) in one of the currents on output lines  310  or  311 , V in  represents differential input signal  328 , and gm represents the small signal transconductance of the input transistors  301  and  302 . 
     The maximum output current occurs when a relatively large differential voltage is applied by differential input signal  328 . Suppose differential input signal  328  is increased to the point where all of the current from current generator  303  flows through transistor  302 , and no current flows through transistor  301 . A low or zero current signal is thus applied to emitter terminal  332  of transistor  308 , and a high current signal is applied to emitter terminal  334  of transistor  309 . Under these conditions, transistor  308  will open since reference voltage signal  344  provides a higher current at base terminal  336  than the low or zero current at emitter terminal  332 , and current will therefore flow through transistor  308 . Transistor  309 , on the other hand, will close since reference voltage signal  344  provides a current at base terminal  338  that is lower than the high current signal at emitter terminal  334 , and current will not be permitted to flow through transistor  309 . 
     The closing of transistor  309  causes the current on output line  311  to approach zero, and the opening of transistor  308  causes the current on output line  310  to double (thus making the current on output line  310  equal to the current generated by current generator  303 ). Accordingly, the maximum output current for circuit  300  in FIG. 3 is limited to the input current provided by current generator  303 . 
     FIG. 4 is a schematic diagram of an input circuit  400  according to an embodiment of the invention where a high current stage, similar to high current stage  202  of FIG. 2, has been added to the system. This high current stage is provided here by a gain stage  418 . Input circuit  400  also includes an input stage  416  and an output stage  420 , similar to input circuit  300  of FIG.  3 . Gain stage  418  is situated between input stage  416  and output stage  420 , and functions as an intermediary to prevent the low current operation of input stage  416  from restricting the current flow in output stage  420 . 
     Input stage  416  includes transistors  401  and  402  that form a simple differential pair, similar to the differential pair in FIG.  3 . Transistors  401  and  402  include base terminals  422  and  424  that are coupled to input lines  426  and  428 , where input lines  426  and  428  carry a differential input signal  430 . Transistors  401  and  402  also include emitter terminals  432  and  434  that are coupled to a current generator  403 . As in FIG. 3, current generator  403  supplies a constant current to input stage  416 , and typically has a fixed maximum limit. Finally, transistors  401  and  402  include collector terminals  436  and  438  that are loaded with resistors  404  and  405  that are in turn coupled to a path to ground  448 . 
     Gain stage  418  is made up of transistors  412  and  413 . The addition of transistors  412  and  413  allows the maximum output current to be greater than the current supplied by current generator  403 . Here, the amplified input signal generated by input stage  416  is used primarily to operate transistors  412  and  413  of gain stage  418 , and is not used to feed current into output stage  420 . This is done by coupling collectors  436  and  438  of resistors  401  and  402  to a pair of base terminals  440  and  442  on transistors  412  and  413 . Now current flowing from collectors  436  and  438  will control transistors  412  and  413  of gain stage  418 . Transistors  412  and  413  also include collector terminals  444  and  446  that are coupled to the path to ground  448 , and emitter terminals  450  and  452  that are coupled to output stage  420 . 
     Output stage  420  is constructed in similar fashion to output stage  314  of FIG.  3 . Output stage  420  includes a pair of transistors  408  and  409  that are coupled to output lines  410  and  411 , and can include a pair of capacitors  406  and  407  (which again may be a combination of parasitic and compensation capacitance). Output lines  410  and  411  can be coupled to additional circuitry that is not relevant to input circuit  400  for the purposes of this description, and which is therefore not shown. Capacitors  406  and  407  are coupled to output lines  410  and  411 , and can be coupled to a path to ground  454  (as shown), or to additional circuitry (not shown). Transistors  408  and  409  can drive capacitors  406  and  407  over output lines  410  and  411 . 
     Transistors  408  and  409  include emitter terminals  456  and  458 , collector terminals  460  and  462 , and base terminals  464  and  466 . Emitter terminals  456  and  458  are coupled to gain stage  418  via emitter terminals  450  and  452  of transistors  412  and  413 . Collector terminals  460  and  462  are coupled to output lines  410  and  411 . Base terminals  464  and  466  are coupled to a reference voltage line  468  that carries a reference voltage  470 . 
     Reference voltage  470  can be set in a variety of ways, depending on the required performance of input circuit  400 . This, as well as the advantages of input circuit  400 , can be demonstrated through some examples. The following examples are intended only as illustrations of the performance advantages of input circuit  400 , and should not be construed as limiting the invention. 
     In the first example, reference voltage  470  can be set so that the transconductance gain of input circuit  400  of FIG. 4 is the same as the transconductance gain of input circuit  300  of FIG.  3 . This can be done by making the sum of the transconductances (gm) of transistors  412  and  408  equal to the inverse of the resistance of resistor  404 . Similarly, the sum of the transconductances of transistors  413  and  409  could be made equal to the inverse of the resistance of resistor  405 . This can make the gains of FIGS. 3 and 4 the same. 
     The maximum output current in FIG. 4 can occur as it does in FIG. 3, when a relatively large differential voltage is applied by differential input signal  430 . Suppose differential input signal  430  is increased to the point where all of the current provided by current generator  403  flows though transistor  402 , and no current flows through transistor  401 . This can cause the voltage drop across resistor  405  to double over its quiescent value, and the voltage drop across resistor  404  to approach zero. This voltage can be applied to transistors  412 ,  413 ,  408 , and  409 , causing the current on output line  411  to approach zero, and the current on output line  410  to increase non-linearly. Here, the current on output line  410  does not have an abrupt limit as it does in input circuit  300  of FIG.  3 . The current on output line  410  is limited only by the amount of voltage applied and the transistor gain at high currents. 
     In the second example, the advantage of input circuit  400  is demonstrated using numerical values. Here, the quiescent value of the voltage drop across resistors  404  and  405  can be set to 250 mV. This is a typical value for a circuit which allows the common mode voltage of differential input signal  430  to become slightly negative and still maintain an amplifier function. The resistance of resistors  404  and  405  can then be set to 10 k ohms. This means that the current provided by current generator  403  has a value of about 50 uA. For the gain in FIG. 4 to equal the gain in FIG. 3, the sum of the transconductances of transistors  412  and  408  must be equal the inverse of the resistance of resistor  404 . This determines the operating current of transistors  412  and  408 . For a bipolar transistor:        gm   =       I   C       V   t                              
     where I C  is the collector current and V t  is the thermal voltage, which is around 26 mV at 27° C. 
     For the sum of gm in transistors  412  and  408  to equal {fraction (1/10)} k ohms, the quiescent collector current of transistors  412  and  408  could be set to 1.3 uA via reference voltage  470 . This can also cause the quiescent collector current in transistors  413  and  409  to be set to 1.3 uA. Note that this results in a significant power savings compared to transistors  308  and  309  in FIG. 3, which would run at 25 uA collector current given the assumption of current generator  303  providing 50 uA current and transistors  308  and  309  having the same currents as transistors  301  and  302 . 
     The maximum output current may be calculated from the current-voltage relation for a bipolar transistor:          I   C     =       I   S     ·     exp        (       V   be       V   t       )                                
     where I C  represents the collector current, I S  represents a current parameter dependent on transistor construction, exp ( ) represents constant e (2.72 . . . ) raised to a power, V be  represents the base emitter voltage, and V t  represents the thermal voltage. 
     It is also useful to calculate the change in collector current due to the change in V be :            I   C1       I   C2       =     exp   (       (       V   be1     -     V   be2       )       V   t                                
     In this numerical example, the change in V be  of each transistor  412  and  408  can be half the change in voltage across resistor  404 . The change in V be  can be 125 mV, as the quiescent drop across resistor  405  doubles from 250 mV to 500 mV, and the drop across resistor  404  reduces from 250 mV to zero. This can give an increase in the current in output line  410  of 122 times the quiescent current of 1.3 uA, resulting in a maximum current output of over 150 uA, three times that of input circuit  300  in FIG.  3 . 
     Again, the values provided above are intended for use as examples and illustrations of the invention only; many values of impedance and current are possible depending on specific requirements in a circuit. 
     Accordingly, systems and methods of the invention have been described for increasing the slew rate of a circuit by minimizing the effect of a current limited stage. Unlike previously developed techniques in which current limited stages have a direct effect on subsequent circuit elements, the present invention utilizes an intervening gain stage that uses the signal from the current limited stage as a control signal, and not as a current providing signal. This results in an output stage that is free to maximize the current that flows through it. The present invention thus provides a method of increasing the slew rate of a circuit without introducing additional noise or decreasing the available gain, provides the possibility of increasing slew rate in a circuit while still allowing for single supply operation on a low voltage supply, and allows for further objectives that are disclosed or understood throughout the specification and drawings. 
     While various embodiments of the invention have been shown and described, it will be apparent to those skilled in the art that numerous alterations may be made without departing from the inventive concepts presented herein. Thus, the invention is not to be limited except in accordance with the following claims and their equivalents.
An amplifier circuit comprising an input stage capable of receiving and amplifying an input signal, a gain stage electrically coupled to the input stage that is capable of further amplifying the input signal, and an output stage electrically coupled to the gain stage that is capable of charging a capacitance of the amplifier circuit and outputting the amplified input signal. The gain stage of the amplifier circuit comprises a pair of gain transistors with base terminals that are electrically coupled to the input stage, collector terminals that are electrically coupled to a path to ground, and emitter terminals that are electrically coupled to the output stage.