CROSS-REFERENCE TO RELATED APPLICATIONS 
       [0001]    This is a continuation of International Application No. PCT/JP2015/003062 filed on Jun. 18, 2015, which claims priority to Japanese Patent Application No. 2014-173729 filed on Aug. 28, 2014. The entire disclosures of these applications are hereby incorporated by reference. 
     
    
     BACKGROUND 
       [0002]    The present disclosure relates to a frequency modulated continuous wave (FMCW) radar. 
         [0003]    An FMCW radar is known as a type of distance measuring radar. An FMCW radar continuously transmits a signal with a frequency varying with time, and receives a reflected wave from an object. The radar can measure the distance to the object and its moving velocity by analyzing the reflected wave. Such a radar includes a transmission system and a reception system. In a radar of this type, however, a signal sometimes leaks from the transmission system to the reception system, resulting in a lower measurement precision. In particular, as a circuit with such a radar decreases its overall size, the distance between the transmission and reception systems also decreases. As a result, the leakage signal comes to have an even more significant negative effect on its performance. 
         [0004]    According to a known technique, a receiver supplies an FM signal to a mixer through a delay device causing the same amount of time delay as that of a leakage signal component from a transmission system to a reception system, thereby converting the leakage signal component into a direct-current (DC) component. Then, the receiver further removes the DC component to attempt to cancel the leakage signal component (see Japanese Unexamined Patent Publication No. H11-183600). 
       SUMMARY 
       [0005]    Actually, the amplitude and phase of the leakage signal component from the transmission system to the reception system change as the transmission frequency changes. Thus, according to the known technique, the leakage signal component remains as an alternating current (AC) signal having the same period as one period of variation of the transmission frequency. This makes it impossible for a DC component removal circuit to totally cancel the leakage signal component. Furthermore, the precision of short-distance measurement, in particular, tends to decline disadvantageously, because the removal of the DC component actually involves removal of low frequency components near the DC components. 
         [0006]    It is therefore an object of the present disclosure to provide an FMCW radar having the ability to effectively prevent the leakage signal from affecting the measurement precision, even if the transmission and reception systems are arranged close to each other, and to precisely measure the distance to the object, no matter how short it may be, even if the amplitude and phase of the leakage signal vary with the transmission frequency, in particular. 
         [0007]    To achieve this object, an FMCW radar according to an aspect of the present disclosure includes: a transmission signal generator configured to generate a frequency-modulated transmission signal; a transmission signal sender configured to send the transmission signal; a receiver configured to receive a reflected wave of the transmission signal; an adjuster configured to adjust an amplitude and phase of a cancel signal, which cancels a leakage signal component in a received signal, in accordance with a variation in a frequency of the transmission signal; and a superimposer configured to superimpose the cancel signal over the received signal to cancel the leakage signal component. 
         [0008]    According to this aspect of the present disclosure, the amplitude and phase of a cancel signal, which cancels a leakage signal component in a received signal, are adjusted in accordance with a variation in the frequency of a transmission signal. Thus, the present disclosure provides an FMCW radar having the ability to effectively prevent the leakage signal from affecting the measurement precision, even if the transmission and reception systems are arranged close to each other, and to precisely measure the distance to the object, no matter how short it may be, even if the amplitude and phase of the leakage signal vary with the transmission frequency, in particular. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS 
         [0009]      FIG. 1  is a block diagram of an FMCW radar according to an exemplary embodiment of the present disclosure. 
           [0010]      FIG. 2  is a block diagram showing an exemplary detailed configuration for a radio frequency phase shift (RFPS) circuit shown in  FIG. 1 . 
           [0011]      FIG. 3  is a block diagram showing an exemplary detailed configuration for a clutter cancellation (CLC) section of an image rejection and clutter cancellation (IMR &amp; CLC) circuit shown in  FIG. 1 . 
           [0012]      FIG. 4  is a vector diagram describing how the CLC section shown in  FIG. 3  operates. 
           [0013]      FIG. 5  is a block diagram of an FMCW radar according to a variation of the exemplary embodiment shown in  FIG. 1 . 
       
    
    
     DETAILED DESCRIPTION 
       [0014]    Embodiments of the present disclosure will now be described in detail with reference to the accompanying drawings. 
         [0015]      FIG. 1  is a block diagram of an FMCW radar according to an exemplary embodiment of the present disclosure. The FMCW radar shown in  FIG. 1  is designed to adjust, in accordance with a variation in the frequency of a transmission signal, the amplitude and phase of a cancel signal for cancelling a leakage signal component (a clutter signal component) from a transmission system to a reception system. The FMCW radar includes a reference (clock) signal generator (CLK)  110 , a local oscillator (LO)  111 , first and second attenuators/low pass filters (ATTs/LPFs)  112 ,  113 , a first mixer  114 , a power amplifier (PA)  115 , a transmission antenna  116 , a hybrid (HYB) circuit  117 , a radio frequency phase shift (RFPS) circuit  118 , a reception antenna  120 , a low noise amplifier (LNA)  121 , a second mixer  122 , an image rejection and clutter cancellation (IMR &amp; CLC) circuit  123 , an intermediate frequency (IF) mixer  124 , analog-to-digital converters (ADCs)  125 ,  126 , and a digital signal processor (DSP)  130 . 
         [0016]    The CLK  110  generates a reference signal for generating a transmission signal. The first ATT/LPF  112  supplies, to the first mixer  114 , the reference signal separated into in-phase (I) and quadrature (Q) components. The second ATT/LPF  113  supplies, to each of the IMR &amp; CLC circuit  123  and the IF mixer  124 , the reference signal separated into I and Q components. The LO  111  supplies, to the first and second mixers  114 ,  122 , a frequency-modulated local oscillation signal, and sequentially supplies, to the DSP  130 , pieces of frequency information FI indicating frequencies at respective points in time during frequency sweeping. The first mixer  114  upconverts the reference signal into an RF signal, based on the associated frequency-modulated local oscillation signal, thereby generating a frequency-modulated transmission signal. The PA  115  amplifies the RF signal from the first mixer  114  to transmit the frequency-modulated transmission signal to the transmission antenna  116 . The transmission antenna  116  radiates an electromagnetic wave toward an object. 
         [0017]    The reception antenna  120  receives a reflected wave of the frequency-modulated transmission signal from the object, and receives a clutter signal component leaking from the transmission antenna  116 . On the other hand, the HYB circuit  117  has an input capacitively coupled to the output of the PA  115 , and separates the transmission signal, which is a single signal, into I and Q signal components having a phase difference of 90°. The RFPS circuit  118  adjusts, in accordance with a first adjustment value ADJ 1  supplied from the DSP  130  based on the frequency information FI, the amplitude of each of the I and Q signal components supplied from the HYB circuit  117 , thereby generating a cancel signal having an opposite phase to the clutter signal component. That is to say, the cancel signal is adjusted, for each frequency, to have the same amplitude as, and an opposite phase to, the clutter signal component. The RFPS circuit  118  has an output connected to the input of the LNA  121 , where the cancel signal is superimposed over a received signal to cancel the clutter signal component. 
         [0018]    The LNA  121  amplifies the received signal including the clutter signal component that may partially remain. The second mixer  122  downconverts the received signal into an intermediate frequency (IF) signal separated into I and Q signal components, based on the associated frequency-modulated local oscillation signal. The IMR &amp; CLC circuit  123  provides image rejection, generates a cancel signal having an opposite phase to the clutter signal component in accordance with a second adjustment value ADJ 2  supplied from the DSP  130  based on the frequency information FI, and superimposes the cancel signal over the received signal to cancel the clutter signal component. The IF mixer  124  downconverts the received signal from which the clutter signal component has been removed. The first and second ADCs  125  and  126  each convert the output of the IF mixer  124 , separated into I and Q signal components, into a digital signal, and then passes it to the DSP  130 . The DSP  130  calculates the distance to the object and the moving velocity of the object by analyzing the reflected wave from the object based on the outputs of the first and second ADCs  125  and  126 . 
         [0019]    The DSP  130  operates in a calibration mode prior to the foregoing normal operation. For example, in a calibration mode for the RFPS circuit  118 , the DSP  130  stores, as a correction value in its internal memory, the first adjustment value ADJ 1  that is set such that the input level to the DSP  130  becomes minimum with no reflected wave input from the object. This correction value corresponds to one of the pieces of frequency information FI received from the LO  111  at this point in time. During the normal operation, the RFPS circuit  118  may be appropriately adjusted based on the first adjustment value ADJ 1  derived from the correction value in the memory. The calibration may be performed again either regularly or irregularly, not only during an initial stage. The correction value may also be obtained sporadically at only some points in time in accordance with a variation in transmission frequency. In the interval between those points in time, an interpolation (e.g., a linear interpolation) may be performed to obtain the correction value. A value obtained by performing an arithmetic operation on (e.g., by calculating a moving average of) the results of calibrations that have been carried out regularly or irregularly a number of times may be used as the correction value. The above statement applies not only to the RFPS circuit  118  but also to the IMR &amp; CLC circuit  123 . Furthermore, if the clutter signal component is suppressed by both of the RFPS circuit  118  and the IMR &amp; CLC circuit  123 , a calibration mode of operation for the RFPS circuit  118  is initially performed, where no cancel signal is output from the IMR &amp; CLC circuit  123 . Next, a calibration mode of operation for the IMR &amp; CLC circuit  123  is performed, where the RFPS circuit  118  outputs a cancel signal in accordance with the first adjustment value ADJ 1  obtained during the calibration mode of operation previously performed. This allows the IMR &amp; CLC circuit  123  to cancel the clutter signal component that has not been completely suppressed by the RFPS circuit  118 . 
         [0020]      FIG. 2  shows an exemplary detailed configuration for the RFPS circuit  118  shown in  FIG. 1 . The RFPS circuit  118  shown in  FIG. 2  includes two balanced-unbalanced converters (baluns (BLNs))  201 ,  202  on its input end, two variable gain amplifiers (VGAs)  203 ,  204 , and one balun (BLN)  205  on its output end. The two BLNs  201 ,  202  each convert an associated one of the I and Q signal components, which are two single signal components, into a differential pair of signal components. Then, one of the VGAs, i.e., the VGA  203 , amplifies the differential pair of I signal components, while the other VGA, i.e., the VGA  204 , amplifies the differential pair of Q signal components. At this time, the amplitudes of the two outputs of each of these VGAs  203 ,  204  are adjusted in accordance with the first adjustment value ADJ 1 . When finally passing through the BLN  205 , these two pairs of amplified I and Q signal components are superimposed one over the other, thereby obtaining a single output signal. In this embodiment, the baluns and the variable gain amplifiers for two differential pairs of signal components are used. However, variable gain amplifiers for single signal components may be used without using baluns. 
         [0021]      FIG. 3  shows an exemplary detailed configuration for a CLC section of the IMR &amp; CLC circuit  123  shown in  FIG. 1 . The CLC section shown in  FIG. 3  includes an operational amplifier  210  serving as an inverting amplifier, first input resistors Rin 1 , second input resistors Rin 2 , and feedback resistors Rf, and synthesizes signals together at a virtual ground node. The second input resistors Rin 2  each have a variable resistance value, which is adjusted in accordance with the second adjustment value ADJ 2 , thereby controlling the respective amplitudes of the cancel signals I and Q. Adjusting the respective amplitudes of these cancel signals I and Q as described above allows this CLC section to produce a cancel signal I+Q having the same amplitude as, and an opposite phase to, the clutter signal component as shown in  FIG. 4 . 
         [0022]    As can be seen from the foregoing description, the configuration shown in  FIG. 1  may curb a decline in the precision of the distance measured, no matter how short it may be, even if a variation in the transmission frequency has triggered a variation in the amplitude and phase of the clutter signal component. In addition, the RFPS circuit  118  and the IMR &amp; CLC circuit  123  may adjust the amplitude and phase of the cancel signal simply by adjusting only the respective amplitudes of the I and Q signal components. This may conveniently reduce the circuit size. 
         [0023]    Optionally, only one of the RFPS circuit  118  or IMR &amp; CLC circuit  123  shown in  FIG. 1  may be implemented. The RFPS circuit  118  is closer to the reception antenna  120  than the IMR &amp; CLC circuit  123 , and therefore, more effectively cancels the clutter signal component. This reduces the power input to the LNA  121  and the second mixer  122 , thus alleviating the distortion characteristic. Nevertheless, the IMR &amp; CLC circuit  123  operates in a lower frequency range and therefore, makes the cancellation control easier, than the RFPS circuit  118  does, which is beneficial. Alternatively, the output of the RFPS circuit  118  may be connected between the LNA  121  and the second mixer  122 . 
         [0024]    Instead of the LO  111 , the CLK  110  may perform the frequency sweeping. In that case, not the frequency information FI of the LO  111  but frequency information of the CLK  110  is provided for the DSP  130 , which supplies the adjustment values ADJ 1 , ADJ 2  in accordance with this frequency information. 
         [0025]      FIG. 5  is a block diagram of an FMCW radar according to a variation of the embodiment shown in  FIG. 1 , and shows an example in which the reference signal is not separated into I and Q signal components. In  FIG. 5 , the IMR &amp; CLC circuit  123  shown in  FIG. 1  is replaced with a simple CLC circuit  123 . In addition, the configuration shown in  FIG. 5  needs neither the ATTs/LPFs  112 ,  113  nor the HYB circuit  117  unlike the configuration shown in  FIG. 1 . 
         [0026]    For both of the configurations shown in  FIGS. 1 and 5 , all of the circuit elements but the transmission antenna  116  and the reception antenna  120  may be integrated together in a single chip. 
         [0027]    As can be seen from the foregoing description, the FMCW radar according to an exemplary embodiment of the present disclosure updates the cancel signal in accordance with a variation in the transmission frequency, thus allowing the cancel signal to follow the variation pattern of the clutter signal component. This may effectively prevent the precision of the distance measured from declining, even if the transmission and reception systems are arranged close to each other.
Disclosed herein is a frequency modulated continuous wave (FMCW) radar. The FMCW radar includes: a transmission signal generator that generates a frequency-modulated transmission signal; a transmission signal sender that sends the transmission signal; a receiver that receives a reflected wave of the transmission signal; an adjuster that adjusts the amplitude and phase of a cancel signal, which cancels a leakage signal component in a received signal, in accordance with a variation in the frequency of the transmission signal; and a superimposer that superimposes the cancel signal over the received signal to cancel the leakage signal component.