CROSS REFERENCE TO RELATED APPLICATION 
       [0001]    This is a continuation of U.S. application Ser. No. 11/033,905, filed Jan. 13, 2005. This application relates to and claims priority from Japanese Patent Application No. 2004-006864, filed on Jan. 14, 2004. The entirety of the contents and subject matter of all of the above is incorporated herein by reference. 
     
    
     BACKGROUND 
       [0002]    This invention relates to a wireless transmitter/receiver having a function of adjusting, by way of closed-loop feedback control, the timing of a transmission signal. In particular, the invention relates to a wireless transmitter/receiver that employs EER to adjust the timing of sending an r (amplitude) component and a θ (phase) component. 
         [0003]    For transmitters installed in base stations and terminals of cellular phone and other wireless communication systems, methods have been developed to separate a transmission signal into two components, process the two separately, and then recombine them as a transmission output. Known examples of such methods include one in which a transmission signal is separated into an I signal and a Q signal to be processed separately, and EER (Envelope Elimination and Restoration) in which a transmission signal is separated into an r (amplitude) component and a θ (phase) component to be processed separately. 
         [0004]    However, signals created by separating one signal into two components to be processed separately differ from each other in signal propagation delay time in the case where processing circuits of the two components are arranged to present different signal path lengths. The separated signals could also differ from each other in signal processing time if different processing circuits are used to process the two. These result in disordered timing of recombining the separated signals into one signal and lowered quality of the recombined signal. 
         [0005]    In EER, an r signal and a θ signal are combined, after receiving supply voltage modulation and frequency conversion, respectively, into one signal by a saturated power amplifier, which has high efficiency (see Je-Kuan Jau, “Linear Interpolation Scheme for Compensation of Path-Delay Difference in an Envelope Elimination and Restoration Transmitter”, pp1072-1075, Proceedings of APMC2001). The power consumption of a power amplifier which amplifies a transmission signal in a communication device constitutes a very large portion of the total power consumption of the communication device, and improving the power amplifier efficiency is a technical objective to be reached. EER addresses this objective by using a saturated power amplifier, which is high in efficiency, and is considered to be effective in reducing the size, cost and power consumption of a communication device. 
         [0006]    A problem of EER is that it is liable to give r and θsignals different delay amounts from each other. While a supply voltage modulation circuit to process the r component is composed of, for example, a DC-DC converter, a frequency converter to process the θ signal is composed of a mixer or the like. Because of the vastly different circuit elements used by the two circuits, the delays caused in the process of signal processing mess up the timing of recombining the signals into one signal and greatly lower the quality of the recombined signal. 
         [0007]      FIG. 7  is a waveform chart showing the principle of degradation in transmission signal quality due to delays of r and θ signals in EER. 
         [0008]    When a sine wave is inputted to a transmitter that employs EER, the waveform of an amplitude component r (θ)  101  is that of a sine wave folded back along the x axis whereas a phase component p (θ)  102  forms a square wave. Normally, the folding back of the r (θ)  101  synchronizes with the phase inversion of the p (θ)  102 . Here, consider the case where the phase inversion of the p (θ)  102  is behind the folding back of the r (θ)  101  by τ. A signal S (θ)  103 , which is obtained by recombining these r (θ)  101  and p (θ)  102 , experiences discontinuous phase inversions for the period τ, causing sharp peaks in an error signal u (θ)  104 . This error signal component turns into spurious output (noise) and lowers the quality of the signal. In order to obtain a desired signal quality, the delay difference between the r signal and the θ signal somehow has to be adjusted to align the folding back and the phase inversion with each other. 
         [0009]      FIG. 8  is a block diagram illustrating a conventional timing adjustment method for a wireless transmitter/receiver that employs EER. In  FIG. 8 , the amount of delay along a signal path for r (amplitude)  201  and the amount of delay along a signal path for θ (phase)  202  are made equal to each other by inserting a delay Δdd, which corresponds to the delay difference, Δdr minus Δdθ, to one of the paths where a delay caused by a circuit element is smaller (here, Δdr is larger than Δdθ and the delay Δdd is given to the θ side). It is a digital region where Δdd is inserted in  FIG. 8  and, when Δdd is an integer multiple of the clock cycle, the adjustment can be made by simply delaying the θ signal by n clocks with the use of a shift register  203  or the like. 
         [0010]    Usually, the delay scale is smaller than one clock and external factors such as a temperature variation make delay fluctuate with time. Je-Kuan Jau proposes a method of adjusting delay at a precision of 1/2 clock with the use of a digital filter, which performs a linear interpolation on a transmission signal. This structure uses, as  FIG. 8  shows, a single master clock source  206  (fixed frequency) to drive digital-to-analog converters (DAC)  204  and  205  of the two paths. 
         [0011]    Described next is an example of a timing adjustment method using a feedback (Fb) circuit for a wireless transmitter/receiver that employs other systems than EER. 
         [0012]      FIG. 9  is a block diagram showing a timing adjustment method for a transmission signal and a feedback signal in a predistortion (distortion compensation) transmitter (see JP 2001-189685 A). 
         [0013]    In  FIG. 9 , a feedback circuit  301  receives a signal that has been amplified by a power amplifier (PA)  302  and compares the amplified signal against the original transmission signal to measure the amount of nonlinear distortion caused along a transmission signal path (signal path for Tx)  303  including the power amplifier  302  and other elements. A pre-distortor co-efficient calculation unit  304  calculates a coefficient that gives a distortion of the reverse characteristic to cancel the non-linear distortion, and sets the obtained coefficient to a pre-distortor  305 . The pre-distortor  305  applies a non-linear distortion determined by the set coefficient to the transmission signal, which is then sent to a frequency-conversion and amplifier unit  307  through a DAC  306 . 
         [0014]    Meanwhile, in order to align the transmission signal and the feedback signal with each other for comparison, a delay time calculation unit  308  detects the delay difference between the two and determines the amount of delay of a shift register  309  (Δd 1 ) and the amount of delay of a variable delay element  310  (Δd 2 ). The delay amount Δd 1  of the shift register  309  corresponds to a delay for the transmission signal by an integer multiple of the clock cycle. The delay amount Δd 2  of the variable delay element  310  corresponds to a delay on the 1/n clock-basis of the clock phase of an analog/digital converter (ADC)  311 , which converts the feedback signal into a digital signal. In this structure, the same single master clock source is used to drive the DAC  306  and the ADC  311 , and the clock phase of the master clock source is fixed. 
       SUMMARY 
       [0015]    While being capable of adjusting a delay smaller than one clock, the method shown in  FIG. 8  has a problem of signal quality degradation since quantization error remains in the adjustment amount, due to its inconsecutiveness, and the group delay characteristics of the filter make the amount of delay vary from frequency to frequency. Although Δdd could be inserted in an analog region with the use of a delay line, the resultant performance is poor because causing a delay of accurate amount is difficult and the analog element is greatly fluctuated in characteristic by external factors such as a temperature variation. In addition, discontinuously changing the delay amount to be inserted causes, at an instant, a discontinuous change in waveform of the transmission signal and it can worsen the spurious output characteristics of the transmission signal. 
         [0016]    The method shown in  FIG. 9  suffers from quantization error remaining in the adjustment amount as does the method of  FIG. 8 . Furthermore, the method of  FIG. 9  is for timing the transmission signal and the feedback signal with each other, not for adjusting a delay difference between the two components of the transmission signal, and therefore is not capable of solving the problem of EER. 
         [0017]    This invention has been made in view of the above problems and it is therefore an object of this invention to provide a method of adjusting, through a simple circuit, with high precision, a delay difference between an r signal and a θ signal in a transmitter/receiver that employs EER. It is another object of this invention to provide a method of adjusting a delay difference between paths of two signals such as an I signal and a Q signal which are originally one signal and recombined after traveling along the paths of different delay amounts for modulation in a transmitter/receiver that employs other systems than EER. 
         [0018]    According to the present invention, there is provided a transmitter comprising: a DA converter unit which converts, into analog signals, two digital signals that are obtained by separating an inputted digital signal or two digital signals that are separately inputted; a combiner which combines the analog signals obtained through the conversion by the DA converter unit with each other; a distributor which extracts a portion of the combined signal as a feedback signal; an AD converter which converts the feedback signal into a digital signal; an oscillator unit which supplies clock signals to operate the DA converter unit and to the AD converter; a first separation unit which separates the feedback signal converted by the AD converter into two signals; and a comparator unit which compares the separated input signal and the feedback signal separated by the first separation unit, wherein the oscillator unit controls the output clock signals based on a result of the comparison by the comparator unit. 
         [0019]    Unlike the conventional methods which directly measure a delay difference between an r signal and a θ signal, this invention judges the timing of the transmission signal and the timing of the feedback signal in relation to each other (one is behind or ahead of the other), and therefore a measure to compare the two can have a simple structure. 
         [0020]    Furthermore, this invention uses a VCO (Voltage Controlled Oscillator) as a clock signal generating unit, thereby making it possible to follow changes in delay amount in a continuous manner and give the transmission signal improved spurious output characteristics compared to the conventional methods. 
         [0021]    As a result, an output signal of a transmitter/receiver that employs EER is improved in quality and the efficiency of its power amplifier is enhanced. The output signal quality is also improved in a transmitter/receiver that employs other systems than EER while keeping the area and power consumption of a control circuit small. 
     
    
     
       BRIEF DESCRIPTION OF THE DRAWINGS  
         [0022]      FIG. 1  is a block diagram showing a common structure of a transmitter/receiver used for a wireless communication system. 
           [0023]      FIG. 2  is a block diagram showing the structure of an RF transmitter unit according to a first embodiment of this invention. 
           [0024]      FIG. 3  is a block diagram showing the structure of a DLL used in this invention. 
           [0025]      FIG. 4  is a waveform diagram showing an example of signal power changes with time of the input of the DLL circuits. 
           [0026]      FIG. 5  is a block diagram showing the structure of an RF transmitter unit according to a second embodiment of this invention. 
           [0027]      FIG. 6  is a block diagram showing the structure of an RF transmitter unit according to a third embodiment of this invention. 
           [0028]      FIG. 7  is a waveform chart showing the principle of degradation in transmission signal quality due to delays of r and θ signals in EER. 
           [0029]      FIG. 8  is a block diagram showing a conventional structure of a transmitter/receiver that employs EER. 
           [0030]      FIG. 9  is a block diagram showing a conventional structure of a predistortion transmitter. 
       
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS  
       [0031]      FIG. 1  is a block diagram showing a common structure of a transmitter/receiver for a wireless base station. 
         [0032]    The transmitter/receiver is composed of an interface unit  401 , which is connected to a public switched telephone network or a packet switching data network, a baseband unit  402 , which performs digital modulation/demodulation processing, an RF transmitter unit  403 , which performs, on a transmission signal, digital-to-analog conversion to convert the transmission signal into an analog signal as well as frequency conversion for conversion from the baseband bandwidth to the RF bandwidth, and which amplifies the output power, a frontend unit  404 , which is composed of a filter, a duplexer and others, an antenna  405 , and an RF receiver unit  406 , which converts the frequency of the transmission signal from the RF bandwidth to the baseband bandwidth and which performs analog-to-digital conversion on the transmission signal after out-of-band noise is removed with the use of a filter. This invention particularly relates to the structure of the RF transmitter unit  403 . 
         [0033]      FIG. 2  is a block diagram showing the structure of the RF transmitter unit  403  according to a first embodiment of this invention. 
         [0034]    A transmission signal inputted from the baseband unit  402  is separated into an r (amplitude) component and a θ(phase) component by an rθ separation unit  501 . The r component and the θ component are converted into analog signals in digital-to-analog converters (DAC)  502  and  503 , respectively. The r signal is converted by a drain voltage modulation unit  504  into a signal to control the supply voltage (drain voltage) of a power amplifier (PA)  505 . The converted signal is inputted to a supply terminal of the power amplifier  505 , with the result that the envelope curve of an output signal of the power amplifier  505  resembles the waveform of the r signal. The θ signal receives, in a frequency-conversion and amplifier unit  506 , frequency conversion to be converted to the RF bandwidth (up conversion) and power amplification. The amplified θ signal is inputted into a signal input terminal of the power amplifier  505 . 
         [0035]    A portion of an output of the power amplifier  505  is distributed to a feedback circuit  507  by a distributor (coupler, or the like, omitted from the drawing). In the feedback circuit  507 , a frequency-conversion and attenuator unit  508  attenuates the power of the distributed output, which then receives frequency conversion to be converted to the baseband bandwidth (down conversion) and is converted into a digital signal by an analog-to-digital converter (ADC)  509 . Thereafter, this feedback signal is separated into an r (amplitude) component and a θ (phase) component by an rθ separation unit  510 . The r component of the feedback signal is compared by an r signal DLL (Delay Locked Loop)  511  against the r component of the transmission signal whereas the θ component of the feedback signal is compared by a θ signal DLL  512  against the θ component of the transmission signal. 
         [0036]    With the circuit structure to be described below, the DLL  511  and  512  judge which one of a phase of a transmission signal and a phase of a feedback signal is ahead of (or behind) the other. Based on the judgment, the DLL circuits control the output phase of a voltage controlled oscillator (VCO)  513 , which supplies clock signals to the r signal DAC  502 , and the output phase of a VCO  514 , which supplies clock signals to the θ signal DAC  503 , in a manner that makes the difference converges to zero (in a manner that makes the timing of the transmission signal and the timing of the feedback signal synchronize with each other). The feedback signal ADC  509  receives clocks from the oscillator (master clock source)  516  of which output frequency is fixed. 
         [0037]    In the case where a delay difference between the path of the r signal and the path of the θ signal is one clock or more, the integer portion of the difference (in the example of  FIG. 2  where Δdr is larger than Δdθ, the integer portion corresponds to the quotient of Δdr minus Δdθ divided by the clock cycle) is adjusted by a shift register  515  whereas the remainder of the division is adjusted by controlling the VCOs  513  and  514 . Therefore, it is not necessary to widen the phase (frequency) control range of the VCOs  513  and  514  and the output signal characteristics (for example, C/N) of the VCOs can be improved. 
         [0038]    The DACs  502  and  503  operate on clock signals created by the VCOs  513  and  514 , independent of the timing at which their input signals change. The DACs  502  and  503  may accordingly take in input signals at the very moment the input signals undergo changes. In this case, bits of the input signals do not necessarily change simultaneously and wrong data could be inputted to the VOCs depending on input timing. As preventative measures, sample-and-hold (S/H) circuits  517  and  518  are provided on the input side of the DAC  502  along the r signal path and on the input side of the DAC  503  along the θ signal path, respectively. The sample-and-hold circuits  517  and  518  hold data to avoid skipping data that is to be inputted to the DACs  502  and  503  despite a change in timing of inputting clocks in the DACs  502  and  503 . 
         [0039]      FIG. 3  is a block diagram showing the structure of the DLL  511  and  512  used in this invention. This circuit structure is disclosed in, for example, JP 2003-273663 A. 
         [0040]    In the DLL  511 , an input  1  corresponds to the r component of the transmission signal, an input  2  corresponds to the r component of the feedback signal, and a VCO control signal output is connected to the VCO  513 . In the DLL  512 , an input  1  corresponds to the θ component of the transmission signal, an input  2  corresponds to the θ component of the feedback signal, and a VCO control signal output is connected to the VCO  514 . 
         [0041]    Each DLL circuit uses shift registers  601  and  602 , which operate in sync with a master clock source  516 , to delay input signals. A correlator  603  multiplies the value of the power of an (n−1)-th sample point ( 701  in  FIG. 4 ) of the input  1  by the value of the power of an n-th sample point ( 702  in  FIG. 4 ) of the input  2 , and integrates the multiplication result over a given interval. A correlator  604  multiplies the value of the power of the (n−1)-th sample point ( 701  in  FIG. 4 ) of the input  1  by the value of the power of an (n−2)-th sample point ( 703  in  FIG. 4 ) of the input  2 , and integrates the multiplication result over a given interval. Thereafter, the difference between the two integration results is calculated by an adder and subtractor  605 , converted by a DAC  606  into an analog signal, and then inputted to the VCO  513  or  514  through a loop filter  607 . 
         [0042]      FIG. 4  is a waveform diagram showing an example of signal power changes with time of the input  1  and of the input  2 . 
         [0043]    After calculating the difference between the two integration results, the adder and subtractor  605  outputs a positive value since the result of the calculation by the correlator  603  is larger than the result of the calculation by the correlator  604  with the value of the sample point  701  being common to both and the value of the sample point  702  being larger than the value of the sample point  703 . This means that the transmission timing of the input  1  is behind the transmission timing of the input  2  as shown in  FIG. 4 . To remedy the situation, the output frequency of the VCOs  513  and  514  is set higher by raising the output voltage of the DLL  511  and  512  (the voltage of VCO control signals). This advances the timing at which the DACs  502  and  503  take in the input  1  (the r signal and θ signal of the transmission signal) toward the timing at which the input  2  (the r signal and θ signal of the feedback signal) is inputted. 
         [0044]    According to the first embodiment, the timing of transmitting the r signal and the timing of transmitting the θ signal are controlled with output phases of two VCOs independently of each other. Therefore, chances are small that the former and the latter affect each other&#39;s stability. On the other hand, the former and the latter are successfully aligned with each other since the timing of the same feedback signal is used as the control guide. 
         [0045]    Furthermore, the first embodiment is capable of making the amount of delay of the r signal and the amount of delay of the θ signal equal to each other despite a temperature variation and a change with time, which the conventional method of presetting a delay amount cannot overcome. 
         [0046]      FIG. 5  is a block diagram showing the structure of an RF transmitter unit  403  according to a second embodiment of this invention. 
         [0047]    In this embodiment, the timing of a feedback signal is adjusted with the timing of transmitting a θ signal as reference and the timing of transmitting an r signal is adjusted based on the thus adjusted timing of the feedback signal. This embodiment shares its basic structure with the first embodiment ( FIG. 2 ) except a timing adjustment circuit of the RF transmitter unit  403 . A detailed description on the structure common to this embodiment and the first embodiment will be omitted here. 
         [0048]    In this embodiment, the master clock source  516  supplies clock signals to the θ signal DAC  503  whereas the VCO  513  and a VCO  801  supply clock signals to the r signal DAC  502  and the feedback signal ADC  509 , respectively. The feedback signal is sampled and digitized by the ADC  509 . Then the value of the feedback signal is held by a sample-and-hold circuit  802  to prevent the rθ separation unit  510 , which operates on master clocks, from taking in a wrong value. 
         [0049]    Thereafter, the feedback signal is separated into an r (amplitude) component and a θ (phase) component by the rθ separation unit  510 . The r component of the feedback signal is compared by the r signal DLL  511  against the r component of the transmission signal whereas the θ component of the feedback signal is compared by a feedback signal DLL  803  against the θ component of the transmission signal. 
         [0050]    With the circuit structure described above, the DLL  511  and  803  judge which one of a phase of a transmission signal and a phase of a feedback signal is ahead of (or behind) the other. Based on the judgment, the DLL circuits control the output phase of a VCO  513 , which supplies clock signals to the r signal DAC  502 , and the output phase of a VCO  801 , which supplies clock signals to the feedback signal ADC  509 , in a manner that makes the difference converges to zero (in a manner that makes the timing of the transmission signal and the timing of the feedback signal synchronize with each other). The θ signal DAC  503  receives clocks from the master clock source  516  of which output frequency is fixed. 
         [0051]    As in the first embodiment, in the case where a delay difference between the path of the r signal and the path of the θ signal is one clock or more, the integer portion of the difference (in the example of  FIG. 5  where Δdr is larger than Δdθ, the integer portion corresponds to the quotient of Δdr minus Δdθ divided by the clock cycle) is adjusted by the shift register  515  whereas the remainder of the division is adjusted by controlling the VCOs  513  and  801 . Therefore, it is not necessary to widen the phase (frequency) control range of the VCOs  513  and  801  and the output signal characteristics (for example, C/N) of the VCOs can be improved. 
         [0052]    The structure of the DLL  511  and  803  is the same as that of the DLL circuits in the first embodiment which is shown in  FIG. 3 . Here, in the DLL  511 , an input  1  corresponds to the r component of the transmission signal, an input  2  corresponds to the r component of the feedback signal, and a VCO control signal output is connected to the VCO  513 . In the DLL  803 , an input  1  corresponds to the θ component of the transmission signal, an input  2  corresponds to the θ component of the feedback signal, and a VCO control signal output is connected to the VCO  801 . 
         [0053]    This embodiment is structured such that the master clock source  516  supplies clocks to the θ signal DAC  503  whereas the VCO  513  supplies clocks to the r signal DAC  502 . Instead, the r signal DAC  502  may receive clocks from the master clock source  516  whereas the θ signal DAC  503  receives clocks from the VCO  513 . However, the structure shown in  FIG. 5  provides higher stability than this modification example since the fluctuation amount of the θ component is generally smaller than the fluctuation amount of the r component. In the modification example, the θ signal separated from the transmission signal and the θ signal separated from the feedback signal are inputted to the DLL  511  whereas the r signal separated from the transmission signal and the r signal separated from the feedback signal are inputted to the DLL  803 . 
         [0054]    According to this embodiment, the demodulation precision of a feedback signal is improved by adjusting the timing of the feedback signal with the θ component, which has less fluctuation amount, as the reference. In addition, the stability is high since master clocks are supplied to one of the two (r and θ) components of a transmission signal (preferably the θ signal whose fluctuation amount is small). 
         [0055]      FIG. 6  is a block diagram showing the structure of an RF transmitter unit  403  according to a third embodiment of this invention. 
         [0056]    This embodiment deals with one of adjustment methods to adjust, in a transmitter/receiver that employs other systems than EER and that modulates an I signal and a Q signal through paths of different delay amounts, the delay difference between the two signal paths, in order to show that this invention is applicable also when a wireless transmitter employs other systems than EER. This embodiment shares its structure, except the RF transmitter unit  403 , with the first embodiment which is shown in  FIG. 1 . A detailed description on the structure common to this embodiment and the first embodiment will be omitted here. 
         [0057]    An I signal and Q signal received from the baseband unit  402  are converted into analog signals by the DACs  901  and  902 , respectively. The analog I and Q signals pass the filters  903  and  904 , respectively, before combined with each other through modulation by a orthogonal mixer (modulator)  905 . The resultant signal receives frequency conversion and then enters a power amplifier  906  from a signal input terminal of the power amplifier  906 . A portion of an output of the power amplifier  906  is distributed to a feedback circuit  907  by a distributor (coupler, or the like, omitted from the drawing). In the feedback circuit  907 , a frequency-conversion and attenuator unit  908  attenuates the power of the distributed output, which then receives frequency conversion to be converted to the baseband bandwidth (down conversion) and is converted into a digital signal by an ADC  909 . Thereafter, the feedback signal is subjected to demodulation by an IQ separation unit  910 . The I component of the feedback signal is compared by an I signal DLL  911  against the I component of the transmission signal whereas the Q component of the feedback signal is compared by a Q signal DLL  912  against the Q component of the transmission signal. 
         [0058]    With the circuit structure described above, the DLL  911  and  912  judge which one of the transmission signal and the feedback signal is ahead of (or behind) the other. Based on the judgment, the DLL circuits control the output phase of a voltage controlled oscillator (VCO)  913 , which supplies clock signals to the I signal DAC  901 , and the output phase of a VCO  914 , which supplies clock signals to the Q signal DAC  902 , in a manner that makes the difference converges to zero (in a manner that makes the timing of the transmission signal and the timing of the feedback signal synchronize with each other). The feedback signal ADC  909  receives clocks from the oscillator (master clock source)  516  of which output frequency is fixed. 
         [0059]    Sample/hold (S/H) circuits to hold signals that are to be inputted to the DACs  901  and  902  are provided on the input side of the DACs  901  and  902 . 
         [0060]    The structure of the DLL  911  and  912  is the same as that of the DLL circuits in the first embodiment which is shown in  FIG. 3 . Here, in the DLL  911 , an input  1  corresponds to the I component of the transmission signal, an input  2  corresponds to the I component of the feedback signal, and a VCO control signal output is connected to the VCO  913 . In the DLL  912 , an input  1  corresponds to the Q component of the transmission signal, an input  2  corresponds to the Q component of the feedback signal, and a VCO control signal output is connected to the VCO  914 . 
         [0061]    Generally speaking, a delay difference between the I component and the Q component is smaller than a delay difference between the r signal and the θ signal in EER, and does not raise a problem in many wireless communication systems. However, considering the fact that the recent advance of broadband has brought about increased use of 64 QAM (Quadrature Amplitude Modulation) and other types of multilevel modulation, an even higher linearity will be demanded from IQ modulation and this embodiment will gain more and more importance from now on. 
         [0062]    This invention can be utilized in a transmitter/receiver for a base station or terminal of cellular phone and other wireless communication systems, and improves the waveform quality (linearity) of an output signal of the transmitter/receiver. The improvement of the signal quality allows the device to employ EER of high efficiency or a highly saturated power amplifier. This invention thus contributes to reduction of power consumption of the device.
A wireless communication device including: a transmitter including two DA converter units which convert two digital signals into analog signals; a combiner which combines the analog signals; a distributor which extracts a portion of the combined signal as a feedback signal; an AD converter which converts the feedback signal; an oscillator unit which supplies clock signals to operate the DA converter units and the AD converter; a first separation unit which separates the feedback signal converted by the AD converter into two signals; and a comparator unit which compares at least one of the two digital signals that are obtained by separating the inputted digital signal, or, at least one of the two digital signals that are separately inputted, with the feedback signal separated by the first separation unit, wherein the oscillator unit controls the output clock signals based on a result of the comparison by the comparator unit.