FIELD OF THE INVENTION 
     The present invention relates to an analog-to-digital converter (ADC). In particular, the present invention is related to the reduction of quantization noise in an ADC. 
     BACKGROUND OF THE INVENTION 
     One problem encountered in the design of first order sigma-delta ADCs is the presence of certain direct current (DC) offset voltages at the analog input of the ADC. Such DC offset voltages are typically generated by external circuitry or internal amplifiers. The DC offset voltages which are particularly problematic have amplitudes slightly offset from, but not equal to, voltages corresponding to multiples of one-half of the voltage represented by the least significant bit (LSB) of the ADC. When such DC offset voltages are present, the ADC will generate digital signals having recurrent patterns that approximate the value of the DC offset. Thus, the digital output of the ADC will be noisy when the repetition rate of the patterns is within the baseband of the ADC. 
     A similar noise problem occurs for certain alternating (AC) voltages when the sampling rate of the ADC is high compared to the baseband. In this situation, the low frequency of the AC signals, relative to the sampling rate, results in recurrent patterns in the digital output of the ADC which are similar to those produced by problematic DC offset voltages. 
     For many application, such as telecommunications, it is important to have low noise at steady state in the absence of an input signal, and a high signal-to-noise ratio when an input signal is present at the ADC. In both situations, it is important to eliminate or reduce the recurrent patterns in the digital output of the ADC. Accordingly, it would be useful to provide an improved ADC which reduces the recurrent patterns in the digital output of ADCs, such as sigma-delta ADCs, caused by certain level DC offsets and certain AC signals. 
     SUMMARY OF THE INVENTION 
     The present invention provides an analog-to-digital converter including an analog-to-digital conversion circuit and an alternating current source. The alternating current, which is supplemental to the signal which is being converted to a representative digital signal, is applied to the input of the analog-to-digital conversion circuit to reduce noise caused by quantization effects such as the DC offset or certain AC signals. 
     The present invention further provides an analog-to-digital converter including an integrator, a counter, a digital-to-analog converter (DAC), and an alternating current source. The integrator is coupled to the counter and applies a signal representative of the integral of an analog signal to the counter. The counter produces a digital signal representative of the integral. The counter is coupled to the digital-to-analog converter which converts the digital signal to an analog signal which is applied to the input of the integrator. The current source is also coupled to the input of the integrator, and applies an alternating current to the input which is subtracted from the analog signal applied to the analog-to-digital converter. The alternating current reduces the noise caused by quantization effects. 
     The present invention further provides an analog-to-digital converter including conversion means coupled to the current means. The conversion means produces a digital signal representative of an analog signal applied to the input of the converter, and the current means applies an alternating current to the analog input. 
     The present invention still further provides an analog-to-digital converter including integrator means, and counter means operatively coupled to form an analog-to-digital converter. The integrator means applies a signal representative of the integral of an analog signal to the counter means. The counter means produces a digital signal representative of the input signal applied to the integrator means. The counter means is coupled to the converter means which converts the digital signal to an analog signal which is applied to the input of the integrator means. The analog-to-digital converter also includes a current means which applies an alternating current to the input of the integrator means. 
    
    
     BRIEF DESCRIPTION OF THE DRAWINGS 
     FIG. 1 is a schematic diagram of the preferred embodiment of an analog-to-digital converter according to the present invention; 
     FIG. 2 is a graphical representation of a waveform for a dither current applied to the input of the analog-to-digital converter illustrated in FIG. 1; 
     FIG. 3a is a graphical representation of signal-to-noise ratios vs. direct current offsets when a 2-level dither current is applied to the input of the analog-to-digital converter illustrated in FIG. 1; and 
     FIG. 3b is a graphical representation of signal-to-noise ratios vs. direct current offsets when a 4-level dither current is applied to the input of the analog-to-digital converter illustrated in FIG. 1. 
    
    
     DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT 
     Referring to FIG. 1, a sigma-delta analog-to-digital converter (ADC) 10 includes an integration circuit 12, a comparator circuit 14, a counter circuit 16, a decimator circuit 18, a dither current circuit 20, a digital-to-analog converter (DAC) 22, an analog input terminal 24, and a digital output 26. Circuits 12, 14, 16, 18, 20 and 22 are preferably fabricated from the same substrate, and disposed within a single integrated circuit package 11. In general, ADC 10 converts analog signals applied to input 24 into digital signals output at output 26. The preferred embodiment of ADC 10 illustrated in FIG. 1 is directed to the conversion of analog signals representative of audio signals having frequencies in the range of 20 Hz to 4 kHz. This type of ADC is typically used in telecommunications applications, where the baseband of 20 Hz to 4 kHz is satisfactory for converting analog signals representative of human voice to digital signals. 
     Integration circuit 12 includes an operational amplifier 28 having its inverting input coupled to analog input 24 by a resistor 30 and also coupled to the output by the series connection of a feedback resistor 32 and feedback capacitor 34. The non-inverting input of amplifier 28 is coupled to an appropriate reference voltage such as ground. The output of amplifier 28 is coupled to the non-inverting input of voltage comparator 14. The inverting input of comparator 14 is connected to an appropriate reference voltage such as ground. Comparator 14 is powered so that when the voltage at the non-inverting input is greater than the voltage at the inverting input, a high voltage (positive) is output at the output of comparator 14, and when the voltage at the non-inverting input is less than the voltage at the inverting input, the output of comparator 14 is negative (low). In general, the output of comparator 14 provides a digital signal to counter 16 which is either high, zero or low. 
     Although the number of bits produced at the output of counter 16 will vary depending on the application, the preferred embodiment provides for a 5 bit counter 16. The input of counter 16 is coupled to the output of comparator 14, and samples the output of comparator 14 at a rate in the range of 2-4 MHz, which results in an oversampling of the output of comparator 14. DAC 22 will have a resolution depending on counter 16 which has a resolution based upon the particular counter 16. In the preferred embodiment, DAC 22 has 5 bits of resolution. The digital input of DAC 22 is connected to the digital output of counter 16. The analog output of DAC 22 is connected to the inverting input of amplifier 28. 
     Dither current circuit 20 includes a multi-level dither current generator 36 and a clock 38. In the present embodiment, clock 38 has a frequency of 256 kHz. The output of circuit 20 is connected to the inverting input of amplifier 28 and applies a dither current to the inverting input of amplifier 28. Referring to FIG. 2, FIG. 2 illustrates the waveform for a 4-level dither current. This current has a fundamental frequency of 32 kHz based upon the 256 kHz clock rate and a step-shaped waveform substantially as shown in FIG. 2, wherein the current changes in steps of magnitude I. By way of modification, the waveform may be modified to have varying levels of dither. For example, the dither current could have a single level which results in an alternating square wave having a frequency of 32 kHz. In general, the dither current, whether 4-level, 2-level, or any other number of levels, is an alternating current which is supplemental to the current provided by the signal being monitored at analog input 24, and depending upon the application, may have varying waveforms. To avoid the introduction of substantial undesired noise, I is set at about 1/64 the value of the full scale current of DAC 22. 
     Decimator 18 is also coupled to the digital output of counter 16 and samples the digital output of counter 16 at a rate which is approximately twice that of the maximum frequency of the signal which is being monitored at analog input 24. To avoid undesired effects such as aliasing, decimator 18 includes two stages. The first stage is a digital low pass filter and the second stage is the circuitry which samples the output of the low pass filter. By way of example, if ADC 10 is designed to convert signals in the baseband of 20 Hz to 4 kHz, the sampling rate of the second stage of decimator 18 would be in the range of 8 kHz and the passband of the first stage would be 4 kHz (i.e., 1/2 the sampling frequency of the first stage). As a result, decimator 18 operates as a low pass filter to remove noise which is above the highest frequency of the signal desired to be sampled by ADC 10. For example, the fundamental frequency of the dither current is 32 kHz. 
     Referring now to the operation of ADC 10, ADC 10 may be used to operate on a continuous analog waveform applied to analog input 24 to produce a digital output comprising a sequence of binary numbers. Each of these numbers approximates a corresponding analog sample by a finite number of bits (e.g., 5 bits). Referring to the inverting input of amplifier 28, this input is basically a summing point at which an error signal is generated. The error signal is the difference between the analog signal applied to input 24, and the sum of the dither current produced by circuit 20 and the analog signal output by DAC 22. The error signal represents how closely the digital output of counter 16 approximates the analog signal at input 24. Integration circuit 12 integrates the error signal to filter out the high frequency components of the error signal and retain the portion of the error signal within the bandwidth of signals for which ADC 10 was designed to properly convert (baseband). For example, for audio telecommunications applications, it is desirable to calculate the values of feedback resistor 32 and feedback capacitor 34 so that components of the error signal above 4 kHz are filtered out and components of the signal below 4 kHz are substantially retained. 
     The output of integration circuit 12 is compared to a reference level (e.g., ground) by comparator 14. Counter 16 counts based upon the status of comparator 14, where counter 16 counts down if the output is low, counts up if the output of comparator 14 is high. DAC 22, as discussed above, converts the digital signals produced by counter 16 to an analog signal applied to the inverting input of amplifier 28. 
     As discussed above, the sampling rate of counter 16 is in the range of 2-4 MHz, which results in oversampling of input signals falling within the baseband of ADC 10 (20 Hz-4 kHz). In other words, the frequency of counter 16 is in the range of 100-200 times the maximum frequency of signals for which ADC 10 is designed to properly convert to digital signals. As a result of oversampling by counter 16, the quantization noise within the baseband of ADC 10 is reduced. However, the noise outside of the baseband of ADC 10 is increased due to the position of integration circuit 12 and the presence of the feedback loop including DAC 22. This outband noise is later removed by the low pass filter of decimator 18. 
     The effect of the dither current produced by dither current circuit 20 will now be discussed by way of example. Assuming that circuit 20 is idle, or nonexistent, and analog input 24 is at ground, the output of counter 16 will be a sequence of +1, 0, -1, 0, +1, 0, -1, . . . Decimation filter 18 operates to filter the output of counter 16 in this situation such that its digital output is a binary number representative of 0. Similarly, if the analog signal at input 24 is a DC signal and equals 1/2 of the voltage corresponding to the least significant bit, the counter output will be a sequence of +1, 0, +1, 0 . . . Decimation filter 18 filters this output from counter 16 and outputs a binary signal representative of 0.5. However, the situation will change where the DC offset is slightly higher than 1/2 of the corresponding voltage of the least significant bit by an amount ΔV. In this situation, the counter output will generally remain in the +1, 0, +1, 0 sequence with the exception of occasional +2 output values (second least significant bit of the counter) to make up the ΔV. The repetition rate of the +2 values produced at the second least significant bit of the counter depends upon the ΔV, and the values of resistor 32 and capacitor 34, and the current output of DAC 22. Where the repetition rate of the change in state of the second least significant bit of counter 16 is within the bandwidth of decimator 18, the output of decimator 18 will be a digital signal representative of the signal at analog input 24 which is considered noisy. Thus, for an analog signal including a DC offset which is slightly higher than 1/2 of the corresponding voltage of the least significant bit, the signal-to-noise ratio for the signals is reduced due to the noise resulting at digital output 26, as discussed above. 
     In summary, the noise induced at digital output 26 varies widely, depending upon the DC offset level. The noise occurs in peaks and is most prominent when the DC component of a signal is around, but not exactly at, a multiple of 1/2 of the voltage corresponding to the least significant bit. 
     The dither current produced by dither current circuit 20 operates to disturb the DC offset level of a signal by adding or subtracting a current value of I or 2I at different times, where the value of I is set at about 1/64 the value of the full scale current of DAC 22. A dither current having a step-shaped waveform or other appropriately shaped waveform as shown in FIG. 2, when applied to the inverting input of amplifier 28, results in a disturbance of the DC offset since the current of the DC offset changes to values alternating by ±I for a 2-level dither and ±2I for a 4-level dither. The effect of the dither current is that DC offsets are disturbed so that the DC offset level is not sustained and the energy applied by a particular DC offset level to integrator 28 is reduced. 
     Referring to FIGS. 3a and 3b, these figures are graphical representations of simulated signal-to-noise ratios (S/N) corresponding to a range of offset signals. Absolute scales on FIGS. 3a and 3b are not provided for the S/N axes since these figures are provided for relative comparison and the scales of FIGS. 3a and 3b are substantially the same. FIGS. 3a and 3b are exemplary only, and are directed to the situation where the voltage corresponding to the LSB is 0.05 volts. Thus, for this example, one-half of the voltage is 0.025 volts, and, as discussed above, a reduced S/N ratio around 0.025 volts and -0.025 volts would be expected. By way of further example, the vertical scales of FIGS. 3a and 3b may be 2 dB per division. 
     FIG. 3a illustrates the S/N ratio when a 2-level dither current is used. Without the dither current, the reduction in the S/N ratio would be even greater than that shown around 0.025 and -0.025 volts. FIG. 3b illustrates the S/N ratio when a 4-level dither current is used. In comparing FIGS. 3a and 3b, it can be seen that the 4-level dither current further reduces the drop in the S/N ratio around 0.025 and -0.025 volts. For a 4-level dither, the noise peaks may be reduced by up to 2/3, thus also reducing the repetitive noise energy by the same factor. By way of example, for ADC 10, operative for a baseband of 20 Hz to 4 kHz, the frequency of the dither current is set at approximately 32 kHz which is approximately 4 times the sampling frequency of the first stage of decimator 18 and 8 times the sampling frequency of the second stage of decimator 18. This allows decimator 18 to filter out the high frequency harmonics of the dither current. 
     It will be understood that the above description is of the preferred exemplary embodiment of the invention, and that the invention is not limited to the specific forms shown. For example, decimator 18 may consist of several decimation filters to permit proper tailoring of the decimation filter to the types and bandwidths of the signals for which ADC 10 is configured to sample. Various other substitutions, modifications, changes and omissions may be made in the design and arrangement of the elements of the preferred embodiment without departing from the spirit of the invention as expressed in the appended claims.
An improved sigma-delta analog-to-digital converter (ADC) is disclosed herein. The digital converter includes a dither circuit fabricated within the package of the ADC. The circuit is configured to apply a dither current to the analog input of the ADC. The frequency of the dither current is selected based upon the bandwidth of the analog signals for which the ADC is designed to sample and convert to digital signals. Application of the dither current to the input of the ADC reduces quantization noises produced as a result of certain ranges of DC offset voltages found within analog signals applied to the ADC.